Filter and leakage inductance collaborative design method for single-stage bidirectional high-frequency isolation inverter

Abstract

The application relates to the technical field of power control, in particular to a filter and leakage inductance collaborative design method of a single-stage bidirectional high-frequency isolation inverter, which comprises the following steps: obtaining basic design parameters and constructing an inverter modulation model according to a modulation strategy, and solving the model to obtain a modulation signal expression; deducing zero-current switching current constraint conditions and zero-voltage switching current constraint conditions of a switching tube based on the modulation strategy; combining the constraint conditions with the inverter modulation model to solve and obtain soft switching constraint conditions; determining a collaborative value range of an inverter side inductance of a filter and transformer leakage inductance which meets the soft switching requirement according to the soft switching constraint conditions and a value range of the modulation signal expression; selecting design parameters of the inverter side inductance of the filter and the transformer leakage inductance, and determining the rest circuit parameters of the filter according to reactive power demand and grid-connected harmonic standards. The application can effectively inhibit the problems of soft switching failure, increased loss and electromagnetic interference caused by parameter mismatch.

Description

Technical Field

[0001] This application relates to the field of power control technology, and in particular to a filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter. Background Technology

[0002] Single-stage bidirectional high-frequency isolated inverters are widely used in various scenarios due to their electrical isolation, bidirectional power transmission capability, and high power density. To improve system power density and dynamic response speed, the switching frequency of these inverters is typically designed to be high, generally in the tens or even hundreds of kHz range. However, with the increase in switching frequency, the switching losses of the switching devices in hard-switching mode increase significantly. This not only leads to excessive temperature rise and shortened lifespan of the devices but also causes severe electromagnetic interference (EMI), affecting system reliability. Therefore, achieving soft switching (i.e., zero-voltage turn-on or zero-current turn-off) over a wide load range becomes crucial for ensuring efficient and stable system operation. In this inverter topology, soft switching relies on the energy exchange and resonance process within the circuit. The filter parameters and transformer leakage inductance, as key inductive components, directly determine the energy distribution and commutation process in the circuit, and are decisive factors affecting the soft-switching effect and overall system performance.

[0003] However, existing technologies typically employ an isolated design approach to determine the parameters of filters and transformer leakage inductance. Specifically, filter design primarily focuses on meeting harmonic suppression requirements of the grid-connected current, while transformer leakage inductance design only addresses maximum energy transfer needs. This independent design approach fails to consider the coupling relationship between the two during circuit resonant commutation, easily leading to parameter mismatch. Especially for inverters with bidirectional power transfer characteristics, parameters selected based on a single power flow direction often fail to meet soft-switching conditions when power flows in the reverse direction. Furthermore, under light load or grid voltage fluctuation conditions, such parameter combinations struggle to cover the soft-switching energy constraints across the entire power range, causing soft-switching failures in the inverter under specific operating conditions, resulting in increased losses and worsened electromagnetic interference. Summary of the Invention

[0004] This application provides a filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter. The designed circuit parameters can meet the soft-switching performance requirements in both bidirectional power transfer modes, and effectively suppress soft-switching failure, increased losses, and electromagnetic interference caused by parameter mismatch. This application provides the following technical solution:

[0005] In a first aspect, this application provides a filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter, the design method comprising the following steps:

[0006] The basic design parameters of a single-stage bidirectional high-frequency isolation inverter are obtained, and an inverter modulation model is constructed based on the modulation strategy of the single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode. The modulation model is then solved to obtain a modulation signal expression that includes the parameters of the filter inverter-side inductance and transformer leakage inductance.

[0007] Based on the modulation strategy, the zero-current switching current constraint and the zero-voltage switching current constraint of the switching transistor are derived; the zero-current switching current constraint and the zero-voltage switching current constraint are combined with the inverter modulation model to solve for the soft-switching constraint.

[0008] Based on the soft-switching constraints and the range of values ​​for the modulation signal expression, determine the combined range of values ​​for the filter inverter-side inductance and transformer leakage inductance that meet the soft-switching requirements; within the combined range, select the design parameters for the filter inverter-side inductance and transformer leakage inductance, and determine the remaining circuit parameters of the filter based on the reactive power requirements and grid harmonic standards.

[0009] In one specific implementation scheme, the construction of the inverter modulation model includes the following steps:

[0010] Based on the modulation strategy, the instantaneous current expression of the transformer leakage inductance within a switching cycle is determined and transformed into the current increment expression for each time segment, and the current increment expression of the filter inverter-side inductor during the demagnetization stage is determined.

[0011] Based on the inductor volt-second balance principle, the leakage inductor flux balance equation is established to describe the sum of the current increment expressions of the transformer leakage inductance within one switching cycle as zero, and the filter inductor flux balance equation is established to describe the sum of the current increment expressions of the filter inverter-side inductance during the magnetization and demagnetization stages as zero.

[0012] Based on the current increment expressions of the transformer leakage inductance and the AC side voltage, a power transfer equation is constructed to characterize that the calculated actual power is equal to the reference power.

[0013] In the bidirectional power transmission mode, the corresponding leakage inductance flux balance equation, the filter inductance flux balance equation, and the power transmission equation are combined to form the inverter modulation model.

[0014] In one specific implementation, solving the inverter modulation model to obtain the modulation signal expression including the parameters of the filter inverter-side inductance and the transformer leakage inductance includes the following steps:

[0015] In DC-AC transmission mode, the combined leakage inductance flux balance equation, the filter inductance flux balance equation and the power transmission equation are solved to obtain a first modulation signal expression that characterizes the duty cycle of the primary circuit, including the transformer leakage inductance and the filter inverter-side inductance variable.

[0016] In AC-DC transmission mode, the combined leakage inductance flux balance equation, the filter inductance flux balance equation, and the power transmission equation are solved to obtain a second modulation signal expression that includes the transformer leakage inductance and the filter inverter-side inductance variable and characterizes the duty cycle of the secondary circuit.

[0017] In a specific feasible implementation, deriving the zero-current switching current constraint condition for the switching transistor includes the following steps:

[0018] In the DC-AC transmission mode, a first constraint condition is established to limit the duty cycle represented by the first modulation signal expression to satisfy a first numerical upper limit.

[0019] In the AC-DC transmission mode, a second constraint condition is established that limits the duty cycle represented by the second modulation signal expression to meet the second numerical upper limit, and a third constraint condition is established that limits the duty cycle represented by the second modulation signal expression to meet the preset timing logic constraint.

[0020] The first constraint, the second constraint, and the third constraint are combined to form the zero-current switching current constraint.

[0021] In a specific feasible implementation, deriving the zero-voltage switching current constraint condition for the switching transistor includes the following steps:

[0022] In DC-AC transmission mode, for the first set of switches on the primary side, a fourth constraint condition is established to ensure that the turn-off current meets the requirements determined based on the parasitic capacitance energy demand.

[0023] In AC-DC transmission mode, for the second set of switches on the primary side, a fifth constraint condition is established to limit the turn-off current to meet the requirements determined based on the parasitic capacitance energy demand, and a sixth constraint condition is established to limit the current of the secondary side switches to meet the preset current polarity logic constraint when the switches are in operation.

[0024] The fourth, fifth, and sixth constraints are combined to form the zero-voltage switching current constraint.

[0025] In a specific feasible implementation, the design parameters for the filter inverter-side inductance and transformer leakage inductance are selected within the range of the coordinated values, including the following steps:

[0026] Based on the influence of transformer leakage inductance on the effective operating range at the peak of the modulation signal, and with the goal of meeting the preset power transmission capacity, the design parameters of transformer leakage inductance are selected.

[0027] Based on the selected design parameters of the transformer leakage inductance, by analyzing the waveform of the modulation signal under different inductance values, the minimum inductance value that can make the modulation signal meet the soft switching constraint condition throughout the entire modulation cycle is determined, and the minimum inductance value is determined as the lower limit of the inductance value on the inverter side of the filter.

[0028] Based on the preset fundamental voltage drop limit requirement on the inductor on the inverter side of the filter, the upper limit of the value of the inductor on the inverter side of the filter is determined by using the fundamental voltage drop calculation formula.

[0029] Within the numerical range formed by the lower limit and the upper limit, the design parameters of the filter inverter-side inductor are selected by comprehensively considering the current ripple magnitude and system cost and volume factors.

[0030] In one specific implementation scheme, determining the remaining circuit parameters of the filter based on reactive power requirements and grid-connected harmonic standards includes the following steps:

[0031] Based on the preset ratio limit of reactive power injected into the filter capacitor to transmission power, and combined with the rated transmission power of the inverter and the grid voltage parameters, the value of the filter capacitor is calculated and determined.

[0032] By comparing the harmonic distribution characteristics under different power transmission directions, the power transmission direction with the largest harmonic content is selected as the design condition. The inverter modulation voltage under this design condition is subjected to Fourier decomposition to determine the main subharmonic with the largest amplitude.

[0033] Based on the determined filter inverter-side inductor and filter capacitor, the grid current harmonic amplitude corresponding to the main subharmonic is calculated using the transfer function of the grid current with respect to the inverter modulation voltage. The value that enables the grid current harmonic amplitude to meet the preset grid harmonic standard is selected as the design parameter of the filter grid-side inductor.

[0034] Secondly, this application provides a filter and leakage inductance co-design system for a single-stage bidirectional high-frequency isolation inverter, the design system comprising:

[0035] The modulation signal solving module is used to obtain the basic design parameters of the single-stage bidirectional high-frequency isolation inverter, and construct the inverter modulation model according to the modulation strategy of the single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode. The module solves the inverter modulation model to obtain the modulation signal expression containing the parameters of the filter inverter-side inductance and transformer leakage inductance.

[0036] The constraint solving module is used to derive the zero-current switching current constraint and the zero-voltage switching current constraint of the switching transistor based on the modulation strategy; and to solve the soft-switching constraint by combining the zero-current switching current constraint and the zero-voltage switching current constraint with the inverter modulation model.

[0037] The design parameter determination module is used to determine the coordinated value range of the filter inverter-side inductance and transformer leakage inductance that meet the soft-switching requirements based on the soft-switching constraints and the value range of the modulation signal expression; select the design parameters of the filter inverter-side inductance and transformer leakage inductance within the coordinated value range; and determine the remaining circuit parameters of the filter based on the reactive power requirements and grid harmonic standards.

[0038] Thirdly, this application provides an electronic device, including a processor and a memory, wherein the memory stores a program, which is loaded and executed by the processor to implement the aforementioned filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter.

[0039] Fourthly, this application provides a computer-readable storage medium storing a program that, when executed by a processor, is used to implement the aforementioned filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter.

[0040] First, based on the basic design parameters of a single-stage bidirectional high-frequency isolation inverter and the modulation strategy under bidirectional power transmission mode, an inverter modulation model is constructed and the modulation signal expression, including the parameters of the filter inverter-side inductance and transformer leakage inductance, is obtained, thus establishing the mathematical relationship between key parameters. Then, based on the modulation strategy, the zero-current and zero-voltage switching current constraints of the switching transistors are derived, and these physical constraints are solved in conjunction with the inverter modulation model to obtain the constraints that allow the system to meet soft-switching operation. Finally, by comprehensively utilizing the soft-switching constraints and the range of values ​​for the modulation signal expression, the coordinated value range of the filter inverter-side inductance and transformer leakage inductance that can simultaneously meet the soft-switching requirements is determined. Within this coordinated range, design parameters are precisely selected, ensuring that the designed circuit parameters meet the soft-switching performance requirements under bidirectional power transmission mode.

[0041] The above description is only an overview of the technical solution of this application. In order to better understand the technical means of this application and to implement it in accordance with the contents of the specification, the preferred embodiments of this application are described in detail below with reference to the accompanying drawings. Attached Figure Description

[0042] Figure 1 This is a schematic diagram of the circuit structure of a single-stage bidirectional high-frequency isolation inverter according to an embodiment of this application.

[0043] Figure 2 This is a flowchart illustrating the filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter according to an embodiment of this application.

[0044] Figure 3 This is a waveform timing diagram of the modulation strategy of the inverter in the DC-AC and AC-DC directions according to the embodiments of this application.

[0045] Figure 4 This is the equivalent circuit diagram of the inverter in the DC-AC direction of the embodiment of this application.

[0046] Figure 5 This is the equivalent circuit diagram of the inverter in the AC-DC direction of the embodiment of this application.

[0047] Figure 6 This is a waveform diagram of the modulation signal of the inverter in the AC-DC direction according to an embodiment of this application.

[0048] Figure 7 This is a waveform diagram of the modulation signal of the inverter in the DC-AC direction according to an embodiment of this application.

[0049] Figure 8 This is a waveform diagram of the modulation signal of the inverter in a specific cross-section according to an embodiment of this application.

[0050] Figure 9 This is a block diagram of an electronic device with a filter and leakage inductance co-designed in a single-stage bidirectional high-frequency isolation inverter according to an embodiment of this application. Detailed Implementation

[0051] The specific embodiments of this application will be described in further detail below with reference to the accompanying drawings and examples. The following examples are used to illustrate this application, but are not intended to limit the scope of this application.

[0052] This application provides a method for the coordinated design of filters and leakage inductance in a single-stage bidirectional high-frequency isolation inverter. This method is used to collaboratively optimize the design of key magnetic components and filter parameters in a single-stage bidirectional high-frequency isolation inverter. (Refer to...) Figure 1 This is a schematic diagram of the circuit structure of a single-stage bidirectional high-frequency isolation inverter in an embodiment of this application. Figure 1 As shown, the inverter includes, from left to right, a DC power supply. Primary circuit, high-frequency isolation transformer, secondary circuit, LCL filter, and AC power supply The primary circuit employs a full-bridge structure, consisting of four switching transistors (S1, S2, S3, and S4), used to convert DC power to high-frequency AC power or vice versa. A high-frequency isolation transformer is connected between the primary and secondary circuits, as shown in the diagram. This represents the equivalent leakage inductance of the transformer. This indicates the transformer's turns ratio. The secondary circuit (labeled LC in the diagram) also consists of four switching transistors (S5, S6, S7, and S8). Specifically, this secondary circuit includes two four-quadrant switches, each consisting of interconnected emitters of two IGBTs to achieve bidirectional current blocking and conduction. The LCL filter is connected between the secondary circuit and the AC power supply. Between, mainly including the inverter-side inductor , grid-side inductor and filter capacitors It is used to filter out high-frequency harmonics and realize grid connection interface functions. It should be noted that... Figure 1 The red markings are mainly used to indicate the direction of current flow, while the blue dashed boxes are mainly used to indicate the functional areas corresponding to the secondary circuit.

[0053] Reference Figure 2 This is a flowchart illustrating a filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter provided in one embodiment of this application. The design method includes at least the following steps:

[0054] Step S1: Obtain the basic design parameters of the single-stage bidirectional high-frequency isolation inverter, and construct the inverter modulation model according to the modulation strategy of the single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode. Solve the inverter modulation model to obtain the modulation signal expression containing the parameters of the filter inverter-side inductance and transformer leakage inductance.

[0055] Step S1 includes at least the following steps:

[0056] Step S101: Obtain the basic design parameters of the single-stage bidirectional high-frequency isolation inverter.

[0057] In step S101, the basic design parameters first need to be determined based on the actual application scenario and design specifications of the inverter. Specifically, based on the inverter's design goals, the DC-side voltage of the inverter is determined. AC side voltage Transmission power and transformer turns ratio .

[0058] Step S102: Construct an inverter modulation model based on the modulation strategy of a single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode.

[0059] Step S102 includes at least the following steps:

[0060] Step S1021: Based on the modulation strategy, determine the instantaneous current expression of the transformer leakage inductance within a switching cycle and convert it into the current increment expression for each time segment, and determine the current increment expression of the filter inverter-side inductor during the demagnetization stage.

[0061] In step S1021, the preset modulation strategy adopted by the inverter in bidirectional power transmission mode is first determined. The modulation strategy specifies the turn-on and turn-off sequence of each switch at different time points, thereby dividing one switching cycle into different operating modes. (Refer to...) Figure 3 , Figure 3 (a) and Figure 3 Figure (b) shows the waveform timing diagrams of the modulation strategies in the DC-AC and AC-DC directions in the embodiments of this application, respectively. This is the base modulation signal in the DC-AC direction. The base modulation signal in the AC-DC direction is represented by g, which is the total duty cycle of the primary and secondary circuits in one switching cycle. 1、4 g 2、3 g5, g6, g7, and g8 are the trigger pulses for the corresponding switching transistors, u p u q u Lδ These represent the DC-side voltage, AC-side voltage, and voltage drop across the leakage inductance of the high-frequency transformer. , They flow through S respectively 5,6 Flowing through S 7,8 The current, This is the grid-connected current.

[0062] It should be noted that, Figure 3 The different color markings are only used to distinguish various signal waveforms and reference lines. The red markings are mainly used to show the modulation signal, the switching transistor trigger pulse signal, and the voltage waveform involved in timing analysis. The blue markings are mainly used to show the current waveform. The green dashed lines are mainly used to show the voltage reference line, the current reference line, or the comparison baseline. The above different color markings are only used for illustration and reading distinction of the figure, and do not indicate any additional parameter limitations, control logic limitations, or other substantial limitations.

[0063] Specifically, refer to Figure 4 This is the equivalent circuit diagram corresponding to each mode in the DC-AC direction in the embodiments of this application. For example... Figure 4 As shown in (a), when t∈[t 10 ,t 11 When the converter is in freewheeling mode 1, the primary side current is discontinuous. ac Freewheeling current is achieved through the channel formed by the conducting switch S8 and the anti-parallel diode of switch S7. For example... Figure 4 (b) and Figure 4 As shown in (c), when t∈[t 11 ,t 13At time t, the converter operates in commutation state 1. 11 At a certain moment, switches S2 and S3 perform zero-current turn-off (ZCS-OFF), and after a dead time, switches S1 and S4 perform zero-current turn-on (ZCS-ON). At this time, the DC bus voltage is applied to the transformer leakage inductance terminals, and the leakage inductance current begins to rise linearly from zero. For example... Figure 4 As shown in (d), when t∈[t 13 , t 14 When the converter is in energy transfer state 1, the grid-connected current i ac Leakage inductance current i entirely coupled from the primary side p As a result, the freewheeling branch on the secondary side is naturally cut off, and the switching transistor S8 remains on to maintain the loop connection. For example... Figure 4 (e) and Figure 4 As shown in (f), when t∈[t 14 ,t 16 At time 2, the converter is in commutation state. 14 Constantly control the switching transistors S1 and S4 to perform zero-voltage turn-off (ZVS-OFF), leakage inductance current i p Continuing the current flow, charge is drawn from the parasitic capacitors of S1 and S4, and the parasitic capacitors of S2 and S3 are discharged; when the voltage across S2 and S3 drops to zero, their anti-parallel diodes naturally conduct, thus providing... t 15 This creates conditions for the zero-voltage turn-on (ZVS-ON) of the switching transistors S2 and S3 at all times.

[0064] Similarly, refer to Figure 5 This is the equivalent circuit diagram corresponding to each mode in the AC-DC direction in the embodiments of this application. For example... Figure 5 As shown in (a), when t∈[t 20 ,t 21 When the converter operates in freewheeling mode 2, the primary current is discontinuous, while the grid-connected current i on the secondary side is... ac Freewheeling current is achieved through the channel formed by the conducting switch S7 and the anti-parallel diode of switch S8. For example... Figure 5 (b) and Figure 5 As shown in (c), when t∈[t 21 ,t 23 At time t, the converter operates in commutation state 3. 21 At time t, control switches S1 and S4 perform zero-current turn-off (ZCS-OFF) in advance. After the dead time, at t 22 The switching transistors S2 and S3 are constantly controlled to conduct with zero current (ZCS-ON). For example... Figure 5 As shown in (d), when t∈[t 23 ,t24 When [t], the converter continues [t] 22 ,t 23 The on / off status during a time period. For example... Figure 5 (e) and Figure 5 As shown in (f), when t∈[t 24 ,t 26 At time t, the converter operates in commutation state 4. 24 At time t, control switches S2 and S3 perform zero-voltage turn-off (ZVS-OFF) in advance; after a brief dead time, at t 25 At that moment, switches S1 and S4 achieve zero-voltage conduction (ZVS-ON). It should be noted that, in order to prevent i q If the oscillation reverses again, the control system should be in [t] 24 ,t 26 Within a specified time period, switch S7 is promptly turned off, and at this time, the voltage across S7 is zero, thus achieving zero-voltage turn-off (ZVS-OFF). Figure 5 As shown in (g), when t∈[t 26 ,t 27 At this time, the converter operates in energy transfer state 2, during which i ac Completely by i p The secondary side continuous current branch is discontinuous. For example... Figure 5 As shown in (h), when t∈[t 27 ,t 28 At time t, the converter operates in commutation state 5. 27 The control switch S7 is constantly re-executed to perform zero-current conduction (ZCS-ON), thereby completing the modulation cycle of this period and entering the freewheeling phase of the next period.

[0065] It should be noted that, Figure 4 and Figure 5 The red markings are mainly used to show the current flow path under the corresponding operating mode, and the blue markings are mainly used to show the switching devices or branches that need to be highlighted under the corresponding operating mode, so as to distinguish the conduction relationship and commutation process under different modes; different colors are only used for illustration in the attached drawings and do not indicate additional structural or parameter limitations.

[0066] Based on the circuit states of each mode in the DC-AC direction described above, and according to circuit principles, the transformer leakage inductance current within one switching cycle can be derived. The instantaneous expression changing with time. Specifically, in commutation state 1 and energy transfer state 1, the DC side voltage is applied to the leakage inductor, and the current rises; in commutation state 2, the leakage inductor energy is released. Combining the characteristics of each stage, the leakage inductor current... The instantaneous expression is shown in formula (1):

[0067] ; (1)

[0068] Subsequently, the above continuous instantaneous current expression is transformed into a discrete current increment expression. By calculating the product of the current slope and the duration in each mode, the leakage inductance current increment expression for each time segment in the DC-AC direction is obtained, as shown in formula (2):

[0069] ; (2)

[0070] in, The current increment corresponding to freewheeling state 1, The current increment corresponding to commutation state 1, The current increment corresponding to energy transfer state 1, The current increment corresponding to commutation state 2; For switching frequency, and These are the duty cycles corresponding to commutation states 1 and 2, respectively.

[0071] Finally, regarding the inverter-side inductor in the LCL filter... Considering the demagnetization process (i.e., energy release phase) after energy transfer, the inverter-side inductor is determined based on the relationship between inductor voltage and current. The expression for the current increment during the demagnetization stage is shown in equation (3):

[0072] ; (3)

[0073] in, Corresponding filter inverter-side inductor The current increment during the demagnetization phase.

[0074] It should be noted that the derivation logic for the AC-DC direction is consistent with the above process. Based on Figure 5 The modulation strategy described in the text can also be used to derive the expression for the incremental leakage inductance current and the expression for the incremental filter inductance current in this direction. The specific derivation process will not be repeated here.

[0075] Among them, leakage inductance current in the AC-DC direction The instantaneous expression is shown in formula (4):

[0076] ; (4)

[0077] The expression for the leakage inductance current increment in each time segment under the AC-DC direction is shown in Equation (5):

[0078] ; (5)

[0079] in, The current increment corresponding to freewheeling state 2, The current increment corresponding to commutation state 3 The current increment corresponding to commutation state 4, The current increment corresponding to energy transfer state 2, The current increment corresponding to commutation state 5; , , These are the duty cycles corresponding to commutation states 3, 4, and 5, respectively.

[0080] In AC-DC inverter side inductor The expression for the current increment during the demagnetization stage is shown in equation (6):

[0081] ; (6)

[0082] Step S1022: Based on the inductor volt-second balance principle, establish the leakage inductance flux balance equation, which describes the sum of the current increment expressions of the transformer leakage inductance within one switching cycle as zero, and the filter inductance flux balance equation, which describes the sum of the current increment expressions of the filter inverter-side inductance during the magnetization and demagnetization stages as zero.

[0083] In step S1022, based on the inductor volt-second balance principle, when the inverter is operating in steady state, the net change in current of each magnetic component within one switching cycle should be zero. Specifically, for the transformer leakage inductance, the sum of its current increments in each stage within one switching cycle must be zero; for the filter inverter-side inductor, the sum of its current increments in the energy transfer stage (magnetizing) and the demagnetizing stage must be zero. Accordingly, the current increment expressions obtained in step S1021 are combined to construct a set of magnetic flux balance equations in the DC-AC direction, as shown in equation (7):

[0084] ; (7)

[0085] Step S1023: Based on the current increment expressions of the transformer leakage inductance and the AC side voltage, construct a power transfer equation that characterizes the calculated actual power as equal to the reference power.

[0086] In step S1023, in order to solve the modulation model in the DC-AC direction, power transmission needs to be introduced as an independent constraint. That is, the actual transmitted power calculated by the inverter's internal state variables (current increment) should be strictly equal to the system's preset reference transmitted power. Based on the leakage inductance current increment derived in step S1021 and combined with the current AC side voltage, the power transmission equation in the DC-AC direction is constructed, as shown in formula (8):

[0087] (8)

[0088] in, This is the reference transmission power.

[0089] Step S1024: In bidirectional power transmission mode, the corresponding leakage inductance flux balance equation, filter inductance flux balance equation and power transmission equation are combined to form an inverter modulation model.

[0090] In step S1024, the physical constraint equations determined in the preceding steps are combined to construct inverter modulation model equation sets applicable to both DC-AC and AC-DC directions.

[0091] In the DC-AC direction, the flux balance equation described in step S1022 and the power transmission equation described in step S1023 are combined to form the inverter modulation model in the DC-AC direction, as shown in formula (9):

[0092] ; (9)

[0093] In the AC-DC direction, using the same method as steps S1022 and S1023, the flux balance equation described in step S1022 and the power transfer equation described in step S1023 are combined to form the inverter modulation model in the AC-DC direction, as shown in formula (10):

[0094] ; (10)

[0095] Step S103: Solve the inverter modulation model to obtain the modulation signal expression that includes the parameters of the filter inverter-side inductance and transformer leakage inductance.

[0096] In step S103, under the DC-AC transmission mode, the combined leakage inductance flux balance equation, filter inductance flux balance equation, and power transmission equation are solved to obtain the first modulation signal expression characterizing the duty cycle of the primary circuit, which includes the transformer leakage inductance and the filter inverter-side inductance variable, as shown in formula (11):

[0097] ; (11)

[0098] in, , , , The specific explanation is shown in formula (12):

[0099] ;(12)

[0100] Similarly, in the AC-DC transmission mode, the combined leakage inductance flux balance equation, filter inductance flux balance equation, and power transmission equation are solved to obtain the second modulation signal expression that characterizes the duty cycle of the secondary circuit, including the transformer leakage inductance and the filter inverter-side inductance variable, as shown in formula (13):

[0101] ; (13)

[0102] in, The expression is shown in formula (14):

[0103] ;(14)

[0104] It should be noted that the principle of the formula is as follows: k times the instantaneous value of the current AC current is set as the target current peak value. The difference between the target value and the current instantaneous value is calculated as the current increment required by the circuit. The product of the DC side voltage and the switching frequency is used as the driving reference for the current change. By dividing the required current increment by the driving reference, the proportion of conduction time that must be experienced to build this specific proportion of current increment under the current voltage driving capability is calculated, thereby obtaining the duty cycle of commutation state 4.

[0105] Step S2: Based on the modulation strategy, derive the zero-current switching current constraint and the zero-voltage switching current constraint of the switching transistor; combine the zero-current switching current constraint and the zero-voltage switching current constraint with the inverter modulation model to solve for the soft-switching constraint.

[0106] Step S2 includes at least the following steps:

[0107] Step S201: Derive the zero-current switching current constraint condition of the switching transistor based on the modulation strategy.

[0108] In the DC-AC transmission mode, a first constraint condition is established, limiting the duty cycle represented by the first modulation signal expression to a first numerical upper limit. The purpose of establishing this first numerical upper limit is to allow necessary physical time for current reset. Because during the DC-to-AC transmission process... The moment of zero-current turn-off is the critical point where energy transfer ends and demagnetization begins. The magnitude of the instantaneous current at this moment directly determines the time required to completely discharge the current to zero. Therefore, to achieve zero-current turn-off, the primary-side duty cycle must be strictly controlled. Its maximum value must be limited, which is the total period minus the value of the period itself. A predetermined minimum demagnetization time is used to ensure that the current has naturally returned to zero before the switch is activated.

[0109] In the AC-DC transmission mode, a second constraint condition is established, requiring the duty cycle represented by the second modulation signal expression to satisfy a second numerical upper limit. This second numerical upper limit is established to ensure zero-current turn-on during reverse transmission. In the AC-to-DC transmission mode, The moment is a critical node for shutting off the commutator and preparing for reset; the current amplitude at this moment... This determines the minimum physical time required for the current to return to zero. Therefore, it is necessary to... The size of the value establishes an upper limit threshold to limit the duty cycle on the secondary side. The range of motion is adjusted to prevent the current from failing to reset before the start of the next cycle due to excessive conduction time.

[0110] Simultaneously, the duty cycle represented by the second modulation signal expression is required to satisfy the third constraint condition of the preset timing logic constraint, because and All are total conduction time Internal sub-processes, to ensure that in To ensure correct zero-current switching occurs at all times, the intermediate commutation state must be guaranteed. The duration can be physically included after deducting the preceding commutation time. Within the remaining total time window.

[0111] Based on the above physical reset requirements and timing logic derivation, the first constraint, the second constraint, and the third constraint are combined to form the zero-current switching current constraint, as shown in formula (15):

[0112] ; (15)

[0113] Step S202: Derive the zero-voltage switching current constraint condition of the switching transistor based on the modulation strategy.

[0114] In step S202, based on the zero-voltage switching mechanism, the zero-voltage switching operations during modulation are first divided into two categories: the first category is zero-voltage turn-off achieved by utilizing the parasitic capacitance of the switching transistor. Since the capacitor voltage cannot change abruptly, this type of operation naturally meets the soft-switching requirements. The second category is zero-voltage clamping achieved by utilizing the freewheeling current of the body diode. This type of operation requires the inductive element to store sufficient energy to remove the charge on the parasitic capacitance, thereby maintaining the conduction of the body diode. This step focuses on establishing constraints for the second type of operation, which requires inductive energy support.

[0115] To quantify the aforementioned inductive energy requirements, we first define the minimum energy threshold required to satisfy soft switching. According to the principle of energy conservation, the energy stored in the inductive element must be greater than the energy stored in the parasitic capacitance, i.e., satisfy the following formula (16):

[0116] ; (16)

[0117] in, The reference energy is the parasitic capacitance of a single switching transistor. This represents the current value used by the inductive element to meet the energy requirements of parasitic capacitance charging and discharging during zero-voltage switching. The expression is shown in formula (17):

[0118] ; (17)

[0119] in, This is the parasitic capacitance of the switching transistor.

[0120] In DC-AC transmission mode, for the first set of switches on the primary side (i.e., S) 1, 4 A fourth constraint condition is established to require the turn-off current to satisfy the energy demand determined by parasitic capacitance; the physical mechanism of this constraint condition lies in: After S1 is turned off, the current in the equivalent inductance must maintain a unidirectional flow to charge the parasitic capacitance of S1 and simultaneously draw charge from the parasitic capacitance of S2. Only when the turn-off current... Only when the voltage is large enough can the drain-source voltage of S2 be completely reduced to zero, thereby turning on its body diode and creating conditions for subsequent zero-voltage turn-on. Therefore, the expression for the fourth constraint condition is derived as shown in equation (18):

[0121] ; (18)

[0122] In AC-DC transmission mode, for the second set of switches on the primary side (i.e., S) 2,3 A fifth constraint is established to ensure that the turn-off current meets the energy requirements determined by parasitic capacitance. The derivation logic of the fifth constraint is consistent with that of the fourth constraint, aiming to ensure that... Time S 2,3 During turn-off, the inductive current energy is sufficient to extract the parasitic capacitance charge of the complementary switch, achieving zero-voltage turn-on. Therefore, the expression for the fifth constraint condition is derived as shown in equation (19):

[0123] ; (19)

[0124] In AC-DC transmission mode, a sixth constraint condition is imposed to ensure that the current of the secondary-side switch meets the preset current polarity logic constraint when the switch operates. This sixth constraint condition requires that... At any given moment, the current polarity must remain positive to ensure that the body diode of the secondary-side switch S7 can properly freewheel. Therefore, the expression for the sixth constraint condition is derived as shown in equation (20):

[0125] ; (20)

[0126] The fourth, fifth, and sixth constraints are combined to form the zero-voltage switching current constraint. It should be noted that, as can be seen from formulas (19) and (20), the fifth constraint in this embodiment includes the sixth constraint.

[0127] Step S203: Combine the zero-current switching current constraint, the zero-voltage switching current constraint, and the inverter modulation model to solve for the soft-switching constraint.

[0128] In step S203, the zero-current switching (ZCS) current constraint conditions derived in step S201 and the zero-voltage switching (ZVS) current constraint conditions derived in step S202 are substituted into the inverter modulation model in step S1. The current variables in the constraint conditions are transformed into corresponding duty cycle variables using the modulation model, and then the final soft-switching constraint inequalities are obtained by solving them simultaneously, as shown in formula (21):

[0129] ; (twenty one)

[0130] Step S3: Based on the soft-switching constraints and the range of values ​​for the modulation signal expression, determine the coordinated range of values ​​for the filter inverter-side inductance and transformer leakage inductance that meet the soft-switching requirements; within the coordinated range, select the design parameters for the filter inverter-side inductance and transformer leakage inductance, and determine the remaining circuit parameters of the filter based on the reactive power requirements and grid harmonic standards.

[0131] Step S3 includes at least the following steps:

[0132] Step S301: Based on the soft-switching constraints and the range of values ​​for the modulation signal expression, determine the combined range of values ​​for the filter inverter-side inductance and the transformer leakage inductance that meet the soft-switching requirements.

[0133] In step S301, firstly, based on the soft-switching constraints obtained in step S2, and combined with the physical constraint that all modulation signals must be between 0 and 1 during the modulation process, the system operating state under different parameters is visually analyzed. For example... Figure 6 and Figure 7 As shown, in the AC-DC and DC-AC directions, time t is the x-axis, the filter inverter-side inductance L1 is the y-axis, and each modulation signal is the z-axis, with the preset transformer leakage inductance L1 as the x-axis and y-axis respectively. δ Within the range of values ​​and within the range of L1, plot the modulation signal waveform of the grid voltage for half a cycle. Figure 6 and Figure 7Based on U dc = 40V, f s =50kHz, U ac The parameters (t) = 311sin(100πt)V, n = 11, and transmission power P = 390W were plotted. During the plotting process, areas that did not meet the soft switching constraints or the range of values ​​were filled with red blocks to identify the non-constrained areas that hindered the normal operation of the system.

[0134] according to Figure 6 and Figure 7 The distribution characteristics of the red blocks indicate that the unconstrained region consists of three parts: unconstrained region 1 and unconstrained region 2 are located near the zero-crossing points on both sides of the sinusoidal signal, and unconstrained region 3 is located at the maximum value of the sinusoidal signal. Unconstrained region 1 and unconstrained region 2 are mainly determined by soft-switching constraints. When the modulation signal is located in this region, it indicates that the commutation time is too short, failing to guarantee smooth zero-current switching operation of the converter, and a voltage spike will be induced on the leakage inductance. Unconstrained region 3 is determined by the range of values ​​of the modulation signal. This region limits the operating range of the converter. When the transmission power is large enough that the modulation signal falls into this region, the duty cycle may exceed the limit, leading to converter failure.

[0135] Finally, the range of values ​​for the unconstrained regions is defined based on the aforementioned pattern of inductance parameter variation. As shown in the diagram, with the increase of the inductance L1 on the inverter side of the filter, the ranges of unconstrained regions 1, 2, and 3 all decrease; while with the increase of the transformer leakage inductance L... δ As the value decreases, the range of the unconstrained region 3 decreases. Therefore, the criterion for determining the range of cooperative values ​​is: within a given L... δ Next, we search for a critical value for L1 such that L1 values ​​above this critical value completely eliminate unconstrained regions 1 and 2, which characterize soft-switching failure. This L1 value satisfies the condition that unconstrained regions 1 and 2 do not exist. δ The parameter combination range is the range of values ​​that meet the soft-switching requirements.

[0136] Step S302: Select the design parameters of the filter inverter-side inductance and transformer leakage inductance within the range of coordinated values.

[0137] Step S302 includes at least the following steps:

[0138] Step S3021: Based on the influence of transformer leakage inductance on the effective operating range at the peak of the modulation signal, and with the goal of meeting the preset power transmission capacity, select the design parameters of transformer leakage inductance.

[0139] In step S3021, firstly, based on the collaborative value range analysis framework determined in step S301, attention is focused on the unconstrained region 3 located at the peak of the modulated signal waveform. For example... Figure 8 As shown, Figure 8 This shows the waveform of the modulated signal under a specific L1 value section, providing a basis for... Figure 6 and Figure 7 L1 takes the value 1×10 -3 H and 1.5×10 -3 H, L δ ∈[2×10 -4 H, 6×10 -4 Waveform diagrams of each modulation signal within the range H] Figure 8 (a) and Figure 8 In (b), L1 is taken as 1×10 -3 Modulation signal waveforms at time H under two power transmission directions Figure 8 (c) and Figure 8 In the middle (d), L1 is taken as 1.5 × 10. -3 The modulation signal waveforms under two power transmission directions at time H; where... Figure 8 (a) and Figure 8 (c) corresponds to the DC-AC direction. Figure 8 (b) and Figure 8 In the middle (d), the AC-DC direction is represented. The unconstrained region 3 is the area surrounded by the red curve. The physical significance of this region is that it limits the maximum transmission power of the converter. When the system attempts to transmit a large power, causing the modulation signal to fall into this region, the calculated duty cycle will exceed the physical limit, resulting in converter failure.

[0140] Next, the leakage inductance L of the transformer is analyzed. δ And the effect of the filter inverter-side inductance L1 on this region. Compare this to L1 being 1×10⁻⁶. -3 H and 1.5×10 -3 The waveform at time H shows that increasing L1 can reduce the range of the unconstrained region 3 to some extent (for example, when L1 increases from 1×10). -3 H increased to 1.5 × 10 -3 At time H, the L corresponding to the unconstrained region 3 under different power transmission directions δ The boundary conditions have been improved, with the lower limit increased from 4.03 × 10⁻⁶. -4 H changed to 4.33 × 10 -4 H), but the transformer leakage inductance L δ The value of L has a more direct impact on improving power transmission capability. Analysis shows that as L... δ The reduction in the value of 3 significantly reduces the range of the unconstrained region, which means that the effective operating range of the converter at the peak is expanded, thus enabling it to support higher power transmission.

[0141] Finally, to ensure the inverter operates stably at the preset output power of 390W and to prevent it from falling into the unconstrained region 3 at high power output due to excessive leakage inductance, a sufficiently small transformer leakage inductance value needs to be selected. Based on the above-mentioned variation of the unconstrained region 3 with parameters, and combined with the actual transformer design process, the design parameter L of the transformer leakage inductance was finally determined. δ 3×10 -4 H.

[0142] Step S3022: Based on the selected transformer leakage inductance design parameters, by analyzing the modulation signal waveform under different inductance values, determine the minimum inductance value that can make the modulation signal meet the soft switching constraint condition throughout the entire modulation cycle, and determine the minimum inductance value as the lower limit of the filter inverter side inductance value.

[0143] In step S3022, based on the selected design parameters for the transformer leakage inductance, the lower limit of the value of the filter inverter-side inductance L1 is further determined. Using... Figure 8 Compare the modulation signal waveforms and the distribution of the unconstrained region under different L1 values.

[0144] First, let's analyze the case where L1 has a small value. When L1 is 1 × 10... -3 At time H, the unconstrained regions 1 and 2, representing soft-switching failure, cover the entire range of transformer leakage inductance. This means that regardless of the leakage inductance matching, the system cannot meet the soft-switching constraint condition near the zero-crossing point of the modulation signal, and voltage spikes are generated. Next, the trend of L1's increase is analyzed. As L1 increases, the ranges of unconstrained regions 1 and 2 gradually decrease; when L1 increases to 1.5 × 10⁻⁶... - 3 At time H, unconstrained regions 1 and 2 under different power transmission directions completely disappear, indicating that the system can meet the soft-switching requirements throughout the entire modulation cycle. Finally, the critical value for satisfying the condition is determined. Based on the above variation patterns, under the selected transformer leakage inductance condition, by finding the critical point where unconstrained regions 1 and 2 just completely disappear, the minimum inductance value that allows the modulation signal to satisfy the soft-switching constraint condition throughout the entire modulation cycle is determined. According to... Figure 8 Based on the waveform analysis results of the modulated signal in different directions under the constraints shown, the lower limit L1_min of the inductance on the inverter side of the filter is determined to be 1.2 × 10⁻⁶. -3 H.

[0145] Step S3023: Based on the preset fundamental voltage drop limit requirement on the inductor on the inverter side of the filter, determine the upper limit of the value of the inductor on the inverter side of the filter using the fundamental voltage drop calculation formula.

[0146] In step S3023, when designing the inverter-side inductor L1 of the filter, in addition to meeting the lower limit constraint to ensure soft switching, the fundamental voltage drop across L1 should also be limited to avoid excessive voltage drop across the inductor affecting the inverter's output performance. This step uses the fundamental voltage drop calculation formula to determine the upper limit of the inductor value on the inverter side of the filter. The calculation formula is shown in formula (22):

[0147] ; (twenty two)

[0148] in, for Voltage drop on the grid and the effective value of the grid voltage The ratio, The fundamental angular frequency of the grid voltage. This is the effective value of the fundamental frequency of the inductor current. Calculating the above formula yields... 1.99×10 -3 H.

[0149] Step S3024: Within the numerical range formed by the lower limit and the upper limit of the value, the design parameters of the filter inverter-side inductor are selected by comprehensively considering the current ripple magnitude and system cost and volume factors.

[0150] In step S3024, based on the determined lower limit and upper limit, the effective value range of the filter inverter-side inductor L1 is determined to be [1.2 × 10⁻⁶]. -3 H, 1.99 × 10 -3 When selecting specific values ​​within this range, a balance must be struck between electrical performance and physical engineering constraints. Based on the aforementioned time-domain model analysis under different power transmission directions, it is known that the value of the filter inverter-side inductor L1 is negatively correlated with the inductor current ripple; that is, the larger L1 is, the smaller the current ripple on the inductor, which is beneficial for improving the output power quality. However, increasing the inductance value usually means increasing the core volume and the number of winding turns, which will lead to a corresponding increase in system cost, space occupation, weight, and losses. Therefore, after comprehensively considering the balance between current ripple suppression effect and system cost, volume, and other factors, the design parameter L1 of the filter inverter-side inductor is ultimately selected as 1.5 × 10⁻⁶ in this embodiment. -3 H.

[0151] Step S303: Determine the remaining circuit parameters of the filter based on the reactive power requirements and grid harmonic standards.

[0152] Step S303 includes at least the following steps:

[0153] Step S3031: Based on the preset ratio limit of reactive power injected into the filter capacitor to transmission power, and combined with the rated transmission power of the inverter and the grid voltage parameters, calculate and determine the value of the filter capacitor.

[0154] In step S3031, when designing the filter capacitor, the impact of reactive power injection on system performance needs to be comprehensively considered. The larger the capacitance of the filter capacitor, the more reactive power it injects into the grid, leading to increased system losses. To balance the filtering effect and system losses, the ratio of reactive power injected by the filter capacitor to the transmitted power is introduced as a design constraint parameter. Based on this ratio constraint, the filter capacitor... The formula for calculating the value is shown in formula (23):

[0155] ; (twenty three)

[0156] in, Reactive power injected into the filter capacitor and transmitted power The ratio, The fundamental angular frequency of the grid voltage. This is the effective value of the grid voltage.

[0157] Step S3032: Compare the harmonic distribution characteristics under different power transmission directions, select the power transmission direction with the largest harmonic content as the design condition, perform Fourier decomposition on the inverter modulation voltage under the design condition, and determine the main secondary harmonic with the largest amplitude.

[0158] In step S3032, since an additional control component d is introduced in the AC-DC direction (rectification condition) in the modulation strategy of this application, 22 (t), resulting in a significantly higher harmonic content in this direction compared to the DC-AC direction. To ensure that the filter design meets the requirements of the worst operating conditions, this embodiment selects the AC-DC direction as the design benchmark.

[0159] Specifically, firstly, the inverter modulation voltage in the next modulation cycle in the AC-DC direction is constructed. The time-domain expression of the voltage is given by equation (24), which is a piecewise function.

[0160] ; (twenty four)

[0161] Among them, the amplitude of the modulation voltage It is not a fixed value, but is determined by the DC bus voltage, the transformer turns ratio, and the inductance parameters determined in the preceding steps. The calculation formula is shown in formula (25):

[0162] (25)

[0163] The modulation voltage described by the above expression Fourier decomposition was performed. The analysis results show that the major subharmonic with the largest amplitude appears near the switching frequency (50.05kHz in this embodiment), and its amplitude is 38.91% of the fundamental voltage.

[0164] Step S3033: Based on the determined filter inverter-side inductor and filter capacitor, the grid current harmonic amplitude corresponding to the main harmonic is calculated using the transfer function of the grid current with respect to the inverter modulation voltage. The value that can make the grid current harmonic amplitude meet the preset grid harmonic standard is selected as the design parameter of the filter grid-side inductor.

[0165] In step S3033, after determining the values ​​of the filter inverter-side inductor and filter capacitor respectively, this step aims to determine the design parameters of the filter grid-side inductor.

[0166] First, a mathematical model of the LCL filter is constructed, and the grid-connected current is established. Modulation voltage of inverter transfer function Based on the circuit topology, the expression for the transfer function is shown in equation (26):

[0167] ; (26)

[0168] in, This represents the Laplace operator. Next, the transfer function is used to calculate the current response generated by the dominant second harmonic on the grid-connected side. Using the determined dominant second harmonic as the input excitation, and substituting it into the aforementioned transfer function, the amplitude of the grid-connected current harmonic at that specific frequency is calculated. Finally, referring to the amplitude limits for current harmonics in this frequency band in standards such as IEEE Std.929-2000 and IEEE Std.1547-2003, a system for... The inequality constraints. Under the premise of satisfying the harmonic constraint, solve and select appropriate... Numerical values. After calculation and selection, this embodiment finally determines the design parameters of the filter network-side inductor. 0.8×10 -3 H.

[0169] In summary, this application provides a method for the coordinated design of filters and leakage inductance in a single-stage bidirectional high-frequency isolation inverter. First, based on the inverter's basic design parameters and modulation strategy under bidirectional power transfer mode, an inverter modulation model is constructed to solve for the modulation signal expression containing the two key parameters: the filter inverter-side inductance and the transformer leakage inductance. Next, based on the modulation strategy, the zero-current switching and zero-voltage switching current constraints of the switching transistors are derived, and these physical constraints are combined with the aforementioned modulation model to calculate the system's soft-switching constraints. Then, combining the range of values ​​for the modulation signal expression and the soft-switching constraints, a coordinated range of values ​​for the filter inverter-side inductance and the transformer leakage inductance that simultaneously meets the design requirements is determined. Finally, within this coordinated range, specific inductor design parameters are selected, and based on reactive power requirements and grid harmonic standards, the remaining circuit parameters in the filter are determined, completing the overall circuit's coordinated design.

[0170] This application first constructs an inverter modulation model, unifying the physical quantities of the filter inverter-side inductance and transformer leakage inductance into a single mathematical expression. This achieves preliminary coupling of the two parameters in mathematical representation, transforming previously independent variables into interrelated functional variables and quantifying their combined influence on the modulation signal. Subsequently, soft-switching constraints are introduced as boundary criteria for parameter selection. The current constraints for zero-current switching and zero-voltage switching are explicitly derived, defining the physical boundaries for stable system operation. These physical boundaries are then substituted into the modulation model for simultaneous solution, searching for feasible solution sets that satisfy physical conservation laws in the multidimensional parameter space. Finally, a cooperative value range is obtained through the solution. This cooperative value range is the set of all parameter combinations that satisfy the soft-switching conditions. This means that any set of parameters falling within this range theoretically necessarily satisfies the preset soft-switching requirements. By selecting parameters within this defined range, designers can ensure that the circuit operating point remains within the soft-switching region under full power range and bidirectional flow conditions, thereby effectively suppressing soft-switching failure, increased losses, and electromagnetic interference caused by parameter mismatch.

[0171] One embodiment of this application also provides a filter and leakage inductance co-design system for a single-stage bidirectional high-frequency isolation inverter. The filter and leakage inductance co-design system for a single-stage bidirectional high-frequency isolation inverter includes at least the following modules:

[0172] The modulation signal solving module is used to obtain the basic design parameters of the single-stage bidirectional high-frequency isolation inverter, and construct the inverter modulation model according to the modulation strategy of the single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode. Solving the inverter modulation model yields the modulation signal expression containing the parameters of the filter inverter-side inductance and transformer leakage inductance.

[0173] The constraint solving module is used to derive the zero-current switching current constraint and the zero-voltage switching current constraint of the switching transistor based on the modulation strategy; and to solve the soft-switching constraint by combining the zero-current switching current constraint and the zero-voltage switching current constraint with the inverter modulation model.

[0174] The design parameter determination module is used to determine the coordinated value range of the filter inverter-side inductance and transformer leakage inductance that meet the soft-switching requirements based on the soft-switching constraints and the value range of the modulation signal expression; within the coordinated value range, the design parameters of the filter inverter-side inductance and transformer leakage inductance are selected, and the remaining circuit parameters of the filter are determined based on the reactive power requirements and grid harmonic standards.

[0175] For relevant details, please refer to the above method implementation examples.

[0176] Figure 9 This is a block diagram of an electronic device provided in one embodiment of the present application, the device including at least a processor 901 and a memory 902.

[0177] Processor 901 may include one or more processing cores, such as a quad-core, octa-core, or other multi-core processor. Processor 901 may be implemented using a general-purpose processor (CPU), a DSP (Digital Signal Processor), an FPGA (Field-Programmable Gate Array), or any combination of these hardware forms. In some embodiments, since this application involves the construction of an inverter modulation model, the solution of complex transcendental equations, and Fourier transform (FFT) analysis, processor 901 is specifically configured with a high-precision floating-point unit for performing cooperative domain calculations of the inductance of the inverter side filter and the leakage inductance of the transformer, soft-switching boundary determination, and harmonic amplitude calculation. Furthermore, processor 901 may also integrate a GPU (Graphics Processing Unit) for drawing and displaying the two-dimensional cooperative value range and modulation signal waveform, assisting designers in visually selecting parameters.

[0178] The memory 902 may include one or more computer-readable storage media, which may be non-transitory and used to store program code, system design parameters (such as rated power, grid voltage, switching frequency, etc.), and intermediate calculation results. The memory 902 may also include high-speed random access memory and non-volatile memory, such as disk or flash memory. In some embodiments, the non-transitory storage media in the memory 902 may store at least one instruction set, which is executed by the processor 901 to implement the filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter provided in this application. Specifically, this includes steps such as constructing an inverter modulation model, solving soft-switching constraints, determining the range of co-design values, and optimizing filter parameters based on harmonic standards.

[0179] In some embodiments, the electronic device may also optionally include a peripheral device interface and at least one peripheral device, with the processor 901, memory 902, and peripheral device interface connected via a bus or signal line. Indicatively, the peripheral device includes, but is not limited to: an input device (for inputting design specifications and constraints), a display interface (for displaying design results and parameter waveforms), and a communication module (for exporting the designed parameters to the inverter controller or simulation software), etc.

[0180] Of course, electronic devices may also include fewer or more components, and this embodiment does not limit this.

[0181] Optionally, this application also provides a computer-readable storage medium storing a program that is loaded and executed by a processor to implement the filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter described in the above method embodiments.

[0182] Optionally, this application also provides a computer product including a computer-readable storage medium storing a program, which is loaded and executed by a processor to implement the filter and leakage inductance collaborative design method of the single-stage bidirectional high-frequency isolation inverter described in the above method embodiments.

[0183] The technical features of the above embodiments can be combined in any way. For the sake of brevity, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0184] The above embodiments merely illustrate several implementation methods of this application, and while the descriptions are relatively specific and detailed, they should not be construed as limiting the scope of the invention patent. It should be noted that those skilled in the art can make various modifications and improvements without departing from the concept of this application, and these all fall within the protection scope of this application. Therefore, the protection scope of this patent application should be determined by the appended claims.

Claims

1. A method for co-designing the filter and leakage inductance of a single-stage bidirectional high-frequency isolation inverter, characterized in that, The design method includes the following steps: The basic design parameters of a single-stage bidirectional high-frequency isolation inverter are obtained, and an inverter modulation model is constructed based on the modulation strategy of the single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode. The modulation model is then solved to obtain a modulation signal expression that includes the parameters of the filter inverter-side inductance and transformer leakage inductance. Based on the modulation strategy, the zero-current switching current constraint and the zero-voltage switching current constraint of the switching transistor are derived; the zero-current switching current constraint and the zero-voltage switching current constraint are combined with the inverter modulation model to solve for the soft-switching constraint. Based on the soft-switching constraints and the range of values ​​for the modulation signal expression, determine the combined range of values ​​for the filter inverter-side inductance and transformer leakage inductance that meet the soft-switching requirements; select the design parameters for the filter inverter-side inductance and transformer leakage inductance within the combined range, and determine the remaining circuit parameters of the filter based on the reactive power requirements and grid harmonic standards. The construction of the inverter modulation model includes the following steps: Based on the modulation strategy, the instantaneous current expression of the transformer leakage inductance within a switching cycle is determined and transformed into the current increment expression for each time segment, and the current increment expression of the filter inverter-side inductor during the demagnetization stage is determined. Based on the inductor volt-second balance principle, the leakage inductor flux balance equation is established to describe the sum of the current increment expressions of the transformer leakage inductance within one switching cycle as zero, and the filter inductor flux balance equation is established to describe the sum of the current increment expressions of the filter inverter-side inductance during the magnetization and demagnetization stages as zero. Based on the current increment expressions of the transformer leakage inductance and the AC side voltage, a power transfer equation is constructed to characterize that the calculated actual power is equal to the reference power. In the bidirectional power transmission mode, the corresponding leakage inductance flux balance equation, the filter inductance flux balance equation, and the power transmission equation are combined to form the inverter modulation model.

2. The design method according to claim 1, characterized in that, Solving the inverter modulation model to obtain the modulation signal expression including the parameters of the filter inverter-side inductance and transformer leakage inductance includes the following steps: In DC-AC transmission mode, the combined leakage inductance flux balance equation, the filter inductance flux balance equation and the power transmission equation are solved to obtain a first modulation signal expression that characterizes the duty cycle of the primary circuit, including the transformer leakage inductance and the filter inverter-side inductance variable. In AC-DC transmission mode, the combined leakage inductance flux balance equation, the filter inductance flux balance equation, and the power transmission equation are solved to obtain a second modulation signal expression that includes the transformer leakage inductance and the filter inverter-side inductance variable and characterizes the duty cycle of the secondary circuit.

3. The design method according to claim 2, characterized in that, The zero-current switching current constraint condition for the switching transistor is derived through the following steps: In the DC-AC transmission mode, a first constraint condition is established to limit the duty cycle represented by the first modulation signal expression to satisfy a first numerical upper limit. In the AC-DC transmission mode, a second constraint condition is established that limits the duty cycle represented by the second modulation signal expression to meet the second numerical upper limit, and a third constraint condition is established that limits the duty cycle represented by the second modulation signal expression to meet the preset timing logic constraint. The first constraint, the second constraint, and the third constraint are combined to form the zero-current switching current constraint.

4. The design method according to claim 1, characterized in that, The zero-voltage switching current constraint condition for the switching transistor is derived through the following steps: In DC-AC transmission mode, for the first set of switches on the primary side, a fourth constraint condition is established to ensure that the turn-off current meets the requirements determined based on the parasitic capacitance energy demand. In AC-DC transmission mode, for the second set of switches on the primary side, a fifth constraint condition is established to limit the turn-off current to meet the requirements determined based on the parasitic capacitance energy demand, and a sixth constraint condition is established to limit the current of the secondary side switches to meet the preset current polarity logic constraint when the switches are in operation. The fourth, fifth, and sixth constraints are combined to form the zero-voltage switching current constraint.

5. The design method according to any one of claims 1-4, characterized in that, Selecting the design parameters for the filter inverter-side inductance and transformer leakage inductance within the range of the coordinated values ​​includes the following steps: Based on the influence of transformer leakage inductance on the effective operating range at the peak of the modulation signal, and with the goal of meeting the preset power transmission capacity, the design parameters of transformer leakage inductance are selected. Based on the selected design parameters of the transformer leakage inductance, by analyzing the waveform of the modulation signal under different inductance values, the minimum inductance value that can make the modulation signal meet the soft switching constraint condition throughout the entire modulation cycle is determined, and the minimum inductance value is determined as the lower limit of the inductance value on the inverter side of the filter. Based on the preset fundamental voltage drop limit requirement on the inductor on the inverter side of the filter, the upper limit of the value of the inductor on the inverter side of the filter is determined by using the fundamental voltage drop calculation formula. Within the numerical range formed by the lower limit and the upper limit, the design parameters of the filter inverter-side inductor are selected by comprehensively considering the current ripple magnitude and system cost and volume factors.

6. The design method according to claim 5, characterized in that, The determination of the remaining circuit parameters of the filter based on reactive power requirements and grid-connected harmonic standards includes the following steps: Based on the preset ratio limit of reactive power injected into the filter capacitor to transmission power, and combined with the rated transmission power of the inverter and the grid voltage parameters, the value of the filter capacitor is calculated and determined. By comparing the harmonic distribution characteristics under different power transmission directions, the power transmission direction with the largest harmonic content is selected as the design condition. The inverter modulation voltage under this design condition is subjected to Fourier decomposition to determine the main subharmonic with the largest amplitude. Based on the determined filter inverter-side inductor and filter capacitor, the grid current harmonic amplitude corresponding to the main subharmonic is calculated using the transfer function of the grid current with respect to the inverter modulation voltage. The value that enables the grid current harmonic amplitude to meet the preset grid harmonic standard is selected as the design parameter of the filter grid-side inductor.

7. A filter and leakage inductance co-design system for a single-stage bidirectional high-frequency isolation inverter, characterized in that, The design system includes: The modulation signal solving module is used to obtain the basic design parameters of the single-stage bidirectional high-frequency isolation inverter, and construct the inverter modulation model according to the modulation strategy of the single-stage bidirectional high-frequency isolation inverter in bidirectional power transmission mode. The module solves the inverter modulation model to obtain the modulation signal expression containing the parameters of the filter inverter-side inductance and transformer leakage inductance. The construction of the inverter modulation model includes the following steps: Based on the modulation strategy, the instantaneous current expression of the transformer leakage inductance within a switching cycle is determined and transformed into the current increment expression for each time segment, and the current increment expression of the filter inverter-side inductor during the demagnetization stage is determined. Based on the inductor volt-second balance principle, the leakage inductor flux balance equation is established to describe the sum of the current increment expressions of the transformer leakage inductance within one switching cycle as zero, and the filter inductor flux balance equation is established to describe the sum of the current increment expressions of the filter inverter-side inductance during the magnetization and demagnetization stages as zero. Based on the current increment expressions of the transformer leakage inductance and the AC side voltage, a power transfer equation is constructed to characterize that the calculated actual power is equal to the reference power. In the bidirectional power transmission mode, the corresponding leakage inductance flux balance equation, the filter inductance flux balance equation, and the power transmission equation are combined to form the inverter modulation model. The constraint solving module is used to derive the zero-current switching current constraint and the zero-voltage switching current constraint of the switching transistor based on the modulation strategy; and to solve the soft-switching constraint by combining the zero-current switching current constraint and the zero-voltage switching current constraint with the inverter modulation model. The design parameter determination module is used to determine the coordinated value range of the filter inverter-side inductance and transformer leakage inductance that meet the soft-switching requirements based on the soft-switching constraints and the value range of the modulation signal expression; select the design parameters of the filter inverter-side inductance and transformer leakage inductance within the coordinated value range; and determine the remaining circuit parameters of the filter based on the reactive power requirements and grid harmonic standards.

8. An electronic device, characterized in that, It includes a processor and a memory, wherein the memory stores a program that is loaded and executed by the processor to implement the filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter as described in any one of claims 1-6.

9. A computer-readable storage medium, characterized in that, The computer-readable storage medium stores a program that, when executed by a processor, is used to implement the filter and leakage inductance co-design method for a single-stage bidirectional high-frequency isolation inverter as described in any one of claims 1-6.

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