Improved parameter design method and system of wide gain clllc dc converter based on fundamental equivalent model

Abstract

The application discloses a wide gain CLLLCl DC converter improved parameter design method and system based on a fundamental equivalent model. First, a fundamental equivalent model of a CLLLCl bidirectional DC converter topology is established; second, based on a multi-effective working area constraint design process, an asymmetric design method is introduced, a multi-constraint condition considering transmission power, voltage gain, current stress and primary and secondary inductance ratio is constructed, and a boundary equation for resonant parameter design is derived; then, the transformer ratio and excitation inductance of the CLLLCl converter are determined according to the multi-constraint condition; finally, in the effective working area defined by the boundary equation, the parameter values of the primary and secondary resonant inductance and resonant capacitance are selected, so that the stable working point of the converter is in the effective working area defined by the multi-constraint condition. The application not only retains the simplicity of the fundamental model method, but also improves the accuracy and reliability of the parameter design through quantitative analysis of the sensitivity of each parameter.

Description

Technical Field

[0001] This invention belongs to the field of power electronic converter technology, and in particular relates to an improved parameter design method and system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model. Background Technology

[0002] Resonant converters are widely used in many industrial fields due to their soft-switching, small size, high power density, and high efficiency. The output voltage of the converter can be changed by altering the operating frequency of the switching transistor. Resonant converters have various topologies, among which the LLC (Inductor-Inductor-Capacitor) topology is widely used due to its simple structure and wide output voltage range. However, the LLC topology only enables unidirectional power transfer. To suit bidirectional power flow scenarios such as energy storage systems and vehicle-to-grid (V2G) applications, two separate hardware platforms must be designed to achieve bidirectional power flow. While this approach is relatively simpler to design, it increases both cost and converter size.

[0003] To address the aforementioned issues, a CLLLC (Capacitor-Inductor-Inductor-Capacitor) topology, balancing high efficiency and bidirectional characteristics, was proposed. On the primary side of the transformer, the CLLLC resonant converter also has an LLC network, but an additional LC (Inductor-Capacitor) resonant cavity is added on the secondary side, which is crucial for reverse power transfer. There are two design methods for the two series-connected LC resonant cavities on the primary and secondary sides: symmetrical and asymmetrical design. In the symmetrical design, the LC value on the secondary side is equal to the LC value on the primary side after a conversion operation. This greatly simplifies the converter gain formula, and only the primary side LC resonant cavity needs to be designed, simplifying the design. However, the symmetrical design results in a wide operating frequency range for the switching transistors. If poorly designed, the theoretical maximum switching frequency may exceed the upper limit of the switching transistor frequency, directly leading to design failure. Furthermore, the LC value is usually calculated under the condition of the minimum switching frequency, which significantly reduces the converter's power density, increases control complexity, and greatly reduces converter efficiency when the stable operating point is not at the resonant operating point. Conversely, asymmetric design takes into account the parameters of the primary and secondary LC resonant cavities. The equivalent value of the secondary LC resonator may not be equal to the value of the primary LC resonator, thus increasing the degree of freedom and effectively reducing the operating frequency range of the switching transistor. Its disadvantage is that the design is more complex and has more parameters.

[0004] Meanwhile, the precise design of resonant parameters directly determines the stability, efficiency, and device stress of the resonant converter. Currently, the mainstream design methods are the time-domain model method and the fundamental wave model method. The time-domain model method, by establishing a set of differential equations for the switching process of the transistor, can accurately describe the transient characteristics of current and voltage. However, this method requires solving high-order nonlinear differential equations, resulting in a long design cycle and requiring numerical simulation iterations, consuming significant computational resources. The fundamental wave model method, by equating the square wave voltage to a sinusoidal voltage at the fundamental frequency and ignoring other harmonic components, makes the analysis and calculation process more convenient and greatly improves design efficiency. However, when the switching frequency deviates from the resonant frequency, the design error caused by higher harmonics will increase. Based on this, a series of optimization design methods, such as the extended harmonic model method considering the influence of higher harmonics, intelligent optimization algorithms, methods considering the influence of the third harmonic, and simplified time-domain methods, have achieved certain optimization effects, but they still cannot simultaneously meet the design requirements of transistor stress, power limit, and efficiency inflection point.

[0005] In summary, while existing resonant converters achieve bidirectional power transmission at the topology level through CLLLC structures and offer improvements in design freedom, current mainstream methods such as the time-domain method, fundamental wave method, and their optimization algorithms all suffer from trade-offs in the crucial resonant parameter design phase. They struggle to simultaneously ensure design efficiency while accurately considering the converter's switching transistor stress, maximum power transmission capability, and efficiency characteristics over a wide operating range. Therefore, providing a resonant parameter design method that comprehensively optimizes these multi-objective requirements has become a pressing technical problem in this field. Summary of the Invention

[0006] To address the above technical problems, this invention provides an improved parameter design method and system for wide-gain CLLLC DC-DC converters based on a fundamental equivalent model.

[0007] The technical solution adopted by this invention to solve its technical problem is: An improved parameter design method for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model includes the following steps: S100: Based on the CLLLC bidirectional DC-DC converter topology, its fundamental equivalent model is established; S200: Based on the multi-effective working area constraint design process, an asymmetric design method is introduced to construct a multi-constraint condition that comprehensively considers transmission power, voltage gain, current stress and primary-secondary inductance ratio, and derives the boundary equation for resonant parameter design. S300: Determine the transformer turns ratio and magnetizing inductance of the CLLLC converter based on multiple constraints; S400: Within the effective operating region defined by the boundary equation, select the parameter values ​​of the primary resonant inductor, primary resonant capacitor, secondary resonant inductor, and secondary resonant capacitor so that the stable operating point of the converter is within the effective operating region jointly defined by multiple constraints.

[0008] Preferably, S100 specifically comprises: S110: Perform a Fourier transform on the input voltage of the resonant network to obtain its fundamental component and effective value; S120: Analyze the voltage at the midpoint of the two arms of the rectifier bridge to obtain the fundamental component and effective value of the resonant cavity output voltage; S130: Analyze the output current of the rectifier network to obtain its fundamental component and average value. Combined with the effective value of the fundamental component of the output voltage, the equivalent resistance of the secondary side of the converter can be obtained. S140: The resonant inductance, resonant capacitance, and equivalent load of the transformer secondary side are transferred to the primary side of the transformer to obtain the CLLLC fundamental equivalent circuit model.

[0009] Preferably, the input voltage of the resonant network in S100 U i ( t ) is the amplitude U in The fundamental component and its effective value of a square wave signal are calculated as follows: (1) (2) in, The instantaneous value of the input voltage to the resonant network. DC input voltage k For harmonic order, For switching frequency, For time variables, This represents the instantaneous value of the input voltage after the fundamental frequency is equivalent. Angular frequency, For the fundamental sine function, This represents the effective value of the input fundamental voltage. Midpoint voltage of the two arms of the rectifier bridge U o ( t The amplitude is ± U o The expression for the fundamental component of the resonant cavity output voltage of a square wave signal with a 50% duty cycle and the same frequency as the switching frequency is as follows: (3) in, This represents the instantaneous fundamental value of the resonant cavity output voltage. This is the DC output voltage. This is the effective value of the output fundamental voltage. The phase difference between the input voltage and the output voltage of the resonant cavity: The expressions for the fundamental component and average value of the output current of the rectifier network are as follows: (4) (5) in, The fundamental instantaneous value of the output current. This is the effective value of the fundamental current. It is the average value of the output current. It is the switching cycle; Equivalent resistance of the secondary side of the converter Calculated using the following formula: (6) in, This represents the fundamental effective value of the output voltage. This represents the fundamental effective value of the output current. This is the DC load resistor.

[0010] Preferably, the transmission power constraint in S200 is determined in the following manner: Set input voltage U i The range of variation is U i(min) ≤ U i ≤ U i(max) Output voltage U o The range of variation is U o(min) ≤ U o ≤ U o(max) The transformer turns ratio is By controlling the input voltage, the voltage conversion ratio is increased. k = U i / ( nU o If ) = 1, the following relation exists: (7) Based on the fundamental equivalent circuit, the output power P o The expression is as follows: (8) in, For input impedance, For the phasor of the output current, For the phasor of the input current, For the efficiency of the converter, For the phasor of the output voltage, For the phasor of the input voltage, The total impedance of the primary-side resonant network is... This is the excitation impedance of the transformer. The equivalent inductance value of the secondary-side resonant network impedance referred to the primary side is given by the following formula: This is the value of the load resistance referred to the primary side. Indicates a parallel relationship. For minimum output power, Minimum input voltage, This represents the maximum input impedance. This indicates that the operating frequency corresponding to this power point is the minimum frequency. This is the minimum frequency.

[0011] Preferably, the magnetizing inductor The value range is determined based on the operating conditions of the soft switch, as shown in the following formula: (9) in, This refers to the switching period when the converter operates at the resonant point. Dead time, For output capacitor The resonant condition of the converter is shown in the following equation: (10) in, The resonant frequency of the CLLLC converter. T =1 / f r , , For primary-side resonant inductance and capacitance, , For secondary resonant inductors and capacitors, It is the resonant angular frequency; Eliminate according to formula (10) , , input impedance Represented as and The function, i.e. Z in = Z in ( L rp , L rs ): (11) in, This is the equivalent capacitance value of the secondary-side resonant capacitor referred to the primary side. For the complex frequency in the Laplace transform, The equivalent inductance value of the secondary resonant inductor referred to the primary side; definition h Calculate the resonant inductance for the secondary side L' rs resonant inductor with primary side L rp Substituting the ratio into the above equation, we obtain the primary-side resonant cavity impedance. Z rp Equivalent impedance of the resonant cavity after secondary side reduction Z' rs Relationship: (12) Substitute the input impedance The expression yields: (13) in, This is the equivalent resistance of the load resistance referred to the primary side. For switching frequency, Minimum switching frequency, It is the imaginary unit in mathematics.

[0012] Preferably, the voltage gain constraint in S200 is determined by defining the forward gain coefficient. M f With reverse gain coefficient M b Its expression is as follows: (14) in, This is the secondary voltage. This is the primary voltage; Considering input voltage ripple, a 10% margin is allowed when designing the converter gain, and the following constraints must be met: (15) The maximum gain coefficient of the converter during forward operation is: M f(max) = U o(max) / U i(min) The minimum gain coefficient is M f(min) = U o(min) / U i(max).

[0013] Preferably, the current stress constraint condition in S200 is determined in the following manner: The converter must satisfy the following equation for normal operation: (16) in For stress current, This is the maximum current on the secondary side. This is the maximum primary current. This is the margin coefficient; Secondary current The calculation expression is as follows: when the switching frequency is at its minimum and the load condition is full, the secondary current reaches its maximum value: (17) in, The primary voltage phasor. This is the minimum value of the load resistance.

[0014] Preferably, the ratio of primary to secondary inductance in S200 h The constraints are determined in the following way: The gain-frequency curve of a CLLLC converter must satisfy the condition that, within the desired operating frequency range of the switching transistor, the converter gain is inversely proportional to the frequency. The constraints are as follows: (18).

[0015] Preferably, S400 specifically involves: making the CLLLC converter operate near its resonant point, at which point the converter's own gain is 1, and the transformer turns ratio is... It is expressed as the ratio of the rated voltages of the primary and secondary sides; combining equations (8), (11), (16), and (18), firstly, the ratio of the primary and secondary inductances is... h After determining the parameters, the power constraint design of the converter is carried out to ensure that the converter output power is twice the design specification. Next, the current stress constraint design is performed, requiring that under full load and maximum output gain, the primary and secondary resonant cavity currents be less than the current stresses of the corresponding high and low voltage side switches. Finally, the parameters after the design are verified to meet the requirements. h The constraint is whether the converter gain is negatively correlated with the switching frequency.

[0016] An improved parameter design system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model includes: The modeling module is used to establish the fundamental equivalent model of the CLLLC bidirectional DC-DC converter topology. The constraint construction module is used to construct multiple constraint conditions that comprehensively consider transmission power, voltage gain, current stress and primary-secondary inductance ratio based on the multi-effective working area constraint design process, introduce the asymmetric design method, and derive the boundary equations for resonant parameter design. The parameter determination module is used to determine the transformer turns ratio and magnetizing inductance of the CLLLC converter based on multiple constraints. The parameter selection module is used to select the parameter values ​​of the primary resonant inductor, primary resonant capacitor, secondary resonant inductor, and secondary resonant capacitor within the effective operating region defined by the boundary equation, so that the stable operating point of the converter is within the effective operating region jointly defined by multiple constraints.

[0017] The aforementioned improved parameter design method and system for wide-gain CLLLC DC-DC converters based on a fundamental equivalent model constructs a multi-effective operating region constraint design framework based on an asymmetric design method. It incorporates key performance indicators such as transmission power, voltage gain, current stress, and the primary-to-secondary inductance ratio into a unified design system. By quantitatively analyzing the sensitivity of parameters such as the resonant inductor to each performance indicator, the constraint boundaries of each parameter are derived and established. This method retains the simplicity of the fundamental model method while improving the accuracy and reliability of parameter design. Attached Figure Description

[0018] Figure 1 This is a flowchart of an improved parameter design method for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model, according to one embodiment of the present invention. Figure 2 This is a schematic diagram of a bidirectional CLLLC DC-DC converter topology in one embodiment of the present invention; Figure 3 This is a schematic diagram of the resonant cavity FHA model in the G2V state according to one embodiment of the present invention; Figure 4 This is a schematic diagram of the resonant cavity FHA model in V2G state according to one embodiment of the present invention; Figure 5 As shown in one embodiment of the present invention L rp and P out The relationship curve is shown in Figure (a). L rp When the variation range is from 10uH to 60uH L rp and P out The relationship curve diagram, (b) is L rp When the variation range is 80uH to 130uH L rp andP out Relationship curve diagram; Figure 6 As shown in one embodiment of the present invention M f and L rp The relationship curve is shown in Figure (a). f s =80kHz M f and L rp Relationship curve diagram, (b) is f s =450kHz M f and L rp Relationship curve diagram; Figure 7 In one embodiment of the present invention, the primary and secondary currents are... L rp The relationship curve is shown in Figure (a). f s At 80kHz, the primary and secondary currents and L rp The relationship curve diagram, (b) is f s At 450kHz, the primary and secondary currents and L rp Relationship curve diagram; Figure 8 Different in one embodiment of the present invention h A graph showing the relationship between switching frequency and forward gain under given conditions; Figure 9 This is a waveform diagram of the output voltage and primary current in a semi-physical simulation of underresonant operation in one embodiment of the present invention. Figure 10 This is a waveform diagram of the output voltage and primary current in a semi-physical simulation of the quasi-resonant operating state in one embodiment of the present invention; Figure 11 The output voltage and primary current waveforms in a semi-physical simulation of the over-resonance operating state are shown in one embodiment of the present invention. Detailed Implementation

[0019] To enable those skilled in the art to better understand the technical solution of the present invention, the present invention will be further described in detail below with reference to the accompanying drawings.

[0020] This invention provides an improved parameter design method and system for wide-gain CLLLC DC-DC converters based on a fundamental equivalent model. It constructs a multi-effective operating region constraint design framework based on an asymmetric design method, incorporating key performance indicators such as transmission power, voltage gain, current stress, and the primary-to-secondary inductance ratio into a unified design system. By quantitatively analyzing the sensitivity of parameters such as the resonant inductor to each performance indicator, the constraint boundaries of each parameter are derived and established. This method retains the simplicity of the fundamental model method while improving the accuracy and reliability of parameter design.

[0021] In one embodiment, such as Figure 1 As shown, an improved parameter design method for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model includes the following steps: S100: Based on the CLLLC bidirectional DC-DC converter topology, its fundamental equivalent model is established; S200: Based on the multi-effective working area constraint design process, an asymmetric design method is introduced to construct a multi-constraint condition that comprehensively considers transmission power, voltage gain, current stress and primary-secondary inductance ratio, and derives the boundary equation for resonant parameter design. S300: Determine the transformer turns ratio and magnetizing inductance of the CLLLC converter based on multiple constraints; S400: Within the effective operating region defined by the boundary equation, select the parameter values ​​of the primary resonant inductor, primary resonant capacitor, secondary resonant inductor, and secondary resonant capacitor so that the stable operating point of the converter is within the effective operating region jointly defined by multiple constraints.

[0022] Specifically, addressing the issue that mainstream CLLLC converter design methods struggle to balance accuracy and efficiency, an asymmetric design method is introduced based on the fundamental equivalent model. A multi-effective operating region constraint design process is constructed. By quantitatively analyzing the transmission power, voltage gain, current stress, and primary-secondary inductance ratio constraint boundaries, boundary equations for resonant parameter design are obtained. This method retains the engineering feasibility of the fundamental model method while solving the performance deficiencies caused by a single constraint method, thus improving the design accuracy of resonant parameters.

[0023] In one embodiment, S100 specifically includes: S110: Perform a Fourier transform on the input voltage of the resonant network to obtain its fundamental component and effective value; S120: Analyze the voltage at the midpoint of the two arms of the rectifier bridge to obtain the fundamental component and effective value of the resonant cavity output voltage; S130: Analyze the output current of the rectifier network to obtain its fundamental component and average value. Combined with the effective value of the fundamental component of the output voltage, the equivalent resistance of the secondary side of the converter can be obtained. S140: The resonant inductance, resonant capacitance, and equivalent load of the transformer secondary side are transferred to the primary side of the transformer to obtain the CLLLC fundamental equivalent circuit model.

[0024] Specifically, such as Figure 2 The diagram shows the fundamental equivalent model of a bidirectional CLLLC DC-DC converter topology. The fundamental equivalent models are shown for the converter operating in the forward direction (charging the battery, Grid-to-Vehicle, G2V) and the fundamental equivalent models for the converter operating in the reverse direction (discharging the battery, Grid-to-Vehicle, V2G), respectively. Figure 3 and Figure 4 As shown in Table 1, the meanings of the parameters in the circuit are given in Table 1, and the corresponding equivalent values ​​can be derived from the following formula: .

[0025] Table 1. Meaning of each parameter in the fundamental wave equivalent model:

[0026] Based on the equivalent circuit model after conversion, the forward and reverse gains of the CLLLC DC-DC converter can be easily calculated, and the specific expressions are shown in the following formula: ; .

[0027] In one embodiment, the input voltage of the resonant network in S100 U i ( t ) is the amplitude U in The fundamental component and its effective value of a square wave signal are calculated as follows: (1) (2) in, The instantaneous value of the input voltage to the resonant network. DC input voltage k For harmonic order, For switching frequency, For time variables, This represents the instantaneous value of the input voltage after the fundamental frequency is equivalent. Angular frequency, For the fundamental sine function, This represents the effective value of the input fundamental voltage. When the power flows in the forward direction, only the full-bridge rectifier circuit composed of the secondary diodes operates. After passing through the filter, the output voltage is [value missing]. U outThe DC voltage output, when the dead time is ignored, is the voltage at the midpoint of the two arms of the rectifier bridge. U o ( t The amplitude is ± U o The expression for the fundamental component of the resonant cavity output voltage of a square wave signal with a 50% duty cycle and the same frequency as the switching frequency is as follows: (3) in, This represents the instantaneous fundamental value of the resonant cavity output voltage. This is the DC output voltage. This is the effective value of the output fundamental voltage. The phase difference between the input voltage and the output voltage of the resonant cavity: Since the load applied to the filter network is a purely resistive load, the phase difference between the output current and the output voltage can be ignored. The expressions for the fundamental component and average value of the output current of the rectifier network in S120 are as follows: (4) (5) in, The fundamental instantaneous value of the output current. This is the effective value of the fundamental current. It is the average value of the output current. It is the switching cycle; Equivalent resistance of the secondary side of the converter in S130 Calculated using the following formula: (6) in, This represents the fundamental effective value of the output voltage. This represents the fundamental effective value of the output current. This is the DC load resistor.

[0028] In one embodiment, the transmission power constraint in S200 is determined in the following manner: During the operation of a CLLLC, changes in system state (such as changes in battery capacity) will cause changes in output voltage. Let the range of change be... U o(min) ≤ U o ≤ U o(max) Since the transformer turns ratio is fixed, the input voltage is usually controlled in order to reduce circulating current and improve output efficiency. U i The change makes the voltage transformation ratio... k = Ui / ( nU o )=1, setting the input voltage variation range to U i(min) ≤ U i ≤ U i(max) Then the following relation exists: (7) As can be seen from the macroscopic characteristics of CLLLC, the operating frequency of the switching transistor is inversely proportional to the output power. Therefore, under the condition of the lowest switching frequency, the output power is... P o It reaches the theoretical maximum value. Based on the fundamental equivalent circuit, the output power... P o The expression is as follows: (8) in, For input impedance, For the phasor of the output current, For the phasor of the input current, For the efficiency of the converter, For the phasor of the output voltage, For the phasor of the input voltage, The total impedance of the primary-side resonant network is... This is the excitation impedance of the transformer. The equivalent inductance value of the secondary-side resonant network impedance referred to the primary side is given by the following formula: This is the value of the load resistance referred to the primary side. Indicates a parallel relationship. For minimum output power, Minimum input voltage, This represents the maximum input impedance. This indicates that the operating frequency corresponding to this power point is the minimum frequency. This is the minimum frequency.

[0029] Specifically, such as Figure 5 As shown, Figure 5 The switching frequency is given. f s =80kHz, different h Under the condition of value, the maximum output power P out With the primary resonant cavity inductance L rp The curve shows the change. It can be seen that when... L rp When smaller P M FollowL rp Increase and increase, when L rp When larger P M Follow L rp Increase and decrease, and L rp Only within a certain range is it satisfied P M(min) Greater than the maximum value required by the theoretical load P max In this application, this area is defined as the Effective Operation Area (EOA).

[0030] In one embodiment, the magnetizing inductor The value range is determined based on the operating conditions of the soft switch, as shown in the following formula: (9) in, This refers to the switching period when the converter operates at the resonant point. Dead time, For output capacitor; The resonant condition of the converter is shown in the following equation: (10) in, The resonant frequency of the CLLLC converter. T =1 / f r , , For primary-side resonant inductance and capacitance, , For secondary resonant inductors and capacitors, It is the resonant angular frequency; Equation (10) gives the resonant condition of the converter. In actual design, since the resonant frequency is given, the following equation can be used to establish a coupling relationship between the inductor and capacitor of the primary and secondary resonant cavities, thereby eliminating the unknown quantity of capacitance and further optimizing the design.

[0031] Eliminate according to formula (10) , , input impedance Represented as and The function, i.e. Z in = Z in ( L rp , Lrs ): (11) in, This is the equivalent capacitance value of the secondary-side resonant capacitor referred to the primary side. For the complex frequency in the Laplace transform, The equivalent inductance value of the secondary resonant inductor referred to the primary side; definition h Calculate the resonant inductance for the secondary side L' rs resonant inductor with primary side L rp Substituting the ratio into the above equation, we obtain the primary-side resonant cavity impedance. Z rp Equivalent impedance of the resonant cavity after secondary side reduction Z' rs Relationship: (12) Substitute the input impedance The expression yields: (13) in, This is the equivalent resistance of the load resistance referred to the primary side. For switching frequency, Minimum switching frequency, It is the imaginary unit in mathematics.

[0032] In one embodiment, the voltage gain constraint in S200 is determined by defining the forward gain coefficient. M f With reverse gain coefficient M b Its expression is as follows: (14) in, This is the secondary voltage. This is the primary voltage; The input voltage typically comes from the PFC circuit output and includes a 10% voltage ripple. Therefore, when designing the converter gain, a 10% margin should also be allowed, and the following constraints must be met: (15) The maximum gain coefficient of the converter during forward operation is: M f(max) = U o(max) / U i(min) The minimum gain coefficient is M f(min)= U o(min) / U i(max) .

[0033] Specifically, such as Figure 6 As shown, Figure 6 Give different h Under the condition of value, the primary resonant inductance L rp With the positive gain of the converter M f The relationship curves are given, with the switching frequencies taken as the minimum and maximum design parameters, respectively. Finally, the EOA is marked on the graph according to the constraint conditions of equation (15). When designing a CLLLC DC-DC converter, the converter needs to operate in the inductive region, that is, the switching frequency is inversely proportional to the converter gain. Therefore, the converter gain reaches its maximum value when the switching frequency is at its minimum, and vice versa when the switching frequency is at its maximum.

[0034] In one embodiment, to ensure safe system operation, it is necessary to ensure that the maximum current stress of the converter during operation is less than the maximum inrush current that the switching transistor can withstand, with a certain margin, taking into account the secondary current. I s Obviously greater than the primary current I p Generally, it meets the requirements. I s = nI p To simplify the system operation constraints, the current stress constraint in S200 is determined in the following way: The converter must satisfy the following equation for normal operation: (16) in For stress current, This is the maximum current on the secondary side. This is the maximum primary current. This is the margin coefficient; Specifically, such as Figure 7 As shown, Figure 7 The maximum value of the primary and secondary currents and the primary resonant inductance are given. L rp The relationship curve is plotted, and the EOA region is marked. Current stress is the maximum current value reached by the converter at a certain moment, characterizing the impact effect of current on the switching devices. Therefore, it is necessary to calculate the maximum current values ​​corresponding to the primary and secondary sides of the converter during stable operation, ensuring that both are less than the maximum current that the switching transistors can withstand. To ensure safe system operation, it is necessary to ensure that the maximum current stress of the converter during operation is less than the maximum inrush current that the switching transistors can withstand, with a certain margin, considering the secondary current...I s Obviously greater than the primary current I p Generally, it meets the requirements. I s = nI p To simplify the system's operational constraints, the converter needs to satisfy equation (16) during normal operation, where... λ This is the margin factor, usually chosen to be 1.5 to 2.

[0035] Secondary current The calculation expression is as follows: when the switching frequency is at its minimum and the load condition is full, the secondary current reaches its maximum value: (17) in, The primary voltage phasor. This is the minimum value of the load resistance.

[0036] In one embodiment, under normal circumstances, the gain-frequency curve of the CLLLC converter falls within the desired operating frequency range of the switching transistor. f s ∈[ f s(min) ，f s(max) The converter gain is inversely proportional to the frequency, but when... h When the value is changed, the converter gain may no longer be inversely proportional to the operating frequency of the switching transistor. That is, when the switching frequency is reduced, the converter gain may actually decrease. To prevent this from happening, the ratio of the primary and secondary inductances in S200 is... h The constraints are determined in the following way: The gain-frequency curve of a CLLLC converter must satisfy the condition that, within the desired operating frequency range of the switching transistor, the converter gain is inversely proportional to the frequency. The constraints are as follows: (18).

[0037] Specifically, such as Figure 8 As shown, Figure 8 Different h Under the given conditions, the relationship curve between the switching frequency and the converter's G2V operating condition can be observed when... h As the value increases, the gain curve no longer decreases monotonically with frequency, causing the converter to fail to meet the high gain requirement. To prevent the above fault, the design should follow the constraint of equation (18) and select an appropriate value. h The value, within the specified switching frequency range, meets the converter gain requirements.

[0038] In one embodiment, S400 specifically involves: operating the CLLLC converter near its resonant point, at which point the converter's gain is 1 and the transformer turns ratio is... This is expressed as the ratio of the rated voltages of the primary and secondary sides (the transformer turns ratio has been determined in S300, and the magnetizing inductance is calculated using equation (9)). L m (value); combining equations (8), (11), (16), and (18), by selecting appropriate L rp 、C rp 、L m 、L rs 、 C rs This value ensures that the converter's stable operating point is within the effective operating region. Firstly, the ratio of the primary to secondary inductance... h Make a determination ( h An excessively large value will cause the converter gain curve and switching frequency relationship curve to no longer decrease monotonically, thus preventing the converter from reaching its maximum gain operating state. However, if... h If the value is too small, it will increase the inductance of the secondary inductor, resulting in a larger converter size and reduced power density. Next, power constraint design of the converter will be performed, and the primary resonant inductor... L rp Within a certain range of variation, the converter output power increases with the increase of the inductance value. Generally, the converter output power is designed with a margin of twice the design specification to compensate for the design error between the fundamental wave model and the actual situation. Subsequently, current stress constraint design is performed, requiring that the primary and secondary resonant cavity currents of the converter under full load and maximum output gain conditions be less than the current stresses of the corresponding high and low voltage side switching transistors. Finally, it is verified whether the parameters after the design are completed meet the requirements. h The constraint is whether the converter gain is negatively correlated with the switching frequency.

[0039] Specifically, Figure 9 , Figure 10 , Figure 11 To verify the simulation results of the hardware-in-the-loop (HIL) simulation platform, the control, data acquisition, and waveform observation of the converter were achieved through multi-module collaboration. Based on this experimental platform, a load was set up. R L =15kHz, and the three states of underresonance, quasi-resonance and overresonance are experimentally verified when the converter is in forward operation. The output voltage and power are observed to see if they meet the design standards. The accuracy of the design method of this invention is verified by detecting the primary and secondary currents. Figure 9The waveforms of the primary current and output voltage when the converter operates in underresonant mode are presented. For ease of observation, the original values ​​are multiplied by a variation factor of 0.1. It can be seen that the converter switching frequency is 15.3kHz, the maximum primary current is 32.6A, the output voltage is 51.2V, and the ripple magnitude is 0.46V, which meets the design requirements. Figure 10 The output voltage and primary current waveforms of the converter are given when it is operating in a quasi-resonant state. The output voltage is stable at 48V with a ripple of 0.48V, the output current is a sinusoidal waveform with an amplitude of 38A, and the switching frequency is 19.94kHz. These parameters meet the design specifications and are consistent with the theoretical design results. Figure 11 The output voltage and primary current waveforms of the converter are given when it is operating in the over-resonance state. When the switching frequency reaches 30kHz, the output voltage reaches the minimum value of 36V, and the output current is close to a triangular wave with an amplitude of 22.6A, which meets the design specifications.

[0040] The aforementioned improved parameter design method for wide-gain CLLLC DC-DC converters based on a fundamental equivalent model introduces an asymmetric design method. This method constructs a multi-effective operating region constraint design process that comprehensively considers factors such as switching transistor voltage and current stress, and the converter's maximum operating power, deriving boundary equations for resonant parameter design. Finally, in the simulation main loop, the CLLLC converter parameters are selected according to requirements to conform to the aforementioned multiple boundary equations. This method maintains the simplicity of the fundamental model method while solving the performance deficiencies caused by single-constraint methods, significantly improving design accuracy. Case studies show that the proposed method meets design requirements in terms of output voltage ripple and device stress, providing a reliable solution for its engineering application in bidirectional power scenarios.

[0041] In one embodiment, an improved parameter design system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model is also provided, including: The modeling module is used to establish the fundamental equivalent model of the CLLLC bidirectional DC-DC converter topology. The constraint construction module is used to construct multiple constraint conditions that comprehensively consider transmission power, voltage gain, current stress and primary-secondary inductance ratio based on the multi-effective working area constraint design process, introduce the asymmetric design method, and derive the boundary equations for resonant parameter design. The parameter determination module is used to determine the transformer turns ratio and magnetizing inductance of the CLLLC converter based on multiple constraints. The parameter selection module is used to select the parameter values ​​of the primary resonant inductor, primary resonant capacitor, secondary resonant inductor, and secondary resonant capacitor within the effective operating region defined by the boundary equation, so that the stable operating point of the converter is within the effective operating region jointly defined by multiple constraints.

[0042] Specific limitations regarding the improved parameter design system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model can be found in the limitations of the improved parameter design method for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model described above, and will not be repeated here. Each module in the aforementioned improved parameter design system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model can be implemented entirely or partially through software, hardware, or a combination thereof. These modules can be embedded in hardware or independent of the processor in a computer device, or stored in software in the memory of a computer device, so that the processor can call and execute the corresponding operations of each module.

[0043] The above provides a detailed description of the improved parameter design method and system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model, as provided by this invention. Specific examples have been used to illustrate the principles and implementation methods of this invention; the descriptions of these embodiments are merely for the purpose of helping to understand the core ideas of this invention. It should be noted that those skilled in the art can make various improvements and modifications to this invention without departing from its principles, and these improvements and modifications also fall within the protection scope of the claims of this invention.

Claims

1. An improved parameter design method for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model, characterized in that, Includes the following steps: S100: Based on the CLLLC bidirectional DC-DC converter topology, its fundamental equivalent model is established; S200: Based on the multi-effective working area constraint design process, an asymmetric design method is introduced to construct a multi-constraint condition that comprehensively considers transmission power, voltage gain, current stress and primary-secondary inductance ratio, and derives the boundary equation for resonant parameter design. S300: Determine the transformer turns ratio and magnetizing inductance of the CLLLC converter based on multiple constraints; S400: Within the effective operating region defined by the boundary equation, select the parameter values ​​of the primary resonant inductor, primary resonant capacitor, secondary resonant inductor, and secondary resonant capacitor so that the stable operating point of the converter is within the effective operating region jointly defined by multiple constraints.

2. The method according to claim 1, characterized in that, S100 specifically refers to: S110: Perform a Fourier transform on the input voltage of the resonant network to obtain its fundamental component and effective value; S120: Analyze the voltage at the midpoint of the two arms of the rectifier bridge to obtain the fundamental component and effective value of the resonant cavity output voltage; S130: Analyze the output current of the rectifier network to obtain its fundamental component and average value. Combined with the effective value of the fundamental component of the output voltage, the equivalent resistance of the secondary side of the converter can be obtained. S140: The resonant inductance, resonant capacitance, and equivalent load of the transformer secondary side are transferred to the primary side of the transformer to obtain the CLLLC fundamental equivalent circuit model.

3. The method according to claim 2, characterized in that, Input voltage of the resonant network in S100 U i ( t ) is the amplitude U in The fundamental component and its effective value of a square wave signal are calculated as follows: (1) (2) in, The instantaneous value of the input voltage to the resonant network. DC input voltage k For harmonic order, For switching frequency, For time variables, This represents the instantaneous value of the input voltage after the fundamental frequency is equivalent. Angular frequency, For the fundamental sine function, This represents the effective value of the input fundamental voltage. Midpoint voltage of the two arms of the rectifier bridge U o ( t The amplitude is ± U o The expression for the fundamental component of the resonant cavity output voltage of a square wave signal with a 50% duty cycle and the same frequency as the switching frequency is as follows: (3) in, This represents the instantaneous fundamental value of the resonant cavity output voltage. This is the DC output voltage. This is the effective value of the output fundamental voltage. The phase difference between the input voltage and the output voltage of the resonant cavity: The expressions for the fundamental component and average value of the output current of the rectifier network are as follows: (4) (5) in, The fundamental instantaneous value of the output current. This is the effective value of the fundamental current. It is the average value of the output current. It is the switching cycle; Equivalent resistance of the secondary side of the converter Calculated using the following formula: (6) in, This represents the fundamental effective value of the output voltage. This represents the fundamental effective value of the output current. This is the DC load resistor.

4. The method according to claim 3, characterized in that, The transmission power constraints in S200 are determined in the following manner: Set input voltage U i The range of variation is U i(min) ≤ U i ≤ U i(max) Output voltage U o The range of variation is U o(min) ≤ U o ≤ U o(max) The transformer turns ratio is By controlling the input voltage, the voltage conversion ratio is increased. k = U i / ( nU o If ) = 1, the following relation exists: (7) Based on the fundamental equivalent circuit, the output power P o The expression is as follows: (8) in, For input impedance, For the phasor of the output current, For the phasor of the input current, For the efficiency of the converter, For the phasor of the output voltage, For the phasor of the input voltage, The total impedance of the primary-side resonant network is... This is the excitation impedance of the transformer. The equivalent inductance value of the secondary-side resonant network impedance referred to the primary side is given by the following formula: This is the value of the load resistance referred to the primary side. Indicates a parallel relationship. For minimum output power, Minimum input voltage, This represents the maximum input impedance. This indicates that the operating frequency corresponding to this power point is the minimum frequency. This is the minimum frequency.

5. The method according to claim 4, characterized in that, Magnetizing inductor The value range is determined based on the operating conditions of the soft switch, as shown in the following formula: (9) in, This refers to the switching period when the converter operates at the resonant point. Dead time, For output capacitor; The resonant condition of the converter is shown in the following equation: (10) in, The resonant frequency of the CLLLC converter. T =1 / f r , , For primary-side resonant inductance and capacitance, , For secondary resonant inductors and capacitors, It is the resonant angular frequency; Eliminate according to formula (10) , , input impedance Represented as and The function, i.e. Z in = Z in ( L rp , L rs ): (11) in, This is the equivalent capacitance value of the secondary-side resonant capacitor referred to the primary side. For the complex frequency in the Laplace transform, The equivalent inductance value of the secondary resonant inductor referred to the primary side; definition h Calculate the resonant inductance for the secondary side L' rs resonant inductor with primary side L rp Substituting the ratio into the above equation, we obtain the primary-side resonant cavity impedance. Z rp Equivalent impedance of the resonant cavity after secondary side reduction Z' rs Relationship: (12) Substitute the input impedance The expression yields: (13) in, This is the equivalent resistance of the load resistance referred to the primary side. For switching frequency, Minimum switching frequency, It is the imaginary unit in mathematics.

6. The method according to claim 5, characterized in that, The voltage gain constraint in S200 is determined by defining the forward gain coefficient. M f With reverse gain coefficient M b Its expression is as follows: (14) in, This is the secondary voltage. This is the primary voltage; Considering input voltage ripple, a 10% margin is allowed when designing the converter gain, and the following constraints must be met: (15) The maximum gain coefficient of the converter during forward operation is: M f(max) = U o(max) / U i(min) The minimum gain coefficient is M f(min) = U o(min) / U i(max) .

7. The method according to claim 6, characterized in that, The current stress constraint conditions in S200 are determined in the following manner: The converter must satisfy the following equation for normal operation: (16) in For stress current, This is the maximum current on the secondary side. This is the maximum primary current. This is the margin coefficient; Secondary current The calculation expression is as follows: when the switching frequency is at its minimum and the load condition is full, the secondary current reaches its maximum value: (17) in, The primary voltage phasor. This is the minimum value of the load resistance.

8. The method according to claim 7, characterized in that, The ratio of primary to secondary inductance in S200 h The constraints are determined in the following way: The gain-frequency curve of a CLLLC converter must satisfy the condition that, within the desired operating frequency range of the switching transistor, the converter gain is inversely proportional to the frequency. The constraints are as follows: (18)。 9. The method according to claim 8, characterized in that, Specifically, S400 enables the CLLLC converter to operate near its resonant point, at which point the converter's gain is 1 and the transformer turns ratio is [value missing]. It is expressed as the ratio of the rated voltages of the primary and secondary sides; combining equations (8), (11), (16), and (18), firstly, the ratio of the primary and secondary inductances is... h After determining the parameters, the power constraint design of the converter is carried out to ensure that the converter output power is twice the design specification. Next, the current stress constraint design is performed, requiring that under full load and maximum output gain, the primary and secondary resonant cavity currents be less than the current stresses of the corresponding high and low voltage side switches. Finally, the parameters after the design are verified to meet the requirements. h The constraint is whether the converter gain is negatively correlated with the switching frequency.

10. An improved parameter design system for a wide-gain CLLLC DC-DC converter based on a fundamental equivalent model, characterized in that, include: The modeling module is used to establish the fundamental equivalent model of the CLLLC bidirectional DC-DC converter topology. The constraint construction module is used to construct multiple constraint conditions that comprehensively consider transmission power, voltage gain, current stress and primary-secondary inductance ratio based on the multi-effective working area constraint design process, introduce the asymmetric design method, and derive the boundary equations for resonant parameter design. The parameter determination module is used to determine the transformer turns ratio and magnetizing inductance of the CLLLC converter based on multiple constraints. The parameter selection module is used to select the parameter values ​​of the primary resonant inductor, primary resonant capacitor, secondary resonant inductor, and secondary resonant capacitor within the effective operating region defined by the boundary equation, so that the stable operating point of the converter is within the effective operating region jointly defined by multiple constraints.

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