Two-stage phase current modulation method for switched reluctance generator based on voltage balance

Abstract

This invention relates to the field of switched reluctance generator (SMR) technology, specifically to a two-stage phase current modulation method for SMRs based on voltage balance, comprising the following steps: S1, modeling the relationship between the moving electromotive force and the demagnetizing voltage; S2, designing the phase current mode switching criterion; S3, current chopping control during the excitation stage; S4, voltage balance modulation during the generation stage; S5, tuning the dual closed-loop control parameters; S6, configuring initial parameters for efficiency optimization; S7, dynamically constructing the optimization interval; S8, dynamically verifying mode switching; and S9, adapting system integration parameters. This invention establishes a quantitative model of the moving electromotive force and the demagnetizing voltage and derives an analytical expression for the duty cycle, achieving calculable and constrained control of the duty cycle D. This provides a precise basis for voltage balance modulation, thereby improving modulation accuracy and adaptability to operating conditions, and reducing errors and instabilities caused by empirical settings.

Description

Technical Field

[0001] This invention relates to the field of switched reluctance generator technology, and more specifically to a two-stage phase current modulation method for switched reluctance generators based on voltage balance. Background Technology

[0002] Switched reluctance generators (SRGs) are widely used in wind power, tidal power and other renewable energy fields due to their advantages such as simple structure, high efficiency and low maintenance cost. However, SRGs face problems in practical applications, such as low phase current control accuracy, large commutation impact, severe voltage fluctuation and low system efficiency. Traditional SRG modulation methods mostly rely on fixed modulation strategies and fail to flexibly adjust the current mode and regulation parameters according to different operating conditions, resulting in unsatisfactory current control accuracy and susceptibility to system disturbances. Existing control algorithms lack sufficient adaptability when dealing with rapidly changing loads and speeds and cannot effectively optimize system efficiency, especially under low speed or high load conditions, where current control and system stability decrease significantly.

[0003] Existing control methods for switched reluctance generators typically employ simple current-mode regulation strategies, failing to effectively consider the dynamic relationship between the kinetic electromotive force and the demagnetizing voltage. This results in unstable modulation accuracy under different speed and load conditions. Current methods often fail to achieve precise mode switching and smooth current waveform transitions; current switching is frequently accompanied by large current surges and voltage ripples caused by commutation, thus affecting system stability and efficiency. During the excitation phase, many methods fail to adaptively adjust the chopper frequency according to the actual speed, leading to excessively high current peaks at low speeds and increased switching losses at high speeds. Voltage balance modulation techniques during the generation phase also fail to fully optimize the duty cycle and current distribution, especially in dual-generator parallel systems, where insufficient current distribution and load balancing control further reduce generation efficiency and system reliability. Therefore, we propose a two-stage phase current modulation method for switched reluctance generators based on voltage balance. Summary of the Invention

[0004] In view of the above-mentioned shortcomings of the prior art, the first objective of the present invention is to provide a two-stage phase current modulation method for a switched reluctance generator based on voltage balance, thereby solving the problems in the background art.

[0005] To achieve the above objectives, the present invention provides the following technical solution:

[0006] A two-stage phase current modulation method for a switched reluctance generator based on voltage balance includes the following steps:

[0007] S1. Modeling the relationship between the moving electromotive force and the demagnetizing voltage;

[0008] S2, design of phase current mode switching criteria;

[0009] S3, Excitation stage current chopper control;

[0010] S4, Voltage balance modulation during power generation;

[0011] S5. Dual closed-loop control parameter tuning;

[0012] S6. Initial parameter configuration for efficiency optimization;

[0013] S7. Dynamic construction of the optimization interval;

[0014] S8, Dynamic verification of mode switching;

[0015] S9, System integration parameter adaptation.

[0016] The present invention is further configured such that: in step S1, the modeling of the relationship between the moving electromotive force and the demagnetizing voltage:

[0017] S1.1. The kinetic electromotive force (es) under different rotational speeds (n) and reference currents (iref) is measured experimentally. The functional relationship between es and iref and ω is obtained by fitting using the least squares method.

[0018]

[0019] Where a=0.06879, b=0.08841, c=3.4444, d=-5.7886e-5, e=-0.36419, g=-6.68518, a quantitative calculation model for the kinetic electromotive force is established;

[0020] S1.2, According to the phase voltage equation By combining the equivalent circuit of the power generation stage, the balance condition between the demagnetizing voltage (Udc) and the moving electromotive force is derived, and the expression for the average demagnetizing voltage (Uave_q) is obtained: Where D is the duty cycle of the lower switch, Ud is the diode voltage drop, and UM is the voltage drop of the main switch.

[0021] S1.3 Let Uave_q=es, solve for the mathematical expression of duty cycle D: D=[Udc+ipha·(Rph+Rline)+2Ud-es] / (Udc+ipha·(Rph+Rline)+2Ud-iph·Rph-2UM), and clarify that the range of D must satisfy 0≤D≤1, which provides a criterion for subsequent modulation strategy switching.

[0022] The present invention is further configured such that: in step S2, the phase current mode switching criterion design:

[0023] S2.1 Define the switching threshold Us = Udc + ipha·(Rph + Rline) + 2Ud. When Us > f(iref,ω) (f(iref,ω) is the fitting function of es), the duty cycle D > 0, and voltage balance modulation is used; otherwise, D ≤ 0, switching to single-pulse control. Since UM is small, the equation... The denominator in the equation is always greater than 0, and the sign of the duty cycle D depends on the value of the numerator:

[0024]

[0025] S2.2 By real-time acquisition of rotational speed, phase current and bus voltage, the difference ΔU between Us and f(iref,ω) is calculated. When ΔU>0, Flag=1 (mode one) is set, otherwise Flag=0 (mode two) is set to realize automatic switching of phase current modulation mode.

[0026] S2.3. At the transition speed (base speed ω1), verify that the phase current change rate diph / dθ≈0 when Us=f(iref,ω) to ensure a smooth transition of the phase current waveform during mode switching and avoid current surge.

[0027] The present invention is further configured such that: in step S3, the current chopping control during the excitation stage:

[0028] S3.1. Based on the rotor position sensor signal, set the turn-on angle (θon) and turn-off angle (θoff) to ensure that the winding inductance is in the rising region during the excitation stage. At this time, the voltage equation satisfies Ue-es=Lph·diph / dt+iph·Rph. The phase current tracks the reference value iref through current hysteresis control.

[0029] S3.2 The chopper frequency is adaptively adjusted according to the speed change. At low speeds, the chopper frequency is increased to limit current peaks, and at high speeds, the frequency is decreased to avoid switching losses. This is combined with... The formula for the rate of change of current is: diph / dt=(Ue-es) / Lph, which ensures a fast response of the excitation current;

[0030] S3.3. A soft switching method with the upper tube normally off and the lower tube chopper is adopted to avoid voltage spikes caused by hard switching. At this time, the upper tube switching signal S3 is normally set to 0, and the lower tube signal S4 is output by the hysteresis controller to achieve stable control of the current during the excitation stage.

[0031] The present invention is further configured such that: in step S4, voltage balance modulation during the power generation stage:

[0032] S4.1 During the power generation stage (θoff≤θ≤θc), by adjusting the duty cycle D of the lower tube switch S2, the average demagnetizing voltage Uave_q is made to track the moving electromotive force es. Combined with equation (3-13) Uave_q=D·Us+(1-D)·(iph·Rph+2UM), the phase current flat-top wave control is realized.

[0033] S4.2 When the moving electromotive force is greater than the demagnetizing voltage, the phase current enters the zero voltage loop by turning off S2. At this time, the voltage equation is 0=es+Lph·diph / dt+iph·Rph, avoiding the risk of overcurrent caused by the continuous rise of current.

[0034] S4.3. During phase commutation between adjacent phases, overlapping control is used, where the preceding phase is delayed to turn off and the following phase is turned on ahead of time, combined with... This reduces voltage ripple during commutation.

[0035] The present invention is further configured such that: in step S5, the dual closed-loop control parameter tuning:

[0036] S5.1. Taking the bus voltage deviation ΔUdc as input, a reference current iref is output through a PI controller, where the proportional coefficient Kp and integral coefficient Ki must meet the system dynamic response requirements, as shown in the equation. Ensure that the steady-state voltage error is less than 2%;

[0037] S5.2. Introduce an FB2PID controller, which adjusts the main control component IRa and the compensation component IRb through feedback, as shown in the equation. This suppresses current loop disturbances and improves current tracking accuracy.

[0038] S5.3. For a dual-machine parallel system, derive the decoupling equations for the bus current changes ΔI1 and ΔI2, voltage changes ΔUdc, and current distribution ratio changes ΔKI, as shown in equation [equation missing]. The decoupling matrix Q = [1 / RL, KI / (1+KI); 1 / RL, 1 / (1+KI)] enables independent control of the voltage loop and the current loop.

[0039] The present invention is further configured such that: in step S6, the initial parameter configuration for efficiency optimization:

[0040] S6.1 Based on efficiency experimental data at different speeds and power levels, the least squares method is used to fit the relationship between the optimal turn-on angle θon and the target power Pref and speed n, as shown in the equation. , where Pref is the target power generation of the SRG, n is the real-time rotational speed, and the coefficients are a = -0.05764, b = -0.00362, c = 5.238e-5, d = -1.19e-6, e = -1.437e-6, f = 26.63, which are used to calculate the optimal turn-on angle in real time;

[0041] S6.2. Analyze the influence of the turn-off angle on the power generation efficiency and power through simulation, and select the optimal turn-off angle θoff = 31° to balance the power generation capacity and efficiency and reduce the complexity of parameter adjustment;

[0042] S6.3. Use the rotational speed ratio of the two generators Kn = n1 / n2 as the initial current distribution ratio KI. When n1 > n2, Kn = n1 / n2; otherwise, Kn = n2 / n1, providing an initial search point for efficiency optimization.

[0043] The present invention is further configured as follows: In the step S7, dynamically constructing the optimization interval:

[0044] S7.1. According to the initial current distribution ratio Kn and the search interval length k, construct the optimization interval [KIa, KId] = [Kn - k, Kn + k]. When Kn < k, adjust it to [1 / Kn - k, 1 / Kn + k] to ensure interval symmetry;

[0045] S7.2. Select a point 30% away from the endpoint in the interval as the interpolation point KIb, such as KIb = KIa + 0.3·(KId - KIa), to reduce the fitting error;

[0046] S7.3. Collect the input powers Pina, Pinb, Pind corresponding to KIa, KIb, KId, fit the quadratic function Pin = a·KI² + b·KI + c, and solve the extreme point KIm = -b / (2a) as the candidate value of the optimal current distribution ratio.

[0047] The present invention is further configured as follows: In the step S8, dynamically verifying the mode switching:

[0048] S8.1. Under the conditions of rotational speed n = 200r / min and iref = 3A, verify the phase current tracking performance of Mode 1 (voltage balance modulation). At this time, the duty cycle D is calculated by Equation (3-15), and the phase current waveform is a flat-topped wave with a reduced attenuation speed;

[0049] S8.2. When the rotational speed n = 900r / min, verify the excitation interval adjustment effect of Mode 2 (single-pulse control). Through the optimized combination of θon and θoff, ensure that the phase current amplitude meets the power demand;

[0050] S8.3 During the process of the speed increasing from 400 r / min to 1000 r / min, monitor the change of ΔU to verify the response time of the Flag signal switching from 1 to 0, and ensure that there is no current impact during mode switching.

[0051] The present invention is further configured such that: in step S9, system integration parameter adaptation:

[0052] S9.1 The control algorithm is implemented based on TMS320F28335DSP. AD7606 is configured to collect phase current and voltage signals, and DA5344 outputs PWM control signals. The sampling frequency is set to 10kHz.

[0053] S9.2. Select FQA160N08 MOSFET and D75E60 freewheeling diode, and design an asymmetric half-bridge topology to ensure a maximum phase current of 10A and a bus voltage of 450V operating range.

[0054] S9.3. Construct a dual SRG parallel experimental platform, set an adjustable load resistor and prime mover, and acquire phase current and bus voltage waveforms using an oscilloscope to verify the actual effect of each step, as shown in the formula. The voltage smoothing coefficient τ = (Umax - Umin) / Uave < 5%.

[0055] Beneficial effects

[0056] Compared with known public technologies, the technical solution provided by this invention has the following beneficial effects:

[0057] 1. This invention establishes a quantitative model of the kinetic electromotive force and demagnetizing voltage and derives an analytical formula for the duty cycle, thereby enabling calculable and constrained control of the duty cycle D. This provides a precise basis for voltage balance modulation, thereby improving modulation accuracy and adaptability to operating conditions, reducing errors and instability factors caused by empirical settings. By setting a switching threshold Us and calculating ΔU in real time to generate a Flag, automatic switching between voltage balance modulation and single-pulse control is achieved. At the transition point, the rate of change of current is approximately zero, thus achieving smooth switching across the entire speed range, avoiding current surges and reducing overcurrent risks.

[0058] 2. This invention achieves rapid tracking and peak limiting of excitation current through current hysteresis chopping during the excitation stage, adaptive chopping frequency with rotational speed, and soft-switching strategy. At the same time, it reduces voltage spikes and switching losses, thereby improving current control stability and enhancing device reliability. During the power generation stage, voltage balance modulation is used to obtain a flat-top wave of phase current, zero-voltage loop suppresses current surge, and commutation overlap reduces voltage ripple. Combined with dual closed loops, efficiency optimization, and parallel decoupling, the invention achieves smoother bus voltage, more accurate current tracking, and more reasonable current distribution in the parallel system, thereby improving power generation efficiency and overall system stability. Attached Figure Description

[0059] Figure 1 This is a schematic flowchart of the two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to the present invention.

[0060] Figure 2 This is a schematic diagram of the equivalent circuit of each operating process of the SRG in the two-stage phase current modulation method of the switched reluctance generator based on voltage balance according to the present invention.

[0061] Figure 3 This is a schematic diagram of the equivalent circuit of the dual SRG parallel power generation system based on the voltage balance-based two-stage phase current modulation method of the switched reluctance generator of the present invention.

[0062] Figure 4 This is a schematic diagram of a novel two-stage phase current modulation strategy based on voltage balance in the two-stage phase current modulation method for switched reluctance generators based on voltage balance according to the present invention.

[0063] Figure 5 This is a schematic diagram of low-speed current chopping control for the two-stage phase current modulation method of the switched reluctance generator based on voltage balance according to the present invention.

[0064] Figure 6 This is a schematic diagram of the simulated waveforms of each signal of the phase winding of the two-stage phase current modulation method of the switched reluctance generator based on voltage balance in this invention under mode one control.

[0065] Figure 7 This is a schematic diagram of the simulated waveforms of each signal of the phase winding under mode two control of the two-stage phase current modulation method of the switched reluctance generator based on voltage balance of the present invention.

[0066] Figure 8 This is a schematic diagram of the control block of a parallel power generation system based on FB2PID dual closed-loop current sharing control for the two-stage phase current modulation method of the switched reluctance generator based on voltage balance according to the present invention.

[0067] Figure 9 This is a schematic diagram of the voltage and current dual closed-loop control block of the two-stage phase current modulation method for switched reluctance generators based on voltage balance according to the present invention.

[0068] Figure 10 This is a schematic diagram showing the optimal efficiency turn-on angle at different power levels and speeds for the two-stage phase current modulation method of the switched reluctance generator based on voltage balance according to the present invention.

[0069] Figure 11 This is a schematic diagram of fitting the optimal efficiency turn-on angle at different power levels and speeds for the two-stage phase current modulation method of the switched reluctance generator based on voltage balance according to the present invention.

[0070] Figure 12This is a schematic diagram illustrating the prediction optimization principle of the two-stage phase current modulation method for a switched reluctance generator based on voltage balance in this invention.

[0071] Figure 13 This is a schematic diagram of the efficiency optimization process based on the prediction method for the two-stage phase current modulation method of the switched reluctance generator based on voltage balance in this invention. Detailed Implementation

[0072] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below. Obviously, the described embodiments are only some, not all, of the embodiments of the present invention. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention without creative effort are within the scope of protection of the present invention.

[0073] The present invention will be further described below with reference to embodiments.

[0074] Example

[0075] like Figure 1-13 As shown, the two-stage phase current modulation method for a switched reluctance generator based on voltage balance includes the following steps:

[0076] S1. Modeling the relationship between the moving electromotive force and the demagnetizing voltage;

[0077] S2, design of phase current mode switching criteria;

[0078] S3, Excitation stage current chopper control;

[0079] S4, Voltage balance modulation during power generation;

[0080] S5. Dual closed-loop control parameter tuning;

[0081] S6. Initial parameter configuration for efficiency optimization;

[0082] S7. Dynamic construction of the optimization interval;

[0083] S8, Dynamic verification of mode switching;

[0084] S9, System integration parameter adaptation;

[0085] In step S1, modeling the relationship between the kinetic electromotive force and the demagnetizing voltage:

[0086] S1.1. The kinetic electromotive force (es) under different rotational speeds (n) and reference currents (iref) is measured experimentally. The functional relationship between es and iref and ω is obtained by fitting using the least squares method.

[0087]

[0088] Where a=0.06879, b=0.08841, c=3.4444, d=-5.7886e-5, e=-0.36419, g=-6.68518, a quantitative calculation model for the kinetic electromotive force is established;

[0089] S1.2, According to the phase voltage equation By combining the equivalent circuit of the power generation stage, the balance condition between the demagnetizing voltage (Udc) and the moving electromotive force is derived, and the expression for the average demagnetizing voltage (Uave_q) is obtained: Where D is the duty cycle of the lower switch, Ud is the diode voltage drop, and UM is the voltage drop of the main switch.

[0090] S1.3 Let Uave_q=es, solve for the mathematical expression of duty cycle D: D=[Udc+ipha·(Rph+Rline)+2Ud-es] / (Udc+ipha·(Rph+Rline)+2Ud-iph·Rph-2UM), and clarify that the range of D must satisfy 0≤D≤1, which provides a criterion for subsequent modulation strategy switching;

[0091] In step S2, the phase current mode switching criterion design is as follows:

[0092] S2.1 Define the switching threshold Us = Udc + ipha·(Rph + Rline) + 2Ud. When Us > f(iref,ω) (f(iref,ω) is the fitting function of es), the duty cycle D > 0, and voltage balance modulation is used; otherwise, D ≤ 0, switching to single-pulse control. Since UM is small, the equation... The denominator in the equation is always greater than 0, and the sign of the duty cycle D depends on the value of the numerator:

[0093]

[0094] S2.2 By real-time acquisition of rotational speed, phase current and bus voltage, the difference ΔU between Us and f(iref,ω) is calculated. When ΔU>0, Flag=1 (mode one) is set, otherwise Flag=0 (mode two) is set to realize automatic switching of phase current modulation mode.

[0095] S2.3. At the transition speed (base speed ω1), verify that the phase current change rate diph / dθ≈0 when Us=f(iref,ω) to ensure a smooth transition of the phase current waveform during mode switching and avoid current surges.

[0096] In step S3, the current chopping control during the excitation stage:

[0097] S3.1. Based on the rotor position sensor signal, set the turn-on angle (θon) and turn-off angle (θoff) to ensure that the winding inductance is in the rising region during the excitation stage. At this time, the voltage equation satisfies Ue-es=Lph·diph / dt+iph·Rph. The phase current tracks the reference value iref through current hysteresis control.

[0098] S3.2 The chopper frequency is adaptively adjusted according to the speed change. At low speeds, the chopper frequency is increased to limit current peaks, and at high speeds, the frequency is decreased to avoid switching losses. This is combined with... The formula for the rate of change of current is: diph / dt=(Ue-es) / Lph, which ensures a fast response of the excitation current;

[0099] S3.3. A soft switching method with the upper tube normally off and the lower tube chopping is adopted to avoid voltage spikes caused by hard switching. At this time, the upper tube switching signal S3 is normally set to 0, and the lower tube signal S4 is output by the hysteresis controller to achieve stable control of the current during the excitation stage.

[0100] In step S4, voltage balance modulation during the power generation stage:

[0101] S4.1 During the power generation stage (θoff≤θ≤θc), by adjusting the duty cycle D of the lower tube switch S2, the average demagnetizing voltage Uave_q is made to track the moving electromotive force es. Combined with equation (3-13) Uave_q=D·Us+(1-D)·(iph·Rph+2UM), the phase current flat-top wave control is realized.

[0102] S4.2 When the moving electromotive force is greater than the demagnetizing voltage, the phase current enters the zero voltage loop by turning off S2. At this time, the voltage equation is 0=es+Lph·diph / dt+iph·Rph, avoiding the risk of overcurrent caused by the continuous rise of current.

[0103] S4.3. During phase commutation between adjacent phases, overlapping control is used, where the preceding phase is delayed to turn off and the following phase is turned on ahead of time, combined with... This reduces voltage ripple during commutation;

[0104] In step S5, during the tuning of the dual closed-loop control parameters:

[0105] S5.1. Taking the bus voltage deviation ΔUdc as input, a reference current iref is output through a PI controller, where the proportional coefficient Kp and integral coefficient Ki must meet the system dynamic response requirements, as shown in the equation. Ensure that the steady-state voltage error is less than 2%;

[0106] S5.2. Introduce an FB2PID controller, which adjusts the main control component IRa and the compensation component IRb through feedback, as shown in the equation. This suppresses current loop disturbances and improves current tracking accuracy.

[0107] S5.3. For the dual - machine parallel system, decoupling equations of the bus - current variation ΔI1, ΔI2 and the voltage variation ΔUdc, current - sharing ratio variation ΔKI are derived, as shown in Equation , where the decoupling matrix Q = [1 / RL, KI / (1 + KI); 1 / RL, 1 / (1 + KI)], to achieve independent control of the voltage loop and current loop;

[0108] In the step S6, initial parameter configuration for efficiency optimization:

[0109] S6.1. Based on the efficiency experimental data at different speeds and power levels, the relationship between the optimal turn - on angle θon and the target power Pref, speed n is fitted by the least - squares method, as shown in Equation , where Pref is the target power generation of the SRG, n is the real - time speed, and the coefficients are a = - 0.05764, b = - 0.00362, c = 5.238e - 5, d = - 1.19e - 6, e = - 1.437e - 6, f = 26.63, which are used to calculate the optimal turn - on angle in real time; [[ID=……]]

[0110] S6.2. Through simulation analysis of the influence of the turn - off angle on the power generation efficiency and power, the optimal turn - off angle θoff = 31° is selected to balance the power generation capacity and efficiency and reduce the complexity of parameter adjustment;

[0111] S6.3. Taking the speed ratio of the two generators Kn = n1 / n2 as the initial current - sharing ratio KI, when n1 > n2, Kn = n1 / n2, otherwise Kn = n2 / n1, to provide an initial search point for efficiency optimization;

[0112] In the step S7, dynamic construction of the optimization interval:

[0113] S7.1. According to the initial current - sharing ratio Kn and the search - interval length k, an optimization interval [KIa, KId] = [Kn - k, Kn + k] is constructed. When Kn < k, it is adjusted to [1 / Kn - k, 1 / Kn + k] to ensure interval symmetry;

[0114] S7.2. Select the interpolation point KIb at 30% from the end point within the interval, such as KIb = KIa + 0.3·(KId - KIa) to reduce the fitting error;

[0115] S7.3. Collect the input powers Pina, Pinb, Pind corresponding to KIa, KIb, KId, fit the quadratic function Pin = a·KI² + b·KI + c, and solve the extreme - point KIm = - b / (2a) as the candidate value of the optimal current - sharing ratio;

[0116] In the step S8, dynamic verification of mode switching:

[0117] S8.1 Under the conditions of speed n=200r / min and iref=3A, verify the phase current tracking performance of mode one (voltage balance modulation). At this time, the duty cycle D is calculated by formula (3-15), the phase current waveform is a flat-top wave, and the attenuation speed is reduced.

[0118] S8.2. At a speed of n=900r / min, verify the excitation range adjustment effect of mode two (single pulse control). Through the optimized combination of θon and θoff, ensure that the phase current amplitude meets the power requirements.

[0119] S8.3 During the process of the speed increasing from 400 r / min to 1000 r / min, monitor the change of ΔU to verify the response time of the Flag signal switching from 1 to 0, and ensure that there is no current impact during mode switching;

[0120] In step S9, system integration parameter adaptation is underway:

[0121] S9.1 The control algorithm is implemented based on TMS320F28335DSP. AD7606 is configured to collect phase current and voltage signals, and DA5344 outputs PWM control signals. The sampling frequency is set to 10kHz.

[0122] S9.2. Select FQA160N08 MOSFET and D75E60 freewheeling diode, and design an asymmetric half-bridge topology to ensure a maximum phase current of 10A and a bus voltage of 450V operating range.

[0123] S9.3. Construct a dual SRG parallel experimental platform, set an adjustable load resistor and prime mover, and acquire phase current and bus voltage waveforms using an oscilloscope to verify the actual effect of each step, as shown in the formula. The voltage smoothing coefficient τ = (Umax - Umin) / Uave < 5%.

[0124] In this embodiment, the voltage balance-based two-stage phase current modulation method for switched reluctance generators establishes a quantitative model of the moving electromotive force and demagnetizing voltage to achieve accurate calculation of the duty cycle and automatic switching of the phase current mode. During the excitation stage, current chopping control is used to ensure fast response and peak limit. During the power generation stage, voltage balance modulation is used to achieve a flat-top phase current. Combined with dual closed-loop control and efficiency optimization strategy, the switching angle, duty cycle and current distribution are dynamically optimized. At the same time, the feasibility of actual hardware is ensured through system integration parameter adaptation. Thus, in actual use, it effectively improves current tracking accuracy, reduces commutation impact, smooths bus voltage, and improves power generation efficiency and system stability.

[0125] Working principle

[0126] like Figure 1-13As shown, in practical applications, a quantitative model between the kinetic electromotive force (EMF) and the demagnetizing voltage was established, enabling an accurate description of the kinetic EMF. By fitting experimental data, the functional relationship between the kinetic EMF (es), the reference current (iref), and the rotational speed (ω) was obtained, providing a theoretical basis for subsequent modulation strategies. The change in the kinetic EMF was used to calculate the duty cycle D. This duty cycle calculation formula, by considering factors such as the demagnetizing voltage, phase current, and voltage drop of the switching transistor, ensures precise control of voltage balance and current modulation.

[0127] In the phase current mode switching section, this method defines a switching threshold Us and calculates the difference ΔU based on the real-time measured speed, phase current and bus voltage, and automatically switches between different modulation modes. When Us>f(iref,ω), voltage balance modulation (mode one) is used; otherwise, it switches to single-pulse control (mode two). Through this mode switching mechanism, the system can automatically adjust the modulation strategy under different operating conditions to ensure a smooth transition of the phase current waveform and avoid current surges caused by mode switching.

[0128] During the excitation phase, this method employs current chopping control to ensure that the current peak is limited at low speeds and to avoid switching losses at high speeds. By dynamically adjusting the chopping frequency and combining it with the rotor position sensor signal, the method ensures that the excitation current can quickly respond to the reference current iref. This control strategy avoids voltage spikes caused by hard switching by using a soft switching method of normally off upper tube and chopping lower tube, thereby achieving stable current control during the excitation phase.

[0129] During the power generation phase, by adjusting the duty cycle D of the lower switch, the average demagnetizing voltage Uaveq is made to track the moving electromotive force es, thus achieving flat-top wave control of the phase current. If the moving electromotive force exceeds the demagnetizing voltage during this process, the system will turn off the lower switch to avoid the risk of overcurrent caused by excessive current. During the commutation process, by delaying the turn-off of the previous phase and advancing the turn-on of the next phase, the voltage ripple during the commutation period is reduced, further improving the voltage smoothness and stability of the system.

[0130] In summary, this method achieves efficient phase current control through precise voltage balance modulation and mode switching mechanisms, ensuring flat-top current, smooth bus voltage, and system stability during power generation. Through dual closed-loop control and efficiency optimization strategies, the system can dynamically adjust operating parameters to improve power generation efficiency and reduce energy loss during system operation. Ultimately, this significantly improves the performance of switched reluctance generators in practical applications.

[0131] The above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions will not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A two-stage phase current modulation method for a switched reluctance generator based on voltage balance, characterized in that, Includes the following steps: S1. Modeling the relationship between the moving electromotive force and the demagnetizing voltage; S2, design of phase current mode switching criteria; S3, Excitation stage current chopper control; S4, Voltage balance modulation during power generation; S5. Dual closed-loop control parameter tuning; S6. Initial parameter configuration for efficiency optimization; S7. Dynamic construction of the optimization interval; S8, Dynamic verification of mode switching; S9, System integration parameter adaptation.

2. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that: In step S1, modeling the relationship between the moving electromotive force and the demagnetizing voltage: S1.

1. The kinetic electromotive force (es) under different rotational speeds (n) and reference currents (iref) is measured experimentally. The functional relationship between es and iref and ω is obtained by fitting using the least squares method. Where a=0.06879, b=0.08841, c=3.4444, d=-5.7886e-5, e=-0.36419, g=-6.68518, a quantitative calculation model for the kinetic electromotive force is established; S1.2, According to the phase voltage equation By combining the equivalent circuit of the power generation stage, the balance condition between the demagnetizing voltage (Udc) and the kinetic electromotive force is derived, and the expression for the average demagnetizing voltage (Uave_q) is obtained: Where D is the duty cycle of the lower switch, Ud is the diode voltage drop, and UM is the voltage drop of the main switch. S1.3 Let Uave_q=es, solve for the mathematical expression of duty cycle D: D=[Udc+ipha·(Rph+Rline)+2Ud-es] / (Udc+ipha·(Rph+Rline)+2Ud-iph·Rph-2UM), and clarify that the range of D must satisfy 0≤D≤1, which provides a criterion for subsequent modulation strategy switching.

3. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that: In step S2, the phase current mode switching criterion design is as follows: S2.1 Define the switching threshold Us = Udc + ipha·(Rph + Rline) + 2Ud. When Us > f(iref,ω) (f(iref,ω) is the fitting function of es), the duty cycle D > 0, and voltage balance modulation is used; otherwise, D ≤ 0, switching to single-pulse control. Since UM is small, the equation... The denominator in the equation is always greater than 0, and the sign of the duty cycle D depends on the value of the numerator: S2.2 By real-time acquisition of rotational speed, phase current and bus voltage, the difference ΔU between Us and f(iref,ω) is calculated. When ΔU>0, Flag=1 (mode one) is set, otherwise Flag=0 (mode two) is set to realize automatic switching of phase current modulation mode. S2.

3. At the transition speed (base speed ω1), verify that the phase current change rate diph / dθ≈0 when Us=f(iref,ω) to ensure a smooth transition of the phase current waveform during mode switching and avoid current surge.

4. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that: In step S3, the current chopping control during the excitation stage: S3.

1. Based on the rotor position sensor signal, set the turn-on angle (θon) and turn-off angle (θoff) to ensure that the winding inductance is in the rising region during the excitation stage. At this time, the voltage equation satisfies Ue-es=Lph·diph / dt+iph·Rph. The phase current tracks the reference value iref through current hysteresis control. S3.2 The chopper frequency is adaptively adjusted according to the speed change. At low speeds, the chopper frequency is increased to limit current peaks, and at high speeds, the frequency is decreased to avoid switching losses. This is combined with... The formula for the rate of change of current is: diph / dt=(Ue-es) / Lph, which ensures a fast response of the excitation current; S3.

3. A soft switching method with the upper tube normally off and the lower tube chopper is adopted to avoid voltage spikes caused by hard switching. At this time, the upper tube switching signal S3 is normally set to 0, and the lower tube signal S4 is output by the hysteresis controller to achieve stable control of the current during the excitation stage.

5. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that: In step S4, voltage balance modulation during the power generation stage: S4.1 During the power generation stage (θoff≤θ≤θc), by adjusting the duty cycle D of the lower tube switch S2, the average demagnetizing voltage Uave_q is made to track the moving electromotive force es. Combined with equation (3-13) Uave_q=D·Us+(1-D)·(iph·Rph+2UM), the phase current flat-top wave control is realized. S4.

2. When the motional electromotive force is greater than the demagnetizing voltage, turn off S2 to make the phase current enter the zero-voltage loop. At this time, the voltage equation is 0 = es + Lph·diph / dt + iph·Rph, avoiding the overcurrent risk caused by continuous current increase; S4.

3. During phase commutation between adjacent phases, overlapping control is used, where the preceding phase is delayed to turn off and the following phase is turned on ahead of time, combined with... This reduces voltage ripple during commutation.

6. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that, In the step S5, the double closed-loop control parameter tuning: S5.

1. Taking the bus voltage deviation ΔUdc as input, a reference current iref is output through a PI controller, where the proportional coefficient Kp and integral coefficient Ki must meet the system dynamic response requirements, as shown in the equation. Ensure that the steady-state voltage error is less than 2%; S5.

2. Introduce an FB2PID controller, which adjusts the main control component IRa and the compensation component IRb through feedback, as shown in the equation. This suppresses current loop disturbances and improves current tracking accuracy. S5.

3. For a dual-machine parallel system, derive the decoupling equations for the bus current changes ΔI1 and ΔI2, voltage changes ΔUdc, and current distribution ratio changes ΔKI, as shown in equation [equation missing]. The decoupling matrix Q = [1 / RL, KI / (1+KI); 1 / RL, 1 / (1+KI)] enables independent control of the voltage loop and the current loop.

7. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that, In the step S6, the initial parameter configuration for efficiency optimization: S6.1 Based on efficiency experimental data at different speeds and power levels, the least squares method is used to fit the relationship between the optimal turn-on angle θon and the target power Pref and speed n, as shown in the equation. In the formula, Pref is the target power generation of the SRG, n is the real-time rotational speed, and the coefficients are a=-0.05764, b=-0.00362, c=5.238e-5, d=-1.19e-6, e=-1.437e-6, f=26.63, which are used to calculate the optimal turn-on angle in real time. S6.

2. Analyze the influence of the turn-off angle on the power generation efficiency and power through simulation, and select the optimal turn-off angle θoff = 31° to balance the power generation capacity and efficiency and reduce the complexity of parameter adjustment; S6.

3. Use the speed ratio of two generators Kn = n1 / n2 as the initial current distribution ratio KI. When n1 > n2, Kn = n1 / n2; otherwise, Kn = n2 / n1, providing an initial search point for efficiency optimization.

8. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that, In the step S7, the dynamic construction of the optimization interval: S7.

1. According to the initial current distribution ratio Kn and the search interval length k, construct the optimization interval [KIa, KId] = [Kn - k, Kn + k]. When Kn < k, adjust it to [1 / Kn - k, 1 / Kn + k] to ensure interval symmetry; S7.

2. Select a point 30% away from the endpoint in the interval as the interpolation point KIb, such as KIb = KIa + 0.3·(KId - KIa) to reduce the fitting error; S7.

3. Collect the input powers Pina, Pinb, Pind corresponding to KIa, KIb, KId, fit the quadratic function Pin = a·KI² + b·KI + c, and solve the extreme point KIm = -b / (2a) as the candidate value of the optimal current distribution ratio.

9. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that, In the step S8, the dynamic verification of mode switching: S8.

1. Under the conditions of rotational speed n = 200 r / min and iref = 3 A, verify the phase current tracking performance of mode 1 (voltage balance modulation). At this time, the duty cycle D is calculated by formula (3 - 15), and the phase current waveform is a flat-topped wave with a reduced attenuation rate; S8.

2. When the rotational speed n = 900 r / min, verify the excitation interval adjustment effect of mode 2 (single-pulse control). Through the optimal combination of θon and θoff, ensure that the phase current amplitude meets the power demand; S8.

3. During the process of the rotational speed stepping from 400 r / min to 1000 r / min, monitor the change of ΔU and verify the response time of the Flag signal switching from 1 to 0 to ensure no current impact during mode switching.

10. The two-stage phase current modulation method for a switched reluctance generator based on voltage balance according to claim 1, characterized in that, In the step S9, the system integration parameter adaptation: S9.

1. Implement the control algorithm based on TMS320F28335 DSP, configure AD7606 to collect the phase current and voltage signals, and DA5344 to output PWM control signals. The sampling frequency is set to 10 kHz; S9.

2. Select FQA160N08 MOSFET and D75E60 freewheeling diodes, design an asymmetric half-bridge topology to ensure the working range of a maximum phase current of 10 A and a bus voltage of 450 V; S9.

3. Construct a dual SRG parallel experimental platform, set an adjustable load resistor and prime mover, and acquire phase current and bus voltage waveforms using an oscilloscope to verify the actual effect of each step, as shown in the formula. The voltage smoothing coefficient τ = (Umax - Umin) / Uave < 5%.

Images