A wide range configurable temperature sensor detection circuit

Abstract

The application discloses a wide-range configurable temperature sensor detection circuit, relates to the technical field of power electronics, and is used for solving the problems of easy ADC saturation, low-resistance segment resolution reduction and inconsistent connection link bias error when a same device is compatible with low-resistance thermal resistance and high-resistance thermistor, the circuit switches the excitation strength in the low-resistance and high-resistance interval through a switchable constant-current gear, and configures a selectable voltage division path in the thermistor mode, so that the sampling voltage in the high-resistance scene is limited in the rear input range through the voltage division attenuation coefficient; when the voltage division output is greater than a preset full-range threshold, a parallel clamping resistance is accessed to reduce the equivalent impedance of the sensor; meanwhile, the adjustable gain instrument amplification module, the low-pass filter and the ADC interface protection are matched to realize stable sampling in different resistance value intervals; and the resistance value conversion consistency and reproducibility before and after the cross-range switching are improved.

Description

Technical Field

[0001] This invention relates to the field of power electronics technology, and more specifically, to a wide-range configurable temperature sensor detection circuit. Background Technology

[0002] Existing temperature acquisition front-ends are typically designed for a single type of sensor or a narrow resistance range. For example, constant current excitation and voltage-to-resistance measurement are used for RTDs, while voltage divider measurement is used for thermistors. When the same device needs to be compatible with both low-resistance and high-resistance thermistors, using fixed excitation and fixed range can easily lead to two types of engineering contradictions: First, in low-temperature, high-resistance thermistor scenarios, the voltage across the sensor may rise significantly, exceeding the ADC input range and causing saturation distortion. Second, blindly reducing the excitation or gain to avoid saturation can result in the low-resistance thermistor having an excessively low output voltage at room temperature, leading to decreased resolution and noise immunity. Simultaneously, terminal contact resistance, lead resistance, and residual bias in the amplifier and sampling link introduce zero-point errors and drift, further amplifying the inconsistency problem in cross-range measurements.

[0003] Therefore, how can we ensure that the ADC input does not saturate and can continuously output an effective sensor resistance value that can be used for temperature conversion when the sensor resistance varies over a wide range from about 100 ohms to about 330 kiloohms in the same temperature sensor detection circuit? Summary of the Invention

[0004] To overcome the above-mentioned defects of the prior art, embodiments of the present invention provide a wide-range configurable temperature sensor detection circuit to solve the problems that it is difficult to simultaneously accommodate low-resistance RTDs and high-resistance RTDs with fixed excitation and fixed range, and that it is easy to cause ADC saturation, low-resistance measurement resolution degradation, and difficulty in offsetting connection link bias errors.

[0005] To achieve the above objectives, the present invention provides the following technical solution:

[0006] A wide-range configurable temperature sensor detection circuit includes:

[0007] Constant current source module, terminal J1, resistance bias calibration circuit, NTC voltage divider and range protection circuit, low-pass filter, high-precision instrument amplification module, ADC interface and protection, microcontroller unit (MCU);

[0008] The constant current source module includes an operational amplifier U1, a power transistor Q1, a sampling resistor Rs1, a sampling resistor Rs2, and an analog switch SW1. The microcontroller unit MCU controls SW1 to select either Rs1 or Rs2 to output the constant current source current Iexc.

[0009] The temperature sensor is connected through pins A, B, and C of terminal J1. The resistance bias calibration circuit includes an analog switch SW2. In the calibration state, SW2 shorts pins A and B to acquire the bias voltage Vbias and calculate the bias resistance Rbias. In the measurement state, SW2 is disconnected and acquires Vsense to calculate the sensor resistance Rsensor.

[0010] The NTC voltage divider and range protection circuit includes analog switches SW3 / SW4, which switch between pass-through and voltage divider / clamping states and output Vdiv. Vdiv is then processed by a low-pass filter and a high-precision instrument amplification module, and is acquired as Vadc by the ADC interface and protection. The microcontroller unit (MCU) obtains the temperature based on the PTC or NTC temperature and resistance table.

[0011] In a preferred embodiment, terminal J1 is provided with pins A, B, and C. Pins A and B are used for 2-wire connection, and pins A, B, and C are used for 3-wire connection. Pins A and B are connected to the same end of the temperature sensor, and pin C is connected to the other end of the temperature sensor. The differential input terminals of the high-precision instrument amplification module are respectively connected to pins B and C.

[0012] In a preferred embodiment, the microcontroller unit (MCU) acquires the bias voltage Vbias in the calibration state and calculates the bias resistance value Rbias based on the constant current source current Iexc. In the measurement state, the MCU acquires Vsense and calculates the sensor resistance value Rsensor based on the bias resistance value Rbias.

[0013] In a preferred embodiment, the NTC voltage divider and range protection circuit includes voltage divider resistors Rdiv1 and Rdiv2, which are connected in series to form a voltage divider output node. The output ratio of the voltage divider output node is Rdiv2 / (Rdiv1+Rdiv2), and is expressed by the voltage divider attenuation coefficient k as k=(Rdiv1+Rdiv2) / Rdiv2. In NTC mode, analog switch SW3 connects the voltage divider output node to a low-pass filter, making Vdiv=Vsense / k. In PTC mode, analog switch SW3 switches to a direct signal path.

[0014] In a preferred embodiment, the NTC voltage divider and range protection circuit includes a parallel clamping resistor Rsh. When the voltage divider output Vdiv approaches the full range of the ADC, the microcontroller unit MCU controls the analog switch SW4 to enter the clamping state, so that the parallel clamping resistor Rsh and the temperature sensor are connected in parallel to form an equivalent resistance Req, where Req is the parallel connection of Rsensor and Rsh.

[0015] In a preferred embodiment, the low-pass filter is a first-order RC structure, comprising a resistor Rlp and a capacitor Clp, and its cutoff frequency fc is equal to one divided by two times pi multiplied by Rlp and Clp.

[0016] In a preferred embodiment, the high-precision instrumentation amplifier module adopts a three-operation amplifier architecture or a dedicated instrumentation amplifier chip U3. The gain of the high-precision instrumentation amplifier module is determined by the gain setting resistor Rg and the internal resistor Rint. The gain G is equal to one plus two times Rint and then divided by Rg.

[0017] In a preferred embodiment, the ADC interface and protection include a series current-limiting resistor Rin and bidirectional clamping diodes D1 and D2. The current-limiting resistor Rin is connected in series between the output terminal of the high-precision instrumentation amplifier module and the ADC input terminal ADC_IN. The bidirectional clamping diodes D1 and D2 clamp ADC_IN to the power supply VDD and ground GND, respectively, and the bidirectional clamping diodes D1 and D2 are Schottky diodes.

[0018] In a preferred embodiment, the microcontroller unit (MCU) processes the acquired Vadc. In NTC mode, Vadc is restored to Vsense based on the voltage division ratio k, and the sensor resistance Rsensor is calculated. In PTC mode, the sensor resistance Rsensor is calculated based on Vsense and the bias resistance Rbias. The temperature is obtained by looking up the table or using linear interpolation according to the PTC or NTC temperature-resistance table.

[0019] In a preferred embodiment, the microcontroller unit (MCU) configures the constant current range SW1, the voltage divider / clamping state SW3 / SW4, and the instrument amplification gain according to the sensor model and wiring method, and controls the analog switch SW2 to switch between calibration and measurement states to refresh the bias resistance value Rbias and track bias changes caused by wiring harness aging.

[0020] The technical effects and advantages of the wide-range configurable temperature sensor detection circuit of the present invention are as follows:

[0021] This invention introduces a switchable constant current range at the front end to switch the excitation intensity between low and high resistance ranges. In the thermistor mode, it configures an optional voltage divider path and a parallel clamping resistor near full scale. This reduces the sensor's equivalent impedance in extremely high-resistance scenarios, limiting the sampling voltage within the usable range of the ADC and preventing unusable readings due to saturation. Simultaneously, an adjustable gain instrumentation amplification link and low-pass filtering ensure more suitable amplification levels and a more stable signal-to-noise ratio across different resistance ranges. Furthermore, by shorting the residual bias in calibration mode and subtracting it in measurement mode, it cancels out zero-point errors in the connection and sampling links, resulting in more consistent and reproducible output resistance values ​​before and after range switching. In addition, current limiting and clamping protection at the ADC end reduce the risk of abnormal inputs to the sampling and devices, thus achieving more stable temperature detection and conversion output over a wide resistance range. Attached Figure Description

[0022] Figure 1 This is a schematic diagram of a wide-range configurable temperature sensor detection circuit according to the present invention.

[0023] Figure 2 is a complete circuit diagram of a wide-range configurable temperature sensor detection circuit according to the present invention.

[0024] Figure 3 is a schematic diagram of the switching state and current flow direction of the resistance bias calibration circuit in the calibration state in this invention.

[0025] Figure 4 is a schematic diagram of the switching state and signal flow of the temperature sensor detection circuit in the measurement state in this invention. Detailed Implementation

[0026] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments.

[0027] This invention provides a wide-range configurable temperature sensor detection circuit, aiming to achieve high-precision measurement of various types of temperature sensors with wide resistance ranges and compatible wiring methods through a unified hardware architecture. The specific embodiments of this invention will be described in detail below with reference to the accompanying drawings to ensure that those skilled in the art can fully reproduce the technical solution of this invention based on this specification.

[0028] like Figure 1As shown, the temperature sensor detection circuit of this invention includes a constant current source module, a temperature sensor access terminal J1, a resistance bias calibration circuit, an NTC voltage divider and range protection circuit, a low-pass filter, a high-precision instrument amplification module, an analog-to-digital converter interface, and a central control unit. All of the above functional modules are interconnected via conductive traces on a printed circuit board and share the same reference ground plane, employing a star grounding strategy to suppress common-mode interference.

[0029] like Figure 2 As shown in the complete circuit schematic, the constant current source module includes operational amplifier U1, power transistor Q1, sampling resistor Rs1, sampling resistor Rs2, and analog switch SW1; terminal J1 includes pins A, B, and C; the resistance bias calibration circuit includes analog switch SW2 connected between pins A and B; the NTC voltage divider and range protection circuit includes voltage divider resistors Rdiv1 and Rdiv2, parallel clamping resistor Rsh, analog switch SW3, and analog switch SW4; the low-pass filter includes resistor Rlp and capacitor Clp; the high-precision instrumentation amplification module includes instrumentation amplifier U3 and gain setting resistor Rg; the ADC interface and protection include current limiting resistor Rin, clamping diode D1, clamping diode D2, and ADC input terminal ADC_IN; the microcontroller unit (MCU) is connected to SW1, SW2, SW3, SW4, and ADC_IN through control ports to complete constant current range selection, calibration state switching, voltage divider path selection, clamping state control, and sampling data processing.

[0030] The constant current source module consists of an operational amplifier U1, a power transistor Q1, and a switchable sampling resistor Rs, forming a closed-loop feedback structure. The non-inverting input of operational amplifier U1 receives the reference voltage Vref, and its inverting input is connected to one end of the sampling resistor Rs, while the other end of Rs is grounded. Power transistor Q1 is an NPN bipolar junction transistor; its base is connected to the output of operational amplifier U1, its emitter serves as the constant current output IOUT, connected to pin A of the temperature sensor's input terminal J1, and its collector is connected to the power supply AVDD_HV. Analog switch SW1 is located in the sampling resistor selection branch, used to select either sampling resistor Rs1 or sampling resistor Rs2 to be connected to the constant current source feedback loop. The non-inverting input of operational amplifier U1 receives a reference voltage Vref, and the inverting input is connected to the upper end of the selected sampling resistor. The lower end of the selected sampling resistor is grounded, so that the constant current output Iexc is determined by Iexc = Vref / Rs. When SW1 selects Rs1, the first constant current range Iexc1 is formed; when SW1 selects Rs2, the second constant current range Iexc2 is formed, or it can switch between two different sampling resistors Rs1 and Rs2. When SW1 selects Vref1 and Rs1, the constant current output Iexc1 is equal to Vref1 divided by Rs1, and its value is approximately 1 mA; when SW1 selects Vref2 and Rs2, the constant current output Iexc2 is equal to Vref2 divided by Rs2, and its value is approximately 0.1 mA. The first range Iexc1 is suitable for low-resistance PTC100 type sensors, and the second range Iexc2 is suitable for high-resistance PTC1000 and NTC5K and NTC10K type sensors.

[0031] The temperature sensor's input terminal J1 has three pins, labeled A, B, and C. Pins A and B form the main excitation circuit, while pin C serves as the auxiliary sensing terminal. In a two-wire application, only pins A and B are used, with both ends of the sensor connected to these two pins respectively. In a three-wire application, pins A and B are connected to the same end of the sensor, forming a redundant path to compensate for lead resistance, and pin C is connected to the other end of the sensor. This design allows the circuit to adapt to two mainstream industrial wiring standards without changing the hardware.

[0032] The resistance bias calibration circuit includes an analog switch SW2, which is controlled by the central control unit. In the measurement state, SW2 is open, and a constant current Iexc flows through the sensor, generating a sampling voltage Vsense across its terminals. In the calibration state, SW2 is closed, directly shorting pins A and B. At this time, the constant current loop does not contain the sensor itself, and the sampled voltage Vbias fully reflects the additional voltage drop introduced by the external connection path, including the equivalent bias voltage generated by lead resistance, contact resistance, and wiring parasitic impedance. The central control unit calculates the bias resistance Rbias based on the current constant current level, expressed as Rbias = Vbias / Iexc. In subsequent measurement states, this bias component is subtracted from the original sampling voltage Vsense to obtain the corrected sensor voltage Vsensor = Vsense − Vbias, and then the actual resistance Rsensor = Vsensor / Iexc is calculated. In a three-wire application, the positive input terminal of the high-precision instrument amplification module is connected to pin B, and the negative input terminal is connected to pin C. Since pin B does not carry the main excitation current or only carries a very small induced current in the three-wire system, the measured differential voltage is closer to the actual voltage across the sensor terminals, while still retaining the ability to periodically enter the calibration state to track changes in contact resistance caused by environmental aging or mechanical vibration.

[0033] like Figure 3 As shown, in the calibration state, the microcontroller unit (MCU) controls analog switch SW2 to close, shorting pins A and B of terminal J1. Analog switch SW4 remains open to prevent the parallel clamping resistor Rsh from participating in bias acquisition. Analog switch SW3 maintains a through path or voltage divider path corresponding to the current sensor type, ensuring that the calibration state and measurement state pass through the same post-stage filtering, amplification, and ADC sampling link. The voltage acquired at this time is used to characterize the equivalent bias component caused by terminal contact resistance, lead resistance, analog switch on-resistance, and residual bias of the sampling link.

[0034] like Figure 4As shown, in measurement mode, the microcontroller unit (MCU) controls analog switch SW2 to open, allowing the constant current source output current Iexc to flow through the temperature sensor to form a sampling voltage Vsense. When the current sensor is a PTC type or a low-resistance thermistor type, analog switch SW3 switches to a direct signal path, allowing Vsense to enter the high-precision instrumentation amplification module through a low-pass filter. When the current sensor is an NTC type or a high-resistance thermistor type, analog switch SW3 switches to a voltage divider path, allowing the voltage divider output Vdiv to enter the low-pass filter. When Vdiv is greater than the preset full-scale threshold Vth, the MCU controls analog switch SW4 to close, connecting the parallel clamping resistor Rsh in parallel with the temperature sensor to reduce the equivalent impedance and limit the input voltage of the subsequent stage.

[0035] The NTC voltage divider and range protection circuit includes a voltage divider resistor network Rdiv1 and Rdiv2, a parallel clamping resistor Rsh, and corresponding analog switches SW3 and SW4. Voltage divider resistors Rdiv1 and Rdiv2 are connected in series between the constant current output terminal IOUT and ground, with the intermediate node serving as the voltage divider output Vdiv. Analog switch SW3 is positioned between the voltage divider output Vdiv and the input terminal of the low-pass filter. In PTC mode, it conducts a direct signal path, meaning the constant current output terminal IOUT is directly connected to the low-pass filter; in NTC mode, it conducts a voltage divider path, meaning Vdiv is fed into the low-pass filter. The parallel clamping resistor Rsh is connected across the sensor via analog switch SW4, bridging pins A and C. When the detected sampling voltage approaches the full-scale threshold of the analog-to-digital converter, the central control unit closes SW4, creating a parallel equivalent impedance Req = Rsensor∥Rsh between the sensor and Rsh, effectively limiting the maximum output voltage. The voltage divider output ratio is set to Rdiv2 / (Rdiv1+Rdiv2), and the corresponding voltage divider attenuation coefficient k is (Rdiv1+Rdiv2) / Rdiv2, so that the voltage divider output Vdiv=Vsense / k; for example, when Rdiv1 is 90 kΩ and Rdiv2 is 10 kΩ, the voltage divider output ratio is 1 / 10 and the voltage divider attenuation coefficient k is 10, thereby ensuring that the signal after voltage division is still within the effective input window of the analog-to-digital converter under the extreme resistance conditions of NTC low temperature.

[0036] The low-pass filter consists of a first-order RC structure composed of a resistor Rlp and a capacitor Clp, positioned after the voltage divider / straight-through path and before the high-precision instrumentation amplifier module. Its cutoff frequency fc is determined by the formula fc = 1 / (2πRlpClp), ranging from 10 Hz to 100 Hz. Here, Rlp is a resistor in the thousands of ohms range, typically 10 kiloohms; Clp is a capacitor in the microfarad range, typically 1 microfarad. This filter suppresses switching transients, electromagnetic interference, and common-mode noise components, ensuring the purity of the signal entering the high-precision amplification stage.

[0037] The high-precision instrumentation amplifier module employs a three-op-amp architecture or a dedicated instrumentation amplifier chip. Its gain is determined by the external gain setting resistor Rg, and the expression for gain G is G = 1 + (2Rint / Rg), where Rint is the internal fixed feedback resistor. The gain range covers 1 to 50 times. The central control unit dynamically adjusts the resistance value of Rg or switches the internal gain level according to the currently used sensor type and constant current range. In PTC100 measurement mode, the gain is set to 5 to 20 times; in PTC1000 measurement mode, the gain is set to 10 to 50 times; in NTC measurement mode, due to voltage division attenuation, the gain is set to 1 to 10 times. The output of the instrumentation amplifier module is connected to the analog-to-digital converter input ADC_IN via a current-limiting resistor Rin.

[0038] The analog-to-digital converter (ADC) interface includes a current-limiting resistor Rin and bidirectional clamping diodes D1 and D2. The current-limiting resistor Rin is connected in series between the output of the instrumentation amplifier module and the input of the ADC, ADC_IN, to limit potential inrush current and, together with the ADC input capacitor, forms a two-stage low-pass filter. Clamping diode D1 is a Schottky diode, with its anode connected to ADC_IN and its cathode connected to the power rail AVDD; clamping diode D2 is also a Schottky diode, with its cathode connected to ADC_IN and its anode connected to ground GND. Together, they provide a fast discharge path when the input voltage exceeds AVDD or falls below GND, protecting the microcontroller from electrostatic discharge or abnormal overvoltage damage.

[0039] The central control unit is an embedded microcontroller that integrates GPIO, ADC, timers and communication interfaces. This unit performs operations in a predetermined sequence: First, it determines the sensor type and wiring method based on user configuration or automatic identification results, and sets the constant current level, voltage divider / clamping state, and instrument amplification gain accordingly; then, it controls analog switch SW2 to enter calibration mode, acquires the bias voltage Vbias, and stores the corresponding bias resistance value Rbias; next, it switches to measurement mode, acquiring the filtered and amplified voltage signal Vadc; then, it performs reverse processing on Vadc according to the current operating mode. If it is in NTC voltage divider mode, it first restores the ADC acquired value Vadc to the pre-amplified voltage divider output Vdiv according to the instrument amplification gain G, Vdiv=Vadc / G; then, it restores the original sampling voltage Vsense according to the voltage divider attenuation coefficient k, Vsense=Vdiv×k, i.e., Vsense=(Vadc / G)×(Rdiv1+Rdiv2) / Rdiv2; if it is in PTC pass-through mode, then Vsense=Vadc / G; finally, it combines the constant current value Iexc and the bias voltage Vbias. The sensor resistance is calculated; finally, based on the pre-stored PTC or NTC temperature-resistance mapping relationship, the final temperature value is obtained through table lookup or linear interpolation. Throughout the operation, the central control unit periodically triggers the calibration process, such as performing a bias calibration every 10 minutes or each power-on initialization, to compensate for lead impedance drift caused by the passage of time.

[0040] All analog switches SW1, SW2, SW3, and SW4 employ low on-resistance, high-isolation CMOS analog switching devices, with typical on-resistance less than 1 ohm and turn-off isolation greater than 80 dB. Their control signals are output from the central control unit via independent GPIO pins. The switching sequence of each switch is precisely controlled by software to ensure state stability before signal acquisition. Electrical connections between modules employ a single-point grounding strategy. The power supply for the constant current source module is regulated by an independent LDO regulator, with output ripple less than 10 microvolts, ensuring the long-term stability of the reference voltage Vref. The entire detection circuit is packaged on a single printed circuit board. The sensor access terminal J1 uses a shielded connector, and the casing is grounded to improve anti-interference capability and field adaptability.

[0041] In one specific embodiment, the system is configured to measure a PTC100 sensor using a two-wire connection. The central control unit first sets the constant current source to the first setting, Iexc1 = 1 mA, closes SW3 (straight-through mode), closes SW4 (no clamping), and sets the instrumentation gain to 10x. Then, SW2 is closed to enter calibration mode, acquiring Vbias = 8 mV, and calculating Rbias = 8 ohms. After switching to measurement mode, Vadc = 1.25 V is acquired. Since the gain is 10x, the differential input voltage of the instrumentation amplifier is 125 mV. After subtracting the bias, Vsensor = 125 mV - 8 mV = 117 mV. Therefore, Rsensor = 117 ohms is calculated, corresponding to the nominal resistance of the PTC100 at approximately 30 degrees Celsius.

[0042] In another embodiment, the system is configured to measure an NTC10K sensor using a three-wire connection. The central control unit sets the constant current source to the second setting, Iexc2 = 0.1 mA, turns on SW3 (enables voltage divider), with a voltage division ratio of 1:10 (i.e., Rdiv1 = 90 kΩ, Rdiv2 = 10 kΩ), and turns off SW4. The instrumentation gain is set to 5x. In calibration mode, Vbias = 2 mV and Rbias = 20 Ω are measured. In measurement mode, Vadc = 0.8 V, corresponding to an instrumentation amplifier input of 160 mV. Due to the voltage divider mode, the original sampling voltage Vsense = 160 mV × 10 = 1.6 V. After deducting the bias, Vsensor = 1.6 V − 0.002 V ≈ 1.598 V. Therefore, Rsensor = 1.598 V / 0.1 mA = 15.98 kΩ. According to the NTC10KR-T table, the corresponding temperature is approximately -10 degrees Celsius.

[0043] To verify the technical effectiveness of this invention, a comparative test was conducted using a standard example. The comparative example employed a conventional detection circuit with a fixed 1 mA constant current source, no bias calibration, and no voltage divider or clamping protection; the remaining signal conditioning was the same as the ADC section. The test object was an NTC10K sensor, with a resistance of approximately 330 kΩ at -40°C. In the comparative example, the sampling voltage Vsense = 330 kΩ × 1 mA = 330 V, far exceeding the ADC input range (0 to 3.3 V), causing signal saturation and preventing the acquisition of valid data. In this embodiment of the invention, the second setting Iexc2 = 0.1 mA is enabled, and the voltage divider ratio is 1:10. The sampling voltage Vsense = 330 kΩ × 0.1 mA = 33 V, which becomes 3.3 V after voltage division, just at the edge of the ADC's full-scale range. If SW4 is further enabled in parallel with Rsh = 50 kΩ, the equivalent impedance Req = 330∥50 ≈ 43.4 kΩ, and the sampling voltage drops to 4.34 V. After voltage division, it becomes 0.434 V, leaving sufficient dynamic margin.

[0044] To verify the technical effect of this invention, a comparative test was conducted using a comparative example. The comparative example used a traditional fixed 1 mA constant current source, a detection circuit without bias calibration, voltage divider, or clamping protection; the remaining signal conditioning was the same as the ADC section. The test results are shown in Table 1. The sensor resistance value was entered according to the nominal or measured resistance value at the corresponding temperature. The sampling voltage of the comparative example was calculated as the product of the fixed constant current source current and the sensor resistance value. The voltage after range protection in this invention is the input voltage of the subsequent stage after constant current range switching, voltage divider path, or parallel clamping processing.

[0045] Table 1

[0046] 1 PTC100 0 100 0.1 no 0.092 yes 0.2 2 PTC100 100 138.5 0.1385 no 0.130 yes 0.3 3 PTC1000 0 1000 1.0 no 0.992 yes 0.4 4 PTC1000 100 1385 1.385 no 1.375 yes 0.5 5 NTC10K 25 10 0.01 no 0.0098 yes 0.3 6 NTC10K −10 16 0.016 no 0.0158 yes 0.4 7 NTC10K −40 330 330 (overflow) yes 3.3 (After voltage division) yes 0.8 8 NTC5K −40 660 660 (overflow) yes 3.0 (after clamping) yes 1.0

[0047] As shown in Table 1, the comparative example easily exhibits sampling voltages far exceeding the full-scale range of the ADC under low-temperature conditions with high-resistance NTC sensors, leading to input saturation in the subsequent stage and the inability to acquire effective temperature data. This invention limits the input voltage of the subsequent stage to the effective range of the ADC by reducing the constant current level, enabling the voltage divider attenuation path, and connecting a parallel clamping resistor near full scale. Simultaneously, in PTC-type low-resistance sensor measurements, this invention reduces zero-point errors caused by lead resistance and contact resistance through bias calibration, thereby improving measurement stability in the low-resistance range.

[0048] Furthermore, in the three-wire connection method, this invention achieves natural suppression of lead resistance by connecting the positive terminal of the instrumentation amplifier to pin B (non-excitation terminal) and the negative terminal to pin C (sensing terminal). Experiments show that with a 10-meter-long copper wire connection (single-wire resistance of approximately 0.35 ohms), the traditional two-wire system for measuring the PTC100 introduces a positive deviation of approximately 0.9°C at 0°C, while the deviation is reduced to less than 0.1°C after the three-wire system of this invention with bias calibration.

[0049] In summary, this invention integrates four functional modules—switchable constant current source, dynamic bias calibration, NTC voltage divider and range protection, and high-precision differential signal conditioning—into a unified hardware platform. The central control unit implements parameter linkage and adaptive state control, enabling wide-range, high-precision, and highly robust measurements of various temperature sensors, including PTC100, PTC1000, NTC5K, and NTC10K, in both two-wire and three-wire connection configurations. This solution requires no replacement of the front-end hardware; it adapts to different sensor types and operating conditions solely through software configuration, demonstrating excellent engineering practicality and industrialization prospects.

[0050] In conclusion, the above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A wide-range configurable temperature sensor detection circuit, characterized in that, include: Constant current source module, terminal J1, resistance bias calibration circuit, NTC voltage divider and range protection circuit, low-pass filter, high-precision instrument amplification module, ADC interface and protection, microcontroller unit (MCU); The constant current source module includes an operational amplifier U1, a power transistor Q1, a sampling resistor Rs1, a sampling resistor Rs2, and an analog switch SW1. The microcontroller unit MCU controls SW1 to select either Rs1 or Rs2 to output the constant current source current Iexc. The temperature sensor is connected through pins A, B, and C of terminal J1. The resistance bias calibration circuit includes an analog switch SW2. In the calibration state, SW2 shorts pins A and B to acquire the bias voltage Vbias and calculate the bias resistance Rbias. In the measurement state, SW2 is disconnected and acquires Vsense to calculate the sensor resistance Rsensor. The NTC voltage divider and range protection circuit includes analog switch SW3 and analog switch SW4. The NTC voltage divider and range protection circuit includes voltage divider resistors Rdiv1 and Rdiv2, which are connected in series to form a voltage divider output node. The output ratio of the voltage divider output node is Rdiv2 / (Rdiv1+Rdiv2), and is expressed by the voltage divider attenuation coefficient k as k=(Rdiv1+Rdiv2) / Rdiv2. In NTC mode, analog switch SW3 connects the voltage divider output node to a low-pass filter. Vdiv passes through the low-pass filter and the high-precision instrument amplification module, and is then acquired by the ADC interface and protection as Vadc, making Vdiv=Vsense / k. In PTC mode, analog switch SW3 switches to the direct signal path. The NTC voltage divider and range protection circuit includes a parallel clamping resistor Rsh. When the voltage divider output Vdiv is greater than the preset full-scale threshold Vth, the microcontroller MCU controls the analog switch SW4 to enter the clamping state, so that the parallel clamping resistor Rsh and the temperature sensor are connected in parallel to form an equivalent resistance Req. Req is the parallel connection of Rsensor and Rsh. The preset full-scale threshold Vth is set according to the full-scale voltage of the ADC. The microcontroller unit (MCU) obtains the temperature based on a PTC or NTC temperature and resistance table.

2. The wide-range configurable temperature sensor detection circuit according to claim 1, characterized in that: Terminal J1 has pins A, B, and C. In a two-wire connection, the two ends of the temperature sensor are connected to pins A and B, respectively. In a three-wire connection, pins A and B are connected to the same end of the temperature sensor, and pin C is connected to the other end of the temperature sensor. The differential input terminals of the high-precision instrument amplification module are connected to pins B and C, respectively.

3. The wide-range configurable temperature sensor detection circuit according to claim 1, characterized in that: In calibration mode, the microcontroller unit (MCU) acquires the bias voltage Vbias and calculates the bias resistance value Rbias based on the constant current source current Iexc. In measurement mode, the MCU acquires Vsense and calculates the sensor resistance value Rsensor based on the bias resistance value Rbias.

4. The wide-range configurable temperature sensor detection circuit according to claim 1, characterized in that: The low-pass filter is a first-order RC structure, including a resistor Rlp and a capacitor Clp. Its cutoff frequency fc is equal to one divided by two times pi multiplied by Rlp and Clp.

5. The wide-range configurable temperature sensor detection circuit according to claim 2, characterized in that: The high-precision instrument amplification module adopts a three-operation amplifier architecture or a dedicated instrument amplifier chip U3. The gain of the high-precision instrument amplification module is determined by the gain setting resistor Rg and the internal resistor Rint. The gain G is equal to one plus two times Rint and then divided by Rg.

6. The wide-range configurable temperature sensor detection circuit according to claim 2, characterized in that: The ADC interface and protection include a series current-limiting resistor Rin and clamping diodes D1 and D2. The current-limiting resistor Rin is connected in series between the output terminal of the high-precision instrument amplification module and the ADC input terminal ADC_IN. Clamping diode D1 clamps ADC_IN to the power supply VDD, and clamping diode D2 clamps ADC_IN to ground GND. Clamping diodes D1 and D2 are Schottky diodes.

7. The wide-range configurable temperature sensor detection circuit according to claim 1, characterized in that: The microcontroller unit (MCU) processes the acquired Vadc data. In NTC mode, it restores Vadc to Vsense based on the instrument amplification gain G and the voltage divider attenuation coefficient k, and calculates the sensor resistance Rsensor. In PTC mode, it restores Vadc to Vsense based on the instrument amplification gain G, and calculates the sensor resistance Rsensor in conjunction with the bias resistance Rbias. The MCU obtains the temperature by looking up a table or using linear interpolation based on the PTC or NTC temperature-resistance table.

8. The wide-range configurable temperature sensor detection circuit according to claim 7, characterized in that: The microcontroller unit (MCU) configures the constant current range SW1 and the instrument amplification gain according to the sensor model and wiring method, and controls the analog switch SW2 to switch between calibration and measurement states to refresh the bias resistance value Rbias and track the bias changes caused by wiring harness aging; it also determines the voltage divider path selection through SW3 and enters the clamping state control through SW4.

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