A method for suppressing high frequency oscillation in a flexible direct current power transmission system

Abstract

This invention discloses a high-frequency oscillation suppression method in flexible DC transmission systems, solving the mid-to-high frequency oscillation instability problem caused by time delay in existing converter stations. To achieve negative resistance suppression in the mid-to-high frequency range of modular multilevel converters, an impedance matrix of the converter is obtained based on a multi-harmonic linearization impedance modeling method. A simplified impedance method for the mid-to-high frequency range is proposed based on the control loop characteristics, separating the high-frequency delay-added impedance and the mid-frequency current loop-added impedance. Appropriate filters are added to the voltage feedforward branch and the feedback branch of the current loop, and reasonable parameters are designed to improve the negative resistance phenomenon in the mid-to-high frequency range, achieving mid-to-high frequency impedance reshaping of the modular multilevel converter. This provides a foundation for stability analysis of flexible DC transmission systems and control optimization design of converter stations, and is of great significance for the safe and stable operation of wind farm-converter station systems. It has significant engineering application value and is suitable for widespread promotion and use.

Description

Technical Field

[0001] This invention relates to the field of power system analysis technology, and specifically to a method for suppressing high-frequency oscillations in a flexible DC transmission system. Background Technology

[0002] With the continuous increase in the installed capacity and proportion of new energy sources, mainly wind and solar power, high-voltage direct current (HVDC) transmission based on modular multilevel converters and multi-terminal DC grids are gradually becoming important methods for large-scale transmission of new energy from areas without synchronous power sources. Wind power, solar power, and other new energy generation systems transmitted via flexible DC transmission all contain power electronic equipment. Large-scale grid-connected new energy systems form localized high-proportion new energy and high-proportion power electronics power systems. The characteristics of these localized high-proportion power systems are dominated by the control characteristics of the aforementioned power electronic devices, and the system's operating characteristics will undergo profound changes compared to traditional power systems.

[0003] In recent years, grid connection failures caused by broadband oscillations have severely impacted the safe and stable operation of new energy systems. Impedance analysis is commonly used to analyze the stability of complex power electronic systems. For interconnected systems, the first step is to establish impedance models for each subsystem, and then analyze the overall system stability by determining whether the impedance ratio satisfies the Nyquist stability criterion. To analyze system oscillation problems, it is necessary to first model and analyze the port impedance of the converter station. Considering the complex dynamic harmonics within modular multilevel converters, a multi-harmonic linearized impedance modeling method is required. Based on its multi-input, multi-output model characteristics, the influence of all frequency coupling paths is fully considered to establish the equivalent impedance model of the AC port of the modular multilevel converter.

[0004] Analysis of port impedance characteristics revealed negative resistance in the mid-to-high frequency impedances of the modular multilevel converter's AC side. This negative resistance, when intersecting with the AC-side wind farm impedance, negatively impacts system stability. Therefore, to ensure the safe and stable operation of the wind farm-to-sending-end converter station system, it is necessary to suppress the negative resistance in the mid-to-high frequency impedance of the modular multilevel converter.

[0005] Studies have shown that due to the suppression effect of bridge arm inductance and submodule capacitance on high-frequency disturbances, frequency coupling mainly acts in the low-frequency range without affecting the mid-to-high frequency impedance of modular multilevel circuits. The port impedance matrix exhibits a diagonally dominant characteristic in the mid-to-high frequency band, thus simplifying the processing of mid-to-high frequency impedances. Some studies have proposed using a combination of passive filtering devices and low-pass filters to improve the negative damping characteristics of converters. Adding passive filtering devices can prevent specific frequency oscillation signals from passing through the converter, but passive filtering devices suffer from high manufacturing costs and large footprint. Other studies have proposed that the AC voltage feedforward stage is the main cause of high-frequency negative damping and oscillations. Filters can be introduced into the voltage feedforward path to block voltage disturbances, thereby eliminating the negative damping caused by voltage feedforward. However, further analysis shows that AC current control in the mid-frequency range also generates significant negative damping. Treating only the voltage feedforward branch leads to a deterioration of the mid-frequency negative resistance range, causing mid-frequency oscillations, and filter design still relies on trial and error. Still other studies have proposed an adaptive suppression strategy: extracting the oscillation frequency through fast Fourier transform and using a notch filter to suppress this frequency component. However, this method cannot handle situations where multiple resonant frequencies occur simultaneously, and notch filters may induce another oscillation while suppressing one. Therefore, researching high-frequency negative impedance suppression methods in modular multilevel converters and reshaping their high-frequency impedances is of great significance for ensuring the stability of interconnected systems. Summary of the Invention

[0006] This invention provides a method for suppressing high-frequency oscillations in a flexible DC transmission system to address the problem of instability caused by time delays in existing converter stations.

[0007] To achieve the above objectives, the technical solution of the present invention is as follows:

[0008] A method for suppressing high-frequency oscillations in a flexible DC transmission system includes the following steps:

[0009] S1. Obtain the AC side port impedance of the modular multilevel converter based on modeling analysis;

[0010] S2 uses impedance frequency segmentation analysis to obtain the high-frequency impedance of the modular multilevel converter, and adds a voltage feedforward filter to the voltage feedforward branch in the control loop to reshape the high-frequency impedance and suppress the high-frequency oscillation caused by the negative damping characteristics in the high-frequency band.

[0011] S3, after high-frequency impedance reshaping, uses impedance frequency segmentation analysis to obtain the intermediate frequency impedance of the modular multilevel converter, and reshapes the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop to suppress the intermediate frequency oscillation caused by the negative damping characteristics in the intermediate frequency band.

[0012] Furthermore, in S2, the high-frequency impedance reshaping achieved by adding a voltage feedforward filter to the voltage feedforward branch in the control loop, thereby suppressing high-frequency oscillations caused by negative damping characteristics in the high-frequency band, satisfies the following constraints:

[0013] 1) The amplitude of the sinusoidal quantity in the real part of the delayed additional admittance obtained from the high-frequency impedance of the modular multilevel converter decays to zero in the high-frequency range, as expressed by:

[0014] ;

[0015] Among them, R ffv X is the real part of the voltage feedforward filter. ffv This represents the imaginary part of the voltage feedforward filter;

[0016] 2) The bandwidth of the voltage feedforward filter is far from the bandwidth of the current loop, expressed as:

[0017] ;

[0018] Among them, W ffv W represents the bandwidth of the voltage feedforward filter. ci This represents the bandwidth of the current loop.

[0019] Furthermore, in S3, the reshaping of the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop satisfies the following constraints:

[0020] 1) The real part of the additional admittance of the current loop obtained from the intermediate frequency impedance of the converter is positive in the intermediate frequency range, expressed as:

[0021] ;

[0022] Where, r vir x is the real part of the voltage feedforward filter. vir This represents the imaginary part of the voltage feedforward filter;

[0023] 2) The phase lag introduced by the current feedback filter at the current loop crossover frequency does not exceed the phase margin of the current loop, expressed as:

[0024] ;

[0025] Where ξ is the damping coefficient of the current feedback filter (0.707), ω ci ω is the crossing angular frequency of the current loop. vir This is the cutoff angular frequency of the current feedback filter.

[0026] Further, in S1, obtaining the AC side port impedance of the modular multilevel converter based on modeling analysis includes the following steps:

[0027] Step (A) Establish a modular multilevel converter control unit and acquire the converter MMC arm modulation signal; Step (B) Derive the equivalent model of the controlled source of the converter based on the converter arm modulation signal.

[0028] Step (C): Based on the controlled source equivalence principle, construct the MMC bridge arm time-domain model according to the modulation signal;

[0029] Step (D): Based on the multi-harmonic linearization method, the internal state variables of the bridge arm are expanded into frequency domain harmonic vectors to establish the MMC frequency domain small-signal model.

[0030] Step (E): Obtain the AC side port impedance matrix of the converter based on the frequency domain small-signal model.

[0031] Further, the establishment of the modular multilevel converter control unit in step (A) to acquire the converter MMC arm modulation signal specifically involves...

[0032] The modular multilevel converter control unit includes a modular multilevel converter and a control unit. The control unit includes a voltage loop, a current loop, and a capacitor voltage equalization module. The control unit samples the AC voltage V. abc Alternating current i abc v is obtained after abc / dq coordinate transformation dq i dq Then v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dq With the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s); Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc The upper and lower arm modulation signals m of the modular multilevel converter are obtained through level modulation and capacitor voltage equalization modules. lu m ld ; where m lu m ld The expression is as follows:

[0033] ;

[0034] Where M1 is the fundamental frequency modulation ratio, The fundamental angular frequency, The phase of the fundamental frequency component of the AC voltage. is the phase shift angle of the three phases, where l represents any one of the three phases;

[0035] The controlled source equivalent model of the converter is derived based on the bridge arm modulation signal of the converter in step (B), specifically as follows:

[0036] Based on the upper and lower arm modulation signals m of the modular multilevel converter lu m ld The controlled source equivalent model of the converter is derived, and the capacitors of all sub-modules connected in series in the bridge arm are represented by an equivalent capacitor C. eq To represent; in the controlled source equivalent model, the current flowing through the equivalent capacitance of the upper and lower bridge arms is controlled by the current source m. lu i lu m ld i ld Equivalent; the output voltage of the upper and lower bridge arms is controlled by the voltage source m lu v lu m ld v ld Equivalent capacitance C eq The expression is as follows:

[0037] Among them, C sm N represents the capacitance value of a single submodule in the bridge arm, and N is the total number of series-connected submodules in the bridge arm.

[0038] In step (C), the construction of the MMC bridge arm time-domain model based on the controlled source equivalence principle and the modulation signal is specifically as follows:

[0039] Based on the equivalent model of the controlled source of the modular multilevel converter, the time-domain model of the bridge arm is derived. Since the six bridge arms of the three phases of the modular multilevel converter satisfy symmetry, only the upper bridge arm of a single phase needs to be modeled. The specific time-domain expression of the upper bridge arm model of the single phase of the modular multilevel converter is as follows.

[0040] ;

[0041] Among them, v lu_c The sum of the capacitor voltages of all submodules in the upper bridge arm, v lO' V is the AC port voltage. dc For DC side voltage, i lu R is the current in the upper arm of the bridge. arm L arm For the upper bridge arm, a resistor and an inductor are connected in series, v O'O This represents the neutral point potential difference between the AC and DC sides. Since the three-phase AC system is balanced, the sum of the three-phase AC voltages and currents is zero. The AC neutral point voltage is the sum of the zero-sequence components of the differential-mode voltages of the three-phase bridge arms. O'O The expression is as follows:

[0042] ;

[0043] in, This represents the zero-sequence component of the differential-mode voltage of a single-phase bridge arm.

[0044] Step (D) describes the use of a multi-harmonic linearization method to expand the internal state variables of the bridge arm into frequency-domain harmonic vectors, establishing an MMC frequency-domain small-signal model. Specifically...

[0045] Impedance modeling based on multi-harmonic linearization modulates the bridge arm modulation signal m of the internal state variable of the modular multilevel converter. lu Bridge arm current i lu Bridge arm equivalent capacitance voltage v lu_c Expanded in the frequency domain into the form of harmonic vectors m lu i lu v lu_c A frequency domain small-signal model of the modular multilevel converter is established based on the bridge arm time-domain model; the frequency sequence corresponding to the harmonic vector is [-nf1,…,-f1,0,f1,…,nf1]. T Where n is a positive integer, f1 is the fundamental frequency, and each element in the harmonic vector is the Fourier coefficient of the corresponding steady-state harmonic. The expression for the harmonic steady-state vector in the frequency domain is as follows.

[0046] ;

[0047] The expression for the small-signal model of the bridge arm is as follows:

[0048] ;

[0049] Here, "^" represents a small perturbation. After introducing the small signal, the center frequency corresponding to each small signal vector is the perturbation frequency f. p At this time, the perturbation frequency sequence corresponding to the small signal vector is [f p -nf1,…,f p -f1,f p ,f p +f1,…,f p +nf1] T Y Ceq Z L The equivalent capacitance admittance matrix and impedance matrix of the bridge arm at each frequency under small-signal perturbation are expressed as follows.

[0050] ;

[0051] , These are small-signal vectors representing the AC port voltage and the AC center point voltage. , , For i lu v lu m luThe corresponding small-signal vector in the frequency domain is expressed as follows:

[0052] ;

[0053] I lu V lu M lu For i lu v lu m lu The corresponding steady-state value matrix is ​​expressed as follows:

[0054] ;

[0055] Because of v O'O In steady state, the zero-sequence differential-mode component of the bridge arm voltage is canceled out, resulting in no zero-sequence current path at the AC port. This corresponds to setting the zero-sequence differential-mode element of the bridge arm impedance matrix in the mathematical model to zero. After introducing small-signal quantities, the phase sequence division of each harmonic is as follows, and the differential-mode component division expression is as follows.

[0056] ;

[0057] The common-mode component partitioning expression is as follows:

[0058] ;

[0059] For the Z in the bridge arm impedance matrix L The zero-order differential modulus component is corrected by setting it to zero to obtain Z'. L Then, in the small signal equation Negligible; Corrected Z' L The expression is as follows:

[0060] ;

[0061] The corrected expression for the small-signal equation in the frequency domain is as follows.

[0062] ;

[0063] In step (E), the AC side port impedance matrix of the converter is obtained based on the frequency domain small-signal model, specifically as follows:

[0064] The AC-side port impedance matrix of the modular multilevel converter is obtained based on the frequency domain small-signal model. Since the positive and negative sequence impedances of the sending-end converter station are symmetrical, only the positive sequence impedance needs to be analyzed. The positive sequence AC side port impedance of the modular multilevel converter is shown in the following formula.

[0065] ;

[0066] Where E is the identity matrix, P represents the transfer function matrix from the small-signal vector of the upper arm current to the small-signal vector of the modulating signal, Q represents the transfer function matrix from the small-signal vector of the AC port voltage to the small-signal vector of the modulating signal, and the small-signal vector of the bridge arm modulating signal... The relationship between the small-signal vectors of the AC port voltage and the bridge arm current is shown in the following equation.

[0067] ;

[0068] The transfer function matrices P and Q contain only non-zero sequence differential mode components, and different harmonic phase sequences correspond to different transfer functions. The specific expression for P is as follows.

[0069] ;

[0070] The specific expression for Q is as follows.

[0071] .

[0072] Furthermore, in S2, the high-frequency impedance of the modular multilevel converter is obtained by impedance frequency band analysis, specifically as follows:

[0073] For high-frequency bands, the voltage feedforward branch plays a major role in the control loop, while the PI controllers in the voltage loop and current loop... v (s), G i Neglecting the effect of (s), when f > 1000Hz, the high-frequency impedance expression for the AC side of the modular multilevel converter is as follows.

[0074] ;

[0075] In the formula, G d R is the system link delay transfer function. arm L is the bridge arm resistance. arm The inductance is the bridge arm inductance, and f is the frequency variable;

[0076] If system control delay is not considered, when f > 1000Hz, the AC side impedance of the converter is mainly composed of the bridge arm impedance Z. arm =0.5(R arm +sL arm Dominated by ) and exhibiting inductive characteristics; when the system has a delay T d At that time, voltage feedforward will delay the transfer function G. d Introduced into the high-frequency impedance, the high-frequency impedance on the AC side of the converter is now equivalent to that of the bridge arm impedance Z. arm And the additional impedance Z introduced by voltage feedforward due to time delay Gd Composed of parallel elements, the expression is as follows:

[0077] ;

[0078] Delayed additional impedance Z Gd When connected in parallel with the bridge arm impedance, a sinusoidal quantity with angular frequency ω appears in the real part of the high-frequency admittance. The expression for the real part of the high-frequency admittance is as follows.

[0079] .

[0080] Furthermore, in S2, the method of suppressing high-frequency oscillations caused by negative damping characteristics in the high-frequency band by adding a voltage feedforward filter to the voltage feedforward branch in the control loop to reshape the high-frequency impedance is specifically as follows:

[0081] To suppress the high-frequency negative damping characteristics, the delayed additional impedance Z is used. Gd From this perspective, a voltage feedforward filter is introduced into the voltage feedforward branch to reduce the effect of delay. Before determining the type of filter, it is expressed as a combination of real and imaginary parts, as shown in the following expression.

[0082] ;

[0083] Among them, G ffv For a voltage feedforward filter, R ffv X is the real part of the voltage feedforward filter. ffv This represents the imaginary part of the voltage feedforward filter;

[0084] After introducing a voltage feedforward filter into the voltage feedforward branch, taking the reciprocal of the delay-added impedance and separating its real part, the real part expression of the delay-added admittance is obtained as follows.

[0085] ;

[0086] According to the above formula, the real part of the delayed additional admittance still contains a sinusoidal quantity with angular frequency ω, and the amplitude of this sinusoidal quantity is... ;

[0087] To satisfy the constraint that the amplitude of the sinusoidal quantity attenuates to zero in the high-frequency range, a second-order low-pass filter is selected to suppress the high-frequency negative damping characteristics of the converter. The specific expression of the voltage feedforward filter is as follows.

[0088] ;

[0089] Where ξ is the filter damping coefficient, ω ffv f is the filter cutoff angular frequency. ffv ω is the filter cutoff frequency. ffv With cutoff frequency f ffv The relationship between them is ω ffv =2πf ffv ;

[0090] Further, in S3, the medium-frequency impedance of the modular multilevel converter obtained by impedance sub-band analysis is specifically as follows

[0091] In the medium-frequency range of 100 Hz < f < 1000 Hz, the effect of the voltage-loop PI controller G v (s) is ignored, and only the effect of the current loop in the control loop is considered. When 100 Hz < f < 1000 Hz, the expression of the additional impedance brought by the current loop is as follows

[0092] ;

[0093] Wherein, is the additional impedance of the current loop, and K pwm is the modulation gain;

[0094] The additional impedance of the current loop is equivalent to a series of an additional loop impedance on the basis of the original impedance. When 100 Hz < f < 1000 Hz, the AC-side impedance of the converter is equivalent to being composed of the high-frequency impedance and the additional impedance introduced by the current loop in series, and the expression is as follows

[0095] ;

[0096] Further, in S3, the reshaping of the medium-frequency impedance by adding a filter to the current feedback branch in the control loop is specifically as follows

[0097] Before determining the type of the filter, it is expressed in the form of the sum of the real part and the imaginary part, and the expression is as follows

[0098] ;

[0099] Wherein, G vir is the current feedback filter, r vir is the real part of the current feedback filter, and x vir is the imaginary part of the current feedback filter;

[0100] After adding the current feedback filter to the current feedback branch, taking the reciprocal of the additional impedance of the current loop and separating the real part, the expression of the real part of the additional admittance of the current loop is obtained as follows

[0101] ;

[0102] Wherein, k ii is the integral coefficient of the current controller; K pwm is the modulation gain; according to the above formula, in the frequency range of 100 Hz < f < 1000 Hz, 3π / 50 < ωT d < 3π / 5, according to the voltage feed-forward low-pass filter Gffv For the design, when 100 Hz < f < 1000 Hz, the conclusion obtained by comparing some expressions of the above formula with zero value is as follows:

[0103] ;

[0104] Considering that the medium-frequency impedance reshaping satisfies the constraint conditions, select the current feedback filter G vir As a second-order low-pass filter, the real part of the second-order low-pass filter in the frequency band after the cut-off frequency satisfies less than or equal to zero, which is in line with the constraint conditions. The specific expression of the second-order low-pass filter is as follows:

[0105] ;

[0106] Where ξ is the damping coefficient of the current feedback filter, and ω vir is the cut-off angular frequency of the current feedback filter, and ω vir and the cut-off frequency f vir The relationship between them is ω vir = 2πf vir .

[0107] The present invention also provides a high-frequency oscillation suppression device in a flexible DC power transmission system. The device includes a modular multilevel converter and a control unit. The control unit includes a voltage loop, a current loop, and a capacitor voltage balancing module, and

[0108] a modeling analysis module, which obtains the AC-side port impedance of the modular multilevel converter based on modeling analysis;

[0109] a high-frequency impedance reshaping module, which uses impedance sub-band analysis to obtain the high-frequency impedance of the modular multilevel converter, and reshapes the high-frequency impedance by adding a voltage feed-forward filter to the voltage feed-forward branch in the control loop to suppress the high-frequency oscillation caused by the negative damping characteristic in the high-frequency band;

[0110] a medium-frequency impedance reshaping module, which is used to, after reshaping the high-frequency impedance, use impedance sub-band analysis to obtain the medium-frequency impedance of the modular multilevel converter, and reshape the medium-frequency impedance by adding a current feedback filter to the current feedback branch in the control loop to suppress the medium-frequency oscillation caused by the negative damping characteristic in the medium-frequency band.

[0111] The present invention has the following beneficial effects:

[0112] This invention addresses the mid-to-high frequency oscillation instability problem caused by time delay in existing converter stations. To achieve negative resistance suppression in the mid-to-high frequency range of modular multilevel converters, an impedance matrix of the converter is obtained based on a multi-harmonic linearization impedance modeling method. According to the control loop characteristics, a simplified impedance method for the mid-to-high frequency range is proposed, separating the high-frequency delay-added impedance and the mid-frequency current loop-added impedance. Appropriate filters are added to the voltage feedforward branch and the feedback branch of the current loop, respectively, and reasonable parameters are designed to improve the negative resistance phenomenon (negative damping characteristics) in the mid-to-high frequency range, thus realizing the mid-to-high frequency impedance reshaping of the modular multilevel converter.

[0113] This invention takes into account both cost and area, ensuring that high-frequency negative resistance is suppressed while no additional resonance points appear in the mid-frequency band. The selection and parameters of the filter can also be reasonably designed. By segmenting and constraining the real part of the converter station admittance to be greater than zero, the precise reshaping of the mid-to-high frequency impedance is achieved, which is of great significance for the stability analysis of the wind farm to the sending-end converter station system.

[0114] This invention effectively divides frequency bands to achieve simplified impedance derivation, impedance reshaping, and oscillation suppression at high frequencies in converter stations. The parameter design process is also more specific and accurate, providing a foundation for the stability analysis of flexible DC transmission systems and the control optimization design of converter stations. It is of great significance for the safe and stable operation of wind farm-converter station systems, has significant engineering application value, and is suitable for widespread promotion and use.

[0115] The present invention discloses a method for suppressing high-frequency oscillations in a flexible DC transmission system, the specific design principle of which is as follows:

[0116] First, sample the AC voltage v. abc Alternating current i abc v is obtained through abc / dq coordinate transformation dq i dq Then v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dq With the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s). Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc After passing through the recent level modulation and capacitor voltage equalization module, the upper and lower arm modulation signals m of the modular multilevel converter are obtained. lu m ld Subsequently, based on the bridge arm modulation signal m of the modular multilevel converter... lu m ldThe controlled source equivalent model of the converter is derived, and the capacitors of all sub-modules connected in series in the bridge arm are represented by an equivalent capacitor C. eq To represent; then, based on the equivalent model of the controlled source of the modular multilevel converter, the time-domain model of the bridge arm is derived; next, based on the impedance modeling method of multi-harmonic linearization, the bridge arm modulation signal m, an internal state variable of the modular multilevel converter, is used to represent... lu Bridge arm current i lu Bridge arm equivalent capacitance voltage v lu_c Expanded in the frequency domain into the form of harmonic vectors m lu i lu v lu_c A frequency domain small-signal model of the modular multilevel converter is established based on the bridge arm time-domain model; then, the AC-side port impedance matrix of the modular multilevel converter is obtained based on the frequency domain small-signal model. Then through the impedance matrix Frequency band analysis yields the simplified high-frequency impedance of the modular multilevel converter. A high-frequency oscillation suppression method is proposed based on high-frequency impedance characteristics: by adding a suitable filter G to the voltage feedforward branch in the control loop. ffv The high-frequency impedance is reshaped to suppress high-frequency oscillations caused by negative damping characteristics in the high-frequency range; finally, the impedance matrix is ​​used... Frequency band analysis yields the simplified intermediate frequency impedance of the modular multilevel converter. Based on the mid-frequency impedance characteristics, a mid-frequency oscillation suppression method is proposed: by adding a suitable filter G to the current feedback branch in the control loop. vir The mid-frequency impedance is reshaped to suppress mid-frequency oscillations caused by negative damping characteristics in the mid-frequency range. Attached Figure Description

[0117] Figure 1 This is a block diagram of the main circuit and control structure of the modular multilevel converter in the converter station system of the present invention;

[0118] Figure 2 This is a block diagram of the single-phase controlled source equivalent circuit of the modular multilevel converter of the present invention;

[0119] Figure 3 This is a block diagram of the control structure of the medium-to-high frequency oscillation suppression method of the present invention;

[0120] Figure 4 This is a schematic diagram of system simulation frequency sweep verification of the AC port impedance of the converter station before and after the addition of the mid-to-high frequency oscillation suppression method of the present invention.

[0121] Figure 5 The present invention relates to the analytical values ​​and frequency sweep points of the AC port impedance characteristic curves of the converter station system before and after the application of the medium- and high-frequency oscillation suppression method in the voltage feedforward and current feedback branches. Detailed Implementation

[0122] The present invention will now be further described with reference to the accompanying drawings.

[0123] The design process of this invention is as follows:

[0124] like Figure 1 As shown, this invention provides a method for suppressing high-frequency oscillations in a flexible DC transmission system, including a sending-end converter station system. The sending-end converter station system includes a modular multilevel converter and a control unit. The control unit includes a voltage loop, a current loop, and a capacitor voltage equalization module. The modular multilevel converter converts AC power from the wind farm into DC power and injects it into the DC line, while maintaining stable operation on the AC side. The voltage loop controls the AC voltage, and its output serves as the reference value for the current loop. dq_ref The current loop is used to control the alternating current, and its output serves as the modulated wave v of the alternating voltage. Mdq The capacitor voltage equalization module is used to suppress capacitor voltage fluctuations in the modular multilevel converter bridge arm submodules. Its output is the modulation signal m of the upper and lower bridge arms of the modular multilevel converter. lu m ld .

[0125] like Figure 2 As shown, the controlled source equivalent circuit of the modular multilevel converter combines all the submodule capacitors connected in series in the bridge arm into a single equivalent capacitor C. eq This indicates that the current flowing through the equivalent capacitance of the upper and lower bridge arms is controlled by the current source m. lu i lu m ld i ld Equivalently, the output voltages of the upper and lower bridge arms are controlled by the voltage source m. lu v lu m ld v ld Equivalent to.

[0126] like Figure 3 As shown, this method connects a voltage feedforward filter G in series in the voltage feedforward branch. ffv The dq component of the AC side voltage is directly compensated for voltage disturbances by a second-order low-pass filter; a current feedback filter G is connected in series in the current feedback branch. vir The dq component of the AC side current is filtered by a second-order low-pass filter and then subtracted from the current reference value before being fed into the PI controller. The modular multilevel converter control unit samples the AC voltage v. abc Alternating current i abc v is obtained through abc / dq coordinate transformation dq i dq Then v dq With voltage reference vdq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dq With the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s). Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc The drive signal for the converter bridge arm submodule is obtained after the recent level modulation and capacitor voltage equalization module.

[0127] This invention includes the following steps:

[0128] Step (A), after the abc / dq coordinate transformation, v is obtained. dq i dq Then v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dq With the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s). Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc After passing through the recent level modulation and capacitor voltage equalization module, the upper and lower arm modulation signals m of the modular multilevel converter are obtained. lu m ld .

[0129] Step (B), based on the bridge arm modulation signal m of the modular multilevel converter lu m ld The controlled source equivalent model of the converter is derived, and the capacitors of all sub-modules connected in series in the bridge arm are represented by an equivalent capacitor C. eq To express.

[0130] Step (C): Based on the equivalent model of the controlled source of the modular multilevel converter, the time-domain model of the bridge arm is derived.

[0131] Step (D), based on the impedance modeling method of multi-harmonic linearization, modulates the bridge arm modulation signal m of the internal state variable of the modular multilevel converter. lu Bridge arm current i lu Bridge arm equivalent capacitance voltage v lu_c Expanded in the frequency domain into the form of harmonic vectors m lu i lu v lu_c A frequency domain small-signal model of the modular multilevel converter is established based on the bridge arm time-domain model.

[0132] Step (E): Obtain the AC side port impedance matrix based on the frequency domain small-signal model of the modular multilevel converter. ;

[0133] Step (F), through the impedance matrix Frequency band analysis yields the simplified high-frequency impedance of the modular multilevel converter. A high-frequency oscillation suppression method is proposed based on high-frequency impedance characteristics: by adding a suitable filter G to the voltage feedforward branch in the control loop. ffv Reshaping the high-frequency impedance suppresses high-frequency oscillations caused by negative damping characteristics in the high-frequency range;

[0134] Step (G), through the impedance matrix Frequency band analysis yields the simplified intermediate frequency impedance of the modular multilevel converter. Based on the mid-frequency impedance characteristics, a mid-frequency oscillation suppression method is proposed: by adding a suitable filter G to the current feedback branch in the control loop. vir The mid-frequency impedance is reshaped to suppress mid-frequency oscillations caused by negative damping characteristics in the mid-frequency range.

[0135] Impedance modeling based on multi-harmonic linearization yields the positive-sequence impedance of the AC side of the sending-end converter station under islanded operation mode. Then, based on the characteristics of the control loop, simplified impedance formulas for the converter station in the medium and high frequency bands are obtained respectively. , Then, filters G are connected in series in the current feedback branch and the voltage feedforward branch, respectively. vir G ffv Based on the constraint that the real part of the converter station admittance is greater than zero in the mid-to-high frequency band, a second-order low-pass filter is selected, and the cutoff frequency ω is designed. vir ω ffv This method achieves high-frequency impedance reshaping and oscillation suppression in converter stations, and the parameter design process is relatively specific and accurate. It provides a foundation for the stability analysis of flexible DC transmission systems and the control optimization design of converter stations. It is of great significance for the safe and stable operation of wind farm-converter station systems, has significant engineering application value, and is suitable for widespread promotion and use.

[0136] Example 1:

[0137] This example provides a method for suppressing high-frequency oscillations in a flexible DC transmission system, including the following steps:

[0138] S1. Obtain the AC side port impedance of the modular multilevel converter based on modeling analysis;

[0139] S2 uses impedance frequency segmentation analysis to obtain the high-frequency impedance of the modular multilevel converter, and adds a voltage feedforward filter to the voltage feedforward branch in the control loop to reshape the high-frequency impedance and suppress the high-frequency oscillation caused by the negative damping characteristics in the high-frequency band.

[0140] S3, after high-frequency impedance reshaping, uses impedance frequency segmentation analysis to obtain the intermediate frequency impedance of the modular multilevel converter, and reshapes the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop to suppress the intermediate frequency oscillation caused by the negative damping characteristics in the intermediate frequency band.

[0141] Example 2:

[0142] The further design of this example involves adding a voltage feedforward filter to the voltage feedforward branch in the control loop to reshape the high-frequency impedance, thereby suppressing high-frequency oscillations caused by the negative damping characteristics in the high-frequency range and satisfying the following constraints:

[0143] 1. The amplitude of the sinusoidal quantity in the real part of the delayed additional admittance obtained from the high-frequency impedance of the modular multilevel converter decays to zero in the high-frequency range, as expressed by:

[0144] ;

[0145] Among them, R ffv X is the real part of the voltage feedforward filter. ffv This represents the imaginary part of the voltage feedforward filter;

[0146] 2. The bandwidth of the voltage feedforward filter is far from the bandwidth of the current loop, as expressed as:

[0147] ;

[0148] Among them, W ffv W represents the bandwidth of the voltage feedforward filter. ci This represents the bandwidth of the current loop.

[0149] In S3, the reshaping of the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop satisfies the following constraints:

[0150] 1. The real part of the additional admittance of the current loop obtained from the intermediate frequency impedance of the converter is positive in the intermediate frequency range, i.e.

[0151] ;

[0152] Where, r vir x is the real part of the voltage feedforward filter. vir This represents the imaginary part of the voltage feedforward filter;

[0153] 2. The phase lag introduced by the current feedback filter at the current loop crossover frequency does not exceed the phase margin of the current loop, as expressed as:

[0154] ;

[0155] Where ξ is the damping coefficient of the current feedback filter (which can be taken as 0.707), ω ci ω is the crossing angular frequency of the current loop. vir This is the cutoff angular frequency of the current feedback filter.

[0156] Example 3:

[0157] The specific design of step S1 in this example is as follows: Figures 1-3 As shown, the AC side port impedance of a modular multilevel converter is obtained based on modeling analysis, including the following steps:

[0158] Step (A) Establish a modular multilevel converter control unit and acquire the converter MMC arm modulation signal; Step (B) Derive the equivalent model of the controlled source of the converter based on the converter arm modulation signal.

[0159] Step (C): Based on the controlled source equivalence principle, construct the MMC bridge arm time-domain model according to the modulation signal;

[0160] Step (D): Based on the multi-harmonic linearization method, the internal state variables of the bridge arm are expanded into frequency domain harmonic vectors to establish the MMC frequency domain small-signal model.

[0161] Step (E): Obtain the AC side port impedance matrix of the converter based on the frequency domain small-signal model.

[0162] Example 4:

[0163] The specific design of Implementation 3 in this example is as follows:

[0164] Step (A) involves establishing a modular multilevel converter control unit and acquiring the converter MMC arm modulation signal, specifically...

[0165] The modular multilevel converter control unit includes a modular multilevel converter and a control unit. The control unit includes a voltage loop, a current loop, and a capacitor voltage equalization module. The control unit samples the AC voltage V. abc Alternating current i abc v is obtained after abc / dq coordinate transformation dq i dq Then v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dqWith the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s); Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc The upper and lower arm modulation signals m of the modular multilevel converter are obtained through level modulation and capacitor voltage equalization modules. lu m ld ; where m lu m ld The expression is shown below.

[0166] ;

[0167] Where M1 is the fundamental frequency modulation ratio, The fundamental angular frequency, The phase of the fundamental frequency component of the AC voltage. is the phase shift angle of the three phases, where l represents any one of the three phases;

[0168] The controlled source equivalent model of the converter is derived based on the bridge arm modulation signal of the converter in step (B), specifically as follows:

[0169] Based on the upper and lower arm modulation signals m of the modular multilevel converter lu m ld The controlled source equivalent model of the converter is derived, and the capacitors of all sub-modules connected in series in the bridge arm are represented by an equivalent capacitor C. eq To represent; in the controlled source equivalent model, the current flowing through the equivalent capacitance of the upper and lower bridge arms is controlled by the current source m. lu i lu m ld i ld Equivalent; the output voltage of the upper and lower bridge arms is controlled by the voltage source m lu v lu m ld v ld Equivalent capacitance C eq The expression is as follows:

[0170] Among them, C sm N represents the capacitance value of a single submodule in the bridge arm, and N is the total number of series-connected submodules in the bridge arm.

[0171] In step (C), the construction of the MMC bridge arm time-domain model based on the controlled source equivalence principle and the modulation signal is specifically as follows:

[0172] Based on the equivalent model of the controlled source of the modular multilevel converter, the time-domain model of the bridge arm is derived. Since the six bridge arms of the three phases of the modular multilevel converter satisfy symmetry, only the upper bridge arm of a single phase needs to be modeled. The specific time-domain expression of the upper bridge arm model of the modular multilevel converter is shown below.

[0173] ;

[0174] Among them, v lu_c The sum of the capacitor voltages of all submodules in the upper bridge arm, v lO' V is the AC port voltage. dc For DC side voltage, i lu R is the current in the upper arm of the bridge. arm L arm For the upper bridge arm, a resistor and an inductor are connected in series, v O'O This represents the neutral point potential difference between the AC and DC sides. Since the three-phase AC system is balanced, the sum of the three-phase AC voltages and currents is zero. The AC neutral point voltage is the sum of the zero-sequence components of the differential-mode voltages of the three-phase bridge arms. O'O The expression is as follows:

[0175] ;

[0176] in, This represents the zero-sequence component of the differential-mode voltage of a single-phase bridge arm.

[0177] Step (D) describes the use of a multi-harmonic linearization method to expand the internal state variables of the bridge arm into frequency-domain harmonic vectors, establishing an MMC frequency-domain small-signal model. Specifically...

[0178] Impedance modeling based on multi-harmonic linearization modulates the bridge arm modulation signal m of the internal state variable of the modular multilevel converter. lu Bridge arm current i lu Bridge arm equivalent capacitance voltage v lu_c Expanded in the frequency domain into the form of harmonic vectors m lu i lu v lu_c A frequency domain small-signal model of the modular multilevel converter is established based on the bridge arm time-domain model; the frequency sequence corresponding to the harmonic vector is [-nf1,…,-f1,0,f1,…,nf1]. T Where n is a positive integer, f1 is the fundamental frequency, and each element in the harmonic vector is the Fourier coefficient of the corresponding steady-state harmonic. The expression for the harmonic steady-state vector in the frequency domain is shown below.

[0179] ;

[0180] The expression for the small-signal model of the bridge arm is shown below.

[0181] ;

[0182] Here, "^" represents a small perturbation. After introducing the small signal, the center frequency corresponding to each small signal vector is the perturbation frequency f. p At this time, the perturbation frequency sequence corresponding to the small signal vector is [f p -nf1,…,f p -f1,f p ,f p +f1,…,f p +nf1] T Y Ceq Z L The expressions for the bridge arm equivalent capacitance admittance matrix and bridge arm impedance matrix at various frequencies under small-signal perturbation are shown below.

[0183] ;

[0184] , These are small-signal vectors representing the AC port voltage and the AC center point voltage. , , For i lu v lu m lu The corresponding small-signal vector in the frequency domain is expressed as follows.

[0185] ;

[0186] I lu V lu M lu For i lu v lu m lu The corresponding steady-state value matrix is ​​expressed as follows:

[0187] ;

[0188] Because of v O'O In steady state, the zero-sequence differential-mode component of the bridge arm voltage is canceled out, resulting in no zero-sequence current path at the AC port. This corresponds to setting the zero-sequence differential-mode element of the bridge arm impedance matrix in the mathematical model to zero. After introducing small-signal quantities, the phase sequence division of each harmonic is as follows, and the differential-mode component division expression is shown below.

[0189] ;

[0190] The common-mode component partitioning expression is shown below.

[0191] ;

[0192] For the Z in the bridge arm impedance matrixL The zero-order differential modulus component is corrected by setting it to zero to obtain Z'. L Then, in the small signal equation It can be directly ignored; the corrected Z' L The expression is as follows:

[0193] ;

[0194] The corrected expression for the small-signal equation in the frequency domain is shown below.

[0195] ;

[0196] In step (E), the AC side port impedance matrix of the converter is obtained based on the frequency domain small-signal model, specifically as follows:

[0197] The AC-side port impedance matrix of the modular multilevel converter is obtained based on the frequency domain small-signal model. Since the positive and negative sequence impedances of the sending-end converter station are symmetrical, only the positive sequence impedance needs to be analyzed here. The positive sequence AC side port impedance of the modular multilevel converter is shown in the following formula.

[0198] ;

[0199] Where E is the identity matrix, P represents the transfer function matrix from the small-signal vector of the upper arm current to the small-signal vector of the modulating signal, Q represents the transfer function matrix from the small-signal vector of the AC port voltage to the small-signal vector of the modulating signal, and the small-signal vector of the bridge arm modulating signal... The relationship between the AC port voltage and the small-signal vector of the bridge arm current is shown in the following equation.

[0200] ;

[0201] The transfer function matrices P and Q contain only non-zero sequence differential mode components, and different harmonic phase sequences correspond to different transfer functions. The specific expression for P is shown below.

[0202] ;

[0203] The specific expression for Q is shown below.

[0204] ;

[0205] Example 5:

[0206] This example further involves using impedance frequency band analysis in S2 to obtain the high-frequency impedance of the modular multilevel converter, specifically...

[0207] For high-frequency bands, the voltage feedforward branch plays a major role in the control loop, while the PI controllers in the voltage outer loop and current loop...v (s), G i The effect of (s) can be ignored. When f > 1000Hz, the expression for the high-frequency impedance of the AC side of the modular multilevel converter is as follows.

[0208] ;

[0209] In the formula, G d R is the system link delay transfer function. arm L is the bridge arm resistance. arm Let f be the inductance of the bridge arm, and f be the frequency variable.

[0210] Without considering system control delay, when f > 1000Hz, the AC side impedance of the converter is mainly determined by the bridge arm impedance Z. arm =0.5(R arm +sL arm Dominated by ) and exhibiting inductive characteristics; according to formula (18), when the system has a delay T d At that time, voltage feedforward will delay the transfer function G. d Introduced into the high-frequency impedance, the high-frequency impedance on the AC side of the converter is now equivalent to that of the bridge arm impedance Z. arm And the additional impedance Z introduced by voltage feedforward due to time delay Gd The parallel configuration yields the simplified high-frequency impedance expression for the MMC, as shown below.

[0211] ;

[0212] Delayed additional impedance Z Gd When connected in parallel with the bridge arm impedance, a sinusoidal quantity with angular frequency ω appears in the real part of the high-frequency admittance. The expression for the real part of the high-frequency admittance is shown below.

[0213] ;

[0214] In S2, the high-frequency impedance reshaping achieved by adding a voltage feedforward filter to the voltage feedforward branch in the control loop to suppress high-frequency oscillations caused by negative damping characteristics in the high-frequency band is specifically as follows:

[0215] To suppress the high-frequency negative damping characteristics, the delayed additional impedance Z is used. Gd From this perspective, a voltage feedforward filter is introduced into the voltage feedforward branch to reduce the effect of delay. Before determining the type of filter, it is expressed as a combination of real and imaginary parts, as shown in the following expression.

[0216] ;

[0217] Among them, G ffv For a voltage feedforward filter, R ffvis the real part of the voltage feed-forward filter, X ffv is the imaginary part of the voltage feed-forward filter;

[0218] After introducing the voltage feed-forward filter into the voltage feed-forward branch, taking the reciprocal of the delay additional impedance and separating the real part, the expression of the real part of the delay additional admittance is obtained as follows,

[0219] ;

[0220] According to the above formula, there is still a sinusoidal quantity with angular frequency ω in the real part of the delay additional admittance, and the amplitude of this sinusoidal quantity is ;

[0221] Select a second-order low-pass filter to make the amplitude of the sinusoidal quantity decay to zero in the high-frequency band and not affect the control dynamics of the current loop. The high-frequency negative damping characteristic of the converter will be better suppressed. The specific expression of the voltage feed-forward filter is as follows,

[0222] ;

[0223] where ξ is the filter damping coefficient 0.707, ω ffv [[ID=2,6]]is the filter cut-off angular frequency, f ffv is the filter cut-off frequency, ω ffv and the cut-off frequency f ffv The relationship between them is ω ffv = 2πf ffv ; Considering that the control dynamics of the current loop are not affected, the bandwidth ω of the voltage feed-forward filter ffv is at least 3 times the bandwidth ω of the current loop ci .

[0224] Example 6:

[0225] The further design of this example is to analyze the medium-frequency impedance of the modular multilevel converter obtained by impedance frequency band analysis in S3, specifically

[0226] In the medium-frequency band range of 100Hz < f < 1000Hz, since the control bandwidth of the voltage outer loop is only about ten-odd hertz, the effect of the voltage outer loop PI controller G v (s) can still be ignored, but the effect of the current loop in the control loop needs to be considered. When 100Hz < f < 1000Hz, the expression of the additional impedance brought by the current loop is as follows,

[0227] ;

[0228] where, is the current loop additional impedance, K pwm is the modulation gain;

[0229] Since the additional impedance of the current loop is equivalent to adding an additional loop impedance in series to the original impedance, when 100 Hz < f < 1000 Hz, the AC-side impedance of the converter is equivalent to being composed of the high-frequency impedance and the additional impedance introduced by the current loop connected in series, and the expression is as follows,

[0230] ;

[0231] In S3, the reshaping of the intermediate-frequency impedance by adding a filter to the current feedback branch in the control loop is specifically

[0232] Before determining the type of the filter, represent it in the form of the real part plus the imaginary part, and the expression is as follows,

[0233] ;

[0234] where, G vir is the current feedback filter, r vir is the real part of the current feedback filter, and x vir is the imaginary part of the current feedback filter;

[0235] After adding the filter to the current feedback branch, take the reciprocal of the additional impedance of the current loop and separate the real part to obtain the real part expression of the additional admittance of the current loop as follows,

[0236] ;

[0237] where, k ii is the integral coefficient of the current controller; K pwm is the modulation gain. According to the above formula, in the frequency range of 100 Hz < f < 1000 Hz, 3π / 50 < ωT d < 3π / 5. According to the design of the voltage feedforward low-pass filter G ffv , when 100 Hz < f < 1000 Hz, the conclusion of comparing some expressions in formula (27) with zero value can be obtained as follows,

[0238] ;

[0239] Therefore, in order to meet the condition that the real part of the additional admittance of the current loop is greater than zero, G vir needs to meet the following constraint conditions:

[0240] ;

[0241] [[ID=⑥1]]Considering the constraint conditions satisfied by the intermediate-frequency impedance reshaping, select the filter G virFor a second-order low-pass filter, the real part of the filter satisfies a condition of less than or equal to zero in the frequency range after the cutoff frequency, which meets the constraint condition. The imaginary part is less likely to satisfy the constraint condition of being greater than or equal to zero. However, after adding a second-order low-pass filter, the portion of the real part of the additional admittance in the current loop that is greater than zero is sufficient to cancel out the portion that is less than zero, making the overall real part of the admittance greater than zero. This characteristic becomes more pronounced as the cutoff frequency decreases. The specific expression for the second-order low-pass filter is shown below.

[0242] ;

[0243] Where ξ is 0.707, ω vir ω is the filter cutoff angular frequency. vir With cutoff frequency f vir The relationship between them is ω vir =2πf vir Although, according to the above constraints, a lower cutoff frequency for the low-pass filter can better suppress negative resistance characteristics, an excessively low cutoff frequency may cause the loop gain of the current loop to fail to meet the stability criterion, leading to system instability. Therefore, G... vir The phase lag introduced at the current loop crossing frequency should meet the following requirements:

[0244] ;

[0245] Where, ω ci Given the current loop crossing angular frequency, when the current loop bandwidth is selected to be 70Hz, the cutoff frequency of the second-order low-pass filter connected in series with the current feedback branch is selected to be 100Hz based on the above conditions.

[0246] Example 7:

[0247] This invention also provides a high-frequency oscillation suppression device for a flexible DC transmission system. The device includes a modular multilevel converter and a control unit. The control unit includes a voltage loop, a current loop, and a capacitor voltage equalization module.

[0248] The modeling and analysis module obtains the AC-side port impedance of the modular multilevel converter based on modeling and analysis.

[0249] The high-frequency impedance reshaping module uses impedance frequency segmentation analysis to obtain the high-frequency impedance of the modular multilevel converter, and reshapes the high-frequency impedance by adding a voltage feedforward filter to the voltage feedforward branch in the control loop to suppress high-frequency oscillations caused by the negative damping characteristics in the high-frequency band.

[0250] The intermediate frequency impedance reshaping module is used to reshape the high-frequency impedance, obtain the intermediate frequency impedance of the modular multilevel converter by impedance frequency segmentation analysis, and reshape the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop to suppress the intermediate frequency oscillation caused by the negative damping characteristics in the intermediate frequency band.

[0251] Test example:

[0252] To better illustrate the effectiveness of the present invention, a specific embodiment of the high-frequency oscillation suppression method in the flexible DC transmission system of the present invention is described below.

[0253] This example employs the method of connecting filters in series in the voltage feedforward branch and the current feedback branch, respectively, to suppress negative resistance in the high-frequency band of the AC port:

[0254] AC side voltage v abc v is obtained through coordinate transformation dq , will v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s) Obtain the loop current reference i dq_ref AC side current i abc i is obtained through coordinate transformation dq After passing through a second-order low-pass filter, it is compared with its current reference i dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s). After passing through a second-order low-pass filter, the voltage v dq Feedforward to the output of the current regulator. Modulation voltage v Mdq v is obtained through inverse coordinate transformation Mabc The bridge arm modulation signal is obtained after recent level modulation and submodule capacitor voltage equalization. Specific electrical parameters are shown in Table 1.

[0255] Table 1

[0256] like Figure 4 The diagram shown illustrates the system simulation frequency sweep verification of the AC port impedance of the converter station before and after the addition of the mid-to-high frequency oscillation suppression method according to the present invention. The main frequency sweep process is as follows: First, a series of disturbance sequences of different frequencies are injected into the AC side, with the amplitude of the current disturbance being one-tenth of the amplitude of the AC side current of the converter, and the phase sequence being positive; then, the AC side voltage response is sampled and measured; finally, the impedance frequency sweep values ​​at different frequency points are calculated.

[0257] like Figure 5As shown, the analytical values ​​and frequency sweep points of the AC port impedance characteristic curves of the converter station system before and after the application of the mid-to-high frequency oscillation suppression method in the voltage feedforward and current feedback branches are presented. Both the voltage loop and current loop employ PI regulators. The main circuit, voltage loop, current loop, and filter parameters are shown in Table 1. The bandwidth of the second-order low-pass filter in the voltage feedforward branch is 210Hz, and the bandwidth of the second-order low-pass filter in the current feedback branch is 100Hz. From... Figure 5 It can be seen that in the mid-to-high frequency range above 100Hz, compared with the original AC port impedance curve analytical value, after adding the mid-to-high frequency oscillation suppression strategy, the impedance phase is suppressed to [value missing]. The impedance analysis value matches the simulated frequency sweep point, indicating that the high-frequency negative resistance in the system is effectively suppressed after adopting the control method in this paper, thus satisfying the system stability condition.

[0258] In summary, the high-frequency oscillation suppression method in a flexible DC transmission system of the present invention first samples the AC voltage v. abc Alternating current i abc v is obtained through abc / dq coordinate transformation dq i dq Then v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dq With the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s). Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc After passing through the recent level modulation and capacitor voltage equalization module, the upper and lower arm modulation signals m of the modular multilevel converter are obtained. lu m ld Subsequently, based on the bridge arm modulation signal m of the modular multilevel converter... lu m ld The controlled source equivalent model of the converter is derived, and the capacitors of all sub-modules connected in series in the bridge arm are represented by an equivalent capacitor C. eq To represent; then, based on the equivalent model of the controlled source of the modular multilevel converter, the time-domain model of the bridge arm is derived; next, based on the impedance modeling method of multi-harmonic linearization, the bridge arm modulation signal m, an internal state variable of the modular multilevel converter, is used to represent... lu Bridge arm current i lu Bridge arm equivalent capacitance voltage v lu_c Expanded in the frequency domain into the form of harmonic vectors m lu i lu v lu_cA frequency domain small-signal model of the modular multilevel converter is established based on the bridge arm time-domain model; then, the AC-side port impedance matrix of the modular multilevel converter is obtained based on the frequency domain small-signal model. Then through the impedance matrix Frequency band analysis yields the simplified high-frequency impedance of the modular multilevel converter. A high-frequency oscillation suppression method is proposed based on high-frequency impedance characteristics: by adding a suitable filter G to the voltage feedforward branch in the control loop. ffv The high-frequency impedance is reshaped to suppress high-frequency oscillations caused by negative damping characteristics in the high-frequency range; finally, the impedance matrix is ​​used... Frequency band analysis yields the simplified intermediate frequency impedance of the modular multilevel converter. Based on the mid-frequency impedance characteristics, a mid-frequency oscillation suppression method is proposed: by adding a suitable filter G to the current feedback branch in the control loop. vir The intermediate frequency impedance is reshaped to suppress intermediate frequency oscillations caused by negative damping characteristics. This effectively achieves impedance reshaping and oscillation suppression in converter stations, and the parameter design process is relatively specific and accurate. It provides a foundation for stability analysis of flexible DC transmission systems and control optimization design of converter stations, which is of great significance for the safe and stable operation of wind farm-converter station systems. It has significant engineering application value and is suitable for widespread promotion and use.

[0259] The foregoing has shown and described the basic principles, main features, and advantages of the present invention. Those skilled in the art should understand that the present invention is not limited to the above embodiments. The embodiments and descriptions in the specification are merely illustrative of the principles of the invention. Various changes and modifications can be made to the invention without departing from its spirit and scope, and all such changes and modifications fall within the scope of the present invention as claimed. The scope of protection of the present invention is defined by the appended claims and their equivalents.

Claims

1. A method for suppressing high-frequency oscillations in a flexible DC transmission system, characterized in that, Includes the following steps S1. Obtain the AC side port impedance of the modular multilevel converter based on modeling analysis; S2 uses impedance frequency segmentation analysis to obtain the high-frequency impedance of the modular multilevel converter, and adds a voltage feedforward filter to the voltage feedforward branch in the control loop to reshape the high-frequency impedance and suppress the high-frequency oscillation caused by the negative damping characteristics in the high-frequency band. S3, after high-frequency impedance reshaping, uses impedance frequency segmentation analysis to obtain the intermediate frequency impedance of the modular multilevel converter, and reshapes the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop to suppress the intermediate frequency oscillation caused by the negative damping characteristics in the intermediate frequency band.

2. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 1, characterized in that, In S2, the high-frequency impedance reshaping achieved by adding a voltage feedforward filter to the voltage feedforward branch in the control loop, thereby suppressing high-frequency oscillations caused by negative damping characteristics in the high-frequency band, satisfies the following constraints: 1) The amplitude of the sinusoidal quantity in the real part of the delayed additional admittance obtained from the high-frequency impedance of the modular multilevel converter decays to zero in the high-frequency range, as expressed by: ； Among them, R ffv X is the real part of the voltage feedforward filter. ffv This represents the imaginary part of the voltage feedforward filter; 2) The bandwidth of the voltage feedforward filter is far from the bandwidth of the current loop, expressed as: ； Among them, W ffv W represents the bandwidth of the voltage feedforward filter. ci This represents the bandwidth of the current loop.

3. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 2, characterized in that, In S3, the reshaping of the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop satisfies the following constraints: 1) The real part of the additional admittance of the current loop obtained from the intermediate frequency impedance of the converter is positive in the intermediate frequency range, expressed as: ； Where, r vir x is the real part of the voltage feedforward filter. vir This represents the imaginary part of the voltage feedforward filter; 2) The phase lag introduced by the current feedback filter at the current loop crossover frequency does not exceed the phase margin of the current loop, expressed as: ； Where ξ is the damping coefficient of the current feedback filter, ω ci ω is the crossing angular frequency of the current loop. vir This is the cutoff angular frequency of the current feedback filter.

4. The high-frequency oscillation suppression method in the flexible DC transmission system according to any one of claims 1-3, characterized in that, In S1, obtaining the AC side port impedance of the modular multilevel converter based on modeling analysis includes the following steps: Step (A): Establish a modular multilevel converter control unit and acquire the converter MMC arm modulation signal; Step (B): The equivalent model of the controlled source of the converter is derived based on the bridge arm modulation signal of the converter. Step (C): Based on the controlled source equivalence principle, construct the MMC bridge arm time-domain model according to the modulation signal; Step (D): Based on the multi-harmonic linearization method, the internal state variables of the bridge arm are expanded into frequency domain harmonic vectors to establish the MMC frequency domain small-signal model. Step (E): Obtain the AC side port impedance matrix of the converter based on the frequency domain small-signal model.

5. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 4, characterized in that, Step (A) involves establishing a modular multilevel converter control unit and acquiring the converter MMC arm modulation signal, specifically... The modular multilevel converter control unit includes a modular multilevel converter and a control unit. The control unit includes a voltage loop, a current loop, and a capacitor voltage equalization module. The control unit samples the AC voltage V. abc Alternating current i abc v is obtained after abc / dq coordinate transformation dq i dq Then v dq With voltage reference v dq_ref The comparison is performed, and the error is fed into the voltage regulator G. v (s), then i dq With the current reference i output by the voltage regulator dq_ref The comparison is performed, and the error is fed into the current regulator G. i (s); Current regulator output modulation voltage v Mdq v is obtained through inverse transformation of dq / abc coordinates Mabc The upper and lower arm modulation signals m of the modular multilevel converter are obtained through level modulation and capacitor voltage equalization modules. lu m ld ; Where, m lu m ld The expression is as follows: ； Where M1 is the fundamental frequency modulation ratio, The fundamental angular frequency, The phase of the fundamental frequency component of the AC voltage. is the phase shift angle of the three phases, where l represents any one of the three phases; The controlled source equivalent model of the converter is derived based on the bridge arm modulation signal of the converter in step (B), specifically as follows: Based on the upper and lower arm modulation signals m of the modular multilevel converter lu m ld The controlled source equivalent model of the converter is derived, and the capacitors of all sub-modules connected in series in the bridge arm are represented by an equivalent capacitor C. eq To represent; in the controlled source equivalent model, the current flowing through the equivalent capacitance of the upper and lower bridge arms is controlled by the current source m. lu i lu m ld i ld Equivalent; the output voltage of the upper and lower bridge arms is controlled by the voltage source m lu v lu m ld v ld Equivalent capacitance C eq The expression is as follows: ； Among them, C sm N represents the capacitance value of a single submodule in the bridge arm, and N is the total number of series-connected submodules in the bridge arm. In step (C), the construction of the MMC bridge arm time-domain model based on the controlled source equivalence principle and the modulation signal is specifically as follows: Based on the equivalent model of the controlled source of the modular multilevel converter, the time-domain model of the bridge arm is derived. Since the six bridge arms of the three phases of the modular multilevel converter satisfy symmetry, only the upper bridge arm of a single phase needs to be modeled. The specific time-domain expression of the upper bridge arm model of the single phase of the modular multilevel converter is as follows. ； Among them, v lu_c The sum of the capacitor voltages of all submodules in the upper bridge arm, v lO' V is the AC port voltage. dc For DC side voltage, i lu R is the current in the upper arm of the bridge. arm L arm For the upper bridge arm, a resistor and an inductor are connected in series, v O'O This represents the neutral point potential difference between the AC and DC sides. Since the three-phase AC system is balanced, the sum of the three-phase AC voltages and currents is zero. The AC neutral point voltage is the sum of the zero-sequence components of the differential-mode voltages of the three-phase bridge arms. O'O The expression is as follows: ； in, This represents the zero-sequence component of the differential-mode voltage of a single-phase bridge arm. Step (D) describes the use of a multi-harmonic linearization method to expand the internal state variables of the bridge arm into frequency-domain harmonic vectors, establishing an MMC frequency-domain small-signal model. Specifically... Impedance modeling based on multi-harmonic linearization modulates the bridge arm modulation signal m of the internal state variable of the modular multilevel converter. lu Bridge arm current i lu Bridge arm equivalent capacitance voltage v lu_c Expanded in the frequency domain into the form of harmonic vectors m lu i lu v lu_c A frequency domain small-signal model of the modular multilevel converter is established based on the bridge arm time-domain model; the frequency sequence corresponding to the harmonic vector is [-nf1,…,-f1,0,f1,…,nf1]. T Where n is a positive integer, f1 is the fundamental frequency, and each element in the harmonic vector is the Fourier coefficient of the corresponding steady-state harmonic. The expression for the harmonic steady-state vector in the frequency domain is as follows. ； The expression for the small-signal model of the bridge arm is as follows: ； Here, "^" represents a small perturbation. After introducing the small signal, the center frequency of each small signal vector is the perturbation frequency f. p At this time, the perturbation frequency sequence corresponding to the small signal vector is [f p -nf1,…,f p -f1,f p ,f p +f1,…,f p +nf1] T Y Ceq Z L The equivalent capacitance admittance matrix and impedance matrix of the bridge arm at each frequency under small-signal perturbation are expressed as follows. ； in, , These are small-signal vectors representing the AC port voltage and the AC center point voltage. , , For i lu v lu m lu The corresponding small-signal vector in the frequency domain is expressed as follows: ； Among them, I lu V lu M lu For i lu v lu m lu The corresponding steady-state value matrix is ​​expressed as follows: ； Because of v O'O In steady state, the zero-sequence differential-mode component of the bridge arm voltage is canceled out, resulting in no zero-sequence current path at the AC port. This corresponds to setting the zero-sequence differential-mode element of the bridge arm impedance matrix in the mathematical model to zero. After introducing small-signal quantities, the phase sequence division of each harmonic is as follows, and the differential-mode component division expression is as follows. ； The common-mode component partitioning expression is as follows: ； For the Z in the bridge arm impedance matrix L The zero-order differential modulus component is corrected by setting it to zero to obtain Z'. L Then, in the small signal equation Negligible; Corrected Z' L The expression is as follows: ； The corrected expression for the small-signal equation in the frequency domain is as follows. ； In step (E), the AC side port impedance matrix of the converter is obtained based on the frequency domain small-signal model, specifically as follows: The AC-side port impedance matrix of the modular multilevel converter is obtained based on the frequency domain small-signal model. Since the positive and negative sequence impedances of the sending-end converter station are symmetrical, only the positive sequence impedance needs to be analyzed. The positive sequence AC side port impedance of the modular multilevel converter is shown in the following formula. ； Where E is the identity matrix, P represents the transfer function matrix from the small-signal vector of the upper arm current to the small-signal vector of the modulating signal, Q represents the transfer function matrix from the small-signal vector of the AC port voltage to the small-signal vector of the modulating signal, and the small-signal vector of the bridge arm modulating signal... The relationship between the small-signal vectors of the AC port voltage and the bridge arm current is shown in the following equation. ； The transfer function matrices P and Q contain only non-zero sequence differential mode components, and different harmonic phase sequences correspond to different transfer functions. The specific expression for P is as follows. ； The specific expression for Q is as follows. 。 6. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 5, characterized in that: In S2, the high-frequency impedance of the modular multilevel converter is obtained by impedance frequency segmentation analysis, specifically as follows: For high-frequency bands, the voltage feedforward branch plays a major role in the control loop, while the PI controllers in the voltage loop and current loop... v (s), G i Neglecting the effect of (s), when f > 1000Hz, the high-frequency impedance expression for the AC side of the modular multilevel converter is as follows. ； In the formula, G d R is the system link delay transfer function. arm L is the bridge arm resistance. arm The inductance is the bridge arm inductance, and f is the frequency variable; If system control delay is not considered, when f > 1000Hz, the AC side impedance of the converter is mainly composed of the bridge arm impedance Z. arm =0.5(R arm +sL arm Dominated by ) and exhibiting inductive characteristics; when the system has a delay T d At that time, voltage feedforward will delay the transfer function G. d Introduced into the high-frequency impedance, the high-frequency impedance on the AC side of the converter is now equivalent to that of the bridge arm impedance Z. arm And the additional impedance Z introduced by voltage feedforward due to time delay Gd Composed of parallel elements, the expression is as follows: ； Delayed additional impedance Z Gd When connected in parallel with the bridge arm impedance, a sinusoidal quantity with angular frequency ω appears in the real part of the high-frequency admittance. The expression for the real part of the high-frequency admittance is as follows. 。 7. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 6, characterized in that: In S2, the high-frequency impedance reshaping achieved by adding a voltage feedforward filter to the voltage feedforward branch in the control loop, thereby suppressing high-frequency oscillations caused by negative damping characteristics in the high-frequency band, specifically involves... To suppress the high-frequency negative damping characteristics, the delayed additional impedance Z is used. Gd From this perspective, a voltage feedforward filter is introduced into the voltage feedforward branch to reduce the effect of delay. Before determining the type of filter, it is expressed as a combination of real and imaginary parts, as shown in the following expression. ； Among them, G ffv For a voltage feedforward filter, R ffv X is the real part of the voltage feedforward filter. ffv This represents the imaginary part of the voltage feedforward filter; After introducing a voltage feedforward filter into the voltage feedforward branch, taking the reciprocal of the delay-added impedance and separating its real part, the real part expression of the delay-added admittance is obtained as follows. ； According to the above formula, the real part of the delayed additional admittance still contains a sinusoidal quantity with angular frequency ω, and the amplitude of this sinusoidal quantity is... ; To satisfy the constraint that the amplitude of the sinusoidal quantity attenuates to zero in the high-frequency range, a second-order low-pass filter is selected to suppress the high-frequency negative damping characteristics of the converter. The specific expression of the voltage feedforward filter is as follows. ； Where ξ is the filter damping coefficient, ω ffv f is the filter cutoff angular frequency. ffv ω is the filter cutoff frequency. ffv With cutoff frequency f ffv The relationship between them is ω ffv =2πf ffv .

8. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 5, characterized in that: In S3, the intermediate frequency impedance of the modular multilevel converter is obtained by impedance frequency segmentation analysis, specifically as follows: In the medium frequency range of 100Hz < f < 1000Hz, the effect of the voltage loop PI controller G v (s) is ignored, and only the effect of the current loop in the control loop is considered. When 100Hz < f < 1000Hz, the expression of the additional impedance brought by the current loop is as follows, ； in, Add impedance to the current loop, K pwm Modulation gain; The additional impedance of the current loop is equivalent to connecting an additional loop impedance in series with the original impedance. When 100Hz < f < 1000Hz, the AC side impedance of the converter is equivalent to being composed of a high-frequency impedance and the additional impedance introduced by the current loop connected in series, and the expression is as follows. 。 9. The high-frequency oscillation suppression method in the flexible DC transmission system according to claim 5, characterized in that: In S3, the reshaping of the intermediate frequency impedance by adding a filter to the current feedback branch in the control loop is specifically as follows: Before determining the type of filter, it is expressed as a combination of real and imaginary parts, as shown in the following expression. ； Among them, G vir For a current feedback filter, r vir x is the real part of the current feedback filter. vir This represents the imaginary part of the current feedback filter; After adding a current feedback filter to the current feedback branch, taking the reciprocal of the added impedance of the current loop and separating its real part, we obtain the real part expression of the added admittance of the current loop as follows. ； where k ii is the integral coefficient of the current controller; K pwm is the modulation gain; according to the above formula, in the frequency range of 100 Hz < f < 1000 Hz, 3π / 50 < ωT d < 3π / 5. According to the design of the voltage feedforward low-pass filter G ffv , when 100 Hz < f < 1000 Hz, the conclusion of comparing some expressions of the above formula with zero value can be obtained as follows ； Considering that the mid-frequency impedance reshaping satisfies the constraints, a current feedback filter G is selected. vir For a second-order low-pass filter, the real part of the filter is less than or equal to zero in the frequency range after the cutoff frequency, which meets the constraint condition. The specific expression of the second-order low-pass filter is as follows. ； Where ξ is the damping coefficient of the current feedback filter, ω vir ω is the cutoff angular frequency of the current feedback filter. vir With cutoff frequency f vir The relationship between them is ω vir =2πf vir .

10. A high-frequency oscillation suppression device in a flexible DC transmission system, comprising a modular multilevel converter and a control unit, wherein the control unit includes a voltage loop, a current loop, and a capacitor voltage equalization module, characterized in that, It also includes, The modeling and analysis module obtains the AC-side port impedance of the modular multilevel converter based on modeling and analysis. The high-frequency impedance reshaping module uses impedance frequency segmentation analysis to obtain the high-frequency impedance of the modular multilevel converter, and reshapes the high-frequency impedance by adding a voltage feedforward filter to the voltage feedforward branch in the control loop to suppress high-frequency oscillations caused by the negative damping characteristics in the high-frequency band. The intermediate frequency impedance reshaping module is used to reshape the high-frequency impedance, obtain the intermediate frequency impedance of the modular multilevel converter by impedance frequency segmentation analysis, and reshape the intermediate frequency impedance by adding a current feedback filter to the current feedback branch in the control loop to suppress the intermediate frequency oscillation caused by the negative damping characteristics in the intermediate frequency band.

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