Coherent lidar system for environmental perception with phase modulation consisting of two periodically nested sequences

Abstract

A coherent lidar system for environmental sensing, which: - emits a phase-modulated signal, the phase of which is generated by switching between discrete values, i.e., from a sequence of phase values, and the times of this phase switching form a subset of an equidistant time grid; - receives the signals reflected back from objects, which are delayed relative to the emitted signal by the distance-dependent propagation time and advanced in frequency by the relative velocity-dependent Doppler effect, and converts them into a low-frequency signal by mixing and digitizes them into a received sequence; and - from this received sequence, determines the variable dimensions of time shift and frequency shift of signals reflected from objects using digital signal processing equipment; - the phase modulation sequence comprises two periodically nested sequences.wherein the first sequence has constant or periodic phase values ​​and primarily serves to determine the frequency shift, while the second sequence alternates essentially irregularly between phase values ​​and serves to determine the time shift, characterized in that: - the phase modulation sequence comprises two periodically nested sequences, wherein this nesting has a period N1 with respect to the sampling time of the received sequence; - to determine the frequency shift of objects, N1 discrete Fourier transforms are calculated over values ​​of the received sequence from the periodic grid of the first sequence, the shift increasing by one grid value each time; - in each of these N1 discrete Fourier transforms, the respective frequencies of magnitude peaks that lie above a detection threshold are determined.- the received sequence is turned back in frequency in the periodic and correspondingly shifted grid of the second sequence for the respective frequencies and grid shifts, - a correlation is determined between the sequence thus generated and the second modulation sequence, and - from values ​​of this correlation and the respective grid shift, the respective time shift and thus object distance, as well as from the respective frequency, the radial relative velocity of the respective object, are determined.

Description

The invention relates to a coherently operating lidar system, in particular for detecting the environment for motor vehicle applications. The coherent lidar system operates with a phase modulation which, according to the invention, contains two sequences which are periodically nested one inside the other, one sequence serving primarily for determining the relative speed of objects and the other for determining their distance.Prior ArtMotor vehicles are increasingly equipped with driver assistance systems which, with the aid of sensor systems, record the environment and derive automatic reactions of the vehicle from the traffic situation detected therefrom and / or instruct, in particular warn, the driver. A distinction is made between comfort and safety functions.However, the developments now go in an even more extensive direction. The driver is no longer only assisted, but the task of the driver is increasingly being done autonomously by the vehicle, i.e. the driver is increasingly being replaced; one speaks of autonomous driving.In particular for autonomous driving, sensors with highly accurate and machine-simple evaluation of environmental information are required. Radar systems are limited in their angular accuracy and separation capability and, alone or in combination with camera systems, cannot yet meet these high detection requirements satisfactorily at least today. Therefore, lidar systems are also used in parallel which have a similarly high angular resolution (horizontal and vertical) as a camera, but additionally still provide distance information and separation capability in each pixel. Nowadays, what are known as time-of-flight lidar systems are mostly used, which treat electromagnetic radiation in the sense of particles and can thus measure only the distance, but not the relative speed, directly. However, it is now increasingly also possible to focus coherently operating lidar systems which treat electromagnetic radiation in the sense of waves (such as radar systems) and thus can also measure the relative speed of objects directly via the Doppler effect. Further advantages of coherent lidar systems are that they are robust on the one hand to interference radiation from other sources (e.g. from other lidar systems or sunlight) and on the other hand have a higher sensitivity at higher distances and thus allow higher ranges. And, coherent lidar systems are also attributed a higher potential of high semiconductor integration, which promising lower manufacturing costs.In coherent lidar systems, the emitted electromagnetic wave is modulated, i.e., it changes over time in at least one of the parameters amplitude, frequency, or phase-otherwise, no distance measurement would be possible. The modulation that is most frequently used in coherent lidar systems is linear frequency modulation (FMCW=frequency-modulated continuous wave), which usually consists of two frequency ramps whose slopes have opposite signs. However, this modulation has ambiguity problems, in particular in the case of a plurality of reflections in the same beam direction, and the generation of a highly linear frequency change is also complicated. These disadvantages do not occur or occur to a lesser extent in the case of phase modulation (e.g. with pseudo-random changes over discrete phase values), however, the digital evaluation of the received signals is more complicated and the approaches proposed in the prior art are associated with disadvantages, in particular with regard to sensitivity and thus range.DE 10 2020 215 039 A1 discloses a coherently operating LiDAR sensor system having a transmission unit comprising a laser source, a phase modulator for modulating a phase of the light from the laser source and a transmission optical unit for transmitting the light modulated by the phase modulator into the environment of the LiDAR sensor system. The transmission unit is configured to emit a plurality of transmission sequences of the light, wherein each transmission sequence has a first section and a second section, wherein the first section is an unmodulated, in particular phase-constant, signal, and the second section is a signal phase-modulated by the phase modulator. Furthermore, an evaluation unit is provided which uses the first section to determine at least an amount of a Doppler frequency and uses the Doppler frequency and the second section to determine a distance to an object in the environment of the LiDAR sensor system.In addition, EP 3 719 537 A1, DE 10 2018 131 580 A1, DE 10 2021 111 820 A1, DE 10 2020 215 041 A1 and EP 2 626 722 A1 also describe lidar systems of the generic type for distance measurement.Object, Solution and Advantages of the InventionIt is an object of the invention to provide a solution for a coherent lidar system with phase modulation that comes at least close to the maximum possible detection sensitivity, accuracy and separation capability without requiring an excessively high digital computing effort.This object is achieved in principle by a lidar system according to claim 1. Practical embodiments of the invention are claimed in the dependent claims. The core idea here is that the phase modulation sequence contains two sequences which are periodically nested one inside the other, one sequence serving primarily for determining the relative speed of objects and the other for determining their distance.The advantages of the invention result from the fact that a lidar system with high performance and low price can be realized.In the coherently operating lidar system for detecting the environment according to the invention, a phase-modulated signal is emitted, wherein the phase of this signal is generated by switching between discrete values, i.e. from a sequence of phase values, and the times of this phase switching form a subset of an equidistant time grid. Furthermore, the signals reflected back from objects, which are delayed with respect to the transmitted signal by the distance-dependent propagation time and are frequency-biased by the relative speed-dependent Doppler effect, are received and converted by mixing into a low-frequency signal and digitized into a reception sequence, from which reception sequence the variable dimensions time shift and frequency shift of signals reflected from objects being determined in digital signal processing means. The phase modulation sequence contains two sequences which are periodically nested one inside the other, wherein the first sequence has constant or periodic phase values and serves primarily for determining the frequency shift, while the second sequence changes substantially irregularly between phase values and serves for determining the time shift.According to a preferred embodiment of the coherently operating lidar system, the first sequence can alternate between two phase values different by approximately 180°.Furthermore, the second sequence can either change pseudo-randomly between discrete phase values or consist of a code whose autocorrelated low side lobes.Preferably, a binary phase modulation, i.e. consisting of only two phase values differing by approximately 180°, is used.Expediently, the two sequences can be sequences interleaved alternately, i.e. with period two in relation to the modulation rate.According to a special embodiment variant, a phase modulation sequence consisting of two sequences periodically nested one inside the other can also be repeated periodically, wherein, in particular in the case of continuous scanning of the laser beam, the cyclic property of the modulation and reception sequence is utilized and detection in respectively different directions is realized.Preferably, the laser beam scans continuously and for the successive detection directions, overlapping sections of a long, optionally periodic phase modulation sequence, which contains two sequences periodically nested one inside the other, are used.Furthermore, the lidar system according to the invention also includes a phase modulation sequence having two sequences which are periodically nested one inside the other, in which the first sequence has constant or periodic phase values and serves primarily for determining the frequency shift, while the second sequence changes substantially irregularly between phase values and serves for determining the time shift. The interleaving of the sequences has the period N1 with respect to the sampling time of the reception sequence. To determine the frequency shift of objects N1, a discrete Fourier transform is calculated, preferably with the aid of fast Fourier transforms, over values of the reception sequence from the periodic grid of the first sequence, optionally interrupted by zeros or extended by zeros, and from N1-1 displacements of this grid, wherein the displacement increases by one grid value in each case. In each of these N1 discrete Fourier transforms, the respective frequencies of magnitude peaks which lie above a detection threshold are determined, and the reception sequence in the periodic and correspondingly shifted raster of the second sequence is rotated back in frequency with respect to the respective frequencies and the respective raster shifts, wherein a correlation is determined in each case between the sequence thus generated and the second modulation sequence, and the respective time shift and thus object distance are determined from values of this correlation, in particular of magnitude peaks lying above a detection threshold, and the respective raster shift, and the radial relative speed of the respective object are determined from the respective frequency, wherein any inconsistencies in frequency are solved with the aid of the phase relationship between values of the discrete Fourier transform and the correlation.Expediently, a first detection threshold can be used to determine the magnitude peaks of the discrete Fourier transforms, the complex value of the discrete Fourier transforms and the respective complex value of the associated correlation at their magnitude peaks are added up several times, if necessary, while compensating for the possible phase shifts, and these sums are checked to a second detection threshold, wherein the first detection threshold is less high above the noise.Brief Description of the DrawingsFIG. 1 shows the coherent lidar system with binary phase modulation. FIG. 2 shows the course of a pseudo-random binary phase modulation.In FIG. 3, the real part of the low-frequency analog received signal and the associated samples for an object are shown. FIG. 4 shows the magnitude of the two-dimensional correlation of the reception sequence for two objects. Fig. 5 shows a modulation sequence according to the prior art, which is sequentially composed of two sub-sequences.FIG. 6 shows a modulation sequence according to the invention with two sub-sequences periodically nested one inside the other. FIG. 7 shows the two FFTs for the first modulation subsequence for the example of two objects. FIG. 8 shows the two correlations to the second modulation subsequence for the example of two objects.FIG. 9 shows an alternative modulation sequence according to the invention with two sub-sequences periodically nested one inside the other.Exemplary EmbodimentsFIG. 1 schematically shows a coherent lidar system 1.1. A coherent signal in the wavelength range of approximately λ=155 nm is generated with the laser source 1.2; the coherence length is at least several microseconds and the frequency is constant. The signal then reaches a switchable inverter 1.3, with which the sign of the signal can be changed, which corresponds to a phase shift of 180°. Sign changes take place only in a fixed grid of e.g. 6.67 ns and according to a prior art are pseudo-random, i.e. the sign is changed only with a frequency or probability of 50% after T m= 6.67 ns. FIG. 2 shows a profile of this modulation sequence b(n), which consists of the values +1 and -1 and is also referred to as binary; it repeats itself with the period N=496, i.e. every 27.3 μs. The modulated signal passes through an amplifier 1.4, a circulator 1.5 (i.e. a transceiver switch), is emitted via a transceiver unit 1.6 and partially reflected back from an object 1.7; with a delay which is dependent on the object distance d and is thus variable, where c=3.10 8 m / s is the speed of light, and with a frequency shift which is dependent on the radial relative speed v and is thus variable and is generated by the Doppler effect, this signal is then picked up by the transceiver unit 1.6 and conducted via the circulator 1.5 into the further reception path. In a complex-valued mixer 1.8, the modulated received signal is superimposed with the unmodulated laser signal and converted into a complex-valued low-frequency signal with the aid of the photodiode unit 1.9; the frequency of this signal corresponds to the Doppler shift, the modulation of this signal is delayed by the signal delay time with respect to that of the transmission signal. FIG. 3 shows the real part of this low-frequency analog received signal e a( t) in the case of an object. Subsequently, the signal is sampled and digitized in an analog-to-digital converter unit 1.10 with the sampling frequency f s= 150 MHz, i.e. all T s= 6.67 ns-the resulting values of the real part are marked as dots in FIG. 3. The complex-valued sampling signal e(n), also referred to as a reception sequence below, can be described as follows: wherein it is assumed here that the delay time t 0 is an integer multiple m 0 of the modulation time T m= 6.67 ns: and that also corresponds to the Doppler shift is integer; a is the complex-valued amplitude of the reception sequence, "exp" denotes the exponential function, and ĵ is the imaginary unit.The complex-valued reception sequence e(n) according to (2) relates to a single object without longitudinal extension and to an ideal receiver. In fact, there may be a plurality of and / or extended objects and an additional noise r(n) is generated in the receiver, in particular by thermal noise; the result is then the reception sequence, wherein "sum i=1,...,l" represents the sum function via the index i=1,...,l of the I non-extended individual objects.From the reception sequence e(n) of the time period n=0, 1,..., N-1, the discrete runtimes m 0,i and the discrete Doppler shifts k 0,i of the I objects can be determined. For the most accurate possible determination, i.e. the best possible separation of signal and noise and thus the maximum sensitivity and range of the lidar system, what is known as optimum filtering is to be used, i.e. the filtering by correlation between the reception sequence e(n) and the two-dimensional space ξ m,k( n) of the possible ideal amplitude-normalized reception sequences of an individual object: M- 1 corresponds to the largest object distance to be assumed or of interest and it is assumed for the Doppler shift k that it can assume all values. This results in the two-dimensional correlation E m,k where "conj" denotes conjugate complex formation and the modulation sequence b(n) does not change because of its relative significance. The correlation E m,k has magnitude peaks (often also referred to as power peaks) at the positions (m,k)=(m 0,i, k 0,i) of objects; FIG. 4 shows the magnitude of the two-dimensional correlation for two objects of the same reception amplitude at (m 0,1, k 0,1) = (150, 500) and (m 0,2, k 0,2) = (51, 3000). In a further signal processing step, these magnitude peaks are determined and subsequently the distance of the objects is determined from their values m 0,i and their radial relative speed is determined from the k 0,i and therefore the distance and relative speed of a plurality of objects can be determined directly and unambiguously from a modulation sequence. This is a great advantage for linear frequency modulation frequently used in coherent lidar systems with two frequency ramps whose slopes have opposite signs--there may be inconsistencies in a single object, and in the case of several objects, these are unavoidable.The calculation of this two-dimensional correlation and its downstream evaluation take place in the digital signal processing unit 1.11. It represents a high outlay with the order N·M·N. However, the above relationship (5) can also be considered as a discrete Fourier transform over the product e(n)·b(n-m), n=0..., N-1 to be determined for each m=0..., M-1; if the discrete Fourier transform (DFT) is calculated via the fast Fourier transform (FFT): where k=0..., N-1 is the output dimension of the FFT, i.e. the discrete frequency, the calculation amount is reduced to the order M·N·log 2( N).The hitherto considered reception sequence e(n) of the time period n=0, 1,..., N-1 and the associated correlation E m,k relate to a single detection direction, that is to say to a pixel with respect to the horizontal and vertical directions. In fact, in each detection cycle of 50 ms, approximately 160,000 detection directions, i.e. pixels, are covered; this is typically realized by a combination of parallel transmitter and receiver, i.e. parallel detection of pixels, and scanning, i.e. sequential detection of pixels. Parallel transmitter and receiver means that there are all elements 1.4-1.10 in FIG. 1 a number of times, e.g. 64 times. The scanning can be effected, for example, by sequential switching or by continuous mechanical movement (for example of a mirror); in the case of continuous scanning, it is also possible for pixels to partially overlap, that is to say that the rear of the N values of the reception sequence e(n) of a pixel are also used as front values of the next pixel. Now, assuming that M=250 (corresponding to 249 m maximum distance in the above configuration), the FFT of length 4096 is ex. (6) 800 million times per second. A realization of these many FFT calculations is not possible via microprocessors or DSPs (Digital Signal Processors); the clock of such processors is typically in the range of 1 GHz, i.e. it would be necessary to calculate almost one complete FFT of the length 4096 per clock, but modern processors can generally also be able to calculate only up to 100 multiplications and additions per clock by orders of magnitude, which is below the requirement for an FFT of the length 4096 by orders of magnitude, even when parallel vectorial arithmetic units are used. Therefore, these many FFTs can be implemented only by a special purpose arithmetic logic implemented in hardware; however, this arithmetic logic is also generally associated with a large implementation outlay (large chip area) and high electrical power consumption.In the article "Phase-Coded-Based Modulation for Coherent Lidar" by Sebastian Banzhaf and Christian Waldschmidt, published in IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 70, NO. 10, OCTOBER 2021, an approach is proposed that makes do with less computational effort. For this purpose, the binary modulation sequence b(n) of the length N is composed of two parts: during the first subsequence b 1( n) of the length N 1 the phase is constant, that is to say for example and the second subsequence b 2( n) of the length N 2= N-N 1 is irregular, for example pseudo-random:The lengths N 1 and N 2 can be the same, i.e. and the modulation sequence can be repeated periodically for a plurality of detection directions, i.e. pixels; in FIG. 5, such a modulation sequence is illustrated, wherein the sequence composed of the subsequences is repeated with the period N=4096.The reception sub-sequence e 1,i( n) generated by an object i to the first, constant modulation sub-sequence b 1( n) applies according to Bez. (2a): i.e. it carries only the Doppler frequency of the respective object, which can be determined via a DFT or FFT. However, because of the unknown time shifts m 0,i one does not know the respective exact time position of e 1,i( n) within the reception sequence e(n); for example, the time shift can assume zero for the sake of simplicity, i.e. the FFT is formed via the reception sub-sequenceHowever, the assumption of the time shift zero leads to the fact that for other actual time shifts m 0,i, i.e. in particular for far-away targets, in the first m 0,i-1 values of the reception subsequence ẽ 1( n), no values are present from the constant modulation subsequence b 1( n), but back values of the modulation subsequence b 2( n) of the preceding period; the height of the respective magnitude peak of the FFT and thus its distance from the noise are thus reduced, so that the sensitivity is reduced. The magnitude peaks of the FFT 1,k of the reception subsequence ẽ 1( n) are checked for a detection threshold. The frequencies k 0,j of the J value peaks lying above the detection threshold are used for the further processing; typically, this value peak is seen in two adjacent FFT values (since they are not-as previously considered-at an integer Doppler index k 0 ), so that by interpolation it is then possible to determine its exact position, i.e. a non-integer frequency k 0,j. These frequencies k 0,j correspond at least approximately to the Doppler frequencies k 0,i of the objects or a subset thereof (for objects with very low reflectivity, they may not lead to a magnitude peak above the detection threshold).The second modulation subsequence b 2( n) leads in the reception sequence e(n) according to Bez. (2a) to components time-shifted by m 0,i and multiplied, i.e. modulated, by the respective Doppler frequency k 0,iIn order to eliminate the respective modulation by the Doppler frequency, the reception sequence e(n) is rotated back by the frequency k 0,j in each case: in this case, the region n=N / 2,..., 'N-1+M-1 of the reception sequence e(n) in which the reception of the second modulation sub-sequence b 2( n) can lie must be considered, wherein M-1 corresponds to the largest object distance to be assumed or of interest. It should be noted that the range n=N,..., N-1+M-1of the received signal can only be used in continuous scanning, but not in switched scanning (then the signal there is not coherent because of another direction of detection).The sequences ẽ 2,j( n) modified in this way contain modulation sub-sequences ai·b 2( n-m 0,i), shifted in the corresponding index j, contributions of objects with other Doppler frequencies k 0,i( i.e. k 0,i ≠k 0,j) represent a modulation sequence uncorrelated to b 2( n), since they are still modulated with the difference frequency k 0,i- k 0,j. Therefore, the sequences ẽ 2,j( n) can now be correlated with the second modulation subsequence b 2( n):In these one-dimensional correlations Ẽ 2,j,m magnitude peaks occur at the points m=m 0,i, i.e. at the discrete runtimes of objects; in the general case of a non-integer discrete runtime m 0,i the magnitude peak extends over two adjacent values of m in the case of suitable measures (e.g. grinding the modulation pulses to an approximately triangular shape) and its non-integer position can be determined by interpolation. The index j of the correlation Δẽ 2,j,m, at which the magnitude peak occurs, and the Doppler frequency k 0,j associated with this j result in the Doppler frequency k 0,i= k 0,j of the object. Thus, from magnitude peaks of the correlations Ẽ 2,j,m lying above a detection threshold, the distances and radial relative speeds of objects in the respective detection direction can be determined. It should be noted that contributions of objects with respectively different Doppler frequencies k 0,i( i.e. k 0,i ≠ k 0,j) due to their uncorrelatedness with b 2( n), only lead to noise in the respective correlations, i.e. to no magnitude peaks above a detection threshold; in the case of reflection signals of objects of approximately the same magnitude, this noise lies clearly below the magnitude peaks of interest and thus does not obscure them-only if reflection signals of objects are greatly different-objects with a strong reflection signal can obscure weak reflection signals with a different Doppler frequency via their noise.With the modulation sequence according to FIG. 5 and the evaluation explained above, in the typical case of an object in a pixel, an FFT is required for the reception sub-sequence e 1( n) to form the first modulation sub-sequence b 1( n) and a correlation between the second modulation sub-sequence b 2( n) and the frequency-rotated reception sub-sequence ẽ 2,1( n) to form the second modulation sub-sequence. This correlation in the time domain can also be realized by multiplying the FFTs in the frequency domain with a subsequent inverse FFT (inverse FFT requires the same computing effort as FFT itself); the FFT of the modulation subsequence b 2( n) can be determined once a priori, so that only the FFT has to be determined by the reception subsequence ẽ 2,1( n). Overall, therefore, the outlay of three FFTs is incurred, while in the two-dimensional correlation E, m,k according to Bez. (6) a total of M=250 FFTs, wherein these FFTs have approximately twice the length if the same duration of a pixel is assumed. This results in a very great reduction of the required computing effort to a magnitude which opens up modern DSPs with parallel vectorial computing units.Disadvantageous in comparison with a modulation sequence according to FIG. 2 and evaluation via a two-dimensional correlation E m,k according to Bez. (5) are in particular the following points:the sensitivity is somewhat more than 3 dB lower if the same duration of a pixel is assumed; Half the length of the FFT (from the reception sub-sequence e 1( n)) as well as half the length of the temporal correlation (from ẽ 2,j( n) to the modulation sub-sequence b 2( n)) results in 3 dB loss, and in addition, as explained above, there is the effect that, in particular for far-distant targets, the first values of the reception sub-sequence ẽ 1( n) do not originate from the constant modulation sub-sequence b 1( n) but from rear values of the modulation sub-sequence b 2( n) of the preceding period, so that these values effectively do not contribute to the respective magnitude peak of the FFT.Because of the half length of the FFT, resolution and accuracy are poorer for the Doppler determination, i.e. the determination of the radial relative velocity by a factor of two.The temporal correlation of ẽ 2,j( n) with b 2( n) for determining the distance is only half the length. Thus, the dynamic range of more than one object per pixel is reduced by 3dB, i.e. an object with a strong reflection signal increases the noise level by 3dB relative to the level of an object with a weaker reflection signal, so that the probability of not detecting such a second object is higher.In continuously scanning systems, pixels can be allowed to overlap in principle, i.e. some of the reception values e(n) are used for two adjacent pixels, in particular in order to achieve a longer data recording time per pixel and thus a better sensitivity. In a modulation sequence according to FIG. 5, however, there is only the possibility of an overlap of 50%, since both partial modulation sequences are required in each pixel. Smaller overlaps, which are frequently preferred, are not possible.These disadvantages of the modulation sequence according to FIG. 5 and the associated evaluation explained above, which are prior art, can be overcome to a significant extent by the modulation sequence according to the invention according to FIG. 6, without the computing power required for the evaluation significantly increasing. In the modulation sequence b(n) according to FIG. 6, the two previous modulation subsequences b 1( n) and b 2( n) are alternately nested one inside the other:Thus, both modulation sub-sequences each extend over the full modulation duration, wherein they each only assume every second raster value.The reception sub-sequence e 1,i( n) generated by an object i to the first, constant modulation sub-sequence b 1( n) can be either at even-numbered or odd-numbered values of n, depending on whether the discrete delay time m 0,i of the respective object is even-numbered or odd-numbered (here it is first assumed that m 0,i is integer): in this region from the index n, the first values originate from the end of the previous modulation period (because of the delay time of the respective object) - in contrast to the previously considered modulation sequence according to FIG. 5, however, these values also originate from the first, constant modulation sequence, so that they also contribute coherently to the later FFT. Because of the two possible positions of the reception sub-sequences e 1,i( n), the FFT for determining the Doppler frequencies of the objects must now be calculated twice-once over the first sequence and once over the second sequenceThe two FFTs relate to the output dimension, that is to say the discrete frequency k=0,..., N / 2-1, which is associated with the discrete frequency k considered hitherto on account of the effectively half sampling rate, wherein "mod(·,N)" represents the modulo function for the module N. In FIG. 7, the two FFTs are according to Bez. (17) is shown in magnitude for two objects of the same reception amplitude at (m 0,1, k 0,1) = (150, 500) and (m 0,2, k 0,2) = (51, 3000) ; the first object at the even-numbered discrete runtime m 0,1= 150 forms a magnitude seat in the first FFT 1,1,k at the Doppler frequency h 0,1= k 0,1= 500, the second object at the odd-numbered discrete runtime m 0,2= 51 forms a magnitude seat in the second FFT 1,2,k at the Doppler frequency k 0,2= k 0,2- N / 2=952.The magnitude peaks of the two FFTs Ẽ 1,1,k and Ẽ 1,2,k are checked for a detection threshold. The frequencies k 0,j of the J value peaks lying above the detection threshold are used for the further processing; typically, this value peak is seen in two adjacent FFT values (since they do not lie at an integer Doppler index k 0,j as viewed in the above example), so that their exact position, i.e. a non-integer frequency k 0,j can then be determined by interpolation. Apart from a possibly missing proportion N / 2 (because of the modulo function according to Bez. (18)), these frequencies k 0,j correspond at least approximately to the Doppler frequencies k 0,i of the objects or a subset thereof (for objects with very low reflectivity, they may not lead to a magnitude peak above the detection threshold). For the further processing, it must also be taken into account in which of the two FFTs the magnitude peak at k 0,j was detected, i.e. whether the associated discrete runtime is even or odd; for this purpose, the variable is introduced, where m is j= 0 if the magnitude peak occurs in first FFT Ẽ 1,1,k and m is j= 1 if the magnitude peak occurs in second FFT Ẽ 1,2,k.The second modulation subsequence b 2( n) according to Bez. (14b) leads in the reception sequence e(n) to components which are time-shifted by m 0,i, and multiplied, that is to say modulated, by the respective Doppler frequency k 0,iIn order to eliminate the respective modulation by the Doppler frequency, the reception sequence e(n) is rotated back by the frequency k 0,j: this takes into account the frequency in Bez. (19) Variable m j, whether the respective reception sub-sequence lies in an odd-numbered or even-numbered raster (i.e. has an even-numbered or odd-numbered discrete delay); and it should be noted that-as in the determination of the FFT-in this region of index n the first reception values originate from the end of the previous modulation period (because of the delay of the respective object), but this does not violate the coherence for the subsequent correlation because of the periodicity of the entire modulation sequence (has period N).The sequences ẽ 2,j( n) modified in this way contain, in the corresponding index j, cyclically shifted modulation sub-sequences a i ·b 2( mod(n-m 0,i, N)). Contributions of objects with other Doppler frequencies k 0,i( i.e. k 0,i ≠ k 0,j) represent a modulation sequence uncorrelated to b 2( n), since they are still modulated with the difference frequency k 0,i- k 0,j; the same applies to contributions with other m j, since these are modulated with b 1( n). Therefore, the sequences ẽ 2,j( n) can now be correlated cyclically with the second modulation sub-sequence b 2( n): in these one-dimensional correlations, magnitude peaks occur at the locations, i.e. at the discrete runtimes m 0,i of objects minus the in Bez. (21) applied displacement m j, so that the discrete transit time is obtained. For the above example with two objects, the amounts of the two correlations to ẽ 2,1( n) with k 0,1= 150 and m 1= 0 and to ẽ 2,2( n) with k 0,2= 952 and m 2= 1 are shown in FIG. 8; the amount peaks lie at the values and at the discrete runtimes and correspond.From the index j of the correlation at which the magnitude peak occurs and the Doppler frequency k 0,j associated with this j, the Doppler frequency k 0,i of the object results only except for a potential offset of N / 2-see Bez. (18), which takes into account that the Doppler frequency is determined from a sequence with half the sampling rate, i.e. an additional half period related to the full sampling rate cannot be detected therewith. However, such a half period leads to the complex values at the respective magnitude peaks in FFT 1,1,k or FFT, respectively. The phase of the phase difference between the two signals is rotated by 180° and the correlation with respect to one another, while in the other case they (i.e. no additional half period in Doppler frequency) have the same phase. With this connection, the ambiguity with respect to the respective Doppler frequency k 0,i can be solved by adding up the two complex values corrected by the possible phase shifts (0° and 180°), i.e. forming the sum and difference of the two complex values-if the sum has a greater magnitude than the difference, then the Doppler frequency k is 0,i= k 0,j, in the other case k is 0,i= h 0,j+ N / 2; In the above example, for the first object (i.e., for k=k, 0,1 and the sum is greater than the difference (due to approximately inphase between the two components), so that the Doppler frequency k is 0,1= k is 0,1= 500 and for the second object (i.e., for k=k, 0,2 and the difference is greater than the sum (due to approximately inphase between the two components), so that the Doppler frequency k is 0,2= k 0,2+ N / 2=3000. Because the Doppler frequency is now determined over the full duration of a modulation period, there is no longer the disadvantage of the modulation sequence of Figure 5 (according to the prior art) where resolution and accuracy for the Doppler determination are reduced by a factor of two, since the FFT is determined only over half the modulation duration.With the relationships and procedures presented above, the distances and radial relative speeds of objects in the respective detection direction can be determined from the magnitude peaks of the correlations lying above a detection threshold. It should be noted that contributions of objects with respectively different Doppler frequencies k 0,i( i.e. k 0,i ≠ k 0,j) and / or another value m j lead, because of their uncorrelatedness with b 2( n), only to noise in the respective correlations, i.e. to no magnitude peaks above a detection threshold; with reflection signals of objects of approximately the same strength, this noise lies clearly below the magnitude peaks of interest and thus not conceals them-only if reflection signals of objects are greatly different-objects with a strong reflection signal can conceal weak reflection signals with a different Doppler frequency via their noise.As an alternative to the approach that detections are generated from magnitude peaks of the correlations lying above a detection threshold, the sums and differences of FFT 1,1,k or FFT can be introduced above. Check 1,2,k with k=k 0,j and correlation for a detection threshold, since that value of sum and difference at which the two components FFT and correlation are added up in the correct phase, i.e. coherently, has a signal-to-noise ratio which is better by 3 dB than the correlation itself; this results from the fact that the sum or difference is integrated over the entire modulation sequence b(n), but only over half of the correlation. The sum or difference therefore also has the same signal-to-noise ratio and thus the same sensitivity as the optimum two-dimensional correlation E m,k according to Bez. (5). However, this applies only if a magnitude peak has already been detected in the FFT at the respective corresponding point, i.e. at k=k 0,j, ; since the FFT has a 3 dB poorer signal-to-noise ratio because of half the integration duration, a correspondingly reduced detection threshold must be applied. This reduced detection threshold has an increased probability of false detection of noise peaks; however, these are then discarded again at the effectively sharper detection threshold of the sum and difference of FFT and correlation, so that a slightly increased computing effort remains as the only disadvantage (because correlation has to be calculated more frequently). It should be noted that the sum and difference need not be determined for each m, but that the correlation can also be checked only once for a detection threshold reduced by approximately 3 dB and sum and difference are formed only at the locations where this detection threshold is exceeded. Because the inventive approach from the modulation sequence according to FIG. 6 and the evaluation described above has the same sensitivity as the optimum two-dimensional correlation E m,k according to Bez. (5), the disadvantage of the modulation sequence according to FIG. 5 (prior art) of a 3 dB poorer sensitivity is eliminated. It will now be discussed briefly whether the approach according to the invention comprising the sum and difference of FFT and correlation could not also be applied to the modulation sequence according to FIG. 5 : this is only the case if the reception phase is stably linear over the entire modulation sequence (that is to say instantaneous frequency constant), which is only conditionally present, in particular, because of speckle effects.The sum and difference approach of FFT and correlation also avoids the disadvantage of the modulation sequence according to FIG. 5 (prior art) of a dynamic range reduced by 3 dB for more than one object per pixel, because the full modulation sequence is now integrated.A further advantage of the modulation sequence according to the invention as shown in FIG. 6 is that any overlap can be realized for two adjacent pixels, since each section of this periodic modulation sequence (period N) has the characteristic property according to Bez. (14).The required computing effort for the approach according to the invention with the modulation sequence according to FIG. 6 differs from that for the modulation sequence according to FIG. 5 (prior art) by an additional FFT of length N / 2 (independent of the number of objects in the respective pixel), whereas the correlations have only half a length, which, however, does not represent an advantage in the realization of the correlation in the frequency range (i.e. via FFT and inverse FFT). In the typical case of an object in a pixel, two FFTs and a correlation are to be determined. In the above-explained approach with a reduced detection threshold for the FFTs, in principle additional calculations of the correlation can also occur. The required computing effort is thus still of a magnitude which opens up modern DSPs with parallel vectorial computing units, which leads to a favourable realization.For the output dimension of the FFT, i.e. the discrete frequency, the non-symmetrically located region k=0,..., N / 2-1 was considered above, as is generally the case, likewise for the discrete frequency after resolution of the ambiguities the non-symmetrically located region k=0,... N-1; the actual relative speeds and thus Doppler frequencies can assume both signs, so that the upper range, e.g. the upper half of k=0,... N-1 is to be mapped to negative values by subtraction of N.As an alternative to the modulation sequence according to the invention as shown in FIG. 6, as considered hitherto, it is also possible to use alternating values (i.e. alternately +1 and -1) for the modulation sub-sequence b 1( n)-this being illustrated in FIG. 9. The only essential difference is that the reception sub-sequences for this b 1( n) have an additional frequency with a period of two (with respect to the modulation rate of b 1( n)), that is to say that the magnitude peaks of the two associated FFTs Ẽ 1,1,k and Ẽ 1,2,k are shifted by half the FFT length N / 4, which is to be corrected accordingly for the further processing.Up to now, the case was considered that the discrete delay time m 0,i is integer and the modulation duration T m is equal to the sampling repetition time T s. For a non-integer discrete delay time m 0,i it could happen, in the case of an ideally rectangular modulation signal which also retains its ideal shape in the reception signal, that sampling takes place exactly at the edge where no meaningful information is to be obtained. In order to avoid this, either the sampling repetition time T s of the reception sequence can be provided smaller, e.g. half as large as the modulation duration T m or the shape of the modulation pulses is ground, for example, to an approximately triangular shape either directly during their generation or in the receiver. In the modulation signal-grinding approach and the general case of a non-integer discrete delay time m 0,i magnitude peaks occur in both FFTs Ẽ 1,1,k and Ẽ 1,2,k so that the correlation for both m j= 0,1 can be determined and the non-integer m 0,i can be determined by interpolating the values of the magnitude peaks of both correlations (it would also be possible to determine the non-integer m 0,i by interpolating the values of the magnitude peaks of the two FFTs or by combining, i.e. sum or difference, the values of FFT and correlation). When a sampling time T s( which is half as long as the modulation duration T m) is used, for example, the interleaving of the two modulation sub-sequences with respect to the sampling time has the period 4 and in each case two successive sampling values originate from the first sequence b 1( n) and the next two sampling values originate from the second sequence b 2( n) in the received signal; Therefore, on the one hand four possible positions must be taken into account for the reception sub-sequences e 1,i( n), i.e. four FFTsẼ 1,1,k,...,Ẽ 1,4,k are to be calculated, and on the other hand the sampling values corresponding to b 2( n) are to be set to zero (because they are no longer in the grid of period 2 and can therefore be omitted simply), so that the FFT is to be calculated over the full number of reception values (in contrast to half the number up to now). In at least a part of these FFTs, three magnitude peaks then occur from a single object; the greatest occurs at the correct position, i.e. the Doppler frequency of the object (in contrast to the above, there are no longer any ambiguities, which is advantageous), and two further magnitude peaks smaller by 3 dB are one quarter of the FFT length before and after it (these are to be ignored for the further processing). Since magnitude peaks occur in several of the four FFTs Fft 1,1,k,..., Fft 1,4,k a non-integer discrete runtime can be determined by interpolation (using the values of the FFTs or the correlations or their combinations).In addition to the case considered hitherto in which the two modulation sub-sequences b 1( n) and b 2( n) are interleaved alternately, that is to say with period two, it is also possible in principle to use relatively long periods for the interleaving and optionally also an unequal number of elements of b 1( n) and b 2( n) per period.In the lidar system according to FIG. 1 considered hitherto, the mixer is designed to be complex-valued, which represents a significant additional outlay (almost twice) in comparison with a real-valued mixer for the reception path. If a real valued mixer is used, one could only determine the magnitude of the relative velocity, not the sign, since there are two magnitude peaks at +k 0 and -k 0 in the FFTs. For ascertaining the sign, approaches via plausibility checking and / or tracking, i.e. tracking over a plurality of recording cycles, would be necessary. Alternatively, a complex valued modulation sequence, e.g. consisting of the four values +1, -1, +j and -j, may be used. For the first modulation subsequence b 1( n), one then periodically rotates over the four values +1, +j, -1, -j, so that for an object with the relative speed zero, the magnitude peak lies at a quarter of the FFT length-negative relative speeds lie below (i.e. have a lower frequency), positive ones lie above it. This first modulation sub-sequence b 1( n), which in turn has a period 4, is furthermore alternately interleaved with the second modulation sub-sequence b 2( n), which can also assume these four values +1, -1, +j and -j, for example in a pseudo-random sequence. Although the generation of such a complex-valued sequence requires an increased circuit complexity, this is only required once, while in the case of a parallel receiver the complexity for a complex-valued receiver is frequently incurred. In the case of a complex-valued modulation sequence, the conjugate complex of the modulation sequence is to be used for the correlation. Alternatively, the conjugated complex of the reception sequence can also be used, provided that this is complex-valued.Up to now, the case was considered that no window function is used for the FFT, i.e. no multiplication of the input values of the FFT by a type of bell curve; this would only be necessary or expedient if two objects with a similar relative speed and significantly different reflection intensity can occur and are to be separated at the same distance in a pixel. In particular, if no window function is used at the input of the FFT, the sensitivity at the output of the FFT is reduced (i.e. the detection capability of objects with weak reflectivity and high distance) if the Doppler index corresponding to the relative speed is not integer, i.e. the magnitude peak is divided between two adjacent FFT values. This effect can be reduced by selecting the length of the FFT to be higher than that of its input signal, i.e. zeros are appended to the input signal, which is referred to as zero padding.With regard to the second modulation subsequence b 2( n), it should also be noted that it can be formed not only from a pseudo-random change between discrete phase values, but also from a code whose auto-correlated low side lobes-for example a Gold code known from the literature. This leads to a higher dynamic range of the correlation, but only when there are no signal components (for example of objects of different relative speed) which represent noise in the signal to be correlated.

Claims

Coherently operating lidar system for detecting the environment, which - emits a phase-modulated signal, the phase of this signal being generated by switching between discrete values, that is to say from a sequence of phase values, and the times of this phase switching forming a subset of an equidistant time pattern, - receives the signals reflected back from objects, which signals are delayed with respect to the emitted signal by the distance-dependent propagation time and are advanced in frequency by the relative-speed-dependent Doppler effect, and converts them into a low-frequency signal by mixing and digitizes them into a reception sequence, and - determines the variable dimensions of time shift and frequency shift of signals reflected from objects from this reception sequence in digital signal processing means, - the phase modulation sequence contains two sequences which are periodically nested one inside the other, wherein the first sequence has constant or periodic phase values and serves primarily for determining the frequency shift, while the second sequence changes substantially irregularly between phase values and serves for determining the time shift, characterized in that - the phase modulation sequence contains two sequences which are periodically nested one inside the other, wherein this interleaving has the period N 1 with respect to the sampling time of the reception sequence, - for determining the frequency shift of objects N 1 discrete Fourier transformations are calculated via values of the reception sequence from the periodic grid of the first sequence, wherein the shift increases by in each case one grid value, - in each of these N 1 discrete Fourier transformations the respective frequencies of magnitude peaks which lie above a detection threshold are determined, the frequency of the reception sequence in the periodic and correspondingly shifted raster of the second sequence is rotated back to the respective frequencies and the respective raster shifts, a correlation is determined in each case between the sequence thus generated and the second modulation sequence, and the respective time shift and thus object distance are determined from values of this correlation and the respective raster shift and the radial relative speed of the respective object is determined from the respective frequency.The lidar system of claim 1, wherein the first sequence alternates between two phase values that are different by about 180°.The lidar system of any of the preceding claims, wherein the second sequence either pseudo-randomly alternates between discrete phase values or consists of a code whose auto-correlated low side lobes.Lidar system according to one of the above claims, in which binary phase modulation, i.e. consisting of only two phase values differing by approximately 180°, is used.Lidar system according to one of the above claims, wherein the two sequences are sequences interleaved alternately, i.e. with period two with respect to the modulation rate.Lidar system according to one of the above claims, in which a phase modulation sequence consisting of two sequences periodically nested one inside the other is periodically repeated, wherein in particular during continuous scanning of the laser beam the cyclic property of the modulation and reception sequence is utilized and detection in respectively different directions is realized.Lidar system according to one of the above claims, wherein the laser beam continuously scans and overlapping sections of a long length, which includes two sequences periodically nested one inside the other, are used for the successive detection directions.Lidar system according to one of Claims 1 - 6, in which a first detection threshold is used to determine the magnitude peaks of the discrete Fourier transforms, the complex value of the discrete Fourier transforms and the respective complex value of the associated correlation are added up several times at the magnitude peaks thereof, if appropriate, with compensation for the possible phase shifts, and these sums are checked to a second detection threshold, wherein the first detection threshold is less high above the noise.

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