Capacitive measuring circuit with increased interference immunity to external alternating fields

Abstract

A capacitive measuring circuit comprising: an antenna electrode; a first AC signal generator (21) configured to generate a first AC voltage signal; a second AC signal generator (24) configured to generate a second AC voltage signal, wherein the second AC voltage signal has a lower frequency than the first AC voltage signal; a first mixer (23) for mixing the first AC voltage signal and the second AC voltage signal to generate a modulated AC voltage signal; a multiplexer (30) having one input, one control input and two outputs, wherein one of the first AC signal generator (21) and one of the first mixer (23) is effectively coupled to the antenna electrode to apply the first AC voltage signal or the modulated AC voltage signal to the antenna electrode.and wherein the other is effectively coupled to the control input of the multiplexer (30), a control and evaluation unit effectively coupled to the antenna electrode or a separate receiver electrode, the control and evaluation unit comprising a current sensing circuit configured to measure current signals, wherein the current signals have an amplitude and / or phase of a current flowing in the antenna electrode or in the separate receiver electrode, the control and evaluation unit being configured to determine a capacitance to be measured based on the measured current signals and to output a signal indicating the determined capacitance; the current sensing circuit comprising a differential transimpedance amplifier circuit (40) having a common-mode voltage setting input, an inverting input (45), a non-inverting input (44), and an output (31).wherein the inverting input (45) and the non-inverting input (44) of the differential transimpedance amplifier circuit (40) are effectively connected to the antenna electrode or the separate receiver electrode via the two outputs of the multiplexer (30) such that the inverting input (45) and the non-inverting input (44) are alternately supplied with the current flowing in the antenna electrode or in the separate receiver electrode, wherein the multiplexer (30) is controlled by the first AC signal generator (21) or first mixer (23) effectively coupled to the control input of the multiplexer (30).

Description

Field of invention

[0001] The present invention relates generally to the technical field of capacitive measuring circuits and in particular to a capacitive measuring system with one or more electrodes, wherein the features of a conductive body, such as the shape and the placement, are determined by capacitive coupling via the electrically conductive body. Background of the invention

[0002] Capacitive measurement and / or detection systems have a wide range of applications and are frequently used, among other things, to detect the presence and / or position of a conductive body near an electrode of the system. A capacitive sensor, sometimes called an electric field sensor or proximity sensor, is a sensor that generates a signal that responds to the influence of something being sensed (a person, a part of a person's body, a pet, an object, etc.) on an electric field. A capacitive sensor generally includes at least one antenna electrode to which an electrical oscillation signal is applied, causing it to radiate an electric field into an area in space near the antenna electrode while the sensor is operating.The sensor has at least one measuring electrode, which itself could have one or more antenna electrodes, at which the influence of an object or a living being on the electric field is detected.

[0003] The technical document entitled “Electric Field Sensing for Graphical Interfaces” by J.R. Smith, published in Computer Graphics I / O Devices, May / June 1998 issue, pages 54–60, describes the concept of electric field sensing as it is used to perform non-contact three-dimensional position measurements, and in particular to measure the position of a human hand for the purpose of inputting three-dimensional positions into a computer. Within the general concept of capacitive measurement, the author distinguishes between distinct mechanisms, which he refers to as “loading mode,” “shunt mode,” and “transmit mode,” corresponding to different possible paths for the electric current. In loading mode, a voltage oscillation signal is applied to a transmitting electrode, which generates an oscillating electric field at ground. The object being measured modifies the capacitance between the transmitting electrode and ground.In "parallel mode," a voltage oscillation signal is applied to the transmitting electrode, creating an electric field at a receiver electrode. The displacement current induced at the receiver electrode is measured, allowing the displacement current through the measured body to be modified. In "transmitting mode," the transmitting electrode is brought into contact with the user's body, who then becomes a transmitter relative to a receiver, either through a direct electrical connection or via capacitive coupling.

[0004] Capacitive coupling is generally achieved by applying an AC voltage signal to a capacitive antenna electrode and measuring the current flowing from the antenna electrode in coupling mode, either to ground (in charging mode) or to the second electrode (receiving electrode). This current is typically measured by a transimpedance amplifier connected to the measuring electrode, which converts the current flowing into the measuring electrode into a voltage proportional to the current flowing into the electrode.

[0005] Fig. Figure 1 shows a typical state-of-the-art circuit configured to measure an unknown capacitance in so-called “charging mode”, which means that the capacitance is measured between an electrode of a capacitive sensor and ground or earth.

[0006] An alternating current source 1It generates an alternating voltage signal of a known frequency and amplitude, for example a periodic sine wave of 100 kHz and a peak amplitude of 1 V. The output node 2 the alternating current source 1 is connected to the non-inverting input of an operational amplifier 3 connected. The operational amplifier 3 is configured as a transimpedance amplifier. The operational amplifier 3 retains through the feedback effect of the associated feedback impedance 4 (preferably a capacitor connected in parallel with a resistor, wherein the impedance of the capacitor at operating frequency is at least 10 times lower than the resistance) at its inverting input essentially the same potential as at its non-inverting input, thereby making the read node 5 is kept at the same potential as the output 2of the AC voltage source. Accordingly, the voltage of the AC voltage source across the unknown capacitance to be measured is 6 laid out over their “plates”.

[0007] The unknown capacity 6 The flowing current is then given by its capacitance and the known voltage of the AC voltage source, whereby the current is also influenced by the feedback impedance. 4 flows when the input current is fed into the non-inverting input of the amplifier. 3 is essentially zero.

[0008] The voltage at the output 7 of the amplifier 3 It therefore responds to the voltage of the AC power source and the unknown capacitance. This amplifier output voltage is then combined with the mixer. 8 (for example, a switching mixer or a multiplier) mixed, with the mixer's local oscillator input 8 from the edition 2The AC voltage source is controlled. The mixer's output 8 is a DC voltage superimposed with multiples of the frequency of the AC voltage source, where the DC voltage level is adjusted to the amplitude of the amplifier output 7 and thus to the output voltage 2 the AC voltage source and the unknown capacitance 6 addresses.

[0009] Since only the DC voltage is desired, the multiples of the frequency of the AC voltage source are filtered out using the low-pass filter. 10 filtered out. The output signal 11 The low-pass filter output is a DC voltage that responds to the voltage of the AC power source and the unknown capacitance. Furthermore, an adjustable phase shift (preferably in selectable steps of 0 and 90 degrees) can be applied between the output and the AC power source. 2 the AC voltage source and the local oscillator input of the mixer 8inserted, thereby enabling the measurement of the complex impedance 6 instead of capacity 6 is made possible.

[0010] Fig. Figure 2 shows a typical state-of-the-art circuit configured to measure an unknown capacitance in so-called “coupling mode”, which means that the capacitance between two electrodes of a capacitive sensor is measured.

[0011] In this variant, an alternating voltage source is generated. 1 An alternating voltage signal of a known frequency and amplitude, for example a periodic sine wave of 100 kHz and a peak amplitude of 1 V. The output node 2 the alternating current source 1 is attached to the first disk of unknown capacity 6 connected. The second disk of unknown capacity 6 is connected to the inverting input of an operational amplifier 3connected. The non-inverting input of the amplifier. 3 is grounded. The operational amplifier 3 retains through the feedback effect of the associated feedback impedance 4 (preferably a capacitor connected in parallel with a resistor, wherein the impedance of the capacitor at operating frequency is at least 10 times lower than the resistance) at its inverting input essentially the same potential as at its non-inverting input, thereby making the read node 5 is held at earth potential. Therefore, the voltage of the AC voltage source is applied to the unknown capacitance being measured. 6 laid out over their “plates”.

[0012] The unknown capacity 6 The flowing current is then given by its capacitance and the known voltage of the AC voltage source, whereby the current is also influenced by the feedback impedance. 4flows when the input current is fed into the non-inverting input of the amplifier. 3 is essentially zero.

[0013] The voltage at the output 7 of the amplifier 3 It therefore responds to the voltage of the AC power source and the unknown capacitance. This amplifier output voltage is then combined with the mixer. 8 (for example, a switching mixer or a multiplier) mixed, with the mixer's local oscillator input 8 from the edition 2 The AC voltage source is controlled. The mixer's output 8 is a DC voltage superimposed with multiples of the frequency of the AC voltage source, where the DC voltage level is adjusted to the amplitude of the amplifier output 7 and thus to the output voltage 2 the AC voltage source and the unknown capacitance 6 addresses.

[0014] Since only the DC voltage is desired, the multiples of the frequency of the AC voltage source are filtered out using the low-pass filter. 10 filtered out. The output signal 11 The low-pass filter's DC voltage responds to the voltage of the AC power source and the unknown capacitance. Furthermore, an adjustable phase shift (preferably in selectable steps of 0 and 90 degrees) can be applied between the output and the AC source. 2 the AC voltage source and the local oscillator input of the mixer 8 inserted, thereby enabling the measurement of the complex impedance 6 instead of a capacity 6 is made possible.

[0015] In both state-of-the-art circuits, the gain of the transimpedance amplifier, which is provided by the operational amplifier 3 is formed, and the feedback impedance 4They are configured to be as large as possible to achieve low-noise performance, and the DC gain of the signal chain stages following the mixer can then be kept comparatively low to avoid DC offset problems. For example, in a practical implementation with an operating frequency of 100 kHz and a source amplitude of 1 V, the feedback impedance would be chosen as a 100 pF capacitor in parallel with a 1 MΩ resistor.

[0016] The output signal range of the operational amplifier 3However, this is limited, for example, to a peak amplitude of 2 V for a current supply of 5 V. This means that a parasitic alternating current with a peak amplitude of more than 126 μA, injected into the reading electrode of the capacitive sensor, will saturate the operational amplifier and introduce an error into the measurement of the unknown capacitance. Such parasitic alternating currents are generated, for example, by external noise sources, one example of which is the so-called "bulk current injection" (BCI) test used in the suitability testing of an occupant detection system. Object of the invention

[0017] The object of the present invention is to provide a robust capacitive measuring circuit that is less susceptible to such parasitic alternating currents. General description of the invention

[0018] To overcome the aforementioned problems, the present invention proposes a capacitive measuring circuit in which the mixer is connected upstream of the amplifier stage. The AC transimpedance amplifier upstream of the mixer in the prior art circuits is removed and replaced by a differential DC transimpedance amplifier or an integrator.

[0019] The mixer's DC offset voltage or current, combined with the high gain now required after the mixer, would result in an unacceptable DC offset at the signal chain's output. To eliminate the mixer offset effect, the gain stages after the mixer are coupled to the mixer output with AC current, and one of the signals entering the mixer is phase- or amplitude-modulated with a known low-frequency signal. An additional mixer after the AC-coupled gain stages is driven with the same low-frequency modulation signal, producing the desired DC output signal that responds to the capacitance being measured.

[0020] In a first preferred embodiment, the capacitive detection system comprises an antenna electrode, a first AC signal generator configured to generate a first AC voltage signal, a second AC signal generator configured to generate a second AC voltage signal, the second AC voltage having a lower frequency than the first AC voltage signal, and a first mixer for mixing the first AC voltage signal and the second AC voltage signal to generate a modulated AC voltage signal. Either the first AC signal generator or the first mixer is effectively coupled to the antenna electrode to apply the first AC voltage signal or the modulated AC voltage signal to the antenna electrode.

[0021] The capacitive detection system further comprises a control and evaluation unit effectively coupled to the antenna electrode or a separate receiver electrode, wherein the control and evaluation unit includes a current measurement circuit configured to measure current signals, the current signals having an amplitude and / or phase of a current flowing in the antenna electrode or in the separate receiver electrode, wherein the control and evaluation unit is configured to determine a capacitance to be measured based on the measured current signals and outputs a signal indicating the determined capacitance.According to one aspect of the invention, the current measuring circuit comprises a differential transimpedance amplifier circuit having a common-mode voltage setting input, an inverting input, a non-inverting input, and an output, wherein the inverting input and the non-inverting input of the transimpedance amplifier circuit are effectively connected to the antenna electrode or the separate receiver electrode by a multiplexer such that the inverting input and the non-inverting input are alternately supplied with the current flowing in the antenna electrode or the separate receiver electrode, wherein the multiplexer is effectively coupled to and controlled by the first AC signal generator and the first mixer.

[0022] In one variant of the above system, the first AC signal generator is effectively coupled to the antenna electrode to apply the first AC voltage signal to the antenna electrode, and the multiplexer is effectively coupled to the first mixer and controlled by the modulated AC voltage signal. In another variant, the first mixer is effectively coupled to the antenna electrode to apply the modulated AC voltage signal to the antenna electrode, and the multiplexer is effectively coupled to the first AC signal generator and controlled by the first AC voltage signal.

[0023] In one embodiment of the capacitive detection system, the multiplexer is effectively coupled to the other of the first AC signal generator and the first mixer by an adjustable phase shifter, so that the multiplexer can be controlled by different phase positions of the first AC signal or the modulated AC signal.

[0024] Finally, an output signal at the output of the differential transimpedance amplifier circuit is preferably filtered by a bandpass filter and then mixed with the second AC voltage signal of the second AC signal generator. Brief description of the drawings

[0025] Further details and advantages of the present invention will become apparent from the following detailed description of non-limiting embodiments with reference to the accompanying drawings, wherein:

[0026] Fig. 1 shows a state-of-the-art measuring circuit in “charging mode”;

[0027] Fig. 2 shows a state-of-the-art measuring circuit in “coupling mode”;

[0028] Fig. 3 shows a first embodiment of a measuring circuit according to the present invention in “charging mode”;

[0029] Fig. 4 shows an alternative embodiment of a measuring circuit according to the present invention in “charging mode”;

[0030] Fig. 5 shows a first embodiment of a measuring circuit according to the present invention in “coupling mode”;

[0031] Fig. 6 shows an alternative embodiment of a measuring circuit according to the present invention in “coupling mode”;

[0032] Fig. 7 a preferred embodiment of the circuit made of Fig. 3 shows. Description of preferred embodiments

[0033] The in Fig. The circuit shown in Figure 3 is a first embodiment that makes it possible to significantly improve the immunity of the capacitive measurement circuit to the injection of external parasitic alternating currents. The capacitive measurement circuit operates in the so-called charging mode.

[0034] The alternating current source 21 It generates an alternating voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and a peak amplitude of 1 V. Its output node 22 is connected to the input of the adjustable phase shifter 32 and a first input of the mixer 23 connected. A second AC power source 24 generates a second AC voltage signal of known frequency and amplitude, but with a lower frequency than the output frequency of the AC voltage source. 21 , for example a periodic square wave of 1 kHz and a peak amplitude of 1 V.

[0035] The outcome 25 the alternating current source 24 is connected to the second input, the local oscillator input, of the mixer 23 connected. The mixer 23 It multiplies the signals at its two inputs. For the specific example signals described above, a phase-modulated sine wave is generated at its output. 26 generated, that is, for the first half of the period of the output signal of the AC voltage source 24 will the outcome 26 identical to the output signal of the AC voltage source 21 be, and during the second half of the period of the alternating voltage source 24 will the outcome 26 the inverted version of the output signal of the AC voltage source 21 be.

[0036] Obviously, different waveforms can be used instead of the square waveform for the alternating voltage source. 24can be used, for example, a so-called binary quasi-random sequence or a wobble frequency or step frequency square wave.

[0037] The outcome 26 the mixer 23 will be added to the Sense-Guard capacity 43 of the capacitive sensor and the input for the common-mode voltage settings of the differential transimpedance amplifier 40 fed in. The differential transimpedance amplifier 40 holds its two entrances 44 and 45 at the same AC voltage level as its input for the common-mode voltage settings, therefore the nodes are 44 and 45 at the same potential as the node 26 Since the multiplexer 30 always one of the left plates of the capacitors 41 or 42 on the reading node 29 switches, the reading node 29 Therefore, the same alternating voltage as the node is always present.26 Therefore, the alternating voltage through the sense-guard capacitance is... 43 essentially zero, which eliminates the need to use one at the nodes 26 connected protective electrode allows for the protection at the nodes 29 connected reading electrode versus unwanted parasitic capacitances between the reading node 29 and to shield the earth.

[0038] Furthermore, the plates have an unknown capacitance or impedance. 28 also the known alternating voltage of the node 26 on. The unknown capacitance or impedance 28 The flowing current is therefore determined by its apparent resistance and the known alternating voltage. The current also flows through the multiplexer. 30 The switching position of the multiplexer 30 is determined by the polarity of the output signal of the phase shifter 32 controlled. Therefore, the current flows through the capacitor. 41into the positive input 44 of the differential transimpedance amplifier 40 or through the capacitor 42 into the negative input 45 of the differential transimpedance amplifier 40 .

[0039] The differential transimpedance amplifier 40 It amplifies the difference between the currents at its positive and negative inputs and outputs a voltage that responds to the input current difference. The multiplexer 30 Therefore, together with the differential transimpedance amplifier, it forms 40 a switching-synchronous rectifier or a switching-synchronous demodulator with a current-switching input and a low AC input resistance, since the AC voltage on the node 29 essentially equal to the tension at the node 26 is and is essentially not affected by the unknown capacitance or impedance 28 depends.

[0040] The output DC voltage of the synchronous rectifier corresponds to the known AC voltage at the node. 29 , the unknown capacitance or impedance 28 and those with the phase shifter 32 The set phase shift is indicated. Typically, the phase shift of the phase shifter is 32 First, the phase shift is set to 0 degrees, then a first measurement is taken, then the phase shift is set to 90 degrees, then a second measurement is taken. By performing two measurements, the complex impedance of the unknown capacitance or impedance can be determined. 28 will be calculated.

[0041] At the exit 31 of the differential transimpedance amplifier 40 A first alternating voltage signal appears with the same frequency as the frequency of the alternating voltage source. 24, which is a mirror signal of the signal from the AC voltage source, derived from a second AC voltage mirror signal 24 is superimposed and at twice the frequency of the output signal of the AC voltage source 21 is shifted. Further images are also generated at the harmonics of the AC output signal. 21 generated.

[0042] Since only the first, low-frequency AC voltage signal is of interest for capacitive measurement, the higher-frequency components are filtered out by the amplifier configured as a bandpass filter. 33 eliminates the first, low-frequency AC signal, which amplifies the first, low-frequency AC signal and simultaneously eliminates any offset DC signal at the mixer's output. 31 It eliminates, and at the same time essentially suppresses any signal that contains components other than the desired, initial low-frequency signal. The amplifier 33This can be used, for example, for the assumed output frequency of the AC voltage source. 24 be configured with a 1 kHz AC-coupled (capacitively coupled) 4-pole Butterworth low-pass filter with a cutoff frequency of 1.5 kHz, which is implemented, for example, with two operational amplifiers in a Sallen-Key configuration.

[0043] The resulting signal 34 from 1 kHz at the output of the bandpass amplifier 33 It is then passed through the mixer again. 35 with the alternating current output signal of the alternating voltage signal source 24 mixed and by the amplifier configured as a low-pass filter 37 amplified and low-pass filtered. The amplifier 37 This can be achieved, for example, with a DC-coupled 2-pole Butterworth low-pass filter with a cutoff frequency of 100 Hz, which is implemented, for example, with an operational amplifier with a Sallen-Key configuration.

[0044] Another preferred, less complex option is to use the amplifier. 37 to replace it with a passive RC filter that has a DC gain of one when the amplifier 33 with sufficient amplification for the application.

[0045] The DC voltage at the final output 38 This is then due to the effect of the mixer. 35 and the low-pass effect of the amplifier 37 on the amplitude of the signal 34 from 1 kHz at the mixer input 35 to address. Ultimately, the DC voltage affects the current through the unknown capacitance or impedance. 28 to.

[0046] By performing the two successive measurements described above (the first with the phase shifter set to a phase of 0 degrees) 32, the second one with the phase shifter set to a phase shift of 90 degrees 32 ) and by combining the two successive DC levels present at the output 38 The impedance of the unknown capacitance or the impedance can be obtained. 28 will be calculated.

[0047] The sequencing of the measurements and the measurement of the DC current level at the output 38 and the calculation of the impedance of the unknown capacitance or the impedance 28 These operations are preferably performed by a microcontroller equipped with an integrated ADC (analog-to-digital converter). In another embodiment, the mixer can 35 and the low-pass filtering amplifier 37 Each of these can be implemented within a microcontroller equipped with an ADC by connecting the ADC input directly to the output. 34 of the amplifier 33connected and the mixer is implemented as software by optionally multiplying the ADC results by the values ​​+1 and -1, with the AC voltage source 24 synchronized and then the resulting values ​​are low-pass filtered or integrated by the software.

[0048] The differential transimpedance amplifier 40 can also be interpreted as a differential integrator with a current-switching input, where the integrator takes the AC voltages of each of the inputs. 44 and 45 maintains the same AC potential as its input for the common-mode voltage settings.

[0049] To optimally suppress injected parasitic alternating currents, it is preferred to first perform a sweep or step sampling of the frequency of the AC voltage source. 21to perform, to identify the frequency or frequencies at which the parasitic alternating currents are located, and then to determine the measurement frequency of the alternating voltage source 21 to set to a frequency at which no parasitic alternating current was detected and where no subharmonic oscillation of a parasitic alternating current is present.

[0050] An alternative to the one in Fig. The circuit shown in 3 is the one in Fig. Circuit 4 shown. The capacitive measuring circuit operates in the so-called charging mode. The difference to the circuit in Fig. 3 consists in the fact that the input for the common-mode voltage settings of the differential transimpedance amplifier 40 and the protective electrode (top plate of the Guard-Sense capacitor) 43 ) now directly with the exit 22 the alternating current source 21 is connected, and that the input of the phase shifter is connected to the output of the mixer. 26is connected. The remaining operation of the circuit is related to the circuit in Fig. 3 identical, except that the input for the common-mode voltage settings of the differential transimpedance amplifier 40 and the unknown capacitance or impedance 28 now with an unmodulated periodic single-frequency signal, and not with a modulated signal as in Fig. 3, is being supplied.

[0051] Fig. Figure 5 shows a circuit similar to the circuit in Fig. 3, with the exception that the capacitive measuring circuit operates in so-called coupling mode. Therefore, only the difference to the circuit from Fig. 3 described.

[0052] The outcome 26 the mixer 23 is related to the unknown capacitance or impedance to be measured 28 connected. The other node of the capacitance or impedance to be measured. 28is connected to the multiplexer 30 connected. Since the input is for the common-mode voltage settings of the differential transimpedance amplifier. 40 when connected to earth, the alternating voltage at the node 29 through the capacitors 41 and 42 and the multiplexer 30 an alternating voltage of zero. Therefore, the known alternating voltage of the node is zero. 26 also due to the unknown capacitance or impedance 28 to.

[0053] Similar to the circuit in Fig. 3. The current is then measured through the unknown capacitance or impedance. 28 through the known alternating voltage of the node 26 and the impedance of the unknown capacitance or the impedance 28 defined. Furthermore, it speaks, as in Fig. 3, the DC voltage of the output 38 depends on the impedance.

[0054] An alternative to the one in Fig. The circuit shown in section 5 is the one in Fig. Circuit shown in 6. The capacitive measuring circuit operates in so-called coupling mode. The difference to the circuit in Fig. 5 consists in the fact that the unknown capacitance or impedance to be measured 28 now directly with the exit 22 the alternating current source 21 is connected and the input of the phase shifter is connected to the output of the mixer. 26 is connected. The remaining operation of the circuit is related to the circuit in Fig. 5 identical, except that the unknown capacitance or impedance 28 now with an unmodulated periodic single-frequency signal, and not with a modulated signal as in Fig. 5, is being supplied.

[0055] Fig. Figure 7 shows a preferred embodiment of the circuit in Fig. 3. Only the differences in the circuitry in Fig. 7 compared to the circuit in Fig. Section 3 is described. The differential transimpedance amplifier 40 in Fig. 3 is through operational amplifiers 411 and 421 , capacitors 413 and 423 and resistors 414 and 424 and the differential amplifier 430 replaced. The differential amplifier 430 is preferably implemented with a suitably configured operational amplifier; an example of this can be found in “The Art of Electronics, 2nd edition, Paul Horowitz and Winfield Hill”, page 185, Fig. 4.18.

[0056] The operational amplifiers 411 and 412 are configured identically, therefore only the operational amplifier configuration is affected. 411 described. The capacitor 41 couples the one from the multiplexer 30 incoming alternating current from the operational amplifier 411and associated components of the existing transimpedance amplifier. The capacitor 413 and the resistance 414 They close the feedback path around the operational amplifier and determine the gain of the transimpedance amplifier. This is due to the action of the feedback components. 413 and 414 The voltage difference between the inputs of the operational amplifiers will be measured. 411 essentially held at zero volts. Since the positive input is connected to the output 26 the mixer 23 The input of the transimpedance amplifier, which is connected around the operational amplifier, is located where the input of the transimpedance amplifier is located. 411 is formed, which is connected to the capacitor 41 is connected, at the same AC voltage potential, and since the operational amplifier 421 The input of the transimpedance amplifier, which is configured identically, is the operational amplifier. 421 is formed, which is connected to the capacitor 42is connected, also at the same AC potential. Preferred values ​​for the capacitors 41 and 42 500 nF is for the capacitors 413 and 423 10 nF and for the resistors 414 and 424 500 kΩ.

[0057] An example of operational amplifiers 411 and 421 The transistor used is the LT1057 from Linear Technology. The voltage gain of the differential amplifier is preferably set to 10. Since there is no amplification device between the input of the capacitive measurement circuit and the mixer, and since the differential transimpedance amplifier is only needed to amplify DC signals, the capacitive measurement circuit is essentially less sensitive to parasitic AC current entering the reading electrode of the capacitive sensor (node). 29 in Fig. 7) is fed in. For example, it is distorted during the circuit in Fig. 7. With the components defined above, an injected parasitic current of 10 mA peak amplitude did not produce a remarkable measurement result, which was conveniently matched to the peak amplitude of 126 μA for the circuit in Fig. 1 is comparable. QUOTES INCLUDED IN THE DESCRIPTION

[0058] This list of documents cited by the applicant was automatically generated and is included solely for the reader's convenience. The list is not part of the German patent or utility model application. The DPMA accepts no liability for any errors or omissions. Cited non-patent literature

[0059] “Electric Field Sensing for Graphical Interfaces” by JR Smith, published in Computer Graphics I / O Devices, May / June 1998 issue, pages 54–60

[0003] “The Art of Electronics, 2nd edition, Paul Horowitz and Winfield Hill”, page 185, Fig. 4.18

[0055]

Claims

[1] Capacitive detection system, comprising: an antenna electrode; a first AC signal generator configured to generate a first AC voltage signal, a second AC signal generator configured to generate a second AC voltage signal, wherein the second AC voltage signal has a lower frequency than the first AC signal, a first mixer for mixing the first AC signal and the second AC signal and for generating a modulated AC signal, wherein either the first AC signal generator or the first mixer is effectively coupled to the antenna electrode in order to apply the first AC voltage signal or the modulated AC voltage signal to the antenna electrode, a control and evaluation unit effectively coupled to the antenna electrode or a separate receiver electrode, wherein the control and evaluation unit has a current measurement circuit configured to measure current signals, wherein the current signals have an amplitude and / or phase of a current flowing in the antenna electrode or in the separate receiver electrode, wherein the control and evaluation unit is configured to determine a capacitance to be measured based on the measured current signals and to output a signal indicating the determined capacitance; wherein the current sensing circuit comprises a differential transimpedance amplifier circuit having a common-mode voltage setting input, an inverting input, a non-inverting input and an output, wherein the inverting input and the non-inverting input of the transimpedance amplifier circuit are effectively connected to the antenna electrode or the separate receiver electrode by a multiplexer such that the inverting input and the non-inverting input are alternately supplied with the current flowing in the antenna electrode or the separate receiver electrode, wherein the multiplexer is effectively coupled to or controlled by the first AC signal generator and the first mixer. [2] Capacitive detection system according to claim 1, wherein the first AC signal generator is effectively coupled to the antenna electrode to apply the first AC voltage signal to the antenna electrode and wherein the multiplexer is effectively coupled to the first mixer and is controlled by the modulated AC voltage signal. [3] Capacitive detection system according to claim 1, wherein the first mixer is effectively coupled to the antenna electrode to apply the modulated AC voltage signal to the antenna electrode, and wherein the multiplexer is effectively coupled to the first AC signal generator and is controlled by the first AC voltage signal. [4] Capacitive detection system according to one of claims 1 to 3, wherein the multiplexer is effectively coupled to the other of the first AC signal generator or the first mixer by an adjustable phase shifter, so that the multiplexer can be controlled by different phase positions of the first AC signal or the modulated AC signal. [5] Capacitive detection system according to one of claims 1 to 4, wherein an output signal at the output of the differential transimpedance amplifier circuit is filtered by a bandpass filter and then mixed with the second AC voltage signal of the second AC signal generator.