Capacitive measuring circuit with increased interference immunity to external alternating fields

Abstract

Capacitive measuring circuit, comprising: a first AC signal generator (21) configured to generate a first AC voltage signal, a second AC signal generator (24) configured to generate a second AC voltage signal, wherein the second AC voltage signal has a lower frequency than the first AC signal, a first mixer (23) for multiplicative mixing of the first AC signal and the second AC signal and for generating a modulated AC signal, a control and evaluation unit effectively coupled to an antenna electrode or a separate receiver electrode, wherein the control and evaluation unit has a current measurement circuit configured to measure current signals, wherein the current signals have an amplitude and / or phase of a current flowing in the antenna electrode or in the separate receiver electrode, wherein the control and evaluation unit is configured to determine a capacitance to be measured based on the measured current signals and to output a signal indicating the determined capacitance; wherein the current measuring circuit comprises a capacitance-to-current or voltage transformer (27) coupled via the capacitance (28) to be determined, wherein the current or voltage transformer (27) has an input and an output (29), and a second mixer (30) with a first and a second input, which is effectively coupled via its first input to the output (29) of the capacitance-to-current or voltage transformer (27), and wherein one of the first AC signal generator (21) and the first mixer (23) is effectively coupled to the capacitance-to-current or voltage converter (27) to supply the first AC voltage signal or the modulated AC voltage signal to the input of the capacitance-to-current or voltage converter (27), and the other is effectively coupled to the second input of the second mixer (30).

Description

Field of invention

[0001] The present invention relates generally to the technical field of capacitive measuring circuits and in particular to a capacitive measuring system with one or more electrodes, wherein the features of a conductive body, such as the shape and the placement, are determined by capacitive coupling via the electrically conductive body. Background of the invention

[0002] Capacitive measurement and / or detection systems have a wide range of applications and are frequently used, among other things, to detect the presence and / or position of a conductive body near an electrode of the system. A capacitive sensor, sometimes called an electric field sensor or proximity sensor, is a sensor that generates a signal that responds to the influence of something being sensed (a person, a part of a person's body, a pet, an object, etc.) on an electric field. A capacitive sensor generally includes at least one antenna electrode to which an electrical oscillation signal is applied, causing it to radiate an electric field into an area in space near the antenna electrode while the sensor is operating.The sensor has at least one measuring electrode, which itself could have one or more antenna electrodes, at which the influence of an object or living being on the electric field is detected. Such a capacitive measuring system is known, for example, from US Patent 6,025,726 A.

[0003] The technical document entitled "Electric Field Sensing for Graphical Interfaces" by J.R. Smith, published in Computer Graphics I / O Devices, May / June 1998 issue, pages 54-60, describes the concept of electric field sensing as it is used to perform non-contact three-dimensional position measurements, and in particular to measure the position of a human hand for the purpose of inputting three-dimensional positions into a computer. Within the general concept of capacitive measurement, the author distinguishes between distinct mechanisms, which he refers to as "loading mode," "shunt mode," and "transmit mode," corresponding to different possible paths for the electric current. In loading mode, a voltage oscillation signal is applied to a transmitting electrode, which generates an oscillating electric field at ground. The object being measured modifies the capacitance between the transmitting electrode and ground.In "parallel mode," a voltage oscillation signal is applied to the transmitting electrode, creating an electric field at a receiver electrode. The displacement current induced at the receiver electrode is measured, allowing the displacement current through the measured body to be modified. In "transmitting mode," the transmitting electrode is brought into contact with the user's body, who then becomes a transmitter relative to a receiver, either through a direct electrical connection or via capacitive coupling.

[0004] Capacitive coupling is generally achieved by applying an AC voltage signal to a capacitive antenna electrode and measuring the current flowing from the antenna electrode in coupling mode, either to ground (in charging mode) or to the second electrode (receiving electrode). This current is typically measured by a transimpedance amplifier connected to the measuring electrode, which converts the current flowing into the measuring electrode into a voltage proportional to the current flowing into the electrode.

[0005] Fig. Figure 1 shows a typical state-of-the-art circuit configured to measure an unknown capacitance in so-called "charging mode", which means that the capacitance is measured between an electrode of a capacitive sensor and ground or earth.

[0006] An AC voltage source 1 generates an AC voltage signal of known frequency and amplitude, for example, a periodic sine wave of 100 kHz and a peak amplitude of 1 V. The output node 2 of the AC voltage source 1 is connected to the non-inverting input of an operational amplifier 3. The operational amplifier 3 is configured as a transimpedance amplifier. Due to the feedback effect of the associated feedback impedance 4 (preferably a capacitor connected in parallel with a resistor, where the impedance of the capacitor at operating frequency is at least 10 times lower than the resistance), the operational amplifier 3 maintains essentially the same potential at its inverting input as at its non-inverting input, thus keeping the read node 5 at the same potential as the output 2 of the AC voltage source.Accordingly, the voltage of the AC voltage source is applied to the unknown capacitance 6 to be measured via its "plates".

[0007] The current flowing through the unknown capacitance 6 is then given by its capacitance and the known voltage of the AC voltage source, with the current also flowing through the feedback impedance 4 when the input current into the non-inverting input of the amplifier 3 is essentially zero.

[0008] The voltage at output 7 of amplifier 3 responds to the voltage of the AC voltage source and the unknown capacitance. This amplifier output voltage is then mixed with mixer 8 (for example, a switching mixer or a multiplier), whereby the local oscillator input of mixer 8 is driven by output 2 of the AC voltage source. The output of mixer 8 is a DC voltage superimposed with multiples of the frequency of the AC voltage source, with the DC voltage level responding to the amplitude of amplifier output 7 and thus to the output voltage 2 of the AC voltage source and the unknown capacitance 6.

[0009] Since only the DC voltage is desired, the frequency multiples of the AC voltage source are filtered out by the low-pass filter 10. The output signal 11 of the low-pass filter is a DC voltage that responds to the voltage of the AC voltage source and the unknown capacitance. Furthermore, an adjustable phase shift (preferably from selectable steps of 0 and 90 degrees) can be inserted between the output 2 of the AC voltage source and the local oscillator input of the mixer 8, thus enabling the measurement of the complex impedance 6 instead of the capacitance 6.

[0010] Fig. Figure 2 shows a typical state-of-the-art circuit configured to measure an unknown capacitance in so-called "coupling mode", which means that the capacitance between two electrodes of a capacitive sensor is measured.

[0011] In this variant, an AC voltage source 1 generates an AC voltage signal of known frequency and amplitude, for example, a periodic sine wave of 100 kHz and a peak amplitude of 1 V. The output node 2 of the AC voltage source 1 is connected to the first plate of the unknown capacitor 6. The second plate of the unknown capacitor 6 is connected to the inverting input of an operational amplifier 3. The non-inverting input of the amplifier 3 is connected to ground. Due to the feedback effect of the associated feedback impedance 4 (preferably a capacitor connected in parallel with a resistor, where the impedance of the capacitor at operating frequency is at least 10 times lower than the resistor), the operational amplifier 3 maintains essentially the same potential at its inverting input as at its non-inverting input, thus keeping the read node 5 at ground potential.Accordingly, the voltage of the AC voltage source is applied to the unknown capacitance 6 to be measured via its "plates".

[0012] The current flowing through the unknown capacitance 6 is then given by its capacitance and the known voltage of the AC voltage source, with the current also flowing through the feedback impedance 4 when the input current into the non-inverting input of the amplifier 3 is essentially zero.

[0013] The voltage at output 7 of amplifier 3 responds to the voltage of the AC voltage source and the unknown capacitance. This amplifier output voltage is then mixed with mixer 8 (for example, a switching mixer or a multiplier), whereby the local oscillator input of mixer 8 is driven by output 2 of the AC voltage source. The output of mixer 8 is a DC voltage superimposed with multiples of the frequency of the AC voltage source, with the DC voltage level responding to the amplitude of amplifier output 7 and thus to the output voltage 2 of the AC voltage source and the unknown capacitance 6.

[0014] Since only the DC voltage is desired, the frequency multiples of the AC voltage source are filtered out by the low-pass filter 10. The output signal 11 of the low-pass filter is the DC voltage responding to the voltage of the AC voltage source and the unknown capacitance. Furthermore, an adjustable phase shift (preferably from selectable steps of 0 and 90 degrees) can be inserted between the output 2 of the AC voltage source and the local oscillator input of the mixer 8, thus enabling the measurement of the complex impedance 6 instead of a capacitance 6.

[0015] In both state-of-the-art circuits, the gain of the transimpedance amplifier, formed by the operational amplifier 3, and the feedback impedance 4 are configured to be as large as possible to achieve low-noise performance, and the DC gain of the signal chain stages following the mixer can then be kept comparatively low to avoid DC offset problems. For example, in a practical implementation with an operating frequency of 100 kHz and a source amplitude of 1 V, the feedback impedance would be chosen as a 100 pF capacitor in parallel with a 1 MΩ resistor.

[0016] However, the output signal range of operational amplifier 3 is limited, for example, to a peak amplitude of 2 V for a current supply of 5 V. This means that a parasitic alternating current with a peak amplitude of more than 126 µA, injected into the reading electrode of the capacitive sensor, will cause the operational amplifier to saturate and introduce an error into the measurement of the unknown capacitance. Such parasitic alternating currents are generated, for example, by external noise sources, one example of which is the so-called "bulk current injection" (BCI) test used in the suitability testing of an occupant detection system. Object of the invention

[0017] The object of the present invention is to provide a robust capacitive measuring circuit that is less susceptible to such parasitic alternating currents. General description of the invention

[0018] To overcome the aforementioned problems, the present invention proposes a capacitive measuring circuit in which the transimpedance amplifier upstream of the mixer in prior art circuits is removed or replaced by an amplifier with low gain and consequently a large dynamic range. The offset DC voltage or offset DC current of the mixer, together with the high gain now required after the mixer, would lead to an unacceptable offset DC voltage at the output of the signal chain. To eliminate the effect of the mixer offset, the gain stages after the mixer are coupled to the mixer output with alternating current, and one of the signals entering the mixer is phase- or amplitude-modulated with a known low-frequency signal.An additional mixer after the AC-coupled amplification stages is driven with the same low-frequency modulation signal, resulting in the desired DC voltage output signal that responds to the capacitance being measured.

[0019] In a first preferred embodiment, the capacitive measuring circuit comprises a first AC signal generator configured to generate a first AC voltage signal, a second AC signal generator configured to generate a second AC voltage signal, wherein the second AC voltage has a lower frequency than the first AC voltage signal, and a first mixer for multiplicatively mixing the first AC voltage signal and the second AC voltage signal to generate a modulated AC voltage signal.

[0020] The capacitive measuring circuit further comprises a control and evaluation unit effectively coupled to an antenna electrode or a separate receiver electrode, wherein the control and evaluation unit includes a current measuring circuit configured to measure current signals, the current signals having an amplitude and / or phase of a current flowing in the antenna electrode or in the separate receiver electrode, wherein the control and evaluation unit is configured to determine a capacitance to be measured based on the measured current signals and outputs a signal indicating the determined capacitance.According to one aspect of the invention, the current measuring circuit comprises a capacitance-to-current or voltage converter coupled via the capacitance to be determined, the current or voltage converter having an input and an output, and a second mixer with a first and a second input, the first of which is effectively coupled to the output of the capacitance-to-current or voltage converter via its first input. Of the first AC signal generator and the first mixer, one is effectively coupled to the capacitance-to-current or voltage converter to supply the first AC voltage signal or the modulated AC voltage signal to the input of the capacitance-to-current or voltage converter, and the other is coupled to the second input, preferably a local oscillator input, of the second mixer.

[0021] In one embodiment of the above system, the first AC signal generator is effectively coupled to the second mixer to apply the first AC signal to the second input, preferably a local oscillator input, of the second mixer, and the capacitance-to-current or voltage converter is effectively coupled to the first mixer to apply the modulated AC signal to the input of the capacitance-to-current or voltage converter. In another embodiment, the first mixer is effectively coupled to the second mixer to apply the modulated AC signal to the second input, preferably a local oscillator input, of the second mixer, and the capacitance-to-current or voltage converter is effectively coupled to the first AC signal generator to apply the first AC signal to the input of the capacitance-to-current or voltage converter.

[0022] In one embodiment of the capacitive measuring circuit, a phase shifter (32) is coupled in front of the second input of the second mixer (30) in order to couple phase-shifted versions of the first AC voltage signal or the modulated AC voltage signal to the second input of the second mixer.

[0023] Finally, an output signal at the output of the second mixer is preferably amplified and filtered through a bandpass filter and then mixed in a third mixer with the second AC voltage signal of the second AC signal generator. Brief description of the drawings

[0024] Further details and advantages of the present invention will become apparent from the following detailed description of non-limiting embodiments with reference to the accompanying drawings, wherein: Fig. 1 shows a state-of-the-art measuring circuit in "charging mode"; Fig. 2 shows a state-of-the-art measuring circuit in "coupling mode"; Fig. 3 shows a first embodiment of a measuring circuit according to the present invention; Fig. 4 shows an alternative embodiment of a measuring circuit according to the present invention; Fig. 5 a preferred embodiment of the circuit made of Fig. 3 shows; Fig. 6 an alternative embodiment of the circuit made of Fig. 3 shows. Description of preferred embodiments

[0025] The in Fig. The circuit shown in Figure 3 is a first embodiment that makes it possible to significantly improve the immunity of the capacitance measurement circuit to the injection of external parasitic alternating currents. The AC voltage source 21 generates an AC voltage signal of known frequency and amplitude, for example, a periodic sine wave of 100 kHz and a peak amplitude of 1 V. Its output node 22 is connected to a first input of the mixer 23.

[0026] A second AC voltage source 24 generates a second AC voltage signal of known frequency and amplitude, but with a lower frequency than the output frequency of the AC voltage source 21, for example, a periodic square wave of 1 kHz and a peak amplitude of 1 V. The output 25 of the AC voltage source 24 is connected to the second input, the local oscillator input, of the mixer 23. The mixer 23 multiplies the signals at its two inputs.

[0027] For the specific example signals described above, a phase-modulated sine wave is generated at its output 26, that is, for the first half of the period of the output signal of the AC voltage source 24, the output 26 will be identical to the output signal of the AC voltage source 21, and during the second half of the period of the AC voltage source 24, the output 26 will be the inverted version of the output signal of the AC voltage source 21.

[0028] Obviously, various waveforms can be used instead of the square waveform for the AC voltage source 24, for example, a so-called binary quasi-random sequence or a sweep frequency or step frequency square wave. Reference numeral 27 is a capacitance-to-current or voltage converter that receives an AC voltage 26 at its input, copies the same AC voltage across the unknown capacitance or impedance 28, measures the AC current through the unknown capacitance or impedance 28, and outputs a current or voltage to its output 29 that responds to the current.

[0029] The capacitance-to-current or voltage converter 27, together with the unknown capacitance or impedance 28, can be configured in either a charging or coupling measurement mode. Details of the implementation are described in the examples below. The capacitance-to-current or voltage converter 27 can be, for example, a so-called current conveyor type II, as described in "Current Conveyor Theory and Practice" by AS Sedra and GW Roberts, published in "Advances in Analog Integrated Circuit Design" by C. Toumazou, FJ Lidgey, and DG Haigh (editors), Peter Peregrinus Limited, London, England, pp. 93–126, 1990, or a transistor operating in a common-base configuration, or a low-gain transimpedance amplifier.

[0030] The output signal 29 of the capacitance-to-current or voltage converter 27 is fed to the first input of the mixer 30. A switching mixer, which multiplies the signal at its input by +1 or -1, or a multiplier mixer for even better rejection of injected parasitic AC currents can be used. The second input, the local oscillator input of the mixer 30, is fed by a phase-shifted version of the output signal of the AC voltage source 21.

[0031] The phase shift is generated by the adjustable phase shifter 32. Typically, the phase shift of the phase shifter 32 is first set to 0 degrees, then a first measurement is taken, then the phase shift is set to 90 degrees, and then a second measurement is taken. By performing two measurements, the complex impedance of the unknown capacitance or impedance 28 can be calculated.

[0032] At output 31 of mixer 30, a first AC signal appears with the same frequency as the frequency of AC voltage source 24. This signal is superimposed on a second AC mirror image of the signal from AC voltage source 24, shifted by twice the frequency of the output signal from AC voltage source 21. Depending on the mixer, further images are generated at the harmonics of the output signal from AC voltage 21.

[0033] Since only the first, low-frequency AC signal is of interest for capacitive measurement, the higher-frequency components are eliminated by the amplifier 33, configured as a bandpass filter. This amplifier amplifies the first, low-frequency AC signal and simultaneously eliminates any offset DC signal at the output of the mixer 31, essentially suppressing any signal containing components of a higher frequency than the desired first, low-frequency signal. For example, the amplifier 33 can be configured for the assumed output frequency of the AC voltage source 24 of 1 kHz, using an AC-coupled (capacitively coupled) 4-pole Butterworth low-pass filter with a cutoff frequency of 1.5 kHz, implemented, for example, with two operational amplifiers in a Sallen-Key configuration.

[0034] The resulting 1 kHz signal 34 at the output of the bandpass amplifier 33 is then mixed again by the mixer 35 with the AC output signal of the AC voltage signal source 24 and amplified and low-pass filtered by the amplifier 37, which is configured as a low-pass filter. The amplifier 37 can, for example, be implemented with a DC-coupled 2-pole Butterworth low-pass filter with a cutoff frequency of 100 Hz, which could be implemented, for example, with an operational amplifier in a Sallen-Key configuration.

[0035] Another preferred, less complex option is to replace the amplifier 37 with a passive RC filter having a DC gain of one, if the amplifier 33 has been chosen with sufficient gain for the application.

[0036] The DC voltage at the final output 38 then responds to the amplitude of the 1 kHz signal 34 at the input of the mixer 35 due to the effect of the mixer 35 and the low-pass effect of the amplifier 37. Finally, the DC voltage responds to the current through the unknown capacitance or impedance 28.

[0037] By performing the two successive measurements described above (the first with the phase shifter 32 set to a phase of 0 degrees, the second with the phase shifter 32 set to a phase shift of 90 degrees) and by combining the two successive DC current levels obtained at the output 38, the impedance of the unknown capacitance or the impedance 28 can be calculated.

[0038] The sequencing of the measurements, the measurement of the DC current level at output 38, and the calculation of the impedance of the unknown capacitance or impedance 28 are preferably performed by a microcontroller equipped with an integrated ADC (analog-to-digital converter). In another embodiment, the mixer 35 and the low-pass filtering amplifier 37 can each be implemented within a microcontroller equipped with an ADC by connecting the ADC input directly to the output 34 of the amplifier 33 and implementing the mixer in software by selectively multiplying the ADC results by the values ​​+1 and -1, synchronizing them with the AC voltage source 24, and then low-pass filtering or integrating the resulting values ​​by the software.

[0039] The reason for the circuit being in Fig. 3 is less sensitive to the injection of a parasitic alternating current, as described in the circuit description in Fig. 5 described in more detail. In order to optimally suppress injected parasitic alternating currents, it is preferred to first perform a sweep or step sampling of the frequency of the AC voltage source 21, to detect the frequency or frequencies at which parasitic alternating currents are located, and then to set the measurement frequency of the AC voltage source 21 to a frequency at which no parasitic alternating current was detected and where there is also no subharmonic oscillation of a parasitic alternating current.

[0040] An alternative to the one in Fig. The circuit shown in 3 is the one in Fig. Circuit shown in 4. The difference to the circuit in Fig. 3 consists in the fact that the input of the capacitance-to-current or voltage converter 27 is directly connected to the output of the AC voltage source 21, and that the input of the phase shifter is connected to the output 26 of the mixer. The remaining operation of the circuit is the same as the circuit in Fig. 3 identical, except that the capacitance-current or voltage converter 27 and the unknown capacitance or impedance 28 are now connected to an unmodulated periodic common-frequency signal, and not to a modulated signal as in Fig. 3, is being supplied.

[0041] Fig. Figure 5 shows a preferred embodiment of the circuit in Fig. 3, which operates in coupling mode. Mixer 23 in Fig. 3 consists of a non-inverting amplifier 231 and an inverting amplifier 232. The AC voltage source 21 generates an AC voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and a peak amplitude of 1 V.

[0042] The low-frequency AC voltage source, using the multiplexer 233, switches one of the two amplifier outputs to node 26. Amplifiers 231 and 232 can preferably be implemented with suitably configured operational amplifiers; the multiplexer 233 can, for example, be a generally available 74HC4053 CMOS multiplexer. In this case, the output of the AC voltage source 24 is preferably a square wave with a low level of 0 V and a high level of 5 V. The operating frequency is set to 1 kHz.

[0043] The current flowing through the unknown capacitance of the impedance 28 flows into the type II current transformer 271. An example of a current transformer is the OPA860 Operational Transconductance Amplifier from Texas Instruments Incorporated. The current transformer replaces the capacitance-to-current or voltage transformer 27 in Fig. 3. The current transformer maintains its X input (node ​​272) at the same AC potential as its Y input (node ​​273), which is at ground potential. Therefore, the voltage across the unknown capacitance or impedance is equal to the voltage at node 26. The current transformer copies the current flowing into its X input to its Z output connected to node 29. The same current flows into the multiplexer 301. The multiplexer 301 and the comparator 302 replace the mixer 30 in Fig. 3. The multiplexer 301 can, for example, be a generally available CMOS multiplexer 74HC4053.

[0044] The comparator 302 converts the phase-shifted sine wave coming from the phase shifter 32 into a square wave with, for example, a low level of 0 V and a high level of 5 V, in order to appropriately drive the control input of the multiplexer 301. The comparator 302 can, for example, be a generally available comparator LM393. Components 331, 332, 333, 334, 335, 336, and 337 form an AC-coupled differential integrator with finite DC gain.

[0045] Capacitors 331 and 332 provide the AC coupling for the integrator. The current flowing from the current transformer output Z, which passes through multiplexer 301 and capacitor 331 into the inverting input of the differential integrator, is integrated and reduces the output voltage of the differential integrator. The current flowing from the current transformer output Z, which passes through multiplexer 301 and capacitor 332 into the non-inverting input of the differential integrator, is integrated and increases the output voltage of the differential integrator. Resistors 334 and 336 prevent saturation of the integrator. Preferred values ​​for capacitors 331 and 332 are, for example, 500 nF, for capacitors 333 and 335, 10 nF, and for resistors 334 and 336, 500 kΩ.

[0046] Another amplifier 33, for example with a voltage gain of 10 and a 4-pole Butterworth low-pass filter with a cutoff frequency of 1.5 kHz, is added after the input of the differential integrator. The details of such an amplifier are not described here, as they can readily be found in a standard book on filter design, for example, the "Electronic Filter Design Handbook" by Arthur B. Williams and Fred J. Taylor.

[0047] The mixer 25 in Fig. Circuit 3 consists of a non-inverting amplifier 351 and an inverting amplifier 352. The low-frequency AC voltage source 24, using the multiplexer 353, switches one of the two amplifier outputs to node 36. The amplifiers 351 and 352 can preferably be implemented with suitably configured operational amplifiers; the multiplexer 353 can, for example, be a generally available 74HC4053 CMOS multiplexer. A further low-pass filter is added to the mixer output by means of resistor 371 and capacitor 372.

[0048] The final DC level output, which responds to the current flowing through the unknown capacitance or impedance 28, is present at output 38. A parasitic AC current entering the reading electrode of the capacitive sensor (X-input 272 in Fig. 5) is essentially suppressed by the mixing effect of the mixer 301 and the low-pass filter effect of the differential integrator.

[0049] The maximum peak amplitude of the parasitic alternating current that can be injected before the measured capacitance is substantially distorted is largely defined by the current range of the current transformer 271, which is 9 mA for the aforementioned OPA860, and which is conveniently matched to the peak amplitude of 126 µA for the prior art circuit. Fig. 1 is comparable.

[0050] Fig. Figure 6 shows a preferred embodiment of the circuit in Fig. 3, which operates in charging mode. The circuit in Fig. 6 is almost identical to the circuit in Fig. 5, except that the capacitance-to-current or voltage converter 27 in Fig. 3 and the mixer 30 in Fig. 3 in Fig. 6 are implemented differently. Therefore, only the implementation in Fig. 6 of the two mentioned parts of Fig. 3 described.

[0051] Transistors 503, 504, 505, 514, 515 and current sources 502 and 512 essentially form a double-tuned mixer, similar to the mixer implemented in the ON Semiconductors MC1496 integrated circuit. A detailed description of the double-tuned mixer is provided in the datasheet "MC1496, MC1496B Balanced Modulators / Demodulators," ON Semiconductors.

[0052] The output 26 of the multiplexer 233 is connected to the base of transistor 503. The DC collector current from transistor 503 is biased with the current sink 502. The AC voltage at node 26 also appears almost completely at the emitter of transistor 503, and therefore through the unknown capacitance or impedance 28. The AC current through the unknown capacitance or impedance 28 also appears almost completely at the collector of transistor 503.

[0053] Since the voltage across the unknown capacitance or impedance 28 is known, the alternating current responds to the impedance of the unknown capacitance or impedance 28. The alternating current is fed into the transistor pair 504, 505. The bias voltage source 531 applies the bias voltage to the bases of transistors 504 and 514, and via resistor 523 also to the bases of transistors 504 and 515.

[0054] The output of phase shifter 32 is AC-coupled to the base of transistor 504 via capacitor 522. During the positive interval of the signal at the output of phase shifter 32, transistors 504 and 515 are conducting, while transistors 505 and 514 are not conducting due to their respective base voltages. During the negative interval of the signal at the output of phase shifter 32, transistors 505 and 514 are conducting, while transistors 504 and 515 are not conducting due to their respective base voltages. This results in an operation analogous to that of multiplexer 301 in Fig. 5 The alternating current from the collector of transistor 503 is led either into resistor 506 and capacitor 331 or into resistor 516 and capacitor 332.

[0055] The transistor pair on the right, transistors 514 and 515, are used to suppress the local oscillator components in the output signal resulting from the switching action of transistors 504 and 505. The current reduced by current sink 512 is therefore set to essentially the same as the current reduced by current sink 502. The DC voltage source 520 supplies power to the dual-tuned mixer. It has a DC voltage of, for example, 15 V. Suitable values ​​for capacitor 522 and resistor 523 are, for example, 100 nF and 1 kΩ, respectively, and 5 mA for each of the current sinks, as well as 1 kΩ for resistors 506 and 516. The differential output currents of the dual-tuned mixer are fed to the same AC-coupled differential integrator as in Fig. 5 led. The remaining operation of the circuit in Fig. 6 corresponds to the operation of the circuit from Fig. 5.

Claims

[1] Capacitive measuring circuit comprising: a first AC signal generator (21) configured to generate a first AC voltage signal, a second AC signal generator (24) configured to generate a second AC voltage signal, wherein the second AC voltage signal has a lower frequency than the first AC signal, a first mixer (23) for multiplicative mixing of the first AC signal and the second AC signal and for generating a modulated AC signal, a control and evaluation unit effectively coupled to an antenna electrode or a separate receiver electrode, wherein the control and evaluation unit has a current measurement circuit configured to measure current signals, wherein the current signals have an amplitude and / or phase of a current flowing in the antenna electrode or in the separate receiver electrode, wherein the control and evaluation unit is configured to determine a capacitance to be measured based on the measured current signals and to output a signal indicating the determined capacitance; wherein the current measuring circuit comprises a capacitance-to-current or voltage transformer (27) coupled via the capacitance (28) to be determined, wherein the current or voltage transformer (27) has an input and an output (29), and a second mixer (30) with a first and a second input, which is effectively coupled via its first input to the output (29) of the capacitance-to-current or voltage transformer (27), and wherein one of the first AC signal generator (21) and the first mixer (23) is effectively coupled to the capacitance-to-current or voltage converter (27) to supply the first AC voltage signal or the modulated AC voltage signal to the input of the capacitance-to-current or voltage converter (27), and the other is effectively coupled to the second input of the second mixer (30). [2] Capacitive measuring circuit according to claim 1, wherein the second input of the second mixer (30) is a local oscillator input of the second mixer (30). [3] Capacitive measuring circuit according to one of claims 1 to 2, wherein a phase shifter (32) is coupled in front of the second input of the second mixer (30) in order to couple phase-shifted versions of the first AC voltage signal or the modulated AC voltage signal to the second input of the second mixer (30). [4] Capacitive measuring circuit according to one of claims 1 to 3, wherein an output signal at the output (31) of the second mixer (30) is amplified and filtered by a bandpass filter (33) and then mixed in a third mixer (35) with the second AC voltage signal of the second AC signal generator (24).

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