Network synthesis design of microwave acoustic wave filters

Abstract

A method for designing an acoustic microwave filter in accordance with frequency response requirements, comprising: selecting an initial filter circuit structure comprising a plurality of circuit elements including at least one resonant element and at least one other reactive circuit element; selecting attenuation-free circuit response variables based on the frequency response requirements; selecting a value for each of the circuit elements based on the selected circuit response variables to generate an initial filter circuit design; transforming the at least one resonant element and the at least one other reactive circuit element of the initial filter circuit design into at least one acoustic resonator model to generate an acoustic filter circuit design; adding parasitic effects to the acoustic filter circuit design to generate a pre-optimized filter circuit design;Optimizing the pre-optimized filter circuit design to produce a final filter circuit design; and building the acoustic microwave filter based on the final filter circuit design.

Description

Field of invention

[0001] The present invention relates generally to microwave filters and in particular to acoustic wave microwave filters. Background of the invention

[0002] Frequency-selective, low-attenuation electrical signal filters for communication applications were developed beginning around 1910 for telegraphy and telephony applications, particularly for multiplexing and demultiplexing communication signal channels transmitted over long-distance lines and wireless links. Filter design methods, known as image or image parameter design methods, were developed by Bell Laboratories in the 1920s (see George A. Campbell, "Physical Theory of the Electric Wave Filter," The Bell System Technical Journal, Vol. I, No. 2 (November 1922); Otto J. Zobel, "Theory and Design of Uniform and Composite Electric Wave-Filters," The Bell System Technical Journal, Vol. II, No. 1 (January 1923)). These filter circuits used circuit elements including inductors, capacitors, and transformers.

[0003] In the 1920s, acoustic wave (AW) resonators, particularly quartz bulk acoustic wave (BAW) resonators, began to be used in some electrical signal filters. The circuit equivalent to an AW resonator has two closely spaced resonances with respect to its frequency, which are called the "resonance" frequency and the "antiresonance" frequency (see K.S. Van Dyke, "Piezo-Electric Resonator and its Equivalent Network," Proc. IRE, Vol. 16, 1928, pp. 742–764). The image filter design methods were applied to filter circuits using these quartz resonators, resulting in two AW filter circuit types: “conductor” and “grate” AW filter designs (see L. Espenschied, “Electrical Wave Filter”, US Patent No. 1,795,204; and W.P. Mason, “Electrical Wave Filters Employing Quartz Crystals as Elements”, The Bell System Technical Journal (1934)).

[0004] In the 1920s and 1930s, a different approach to the design of frequency-selective electrical signal filters for communication applications was developed, which became known as "network synthesis." Pioneering work on this new filter circuit design method was carried out by, among others, Foster and Darlington in the United States (see Ronald M. Foster, "A Reactance Theorem," Bell Syst. Tech. J., Vol. 3, 1924, pp. 259–267, and S. Darlington, "Synthesis of Reactance 4-poles which produce prescribed insertion loss characteristics," J. Math. Phys., Vol. 18, 1939, pp. 257–353) and Cauer in Germany (see W. Cauer, US Patent No. 1,989,545; 1935).

[0005] In network synthesis, after selecting an initial circuit structure that includes the circuit element types and how they are interconnected, the desired attenuation-free filter response is translated into a ratio of complex polynomials in the form of complex frequency-dependent circuit response parameters such as scattering parameters, e.g., S21 and S11. The S21 scattering parameters can be represented as follows: where N(s) is the numerator polynomial, D(s) is the denominator polynomial, the z i 's are the roots (or zeros) of the equation N(s)=0, which p iLet 's be the roots (or reflection zeros) of D(s) = 0, m is the number of transmission zeros, n is the number of reflection zeros, and K is a scale factor. (Note: For the attenuation-free case, transmission zeros are the zeros of S21 and reflection zeros are the zeros of S11. When the small but finite real attenuations are added later to the circuit design process, these zeros can become small but no longer exactly zero and correspond to the natural frequencies or resonances of the final filter.) The filter circuit element values ​​can then be exactly "synthesized" (calculated) from the ratio of the complex polynomials in the attenuation-free case. Neglecting attenuations, which are kept small in practice, the "synthesized" circuit matches the desired response function.

[0006] In the 1950s and 1960s, network synthesis was successfully applied to the design of microwave filters for communication and other applications. These new filters use high-Q (low-attenuation) electromagnetic resonators and electromagnetic couplings between these resonators as circuit elements (see George L. Matthaei et al., “Microwave Filters, Impedance-Matching Networks, and Coupling Structures”, McGraw-Hill Book Company, pp. 95–97, 438–440 (1964); and Richard J. Cameron et al., “Microwave Filters for Communication Systems: Fundamentals, Design and Applications”, Wiley-Interscience (2007)). Starting in the 1960s, network synthesis was also applied to the design of acoustic wave filters for communication and other applications (see Anatol I. Zverev, “Handbook of Filter Synthesis”, John Wiley & Sons, pp. 414–498 (1967); and Robert G. Kinsman, “Crystal Filters: Design, Manufacture, and Application”, John Wiley & Sons, p.37–105 and 117–155, (1987)). In this work, only the resonance of the acoustic wave resonator in the initial circuit structure is included. The antiresonance is treated as a parasitic effect added to the circuit after the element values ​​of the initial circuit have been calculated by the network synthesis method.

[0007] Beginning around 1992, thin-film SAW and BAW resonators were developed and their use in microwave applications (frequencies > 500 MHz) began. AW impedance element filter (IEF) designs were employed, which is an example of an Espenschied-type conductor acoustic filter design (see O. Ikata, et al., “Development of Low-Loss Bandpass Filters Using Saw Resonators for Portable Telephones”, 1992 Ultrasonics Symposium, pp. 111–115, and Ken-ya Hashimoto, “Surface Acoustic Wave Devices in Telecommunications: Modeling and Simulation”, Springer (2000), pp. 149–161). Image-designed AW-IEF bandpass filters in SAW and BAW implementations are often used for microwave filtering applications in the high-frequency (HF) front end of mobile communication devices. The frequency range of approximately 500–3500 MHz is of particular importance in the mobile communications industry.In the United States, there are a number of standard bands used for mobile communications. These bands include Band 2 (~1800–1900 MHz), Band 4 (~1700–2100 MHz), Band 5 (~800–900 MHz), Band 13 (~700–800 MHz), and Band 17 (~700–800 MHz), with other bands being developed.

[0008] The duplexer, a special type of filter, is a key component in the front end of mobile devices. Modern mobile communication devices transmit and receive simultaneously (using LTE, WCDMA, or CDMA) and utilize the same antenna. The duplexer separates the transmit signal, which can have a power of up to 0.5 watts, from the receive signal, which can be as low as picowatts. The transmit and receive signals are modulated on carriers at different frequencies, allowing the duplexer to select them. However, the duplexer must provide low insertion loss, high selectivity, a small circuit area, high power handling capacity, high linearity, and low cost.The image-designed bandpass AW-IEF filter is consistently preferred for use in a duplexer because it meets these requirements significantly better than alternatives such as the tapped-delay line (due to its higher attenuation) and the resonant single-phase unidirectional transducer (SPUDT) filter (because the narrow lines required prevent scaling to microwave frequencies), although the dual-mode SAW (DMS) (also called longitudinally coupled resonator (LCR)) filter is sometimes used as a receiver filter in a duplexer because of the balanced output and improved rejection it provides (see David Morgan, "Surface Acoustic Wave Filters With Applications to Electronic Communications and Signal Processing," Morgan, pp. 335–339, 352–354 (2007)).

[0009] Minor modifications to these traditional AW-IEF filter designs have also been considered for these applications (see U.S. Patents No. 8,026,776 and U.S. Patent No. 8,063,717), typically adding one or more circuit elements (e.g., capacitors, inductors, or an AW resonator) to the IEF design to enhance a specific circuit feature. This can be achieved if the effects of the basic AW-IEF circuit are sufficiently minor that standard computer optimization tools converge and produce an improved design compared to the optimized IEF after the additional element(s) have been added. This is a critical requirement for any circuit containing AW resonators with their closely spaced resonances and antiresonances, and therefore permits only very minor modifications to the basic AW-IEF design and function.

[0010] There is a need for improved microwave acoustic wave filters to provide enhanced performance, smaller size, and lower cost, as well as to incorporate tunability. Network synthesis offers a solution when the composite nature of the acoustic wave resonator is directly integrated into the network synthesis process – which is the subject of this invention. Summary of the invention

[0011] In accordance with the present inventions, a method for designing an acoustic microwave filter in accordance with frequency response requirements (e.g., one or more frequency-dependent return loss, insertion loss, rejection, and linearity, or a passband (e.g., in the 500–3500 MHz range) and a stopband) is provided. The method comprises selecting an initial filter circuit structure that includes a plurality of circuit elements comprising at least one resonant element (e.g., a parallel LC resonator combination of a capacitor and an inductor) and at least one other reactive circuit element (e.g., a capacitor). The initial circuit element may, for example, be a series non-resonant node circuit structure.

[0012] An optional procedure further involves selecting the structural type of each of the resonant element(s) from a surface acoustic wave (SAW) resonator, a bulk acoustic wave (BAW) resonator, a layered bulk acoustic resonator (FBAR), and a micromechanical system (MEMS) resonator. Another optional procedure further involves mapping the frequency response requirements into a normalized design space, where in this case the circuit element values ​​are normalized values ​​determined based on the mapped frequency response requirements, and mapping the normalized circuit element values ​​of the acoustic filter circuit design back into a real design space.

[0013] The procedure further involves selecting attenuation-free circuit response variables based on the frequency response requirements (e.g., in the form of a ratio between numerator polynomials defining pass zeros and denominator polynomials defining reflection zeros, multiplied by a scale factor) and selecting a value for each of the circuit elements based on the selected circuit response variables to generate an initial filter circuit design.

[0014] The method further comprises transforming the resonant element(s) and the other reactive circuit element(s) of the initial filter circuit design into at least one acoustic resonator model to generate an acoustic filter circuit design. In one embodiment, the acoustic resonator model is a Butterworth-Van Dyke (BVD) model. In this case, the other reactive circuit element(s) may include an admittance inverter in series with the parallel LC resonator combination in parallel and a non-resonant susceptance in parallel with the parallel LC resonator combination in parallel, and the parallel LC resonator combination in parallel, the admittance inverter in parallel, and the non-resonant susceptance in parallel may be transformed into one of the BVD model(s).For example, the parallel shunted LC resonator combination and the shunted admittance inverter can be transformed into a series shunted LC resonator combination, and the series shunted LC resonator combination and the shunted non-resonant susceptance can be transformed into a single BVD model. In this case, the single BVD model can be transformed into a shunted BVD model. In this embodiment, the reactive circuit element can further comprise two series admittance inverters connected via a node between the parallel shunted LC resonator combination and the shunted non-resonant susceptance, and the shunted BVD model and the two series admittance inverters can be transformed into a series BVD model and a series reactance with the series BVD model.

[0015] In one embodiment, a plurality of resonant elements, a plurality of reactive circuit elements, and a plurality of resonator models are provided. In this case, the method can further comprise dividing the initial filter circuit design into a plurality of subgroup circuit designs, each of which comprises one of the resonant elements and one or more of the plurality of reactive circuit elements, wherein, for each subgroup circuit design, the resonant element and the reactive circuit element(s) are transformed into a corresponding acoustic resonator model.

[0016] The method further involves adding parasitic effects to the acoustic filter circuit design to generate a pre-optimized filter circuit design, optimizing the pre-optimized filter circuit design to generate a final filter circuit design (e.g., by inputting the pre-optimized filter circuit design into a filter optimizer and generating the final filter circuit design), and constructing the acoustic microwave filter based on the final filter circuit design. An optional method further includes performing element removal optimization of the pre-optimized filter circuit designs to generate the final filter circuit design.

[0017] If a plurality of resonant elements are provided, the procedure may optionally include changing the order in which the plurality of resonant elements are arranged in the pre-optimized filter circuit design along a signal path to generate a plurality of filter solutions, calculating a performance parameter of each of the filter solutions, comparing the performance parameters with each other, and selecting one of the filter solutions as the pre-optimized circuit design based on the comparison of the calculated performance parameters.In one method, the final filter circuit design comprises a plurality of acoustic resonators, and the difference between the lowest and highest resonant frequencies of the plurality of acoustic resonators in the final filter circuit design is at least once, preferably at least twice, and more preferably at least three times the maximum frequency spacing of a single resonator in the plurality of acoustic resonators.

[0018] Other and further aspects and features of the invention will become apparent from reading the following detailed description of preferred embodiments, which are intended to describe but not limit the invention. Brief description of the drawings

[0019] The drawings illustrate the design and usability of preferred embodiments of the present invention, with common reference numerals indicating similar elements. To better understand how the aforementioned and other advantages and objectives of the present invention are achieved, a more detailed description of the present invention, briefly described above, will be given by reference to specific embodiments illustrated in the accompanying drawings. Understood that these drawings represent only typical embodiments of the invention and should therefore not be considered to limit its scope, the invention will now be described and explained with additional precision and detail by means of the accompanying drawings, which:

[0020] Fig. 1. A block diagram of a wireless telecommunications system;

[0021] Fig. 2 is a flowchart illustrating a network synthesis technique used to design an acoustic filter in accordance with a method of the present inventions;

[0022] Fig. Figure 3 is a schematic diagram of an in-series non-resonant node filter, which serves as an initial filter circuit structure for the network synthesis technique of Fig. 2 is used;

[0023] Fig. 4 A schematic diagram of a parallel LC resonator combination of the initial filter circuit structure of Fig. 3 is;

[0024] Fig. 5 is an equivalent circuit schematic diagram for a Butterworth-Van Dyke (BVD) acoustic resonator model;

[0025] Fig. 6 is a subgroup circuit that differs from the initial filter circuit structure (design) of Fig. 3 in accordance with the network synthesis technique of Fig. 2 was taken, with a series acoustic resonator being incorporated into the initial filter circuit design of Fig. 3 is recorded;

[0026] Fig. 7– Fig. There are 9 circuit transformations that are applied sequentially to the subgroup circuit design of Fig. 6 in accordance with the network synthesis technique of Fig. 2 were carried out;

[0027] Fig. 10 is a different subgroup circuit design, which differs from the initial filter circuit structure of Fig. 3 in accordance with the network synthesis technique of Fig. 2 was taken;

[0028] Fig. 11– Fig. There are 13 circuit transformations that are successively applied to the subgroup circuit design of Fig. 10 in accordance with the network synthesis technique of Fig. 2 were carried out, whereby an acoustic resonator was shunted into the initial filter circuit design of Fig. 3 is recorded;

[0029] Fig. Figure 14 is a schematic diagram of an acoustic filter circuit design, derived from the subgroup acoustic circuit designs of Fig. 9 and Fig. 13 in accordance with the network synthesis technique of Fig. 2 was produced;

[0030] Fig. Figure 15 is a schematic diagram of a pre-optimized filter circuit design derived from the acoustic filter circuit design of Fig. 14 in accordance with the network synthesis technique of Fig. 2 was produced;

[0031] Fig. 16 is a table that lists the element values ​​of the pre-optimized filter circuit design of Fig. 15 illustrated;

[0032] Fig. 17 an S21 frequency response plot of the pre-optimized filter circuit design of Fig. 15 is;

[0033] Fig. 18 is a schematic diagram of an optimized filter circuit design, which is obtained by inputting the pre-optimized filter circuit design into a computer-aided filter optimizer and performing an element removal technique in accordance with the network synthesis technique of Fig. 2 has been produced;

[0034] Fig. 19 is a table that lists the element values ​​of the optimized filter circuit design of Fig. 18 illustrated;

[0035] Fig. 20 an S21 frequency response plot of the optimized filter circuit design of Fig. 18 is;

[0036] Fig. 21A and Fig. 21b S11 frequency response plots of the optimized filter circuit design of Fig. 18 are;

[0037] Fig. 22– Fig. There are 24 circuit transformations that are applied sequentially to the subgroup circuit design of Fig. 10 in accordance with the network synthesis technique of Fig. 2 were carried out, whereby acoustic resonators were shunted into the resonant branches of the initial filter circuit design of Fig. 3 were recorded;

[0038] Fig. Figure 25 is a schematic diagram of an acoustic filter circuit design, derived from the subgroup acoustic circuit designs of Fig. 24 in accordance with the network synthesis technique of Fig. 2 was produced;

[0039] Fig. Figure 26 is a schematic diagram of a pre-optimized filter circuit design, derived from the acoustic filter circuit structure of Fig. 25 in accordance with the network synthesis technique of Fig. 2 was produced;

[0040] Fig. 27 is a table that lists the element values ​​of the pre-optimized filter circuit design of Fig. 26 illustrated;

[0041] Fig. 28 an S21-band 5 frequency response representation of the filter circuit design of Fig. 25 after optimization;

[0042] Fig. 29 an S21-band 8 frequency response representation of the filter circuit design of Fig. 25 after optimization;

[0043] Fig. Figure 30 is a schematic diagram of yet another pre-optimized filter circuit design, which is in accordance with the network synthesis technique of Fig. 2 was produced;

[0044] Fig. 31 an S21-band 5 frequency response representation of the filter circuit design of Fig. 30 after optimization; and

[0045] Fig. 32 an S21-band 8 frequency response representation of the filter circuit design of Fig. 30 after optimization. Detailed description of the embodiments

[0046] The present disclosure describes a network synthesis technique for designing acoustic wave (AW) microwave filters (such as surface acoustic wave (SAW), bulk acoustic wave (BAW), layer bulk acoustic resonator (FBAR), and micromechanical system (MEMS) filters). This network synthesis technique yields more powerful and / or more cost-effective AW microwave filters (i.e., at frequencies greater than 500 MHz) compared to previous AW microwave filter design methods. Such AW microwave filters can be either fixed-frequency filters and / or tunable filters (tunable with respect to frequency and / or bandwidth and / or input impedance and / or output impedance), and these can be used for a single-band filter or a multiple-bandpass filter and / or a bandstop.Such AW microwave filters are advantageous in applications with challenging electrical and / or environmental performance requirements and / or strict cost / size constraints, such as those found in the high-frequency (HF) front ends of mobile communication devices including cell phones, smartphones, laptop computers, tablet computers, etc., or the HF front ends of fixed communication devices including M2M devices, cellular base stations, satellite communication systems, etc.

[0047] Examples of AW microwave filters described herein (e.g. Fig. 28– Fig. 29) show a frequency response with a single passband and a single stopband, which is particularly useful in telecommunications duplexers where a passband with a closely spaced stopband is required. For example, and with reference to Fig. 1. A telecommunications system 10for use in a mobile communications device, a transceiver 12 include a device capable of sending and receiving wireless signals, and a controller / processor. 14 , who is capable of performing the functions of the transmitter-receiver 12 to control. The transmitter-receiver 12 generally includes a broadband antenna 16 , a duplexer 18 , which has a transmitter filter 24 and a receiver filter 26 has a transmitter 20 , the one with the antenna 16 via the transmitter filter 24 of the duplexer 18 is connected, and a receiver 22 , the one with the antenna 16 via the receiver filter 26 of the duplexer 18 is connected.

[0048] The transmitter 20 includes a high converter 28 one that is configured to receive a baseband signal from the controller / processor 14is provided to convert to a high-frequency (HF) signal, a variable gain amplifier (VGA) 30 , which is configured to amplify the RF signal, a bandpass filter 32 , which is configured to output the RF signal at an operating frequency specified by the controller / processor 14 was selected to output, and a final stage 34 , which is configured to amplify the filtered RF signal, which then passes through the transmitter filter 24 of the duplexer 18 the antenna 18 is provided.

[0049] The recipient 22 closes a bottleneck or stopband filter 36 one that is configured to prevent transmission signal interference from the RF signal input from the antenna 16 via the receiver filter 26 to reject, a low noise amplifier (LNA) 38, which is configured to amplify the RF signal from the stopband filter with relatively low noise, a tunable bandpass filter 40 , which is configured to receive the amplified RF signal when the controller / processor is activated. 14 to output the selected frequency, and a downconverter 42 , which is configured to convert the RF signal to a baseband signal 14 to convert down to a size suitable for the controller / processor 14 is provided. Alternatively, the function of rejecting transmission signal interference, which is provided by the stopband filter, can be used. 36 The power is provided by the duplexer instead. Or the final amplifier. 34 of the transmitter 20 It can be designed to reduce transmission signal interference.

[0050] It should be acknowledged that the block diagram, which is in Fig. As illustrated in Figure 1, it is functional by its very nature, and that multiple functions can be performed by just one electronic component, or that one function can be performed by multiple electronic components. For example, the upconverter 28 VGA 30 , bandpass filter 40 , downconverter 42 and the controller / processor 14 These functions are often performed by a single transceiver chip. The function of the bandpass filter 32 can within the final stage 34 and the transmitter filter 24 of the duplexer 18 be.

[0051] The exemplary network synthesis technique described herein is used to create acoustic microwave filters for the front end of the telecommunications system. 10 to design and especially for the transmitter filter 24 of the duplexer 18, although the same technique for designing acoustic microwave filters is used for the receiver filter 26 of the duplexer 18 and can be used for other RF filters.

[0052] With reference to Fig. Section 2 will present an exemplary network synthesis technique. 50 The design of an AW microwave filter will be described. First, the filter requirements are determined by using the filter (step 1). 52) established, which include frequency response requirements (including throughband, return loss, insertion loss, rejection, linearity, noise figure, input and output impedances, etc.) as well as size and cost requirements and environmental requirements such as operating temperature range, vibration, failure rate, etc. In the described embodiment, the design aims to meet the following requirements: A throughband from 1850 MHz to 1910 MHz with a maximum insertion loss requirement of 2 dB and three stopbands, a first from 1930 MHz to 1990 MHz with a minimum rejection of 44 dB, a second from 2010 MHz to 2025 MHz with a minimum rejection of 20 dB, and a third from 2110 MHz to 2155 MHz with a minimum rejection of 45 dB.

[0053] Next, the structural types of the circuit elements to be used in the AW filter are selected, for example, the structural type of the resonator (SAW, BAW, FBAR, MEMS, etc.) and the types of inductor, capacitor, and switch are selected, along with the materials to be used to manufacture these circuit elements, including the packaging and assembly technique for manufacturing the filter (step 54 In the specific example described herein, the circuit element types selected are SAW resonators and capacitors built on a substrate consisting of 42-degree XY-cut LiTaO3.

[0054] Then, an initial circuit structure such as a series-non-resonant node, or a series, or a series with cross-couplings, or a series-non-resonant node with cross-couplings, etc., is selected based on the passband(s) and / or stopband(s) obtained from the frequency response requirements (step 56 In the illustrated embodiment, the selected initial circuit structure is the series-nonresonant node structure as described in U.S. Patents Nos. 7,719,382, 7,639,101, 7,863,999, 7,924,114, 8,063,714 and U.S. Provisional Patent Application No. 61 / 802,114, entitled “Element Removal Design in Microwave Filters.” For the purposes of this description, the term “structure” shall refer to the element types and their interconnections without regard to the values ​​of the elements.

[0055] Referring to Fig. 3 includes such an embodiment of a filter circuit structure initializing in series non-resident nodes 100 generally a signal transmission path 102 , which has an entrance 104 (represented by node S) and has an output (represented by node L), a plurality of nodes 108 (represented by nodes S, 1, 2...n), which are located along the signal transmission path 102 are arranged, a plurality of resonant branches 110 , each of which is the node 108 connect to the earth, and a plurality of non-resonant branches 112 , each of which is the node 108 with the earth in their respective parallels to the resonant branches 110 connect.

[0056] The initial filter circuit structure 100 further includes a plurality of resonant elements in the parallel circuit. 114 (represented by susceptances B) R1 , B R2 ...BRn ), which are each in the resonant branches 110 are arranged, and a plurality of non-resonant elements in the parallel circuit 116 (represented by admittance inverter J) 11 , J 22 ...J nn ) in series with the resonant elements 114 The initial filter circuit structure 100 further includes a plurality of non-resonant elements in the side accord. 118 , two of which connect node S and node L to the earth (represented by susceptances B) NS or B NL ), and of which four are in the non-resonant branches 110 (represented by B) N1 , B N2 ...B Nn ) are arranged. The initial filter circuit structure 100 further includes a plurality of non-resonant elements in series 120 (represented by admittance inverter J) S1 , J 12 , J 23 ...J n-1,n , J nL), which each couple the nodes S, 1, 2...n, L together.

[0057] The initial filter circuit structure 100 A plurality of tuning elements (not shown) can be used to adjust the frequencies of the resonant elements. 114 and / or other values ​​of the non-resonant elements 120 include and an electrical controller (not shown) for tuning the initial filter circuit structure 100 to a selected narrowband within a desired frequency range by varying selected non-resonant elements 116 – 120 is configured. Therefore, the initial filter circuit structure is 100 useful for network synthesis of reconfigurable bandpass filters, provided that the resonant elements with a high Q factor are used to control the susceptance B RTo achieve these values, a parallel LC resonator combination is used. This means a resonant circuit as in... Fig. 4 shown, well described.

[0058] The resonant elements with a high Q factor 114 results are better when using a Butterworth-Van Dyke (BVD) model 122 described that in Fig. 5 is illustrated. BVD models 122 SAW resonators can also be described, which can be fabricated by arranging interdigital transducers (IDTs) on a piezoelectric substrate such as crystalline quartz, lithium niobate (LiNbO3), lithium tantalate (LiTaO3) crystals, or BAW (including FBAR) resonators fabricated in materials such as quartz or aluminum nitride, or MEMS resonators. The BVD model 112 includes a movement capacity C m 124 , a static capacitance C0 126 and a motion inductance L m 128one. The movement capacity C m 124 and the motion inductance L m 128 can arise from the interactions of electrical and acoustic behavior and can therefore be considered the motion arm of the BVD model. 122 be designated as the static capacitance C0 126 can result from the electrical behavior of the structure alone (conductors, dielectrics and gaps) and can therefore be considered the static (non-movement) capacitance of the BVD model. 122 The parameters are defined as follows: where ω R and ω A the respective resonance and antiresonance frequencies for any given acoustic resonator can be and gamma γ can depend on a material property which can be further defined by:

[0059] Typical γ values ​​can range from approximately 12 to approximately 18 for 42-degree XY cut LiTaO3. The frequency separation of an acoustic resonator refers to the difference between its resonant frequency and its antiresonant frequency. The percentage separation of an acoustic wave resonator is the percentage frequency separation between its resonant frequency and its antiresonant frequency and can be calculated as follows: where γ is the ratio of the static to the motional capacity of the resonator (equation [4]), as determined by the material properties of the piezoelectric material and modified by the geometry of the device.

[0060] The resonant frequency ω R In an acoustic resonator, this is the frequency at which the magnitude of the impedance reaches a local minimum and the phase of the impedance passes through zero. The antiresonance frequency ω AIn the context of an acoustic resonator, this is the frequency where the magnitude of the impedance reaches a local maximum and the phase of the impedance passes through zero.

[0061] It can be understood from equation [1] that the resonance frequency of each of the acoustic resonators depends on the movement arm of the BVD model. 122 will depend on, while the filter characteristics (e.g., the bandwidth) are strongly influenced by γ in equation [2]. The quality factor (Q) for an acoustic resonator 122The Q factor can be an important quality factor in acoustic filter design, relating to the damping of the element within the filter. The Q of a circuit element represents the ratio of energy stored per cycle to energy dissipated per cycle. The Q factor models the true damping in any acoustic resonator, and generally, more than one Q factor may be required to describe the damping in an acoustic resonator. Q factors can be defined for the filter examples as follows. The motional capacity C m 124 can have an associated Q, which is defined as Q cm = 10 8 ; the static capacitance C0 12 can have an associated Q, which is defined as Q c0 = 200; and the motion inductance L m 128 can have an associated Q, which is defined as Q Lm= 1000. (For simplicity, the damping in the motion resonance is summarized in the motion inductance, and the motion capacitance is considered essentially undamped.) Circuit designers typically characterize SAW resonators by their resonant frequency ω. R , the static capacitance C0, gamma γ and the quality factor QL m For commercial applications, QL m Approximately 1000 for SAW resonators and approximately 3000 for BAW resonators.

[0062] Referring again to Fig. 2. The frequency response requirements are then mapped into a normalized design space (step 2). 58). Mapping can be performed using a suitable algorithm such as a square root / quadratic mapping technique (see George L. Matthaei, “Microwave Filters, Impedance-Matching Networks, and Coupling Structures”, McGraw-Hill Book Company, pp. 95–97, 438–440 (1964)) or a logarithmic / exponential mapping technique, which is more suitable for acoustic wave resonators.

[0063] An attractive logarithmic mapping technique uses the following equations: where 2πω p the geometric center frequency of the transmission band or stopband is , 2πω is the real frequency, Ω is the mapped frequency, γ is the ratio of the static to the motional capacitance of the resonator and Ω R is the depicted resonant frequency of the resonator and Ω A is the depicted antiresonance frequency of the resonator.

[0064] Next, attenuation-free circuit response variables are provided in the form of a ratio of numerator polynomials defining pass zeros and denominator polynomials defining reflection zeros, multiplied by a scale factor as provided in equation [1] (step 60 In general, the total number of transmission zeros can be greater than or equal to the total number of reflection zeros, and often one or more reflection zeros will lie outside any transmission band of the filter.

[0065] Next, the depicted and normalized circuit element values ​​are used for the initial filter circuit structure. 100 starting from these polynomials using a coupling matrix or parameter extraction methods or equivalent circuit synthesis techniques (step 62) calculated to generate an initial lossless circuit design. For the purposes of this description, a “circuit design” shall refer to the circuit structure taking into account the values ​​of the elements that constitute the circuit structure.

[0066] Next, equivalent circuit transformations can be performed to reduce the number of circuit elements or to change the type of circuit elements, the size of the circuit, or the feasibility of individual circuit elements to create an acoustic filter circuit design (step 64These transformations do not substantially alter the response of the initial zero-loss circuit design and can employ equivalent circuit transformations such as equating a J-inverter with an equivalent capacitive or inductive PI or T-network. For example, a shunt / two-J-inverter resonator can be transformed into a single series resonator;A series resonator / two J-inverter combination can be transformed into a single shunt resonator; multiple parallel capacitors can be combined into a single capacitor; or, to eliminate capacitors in any other way, negative capacitors can be removed by combining them with positive parallel capacitors to give a single positive capacitor; multiple series inductors can be combined into a single inductor; or, to eliminate inductors in any other way, negative inductors can be removed by combining them with positive series inductors to give a single positive inductor; or other equivalent circuit transformations can be performed to obtain a zero-loss circuit that can have the target circuit response but be smaller, less expensive, and / or more feasible than the initial zero-loss circuit design.

[0067] Significantly, although the acoustic resonance elements B R best through the in Fig. 5 illustrated BVD models 122 being described is a challenge, as the BVD model 122 due to its additional static capacitance C0, it cannot be directly incorporated into the LC equivalent initial filter circuit design. 100 can be integrated, which is in Fig. Figure 4 illustrates this. Therefore, a specific type of circuit transformation involves transforming the initial filter circuit design. 100 into a suitable structure, into which an acoustic resonator model and in this case a BVD model is incorporated. 122 can be integrated. This circuit transformation is best accomplished by modifying the initial filter circuit design. 100 into several subgroups equal to the number of resonating elements 114is divided. Each subgroup includes circuit elements connected to each node with which a resonant branch is connected. 110 and a non-resonant branch 112 are coupled. The nature of each subgroup will depend on whether a shunted or series acoustic resonator is desired.

[0068] For example, in a transformation technique that incorporates a series acoustic resonator into the initial filter circuit design 100 integrated, a special subgroup circuit design, a resonant element 114 (Susceptance B R ), which is from the respective node 108 connected to the earth, a non-resonant element 116 (Admittance inverter J), which is in series with the resonant element 114 coupled, a non-resonant element 118 (Susceptance B N ), which is from the respective node 108coupled to the Earth, parallel to the resonant element 114 (Susceptance B R ) and two non-resonant elements 120 (Admittance inverter J), which is in series with the respective node 108 are connected, one. For example, and as in Fig. Figure 6 illustrates a subgroup 130a the knot 1 one and therefore the resonant element B R1 from the respective node 108 The admittance inverter element J is connected to the Earth. 11 in series with the resonant element B R1 coupled, the non-resonant element B N1 from the respective node 108 out with the earth parallel with the resonant element B R1 connected, and two admittance inverters J S1 and J 12 are in series with the respective node 108 tied together.

[0069] As in Fig. As shown in 7, the admittance inverter J 11through a capacitive PI network (capacitors –C 11 , C 11 , and –C 11 ) replaced, and the resonating element B1 R is replaced by a parallel LC resonator combination of an inductor (coil L). R1 ) and a capacitance (capacitor C) R1 The circuit substructure 132 , which are from the PI network, which consists of capacitors -C 11 , C 11 , and –C 11 and the parallel LC resonator combination of the coil L R1 and of capacitor C R1 consists of, can be combined into a series LC resonator combination 134 from an inductor (coil L) R1’ ) and a capacitance (capacitor C) R1’ ) can be transformed. Significantly, this LC combination can 134 through the series resonance section of a BVD model 122 to be implemented so that they are better integrated into the circuit substructure 132 can be integrated.

[0070] To the BVD model 122 into the circuit substructure 132 To integrate, the static capacity C0 of the BVD model must be considered. 122 be accommodated. This can be done, as in Fig. As shown in Figure 8, this can be achieved by replacing the parallel susceptance B1. N through a capacity (C0 R1’ and susceptibility B N1’ ). C0 R1’ represents the static capacity of the BVD model 122 and B N1’ is given by the relationship B N1 – ω(C0 R1 The susceptance B N1’ , two series admittance inverters J S1 and J 12 and the acoustic resonator 122 In the parallel circuit, then, as in Fig. 9 illustrates, in a series acoustic resonator 122a and a series reactance 136 (denotes X1) will be transformed.

[0071] In a transformation technique that integrates an acoustic resonator in parallel into the initial filter circuit design100 Integrated, a special sub-circuit design includes a resonant element. 114 (Susceptance B R ), which is from the respective node 108 connected to the earth, a non-resonant element 116 (Admittance inverter J), which is in series with the resonant element 114 is coupled, and a non-resonant element 118 (Susceptance B N ), which is from the respective node 108 coupled to the Earth, parallel to the resonant element 114 (Susceptance B R ) one. For example, and as in Fig. 10 illustrates a subgroup 130b the knot 2 one and therefore the resonant element B R2 from the respective node 108 When connected to the earth, the admittance inverter element J2 is in series with the resonant element B. R2 coupled and is the non-resonant element B N2 from the respective node108 out with the earth parallel with the resonant element B R2 tied together.

[0072] As in Fig. Figure 11 shows the admittance inverter J 22 through a capacitive PI network (capacitors –C 22 , C 22 , and –C2) replaced, and the resonating element B R2 is replaced by a parallel LC resonator combination of an inductor (coil L). R2 ) and a capacitance (capacitor C) R2 The circuit substructure 132 , which are from the PI network, which consists of capacitors -C 22 , C 22 , and –C 22 and the parallel LC resonator combination of the coil L R2 and of capacitor C R2 consists of, can be combined into a series LC resonator combination 134 from an inductor (coil L) R2’ ) and a capacitance (capacitor C) R2’ ) can be transformed. Significantly, this LC combination can 134through the series resonance section of a BVD model 122 to be implemented so that they are better integrated into the circuit substructure 132 can be integrated.

[0073] To the BVD model 122 into the circuit substructure 132 To integrate, the static capacity C0 of the BVD model must be considered. 122 be accommodated. This can be done, as in Fig. 12 shown, can be achieved by replacing the parallel susceptance B 2N through a capacity (C0 R1’ and susceptibility B N1’ ). C0 R2’ represents the static capacity of the BVD model 122 , and B N2’ is given by the relationship B N2 – ω(C0 R2 Thus, as in Fig. 13 shows an acoustic resonator in a parallel circuit 122b to be realized.

[0074] It can be seen that the initial filter circuit design 100 into two alternating subgroups130a and 130b can be subdivided in such a way that a filter circuit design can be created that uses alternating series acoustic resonators 122a and resonators 122b in a parallel circuit. For example, an initial filter circuit design 100 with nine resonators B R into an acoustic filter circuit structure 150a are transformed, which, as in Fig. 14 illustrated, five series acoustic resonators 122a and four acoustic resonators in parallel 122b has, which are arranged in an alternating manner.

[0075] Although the circuit transformation step is described as being performed on the initial filter circuit design (i.e., after calculating the mapped and normalized circuit element values), it should be acknowledged that the circuit transformation step can be performed on the initial filter circuit structure (i.e., before calculating the mapping and normalized circuit element values) to generate an acoustic filter circuit structure, in which case the mapped and normalized circuit element values ​​for the acoustic filter circuit structure can be calculated to generate an acoustic filter circuit design.

[0076] Back to Fig. 2. Referring to the circuit element values ​​of the acoustic filter circuit design 150aIn accordance with the inverse of the mapping technique initially used to map the frequency response requirements into the normalized design space, mapped back into the real design space (i.e., attenuation-free circuit elements (L's and C's) with real frequencies) (step 66 If, for example, the logarithmic mapping technique of equation [5] was used to map the frequency response requirements into the normalized space, then the following logarithmic back-mapping equation can be used to map the normalized circuit element values ​​back into the real design space: ω = ω p (1 + 1 γ ) Ω / 2 [7]

[0077] Remarkably, any B-value can be implemented by either an L or a C, depending on the sign of B. Mapping the normalized circuit values ​​back yields the realized circuit, which is shown in Fig. Figure 15 shows, along with the values ​​of the resonance frequencies ω R and static capacitances C0 for each resonator and, as in Fig. Figure 16 shows the capacitances and inductances of the capacitors and coils, which, when simulated, are in the Fig. The frequency response shown in Figure 17 resulted. (Note: Inductor L1 and capacitor C1 are added at the end of the synthesis by pole extraction to provide equal input and output impedances to the network.)

[0078] Next, parasitic effects on the acoustic filter circuit design will be considered. 150a using an electromagnetic simulator such as the Sonnet ® Software is added, and busbar (connection) attenuation is added to arrive at a pre-optimized filter circuit design (step 68The damping of the acoustic resonators can be incorporated by assigning a Q-factor to the respective circuit elements. In this embodiment, the motion capacitance C m 124 an assigned Q defined as Q cm = 10 8 , the static capacitance C0 126 has an associated Q defined as Q c0 = 200 and the motion inductance L m 128 has an associated Q defined as Q Lm = 1000. The remaining coils have an assigned Q defined as Q u = 60, and the remaining capacitors have an assigned Q defined as Q u = 200. A busbar (connection) resistor of R S A resistance of 0.5 ohms is also added for each acoustic resonator.

[0079] The pre-optimized filter circuit design is then fed into a computer-aided filter optimizer to generate a final filter circuit design (step 1). 70 In an optional procedure, an element removal optimization (ERO) technique is implemented during optimization, removing unnecessary or “vanishing” filter circuit elements or reducing them to simpler circuit elements, as described in Fig. The final filter circuit design is illustrated in Figure 18. The ERO technique is described in provisional US patent application serial number 61 / 802,114 entitled "Element Removal Design in Microwave Filters." The optimization and ERO technique resulted in resonant frequencies ω R and static capacitances C0 for each resonator and capacitances for the capacitors as in Fig. 19 showed what was shown in the simulation in the Fig. The resulting frequency response of 20 illustrated parameters met the target frequency response requirements.

[0080] Remarkably, the multi-band filters, which are in accordance with the in Fig. 2 illustrated network synthesis techniques have been designed, resonances and resonators with resonance frequencies that span a relatively large range, in contrast to microwave acoustic filters designed according to prior art image design techniques and simple extensions thereof.

[0081] For example, one measure against which the range of resonant frequencies of a filter or its resonators can be compared is the frequency spacing of the resonator in the filter with the highest resonant frequency. For a 42-degree XY-cut LiTaO3 substrate, γ is greater than approximately 12. Any parasitic capacitance from the acoustic resonator realization can increase γ and thus decrease the percentage spacing, while parasitic inductance can effectively decrease γ. In this example, for γ = 12, the percentage spacing is 4.0833%, and therefore the spacing of the resonator with the highest resonant frequency is approximately 88.1 MHz (i.e., a resonant frequency of 2151.57 MHz times the percentage spacing of 4.0833%). Another measure against which the range of resonant frequencies of a filter or its resonators can be compared is the mean frequency spacing of its resonators, in this case 77.32 MHz.

[0082] In contrast to the frequency spacing of an acoustic resonator, the "frequency difference" between two acoustic resonators means the absolute frequency difference between the resonance frequencies of the two resonators, and the frequency difference between two resonances of a filter is the absolute frequency difference between the two resonances. Fig. 21(a) and Fig. Figure 21(b) shows the return loss (S11) of the filter, which is in Fig. 18– Fig. 19 is defined. Return loss minima correspond to resonances of the filter circuit and also correspond to reflection zeros of the initial filter circuit design. Fig. Figure 21(a) shows the resonance of the filter, which is primarily responsible for the formation of the filter's passband, N1 to N7. The frequency difference between the in Fig. The highest and lowest resonances shown in 21(a) are 102 MHz, or approximately 1.32 times the mean frequency difference of the resonators. In addition, the frequency difference between the highest and lowest resonances of the combined Fig. 21(a) and Fig. 21(b) 349 MHz (2173–1824 MHz) or approximately 4.51 times the mean frequency spacing of the resonators, while the frequency difference between the highest and lowest resonance in the filter is 459.37 MHz (2151.57–1892.2 MHz) or approximately 5.94 times the mean frequency spacing of the resonators.

[0083] It is therefore expected that the difference between the lowest resonant frequency and the highest resonant frequency of the through-band resonances in the final filter circuit design will be at least 1.25 times the mean distance between the resonators.

[0084] It is expected that multiband filters, which comply with the in Fig. 2 illustrated network synthesis techniques are designed, resonators as well as resonances corresponding to the reflection zeros will be located relatively far away from the transmission band, in contrast to filters designed in accordance with prior art image design techniques, where the resonators and resonances corresponding to reflection zeros are restricted to the transmission band or very close to it.

[0085] Resonances corresponding to reflection zeros occur particularly at frequencies where the local return loss (and / or S11) minima and the local insertion loss (and / or S21) maxima coincide within less than approximately five percent of the maximum frequency separation—less than approximately 4.405 MHz for this example. Alternatively, resonances corresponding to reflection zeros occur at local minima and at local maxima of the delay (or retardation) of S11. As can be seen from Fig. As can be seen in Figure 21b, some resonances corresponding to reflection zeros (especially the reflection zeros corresponding to the markers N1, N2, and N6–N9) are located outside and far from the through-band (1850 MHz to 1910 MHz). The frequency difference between a resonance corresponding to a reflection zero and the nearest edge of the through-band can be greater than once, perhaps greater than 1.25 times, and perhaps greater than twice the maximum frequency spacing (approximately 88.1 MHz in this example). In this particular example, the reflection zeros are located up to 3.40 times the mean frequency spacing (77.32 MHz) from the edge of the through-band. Relative to the transmission bandwidth (60 MHz), the reflection zeros N1, N2 are 43.33% and 28.33% below the lower edge of the transmission band, and the reflection zeros N6, N7 are 13.33% and 26.67% above the upper edge of the transmission band.The reflection zeros N1, N2, N6, and N7 are adjacent to each other. The reflection zeros N8 and N9, which are not adjacent to the passband zeros N1, N2, N6, and N7, are located 311.67% and 438.33% above the upper edge of the passband, respectively. The insertion loss of the final filter circuit design is preferably less than 3 dB and more preferably less than 2 dB.

[0086] Referring back to Fig. 2. Once the final filter circuit design is achieved, a real microwave filter will be created based on the final filter circuit design (step 2). 72 Preferably, the circuit element values ​​of the actual microwave filter will match the corresponding circuit element values ​​of the final filter circuit design.

[0087] Remarkably, a survey of different frequency responses can be analyzed and applied at various points in the network synthesis technique. 50 They can be compared. In one embodiment, a survey of different frequency responses can be analyzed and compared based on different versions of the acoustic filter circuit design. 150a , that at step 68 The generated data is compared to arrive at a pre-optimized acoustic filter circuit design, which is then entered into the computer-aided filter optimizer to be used at step 1. 70to generate the final filter circuit design. For example, different acoustic resonator frequency arrangements can be implemented between the input and output. In particular, the order in which the acoustic resonators are arranged along the signal transmission path can be changed to generate multiple filter solutions. One or more performance parameters can be calculated for each of the filter solutions, the performance parameter(s) for the different filter solutions can be compared, and the best filter solution (and therefore resonator order) can be selected based on this comparison. This data collection method can address all possible permutations of the acoustic resonator frequency order in the actual filter circuit design. The performance parameters can be, for example,Either at a specific frequency or at several frequencies, one or more parameters such as insertion loss, return loss, group delay, node voltages, and branch currents are measured to evaluate each circuit's response against the desired performance characteristics in the filter requirement. This measurement can yield quantitative or qualitative performance evaluation scores that indicate how well a specific circuit performs against the filter requirement.

[0088] In other embodiments, the elicitation method can also address all realizable values ​​of the static capacitances C0 of the resonators, all permutations of positive (inductive) or negative (capacitive) values ​​(parities) of J-inverters, and all other parameters that can be varied in the dampingless design, which, while not altering their response function, can modify the response of a realizable low-damping circuit. Further details discussing an elicitation method that rearranges resonant frequencies are disclosed in U.S. Patent No. 7,924,114.

[0089] Although the filter requirements in this embodiment have been described as defining fixed passbands and stopbands, it should be noted that the filter requirements can define multiple reconfigurable passbands and / or stopbands. For example, in one embodiment, the design can be reconfigurable between two states: a first state (called Band 5) that passes frequencies between 824 MHz and 849 MHz with less than 3.5 dB insertion loss and rejects frequencies between 869 MHz and 894 MHz with at least 40 dB, and a second state (called Band 8) that passes frequencies between 880 MHz and 915 MHz with less than 3.5 dB insertion loss and rejects frequencies between 925 MHz and 960 MHz with at least 40 dB (step 1). 52). The circuit element type is selected as SAW resonators built on 15-degree Y-cut LiTaO3 substrates and capacitors integrated on the 15-degree Y-cut LiTaO3 substrate (step 54 ).

[0090] Then the in Fig. 3 Illustrated initial filter circuit structure 100 selected on the basis of the passband(s) and / or stopband(s) obtained from the frequency response requirements (step 56 In this case, the number of resonators is six. The frequency requirements are then mapped into normalized space (step ). 58 ), a zero-damping circuit response is selected in the form of a polynomial ratio (step 60 ) and the depicted and normalized circuit element values ​​in the initial filter circuit structure 100These polynomials are then used to calculate an initial filter circuit design using a coupling matrix, parameter extraction techniques, or equivalent circuit synthesis techniques (step 1). 62 ).

[0091] Next, equivalent circuit transformations will be performed on the initial filter circuit design. 100 carried out to accommodate acoustic resonators (step 64 In the same way as described above, the circuit transformation divides the initial filter circuit design. 100 into several subgroup circuit designs, equal to the number of resonating elements 114 (in this case six) are, which results in six acoustic resonators in parallel.

[0092] In a transformation technique that integrates an acoustic resonator in parallel into the initial filter circuit design 100 integrated, the in Fig. 6 illustrated subgroup 130 can be transformed by replacing the admittance inverter J S1 through a capacitive PI network (capacitors –C S1 , C S1 and –C S1 ), the admittance inverter J 12 through a capacitive PI network (capacitors –C 12 , C 12 and –C 12 ), the admittance inverter J 11 through a capacitive PI network (capacitors –C 11 , C 11 and –C 11 ) and the resonating element B1 R through a parallel LC resonator combination of an inductor (coil L) R1 ) and a capacitance (capacitor C) R1 ), as in Fig. 22 is illustrated. In the same way that above refers to Fig. As described in section 7, the circuit substructure can be 132 , which is represented by the PI network, consisting of capacitors –C 11 , C 11 and –C 11and the parallel LC resonator combination of the coil L R1 and of capacitor C R1 consists of a series LC combination 134 from an inductor (coil L) R1’ ) and a capacitance (capacitor C) R1’ ) are transformed. To determine the static capacity C0 of the BVD model 122 To accommodate the three adjacent parallel capacitors and inductors (–C S1 , –C 12 and B1 N ) by a capacity (C0 R1’ and susceptibility B1 N’ ), as in Fig. 23 shown, replaced. C0 R1’ represents the static capacity of the BVD model 122 and B N1’ is given by the relationship B N1 – ω(C S1 + C 12 + CO R1 Thus, an acoustic resonator can be operated in a parallel circuit. 122 to be realized, as he is in Fig. 24 is illustrated. The other subgroups 130of the initial filter circuit design 100 can be transformed in the same way to form an acoustic filter circuit structure 150b to reach the, as in Fig. 25 illustrated, six acoustic resonators 122 in the parallel loop.

[0093] The circuit elements of the acoustic filter circuit structure 150b are then mapped back into real space (step 66 ) and parasitic effects become part of the acoustic filter circuit structure 150b added to arrive at the pre-optimized circuit design (step 68 As discussed above, the damping of the circuit elements can be included by assigning a Q-factor to the respective circuit elements. In this embodiment, the motion capacitance C m an assigned Q defined as Q cm = 10 8 The static capacity C0 has an associated Q defined as Q c0= 140 and the motion inductance L m has an associated Q defined as Q Lm = 3000. The remaining coils have an assigned Q defined as Q u = 200, and the remaining capacitors have an assigned Q defined as Q u = 200. A busbar (connection) resistor of R S A resistance of 0.5 ohms is also added for each acoustic resonator. In this embodiment, a switching parasitic load of 3 pF / (mm gate width) and 1.0 ohm*(mm gate width) is also added.

[0094] Next, the pre-optimized filter circuit design is fed into a computer filter optimizer using the optional ERO technique to generate a final filter circuit design (step 70 Before optimization, switches are added to each branch where the impedance between the two bands differs, so that, as in Fig. Figure 26 illustrates how a single circuit is generated from the two separate designs to be optimized. The gate width of each switch, the value of an inductor or capacitor (if required), and the circuit configuration of the branch are selected such that the impedance of a given branch will be the required Band 5 impedance in one switching state and the required Band 8 impedance in the other switching state. The ERO technique is then repeated on the combined circuit. The optimization procedure yields the resonant frequencies ω R and the static capacitances C0 for each resonator and, as in Fig. Figure 27 shows the capacitances and inductances of the capacitors and coils that appear in the simulation in the frequency response for band 5. Fig. 28 is illustrated, and the frequency response for band 8, which is in Fig. The results were illustrated in 29.

[0095] As previously discussed, a survey of different frequency responses can be analyzed and applied at various points in the network synthesis technique. 50 They can be compared. In one embodiment, a survey of different frequency responses can be analyzed and compared based on different versions of the acoustic filter circuit design. 150a , that at step 68 The generated data is compared to arrive at a pre-optimized acoustic filter circuit design, which is then entered into the computer-aided filter optimizer to be used at step 1. 70to generate the final filter circuit design. For example, pairs of circuits (one band 5 and one band 8) are created with every possible resonator frequency order, every possible J-inverter parity (inductor or capacitive), and a selection of static capacitance C0 values ​​for the resonators. This survey method uses all possible permutations of the resonator frequency order, all possible parities, and a range of practical static capacitance C0 values ​​of 0.95, 1.9, 3.8, and 7.6 pF to calculate the insertion loss at the center frequency of the passband for each design. A pair of designs (one band 5 and one band 8 with the same resonator order and the same static capacitance C0 values) can then be selected.

[0096] Although the preceding embodiment includes passbands and / or stopbands that are dynamically reconfigurable, it should be noted that a filter constructed in accordance with network synthesis techniques may have fixed passbands and / or stopbands that are reconfigurable before the final completion of the filter, but are fixed after completion. For example, in a Fig. In the illustrated embodiment 30, a zero-loss circuit model is implemented to create a filter that has either a passband centered at 836.5 MHz (band 5) or at 897.5 MHz (band 8). This zero-loss circuit was created by transforming the initial filter circuit design. 100 , which in Fig. Figure 3 illustrates the use of three SAW resonators.

[0097] In a transformation technique that connects three acoustic resonators in parallel to the initial filter circuit design 100 integrated, can be incorporated into the Fig. 10– Fig. 13 illustrated transformation techniques can be used to create circuit subgroups (each containing a resonant element). 114 (Susceptance B R ), which is from the respective node 108 connected to the earth, a non-resonant element 116 (Admittance inverter J), which is in series with the resonant element 114 is connected, and a non-resonant element 118 (Susceptance B N ), which corresponds to the respective node 108 with the earth parallel to the resonant element 114 (Susceptance B R) connected, including) to transform into three acoustic resonators in shunt. The circuit element type is selected as SAW resonators built on 42-degree Y-cut LiTaO3 substrates, and capacitors integrated on the 42-degree Y-cut LiTaO3 substrate.

[0098] The filter can be reconfigured before completion by changing the values ​​of the row elements between the resonators (in this case C). S1 , C 12 , C 23 , C 3L ) and the non-resonant shunt elements (in this case L) S , L1, L2, L3, L L The filter can then be constructed using either the values ​​of the non-resonant elements for band 5 or the values ​​of the non-resonant elements for band 8. The optimization procedure yields the static capacitances C0 for each resonator and the capacitances and inductances of the capacitors and inductors, as shown in Fig. Figure 30 shows the frequency response for band 5 in the simulation. Fig. 31 is illustrated, and the frequency response for band 8, which is in Fig. The results were illustrated in section 32.

[0099] Although various specific embodiments of the present invention have been shown and described, it should be understood that the above discussion is not intended to limit the present invention to these embodiments. It will be obvious to those skilled in the art that various changes and modifications can be made without departing from the spirit and scope of the present invention. For example, the present invention certainly has applications beyond single-input and single-output filters, and specific embodiments of the present invention can be used to form duplexers, multiplexers, channelizers, responsive switches, etc., where low-loss selective circuits can be used.The present invention therefore aims to cover alternatives, modifications and equivalents that fall within the spirit and scope of the present invention as defined in the claims.

Claims

[1] A method for designing an acoustic microwave filter in accordance with frequency response requirements, comprising: Selecting an initial filter circuit structure comprising a plurality of circuit elements, including at least one resonant element and at least one other reactive circuit element; Selecting attenuation-free circuit response variables based on frequency response requirements; Selecting a value for each of the circuit elements based on the selected circuit response variables to generate an initial filter circuit design; Transforming the at least one resonant element and the at least one other reactive circuit element of the initial filter circuit design into at least one acoustic resonator model to generate an acoustic filter circuit design; Adding parasitic effects to the acoustic filter circuit design to create a pre-optimized filter circuit design; Optimizing the pre-optimized filter circuit design to produce a final filter circuit design; and Building the acoustic microwave filter based on the final filter circuit design. [2] The method of claim 1, wherein the frequency requirements comprise one or more frequency-dependent return loss, insertion loss, rejection and linearity. [3] The method of claim 1, wherein the frequency response requirements comprise a throughband in the 500–3500 MHz range. [4] The method of claim 1, wherein the frequency response requirements comprise a passband and a stopband. [5] The method of claim 1, wherein each of the at least one resonator comprises a parallel LC resonator combination of a capacitor and an inductor. [6] The method of claim 1, wherein the at least one other reactive circuit element comprises a capacitor. [7] The method of claim 1, wherein the initial filter circuit structure is a series non-resonant node filter circuit structure. [8] The method of claim 1, wherein the attenuation-free circuit response variables are in the form of a ratio between numerator polynomials defining transmission zeros and denominator polynomials defining reflection zeros, multiplied by a scale factor. [9] The method of claim 8, wherein the total number of transmission zeros is equal to or greater than the total number of reflection zeros. [10] The method of claim 1, wherein each of the at least one acoustic resonator model is a Butterworth-Van-Dyke (BVD) model. [11] The method of claim 10, wherein the at least one resonator comprises a parallel shunted LC resonator combination, the at least one other reactive circuit element comprises a shunted admittance inverter in series with the parallel shunted LC resonator combination and a non-resonant susceptance in parallel with the parallel shunted LC resonator combination, and wherein the parallel shunted LC resonator combination, the shunted admittance inverter and the shunted non-resonant susceptance are transformed into one of the at least one BVD model. [12] The method of claim 11, wherein the parallel LC resonator combination in shunt and the admittance inverter in shunt are transformed into a series LC resonator combination in shunt and the series LC resonator combination in shunt and the non-resonant susceptance in shunt are transformed into the one BVD model. [13] The method of claim 11, wherein the BVD model is a BVD model in a side circuit. [14] The method of claim 13, wherein the at least one reactive circuit element further comprises two admittance inverters in series, which are connected to a node between the parallel LC resonator combination in shunt and the non-resonant susceptance in shunt, and wherein the BVD model in shunt and the two admittance inverters in series are transformed into a BVD model in series and a reactance in series with the BVD model in series. [15] The method of claim 1, wherein the at least one resonant element comprises a plurality of resonant elements, the at least one other reactive element comprises a plurality of reactive circuit elements, and the at least one acoustic resonator model comprises a plurality of resonator models. [16] The method of claim 15, further comprising dividing the initial filter circuit design into a plurality of subgroup circuit designs, each of which has one of the resonant elements and one or more of the plurality of reactive circuit elements, wherein for each subgroup circuit design the resonant element and the one or more reactive circuit element(s) are transformed into a corresponding acoustic resonator model. [17] The method of claim 1, further comprising selecting the structural type of each of the at least one resonant element from a surface acoustic wave (SAW) resonator, a bulk acoustic wave (BAW) resonator, a layered bulk acoustic resonator (FBAR) and a micromechanical system (MEMS) resonator. [18] The method of claim 1, further comprising: Mapping the frequency response requirements into a normalized design space, where the circuit element values ​​are normalized values ​​determined based on the mapped frequency response requirements; and Mapping the normalized circuit element values ​​of the acoustic filter circuit design back into a real design space. [19] The method of claim 1, wherein the at least one resonance element comprises a plurality of resonance elements. [20] The method of claim 19, Selecting the order in which the majority of resonant elements in the pre-optimized filter circuit design are arranged along a signal path to generate a plurality of filter solutions; Calculating a performance parameter for each of the filter solutions; Comparing the performance parameters with each other; and Selecting one of the filter solutions as the pre-optimized circuit design based on a comparison of the calculated performance parameters. [21] The method of claim 1, further comprising performing an element removal optimization of the pre-optimized filter circuit design to generate the final filter circuit design. [22] The method of claim 1, wherein the final filter circuit design comprises a plurality of acoustic resonators and wherein the difference between the lowest resonant frequency and the highest resonant frequency of the plurality of acoustic resonators in the final filter circuit design is at least once the maximum frequency spacing of a single resonator in the plurality of acoustic resonators. [23] The method of claim 22, wherein the difference between the lowest resonant frequency and the highest resonant frequency of a plurality of resonators in the final filter circuit design is at least twice the maximum frequency spacing of a single resonator in the plurality of resonators. [24] The method of claim 22, wherein the difference between the lowest resonant frequency and the highest resonant frequency of a plurality of resonators in the final filter circuit design is at least three times the maximum frequency spacing of a single resonator in the plurality of resonators. [25] The method of claim 1, wherein the optimization of the pre-optimized filter circuit design comprises inputting the pre-optimized filter circuit design into a filter optimizer to generate the final filter circuit design.