Pulse radar using hybrid pulse and frequency modulation approach

Abstract

Pulse radar by hybrid pulse approach and frequency modulation Radar device comprising: - a transmission channel (1) configured to generate a radar signal (IE); and - a reception channel (2) configured to receive echoes (SE) and demodulate them synchronously so as to extract time-of-flight information therefrom; characterized in that the transmission channel (1) and the reception channel (2) comprise: - a shared local oscillator (1204), having a variable frequency as a function of a control signal; - a generator (1202) of said control signal, adapted so that the frequency of said local oscillator varies over time either linearly or linearly by steps, with possible temporal interference;and- pulse shaping means (101, 201) for generating a series of transmission pulses (IE) and a corresponding series of reception demodulation pulses (IDR), each reception demodulation pulse having a duration greater than that of the corresponding transmission pulse and defining a respective time range for reception of the echoes. Figure for the abstract: Fig. 1;

Description

Title of the invention: Pulse radar using a hybrid pulse and frequency modulation approach

[0001] The invention lies in the field of radars, and more particularly radars on a chip, which can be used in particular for measuring vital signals of a patient or as a presence detector, see for example (Antide 2020). These radars must have both a high spatial resolution (of the order of a few centimeters) and low consumption (a few tens of mW).

[0002] A commonly used technique for these applications is that of frequency-modulated continuous-wave, duty-cycle (FMCW-DC), see for example (Liu 2019) and (Siligaris 2023). Indeed, this technique exploits the principle of compressing a wave emitted with a wide bandwidth to a narrow bandwidth in intermediate frequency, which makes it possible to use analog-to-digital converters (ADCs) with a relatively low acquisition rate, of a few MSps or tens of MSps (IMSps = 10 6 samples per second - acronym for "Mega Samples per second").However, in the presence of targets with very diverse radar cross sections (RCS), it will be necessary to use ADCs with high dynamic range, for example 9 to 12 bits or more, and therefore relatively high power consumption.

[0003] An alternative is to use the ultra-wideband radio pulse (IR-UWB) technique, in which a short pulse is emitted and the time of flight of its echo is measured to determine the target distance. See for example (Andersen 2017). An advantage of this technique is that, since the echo signals are separated in time, it is possible to apply automatic gain control (AGC) to make it possible to acquire a highly contrasted environment in the presence of objects of very diverse RCS while limiting the dynamics of the ADCs. On the other hand, obtaining good spatial resolution relies on the use of a high acquisition rate, of several GSpS (IGSps = 109 samples per second - acronym for "Giga Samples per second").

[0004] The invention aims to overcome, in whole or in part, the aforementioned drawbacks of the prior art. More particularly, it aims to make it possible to use ADCs having a lower dynamic range than in FMCW-DC radars and a lower acquisition rate than in IR-UWB radars, without sacrificing either the spatial resolution or the RCS dynamic range of the detectable targets.

[0005] According to the invention, this aim is achieved by a hybrid approach combining the use of a pulsed radar signal and the principle of frequency modulation. More particularly, the invention uses a radar signal consisting of a series of transmission pulses in which the frequency of the carrier varies, for example linearly, from one pulse to another. Demodulation pulses, generated at the same time as the transmission pulses by a shared local oscillator, are used to demodulate the echoes of the transmission pulses. Since the radar signal is pulsed, the echo signals are separated in time, as in the case of the IR-UWB technique, allowing a gain adjustment which authorizes the use of ADCs with relatively low dynamic range even in the presence of a contrasting environment.Furthermore, the acquisition rate of the ADC depends on the pulse repetition frequency and not on the spatial resolution; it can therefore be lower than in the conventional IR-UWB technique. Advantageously, the technique of the invention is compatible with the use of frequency, pulse repetition period and / or phase jamming, which is desirable in particular for security reasons (immunity to interference and electromagnetic attacks).

[0006] Also, an object of the invention is a radar device comprising: - a transmission channel configured to generate a radar signal; and - a reception channel configured to receive echoes of said radar signal and to demodulate them synchronously with their generation so as to extract time-of-flight information therefrom; characterized in that the transmission channel and the reception channel comprise: - a shared local oscillator, having a variable frequency as a function of a control signal; - a generator of said control signal, adapted so that the frequency of said local oscillator varies over time either linearly, or linearly by steps, or by taking a plurality of discrete values ​​which can be obtained by jamming a linear variation by steps; and - means for shaping a signal generated by said local oscillator at said variable frequency, adapted to generate a series of transmission pulses and a corresponding series of reception demodulation pulses, each reception demodulation pulse having a duration greater than that of the corresponding transmission pulse and defining a respective time range for reception of the echoes.

[0007] According to particular embodiments of such a device:

[0008] - The shaping means may comprise: - a first pulse shaper, belonging to said emission channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and provide at its output said transmission pulses; and - a second pulse shaper, belonging to said reception channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and provide at its output said reception demodulation pulses.

[0009] - The radar device may also comprise a device for controlling said bone. local oscillator configured to restart said local oscillator upon each generation of a transmission pulse and the corresponding reception demodulation pulse.

[0010] - The shaping means may comprise: - said device for controlling said local oscillator, which is then configured to shape rising edges of the signal generated by the latter at said variable frequency; and - a falling edge shaping device, arranged on said transmission channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and to provide at its output said transmission pulses; said falling edge shaping device also being configured to generate a reset signal for said local oscillator.

[0011] - The radar device may also comprise a period jammer. pulse petition, configured to control said local oscillator control device so as to pseudo-randomly vary the interval between two successive restarts of said local oscillator.

[0012] - The radar device may also include a phase jammer for apply the same phase interference to the transmission pulses and to the reception demodulation pulses.

[0013] - The receiving path may comprise a mixer configured to receive in input of said echoes of said radar signal and one of said reception demodulation pulses and outputting a mixing signal, a low-pass filter for filtering said mixing signal and an analog-to-digital converter for converting the filtered mixing signal to digital format. More particularly, said low-pass filter may comprise a windowed integrator, configured to integrate said mixing signal over successive integration time windows, each said integration time window being contained in the reception time range of the echoes defined by a said reception demodulation pulse, and to be reset at the end of said integration window. Said windowed integrator may also have an integration gain that varies over time in a predetermined or adaptive manner. Furthermore, said control signal generator may be adapted so that the frequency said local oscillator varies in time by taking a plurality of discrete values ​​which can be obtained by scrambling a linear variation by steps; and the reception channel can also comprise a means for frequency descrambling of the filtered mixing signal converted to digital format.

[0014] - The reception path may comprise an amplifier equipped with a device automatic gain adjustment to compensate for variations in the intensity of received echoes.

[0015] - Said local oscillator may be a microwave oscillator.

[0016] Other characteristics, details and advantages of the invention will emerge on reading the description given with reference to the appended drawings given by way of example and which represent, respectively:

[0017] [Fig. 1], the functional diagram of a device according to a first embodiment of the invention;

[0018] [Fig.2], transmission pulses and reception demodulation pulses corresponding;

[0019] [Fig.3], a graph illustrating the temporal sequence of the emission pulses and demodulation in reception, as well as the signal generated by the local oscillator;

[0020] [Fig.4], the functional diagram of a device according to a second embodiment of the invention;

[0021] [Fig.5A], the spectrum of a transmission signal;

[0022] [Fig.5B] and [Fig.5C], the spectrum of the mixing signal in reception before and after low-pass filtering, respectively, in the presence of a single target;

[0023] [Fig.6A], [Fig.6B], [Fig.6C] and [Fig.6D], the mixing signal in the domain of the time in the presence of a single target before and after low-pass filtering ([Fig.6A]: in-phase component before filtering; [Fig.6B]: filtered in-phase component; [Fig.6C]: quadrature component before filtering; [Fig.6D]: filtered quadrature component];

[0024] [Fig.7A], [Fig.7B], [Fig.7C], [Fig.7D], the spectrum of the mixing signal in reception in the presence of two targets, respectively: unfiltered and without gain profiling ([Fig.7A]); after low-pass filtering but without gain profiling ([Fig.7B]); unfiltered but with gain profiling ([Fig.7C]); after low-pass filtering and with gain profiling ([Fig.7C]);

[0025] [Fig.8], three different types of local oscillator control signals;

[0026] [Fig.9], the functional diagram of the reception path of a device according to a third embodiment of the invention; and

[0027] [Fig. 10], the use of a windowed integrator to simultaneously perform low-pass filtering, gain profiling and suppression of unwanted signals.

[0028] The radar device of [Fig.l] consists of a transmission channel 1, for generating a radar signal consisting of so-called emission pulses IE and a reception channel 2, for receiving echoes of these pulses reflected by targets and processing them so as to extract time-of-flight information. These two channels have shared components 12.

[0029] The components shared between the transmission channel and the reception channel comprise a clock 1200 generating a timing signal for the device determining the repetition period PRP of the transmission pulses. Optionally, a time jammer 1201 introduces pseudo-random variations of this repetition period; the interval between the transmission pulse of rank n and the following one is then designated by PRPn (see [Fig.2]). In this case, it is the smallest possible PRP which determines the maximum target distance which can be measured.

[0030] The timing signal, possibly temporally scrambled, drives a device 1203 for triggering a local oscillator 1204 of the frequency-controlled type. At each timing signal, the device 1203 restarts the local oscillator 1204, then stops it before the arrival of the next timing signal. The initial phase of the oscillator 1204 is the same at each restart. Its oscillation frequency, for its part, varies linearly from one restart to the next. This is made possible by a generator 1202 of an oscillator control signal, driven by the timing signal (possibly temporally scrambled). On the other hand, the frequency of the local oscillator remains constant between its triggering and its stopping. Generally speaking, the frequency of the local oscillator is preferably in the microwave range.This refers to frequencies between 300 MHz and 300 GHz or, more restrictively, the range 1 GHz - 100 GHz.

[0031] The oscillator signal SO generated by the local oscillator 1204 - consisting of an in-phase component I and a quadrature component Q - is supplied as input to a first pulse shaper 101 belonging only to the transmission channel 1, configured to generate, at each triggering of the local oscillator, a transmission pulse IE typically having a duration of the order of a few nanoseconds (ns). The upper part of [Fig.2] illustrates a transmission pulse IE n comprising a rising edge FME and a falling edge FDE, followed - after an interval PRP - by another transmission pulse IEn+i. The latter has an envelope identical to that of IEn, but a carrier of different frequency, as explained above.

[0032] The I and Q components of the transmission pulses IE are combined by a combiner 102, then supplied as input to a power amplifier 103 to then be transmitted by an antenna not shown.

[0033] The oscillator signal SO is also supplied as input to a second pulse shaper 201 belonging only to the reception channel 2, configured to generate, at each triggering of the local oscillator, a so-called “receive demodulation” IDR pulse, preferably of a duration significantly greater (for example by at least a factor of 10) than that of the transmission pulses IE. The duration of the IDR pulse gives an upper limit to the pulse repetition frequency and determines the acquisition “depth”, i.e. the maximum target distance that can be detected.

[0034] The lower part of [Fig.2] illustrates a reception demodulation pulse IDRn comprising a rising edge FMR and a falling edge FDR, followed - after a PRP interval - by another reception demodulation pulse IDRn+i. The latter has an envelope identical to that of IDRn, but a carrier of different frequency, as explained above. On the other hand, the carriers of the reception demodulation pulses IDRn, IDRn+i are identical to those of the corresponding transmission pulses IEn, IEn+i.

[0035] A phase jammer 1205 may optionally be provided to apply identical polarity inversions, according to a pseudo-random sequence, to the IE and IDR pulses. If the inversions are applied independently to the two quadratures of these pulses, this results in QPSK modulation by a pseudo-random signal. Alternatively, the jammer 1205 may apply phase jumps that can take more than two values.

[0036] The IDR pulses are supplied as input to a mixer 203 which also receives, on another input, an echo signal SE, picked up by an antenna not shown and amplified by a low-noise amplifier 202. It is initially considered that the echo signal SEn, which is mixed with the reception demodulation pulse IDRn, comes from the reflection of the transmission pulse IEn by a target at a distance d and therefore corresponds to a replica of said transmission pulse delayed by T = 2d / C, c being the speed of light. It is understood that, under these conditions, the reception demodulation pulse defines a reception window for the echo signals.

[0037] [Fig. 3] illustrates the temporal sequence of the envelopes of the transmission pulses IE and reception demodulation pulses IDR, as well as of the signal generated by the local oscillator SO. At an initial instant t0, the local oscillator 1204 receives the control signal which defines its oscillation frequency then, at an instant ti approximately 1 ns later, it receives a trigger signal which starts the oscillation. It is noted that the amplitude of the signal SO increases progressively before stabilizing at an instant t2 and maintaining a constant value until the oscillator stops at an instant t3, several tens of nanoseconds later. Said stop occurs approximately 10 ns before the arrival of the next trigger signal. The rising edges of the transmission pulses IE and reception demodulation pulses IDR start once the amplitude of the signal SO is stabilized. While the IE transmission pulse is very brief (eg 6 ns), the demodulation transmission in reception continues until a short time (eg 1 ns) before the oscillator stops.

[0038] The mixing signal SM from the mixer 203 is first filtered by a high-pass filter (optional) whose purpose is to remove continuous or very low frequency components resulting from direct coupling between transmitter and receiver and other parasitic effects. Then, the signal is preferably amplified by a variable gain amplifier 205 which performs a gain adjustment to compensate for the differences in intensity of the different echoes received. The variation can follow a predefined profile - for example increasing over time, because later echoes correspond to a more distant target and therefore to a greater attenuation - or be adaptive.

[0039] The mixing signal SM then passes through a low-pass filter 206, with a bandwidth of the order of a few MHz (depending on the frequency slope used in transmission and the greatest desired echo distance). The filtered mixing signal SMF at the output of the filter 206 is then sampled and converted to digital format by an analog-to-digital converter (ADC) 207. Given the low bandwidth of the filter 206, the converter 207 can have a relatively low acquisition frequency, for example 20 MSpS (1 MSpS = 106 samples / second). The use of a variable gain amplifier makes it possible to limit the resolution of the converter - for example 12 bits - while maintaining an acceptable dynamic range.

[0040] The signal converted to digital format is finally processed by a processor 208 in order to extract time-of-flight information from the echo signals. As will be explained later, this processing may consist of the simple calculation of a Fourier transform.

[0041] [Fig. 4] shows the functional diagram of a device according to a second embodiment of the invention. This device differs from that of [Fig. 1] in that the local oscillator 1204 is triggered by a device 1206 which, at the same time, shapes the rising edge of the IE and IDR pulses. The falling edge of the IE pulses is shaped by a device 105 receiving the oscillator signal as input, while the falling edge of the IDR pulses is simply defined by stopping the oscillator. In other words, the generation of the pulses is done partially in baseband, instead of entirely in the microwave domain as in the case of the first embodiment. Spectrum control is, however, more difficult.

[0042] Document FR 3 099 910 discloses a circuit that can be used to produce pulse shapers 101, 201 and 105. This circuit is compatible with binary phase scrambling.

[0043] Document FR 3 015 153 describes a microwave oscillator triggered and controlled in oscillation frequency, which can be used to implement the local oscillator 1204.

[0044] The operation of the device of [Fig.l] and [Fig.4] will now be illustrated using an analytical model and numerical simulations.

[0045] We first define a reference transmission pulse in baseband p0(t) whose frequency characteristics are compatible with the regulatory templates and the minimum bandwidth sought, as well as having a temporal support as low as possible. We can for example use a Gaussian pulse of expression:

[0046]

[0047] Where the duration parameter Tp is related to the BXrfü band of the pulse by

[0048] 2,OÏWÏ (2) T — * v 7 lp"

[0049] xdB being the reference level for measuring the BxdB band. Typically, xdB=-10 dB and BXc1b=500 MHz are taken, which gives 7P~ 1.367^5 and an effective duration of the pulse at 99% of its amplitude equal to 4.292*7”p~5.867ns. A significantly shorter pulse is possible; the order of magnitude to be retained in practice is -2—.

[0050] The reference emission pulse is then time-shifted so that its effective support is on [0, pT P\, with for example P — 4.292.

[0051] We now consider the case where the local oscillator 1204 is restarted at each pulse from the same initial being, with a phase conventionally taken equal to 0 at each pulse start and a frequency updated at each pulse following a linear ramp rising with slope $ _ where Bchirp cst the frequency excursion of the chirp and T^irp the duration of the chirp, the frequency remaining constant between two restarts. We also consider that the PRP interval between two restarts of the oscillator is constant and equal to T; therefore, the number of pulses is _ Lÿ. The demodulation pulse in reception is taken from length equal to T and phase interference is not taken into account. The analytical expression of the emitted radar signal, in the form of a sequence of emission pulses, is therefore given by

[0052] s(t) (3)

[0053] With œ(n) = + ^■27rBchirp = + n2naT = wm + narT with ar = 2na

[0054] The analytical expression of the sequence of demodulation pulses in IDR reception is written

[0055] LORx(t) = Y^T(t~nT)e^^^ (4)

[0056] With RT(t) a unitary rectangular time gate of duration T centered on I.

[0057] Considering the case of a reflection of the radar signal by a single target at a distance d, the mixing signal SM at the output of the mixer is obtained by calculating the product of the delayed radar signal Aer-Zdlc and the signal LORx ( / ) • We obtain

[0058] SMf = s(t_L0^-nT-r)e>"^ ()er^ (5)

[0059] It is considered that there is no interference between consecutive PRPs, that is to say that the transmission pulse TEn is received during the duration of the reception demodulation pulse TDRn, which assumes r + / 3.Tp < T. We then have:

[0060] SM(t) = - nT-T)RT(t-nT)ej,,^t!^t^ (6)

[0061] The left term is a fixed phase term that can be neglected. The sum term is a pulse signal whose phase of consecutive pulses rotates from a'T which amounts in the frequency domain to having a first harmonic at F — i _ nT- The other frequency components are of period 1 and therefore at i q. Pt for all relative integers k. The distance information of the object is therefore present in this first harmonic, which is extracted by the low-pass filter 206, typically having a bandwidth less than 1. The sampling rate of the analog-digital converter 207 is determined by this bandwidth. In the presence of several echoes at different distances, we obtain as many harmonics, resolved or not.

[0062] [Fig.5A] illustrates the power spectrum of a radar signal having a bandwidth of 500 MHz, a pulse repetition frequency of 10 MHz, a chirp defined by a = Q . In dotted line, the spectral template which must be respected, and which actually is.

[0063] [Fig.5B] illustrates the spectrum of the SM mixing signal in the presence of a target at a distance such that the delay T is 20 ns. The presence of discrete harmonics is noted, the lowest frequency of which is extracted by the low-pass filter 206. [Fig.5C] shows the spectrum of the SMF filtered signal, in which the higher order harmonics are attenuated by 20 dB or more, and thus effectively suppressed. [Fig.6A] and [Fig.6C] show, respectively, the I and Q components of the SM signal in the time domain, and [Fig.6B] and [Fig.6C] the I and Q components of the SMF filtered signal.

[0064] [Fig.7A] and [Fig.7B] show the spectra of the SM and SMF signals, respec tively, in the case of two reflections at 20 ns and 60 ns, with a power difference of 19 dB and in the absence of gain profiling. We can see the splitting of the harmonics, but the component corresponding to the greatest delay is difficult to detect. As illustrated in [Fig.7C] and [Fig.7D], the use of amplifier 205 with automatic gain adjustment makes it possible to compensate for the variations in intensity of the received echoes.

[0065] Certain assumptions of the analytical model can now be relaxed.

[0066] First of all, it is assumed that the condition that there is no interference between consecutive PRPs, i.e. t + PTP < T, is not met. In this case, there is aliasing of the spectrum of the mixing signal, because an echo is demodulated by a signal of a frequency different from that of its carrier.

[0067] First we notice that 0 < t < T is equivalent to OtT < FT < 0 and since _ Bd^p we have NT Ba-nrp p < 0. There is therefore a degree of freedom offered by the band covered by the chirp between the intermediate frequency range (i.e. at the mixer output) covered and the PRF. In the example given previously (PRF = 1 OMHz. a = 0.025GHz / us and Bchiip = 500MHz), N = 200 and the minimum IF is -2.5MHz or PRF / 4, harmonic filtering is possible because the lowest is at 7.5MHz. If Bchirp = 1GHz then the minimum IF is -5MHz and the lowest harmonic +5MHz which would require complex filtering.

[0068] Let us return to the case (k + 1)T > T > kT with k > I and suppose that there is no phase interference. Then, apart from the edge effect caused by the “lost” pulses, the spectrum of the SM signal is “folded” to FT_kT. We verify that Ft Ft + ^ • Put differently, FT is not a continuous function of T, it is not a “classical” spectral folding.

[0069] We always assume (k+l)T>T> kT with k > 1 and that there is a BPSK type phase interference. In this case the descrambling sequence is misaligned since in the basic architecture it is synchronous with the jamming sequence. Therefore, the pulses will not be summed coherently.

[0070] At this point, it is appropriate to enrich the model of the SM signal at the mixer output by taking into account the BPSK scrambling and unscrambling operations with the binary code b(n) e{-1,1}: [007 i] SM ( t ) = ( n ) ( t - nT - t) n> (E^b ( n),R T ( t-nT ) < '<) (t-nT) (7)

[0072] If 0 < T < T, the operation is transparent:

[0073] SM(t) - s(tT).LO Rx (t) = E^b(n)bU^^^ i

[0074] If (k+1)T > T> kT with k> 1;

[0075] =E^b(n)b(n + k)po(t-nT'T)ebw(^

[0076] Since b(n) is ideally a white random process (sequence), then E[b(n)b(n+k) ] = 0, V k?t 0 and E[m(t) ] = 0 so by ergodicity m(t) has zero mean. In practice, b ( n ) is not perfectly white and m ( t ) is similar to an interference noise spreading the power across the entire band as a function of the covariance of b ( n ).

[0077] In an embodiment making it possible to handle cases where (k+l)T>T> kT with k > 1 it is necessary to parallelize the descrambling operation for all the desired values ​​of k. In this case it would be preferential to produce a demodulation pulse in reception without phase interference, which requires a minor modification of the architecture of [Fig.l] and [Fig.4], and to carry out the phase descrambling after the reception mixer by parallelizing the baseband chain and delaying the descrambling sequence b(n) by k values ​​• This constitutes an additional option.

[0078] Finally, there are portions of overlapping PRP where the mixer output is corrupted: ( kT - |3Tp < t < kT ) the received pulse being mixed with a phase-discontinuous demodulation signal. This concerns by extension the portions of PRP in which the demodulation signal is voluntarily switched off. A change of PRP can make it possible to treat these "dead" zones.

[0079] We now consider the case of temporal interference (pseudo-random variations in the spacing between pulses). Numerical simulations make it possible to verify that this has no impact on the frequency of the fundamental harmonic as long as the amplitude of the spacing variation is not excessive.

[0080] The following assumption concerns the restart of the local oscillator and its initial phase. In the proposed analytical model, the phase is reset to 0 (or any constant value) at each pulse, regardless of the generated frequency. The analytical model clearly shows that if this assumption is lifted then the expression of the mixing signal becomes: [0 ° 81] S M(t) =E^p0(t-nT-T)R T (t^ (8)

[0082] which gives the same result, the phase term being cancelled at the mixer output.

[0083] Finally, we are interested in the case where the frequency of the oscillator evolves continuously and linearly, including during the generation of the IE and IDR pulses, which present so a “chirp”. We can then write 100841 s(t)=000-"T)eM.)(>-»T)(9) loess] lOkJO = d°)

[0086] v chirp

[0087] After mixing, we obtain (still with the other initial hypotheses including phase resetting):

[0088] SM(t) = (11)

[0089] This signal is therefore a sinusoid of frequency FT = - aT "sampled" by a finite comb of pulses spaced by T. The result is indeed a spectrum of lines of frequencies FT(]<) = - «t + y weighted by the spectrum of the pulse. The difference with the case considered previously only comes from the evolution of the frequency of the local oscillator during the duration of the pulse, which can be calculated: if the pulse lasts PTp, the frequency changes by 0.147MHz with 0TP = 5.867ns and a = 0.025GHz / us and the difference is limited by 0.3°. We see that this difference is marginal given the bandwidths considered.

[0090] Up to now we have considered the case where the oscillator control signal generated by the device 1202 is a linear ramp, increasing or decreasing. In fact, the frequency excursion must be limited, the control signal will therefore be rather sawtooth or triangular (alternating an increasing ramp and an increasing ramp). As, preferably, the frequency of the oscillator varies discretely, from one trigger to the next, the control signal can also vary in steps. In [Fig.8], the reference 1202' designates a device for generating such a control signal, linear in steps. The values ​​taken by the oscillator control voltage are designated by the integers from 0 to 5; a linear ramp in steps therefore corresponds to the sequence {0, 1, 2, 3, 4, 5}.According to an alternative embodiment of the invention, these discrete values ​​may be subject to jamming, which results in frequency jamming of the radar signal; in [Fig.8], the reference 1202” designates a device for generating such a control signal, corresponding to the sequence {4, 1, 2, 5, 3, 0}. Alternatively, certain values ​​may be omitted, for example the frequency {4, 1, 5, 3, 0} may be used.

[0091] The primary interest of this variant relates to the problems of coexistence with other radars of the same type, or to security (sequence not known to an attacker and modifiable at will). Another interest is the possibility of transmitting on several antennas (MIMO operation) using orthogonal frequency hopping sequences between antennas. In doing so, the transmission can be done simultaneously on several sub- frequency bands in parallel and each receiving channel only retrieves the correlation results with the sequence of its corresponding transmitter.

[0092] The difficulty induced by frequency interference is the need to compensate for it in the receiver, otherwise the mixing signal will not be obtained because there will be random phase jumps between the low-frequency components of the mixing signal corresponding to successive pulses. This compensation cannot be achieved in the analog domain. To achieve it in the digital domain without having to significantly increase the acquisition rate of the analog-to-digital converter 207, it is possible to implement low-pass filtering by means of a windowed integrator 210 driven by the timing signal, as illustrated in [Fig.9]. The integrator being reset to zero at each reset of the local oscillator, each digitized sample contains the useful information, low-pass filtered by the integrator over a time PRPn.A descrambling module 211 applies to the samples the permutation which puts them back in the “correct” order (the one in which the frequency of the oscillator signal evolves linearly).

[0093] Even independently of the case of frequency interference, the use of a windowed integrator can prove advantageous because such a device can replace, in whole or in part, the variable gain amplifier 205 or even the high-pass filter 204. Indeed, as illustrated in [Fig. 10], the integrator has an integration gain G which can vary in a predefined or adaptive manner to compensate for variations in the intensity of the echoes received from different targets. The example in [Fig. 10] corresponds to the case where the echo signal consists of reflections coming from targets with substantially equivalent radar cross-sections, but located at different distances. In this case, the later reflections are generally weaker because they undergo greater attenuation, which can be compensated by an increasing integration gain G within the integration window FI.Furthermore, unwanted SEI signals intended to be suppressed by the high-pass filter 204 arrive at the receiving path very early, since they are mainly due to direct coupling with the transmitter. These signals can therefore be suppressed, at least in part, by shifting the start of the integration window by a few nanoseconds relative to the start of the PRP period.

[0094] The invention has been described with reference to particular embodiments, but variations are possible. For example, the generation of two I and Q components of the signals is not essential (but advantageous, since it makes it possible to reduce the acquisition rate). Similarly, the frequency and time ranges indicated are only given as a non-limiting example. Furthermore, the use of two quadrature components is not essential; if only the I component is used, the combiner 102 is not necessary.

[0095] The condition duration(IE)«duration(IDR) is not essential. In the limit, it is even possible to take the same duration for the two pulses.

[0096] In some embodiments, particularly operating at THz (Terahertz) frequencies, the low noise amplifier 202 at the input of the receive path may be omitted.

[0097] In the presence of temporal interference of the PRP, the analog-digital converter 207 and / or the possible windowed integrator 210 can be driven by the scrambled clock signal. This is however not essential because the frequency of the filtered mixing signal only depends on the delay between the echo signal and the demodulation pulse on reception, and is therefore not affected by possible temporal interference.

[0098] The processor 208 and, where appropriate, the frequency descrambling means 211 may be implemented by means of dedicated digital integrated circuits (FPGA, ASIC) or, more advantageously, a suitably programmed microprocessor. In the latter case, the frequency descrambling means 211 may be implemented by software.

[0099] The invention is particularly suitable for producing “on-chip” radars in which the radar device (or at least the transmission channel and / or the acquisition channel) is monolithically integrated, but it is not limited to this scenario. References

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Claims

Claims

1. Radar device comprising: - a transmission channel (1) configured to generate a radar signal (IE); and - a reception channel (2) configured to receive echoes (SE) of said radar signal and to demodulate them synchronously with their generation so as to extract time-of-flight information therefrom; characterized in that the transmission channel (1) and the reception channel (2) comprise: - a shared local oscillator (1204), having a variable frequency as a function of a control signal; - a generator (1202) of said control signal, adapted so that the frequency of said local oscillator varies over time either linearly, or linearly by steps, or by taking a plurality of discrete values ​​which can be obtained by jamming a linear variation by steps;and - shaping means (101, 201, 1206, 105) of a signal generated by said local oscillator (1204) at said variable frequency, adapted to generate a series of transmission pulses (IE) and a corresponding series of reception demodulation pulses (IDR), each reception demodulation pulse having a duration greater than that of the corresponding transmission pulse and defining a respective time range for reception of the echoes.;

2. Radar device according to claim 1 wherein the shaping means comprise: - a first pulse shaper (101), belonging to said transmission channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and provide at its output said transmission pulses (IE); and - a second pulse shaper (201), belonging to said reception channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and provide at its output said reception demodulation pulses (IDR).

3. Radar device according to one of the preceding claims, comprising a control device (1203, 1206) for said local oscillator configured to restart said local oscillator (1204) during each generation of a transmission pulse and of the demodulation pulse in corresponding reception.

4. Radar device according to claim 3 when it depends directly on claim 1 in which the shaping means comprise: - said control device (1206) of said local oscillator (1204), which is configured to shape rising edges of the signal generated by the latter at said variable frequency; and - a falling edge shaping device (105), arranged on said transmission path, configured to receive at its input said signal generated by said local oscillator at said variable frequency and provide at its output said transmission pulses; said falling edge shaping device (105) also being configured to generate a reset signal of said local oscillator.

5. Radar device according to one of claims 3 or 4 also comprising a pulse repetition period jammer (1201), configured to control said local oscillator driving device (1204) so ​​as to pseudo-randomly vary the interval between two successive restarts of said local oscillator.

6. Radar device according to one of the preceding claims also comprising a phase jammer (1025) for applying the same phase jamming to the transmission pulses (IE) and to the reception demodulation pulses (IDR).

7. Radar device according to one of the preceding claims wherein the reception channel (2) comprises a mixer (203) configured to receive as input said echoes (SE) of said radar signal and one of said reception demodulation pulses (IDR) and provide as output a mixing signal (SM), a low-pass filter (206, 210) for filtering said mixing signal and an analog-digital converter (207) for converting the filtered mixing signal (SMF) to digital format.

8. Radar device according to claim 7 wherein said low-pass filter comprises a windowed integrator (210), configured to integrate said mixing signal over successive integration time windows (IF), each said integration time window being contained in the echo reception time range defined by a said reception demodulation pulse (IDR), and to be reset at the end of said integration window.

9. A radar device according to claim 8 wherein said integrator windowed (210) has a time-varying integration gain in a predetermined or adaptive manner.

10. Radar device according to one of claims 8 or 9 wherein said control signal generator (1202”) is adapted so that the frequency of said local oscillator varies over time by taking a plurality of discrete values ​​obtainable by scrambling a linear variation by steps; and wherein the reception channel (2) also comprises means (211) for frequency descrambling of the filtered mixing signal converted to digital format.

11. Radar device according to one of the preceding claims in which the reception path comprises an amplifier (205) equipped with an automatic gain adjustment device to compensate for variations in intensity of the received echoes.

12. Radar device according to one of the preceding claims wherein said local oscillator (1204) is a microwave oscillator.

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