Motor control device and motor control method

Abstract

To enable stable motor drive.SOLUTION: A motor controller according to an embodiment includes a current detection element that is connected with a DC side of an inverter circuit and generates a signal corresponding to a current value, and determines a rotor position at least based on a phase current of a motor and tracks the position to generate a 2-phase or 3-phase PWM signal. A PWM signal generating unit generates 3-phase phase shift PWM signal by which a current detection unit can detect a 2-phase current at two timings definitely-located in a carrier period of the PWM signal. An angle and a speed of revolving magnetic field of the motor are estimated based on linkage magnetic flux of an armature coil of the estimated motor and a switching command for generating a different PWM signal depending on the level of percent modulation of voltage applied to the motor is outputted to the PWM signal generating unit. Unless motor current cannot be detected in one period of an electric angle when generating a 2-phase or 3-phase PWM signal pattern, a computed angle is generated using a speed estimated in the previous control period and based on the speed.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] An embodiment of the present invention relates to a control device that controls a motor via an inverter circuit by PWM controlling a plurality of switching elements connected in a three-phase bridge, and in particular to sensorless control using a single-shunt current detection method. [Background technology]

[0002] When detecting the currents of the U, V, and W phases to control a motor, there is a technology that uses a single shunt resistor inserted into the DC part of the inverter circuit to detect the currents. To detect all three phases of current with this method, it is necessary to generate a three-phase PWM signal pattern so that two or more phases of current can be detected within one cycle of the PWM (Pulse Width Modulation) carrier.

[0003] Therefore, Patent Document 1 proposes a technique for detecting currents of two or more phases at all times, even in a region where the modulation rate of the voltage applied to the motor is low, without increasing noise, by shifting the phase of the PWM signal within one period. Meanwhile, as a method for estimating the motor speed and angle from the estimated magnetic flux, for example, Non-Patent Documents 1 and 2 propose a magnetic flux observer. In the magnetic flux observer method, the α-axis component Ψα and the β-axis component Ψβ of the flux linkage of the motor winding are estimated based on, for example, two-phase currents Iα and Iβ obtained from a current sensor, two-phase voltages Vα and Vβ, and the motor winding resistance R, and the rotating magnetic field angle of the motor, and therefore the rotor phase angle and generated torque T are estimated. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] Patent No. 5178799 [Non-patent literature]

[0005] [Non-Patent Document 1] Inoue, et al., 3 others, Inoue, et al., "Actual machine verification of a magnetic flux estimation method to expand the operating range of direct torque control in PMSM," Proceedings of the 2021 National Conference of the Institute of Electrical Engineers of Japan, Institute of Electrical Engineers of Japan, March 1, 2021, 5-095 Summary of the Invention [Problem to be solved by the invention]

[0006] In the magnetic flux observer shown in Non-Patent Document 1, it is assumed that the current used for estimating the magnetic flux is a current sensor such as a CT or a three-shunt current detection method. However, in home appliances, a one-shunt current detection method is often used to reduce the cost of inverters. When a one-shunt current detection method is used, if the modulation rate of the voltage applied to the motor from start-up to low speed is low, the current cannot be detected, and an error may occur in the estimation of the magnetic flux. In particular, when the currents of the α-axis and β-axis, which vary in a sinusoidal wave, are used for the calculation of magnetic flux estimation, if the previous value is used when the current is not detected, there is a problem that the estimation error of the interlinkage magnetic flux becomes large.

[0007] Therefore, by combining a flux observer method, which estimates the phase angle and speed of the motor's rotating magnetic field from the estimated flux linkage, with a one-shunt current detection method, the present invention provides a motor control device that suppresses estimation errors in the flux linkage that may occur from start-up in the low-speed region, enabling stable motor drive. [Means for solving the problem]

[0008] The motor control device according to the embodiment drives a motor via an inverter circuit that converts direct current into three-phase alternating current by controlling on / off of a plurality of switching elements connected in a three-phase bridge in accordance with a predetermined PWM signal pattern, a current detection element connected to a DC side of the inverter circuit and generating a signal corresponding to a current value; a PWM signal generating unit that determines a rotor position based on at least a phase current of the motor and generates a two-phase or three-phase PWM signal pattern so as to follow the rotor position; a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal pattern, the PWM signal generation unit generates a three-phase PWM signal pattern so that the current detection unit can detect two-phase currents at two fixed timing points within a carrier wave period of the PWM signal; a signal switching output unit that estimates a rotational magnetic field angle and a speed of the motor based on the interlinkage magnetic flux and outputs a switching command to the PWM signal generating unit to generate different PWM signal patterns according to the level of the modulation rate of the voltage applied to the motor; The motor control device further includes an angle correction unit that, when a two-phase or three-phase PWM signal pattern is generated and the motor current cannot be detected within one electrical angle cycle, uses the speed estimated in the previous control cycle and generates an angle calculated based on the speed estimated in the previous control cycle. [Brief description of the drawings]

[0009] [Figure 1] FIG. 1 is a functional block diagram illustrating a configuration of a motor control device according to a first embodiment. [Diagram 2] Diagram showing vector control block using magnetic flux observer [Diagram 3] A functional block diagram showing a detailed configuration of a position estimation control unit. [Figure 4] Functional block diagram showing the configuration of an integrator used in the magnetic flux estimation section [Diagram 5] Flowchart showing angle correction processing in the single shunt current detection method [Figure 6] A diagram showing an example of current detection rate according to the PWM output method [Figure 7] FIG. 2 is a functional block diagram showing a configuration of a motor control device according to a second embodiment. [Figure 8] FIG. 13 is a functional block diagram showing the configuration of an integrator used in a magnetic flux estimation unit according to a third embodiment. [Figure 9] FIG. 13 is a diagram showing the relationship between the actual angle and the estimated angle in a low speed range in the first embodiment. [Figure 10]FIG. 13 is a diagram showing the relationship between the actual angle and the estimated angle in a low speed range in the third embodiment. [Figure 11] 13 is a flowchart showing a process at the time of starting a motor according to a fourth embodiment. [Figure 12] A diagram showing each signal waveform [Figure 13] FIG. 13 is an enlarged view of a part of FIG. 12. [Figure 14] FIG. 13 is a functional block diagram showing the configuration of a motor control device according to a fifth embodiment. [Figure 15] A flowchart showing a process for switching the carrier frequency and output pattern of a PWM signal according to the level of the modulation rate. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0010] (First embodiment) FIG. 1 is a functional block diagram showing the configuration of the motor control device of this embodiment, which is obtained by adding some functional blocks to FIG. 1 of Patent Document 1. The DC power supply unit 1 is shown by a DC power supply symbol, but includes a rectifier circuit, a smoothing capacitor, and the like when generating DC power from a commercial AC power supply. The DC power supply unit 1 is connected to an inverter circuit 3 via a positive bus 2a and a negative bus 2b, and a shunt resistor 4, which is a current detection element, is inserted on the negative bus 2b side. The inverter circuit 3 is configured by connecting, for example, N-channel type power MOSFETs 5 (U+, V+, W+, U-, V-, W-) which are switching elements in a three-phase bridge connection, and the output terminals of each phase are connected to each phase winding of a motor 6 which is, for example, a brushless DC motor.

[0011] The terminal voltage of the shunt resistor 4 is detected by a current detection unit 7. The current detection unit 7 detects currents Iu, Iv, Iw of the U, V, and W phases based on the terminal voltage and the three-phase PWM signal pattern output to the inverter circuit 3. The phase currents detected by the current detection unit 7 are provided to a duty generation unit 8, which performs A / D conversion and reads them, and then performs calculations based on the control conditions of the motor 6, etc. As a result, the duties U_DUTY, V_DUTY, and W_DUTY for generating the PWM signals of the phases are determined.

[0012] For example, in the case of vector control, when a rotation speed command ωref of the motor 6 is given to the DUTY generating unit 8 from a microcomputer or the like that sets the control conditions, a torque current command Iqref is generated based on the difference from the estimated actual rotation speed of the motor 6. When the rotor position θ of the motor 6 is determined from each phase current Iu, Iv, Iw of the motor 6, the torque current Iq and the excitation current Id are calculated by vector control calculation using the rotor position θ. For example, a PI control calculation is performed on the difference between the torque current command Iqref and the torque current Iq to generate a voltage command Vq. The excitation current Id side is also processed in the same manner to generate a voltage command Vd, and the voltage commands Vq, Vd are converted to three-phase voltages Vu, Vv, Vw using the rotor position θ. Then, based on these three-phase voltages Vu, Vv, Vw, the phase duties U, V, W_DUTY are determined.

[0013] The phase duties U, V, W_DUTY are given to a PWM signal generating unit 9, and a three-phase PWM signal is generated by comparing the level with the carrier wave. A lower arm signal is also generated by inverting the three-phase PWM signal, and after adding dead time as necessary, the lower arm signal is output to a drive circuit 10. In accordance with the given PWM signal, the drive circuit 10 outputs a gate signal to each gate of six power MOSFETs 5 (U+, V+, W+, U-, V-, W-) that constitute the inverter circuit 3. For the upper arm side, a potential boosted by the required level is output.

[0014] The DC voltage detection unit 11 detects the voltage of the DC power supply 1 and outputs the detection result to the motor applied voltage modulation rate calculation unit 12. The motor applied voltage modulation rate calculation unit 12 calculates the modulation rate of the voltage applied to the motor 6 via the inverter circuit 3 based on the duty information and the like input from the DUTY generation unit 8. The calculated modulation rate is output to the PWM output method selection unit 13. The PWM output method selection unit 13, which is a signal switching output unit, outputs a switching signal that switches the output method of the PWM signal by the PWM signal generation unit 9 according to the input modulation rate.

[0015] Fig. 2 shows a vector control block using a magnetic flux observer. In Fig. 2, each phase current detected by a current detection unit 7 is converted into Iα and Iβ, which are the α-axis component and β-axis component of the motor current, in an abc / αβ conversion unit 21 of a duty generation unit 8. The currents Iα and Iβ obtained by this conversion are provided to a position estimation control unit 23. The position estimation control unit 23 estimates the magnetic flux based on the currents Iα and Iβ and voltage commands Vα, Vβ input from a dq / αβ conversion unit 26 described later.

[0016] The position estimation control unit 23 includes a magnetic flux estimation unit 23a and a speed / position estimation unit 23b. The magnetic flux estimation unit 23a estimates the α-axis and β-axis components φα and φβ of the interlinkage magnetic flux according to the following formulas (1) and (2). Mutual inductance is used for L. Self-inductance, d-axis inductance Ld, and q-axis inductance Lq may be substituted. φα=∫(Vα-R×Iα)dt-LIα …(1) φβ=∫(Vβ-R×Iβ)dt-LIβ …(2)

[0017] First, the speed position estimator 23b estimates the phase θ of the rotating magnetic field based on the α-axis and the torque T based on the estimated magnetic fluxes φα and φβ in accordance with the following equations (3) and (4), respectively. θ = ATAN (φβ / φα) … (3) T = 3 / 2 × (number of pole pairs) × (φα × Iβ-φβ × Iα) … (4)

[0018] The integrators on the right-hand side of each of equations (1) and (2) integrate the magnetic flux as shown in Fig. 4 using an incomplete integration method using a low pass filter (LPF) with a cutoff angular frequency ωc as shown in the transfer function of equation (5). "s" is a differential operator. G(S)=1 / (s+ωc) …(5) When the frequency of the magnetic flux is sufficiently higher than the cutoff angular frequency ωc, a good estimation result can be obtained. ω is estimated by differentiating θ estimated by equation (3). In addition to a general LPF, an IIR (Infinite Impulse Response) filter or an FIR (Finite Impulse Response) filter may be used as the LPF.

[0019] FIG. 3 is a functional block diagram showing the internal configuration of the position estimation control unit 23 in more detail in response to the above calculation. The integrator 29 of the magnetic flux estimation unit 23a shown in FIG. 3 is actually composed of an integrator 29a and a low-pass filter (LPF) 29b as shown in FIG. 4, and adopts a so-called incomplete integration method. The output signal of the integrator 29a contains an offset. The output signal is filtered by the LPF 29b to extract the offset component, and the offset component is cancelled by subtracting it in a subtractor at a later stage. The LPF 29b and the subtractor at the later stage may be composed of an HPF (High Pass Filter). Information on the rotor speed of the motor 6 is required for speed control. When the magnetic flux observer is used in a vector control configuration, the fact that the rotating magnetic field speed of the motor 6 and the rotor speed are steadily the same is utilized.

[0020] Referring again to Fig. 2, a rotation speed command ωref for the motor 6 is provided by a host control device such as a microcomputer that sets control conditions. A speed control unit 24 generates a torque current command Iqref based on the difference between the rotation speed command ωref and the rotation speed ω estimated by a position estimation unit 23. An αβ / dq conversion unit 22 calculates a torque current Iq and an excitation current Id from the currents Iα and Iβ by vector control calculation using the rotor position θ.

[0021] In the current control unit 25, for example, a PI control calculation is performed on the difference between the torque current command Iqref and the torque current Iq to generate a voltage command Vq. The same processing is performed on the excitation current Id side to generate a voltage command Vd. The space vector generating unit 27 converts the voltage commands Vq and Vd into three-phase voltages Vu, Vv, and Vw using the rotor position θ. Then, the duties U_DUTY, V_DUTY, and W_DUTY for generating the PWM signals of each phase are determined based on the three-phase voltages Vu, Vv, and Vw.

[0022] The phase duties U, V, W_DUTY are given to a PWM forming unit 28, and a two-phase or three-phase PWM signal is generated by comparing the level with the carrier. A lower arm signal is also generated by inverting the two-phase or three-phase PWM signal, and after adding a dead time as necessary, the signal is output to the drive circuit 10. The method of the PWM forming unit 28 to generate a three-phase PWM signal with shifted phases is, for example, the method of the fourth embodiment disclosed in Patent Document 1.

[0023] The motor applied voltage modulation rate calculation unit 12 shown in FIG. 1 calculates the modulation rate of the motor applied voltage for each carrier period based on Vα and Vβ calculated by the DUTY generation unit 8, as shown in equation (6). (Modulation rate) = 100 × Vdc / (√3 × √(Vq 2 +Vd 2 )) …(6) formula The calculation result is output to the PWM output method selection unit 13. Based on this information, the PWM output method selection unit 13 outputs a signal for switching the PWM output signal to the PWM signal generation unit 9. Furthermore, the PWM signal generation unit 9 outputs a current detection timing signal to the current detection unit 7. Note that the modulation rate of the motor applied voltage may simply be substituted by the motor rotation speed or the like.

[0024] Since the currents Iα and Iβ change in a sinusoidal manner over time, if the phase currents of the motor 6 cannot be detected in the one-shunt current detection method, the estimation accuracy may deteriorate if the magnetic flux is estimated using the currents Iα and Iβ estimated in the previous control cycle. Also, since the angle changes in a sawtooth waveform, if the phase currents of the motor 6 cannot be detected, an error will occur in the calculation of the vector control system if the angle estimated in the previous control cycle is used.

[0025] Fig. 5 shows a flowchart of the angle correction process in the one-shunt current detection method using a magnetic flux observer. If the phase current can be detected (S1; OK), normal control is performed, and the currents and voltages of the α-axis and β-axis are calculated (S2) to perform magnetic flux observer control (S3). Then, calculations are sequentially performed to estimate the angle θ, the load torque T, and the speed ω (S4 to S6). On the other hand, if the phase current cannot be detected (S1; NG), the speed ω estimated in the previous control cycle is used (S7), and the angle θ obtained by integrating the speed ω is used (S8). This process is performed in the magnetic flux estimation unit 23a, which is also the angle correction unit.

[0026] FIG. 6 shows an example of the current detection rate for each PWM output method. The higher the motor rotation speed and load torque, the closer the modulation rate of the motor applied voltage approaches 100%. When the modulation rate is in a low range, the PWM output method selection unit 13 generates a three-phase PWM signal pattern in which the output phase of the PWM signal pulse of each phase is shifted by a method different from the conventional method described in Patent Document 1. On the other hand, when the modulation rate is in a high range, it outputs a switching command to generate a two-phase or three-phase PWM signal pattern of pulse signals that are symmetrical with respect to the midpoint of the PWM period, as shown in FIG. 7 of Patent Document 1, for example.

[0027] As described above, according to this embodiment, the PWM signal generating unit 9 determines the rotor position based on at least the phase current of the motor 6, and generates a two-phase or three-phase PWM signal pattern so as to follow the rotor position. The current detecting unit 7 detects the phase current of the motor 6 based on the signal generated in the shunt resistor 4 and the PWM signal pattern. The PWM signal generating unit 9 generates a three-phase phase-shifted PWM signal pattern so that the current detecting unit 7 can detect two-phase currents at two fixed timing points within the carrier cycle of the PWM signal. At that time, the duty of one of the three phases is increased or decreased in both directions on the lag side and the lead side based on an arbitrary phase of the carrier cycle, the duty of the other phase is increased or decreased in one direction on the lag side or the lead side, and the duty of the remaining phase is increased or decreased in the opposite direction to the above direction.

[0028] The magnetic flux estimation unit 23a estimates the interlinkage magnetic flux of the armature winding of the motor 6 based on the phase current and output voltage command of the motor 6, and estimates the rotating magnetic field angle and the speed of the motor 6 based on the interlinkage magnetic flux. The PWM output method selection unit 13 outputs a switching command to the PWM signal generation unit 9 to generate a symmetrical PWM signal pattern in two or three phases when the modulation rate of the voltage applied to the motor is in a high region, and to generate a phase-shifted PWM signal pattern when the modulation rate is in a low region. When the motor current cannot be detected, the magnetic flux estimation unit 23a uses the previously estimated speed and generates an angle calculated based on the previous speed.

[0029] Here, the condition under which the motor current cannot be detected is when, when a two-phase or three-phase PWM signal pattern is generated, the duration of the PWM signal for which the current is to be detected within one electrical angle period is shorter than the time during which the current can be detected, for example, 5 to 10 μsec, taking into account the current ripple, A / D conversion time, etc. Therefore, even if a three-phase phase-shifted PWM signal pattern is generated so that two-phase current can be detected at two fixed timing points, the detection rate does not always become 100%, and the actual detection rate is generally in the range of 70% to 100%.

[0030] With this configuration, even when the one-shunt current detection method is applied, the three-phase currents Iu, Iv, and Iw can be detected from low to high modulation rates of the motor applied voltage, and the magnetic flux can be estimated based on the α-axis current and β-axis current and the voltage command vector. In addition, even when the phase currents of the motor 6 cannot be detected, the previously estimated speed value can be used to prevent deterioration of the position estimation accuracy.

[0031] Second embodiment Hereinafter, the same parts as those in the first embodiment are denoted by the same reference numerals and their explanations are omitted, and only the different parts are explained. In the first embodiment, the angle θ and the speed ω of the motor 6 are estimated based on the estimated magnetic flux, and the estimation is applied to vector control. In the second embodiment, the magnetic flux observer control is applied to the direct torque control.

[0032] As shown in Fig. 7, in direct torque control using a flux observer, instead of the αβ / dq conversion unit 22, the speed estimation unit 24 to the space vector formation unit 27, a UVW / αβ conversion unit 31, a torque calculation unit 32 which is a direct torque control execution unit, a binary level output unit 33, and a switching table 34 are used. Instead of the rotation speed command ωref, a target torque command Tref and a target magnetic flux command φref are input from a higher-level control device. Then, a three-phase PWM signal pattern is generated by referring to the switching table 34. Since direct torque control is a known technique, a detailed description will be omitted. Note that the motor can be applied to a synchronous reluctance motor or an induction motor in addition to a permanent magnet motor.

[0033] Third embodiment In the first embodiment, an inexact integration method is used for the integrators on the right-hand sides of equations (1) and (2). In the second embodiment, a second-order generalized integration method for the transfer function shown in FIG. 8 and equation (7) is used for integrating the magnetic flux in the same case. G(S)=kω' / (s 2 + Kω'S+ω' 2 ) …(7) ω' is the natural angular frequency of the second-order filter, and k is a coefficient that determines the damping.

[0034] The incomplete integration method can estimate the magnetic flux well when the magnetic flux frequency is sufficiently higher than ωc, but the use of LPF 29b reduces the accuracy in the low speed range indicated by the double-headed arrow in Fig. 9. In contrast, the second-order generalized integration method shown in Fig. 10 improves the frequency characteristics in the low speed range, making it possible to expand the operable range.

[0035] As described above, according to the third embodiment, when the modulation rate of the voltage applied to the motor is in a low region, the output phase of the PWM signal pulse of each phase is shifted by the method of Patent Document 1 to generate a three-phase PWM signal pattern for sensorless operation. By integrating the magnetic flux using the second-order generalized integral method, the estimation accuracy of the motor magnetic flux can be maintained even in slower speed regions, making sensorless control possible.

[0036] (Fourth embodiment) According to the third embodiment, the estimation accuracy of the motor magnetic flux can be maintained even at low speeds. However, when the motor 6 is started, the motor magnetic flux cannot be estimated with sufficient accuracy. For this reason, it is conceivable to perform forced commutation at the time of start-up, in which a d-axis current is applied at an angle according to a command speed, and then switch to sensorless operation after increasing the rotation speed of the motor 6. However, if the load torque of the motor 6 is excessive, there is a risk of loss of synchronism. If a sufficiently large d-axis current is applied during forced commutation, it is possible to handle the load at the time of start-up, but the power at the time of start-up will be large.

[0037] Therefore, in the fourth embodiment, forced commutation is performed at the start of the motor 6, in which a d-axis current is applied according to the estimated motor magnetic flux angle. In addition, a torque current command Iqref is also applied based on the difference between the command rotation speed and the estimated rotation speed of the motor 6. As a result, at the time of forced commutation at start, if the motor load is small, a d-axis current is applied, and if the motor load is large, a q-axis current is applied. Thus, even at the time of forced commutation at start, the motor current is changed according to the load, making it possible to start the motor 6 without wasting power.

[0038] In the start-up control sequence shown in FIG. 11, first, positioning control is performed (S11). Here, the excitation current command Idref is a predetermined value, the torque current command Iqref is zero, and the angle is a target angle. Next, "forced commutation control 1" is performed (S12). The current commands Idref and Iqref, and the angle are the same as in step S11. Then, the rotation speed command ωref is increased, and when the rotation speed command ωref exceeds the rotation speed threshold value 1 in step S13, the process transitions to "forced commutation control 2" (S14).

[0039] In "forced commutation control 2", the excitation current command Idref is set to, for example, about half the predetermined value in "forced commutation control 1". The torque current command Iqref uses the result of speed control in vector control, and the angle is set to a value estimated by a flux observer. If the rotation speed command ωref exceeds the rotation speed threshold 2 in the following step S15, the process transitions to sensorless control (S16). Here, the excitation current command Idref is set to zero. Then, a similar determination is made to step S15 (S17), and if the rotation speed command ωref exceeds the rotation speed threshold 2, the start-up process is terminated, and if it is equal to or less than the rotation speed threshold 2, the process returns to step S14.

[0040] Fig. 12 shows the operating waveforms of the motor when forced commutation is performed according to the estimated motor magnetic flux angle, and Fig. 13 shows an enlarged view of the rectangular area within the load application section in Fig. 12. The torque current component Iq also increases when the load increases during forced commutation. This shows that the motor output torque can be varied according to the load torque, even during forced commutation.

[0041] As described above, according to the fourth embodiment, when starting the motor 6, forced commutation is performed to apply a d-axis current according to the estimated motor magnetic flux angle, and a torque current command Iqref is also applied based on the difference between the estimated rotation speed of the motor 6 and the command rotation speed, so that the output torque of the motor 6 can be varied according to the load torque even during forced commutation. This allows the motor 6 to be started without wasting power.

[0042] Fifth embodiment As in the above-mentioned embodiment, when the output phase of the PWM signal pulse of each phase is shifted, if the frequency of the carrier wave is within the human audible range, such as 4 kHz, noise may become a problem. On the other hand, if the carrier wave frequency is increased for the entire range of motor drive, there is a concern that the switching loss of the inverter circuit 3 increases and the efficiency decreases overall. Therefore, in the fifth embodiment, the PWM frequency changer 41 shown in FIG. 14 changes the frequency of the PWM carrier wave according to the modulation rate range. In addition, the PWM frequency output method selector 13 also changes the output pattern of the PWM signal in synchronization with the change in frequency.

[0043] As shown in FIG. 15, if the modulation rate is less than the threshold (S21; YES), the carrier frequency is set to, for example, 8k to 16kHz or more (S22), and a phase-shifted PWM signal is output (S23). On the other hand, if the modulation rate is greater than the threshold (S21; NO), a symmetrical PWM signal pattern is generated in two or three phases (S24), and the carrier frequency is set low, for example, to 4kHz (S25). In the case of this output pattern, noise caused by the carrier frequency is smaller than when the output phase is shifted. It is not important to change the output pattern of the PWM signal or the carrier frequency in either order. Also, the carrier frequency may be changed in stages or all at once. In addition, the modulation rate of the motor applied voltage may be simply substituted by the motor rotation speed or the like.

[0044] (Other embodiments) The arrangement of the duty pulses for each phase may be determined by applying the first to third embodiments of Patent Document 1. The magnetic fluxes φα and φβ may be estimated using the calculations shown in (8) and (9). φα=∫(Vα-R×Iα)dt …(8) φβ=∫(Vβ-R×Iβ)dt …(9)

[0045] Although some embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, substitutions, and modifications can be made without departing from the spirit of the invention. These embodiments and their modifications are included in the scope and spirit of the invention, and are included in the scope of the invention and its equivalents described in the claims. [Explanation of symbols]

[0046] In the drawing, 3 indicates an inverter circuit, 4 indicates a shunt resistor, 5 indicates a power MOSFET, 6 indicates a motor, 7 indicates a current detection unit, 8 indicates a duty generation unit, 9 indicates a PWM signal generation unit, 12 indicates a motor applied voltage modulation rate calculation unit, 13 indicates a PWM output method selection unit, and 23 indicates a position estimation control unit.

Claims

1. 1. A motor control device that drives a motor via an inverter circuit that converts direct current into three-phase alternating current by controlling the on / off of a plurality of switching elements connected in a three-phase bridge in accordance with a predetermined PWM signal pattern, a current detection element connected to a DC side of the inverter circuit and generating a signal corresponding to a current value; a PWM signal generating unit that generates a two-phase or three-phase PWM signal pattern so as to follow a rotor position determined based on at least a phase current of the motor; a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal pattern, the PWM signal generating unit generates a three-phase phase-shifted PWM signal pattern so that the current detecting unit can detect two-phase currents at two fixed timing points within a carrier wave period of the PWM signal; a magnetic flux estimator that estimates a magnetic flux linkage of an armature winding of the motor based on a phase current and an output voltage command of the motor; a signal switching output unit that estimates a rotating magnetic field angle and a speed of the motor based on the interlinkage magnetic flux and outputs a switching command to the PWM signal generating unit to generate different PWM signal patterns according to the level of the modulation rate of the voltage applied to the motor; and an angle correction unit that, when a two-phase or three-phase PWM signal pattern is generated and the motor current cannot be detected within one electrical angle cycle, uses the speed estimated in the previous control cycle and generates an angle calculated based on the speed estimated in the previous control cycle.

2. 2. The motor control device according to claim 1, wherein the PWM signal generating unit generates the phase-shifted PWM signal pattern by increasing or decreasing a duty for one phase of a three-phase PWM signal in both directions, toward the lag side and the lead side, based on an arbitrary phase of the carrier wave period, while increasing or decreasing a duty for another phase in one direction, toward the lag side or the lead side, based on an arbitrary phase of the carrier wave period, and increasing or decreasing a duty for the remaining phase in a direction opposite to the direction based on an arbitrary phase of the carrier wave period.

3. 2. The motor control device according to claim 1, wherein the signal switching output unit outputs a switching command to the PWM signal generating unit to generate a symmetrical PWM signal pattern in two or three phases when the modulation rate of the voltage applied to the motor is in a high region, and to generate the phase-shifted PWM signal pattern when the modulation rate of the voltage applied to the motor is in a low region.

4. The motor control device according to claim 1 , wherein the magnetic flux estimator estimates the interlinkage magnetic flux by performing time integration based on the motor current and the output voltage command on the αβ coordinates.

5. 5. The motor control device according to claim 4, wherein the magnetic flux estimator estimates the interlinkage magnetic flux by time-integrating a value calculated from the motor current and output voltage command on the αβ coordinates and a winding resistance value of the motor using a double integrator.

6. 2. The motor control device according to claim 1, wherein the signal switching output section changes the frequency of the PWM carrier wave in accordance with a modulation rate region.

7. 7. The motor control device according to claim 1, further comprising a forced commutation execution unit that performs forced commutation by applying a d-axis current, which is a field current component, using an estimated motor angle when starting the motor.

8. 8. The motor control device according to claim 7, wherein the forced commutation execution unit also applies a q-axis current, which is a torque current component, by using a result of speed control when performing the forced commutation.

9. 2. The motor control device according to claim 1, further comprising a vector control execution unit that performs vector control for controlling the motor by using a motor angle and a motor speed calculated from the magnetic flux estimated by the magnetic flux estimation unit.

10. 2. The motor control device according to claim 1, further comprising a direct torque control execution unit that performs direct torque control of the motor using a motor angle and a motor speed calculated from the magnetic flux estimated by the magnetic flux estimation unit.

11. A method for controlling a motor via an inverter circuit that converts direct current into three-phase alternating current by controlling on / off of a plurality of switching elements connected in a three-phase bridge in accordance with a predetermined PWM signal pattern, comprising: A current detection element connected to the DC side of the inverter circuit generates a signal corresponding to a current value; generating a two-phase or three-phase PWM signal pattern to track a rotor position determined based at least on phase currents of the motor; Detecting a phase current of the motor based on the signal generated by the current detection element and the PWM signal pattern; generating a three-phase phase-shifted PWM signal pattern so that two-phase currents can be detected at two fixed timing points within a carrier wave period of the PWM signal; Estimating a flux linkage of an armature winding of the motor based on a phase current and an output voltage command of the motor; A rotational magnetic field angle and a speed of the motor are estimated based on the interlinkage magnetic flux, and a switching command is output to the PWM signal generating unit so as to generate a different PWM signal pattern according to a high or low modulation rate of the voltage applied to the motor; A motor control method in which, when a two-phase or three-phase PWM signal pattern is generated, if the motor current cannot be detected within one electrical angle cycle, a speed estimated in the previous control cycle is used, and an angle calculated based on the speed estimated in the previous control cycle is generated.

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