Power conversion device

Abstract

To provide a power conversion device capable of achieving stable and high-precision control of a magnet motor.SOLUTION: A processor of a power conversion device 20 calculates first power (first reactive power Qc) from a voltage (voltage command values vdc**, vqc**) and a current (current detection values idc, iqc) of a magnet motor 1. The processor calculates second power (second reactive power Qc^) from electrical circuit parameters (Ld*, Lq*, Ke*) of the magnet motor 1, steady-state components (current detection values idc, iqc) of the current of the magnet motor 1, transient components dt / d(idc), dt / d(iqc), and a frequency estimation value ωdc of the magnet motor 1. The processor estimates a phase deviation (phase error Δθ) which represents a deviation (difference) between a phase θdc of control and a phase (phase θd of magnet) of a magnetic flux of the magnet motor such that the first power follows the second power. The processor calculates the frequency estimation value ωdc from an estimation value of the phase deviation.SELECTED DRAWING: Figure 3

Description

[Technical field]

[0001] The present invention relates to a power conversion device. [Background technology]

[0002] As a stable and highly accurate control method in the low-speed range of position sensorless control, there is a description of a technology in which reactive power is calculated and the frequency of the magnet motor is estimated based on a voltage command value to a power converter, a current detection value, and electric circuit parameters and frequency estimate values ​​of the magnet motor, as described in Patent No. 4402600. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 4402600 Summary of the Invention [Problem to be solved by the invention]

[0004] The method described in Patent No. 4402600 uses two types of reactive power (Q and Q ^ ) and calculates the inverter frequency estimate so as to make the deviation ΔQ zero. The frequency estimate can be made less sensitive to temperature changes in the magnet motor winding resistance, achieving highly accurate control characteristics. In the low-speed range, reactive power is used to estimate the first motor frequency through PI (proportional integral) control, while in the medium-to-high speed range, the phase error (control phase and magnet motor phase) is directly estimated using the extended induced voltage, and the second motor frequency is estimated through PI control so as to follow zero. For this reason, when switching between the first motor frequency and the second motor frequency, if there is a difference between the two frequencies, a torque shock due to current changes can be expected.

[0005] An object of the present invention is to provide a power conversion device capable of achieving stable and highly accurate control of a magnet motor. [Means for solving the problem]

[0006] In order to achieve the above object, the present invention provides a power conversion device comprising a processor that calculates a first power from a voltage and current of a magnet motor, calculates a second power from electrical circuit parameters of the magnet motor, steady-state components and transient components of the current of the magnet motor, and an estimated frequency value of the magnet motor, estimates a phase deviation indicating a deviation between a control phase and a phase of the magnetic flux of the magnet motor so that the first power tracks the second power, and calculates the estimated frequency value from the estimated value of the phase deviation. Effect of the Invention

[0007] According to the present invention, it is possible to realize stable and highly accurate control of a magnet motor. Problems, configurations and effects other than those described above will become apparent from the following description of the embodiments. [Brief description of the drawings]

[0008] [Figure 1] FIG. 1 is a configuration diagram of a power conversion device according to an embodiment. [Diagram 2] FIG. 4 is a diagram showing the configuration of a phase error estimation calculation unit using an extended induced voltage in the high-speed range. [Diagram 3] FIG. 4 is a configuration diagram of a phase error estimation calculation unit in a low speed range according to an embodiment. [Figure 4] FIG. 4 is a configuration diagram of a frequency and phase estimation calculation unit according to the embodiment. [Diagram 5] Control characteristics when the extended induced voltage method for the medium to high speed range is used in the low speed range. [Figure 6] Control characteristics when the present invention is used in the low speed range. [Figure 7] Control characteristics when the present invention is used for switching between low speed range and medium / high speed range. [Figure 8] FIG. 1 is a configuration diagram for confirming the feasibility of the present invention. [Figure 9] FIG. 13 is a configuration diagram of a power conversion device according to another embodiment. [Figure 10] FIG. 13 is a configuration diagram of another phase error estimation calculation unit in a low speed range according to the embodiment. [Figure 11]FIG. 13 is a configuration diagram of a power conversion device according to another embodiment. [Figure 12] FIG. 13 is a configuration diagram of another phase error estimation calculation unit in the high speed range according to the embodiment. [Figure 13] FIG. 13 is a configuration diagram of a power conversion device according to another embodiment. [Figure 14] FIG. 13 is a configuration diagram of another phase error estimation calculation unit in the high speed range according to the embodiment. [Figure 15] FIG. 13 is a configuration diagram of another power conversion device according to the embodiment. [Figure 16] FIG. 13 is a configuration diagram of another power conversion device according to the embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0009] This embodiment realizes stable and highly accurate control characteristics even in the low-speed range, for example, at about 10% of the base frequency from a stop, in position sensorless control that omits an encoder that detects the magnet phase of a magnet motor. In this embodiment, in the low-speed range, the phase error is estimated from the deviation of two types of reactive power in the same way as in the medium-high speed range. By estimating the motor frequency so that the estimated value of the phase error follows its command value, torque shock is prevented, and stable and highly accurate control characteristics are realized without adjusting the electric circuit parameters and control gains of the magnet motor set in the controller.

[0010] The present embodiment will be described in detail below with reference to the drawings. The same reference numerals (symbols) are used to designate common components in the various drawings. Furthermore, the embodiments described below are not limited to the examples shown in the drawings.

[0011] <First Example> FIG. 1 is a configuration diagram of a power conversion device according to an embodiment.

[0012] The magnet motor 1 outputs a motor torque that is a combination of a torque component due to the magnetic flux of the permanent magnet and a torque component due to the inductance of the armature winding.

[0013] The power converter 2 outputs a three-phase AC voltage command value v u* , v v * , v w * and varies the output voltage value and output frequency value to the magnet motor 1. The power converter 2 in this embodiment is equipped with a controller (such as a microcomputer), but the power conversion device 20 described below may also be equipped with a controller. The controller is composed of, for example, a memory (storage device), a CPU (processor), an input / output circuit (communication device), etc.

[0014] The DC power supply 3 supplies a DC voltage to the power converter 2 .

[0015] The current detector 4 detects the three-phase AC current i u , i v , i w The detected value of i uc , i vc , i wc The current detector 4 detects AC currents in two of the three phases of the magnet motor 1, for example, the u and w phases, and the v phase AC current is output under the AC condition (i u +i v +i w = 0), i v =-(i u +i w ) may also be calculated.

[0016] The coordinate conversion unit 5 converts the three-phase AC current i u , i v , i w Detection value i uc , i vc , i wc and the phase estimate θ dc to the d-axis and q-axis current detection values ​​i dc , i qc Output.

[0017] The speed control calculation unit 6 calculates a frequency command value ω r * and the frequency estimate ω dc Based on the torque command value t * The current command value i of the q axis is calculated by dividing it by the torque coefficient.q * Output.

[0018] The vector control calculation unit 7 calculates the d-axis and q-axis current command values ​​i d * , i q * , current detection value i dc , i qc , the frequency estimate ω dc and the d-axis and q-axis voltage command values ​​v dc ** , v qc ** Output.

[0019] The phase error estimation calculation unit 8 for the medium to high speed range is c axis and q c Axis voltage command value v dc ** , v qc ** , the frequency estimate ω dc , current detection value i dc , i qc Using the electric circuit parameters of the magnet motor 1, the phase θ dc and the phase θ of the magnet of magnet motor 1 d The estimated value Δθ of the phase error Δθ, which is the deviation from c _ H Output.

[0020] The phase error estimation calculation unit 9 for the low speed region is c axis and q c Axis voltage command value v dc ** , v qc ** , the frequency estimate ω dc , current detection value i dc , i qc and the electric circuit parameters of the magnet motor 1 are used to determine the phase θ dc and the phase θ of the magnet of magnet motor 1 d The estimated value Δθ of the phase error Δθ, which is the deviation from c _ L Calculate the following.

[0021] The frequency and phase estimation calculation unit 10 calculates an estimated value Δθ of the phase error in the low speed range. c _ L Or the estimated phase error in the medium to high speed range Δθ c _ H Based on dc and the phase estimate θ dc Output.

[0022] The coordinate conversion unit 11 is d c axis and q c Axis voltage command value v dc ** , v qc ** and the phase estimate θ dc to the three-phase AC voltage command value v u * , v v * , v w * Output.

[0023] First, the basic operation of the sensorless vector control method using the phase error estimation calculation unit 9 for the low speed range, which is a feature of this embodiment, will be described.

[0024] The speed control calculation unit 6 calculates a frequency command value ω r * The frequency estimate ω r ^ (=ω dc ) is controlled by proportional and integral control according to equation (1). * and the q-axis current command value i q * In the various equations below, the frequency estimate is ω r ^ Or ω dc It is sometimes written as:

[0025]

number

[0026] Here, K. sp: Proportional gain of speed control, K si : Speed ​​control integral gain, P m : Number of pole pairs, K e : Induced voltage coefficient, L d : d-axis inductance, L q :q-axis inductance, *: set value.

[0027] First, the vector control calculation unit 7 calculates the set value R of the winding resistance, which is an electric circuit parameter of the magnet motor 1. * , d-axis inductance setting value L d * , the set value of the q-axis inductance L q * , the value of the induced voltage coefficient K e * , d c axis and q c Axis current command value i d * , i q * and the frequency estimate ω r ^ (=ω dc ) according to equation (2), c axis and q c Axis voltage reference value v dc * , v qc * Output.

[0028]

number

[0029] Here, T acr : Response time constant of current control.

[0030] Secondly, the vector control calculation unit 7 calculates d c axis and q c Axis current command value i d * , i q * The current detection value of each component i dc , i qc By proportional and integral control, d is controlled according to equation (3).c axis and q c Axis voltage correction value Δv dc , Δv qc Calculate the following.

[0031]

number

[0032] Here, K. pd :d c Axis current control proportional gain, K id :d c Axis current control integral gain, K pq :q c Axis current control proportional gain, K iq :q c The vector control calculation unit 7 further calculates the integral gain of the current control of the d c axis and q c Axis voltage command value v dc ** , v qc ** Calculate the following.

[0033]

number

[0034] FIG. 2 shows a block diagram of the medium-to-high-speed phase error estimation calculation unit 8. The medium-to-high-speed phase error estimation calculation unit 8 is c axis and q c Axis voltage command value v dc ** , v qc ** , current detection value i dc , i qc and the electric circuit parameters of magnet motor 1 (R * , L q * ) based on the extended induced voltage equation (5) of the symbol 81, the estimated phase error Δθ in the medium to high speed range is calculated. c_H Calculate the following.

[0035]

number

[0036] Here, the low speed range phase error estimation calculation unit 9, which is a feature of the embodiment of the present invention, will be described.

[0037] FIG. 3 shows a block diagram of the low-speed phase error estimation calculation unit 9. The low-speed phase error estimation calculation unit 9 calculates the phase error d c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc Using the above, the first reactive power Q c is calculated according to equation (6).

[0038]

number

[0039] The second reactive power calculation unit 92 calculates d c axis and q c Steady-state component of the axis current detection value i dc , i qc , transient component dt / d(i dc ), dt / d(i qc ), frequency estimate ω dc , the electric circuit parameters of magnet motor 1 (L d * , L q * , K e * ) to calculate the second reactive power Q c ^ is calculated according to equation (7).

[0040]

number

[0041] The subtractor 93 calculates the first reactive power Q c and the second reactive power Q c^ is input, and its deviation, ΔQ c Calculate the reactive power deviation ΔQ c is input to the PI control calculation unit 95 so as to follow the command value 94 of "0", and the estimated value Δθ of the phase error Δθ in the low speed range is calculated according to the formula (8) by the P (proportional) + I (integral) control calculation. c_L Calculate the following.

[0042]

number

[0043] Here, K pθ : Proportional gain of phase error estimation calculation, K iθ : is the integral gain of the phase error estimation calculation.

[0044] Next, a description will be given of the frequency and phase estimation calculation unit 10, which is a feature of the embodiment of the present invention.

[0045] The switching unit 101 selects an estimated value Δθ of the phase error in the low speed range. c_L and the estimated phase error in the medium to high speed range, Δθ c_H and the frequency command value ω r * is input, and the frequency command value ω r * Depending on the size of Δθ c = Δθ c_L、 In the medium to high speed range, Δθ c = Δθ c_H is the estimated phase error Δθ c will be output as:

[0046] The subtractor 103 subtracts the estimated phase error Δθ c is the command value Δθ c * is input to the PI control calculation unit 104 so as to follow the dc In the I control calculation unit 105, the phase estimation value θ dcare calculated respectively.

[0047]

number

[0048]

number

[0049] Here, Kp pll : Proportional gain of PLL control, Ki pll : is the integral gain of the PLL control.

[0050] Next, the principle by which the embodiment of the present invention provides stable and highly accurate control characteristics will be described. c_H The control characteristics when the frequency command value ω r * is set to 2% of the base frequency. c axis and q c Axis voltage command value v dc ** , v dc ** The set value R of the resistance R included in the calculation formula for the phase error in the medium to high speed range shown in Equation (5) is * When there is no error in, (a)R * / R=1 (standard), when there is an error (b) R * This is the simulation result for / R=0.5.

[0051] In the figure, the upper part shows the load torque τ L , the middle row is the frequency command ω r * and the motor frequency ω r The lower part shows the phase error Δθ. A ramp-like load torque is applied from point A in the figure, and is changed to 100% at point B. From point B to the right, the load torque is maintained.

[0052] (a)R * / R=1 (standard) setting, phase error Δθ is zero in steady state, motor frequency ω r is the frequency command ω r * (b) R * At / R=0.5, the phase error Δθ increases negatively and the motor frequency ω r is stagnating near zero, and magnet motor 1 is out of step.

[0053] The embodiment of the present invention is c axis and q c Axis voltage command value v dc * , v qc * and the current detection value i dc , i qc Using the above, the first reactive power Q c Calculate from equation (6), d c axis and q c Steady-state component of the axis current detection value i dc , i qc , transient component dt / d(i dc ), dt / d(i qc ), frequency estimate ω dc , the setting value of the electric circuit parameters of the magnet motor 1 (L d * , L q * , K e * ) to calculate the second reactive power Q c ^ is calculated using equation (7). c ^ and Q c In order to track the deviation of the phase error in the low-speed range, the estimated value Δθ c _ L is automatically adjusted using equation (8) to obtain the estimated value Δθ c _ L By using this in the frequency and phase estimation calculation unit 9, it is possible to reduce sensitivity to the resistance value and improve the control characteristics.

[0054] The control characteristics in the low speed range according to this embodiment are shown in FIG. *With the setting of / R=0.5, the low-speed phase error estimation calculation unit 9 and the frequency and phase estimation calculation unit 10 are operated, and a load torque similar to that in FIG. 4 is applied. The estimated value Δθ of the phase error is calculated based on the reactive power, which is insensitive to the resistance value R. c _ L To calculate R * Even with a setting of / R=0.5, the actual phase error Δθ can be suppressed to zero.

[0055] Furthermore, in FIG. 7, the frequency command value ω r * The control characteristics are shown when ω is accelerated from 2% to 20% of the base frequency and then decelerated from 20% to 2%. r * is 10% and switches the estimated phase error between low speed and medium speed ranges. ω r * In the low-speed range where the phase error is less than 10%, the estimated phase error is calculated using equations (6) to (8), and in the medium-to-high-speed range where the phase error is 10% or more, the estimated phase error is calculated using equation (5).

[0056] In the figure, the C region switches from the low speed region to the medium to high speed region, and the D region switches from the medium to high speed region to the low speed region. Looking at the phase error Δθ in the lower part, the magnitude changes slightly depending on the timing of the switch, but the motor frequency ω r There was no shock, and it can be seen that the effect of the embodiment of the present invention is clear.

[0057] In this embodiment, ω r * The estimated value of the phase error in the low speed range and the medium to high speed range is switched at a magnitude of 10%, but there is no problem if the value is greater than zero and less than 10%. c _ L The taper gain, which changes between "1" and "0", is G _L , the estimated phase error in the medium to high speed range Δθ c _ H The taper gain G changes between "0" and "1". _H The average phase error estimate is multiplied by Δθ c It is also possible to use the following.

[0058] Here, a verification method when this embodiment is adopted will be described with reference to Fig. 8. A voltage detector 21 and a current detector 22 are attached to a power conversion device 20 that drives a magnet motor 1, and an encoder 23 is attached to the shaft of the magnet motor 1. The power conversion device 20 is composed of, for example, a power converter 2, a digital operator 20b (a user interface that serves both as an input device and a display device), etc.

[0059] The voltage and current vector component calculation unit 24 receives the three-phase AC voltage detection value (v uc , v vc , v wc ), three-phase AC current detection value (i uc , i vc , i wc ) and the encoder output position θ are input, and the vector voltage component v dc , v qc , vector current component i dc , i qc and the detected value ω obtained by differentiating the position θ rc Calculate the following.

[0060] In the observation section 25 of each waveform, the phase error Δθ_ is calculated using equation (11). cal Calculate the following.

[0061]

number

[0062] The parameters to be set in the controller of power converter 2 (R * ,L d * ,L q * ,K e * ) to change the size of Δθ_ in equation (11). cal If it coincides with the actual phase error Δθ, it is clear that the present invention is employed.

[0063] The main features of the first embodiment can be summarized as follows.

[0064] As shown in FIG. 3, the processor of the power conversion device 20 calculates the voltage of the magnet motor 1 (voltage command value v dc ** , v qc ** ) and current (current detection value i dc , i qc ) to the first power (first reactive power Q c ) (Equation (6)). The processor calculates the electric circuit parameters (L d * , L q * , K e * ) and the steady-state component of the current of magnet motor 1 (current detection value i dc , i qc ) and the transient component dt / d(i dc ), dt / d(i qc ) and the estimated frequency value ω of magnet motor 1 dc and the second power (second reactive power Q c ^ ) (Equation (7)). The processor adjusts the control phase θ dc and the phase of the magnetic flux of the magnet motor (magnet phase θ d 4, the processor estimates the phase deviation (phase error Δθ) indicating the deviation (difference) from the phase deviation (Δθ c _ L , Δθ c _ H ) to the frequency estimate ω dc In this embodiment, the first power and the second power are reactive powers.

[0065] Since the second power includes a transient component of the current of the magnet motor 1, the phase deviation (phase error Δθ) and the frequency estimate value ω dcThis ensures the accuracy of the frequency estimation. As a result, stable and highly accurate control characteristics can be realized without adjusting the electrical circuit parameters or control gains of the magnet motor 1. Furthermore, even if the responsiveness of the speed control is improved, the vibration of the current (q-axis current) when the frequency of the magnet motor 1 is switched is suppressed. This makes it possible to suppress torque shock. Since the first power and the second power do not include the winding resistance value, the phase deviation (phase error Δθ) and the frequency estimated value ω caused by the change in the winding resistance value associated with the temperature change in the low speed range can be reduced. dc This makes it possible to achieve highly accurate control of the magnet motor 1 regardless of temperature changes.

[0066] In detail, as shown in FIG. 3, the processor calculates the difference between the product of the voltage command value and the current detection value of the different components of the d-axis, which is the magnetic flux axis of the magnet motor 1, and the q-axis, which is the torque axis of the magnet motor 1 (v dc ** i qc -v qc ** i dc ) to the first power (first reactive power Q c The processor calculates the electric circuit parameters (L d * , L q * , K e * ) and the steady-state components of the d-axis and q-axis current detection values ​​or current command values ​​(current detection value i dc , i qc ) and the transient component dt / d(i dc ), dt / d(i qc ) and the frequency estimate ω dc and the second power (second reactive power Q c ^ ) is calculated. This makes it possible to realize stable and highly accurate control of the magnet motor 1 in a vector control power conversion device.

[0067] In this embodiment, the processor calculates the first power (first reactive power Q c ) and the second power (the second reactive power Q c ^ ) deviation of the reactive power (ΔQc The estimated value Δθ of the phase deviation (phase error Δθ) is calculated by performing proportional and integral control so that c_L This calculates the estimated value Δθ of the phase deviation (phase error Δθ) based on the power deviation. c_L can be calculated.

[0068] As shown in FIG. 3, the processor calculates a frequency command value ω r * In the low-speed range where is less than the threshold, the power deviation of reactive power (deviation ΔQ c The estimated value of the phase deviation Δθ is calculated by performing proportional and integral control so that c_L As shown in FIG. 2, the processor calculates a frequency command value ω r * In the medium to high speed range where is equal to or greater than the threshold, the estimated value Δθ of the phase deviation is calculated by the extended induced voltage method. c _ H This makes it possible to realize stable and highly accurate control of the magnet motor 1 over the entire speed range.

[0069] <Second Example> FIG. 9 is a configuration diagram of a power conversion device according to an embodiment.

[0070] In the first embodiment, the phase error estimation calculation unit 9 for the low speed range is set to d c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc to the first reactive power Q c In this embodiment, the amplitude value V of the three-phase AC voltage command is calculated. 1 * and the amplitude value of the current detection value i 1 and phase θ vi The first reactive power Q c The reference numerals 1 to 8, 10, and 11 in this figure are the same as those in FIG.

[0071] Fig. 10 is a block diagram of a phase error estimation calculation unit 9a in the low speed range according to an embodiment of the present invention. This figure corresponds to the reference numeral 9 in Fig. 1. Reference numerals 9a2, 9a3, 9a4, and 9a5 in Fig. 10 are the same as the reference numerals 92, 93, 94, and 95 in Fig. 3. In this figure, 9a1 indicates the amplitude value V 1 * (12), the amplitude value of the current detection value i 1 (13), and the phase θ vi is calculated from equation (14), and the reactive power Q c Calculate the following.

[0072]

number

[0073]

number

[0074]

number

[0075]

number

[0076] Even if this embodiment, which is an AC quantity, is used, it is possible to realize highly accurate control characteristics, similar to the first embodiment.

[0077] The main features of the second embodiment can be summarized as follows.

[0078] As shown in FIG. 10, the processor of the power conversion device 20 calculates the voltage amplitude value of one phase of the three-phase AC (the amplitude value V 1 * ) and the current amplitude value of that phase (amplitude value of current detection value i 1 ) and the sine signal sin[θ vi] to obtain the first power (first reactive power Q c ) (Equation (15)). The processor calculates the electric circuit parameters (L d * , L q * , K e * ) and the steady-state components of the d-axis and q-axis current detection values ​​or current command values ​​(current detection value i dc , i qc ) and the transient component dt / d(i dc ), dt / d(i qc ) and the frequency estimate ω dc and the second power (second reactive power Q c ^ ) is calculated. Stable and highly accurate control of the magnet motor 1 can be achieved based on the three-phase AC sine signal.

[0079] <Third Example> FIG. 11 is a configuration diagram of a power conversion device according to an embodiment.

[0080] In the first embodiment, the high-speed range phase error estimation calculation unit 8 calculates the estimated value Δθ of the phase error in the medium to high speed range according to the formula (5). c_H In the third embodiment, c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc to the first effective power P c This is a method of calculating the following.

[0081] 12 shows a phase error estimation calculation unit 8a for medium to high speed ranges according to an embodiment of the present invention. The figure corresponds to the reference numeral 8 in FIG 1. The reference numerals 1 to 7 and 9 to 11 in the figure are the same as those in FIG 1.

[0082] In the phase error estimation calculation unit 8a in the figure, the first active power calculation unit 8a1 calculates d c axis and q c Axis voltage command value v dc** , v qc ** and the current detection value i dc , i qc The first active power P c is calculated according to equation (16).

[0083]

number

[0084] The second active power calculation unit 8a2 calculates d c axis and q c Steady-state component of the axis current detection value i dc , i qc , transient component dt / d(i dc ), dt / d(i qc ), frequency estimate ω dc , the setting value of the electric circuit parameters of the magnet motor 1 (R * , L d * , L q * , K e * ) to obtain the second active power P c ^ is calculated according to equation (17).

[0085]

number

[0086] The subtraction unit 8a3 calculates the first active power P c and the second active power P c ^ is input, and the deviation is ΔP c Calculate the deviation of active power ΔP c is input to the PI control calculation unit 8a5 so as to follow the command value 8a4 of "0", and the estimated value Δθ of the phase error Δθ in the high-speed range is calculated according to the equation (18) by P (proportional) + I (integral) control. c_H Calculate the following.

[0087]

number

[0088] Here, K. pθ : Proportional gain of phase error estimation calculation, K iθ : is the integral gain of the phase error estimation calculation.

[0089] The main features of the third embodiment can be summarized as follows.

[0090] As shown in FIG. 12, the processor of the power conversion device 20 calculates the sum of the products of the voltage command values ​​and the current detection values ​​of the same components of the d-axis, which is the magnetic flux axis of the magnet motor 1, and the q-axis, which is the torque axis of the magnet motor 1 (v dc ** i dc +v qc ** i qc ) to the first power (first active power P c ) (Equation (16)). The processor calculates the electric circuit parameters (R * , L d * , L q * , K e * ) and the steady-state components of the d-axis and q-axis current detection values ​​or current command values ​​(current detection value i dc , i qc ) and the transient component dt / d(i dc ), dt / d(i qc ) and the frequency estimate ω dc and the second power (second active power P c ^ ) is calculated (Equation (17)). In this embodiment, the first power and the second power are active powers.

[0091] As a result, in a vector-controlled power conversion device, the estimated value Δθ of the phase deviation in the medium-to-high speed range is calculated based on the effective power instead of the conventional extended induced voltage method. c _ H can be calculated.

[0092] As in the first embodiment, the processor calculates a frequency command value ω r* In the low-speed range where is less than the threshold, the power deviation of reactive power (deviation ΔQ c The estimated value of the phase deviation Δθ is calculated by performing proportional and integral control so that c_L As shown in FIG. 12, the processor calculates the frequency command value ω r * In the medium to high speed range where is equal to or greater than the threshold, the power deviation of the active power (deviation ΔP c The estimated value Δθ of the phase deviation (phase error Δθ) is calculated by performing proportional and integral control so that c_H Calculate the following.

[0093] This makes it possible to realize stable and highly accurate control of the magnet motor 1 over the entire speed range.

[0094] <Fourth Example> FIG. 13 is a configuration diagram of a power conversion device according to an embodiment.

[0095] In the third embodiment, d c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc to the first active power P c In the fourth embodiment, the amplitude value V of the three-phase AC voltage command is calculated. 1 * and the amplitude value of the current detection value i 1 and phase θ vi The first active power P c This is a method of calculating the following.

[0096] 14 shows a phase error estimation calculation unit 8b in the medium to high speed range according to an embodiment of the present invention. This figure corresponds to reference numeral 8a in FIG 2. Reference numerals 1 to 7 and 9 to 11 in this figure are the same as those in FIG 1.

[0097] Reference characters 8b2, 8b3, 8b4, and 8b5 in Fig. 14 are the same as reference characters 8a2, 8a3, 8a4, and 8a5 in Fig. 12. In Fig. 14, 8b1 indicates the amplitude value V 1 * The amplitude value of the current detection value i 1 (13) above, and the phase θ vi is calculated from the above equation (14), and the effective power P c Calculate the following.

[0098]

number

[0099] Similarly to the first embodiment, this embodiment can also realize highly accurate control characteristics.

[0100] The main features of the fourth embodiment can be summarized as follows.

[0101] As shown in FIG. 14, the processor of the power conversion device 20 calculates the voltage amplitude value of one phase of the three-phase AC (the amplitude value V 1 * ) and the current amplitude value of that phase (amplitude value of current detection value i 1 ) and the cosine signal of the phase difference between the voltage command value and the current detection value of that phase, cos[θ vi ] to obtain the first power (first active power P c ) (Equation (19)). The processor calculates the electric circuit parameters (R * , L d * , L q * , K e * ) and the steady-state components of the d-axis and q-axis current detection values ​​or current command values ​​(current detection value i dc , i qc ) and the transient component dt / d(i dc ), dt / d(i qc ) and the frequency estimate ω dc and the second power (second active power P c ^) is calculated. Based on the three-phase AC cosine signal, stable and highly accurate control of the magnet motor 1 can be achieved.

[0102] <Fifth Example> FIG. 15 is a configuration diagram of a power conversion device according to an embodiment.

[0103] The first to fourth embodiments are configured to set the electrical circuit parameters of the magnet motor 1 in the controller (such as a microcomputer) of the power converter 2, but this embodiment uses a method in which the control state quantities are fed back to a higher-level IOT controller and the machine-learned electrical circuit parameters are reset in the power converter controller.

[0104] Reference numerals 1 to 11 in this figure are the same as those in Figure 1. Reference numeral 12 is an IOT controller that executes machine learning.

[0105] In this embodiment, the voltage command value v dc ** , v qc ** and the current detection value i dc , i qc , the estimated phase error Δθ c The electric circuit parameters (R * , L d * , L q * , K e * ) in the controller of the power converter 2.

[0106] Similarly to the first embodiment, this embodiment can also achieve more stable and highly accurate control characteristics.

[0107] The main features of the fifth embodiment can be summarized as follows.

[0108] The processor of the power conversion device 20 calculates the voltage command value v dc ** , v qc **and the current detection value i dc , i qc and the estimated phase deviation Δθ c The above information is transmitted to the controller (IOT controller) of the higher-level device via the communication device.

[0109] The controller of the higher-level device analyzes (machine learning) the values ​​received from the power electronics device 20 and transmits the analysis results to the power electronics device 20.

[0110] As shown in FIG. 15, the processor of the power conversion device 20 calculates a voltage command value v dc ** , v qc ** and the current detection value i dc , i qc and the estimated phase deviation Δθ c The d-axis and q-axis inductances L of the magnet motor analyzed (by machine learning) based on d * , L q * or induced voltage coefficient K e * The power conversion device 20 receives the electric circuit parameters stored in the storage device of the power conversion device 20 from the controller (IOT controller) of the higher-level device via the communication device. The processor of the power conversion device 20 updates the electric circuit parameters stored in the storage device of the power conversion device 20 with the received values. This makes it possible to automatically update the electric circuit parameters.

[0111] <Sixth Example> 16 is a configuration diagram of a power converter according to an embodiment. This embodiment is applied to a magnet motor drive system. In the figure, components 1 and 5 to 11 are the same as those in FIG.

[0112] The magnet motor 1, which is a component of FIG. 1, is driven by a power conversion device 20. In the power conversion device 20, the reference numerals 5 to 11 in FIG. 1 are implemented as software 20a, and the reference numerals 2, 3, and 4 in FIG. 1 are implemented as hardware. In addition, the "low speed range / medium-high speed range switching frequency 26, ω" of the software 20a is controlled by a higher-level device such as a digital operator 20b, a personal computer 28, a tablet 29, or a smartphone 30. chg ", "ω c " can be set and changed.

[0113] If this embodiment is applied to a magnet motor drive system, it is possible to realize highly accurate control characteristics in position sensorless vector control. chg ", "ω c " may be configured on a fieldbus such as a programmable logic controller, a local area network connecting to a computer, or an IOT controller.

[0114] Furthermore, although the first embodiment has been disclosed in this embodiment, the second to fifth embodiments may also be used.

[0115] The main features of the sixth embodiment can be summarized as follows.

[0116] The power conversion device 20 shown in FIG. chg ) and the control response 27(ω c The power conversion device 20 includes a storage device (e.g., a non-volatile memory) that stores the thresholds (switching frequency 26, ω chg ) and control response 27(ω c The device is equipped with an input device (such as a digital operator 20b) for setting thresholds and control responses, or a communication device for communicating with an external device (such as a personal computer 28, a tablet 29, a smartphone 30, etc.) for setting thresholds and control responses.

[0117] This allows the user to set the threshold (switching frequency 26, ω chg ) and control response 27(ω c etc.) can be easily set.

[0118] In the first to fifth embodiments described above, the first reactive power Q c Equation (6) and the second reactive power Q c ^ In equation (7), the current detection value i dc , i qc was used, but the current command value i d *, i q * may also be used. c (16) and the second reactive power P c ^ In equation (17), the current detection value i dc , i qc was used, but the current command value i d * , i q * may also be used.

[0119] Furthermore, in the first to fifth embodiments, the current command value i d * , i q * and the current detection value i dc , i qc to voltage correction value Δv dc , Δv qc The voltage correction value and the voltage reference value for vector control are added to the equation (4), and the current command value i d * , i q * and the current detection value i dc , i qc The intermediate current command value i shown in equation (20) used for vector control calculation is d ** , i q ** and calculate the frequency estimate ω dc And, the electric circuit parameters of the magnet motor 1 may be used to perform a vector control calculation shown in the formula (21).

[0120]

number

[0121]

number

[0122] Here, K. pd1 :d c Axis current control proportional gain, K id1 :d c Axis current control integral gain, K pq1 :q c Axis current control proportional gain, K iq1 :q c Axis current control integral gain, T d : Electrical time constant of d axis (L d / R), T q : q-axis electrical time constant (L q / R).

[0123] Or the current command value i d * , i q * and the current detection value i dc , i qc d used for vector control calculation c Voltage correction value Δv of the proportional calculation component of the axis d_p * , d c Voltage correction value Δv of the integral calculation component of the axis d_i * , q c Voltage correction value Δv of the proportional calculation component of the axis q_p * , q c Voltage correction value Δv of the integral calculation component of the axis q_i * is created using equation (22), and the frequency value estimation ω dc Moreover, a vector control calculation shown in equation (23) using the electric circuit parameters of the magnet motor 1 may be performed.

[0124]

number

[0125]

number

[0126] Here, K pd2 :d c Axis current control proportional gain, K id2 :d c Axis current control integral gain, K pq2 :q c Axis current control proportional gain, K iq2 :q c The integral gain of the axis current control, also called d c Axis current command value i d * and q c Axis current detection value i qc First-order lag signal i qctd , the frequency estimate ω dc Then, the electric circuit parameters of the magnet motor 1 may be used to perform a vector control calculation shown in equation (24).

[0127]

number

[0128] In the first to sixth embodiments, the switching elements constituting the power converter 2 may be Si (silicon) semiconductor elements or wide band gap semiconductor elements such as SiC (silicon carbide) and GaN (gallium nitride).

[0129] The above-mentioned embodiment is an example for explaining the present invention, and has been omitted and simplified as appropriate for the sake of clarity. The present invention can be implemented in various other forms. Unless otherwise limited, each component may be singular or plural. The position, size, shape, range, etc. of each component shown in the drawings may not represent the actual position, size, shape, range, etc. in order to facilitate understanding of the invention. Therefore, the present invention is not necessarily limited to the position, size, shape, range, etc. disclosed in the drawings. As examples of various information, expressions such as "table", "list", and "queue" may be used, but various information may be expressed in other data structures. For example, various information such as "XX table", "XX list", and "XX queue" may be expressed as "XX information". When explaining identification information, expressions such as "identification information", "identifier", "name", "ID", and "number" are used, but these are mutually interchangeable. When there are multiple components having the same or similar functions, they may be explained by adding different subscripts to the same symbol. In addition, when it is not necessary to distinguish between these multiple components, the subscripts may be omitted. In the embodiment, a process performed by executing a program may be explained. Here, the computer executes the program using a processor (e.g., CPU, GPU), and performs processing defined by the program while using storage resources (e.g., memory) and interface devices (e.g., communication ports). Therefore, the subject of the processing performed by executing the program may be the processor. Similarly, the subject of the processing performed by executing the program may be a controller, device, system, computer, or node having a processor. The subject of the processing performed by executing the program may be a calculation unit, and may include a dedicated circuit that performs specific processing. Here, the dedicated circuit is, for example, a Field Programmable Gate Array (FPGA), an Application Specific Integrated Circuit (ASIC), or a Complex Programmable Logic Device (CPLD). The program may be installed in the computer from a program source.The program source may be, for example, a program distribution server or a storage medium readable by a computer. When the program source is a program distribution server, the program distribution server may include a processor and a storage resource for storing a program to be distributed, and the processor of the program distribution server may distribute the program to be distributed to other computers. In addition, in the embodiment, two or more programs may be realized as one program, and one program may be realized as two or more programs.

[0130] The present invention is not limited to the above-described embodiments, and includes various modified examples. For example, the above-described embodiments have been described in detail to clearly explain the present invention, and the present invention is not necessarily limited to those including all of the configurations described. In addition, it is possible to replace a part of the configuration of one embodiment with the configuration of another embodiment, and it is also possible to add the configuration of another embodiment to the configuration of one embodiment. In addition, it is possible to add, delete, or replace a part of the configuration of each embodiment with another configuration.

[0131] The embodiment of the present invention may have the following aspects.

[0132] (C1) A power conversion device that controls the output frequency, output voltage, and output current of a magnet motor, characterized in that the power conversion device estimates a deviation between the control phase and the phase of the magnetic flux of the motor so that a first power calculated from the output voltage and output current of the magnet motor tracks a second power calculated from the electric circuit parameters of the magnet motor, steady-state components and transient components of the output current, and a frequency estimate value, and controls the frequency estimate value using the estimated value of the phase deviation.

[0133] (C2) A power conversion device that controls the output frequency, output voltage, and output current of a magnet motor, characterized in that a deviation between the control phase and the phase of the magnetic flux of the motor is estimated and the estimated value of the phase deviation is used to control the frequency estimate so that a first reactive power calculated from the output voltage and output current of the magnet motor follows a second reactive power calculated from the electric circuit parameters of the magnet motor, the steady-state components and transient components of the output current, and a frequency estimate.

[0134] (C3) A power conversion device that controls the output frequency, output voltage, and output current of a magnet motor, characterized in that a deviation between the control phase and the phase of the magnetic flux of the motor is estimated and the estimated value of the phase deviation is used to control the frequency estimate so that a first active power calculated from the output voltage and output current of the magnet motor tracks a second active power calculated from the electric circuit parameters of the magnet motor, the steady-state components and transient components of the output current, and a frequency estimate.

[0135] (C4) In a vector control power conversion device that calculates voltage command values ​​for the d-axis and q-axis using the current command value, current detection value, and frequency estimate value of the d-axis, which is the magnetic flux axis, and the q-axis, which is the torque axis, of a magnet motor, the power conversion device estimates the deviation between the control phase and the phase of the magnetic flux of the motor so that a first reactive power obtained by multiplying the voltage command values ​​and current detection values ​​of different components of the d-axis and q-axis and adding them (expressed as addition taking into account the positive and negative of the values) follows a second reactive power calculated from the electrical circuit parameters of the magnet motor, the steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimate value, and controls the frequency estimate value using the estimated value of the phase deviation.

[0136] (C5) A vector control power conversion device that calculates a three-phase AC voltage command value using a three-phase AC current command value, current detection value, and frequency estimate value of a magnet motor, characterized in that a deviation between the control phase and the phase of the magnetic flux of the motor is estimated so that a first reactive power obtained by multiplying the voltage amplitude values ​​and current amplitude values ​​of one phase of the three-phase AC and a sine signal of the phase difference between the voltage command value and the current detection value follows a second reactive power calculated from the electric circuit parameters of the magnet motor, the steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimate value, and the power conversion device controls the frequency estimate value using the estimated value of the phase deviation.

[0137] (C6) A vector control power conversion device that calculates d-axis and q-axis voltage command values ​​using the current command value, current detection value, and frequency estimate value of the d-axis, which is the magnetic flux axis, and the q-axis, which is the torque axis, of a magnet motor, characterized in that the power conversion device estimates the deviation between the control phase and the phase of the magnetic flux of the motor so that a first active power obtained by multiplying and adding up the voltage command value and current detection value of the same components of the d-axis and q-axis follows a second active power calculated from the electric circuit parameters of the magnet motor, the steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimate value, and controls the frequency estimate value using the estimated value of the phase deviation.

[0138] (C7) A vector control power conversion device that calculates a three-phase AC voltage command value using a three-phase AC current command value, current detection value, and frequency estimate value of a magnet motor, characterized in that a phase error (phase deviation), which is the deviation between the control phase and the phase of the magnetic flux of the motor, is estimated so that a first active power obtained by multiplying the voltage amplitude values ​​and current amplitude values ​​of one phase of the three-phase AC and a cosine signal of the phase difference between the voltage command value and the current detection value, follows a second active power calculated from the electric circuit parameters of the magnet motor, the steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimate value, and the frequency estimate value is controlled.

[0139] (C8) A power conversion device according to any one of (C1) to (C7), characterized in that the phase error estimation calculation estimates a phase error, which is the deviation between the control phase and the phase of the magnetic flux of the motor, by proportional control and integral control so as to make the deviation between the first and second reactive powers, or the deviation between the first and second active powers, zero.

[0140] (C9) A power conversion device characterized in that, in the power conversion device of (C8), a phase error is estimated and calculated by performing proportional control and integral control so that the deviation between the first and second reactive power information is set to zero when the magnet motor is in a low speed range, and the deviation between the first and second active power information is set to zero when the magnet motor is in a medium to high speed range, based on the relationship between the electrical circuit parameters of the magnet motor and the frequency estimation value.

[0141] (C10) A power conversion device characterized in that, in the power conversion device of (C8), proportional control and integral control are performed so that the deviation between the first and second reactive power information is zero when the magnet motor is in the low speed range, based on the relationship between the electrical circuit parameters of the magnet motor and the frequency estimated value, and when the magnet motor is in the medium to high speed range, the power conversion device directly calculates the phase error using an extended induced voltage method.

[0142] (C11) A power conversion device characterized in that, in the power conversion device of (C9) and (C10), the control response set for the frequency value for switching between low-speed and medium-high-speed control and the proportional control or integral control for estimating a phase error is set in the internal memory of a microcomputer mounted in the power conversion device including a power converter, and can be freely set and changed by connecting a digital operator, a personal computer, a tablet, or a smartphone device.

[0143] (C12) A power conversion device characterized in that, in the power conversion device of (C11), the voltage command value, current detection value, and estimated value of phase error are fed back to an IOT controller, which is a higher-level device, for analysis, and the d-axis and q-axis inductances and induced voltage coefficient of the magnet motor are automatically corrected.

[0144] According to (C1)-(C12), by estimating a phase error using reactive power calculated using the steady-state component and transient component of the current detection value and estimating the output frequency of the magnet motor using this phase error, it becomes possible to prevent torque shock when switching from a low speed range to a medium to high speed range, and a power converter can be provided that achieves highly accurate control characteristics without adjusting the electrical circuit parameters and control gains of the magnet motor that are set in the controller. [Explanation of symbols]

[0145] 1...Magnet motor, 2...Power converter, 3...DC power supply, 4...Current detector, 5...Coordinate conversion section, 6...Speed ​​control calculation section, 7...Vector control calculation section, 8, 8a, 8b...Medium-high speed range phase error estimation calculation section, 9, 9a, 9b...Low speed range phase error estimation calculation section, 10...Frequency and phase estimation calculation section, 11...Coordinate conversion section, 12...IOT controller, 20...Power conversion device, 20a...Power conversion device software (software section), 20b...Power conversion device digital operator, 21...Voltage detector, 22...Current detector, 23...Encoder, 24...Vector component voltage and current calculation section, 25...Waveform observation section for each section, 26...Low speed range / medium-high speed range switching frequency setting value, 27...Low speed range control response setting value, 28...Personal computer, 29...Tablet, 30...Smartphone, i d * …d-axis current command value, i q * …q-axis current command value, i dc …d-axis current detection value (steady-state component), i dc …q-axis current detection value (steady-state component), d / dt(i dc )…Transient component of the d-axis current detection value, d / dt(i qc )…Transient component of the q-axis current detection value, ω dc …frequency estimate, ω r …speed of magnet motor 1, v dc * , v dc ** , v dc ** , v dc *** , v dc **** , vdc ***** …d-axis voltage command value, v qc * , v qc ** , v qc *** , v qc **** , v qc ***** …q-axis voltage command value, Q C …First reactive power, Q C ^ …Second reactive power, P C …First active power, P C ^ …Second active power, Δθ c_L …Estimated phase error in the low-speed range, Δθ c_H …Estimated phase error in the medium to high speed range, Δθ c …Estimated phase error

Claims

1. Calculating a first power from a voltage and a current of the magnet motor; Calculating a second power from electrical circuit parameters of the magnet motor, steady-state and transient components of a current of the magnet motor, and an estimated frequency value of the magnet motor; estimating a phase deviation indicating a deviation between a control phase and a phase of a magnetic flux of the magnet motor so that the first power follows the second power; A power conversion apparatus comprising a processor for calculating the frequency estimate from the phase deviation estimate.

2. The power conversion device according to claim 1, The first power and the second power are reactive powers. A power conversion device comprising:

3. The power conversion device according to claim 1, The first power and the second power are active powers. A power conversion device comprising:

4. The power conversion device according to claim 2, The processor, calculating the first power from a difference between a product of a voltage command value and a current detection value of a different component of a d-axis which is a magnetic flux axis of the magnet motor and a q-axis which is a torque axis of the magnet motor; The second power is calculated from the electric circuit parameters, steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimation value. A power conversion device comprising:

5. The power conversion device according to claim 2, The processor, calculating the first power from a product of a voltage amplitude value of one phase of a three-phase AC, a current amplitude value of that phase, and a sine signal of a phase difference between a voltage command value and a current detection value of that phase; The second power is calculated from the electric circuit parameters, steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimation value. A power conversion device comprising:

6. The power conversion device according to claim 3, The processor, calculating the first power from a sum of products of voltage command values ​​and current detection values ​​of the same components of a d-axis which is a magnetic flux axis of the magnet motor and a q-axis which is a torque axis of the magnet motor; The second power is calculated from the electric circuit parameters, steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimation value. A power conversion device comprising:

7. The power conversion device according to claim 3, The processor, calculating the first power from a product of a voltage amplitude value of one phase of a three-phase AC, a current amplitude value of that phase, and a cosine signal of a phase difference between a voltage command value and a current detection value of that phase; The second power is calculated from the electric circuit parameters, steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimation value. A power conversion device comprising:

8. The power conversion device according to claim 1, The processor, A proportional control and an integral control are performed so as to make a power deviation indicating a deviation between the first power and the second power zero, thereby calculating an estimated value of the phase deviation. A power conversion device comprising:

9. The power conversion device according to claim 8, The processor, In a low speed region where a frequency command value is less than a threshold value, the proportional control and the integral control are performed so as to make the power deviation of reactive power zero, thereby calculating an estimate value of the phase deviation; In a medium to high speed range where the frequency command value is equal to or greater than the threshold value, an estimate of the phase deviation is calculated by performing the proportional control and the integral control so as to make the power deviation of the effective power zero. A power conversion device comprising:

10. The power conversion device according to claim 8, The processor, In a low speed region where a frequency command value is less than a threshold value, the proportional control and the integral control are performed so as to make the power deviation of reactive power zero, thereby calculating an estimate value of the phase deviation; In a medium to high speed range where the frequency command value is equal to or greater than the threshold value, an estimate of the phase deviation is calculated by an extended induced voltage method. A power conversion device comprising:

11. The power conversion device according to claim 9, a storage device that stores the threshold value and a control response used in the proportional control or the integral control; an input device for setting the threshold and the control response; or a communication device for communicating with an external device that sets the threshold and the control response; A power conversion device comprising:

12. The power conversion device according to claim 11, The processor, Transmitting a voltage command value, a current detection value, and the estimated value of the phase deviation to a controller of a higher-level device via the communication device; receiving, via the communication device, from a controller of the higher-level device, inductances of the d-axis and q-axis of the magnet motor or induced voltage coefficients analyzed based on the voltage command value, the current detection value, and the estimated value of the phase deviation; Update the electrical circuit parameters with the received values. A power conversion device comprising:

13. The power conversion device according to claim 1, The processor, The second power is calculated from the electric circuit parameters, steady-state components and transient components of the d-axis and q-axis current detection values ​​or current command values, and the frequency estimation value. A power conversion device comprising:

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