Welding power supply apparatus

Abstract

To provide a welding power supply device which can be adjusted to desired welding conditions.SOLUTION: In the welding power supply device, the power storage unit stores power when any one of the three switching elements is in a conductive state, and outputs a current from at least one switching element to a welding electrode in contact with an object to be welded when any one of the three switching elements is in a non-conductive state. A first signal generator generates, based on a reference clock, three pulses in which a phase of a switching control cycle in each phase is shifted by 120 degrees, and generates three first signals for pulse modulation corresponding to each phase from the generated three pulses. A second signal generation unit generates three second signals for adjusting a conduction time, which is a period of a conduction state of each switching element, based on the three first signals. The drive unit outputs a third signal for applying a predetermined voltage to each switching element to each switching element based on the three second signals, and brings each switching element into a conductive state or a non-conductive state.SELECTED DRAWING: Figure 2

Description

[Technical Field]

[0001] The present invention relates to a welding power supply. [Background technology]

[0002] Resistance welding is one of the metal welding methods. Resistance welding is a welding method in which electrodes sandwich the material to be welded and current is passed through them, generating heat due to the electrical resistance of the metal, which then melts the material. For example, there is a resistance welding machine that is equipped with multiple inverter-type welding power supplies to increase welding capacity and welding output (see Patent Document 1). [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Japanese Patent Application Publication No. 09-085458 Summary of the Invention [Problem to be solved by the invention]

[0004] Good welding quality depends on the characteristics of the materials to be welded. For example, when joining laminated metal foils using resistance welding, the welding quality is unstable due to variations in the properties and condition of the surfaces of the metal foils to be welded. Therefore, good welding quality is achieved by adjusting the welding conditions (e.g., current application time) according to the characteristics of the materials to be welded. Therefore, it is desirable for welding power supplies to be able to appropriately adjust the welding conditions.

[0005] The technique of the present disclosure aims to provide a welding power supply that can be adjusted to desired welding conditions. [Means for solving the problem]

[0006] The technology disclosed herein employs the following technical solutions to solve the above-mentioned problems. A welding power supply according to one aspect of the technology disclosed herein is a welding power supply used for resistance welding. The welding power supply includes a switching element, a power storage unit, a first signal generating unit, a second signal generating unit, and a drive unit. The switching elements are three switching elements corresponding to the first to third phases, respectively. The power storage unit stores power when any one of the three switching elements is in a conductive state, and when the power storage unit is in a non-conductive state, outputs current from at least one switching element to a welding electrode in contact with a workpiece. The first signal generating unit generates three pulses corresponding to each phase based on a reference clock, and generates three first signals for pulse modulation corresponding to each phase from the three generated pulses. Three second signal generating units are provided, one for each of the first to third phases. The second signal generating unit generates three second signals, based on the three first signals, to adjust the conduction time, which is the duration of the conduction of each switching element. Three drive units are provided corresponding to the first to third phases, respectively. Furthermore, the drive units output third signals to the switching elements based on the three second signals to apply predetermined voltages to the switching elements, thereby turning the switching elements into a conductive state or a non-conductive state.

[0007] The first signal generating unit is configured to generate three pulses with the phases of the switching control cycles for each phase shifted by 120 degrees. [Effects of the Invention]

[0008] According to one aspect of the technique of the present disclosure, it is possible to adjust the current application time to obtain a desired welding quality. In other words, the technique of the present disclosure can provide a welding power supply that can be adjusted to desired welding conditions. [Brief explanation of the drawings]

[0009] [Figure 1] 1 is a schematic configuration diagram of a welding machine according to a first embodiment. [Figure 2] 1 is a diagram showing an example of a circuit configuration of a welding power supply according to a first embodiment. [Figure 3] 2 is a diagram illustrating an example of the configuration of a three-phase carrier waveform generating circuit according to the first embodiment. FIG. [Figure 4] FIG. 2 is a diagram illustrating an example of a circuit configuration of a ring counter according to the first embodiment. [Figure 5] FIG. 3 is a diagram showing a timing chart of three-phase pulses according to the first embodiment. [Figure 6] 1 is a diagram illustrating an example of the configuration of a sawtooth wave generating circuit according to a first embodiment. [Figure 7] FIG. 3 is a diagram showing a timing chart of a three-phase sawtooth wave according to the first embodiment. [Figure 8] FIG. 2 is a diagram showing pulse waveforms of a three-phase interleaving system according to the first embodiment. [Figure 9] FIG. 2 is a diagram showing output waveforms of the three-phase interleaved welding power supply according to the first embodiment. [Figure 10] FIG. 2 is a diagram showing an output waveform of a single-phase inverter type welding power supply; [Figure 11] FIG. 10 is a flowchart showing a process for determining whether welding is good or bad in a welding power supply according to a modified example. [Figure 12] 10A is a diagram showing an example of an output waveform when the welding quality according to the modified example is acceptable, and FIG. 10B is a diagram showing an example of an output waveform when the welding quality according to the modified example is unacceptable. DETAILED DESCRIPTION OF THE INVENTION

[0010] Hereinafter, embodiments of the technology of the present disclosure will be described with reference to the drawings. The same or equivalent components, parts, and processes shown in each drawing will be denoted by the same reference numerals, and redundant descriptions will be omitted as appropriate.

[0011] First Embodiment (resistance welding machine) FIG. 1 is a schematic diagram of a welding machine 100 according to this embodiment, illustrating an example of a resistance welding machine. The welding machine 100 according to this embodiment primarily includes a welding power supply 10 and a welding head 90. The welding power supply 10 controls the current (welding current) and voltage (welding voltage) required for welding. The welding power supply 10 is electrically connected to the welding head 90 via predetermined conductive wires Wa and Wb. The welding head 90 primarily includes a drive unit (not shown), a pressure unit (not shown), and two electrodes 91a and 91b. The welding head 90 is driven by the current and voltage supplied from the welding power supply 10 as follows: The welding head 90 brings the two electrodes (welding electrodes) 91a and 91b into contact with a workpiece M placed between them, applies pressure to the workpiece M with a constant force, and passes a welding current through the workpiece M. The workpieces M are two or more metal materials (welding materials) that are not yet joined. Therefore, in the workpieces M, resistance heat (Joule heat) causes some of the metal materials to melt and solidify. As a result, the two or more metal materials are joined.

[0012] The welding machine 100 according to this embodiment is configured as described above and is capable of performing metal foil layer welding. Metal foil layer welding is characterized by inconsistent welding quality due to variations in the surface properties and conditions of the metal foil. Aluminum foil layer welding is an example of metal foil layer welding that has such characteristics. Aluminum foil layer welding is performed, for example, in the manufacturing process of conductive polymer aluminum electrolytic capacitors.

[0013] (Welding quality during resistance welding) The present inventors have conducted extensive research into the conditions for obtaining stable welding quality in the manufacturing process of conductive polymer aluminum electrolytic capacitors, etc. As a result, the present inventors have determined that in order to obtain stable welding quality, a welding current in the range of 500 A to 1000 A and a current application time of 30 msec or more (ensuring an appropriate current application time) are necessary.

[0014] (Difficulty in ensuring the appropriate power-on time) Linear control welding power supplies have a current control characteristic that causes the welding current to rise in a straight line. Such welding power supplies have a current flow time of 30 msec or less (for example, a maximum of 25 msec). Therefore, linear control welding power supplies cannot ensure an appropriate current flow time to achieve stable welding quality when welding aluminum foil layers. As a result, linear control welding power supplies lack the capacity of a welding power supply.

[0015] In a linear-control welding power supply, the control circuit controls the voltage applied to the gate (G) of a power metal-oxide-semiconductor field-effect transistor (MOSFET) so that the output voltage matches a preset target value. When the voltage applied to the gate (G) is low, the resistance between the drain (D) and source (S) of the power MOSFET increases. When the voltage applied to the gate (G) is high, the resistance between the drain (D) and source (S) of the power MOSFET decreases. In this way, a linear-control welding power supply changes the resistance between the drain (D) and source (S) of the power MOSFET, like a variable resistor, to track the output voltage to the target value. When this type of linear control is performed, conduction loss (conduction loss = ((storage voltage) - (output voltage)) x (output current)) occurs due to the resistance between the drain (D) and source (S) of the power MOSFET. Conduction loss can cause the power MOSFET to heat up and cause a voltage drop in the storage voltage. As a result, linearly controlled welding power supplies cannot output welding current for long periods of time. To extend the current flow time, it is possible to increase the number of parallel connections between power storage devices (capacitors) and power MOSFETs. However, this configuration increases the number of components required for the welding power supply. This increases the size, weight, and manufacturing costs of the welding power supply. Therefore, the proposed configuration is not suitable for mass-produced products.

[0016] (Difficulty in adjusting welding current) One of the factors that affect welding quality is the heat generated by resistance welding. Heat generated by resistance welding is calculated as (contact resistance) x (output current) 2 × (energization time). In other words, the amount of heat generated by resistance welding varies in proportion to the parameter values ​​of welding current and energization time (welding condition values). Therefore, in order to suppress the heat generated by resistance welding, it is necessary to fine-tune the values ​​of each parameter. Note that, because the amount of heat generated by resistance welding is proportional to the square of the welding current, it is difficult to achieve good welding quality by adjusting the welding current. Therefore, a preferred method for achieving high welding quality is to suppress the heat generated by resistance welding by adjusting the energization time, which is one of the welding conditions.

[0017] As mentioned above, linear control welding power supplies cannot ensure a long welding time, making it difficult to adjust the welding time to achieve good welding quality. Examples of welding power supplies other than linear control include single-phase inverter-type welding power supplies. These welding power supplies rectify input alternating current (AC) to direct current (DC), then convert it to high-frequency AC, and are configured to supply current at a constant value. Therefore, these welding power supplies include an inverter circuit with multiple switches that convert DC to AC by switching at a predetermined frequency. In such welding power supplies, the switching frequency is approximately 10 kHz. Therefore, the welding time in single-phase inverter-type welding power supplies is adjustable in increments of 100 μsec. Thus, the welding time adjustment unit for single-phase inverter-type welding power supplies is not suitable for fine adjustment, making it difficult to achieve good welding quality.

[0018] (Three-phase interleaved welding power supply) In view of the above, welding power supply 10 according to this embodiment is provided with three-phase switching nodes and employs an interleaved switching control method as a method for adjusting welding conditions. Welding power supply 10 operates three switching nodes at a fundamental frequency of 10 kHz and performs PWM (Pulse Width Modulation) control so as to shift the phase of the switching cycle of each switching node by 120 degrees.

[0019] As a result, welding power supply 10 according to this embodiment operates at 30 kHz as a whole, even though the fundamental frequency of each switching node is 10 kHz. Therefore, welding power supply 10 according to this embodiment can ensure an appropriate welding time, which is not possible with linear-control welding power supplies. Furthermore, welding power supply 10 can also fine-tune the welding time, which is not possible with single-phase inverter-type welding power supplies. In other words, welding power supply 10 can adjust to desired welding conditions and achieve good welding quality.

[0020] (Circuit configuration) FIG. 2 is a diagram showing an example of the circuit configuration of welding power supply 10 (a three-phase interleaved welding power supply) according to this embodiment. As shown in FIG. 2, welding power supply 10 mainly includes switching power supply 11, bulk capacitor 12, MOSFETs 13a-13c, power inductors 15a-15c, welding command circuit 16, feedback circuits 171-173, and error amplifier 18. Furthermore, welding power supply 10 includes three-phase carrier waveform generating circuit 20, PWM circuits 30a-30c, gate drive circuits 31a-31c, and active clamps 32a-32c. Furthermore, welding power supply 10 includes control circuit (CPU: Central Processing Unit) 70, ADC (AD converter) 60, and the like.

[0021] The switching power supply 11 is an inverter circuit that converts a single-phase alternating current voltage (AC: voltage in the range of 200 V to 240 V) into a direct current voltage (DC: 20 V). The switching power supply 11 is also connected to a bulk capacitor 12. The switching power supply 11 outputs the converted direct current voltage to the bulk capacitor 12. Therefore, the switching power supply 11 corresponds to a power supply circuit.

[0022] Bulk capacitor 12 is a high CV capacitor with a large capacity of, for example, about 1,200,000 μF (= 150,000 μF × 8 pieces). Bulk capacitor 12 is installed for the purpose of stabilizing the power line. The charge stored in bulk capacitor 12 is discharged during welding output.

[0023] The MOSFETs 13a to 13c are field-effect transistors, and are switching elements that turn on (conductive state) between the drain (D) and the source (S) when a voltage is applied between the gate (G) and the source (S). The MOSFETs 13a to 13c also correspond to switching nodes. FIG. 2 shows an example in which N-channel (N-Ch) MOSFETs 13a to 13c are used as switching nodes. The N-channel MOSFETs 13a to 13c turn on when a positive voltage with respect to the source (S) is applied to the gate (G).

[0024] Although the present embodiment shows an example in which MOSFETs are used as switching elements, the present invention is not limited to this, and semiconductor switches such as IGBTs (Insulated Gate Bipolar Transistors) may also be used as switching elements.

[0025] In this embodiment, assuming three-phase PWM control, three MOSFETs 13a to 13c are provided. Specifically, a MOSFET 13a corresponding to a first phase (A-phase), a MOSFET 13b corresponding to a second phase (B-phase), and a MOSFET 13c corresponding to a third phase (C-phase) are provided. In the following description, the first-phase MOSFET 13a will be referred to as the "A-phase MOSFET 13a," the second-phase MOSFET 13b will be referred to as the "B-phase MOSFET 13b," and the third-phase MOSFET 13c will be referred to as the "C-phase MOSFET 13c." Furthermore, the first- to third-phase MOSFETs 13a to 13c will be collectively referred to as "MOSFET 13n."

[0026] In the MOSFET 13n, the gate terminal (G terminal) is connected to the output terminal of the gate drive circuits 31a to 31c, the drain terminal (D terminal) is connected to the output terminal of the bulk capacitor 12, and the source terminal (S terminal) is connected to the input terminal of the power inductors 15a to 15c.

[0027] The MOSFETs 13n are switched based on an input signal to their gate terminals. Specifically, the A-phase MOSFET 13a is ON / OFF controlled (its conductive / non-conductive state is controlled) in accordance with an A-phase PWM signal input to its gate terminal from a first-phase (A-phase) gate drive circuit 31a. The B-phase MOSFET 13b is ON / OFF controlled in accordance with a second-phase (B-phase) PWM signal input to its gate terminal from a second-phase (B-phase) gate drive circuit 31b. The C-phase MOSFET 13c is ON / OFF controlled in accordance with a third-phase (C-phase) PWM signal input to its gate terminal from a third-phase (C-phase) gate drive circuit 31c. The first- to third-phase gate drive circuits 31a to 31c will be described later.

[0028] Diodes 14a to 14c are provided in paths between the source terminal of MOSFET 13n and the input terminals of power inductors 15a to 15c. Specifically, diode 14a corresponding to the first phase (A phase), diode 14b corresponding to the second phase (B phase), and diode 14c corresponding to the third phase (C phase) are provided. In the following description, first-phase diode 14a will be referred to as "A-phase diode 14a," second-phase diode 14b will be referred to as "B-phase diode 14b," and third-phase diode 14c will be referred to as "C-phase diode 14c." Furthermore, when referring to the first-phase to third-phase diodes 14a to 14c, they will be collectively referred to as "diodes 14n."

[0029] Diode 14n is a rectifier that unidirectionally directs the flow of electricity. Diode 14n is conductive when MOSFET 13n is OFF (non-conductive state) and supplies the welding current to workpiece M. Specifically, A-phase diode 14a is conductive when A-phase MOSFET 13a is OFF and supplies the welding current to workpiece M. B-phase diode 14b is conductive when B-phase MOSFET 13b is OFF and supplies the welding current to workpiece M. C-phase diode 14c is conductive when C-phase MOSFET 13c is OFF and supplies the welding current to workpiece M.

[0030] The diode 14n may be replaced with a MOSFET. For example, a synchronous rectification converter may be configured by inputting signals complementary to the gate signals of the MOSFETs 13a to 13c of phases A to C to the gate terminals of the MOSFETs after the replacement.

[0031] Power inductors 15a to 15c are provided in a path between the source terminal of MOSFET 13n and the connection terminal of conductive wire Wa that passes current to workpiece M. Specifically, power inductor 15a corresponding to the first phase (A phase), power inductor 15b corresponding to the second phase (B phase), and power inductor 15c corresponding to the third phase (C phase) are provided. In the following description, first-phase power inductor 15a will be referred to as "A-phase power inductor 15a," second-phase power inductor 15b will be referred to as "B-phase power inductor 15b," and third-phase power inductor 15c will be referred to as "C-phase power inductor 15c." Furthermore, power inductors 15a to 15c of the first to third phases will be collectively referred to as "power inductor 15n."

[0032] The input terminal of A-phase power inductor 15a is connected to the source terminal of A-phase MOSFET 13a. The input terminal of B-phase power inductor 15b is connected to the source terminal of B-phase MOSFET 13b. The input terminal of C-phase power inductor 15c is connected to the source terminal of C-phase MOSFET 13c. The output terminals of each of A-phase to C-phase power inductors 15a to 15c are connected together to the connection terminal of conductive wire Wa that passes current to workpiece M.

[0033] Power inductor 15n is a power storage device that charges the input current and discharges it at a predetermined timing to smooth the pulse waveform. Specifically, A-phase power inductor 15a charges the current that flows when A-phase MOSFET 13a is ON (conducting state) and discharges the current when it is OFF, thereby smoothing the A-phase waveform. B-phase power inductor 15b charges the current that flows when B-phase MOSFET 13b is ON and discharges the current when it is OFF, thereby smoothing the B-phase waveform. C-phase power inductor 15c charges the current that flows when C-phase MOSFET 13c is ON and discharges the current when it is OFF, thereby smoothing the C-phase waveform.

[0034] The power inductor 15n according to this embodiment has an inductance reduction rate of about 25% at an oscillation frequency of 10 kHz and a peak current of 500 A. In other words, this embodiment uses a power inductor with good DC superimposition characteristics that prevent a sudden drop in inductance value even when a large current flows.

[0035] The welding command circuit 16 is provided on a path between the input terminal of the error amplifier 18 and the output terminal of the control circuit 70 .

[0036] An input terminal of welding command circuit 16 is connected to an output terminal of control circuit 70. An output terminal of welding command circuit 16 is connected to an input terminal of error amplifier 18.

[0037] Welding command circuit 16 generates a reference voltage according to the current value, voltage value, and welding time in accordance with a welding command. The welding command is input from control circuit 70. Therefore, welding command circuit 16 is set based on the input signal from control circuit 70, and generates a reference voltage in accordance with the set welding command.

[0038] The feedback circuits 171 to 173 are circuits that perform target control by returning an output signal as an input signal. In this embodiment, three feedback circuits 171 to 173 are provided corresponding to constant current control, constant voltage control, and constant power control. Specifically, a current feedback circuit 171 corresponding to constant current control, a voltage feedback circuit 172 corresponding to constant voltage control, and a power feedback circuit 173 corresponding to constant power control are provided. Note that the feedback circuits 171 to 173 for constant current control, constant voltage control, and constant power control are collectively referred to as "feedback circuits 17n."

[0039] Feedback circuit 17n performs target current control, target voltage control, or target power control by inputting the output current, output voltage, or output power to error amplifier 18. Current feedback circuit 171 is provided in a path between the input terminal of error amplifier 18 and the connection terminal of CT (Current Transformer) 40. CT 40 is provided in a path between power inductor 15n and the connection terminal of conductive wire Wa that passes current to workpiece M. CT 40 is a current transformer that converts (transforms) a large current into a small current (low current).

[0040] The input terminal of the current feedback circuit 171 is connected to the connection terminal of the CT 40. The output terminal of the current feedback circuit 171 is connected to the input terminal of the error amplifier 18. The CT 40 converts the output current from the power inductor 15n and outputs the converted current to the current feedback circuit 171. The current feedback circuit 171 inputs the current converted by the CT 40 to the error amplifier 18.

[0041] Voltage feedback circuit 172 is provided in the path between the input terminal of error amplifier 18 and the connection terminal of RC filter 50. RC filter 50 is provided between the connection terminals of conductive wires Wa, Wb, which pass current to workpiece M. When the load end is in an open state (high impedance state), no current flows, and the output voltage oscillates between 0 V and the charging voltage. When this is fed back as voltage, the oscillation state does not stop, and the control system enters an oscillating state. RC filter 50 prevents this, smoothing the output when the load end is open, suppressing noise, and stabilizing the control system.

[0042] An input terminal of the voltage feedback circuit 172 is connected to a connection terminal of the RC filter 50. An output terminal of the voltage feedback circuit 172 is connected to an input terminal of the error amplifier 18. The RC filter 50 suppresses, for example, noise that appears in the output voltage during welding (for example, ripple that depends on the switching frequency) and outputs the suppressed voltage to the voltage feedback circuit 172. The voltage feedback circuit 172 inputs the voltage suppressed by the RC filter 50 to the error amplifier 18.

[0043] The power feedback circuit 173 is provided on a path between the input terminal of the error amplifier 18 and each output terminal of the current feedback circuit 171 and the voltage feedback circuit 172 .

[0044] An input terminal of the power feedback circuit 173 is connected to each output terminal of the current feedback circuit 171 and the voltage feedback circuit 172. An output terminal of the power feedback circuit 173 is connected to an input terminal of the error amplifier .

[0045] The power feedback circuit 173 inputs an output power based on the output current of the current feedback circuit 171 and the output voltage of the voltage feedback circuit 172 to the error amplifier 18 .

[0046] The circuits located after the feedback circuits 171 to 173 are provided with feedback signal switching circuits 17s.

[0047] The switching circuit 17s is a switching circuit that switches between three control methods: constant current control, constant voltage control, and constant power control, in accordance with a control method switching signal. The control method switching signal is input from the control circuit 70. Therefore, the control method is set in the switching circuit 17s based on the input signal from the control circuit 70, and the control method is switched in accordance with the set control method.

[0048] When the set control method is the constant current control method, switching circuit 17s connects the output terminal of current feedback circuit 171 to the input terminal of error amplifier 18, and switches the circuit connection so that a feedback signal of the output current is input to error amplifier 18. When the set control method is the constant voltage control method, switching circuit 17s connects the output terminal of voltage feedback circuit 172 to the input terminal of error amplifier 18, and switches the circuit connection so that a feedback signal of the output voltage is input to error amplifier 18. When the set control method is the constant power control method, switching circuit 17s connects the output terminal of power feedback circuit 173 to the input terminal of error amplifier 18, and switches the circuit connection so that a feedback signal of the output power is input to error amplifier 18.

[0049] The error amplifier 18 is provided on the path between the feedback circuit 17n and the PWM circuits 30a to 30c.

[0050] The input terminal of error amplifier 18 is connected to the output terminal of welding command circuit 16 and, via switching circuit 17s, to the output terminal of feedback circuit 17n. The output terminal of error amplifier 18 is connected to the input terminals of first-phase (A-phase) to third-phase (C-phase) PWM circuits 30a to 30c, respectively. The first-phase to third-phase PWM circuits 30a to 30c will be described later.

[0051] Error amplifier 18 is an error amplifier that compares a reference voltage (target value) generated in accordance with a welding command with a feedback signal and amplifies the error resulting from the comparison. Specifically, when the control method is constant current control, error amplifier 18 receives a feedback signal from current feedback circuit 171 and outputs the amplified error to each of PWM circuits 30a to 30c. When the control method is constant voltage control, error amplifier 18 receives a feedback signal from voltage feedback circuit 172 and outputs the amplified error to each of PWM circuits 30a to 30c. When the control method is constant power control, error amplifier 18 receives a feedback signal from power feedback circuit 173 and outputs the amplified error to each of PWM circuits 30a to 30c.

[0052] The three-phase carrier waveform generating circuit 20 is provided on the path between the control circuit 70 and the PWM circuits 30a to 30c.

[0053] The input terminals of the three-phase carrier waveform generating circuit 20 are connected to the output terminals of the control circuit 70. The output terminals of the three-phase carrier waveform generating circuit 20 are connected to the input terminals of the first phase (A phase) to third phase (C phase) PWM circuits 30a to 30c.

[0054] The three-phase carrier waveform generation circuit 20 is a first signal generation circuit that generates carrier clocks A to C (first signals, carriers A to C) of the first phase (A phase) to the third phase (C phase). Specifically, a control signal (ON / OFF control signal) that controls the output of the reference clock is input from the control circuit 70 to the three-phase carrier waveform generation circuit 20.

[0055] The three-phase carrier waveform generation circuit 20 generates three 10 kHz pulses, each with a switching control period (switching frequency) phase shifted by 120 degrees for each phase, based on a 60 kHz reference clock. The three-phase carrier waveform generation circuit 20 also generates three sawtooth waves for PWM modulation, one for each phase, from the generated pulses. Each sawtooth wave corresponds to the carrier clocks A to C for each phase.

[0056] The three-phase carrier waveform generation circuit 20 outputs the generated first-phase (A-phase) carrier clock A to the first-phase (A-phase) PWM circuit 30a. The three-phase carrier waveform generation circuit 20 also outputs the generated second-phase (B-phase) carrier clock B to the second-phase (B-phase) PWM circuit 30b. The three-phase carrier waveform generation circuit 20 also outputs the generated third-phase (C-phase) carrier clock C to the third-phase (C-phase) PWM circuit 30c. The three-phase carrier waveform generation circuit 20 will be described later with reference to FIG. 3.

[0057] The PWM circuits 30a to 30c are provided on paths between the gate drive circuits 31a to 31c and the error amplifier 18 and the three-phase carrier waveform generation circuit 20. In this embodiment, assuming three-phase PWM control, three PWM circuits 30a to 30c are provided. Specifically, the PWM circuits 30a correspond to the first phase (A phase), the PWM circuit 30b correspond to the second phase (B phase), and the PWM circuit 30c correspond to the third phase (C phase).

[0058] In the following description, the first-phase PWM circuit 30a will be referred to as the "A-phase PWM circuit 30a," the second-phase PWM circuit 30b will be referred to as the "B-phase PWM circuit 30b," and the third-phase PWM circuit 30c will be referred to as the "C-phase PWM circuit 30c." Furthermore, the first- to third-phase PWM circuits 30a to 30c will be collectively referred to as the "PWM circuit 30n."

[0059] An input terminal of the A-phase PWM circuit 30a is connected to the output terminal of the error amplifier 18 and an output terminal for outputting the carrier clock A of the three-phase carrier waveform generation circuit 20. An output terminal of the A-phase PWM circuit 30a is connected to the input terminal of the first-phase (A-phase) gate drive circuit 31a. An input terminal of the B-phase PWM circuit 30b is connected to the output terminal of the error amplifier 18 and an output terminal for outputting the carrier clock B of the three-phase carrier waveform generation circuit 20. An output terminal of the B-phase PWM circuit 30b is connected to the input terminal of the second-phase (B-phase) gate drive circuit 31b. An input terminal of the C-phase PWM circuit 30c is connected to the output terminal of the error amplifier 18 and an output terminal for outputting the carrier clock C of the three-phase carrier waveform generation circuit 20. An output terminal of the C-phase PWM circuit 30c is connected to the input terminal of the third-phase (C-phase) gate drive circuit 31c.

[0060] The PWM circuit 30n is a second signal generating circuit that adjusts the time (conduction state time: duty) that the MOSFET 13n is turned on, and outputs a PWM signal (second signal) for adjustment. PWM control is a control that changes the ratio (duty ratio) between the H level (ON) and L level (OFF) of a pulse signal. PWM control also performs repeated ON and OFF switching to generate a pulse waveform (pulse signal, i.e., PWM signal) with a constant ON and OFF cycle from a constant voltage, and changes the ON time width. Therefore, the PWM circuit 30n outputs a PWM signal to each of the gate drive circuits 31a to 31c, for changing the ON time width of the MOSFET 13n. Specifically, the A-phase PWM circuit 30a receives the amplified error from the error amplifier 18 and the carrier clock A from the three-phase carrier waveform generation circuit 20, and outputs an A-phase PWM signal to the first-phase (A-phase) gate drive circuit 31a to vary the ON duration of the A-phase MOSFET 13a. The B-phase PWM circuit 30b receives the amplified error from the error amplifier 18 and the carrier clock B from the three-phase carrier waveform generation circuit 20, and outputs a B-phase PWM signal to the second-phase (B-phase) gate drive circuit 31b to vary the ON duration of the B-phase MOSFET 13b. The C-phase PWM circuit 30c receives the amplified error from the error amplifier 18 and the carrier clock C from the three-phase carrier waveform generation circuit 20, and outputs a C-phase PWM signal to the third-phase (C-phase) gate drive circuit 31c to vary the ON duration of the C-phase MOSFET 13c.

[0061] With this configuration, the B-phase PWM signal is output based on the B-phase carrier clock B, whose switching cycle phase is shifted by 120 degrees relative to the A-phase carrier clock A. The C-phase PWM signal is output based on the C-phase carrier clock C, whose switching cycle phase is shifted by 120 degrees relative to the B-phase carrier clock B. The A-phase PWM signal is output based on the A-phase carrier clock A, whose switching cycle phase is shifted by 120 degrees relative to the C-phase carrier clock C.

[0062] As a result, the PWM signals for phases A to C are output at timings that are phase-shifted by 120 degrees. In this embodiment, the carrier clocks for phases A to C have a basic control frequency of 10 kHz. The phase of the output cycle of each carrier clock is shifted by 120 degrees for each phase. Therefore, welding power supply 10 according to this embodiment operates at a control frequency of 30 kHz.

[0063] The gate drive circuits 31a to 31c are provided in a path between the MOSFET 13n and the PWM circuit 30n. In this embodiment, assuming three-phase PWM control, three gate drive circuits 31a to 31c are provided. Specifically, the gate drive circuits 31a correspond to the first phase (A phase), the gate drive circuit 31b correspond to the second phase (B phase), and the gate drive circuit 31c correspond to the third phase (C phase).

[0064] In the following description, the first-phase gate drive circuit 31a will be referred to as the "A-phase gate drive circuit 31a," the second-phase gate drive circuit 31b will be referred to as the "B-phase gate drive circuit 31b," and the third-phase gate drive circuit 31c will be referred to as the "C-phase gate drive circuit 31c." Furthermore, the first-phase to third-phase gate drive circuits 31a to 31c will be collectively referred to as the "gate drive circuit 31n."

[0065] The input terminal of the A-phase gate drive circuit 31a is connected to the output terminal of the A-phase PWM circuit 30a. The output terminal of the A-phase gate drive circuit 31a is connected to the gate terminal of the A-phase MOSFET 13a. The input terminal of the B-phase gate drive circuit 31b is connected to the output terminal of the B-phase PWM circuit 30b. The output terminal of the B-phase gate drive circuit 31b is connected to the gate terminal of the B-phase MOSFET 13b. The input terminal of the C-phase gate drive circuit 31c is connected to the output terminal of the C-phase PWM circuit 30c. The output terminal of the C-phase gate drive circuit 31c is connected to the gate terminal of the C-phase MOSFET 13c.

[0066] The gate drive circuits 31a to 31c of the A to C phases are connected to a drive power supply V PA ~VPC The gate drive circuits 31a to 31c for the A to C phases are supplied with the driving reference potentials COMA to COMC from connection terminals (connection terminals between the source terminal of the MOSFET 13n and the connection terminal of the diode 14n) provided in the subsequent stages of the MOSFETs 13a to 13c for the A to C phases, respectively.

[0067] The gate drive circuit 31n is a drive circuit (third signal generation circuit) that outputs a gate signal (third signal) for applying a voltage between the gate (G) and source (S) of the MOSFET 13n and controls the ON / OFF of the MOSFET 13n. Specifically, the A-phase gate drive circuit 31a outputs a gate signal (A-phase gate signal A) for applying a positive voltage to the gate (G) of the A-phase MOSFET 13a with respect to the source (S) based on the PWM signal (A-phase PWM signal) from the A-phase PWM circuit 30a. As a result, the A-phase MOSFET 13a is turned ON between the drain (D) and the source (S) while a voltage is applied to the gate (G). The B-phase gate drive circuit 31b outputs a gate signal (B-phase gate signal B) for applying a positive voltage to the gate (G) of the B-phase MOSFET 13b with respect to the source (S) based on the PWM signal (B-phase PWM signal) from the B-phase PWM circuit 30b. As a result, the drain (D) and source (S) of the B-phase MOSFET 13b are ON while a voltage is applied to the gate (G). The C-phase gate drive circuit 31c outputs a gate signal (C-phase gate signal C) for applying a positive voltage to the gate (G) of the C-phase MOSFET 13c with respect to the source (S) of the C-phase MOSFET 13c, based on the PWM signal (C-phase PWM signal) from the C-phase PWM circuit 30c. As a result, the drain (D) and source (S) of the C-phase MOSFET 13c are ON while a voltage is applied to the gate (G).

[0068] According to this configuration, after the A-phase MOSFET 13a is turned ON, the B-phase MOSFET 13b is turned ON. Next, after the B-phase MOSFET 13b is turned ON, the C-phase MOSFET 13c is turned ON. Then, after the C-phase MOSFET 13c is turned ON, the A-phase MOSFET 13a is turned ON again.

[0069] As a result, in welding power supply 10 according to this embodiment, the ON timing / OFF timing of each of MOSFETs 13a to 13c from phase A to phase C is shifted by a predetermined period due to the phase of the switching frequency being shifted by 120 degrees for each phase.

[0070] In the welding power supply 10, the above-mentioned PWM control is used to control the supply of welding current, and when any one of the MOSFETs 13n of A-phase to C-phase MOSFETs 13a to 13c is ON, the power inductor 15n is charged, and when any one of the MOSFETs 13n is OFF, the welding current is supplied from the power inductor 15n to the workpiece M.

[0071] As a result, welding power supply 10 can extend the current flow time, which is one of the welding conditions, to an appropriate time. Furthermore, even if welding power supply 10 is single-phase and the control frequency is 10 kHz, it operates at a control frequency of 30 kHz through three-phase PWM control. As a result, welding power supply 10 allows for finer adjustment of the current flow time than single-phase.

[0072] (Noise Countermeasure #1) The path between the source terminal of MOSFET 13n and the input terminal of power inductor 15n is called the phase node. The gate (G) of MOSFET 13n is driven based on the phase node. A bootstrap circuit is generally used as the gate drive circuit for the high-side (upper leg) switching element (gate drive circuit for the high-side MOSFET) in a switching-drive power supply (hereafter referred to as a "switching power supply"). The bootstrap circuit is composed of a diode and a bootstrap capacitor.

[0073] However, when a bootstrap circuit is applied to welding power supply 10 according to this embodiment, a common power supply is used for multiple bootstrap circuits corresponding to MOSFETs 13a-13c of phases A to C. In such a configuration, surge noise generated in a specific phase may affect other phases.

[0074] Therefore, welding power supply 10 according to this embodiment employs gate drive circuit 31n in which specific components are insulated as a countermeasure against surge noise. Specifically, welding power supply 10 employs gate drive circuits in which the specific components ((a): first output component, and (b)-(d): first-third input components) relating to the following (a)-(d) are insulated as gate drive circuits 31a-31c for MOSFETs 13a-13c of phases A to C. Note that "insulation" here means electrically and physically separating signals between individual components in the circuit. (a): Output signals (PWM signals) from the PWM circuits 30a to 30c for phases A to C; (b): Input signals (gate signals) to the MOSFETs 13a to 13c of phases A to C; (c): A driving power supply (V PA ,V PB ,V PC ); (d): Drive reference potentials (COMA, COMB, COMC) for the gate drive circuits 31a to 31c of the A phase to C phase, respectively.

[0075] (Noise Countermeasure #2) At the phase node, surge noise occurs due to the influence of the power inductor when the MOSFET is turned off. Therefore, in switching power supplies, one method of dealing with surge noise is to add an RC snubber circuit to the phase node. The RC snubber circuit consists of a resistor and a capacitor.

[0076] However, when an RC snubber circuit is added to welding power supply 10 according to this embodiment (when an RC snubber circuit is added to each phase node from phase A to phase C), the capacitances from phase A to phase C are coupled via the RC snubber circuit, and surge noise generated in a specific phase may be transmitted to another phase via the RC snubber circuit, possibly resulting in malfunction.

[0077] The output reactor stores energy proportional to the square of the output current. The energy stored when each of MOSFETs 13a to 13c for phases A to C is turned off appears as a surge voltage between the drain (D) and source (S) of MOSFET 13n. If the surge voltage exceeds the rated withstand voltage between the drain (D) and source (S), MOSFET 13n will break down.

[0078] In switching power supplies with a small output current (for example, 100 A or less), surges can be suppressed by the power supply wiring or the wiring on the printed circuit board. As mentioned above, surges can also be suppressed by adding an RC snubber circuit between the drain (D) and source (S) of the MOSFET 13n. However, in the case of welding power supplies, the output current is large, and therefore the surge voltage is also large. As a result, a surge voltage several times higher than the power supply voltage is generated, causing the MOSFET 13n to fail.

[0079] Furthermore, when adding an RC snubber circuit as a surge noise countermeasure, the capacitance of the resistor and capacitor cannot be determined unless the output inductance of the welding power source is known. The output inductance of a welding power source includes not only the output reactor but also the parasitic components of the conductive wire (strand wire) to the welding electrode. The thickness and length of the conductive wire vary depending on the welding equipment that the welding power source is equipped with. Therefore, the output inductance cannot be uniquely determined. As such, adding an RC snubber circuit is not an effective measure to counter surge noise in a welding power source (the constants of the RC snubber circuit must be changed depending on the conductive wire to the welding electrode).

[0080] Therefore, in welding power supply 10 according to this embodiment, a surge noise countermeasure is taken by insulating the phase nodes from each other, rather than by adding an RC snubber circuit to the phase nodes. Specifically, welding power supply 10 adopts a countermeasure of adding the following active clamp circuit.

[0081] The active clamps 32a to 32c are made up of diodes and Zener diodes. Zener diodes are constant voltage diodes that can obtain a constant voltage even when the current changes.

[0082] The active clamps 32a to 32c are provided between the drain (D) and gate (G) of the MOSFET 13n. Specifically, the active clamps 32a correspond to the first phase (A phase), the active clamp 32b correspond to the second phase (B phase), and the active clamp 32c correspond to the third phase (C phase).

[0083] In the following description, the first-phase active clamp 32a will be referred to as the "A-phase active clamp 32a," the second-phase active clamp 32b will be referred to as the "B-phase active clamp 32b," and the third-phase active clamp 32c will be referred to as the "C-phase active clamp 32c." Furthermore, the first-phase to third-phase active clamps 32a to 32c will be collectively referred to as "active clamps 32n."

[0084] The active clamp 32n operates to prevent the drain voltage of the MOSFET 13n from exceeding the Zener voltage, which is set to a value greater than the charging voltage of the bulk capacitor 12 and less than the rated voltage between the drain (D) and source (S) of the MOSFET 13n.

[0085] In this way, welding power supply 10 according to this embodiment provides surge noise countermeasures without adding an RC snubber circuit to the phase node. Therefore, welding power supply 10 does not malfunction due to surge noise generated in a specific phase being transmitted to another phase via an RC snubber circuit. Furthermore, welding power supply 10 provides surge noise countermeasures that are not dependent on output inductance.

[0086] (Configuration of three-phase carrier waveform generation circuit) 3 is a diagram showing an example of the configuration of the three-phase carrier waveform generation circuit 20 according to this embodiment. As shown in FIG. 3, the three-phase carrier waveform generation circuit 20 mainly includes an oscillator 21, a ring counter 22, and sawtooth wave generation circuits 23a to 23c.

[0087] An input terminal of oscillator 21 is connected to an output terminal of control circuit 70. An output terminal of oscillator 21 is connected to an input terminal of ring counter 22. An output terminal of ring counter 22 is connected to input terminals of sawtooth wave generating circuits 23a to 23c. An output terminal of sawtooth wave generating circuits 23a to 23c is connected to an input terminal of PWM circuit 30n.

[0088] Oscillator 21 outputs a 60 kHz reference clock. Oscillator 21 is driven in accordance with an ON / OFF control signal from control device 70. Oscillator 21 outputs the reference clock in accordance with an ON signal, and stops outputting the reference clock in accordance with an OFF signal.

[0089] The ring counter 22 generates three 10 [kHz] pulses, each with a phase shift of 120 degrees in the switching control cycle, based on the 60 [kHz] reference clock output from the oscillator 21. Specifically, the ring counter 22 generates a 10 [kHz] B-phase pulse that is 120 degrees phase shifted from the A-phase. The ring counter 22 also generates a 10 [kHz] C-phase pulse that is 120 degrees phase shifted from the B-phase. The ring counter 22 also generates a 10 [kHz] A-phase pulse that is 120 degrees phase shifted from the C-phase.

[0090] 4 is a diagram illustrating an example of a circuit configuration of the ring counter 22 according to this embodiment. As illustrated in FIG. 4, the ring counter 22 mainly includes a D-FF (D flip-flop circuit) 221, an OR circuit 222, and an inverter circuit (Inverter) 223.

[0091] The D-FF221 is a logic circuit capable of storing one bit of information, and constitutes a sequential circuit that determines the current output based on past inputs. The D-FF221 also has two input terminals, a D terminal and a CK terminal, and one output terminal, a Q terminal. The information to be stored (0 or 1) is input as a D signal from the D terminal. A driving clock signal (Low or High) is input from the CK terminal. The stored information is output as a Q signal from the Q terminal. The CK terminal of the D-FF221 is connected to the output terminal of the oscillator 21. Therefore, the driving clock signal corresponds to a 60 kHz reference clock.

[0092] The ring counter 22 of this embodiment has five stages of D-FFs 2211 to 2215 to generate three 10 [kHz] pulses, with the phase of the switching control cycle shifted by 120 degrees for each phase, based on a 60 [kHz] reference clock.

[0093] The five stages of D-FF2211 to 2215 are connected in such a way that their Q terminals and D terminals are linked in sequence. Specifically, the Q terminal of D-FF2211 is connected to the D terminal of D-FF2212. The Q terminal of D-FF2212 is connected to the D terminal of D-FF2213. The Q terminal of D-FF2213 is connected to the D terminal of D-FF2214. The Q terminal of D-FF2214 is connected to the D terminal of D-FF2215.

[0094] The D-FF 221 stores the D signal when the reference clock rises. On the other hand, when the reference clock is in one of the three states of High, Falling, or Low, the D-FF 221 outputs the immediately preceding stored value as the Q signal regardless of the signal value of the D signal. As a result, in the configuration of this embodiment, the Nth-stage D-FF 221 n (For example, D-FF2211) outputs the previous stored value from the Q terminal to the N+1th stage D-FF221 when the reference clock is in one of the three states: High, Falling, or Low. n+1 In other words, when the reference clock is in one of the three states of High, Falling, and Low, the five-stage D-FFs 2211 to 2215 of this embodiment output the immediately preceding stored value to the D terminal of the preceding D-FF 221. n From the later stage D-FF221 n+1 The structure is such that the information is transmitted to the next

[0095] The Q signal of the first-stage D-FF 2211 is input as an A-phase pulse to the A-phase sawtooth wave generating circuit 23a. The Q signal of the third-stage D-FF 2213 is input as a B-phase pulse to the B-phase sawtooth wave generating circuit 23b. The Q signal of the fifth-stage D-FF 2215 is input as a C-phase pulse to the C-phase sawtooth wave generating circuit 23c.

[0096] The OR circuit 222 is a logic circuit that outputs a signal value of "1" (Low signal) when any of a plurality of input signals has a signal value of "1" (High signal). An input terminal of the OR circuit 222 is connected to the Q terminals of the five stages of D-FFs 2211 to 2215. An output terminal of the OR circuit 222 is connected to the input terminal of the inverter circuit 223.

[0097] The Q signals of the five stages of D-FFs 2211 to 2215 are all input to the OR circuit 222. Therefore, when all five input signals (Q signals) have a signal value of "0 (zero)", the OR circuit 222 outputs a signal value of "0" as an output signal. When any one of the five input signals has a signal value of "1", the OR circuit 222 outputs a signal value of "1" as an output signal.

[0098] The inverter circuit 223 is a logic circuit that inverts an input signal. The inverter circuit 223 has one input terminal and one output terminal. When the signal value of the input signal from the input terminal is "0" (Low signal), the inverter circuit 223 outputs a signal value of "1" (High signal) as an output signal from the output terminal. The inverter circuit 223 is provided on a path between the OR circuit 222 and the first-stage D-FF 2211. Therefore, when the signal value of the input signal from the OR circuit 222 is "0", a signal value of "1" is input from the D terminal to the first-stage D-FF 2211.

[0099] Fig. 5 is a timing chart of three-phase pulses PLa to PLc according to this embodiment. Fig. 5 shows a 10 [kHz] B-phase pulse PLb that is 120 degrees phase-shifted relative to the A-phase. Fig. 5 shows a 10 [kHz] C-phase pulse PLc that is 120 degrees phase-shifted relative to the B-phase. Fig. 5 shows a 10 [kHz] A-phase pulse PLa that is 120 degrees phase-shifted relative to the C-phase.

[0100] The ring counter 22 according to this embodiment generates three-phase pulses PLa to PLc as shown in FIG. 5 with the above configuration.

[0101] When the first reference clock CL rises, D-FF2211 stores the D signal. After that, D-FF2211 outputs the stored value "1" as the Q signal while the reference clock CL is in one of the three states: high, falling, and low. As a result, D-FF2211 outputs an A-phase pulse (high signal) until the reference clock CL rises again. Therefore, the period from the first rise of the reference clock CL to the second rise of the reference clock CL corresponds to the period during which the A-phase pulse is a high signal, i.e., the A-phase H pulse width.

[0102] When the reference clock CL rises for the second time, D-FF2211 stores the signal value "0" from the inverter circuit 223 as the D signal. Thereafter, D-FF2211 outputs the stored value "0 (zero)" as the Q signal while the reference clock CL is in the three states of High, falling, and Low. As a result, an A-phase pulse (Low signal) is output from D-FF2211 until the reference clock CL rises again. Furthermore, when the reference clock CL rises for the second time, D-FF2212 stores the signal value "1" of the previous input from D-FF2211 as the D signal. Thereafter, D-FF2212 outputs the stored value "1" as the Q signal while the reference clock CL is in the three states of High, falling, and Low.

[0103] When the reference clock CL rises for the third time, D-FF2211 stores the signal value "0" from the inverter circuit 223 as the D signal. Thereafter, D-FF2211 outputs the stored value "0" as the Q signal while the reference clock CL is in the three states of High, falling, and Low. As a result, an A-phase pulse (Low signal) is output from D-FF2211 until the reference clock CL rises again. Furthermore, when the reference clock CL rises for the third time, D-FF2212 stores the signal value "0" of the previous input from D-FF2211 as the D signal. Thereafter, D-FF2212 outputs the stored value "0" as the Q signal while the reference clock CL is in the three states of High, falling, and Low. As a result, a Q signal with a signal value of "0" is input from D-FF2212 to the D terminal of D-FF2213. Furthermore, when the reference clock CL rises for the third time, D-FF2213 stores the signal value "1" of the previous input from D-FF2212 as the D signal. After that, D-FF2213 outputs the stored value "1" as the Q signal while the reference clock CL is in the three states of High, Falling, and Low. As a result, D-FF2213 outputs a B-phase pulse (High signal) until the reference clock CL rises again.

[0104] Therefore, the period from the rising edge of the third reference clock CL to the rising edge of the fourth reference clock CL corresponds to the period during which the B-phase pulse is a High signal, that is, the B-phase H pulse width.

[0105] As described above, the ring counter 22 according to this embodiment includes five stages of D-FFs 2211 to 2215. The ring counter 22 outputs a signal value of "1" (High signal) to the Nth stage D-FF 2211 in synchronization with the rising edge of the 60 kHz reference clock CL. n D-FF221 on the N+1th row n+1 Furthermore, the ring counter 22 outputs the Q signal of the D-FF 2211 in the first stage as an A-phase pulse, outputs the Q signal of the D-FF 2213 in the third stage as a B-phase pulse, and outputs the Q signal of the D-FF 2215 in the fifth stage as a C-phase pulse.

[0106] As a result, the output period of the High signal in the pulse becomes 10 kHz, and the phase of the output period is shifted by 120 degrees for each of the three phases. As a result, welding power supply 10 according to this embodiment operates at 30 kHz as a whole.

[0107] Returning to the explanation of Fig. 3, the sawtooth wave generating circuits 23a to 23c generate three sawtooth waves for PWM modulation corresponding to each phase from the pulses PLa to PLc of phases A to C generated by the ring counter 22.

[0108] Therefore, in this embodiment, three sawtooth wave generating circuits 23a to 23c are provided in the circuit located in the subsequent stage of the ring counter 22. Specifically, the sawtooth wave generating circuit 23a corresponds to the first phase (A phase), the sawtooth wave generating circuit 23b corresponds to the second phase (B phase), and the sawtooth wave generating circuit 23c corresponds to the third phase (C phase).

[0109] In the following description, the first-phase sawtooth wave generating circuit 23a will be referred to as the "A-phase sawtooth wave generating circuit 23a," the second-phase sawtooth wave generating circuit 23b will be referred to as the "B-phase sawtooth wave generating circuit 23b," and the third-phase sawtooth wave generating circuit 23c will be referred to as the "C-phase sawtooth wave generating circuit 23c." Furthermore, the first- to third-phase sawtooth wave generating circuits 23a to 23c will be collectively referred to as the "sawtooth wave generating circuit 23n."

[0110] The A-phase sawtooth wave generating circuit 23a outputs a sawtooth wave for A-phase PWM modulation to the A-phase PWM circuit 30a as the A-phase carrier clock A. The B-phase sawtooth wave generating circuit 23b outputs a sawtooth wave for B-phase PWM modulation to the B-phase PWM circuit 30b as the B-phase carrier clock B. The C-phase sawtooth wave generating circuit 23c outputs a sawtooth wave for C-phase PWM modulation to the C-phase PWM circuit 30c as the C-phase carrier clock C.

[0111] 6 is a diagram showing an example of the configuration of the sawtooth wave generating circuit 23n according to this embodiment. As shown in Fig. 6, the sawtooth wave generating circuit 23n mainly includes a transistor (Tr) 231, a capacitor 232, a constant current source 233, and a buffer 234.

[0112] The transistor 231 is a switching element that is turned on (conductive) when a voltage equal to or greater than a predetermined value is applied to its base terminal, causing a current to flow between the collector and emitter. The capacitor 232 is a charger that stores electric charge when a DC voltage is applied. The constant current source 233 is a power supply that can supply a constant current regardless of the magnitude of the load. The buffer 234 has a function of lowering the output impedance of the sawtooth wave generating circuit 23n. The buffer 234 also has a function of preventing leakage from the constant current source 233 to the PWM circuit 30n.

[0113] A base terminal of the transistor 231 is connected to an output terminal of the ring counter 22. A collector terminal of the transistor 231 is connected to a first terminal of the capacitor 232 and a constant current source 233. An emitter terminal of the transistor 231 is connected to a second terminal of the capacitor 232 and a constant current source 233.

[0114] The sawtooth wave generating circuit 23n stores electricity in a capacitor 232 by ON / OFF control of a transistor 231 and current supply from a constant current source 233, and outputs a sawtooth wave via a buffer 234 by discharging from the capacitor 232.

[0115] 7 is a timing chart of three-phase sawtooth waves SAWa to SAWc according to this embodiment. In the following description, the sawtooth waves SAWa to SAWc of phases A to C are collectively referred to as "sawtooth wave SAWn." Furthermore, the pulses PLa to PLc of phases A to C are collectively referred to as "pulse PLn."

[0116] The sawtooth wave generating circuit 23n initializes the voltage of the sawtooth wave SAWn (resets the voltage value to "0 (zero)") when the pulse PLn is input from the ring counter 22. Thereafter, when the transistor 231 is in a non-conductive state (OFF), the sawtooth wave generating circuit 23n supplies a constant current from the constant current source 233 to the capacitor 232, and electric charge is stored therein.

[0117] As a result, the sawtooth wave generating circuit 23n changes (increases) the voltage at a constant rate up to the peak voltage Vsaw of the sawtooth wave SAWn. At this time, the sawtooth wave generating circuit 23n adjusts the voltage value of the peak voltage Vsaw of the sawtooth wave so that it is greater than the voltage value of the feedback signal, so that the duty ratio of the PWM signal does not become 100%.

[0118] The duty ratio of a PWM signal can be expressed as (H pulse width / pulse period), and the pulse period can be expressed as (H pulse width + L pulse width). The H pulse width is the period during which the pulse is a High signal, and the L pulse width is the period during which the pulse is a Low signal. Therefore, a PWM signal with a duty ratio of 100% means that the pulse period is the same as the H pulse width, and is in a constantly energized state.

[0119] The sawtooth wave generating circuit 23n adjusts the voltage value of the peak voltage Vsaw of the sawtooth wave by changing the current value output from the constant current source 233. The current value can be changed by setting the resistance value of a cermet trimmer (trimmer resistor / semi-fixed resistor) included in the constant current source 233.

[0120] As described above, welding power supply 10 according to this embodiment employs a three-phase interleaving system and operates as follows. Based on a 60 kHz reference clock, welding power supply 10 generates three 10 kHz pulses PLa-PLc, each with a switching control cycle phase shifted by 120 degrees. From pulses PLa-PLc, welding power supply 10 generates three carrier clocks A-C for PWM modulation, each corresponding to a phase. Based on carrier clocks A-C with a switching cycle phase shifted by 120 degrees, welding power supply 10 outputs a PWM signal for each phase in accordance with carrier clocks A-C. Based on the PWM signal for each phase, welding power supply 10 controls the ON / OFF of the switching elements for each phase.

[0121] According to this configuration, in the three-phase interleaved welding power supply 10 of this embodiment, the ON timing / OFF timing of each switching element of the three phases is shifted by a predetermined time due to the phase of the switching frequency being shifted by 120 degrees for each phase.

[0122] Three-phase interleaved welding power supply 10 uses the above-mentioned PWM control to control the supply of welding current, and is configured so that when any of the three-phase switching elements is ON, power inductor 15n is charged, and when any of MOSFETs 13n is OFF, welding current is supplied from power inductor 15n to workpiece M. As a result, welding power supply 10 can extend the current flow time, which is one of the welding conditions, to an appropriate time. In other words, three-phase interleaved welding power supply 10 can achieve a long current flow time that cannot be output by a linear control welding power supply.

[0123] Furthermore, even if welding power supply 10 is single-phase and has a control frequency of 10 kHz, it operates at a control frequency of 30 kHz through three-phase PWM control. As a result, welding power supply 10 allows for finer adjustment of the current flow time compared to single-phase.

[0124] The control frequency (30 kHz) of three-phase interleaved welding power supply 10 according to this embodiment is higher than the control frequency (10 kHz) of single-phase inverter welding power supplies. Therefore, three-phase interleaved welding power supply 10 allows fine adjustment of the current flow time. In other words, three-phase interleaved welding power supply 10 allows fine adjustment of the current flow time, which is not possible with single-phase interleaved welding power supplies.

[0125] Furthermore, the three-phase interleaved welding power supply 10 allows for fine adjustment of welding conditions, improving tracking performance in aluminum foil lamination welding, where contact resistance changes rapidly. Here, tracking performance refers to the tracking performance of the control function of the welding power supply 10. When the control frequency is 10 kHz, the control period is 100 μsec. In contrast, the three-phase interleaved welding power supply 10 according to this embodiment is configured to operate at a control frequency of 30 kHz. As a result, the control period is 33 μsec, which is shorter than the control period when the control frequency is 10 kHz. For example, as the joining of the workpieces M progresses and the contact resistance decreases, the current increases. Therefore, control to suppress the current increase is necessary. In this case, the three-phase interleaved welding power supply 10 can perform control at a period shorter than 100 μsec. Therefore, three-phase interleaved welding power supply 10 has improved follow-up (response) to fluctuations in workpiece M (load).

[0126] As described above, the three-phase interleaved welding power supply 10 according to this embodiment allows the current flow time to be extended and finely adjusted, ensuring an appropriate current flow time and providing stable welding quality.

[0127] (Verification of three-phase interleaving method) FIG. 8 is a diagram showing pulse waveforms in a three-phase interleaving scheme according to this embodiment. To verify the operation of the three-phase interleaving scheme according to this embodiment, the output of the pulse waveforms generated in each phase was verified. FIG. 8 shows the verification results. CH1 shows the output waveform when the reference clock CL is 60 kHz. CH2 shows the output waveform of an A-phase pulse PLa generated based on the 60 kHz reference clock CL. CH3 shows the output waveform of a B-phase pulse PLb generated based on the 60 kHz reference clock CL. CH4 shows the output waveform of a C-phase pulse PLc generated based on the 60 kHz reference clock CL.

[0128] As shown in Fig. 8, the pulses PLa to PLc of each phase are single-phase and have a control frequency of 10 kHz. The B-phase pulse PLb has a switching control period phase shifted by 120 degrees relative to the A-phase pulse. The C-phase pulse PLc has a switching control period phase shifted by 120 degrees relative to the B-phase pulse. The A-phase pulse PLa has a switching control period phase shifted by 120 degrees relative to the C-phase pulse. In the three-phase interleaving method according to this embodiment, as described above, the phase of the switching control period is shifted by 120 degrees for the pulses of each phase, and therefore the three phases operate at a control frequency of 30 kHz combined.

[0129] FIG. 9 is a diagram showing output waveforms of the three-phase interleaved welding power supply 10 according to this embodiment. To verify the operation of the three-phase interleaved welding power supply according to this embodiment, we verified each output during constant current control with a current flow time of 100 msec. FIG. 9 shows the verification results. CH1 shows the waveform of the welding command. CH2 shows the waveform of the output current. CH3 shows the waveform of the charging voltage. CH4 shows the waveform of the output voltage.

[0130] Figure 10 shows the output waveform of a single-phase inverter welding power supply. Specifically, Figure 10 shows the outputs during constant current control with a current flow time of 30 msec. CH1 shows the output voltage waveform. CH2 shows the output current waveform. As shown in Figure 10, with the single-phase inverter method, when the current flow time setting is small, the output waveform has a large ripple component and a large fluctuation range. Note that the ripple component here refers to the pulsating component of the current or voltage.

[0131] In contrast to this, as shown in FIG. 9, in the three-phase interleaving method according to this embodiment, the phase of the switching frequency is shifted by 120 degrees for each phase, so that the generation of ripple components that appear in the output voltage during welding and depend on the switching frequency can be suppressed (low noise can be achieved).

[0132] If the ripple component is large, the current to the workpiece M will fluctuate in strength. Therefore, if the generation of the ripple component can be suppressed, a stable welding current can be passed through the workpiece M. As a result, if a stable welding current can be passed through the workpiece M, the workpiece M can be heated more efficiently, improving the welding quality.

[0133] In this way, the three-phase interleaved welding power supply 10 according to this embodiment can adjust to desired welding conditions and obtain good welding quality.

[0134] (Variation: welding quality judgment function) In resistance welding, the welding result, i.e., the quality of the weld, is determined by welding conditions such as welding output (current value, voltage value, power value, etc.), welding time, and welding pressure. The waveform of the welding output is an important criterion for determining the quality of the weld.

[0135] Therefore, in this modification, a method of monitoring the waveform of the welding output and determining whether the welding is good or bad based on the monitoring results (welding quality determination function) is disclosed.

[0136] In the method according to this modification, the fluctuation range of the welding output that is acceptable for the welding quality is defined as a tolerance range (mask). Then, it is determined whether the fluctuation of the output waveform is within the tolerance range. If the fluctuation of the output waveform is within the tolerance range, the welding quality is determined to be "good (accepted)."

[0137] The difference between the first embodiment and this modification is whether or not it has a function for determining whether welding is good or bad. Therefore, the welding power supply according to this modification is the same as welding power supply 10 according to the first embodiment. Therefore, in the following description, the welding power supply according to this modification will be referred to as "welding power supply 10," just like the first embodiment.

[0138] In this modified example, a function for determining whether welding is good or bad is realized by executing a predetermined program by control circuit 70 included in welding power supply 10. Control circuit 70 is one or more processors. Control circuit 70 also realizes the predetermined function by, for example, reading a program stored in advance in nonvolatile memory (ROM: Read Only Memory) into volatile memory (RAM: Random Access Memory) and executing instruction codes defined as the program.

[0139] 11 is a flowchart of a welding quality determination process in a modified welding power supply 10. In this modified welding power supply 10, control circuit 70 executes the processing procedure shown in FIG. 11 to realize the welding quality determination function.

[0140] The control circuit 70 acquires the value of the welding output (current value) (step S11). The control circuit 70 also acquires the welding current value from the ADC 60, for example.

[0141] The control circuit 70 determines whether the acquired value is within an allowable range (step S12). The control circuit 70 also determines whether the welding current value is within a range of fluctuations in the current value that are allowable for welding quality. The range of fluctuations in the current value that are allowable for welding quality is set in advance in the control device 70. The range can be set, for example, by using two critical values, the upper and lower limits of the fluctuation range, as a set of data, and storing multiple sets of this data in units of welding time, and setting it in the control circuit 70. In this way, the control device 70 compares the welding current value with the upper and lower limits of the fluctuation range that correspond to the elapsed time of welding, and determines whether the welding current value is within the allowable range from the comparison result.

[0142] When the control circuit 70 determines that the acquired value is within the allowable range (step S12: YES), it determines that the welded product is non-defective and the welding quality of the welded product is "acceptable" (step S13).

[0143] FIG. 12(A) is a diagram showing an example of an output waveform when the welding quality according to this modified example is acceptable. FIG. 12(B) is a diagram showing an example of an output waveform when the welding quality according to this modified example is unacceptable. FIGS. 12(A) and 12(B) show the allowable range MSK of welding output. The allowable range MSK is a range having two critical values ​​at the same time. As shown in FIG. 12(A), the allowable range MSK has different critical values ​​(upper and lower limits of the fluctuation range) for elapsed welding time A and elapsed welding time B. Therefore, multiple pieces of data for the allowable range MSK are stored for each welding time.

[0144] As shown in the OK region of Fig. 12(A), when the welding current value is equal to or less than the upper limit and equal to or greater than the lower limit of the fluctuation range corresponding to the elapsed time of welding, the control circuit 70 determines that the welding current value is within the allowable range. On the other hand, when the control circuit 70 determines that the acquired value is outside the allowable range (step S12: NO), the control circuit 70 determines that the welded product is defective and classifies the welding quality of the welded product as "fail" (step S14).

[0145] As shown in the NG area of ​​Figure 12(B), control circuit 70 determines that the welding current value is outside the allowable range if it is greater than the upper limit of the fluctuation range corresponding to the elapsed time of welding. Alternatively, it determines that the welding current value is outside the allowable range if it is less than the lower limit of the fluctuation range corresponding to the elapsed time of welding. Thereafter, control circuit 70 displays on a display device (not shown) connected to welding power supply 10 whether the weld quality of the weld is "pass" or "fail."

[0146] As described above, welding power supply 10 according to this modification has a function for determining whether welding is good or bad. This allows welding power supply 10 to monitor the waveform of the welding output during welding and indicate whether the welding is good or bad based on the monitoring results.

[0147] Furthermore, the technology disclosed herein proposes a welding power supply 10 employing a three-phase interleaving method. As described above, the welding quality determination function proposed in this modification monitors whether the welding output contains little noise. As explained in the first embodiment, the three-phase interleaving method can achieve low noise by mutually complementing the OFF times of MOSFETs 13a-13c in the three-phase output. Therefore, the three-phase interleaving method is highly likely to produce good results in determining the quality of welding. Therefore, the three-phase interleaving welding power supply 10 disclosed herein can achieve stable welding quality in the manufacturing process, and a high product yield can be expected.

[0148] The present invention is not limited to the above-described embodiments and modifications, and various modifications are possible within the scope of the gist thereof. Furthermore, the configurations shown in the embodiments and modifications can be combined in any manner. That is, although the present invention has been described based on the embodiments, it is understood that the present invention is not limited to the embodiments, structures, etc. The technology of the present disclosure encompasses various modifications and modifications within the scope of equivalents. In addition, various combinations and forms, as well as other combinations and forms including only one element, more than one element, or less than one element, are also within the scope and concept of the technology of the present disclosure. [Explanation of symbols]

[0149] 10: Welding power supply; 12: bulk capacitor; 13a-13c, 13n: MOSFET; 171-173,17n:feedback circuits; 18: Error amplifier; 20: Three-phase carrier waveform generation circuit; 30a~30c,30n:PWM circuit; 31a to 31c, 31n: gate drive circuit; 32a~32c,32n: Active clamp; 50:RC filter; 91a, 91b: welding electrodes; PLa~PLc,PLn: Pulse

Claims

1. A welding power supply device used for resistance welding, Three switching elements corresponding to the first to third phases, respectively; a power storage unit that stores electricity when any one of the three switching elements is in a conductive state and outputs current from at least one of the switching elements to a welding electrode in contact with a workpiece when the switching element is in a non-conductive state; a first signal generating unit that generates three pulses corresponding to each phase based on a reference clock and generates three first signals for pulse modulation corresponding to each phase from the three generated pulses; a second signal generating unit that generates three second signals based on the three first signals to adjust a conduction time, which is a period during which each of the switching elements is in the conduction state; three drive units that output third signals to the switching elements based on the three second signals, for applying predetermined voltages to the switching elements, and thereby bring the switching elements into the conductive state or the non-conductive state; Equipped with The first signal generation unit a welding power supply configured to generate three of the pulses with the phases of the switching control cycles of the phases shifted by 120 degrees from each other.

2. The drive unit is a first input portion from the second signal generating unit; an output portion to the switching element; a second input portion of the driving power supply; a third input portion for a driving reference potential; 2. The welding power supply of claim 1, wherein each of the first input portion, the second input portion, the third input portion, and the output portion is configured to be electrically isolated.

3. the switching element is a field effect transistor, an active clamp between the drain and gate of the field effect transistor; The active clamp is It consists of a diode and a Zener diode.

3. The welding power supply according to claim 1, wherein the field effect transistor operates so that the drain voltage is smaller than a Zener voltage.

4. a feedback unit that feeds back an output signal corresponding to each of the control methods of constant current control, constant voltage control, and constant power control as an input signal; a switching unit that switches the input signal to be used as a feedback signal in accordance with each of the control methods; an error amplifier that compares a target value generated in accordance with a welding command with the feedback signal and amplifies an error that is a comparison result; 2. The welding power supply according to claim 1, wherein the error amplifier inputs the amplified error to the second signal generator and performs any one of current target control, voltage target control, and power target control.

5. Further, a filter unit is provided to suppress noise appearing in the output voltage during the resistance welding. When the control method is the constant voltage control, The feedback unit 5. The welding power supply according to claim 4, wherein the output voltage from the filter section after noise suppression is fed back as the input signal.

6. A welding power supply used for lamination welding of metal foils, Three field effect transistors corresponding to the first to third phases, respectively; a power inductor that stores electricity when any one of the three field effect transistors is in a conductive state and outputs current from at least one switching element to a welding electrode in contact with a workpiece when the field effect transistor is in a non-conductive state; a three-phase carrier waveform generating circuit that generates three pulses corresponding to each phase based on a reference clock and generates three carrier clocks for pulse modulation corresponding to each phase from the three generated pulses; three PWM circuits that generate three PWM signals that adjust the conduction time, which is the period during which each of the field-effect transistors is in the conducting state, based on the three carrier clocks; three gate drive circuits that output gate signals to the field effect transistors based on the three PWM signals to apply predetermined voltages to the field effect transistors, thereby turning the field effect transistors into the conductive state or the non-conductive state; Equipped with three output terminals of the three-phase carrier waveform generating circuit are connected to input terminals of the PWM circuits, an output terminal of each of the PWM circuits is connected to an input terminal of each of the gate drive circuits; an output terminal of each of the gate drive circuits is connected to a gate terminal of each of the field effect transistors; a source terminal of each of the field effect transistors is connected to an input terminal of the power inductor; The three-phase carrier waveform generating circuit includes: generating the pulses corresponding to the second phase, in which the phase of a switching control period of the field effect transistor of the second phase is shifted by 120 degrees relative to the phase of a switching control period of the first phase; generating the pulses corresponding to the third phase, the phase of a switching control period of the field effect transistor of the third phase being shifted by 120 degrees relative to the phase of a switching control period of the second phase; generating the pulse corresponding to the first phase, the phase of the switching control period of the field effect transistor of the first phase being shifted by 120 degrees with respect to the phase of the switching control period of the third phase; The welding power supply is configured as follows.

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