Motor control device

Abstract

The power factor and distortion of the power supply's input current are improved by suppressing and controlling LC resonance. [Solution] A motor control device comprising a rectifier circuit that rectifies and outputs AC power output from a three-phase AC power supply, an inverter circuit, a DC link capacitor whose capacitance value is set to suppress ripple voltage, and a reactor whose capacitance value is set to suppress ripple current, wherein the DC link capacitor and reactor function as a filter, and is characterized in that, with respect to the phase Θ of one line voltage of the three-phase AC power supply as a reference, the motor control is controlled so that the magnitude of the output voltage vector changes from positive to negative, or from negative to positive, with respect to the average value of the output voltage vector of the inverter circuit when the phase Θ is within the range that satisfies Equation 1, at a timing starting from phase Θ π / 3 × n (where n is an integer of 0 or more). (Equation 1) n × π / 3 ≤ Θ < (n+1) × π / 3 (where n is an integer of 0 or more).

Description

[Technical Field]

[0001] This disclosure relates to a motor control device. [Background technology]

[0002] In a motor control device that functions as an LC filter, comprising a rectifier circuit connected to an AC power supply, an inverter, a capacitor (C), and a reactor (L) connected between the AC power supply and the capacitor, the inverter is controlled to reduce the harmonic components of the DC voltage caused by the resonance of the LC filter. This reduces the harmonic components of the voltage across the capacitor, and consequently reduces the distortion of the current input to the motor control device. Patent Document 1 describes a control method in which a voltage detection device detects the DC link voltage between capacitors, extracts the AC component of the detected voltage using a filter, and suppresses resonance with a compensation amount based on the amount of pulsation. However, depending on the magnitude of the motor load, the phase angle of the inverter output may not be appropriate, causing voltage saturation and thus insufficient resonance suppression. Therefore, the patent document describes a control method that includes a phase-advancing means that outputs a phase-advancing AC component obtained by advancing the AC component extracted by the filter by 90 degrees, a weighting unit that outputs a value obtained by adding a value obtained by multiplying the AC component by a gain p (p is a variable that increases in accordance with the increase in the inverter output and varies in the range of 0 ≤ p ≤ 1) and a value obtained by multiplying the phase-advancing AC component by a gain (1-p), and a gain that outputs a correction signal obtained by multiplying the output of the weighting unit by a predetermined gain. By adjusting p according to the output of the inverter, a resonance suppression effect can be obtained regardless of the magnitude of the motor load and with a small amount of compensation. Patent Document 2 describes a control method for a power converter in which the voltage (VL) of a reactor is detected with the potential on the capacitor side as the reference potential, and a compensation amount based on the magnitude of VL is subtracted from a voltage control rate command, which is the ratio of the amplitude of the AC voltage output by the inverter circuit to the average value of the DC voltage, and a switching signal generated based on that voltage control rate command is supplied to the inverter circuit. The compensation amount is larger as the reactor voltage increases, and the absolute value of the compensation amount is smaller as the amplitude of the AC current output from the inverter circuit increases. [Prior art documents] [Patent Documents]

[0003] [Patent Document 1] Patent No. 5591215 [Patent Document 2] Patent No. 5712987 [Overview of the project] [Problems that the invention aims to solve]

[0004] In a motor control device that controls the power supplied by an LC filter consisting of a rectifier circuit, a capacitor (C) that tolerates DC voltage pulsation, and a reactor (L), as well as an inverter circuit, the filter extracts the pulsating component of the DC voltage and compensates for it in the inverter control amount. This compensation control extends to pulsations up to 6 × m times the power supply frequency (where m is an integer greater than or equal to 0), which can lead to a decrease in the power factor and increased distortion of the power supply's input current. This disclosure aims to improve the power factor and distortion of the power supply input current by suppressing LC resonance in an LC filter and inverter circuit, which consists of a rectifier circuit, a capacitor (C) that allows DC voltage pulsation, and a reactor (L). [Means for solving the problem]

[0005] The motor control device of the first aspect comprises a rectifier circuit that rectifies and outputs AC power output from a three-phase AC power supply, an inverter circuit having a plurality of switching elements that converts the DC power output from the rectifier circuit into AC power through the switching operation of the plurality of switching elements and supplies it to the motor, a capacitor connected between the input nodes of the inverter circuit and set to a capacitance value that allows fluctuations in the voltage output from the rectifier circuit and suppresses ripple voltage caused by the switching operation of the inverter circuit, and a reactor between the three-phase AC power supply and the capacitor set to a capacitance value that suppresses ripple current caused by the switching operation, wherein the capacitor and reactor function as a filter, and is characterized in that, with respect to the phase Θ of one line voltage of the three-phase AC power supply as a reference, the motor control device controls the magnitude of the output voltage vector from positive to negative or from negative to positive, with respect to the average value of the output voltage vector of the inverter circuit when the phase Θ is within the range that satisfies Equation 1, at a timing starting from phase Θ π / 3 × n (where n is an integer greater than or equal to 0) with respect to the phase Θ of one line voltage of the three-phase AC power supply. Equation 1 satisfies n × π / 3 ≤ Θ < (n+1) × π / 3 (where n is a non-negative integer). In this case, the power factor and distortion of the power supply's input current can be improved by suppressing LC resonance through an LC filter and inverter circuit consisting of a rectifier circuit, a capacitor (C) that allows DC voltage pulsation, and a reactor (L). The motor control device according to the second perspective is the motor control device according to the first perspective, characterized in that, with respect to the phase Θ of one line voltage of a three-phase AC power supply, the output voltage of the inverter circuit is controlled such that the output voltage vector pulsates at least once in synchronization with the period of LC resonance by the reactor and capacitor, within the range in which the phase Θ satisfies Equation 2. Equation 2 is n × π / 3 ≤ Θ ≤ n × π / 3 + π / 6 (where n is an integer greater than or equal to 0). A motor control device according to a third perspective is a motor control device according to a first perspective, characterized in that, when controlling the magnitude of the output voltage vector of an inverter circuit from positive to negative or negative to positive, the magnitude of the change in the output voltage vector of the inverter circuit from positive to negative or negative to positive is inversely proportional to the effective value of the current flowing to the motor supplied with AC from the inverter circuit. The motor control device according to the fourth aspect is a motor control device according to the first aspect, characterized in that the DC voltage across the capacitors contains 6 × m (where m is an integer greater than or equal to 0) harmonic components with the frequency of the line voltage as the fundamental wave, the amplitude of the 6 × m (where m is an integer greater than or equal to 0) harmonics is 6% or less of the DC component of the DC voltage, and the output voltage of the inverter circuit is controlled such that the phase of the 6 × m (where m is an integer greater than or equal to 0) harmonics is π ± π / 16 with respect to the fundamental wave phase of the line voltage. The motor control device according to the fifth aspect is a motor control device according to the first aspect, characterized in that the transfer characteristic consisting of the DC voltage between the capacitors and the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply becomes a second-order lag element, and the output voltage of the inverter circuit is controlled such that the damping coefficient of the transfer function of the second-order lag element is greater than 0.7. A motor control device according to the sixth aspect is a motor control device according to the first or fifth aspect, characterized in that it controls the output voltage of an inverter circuit such that the value of the DC voltage between capacitors is less than or equal to the maximum value among the three absolute values ​​of the line voltages of the three-phase AC power supply. [Brief explanation of the drawing]

[0006] [Figure 1] This figure shows the overall configuration of the motor control device according to this embodiment. [Figure 2] This figure shows the line-to-line voltage values ​​output from an AC power supply. [Figure 3] This is a diagram illustrating the output voltage vector. [Figure 4] This is a control block diagram of the control unit. [Figure 5]This diagram shows the timing of the control process. [Figure 6] This is an example of a model for controlling a motor control device. [Figure 7] This is the original equivalent control model for motor control devices. [Figure 8] This is a circuit diagram showing a simplified equivalent circuit of the motor control device shown in Figure 1. [Figure 9] This figure shows comparative example 1 of a control block that controls the circuit shown in Figure 8. [Figure 10] This figure shows a comparative example 2 of a control block that controls the circuit shown in Figure 8. [Figure 11] This diagram shows the control block of this embodiment. [Figure 12] This figure shows the frequency characteristics of the transfer function when there is no power supply impedance (hereinafter referred to as power supply Z). [Figure 13] This figure shows the frequency characteristics of the transfer function when a power supply Z is present. [Figure 14] This figure shows the frequency characteristics of the transfer function of a second-order lag element. [Figure 15] This figure shows the relationship between the magnitude of the power supply Z and the magnitude of the damping coefficient. [Figure 16] This figure shows the waveform of a DC voltage. [Figure 17] This figure shows the harmonic components in this embodiment. [Modes for carrying out the invention]

[0007] The embodiments of this disclosure will be described in detail below with reference to the attached drawings. Figure 1 is a diagram showing the overall configuration of the motor control device 1 according to this embodiment. The motor control device 1 receives three-phase AC power from the AC power supply 20, converts the supplied AC, and supplies it to the motor 65. The motor control device 1 comprises a rectifier circuit 2, an inverter circuit 3, a capacitor 4, a reactor 5, and a control unit 10.

[0008] The rectifier circuit 2 is connected to the AC power supply 20 and rectifies and outputs the AC output from the three-phase AC power supply. The rectifier circuit 2 is a three-phase full-wave rectifier circuit. The inverter circuit 3 converts the DC output by the rectifier circuit 2 into three-phase AC and supplies it to the motor 65. The three-phase AC output by the inverter circuit 3 is referred to as the u-phase, v-phase, and w-phase from the upper line in FIG. 1. Also, the voltage between the u-phase and the v-phase is the line voltage V uv and the voltage between the v-phase and the w-phase is the line voltage V vw and the voltage between the w-phase and the u-phase is the line voltage V wu is defined as such.

[0009] The capacitor 4 is provided in the DC link section to which the rectifier circuit 2 and the inverter circuit 3 are connected. The capacitance value of the capacitor 4 is set to allow fluctuations in the voltage output by the rectifier circuit 2 and suppress the ripple voltage caused by the switching operation of the inverter circuit 3. Let the capacitance of the capacitor be C, the voltage between the capacitors 4 be V dc , the current flowing through the capacitor 4 be i c is defined as such. Also, let the current flowing from the DC link section to the inverter circuit 3 be i inv is defined as such. The capacitor 4 is an example of a film capacitor. )

[0010] The reactor 5 is provided between the three-phase AC power supply and the capacitor 4. The capacitance value of the reactor 5 is set to suppress the ripple current caused by the switching operation of the inverter circuit 3. Let the capacitance value of the reactor 5 be L. In the illustrated example, the reactor 5 is provided between the rectifier circuit and the DC link section, but the reactor 5 may be provided between the three-phase power supply and the rectifier circuit. The reactor 5 and the capacitor 4 constitute an LC filter.

[0011] The AC power supply 20 is a three-phase AC power supply and is referred to as the r-phase, s-phase, and t-phase from the upper line in FIG. 1. Here, the voltage between the r-phase and the s-phase is the line voltage V rs and the voltage between the s-phase and the t-phase is the line voltage V st and the voltage between the t-phase and the r-phase is the line voltage V trLet's assume that. Furthermore, the impedances contained in the AC power supply 20 itself and the transmission line from the AC power supply 20 to the rectifier circuit 2 are shown in Figure 1 as power supply impedances 21r, 21s, and 21t. The impedance value per phase varies depending on the surrounding environment. Here, the power supply impedance value per phase is L a Let's call it [H].

[0012] Motor 65 is, for example, an IPM motor (Interior Permanent Magnet Motor). Motor 65 drives the compressor in the refrigerant circuit.

[0013] The control unit 10 controls the inverter circuit 3 and converts the AC power supplied to the motor 65 to operate the motor 65 efficiently. The control unit 10 is equipped with a device for processing various types of information. The control unit 10 can be configured using a microcomputer and a memory device containing software for operating it. The control unit 10 includes a phase detection unit 11 and a voltage detection unit 12. The phase detection unit 11 detects at least one line voltage of the three-phase AC power supply, processes it with a phase detection circuit to convert it into a pulse, and transmits it to the control unit 10. In the illustrated example, the phase detection unit 11 detects the line voltage between the r phase and the s phase. The voltage detection unit 12 detects the voltage across the capacitor 4.

[0014] Figure 2 shows the line voltage values ​​output from the AC power supply 20. In Figure 2, the vertical axis shows the line-to-line voltage values ​​output from the three-phase AC power supply. In Figure 2, the solid line represents the line-to-line voltage V rs The value of the line voltage V is shown by the dashed line. st The value of the line voltage V is shown by the dashed line. tr The value is shown. Also, the maximum absolute value of the three line voltages is shown by a dashed line. In the example shown in Figure 2, the line voltage V rs The value of is 0V, and the line voltage V rs The timing of the change from negative to positive is defined as 0°. Line voltage V rsand line voltage V st The phase difference is 240 degrees, and the line voltage V rs and line voltage V tr The phase difference is 120 degrees. As mentioned above, capacitor 4 (see Figure 1), shown in Figure 1, allows for fluctuations in the voltage output by the rectifier circuit 2 (see Figure 1). Therefore, capacitor 4 does not smooth the three-phase full-wave rectification output from the rectifier circuit 2, and a potential equal to the maximum of the absolute values ​​of the three line voltages is generated at capacitor C, pulsating at a frequency six times that of the AC power supply 20.

[0015] Here, we define the output voltage vector of the inverter. Figure 3 is a diagram illustrating the output voltage vector. The circuit diagram in Figure 3 shows the state in which the capacitance C of capacitor 4 in Figure 1 is divided into two capacitors 4a, with the capacitance C being 2C. The voltage vector of the u phase is defined as V, with the potential between these two divided capacitors 4a as the reference. un Similarly, using the potential across capacitor 4a as a reference, the voltage vector of the v phase is V vn Let the voltage vector of the w phase be V wn Let's assume this voltage vector V un And, V vn And, V wn Using these, the output voltage vector of the inverter is defined by equation (1-1). Also, the magnitude of the output voltage vector V o This is defined by equation (1-2).

[0016]

number

[0017] Figure 4 is a control block diagram of the control unit 10 (see Figure 1). The control unit 10 includes a phase estimation unit 101, a voltage averaging unit 102, a voltage command generation unit 103, a comparison unit 104, a gain correction unit 105, a multiplier 106, a manipulated variable limiter 107, and a modulation rate command generation unit 108.

[0018] [Phase estimation section] The phase estimation unit 101 estimates the phase of the AC power supply 20. The phase estimation unit 101 acquires pulses transmitted from the phase detection unit 11 and calculates the phase. The method for estimating the phase is not particularly limited, and the voltage V across the capacitor 4 is also used. dc Detects voltage V dc By filtering the voltage V dc Alternatively, pulsations may be extracted, and their frequency and phase may be calculated from the pulsation components.

[0019] [Voltage averaging section] The voltage averaging unit 102 measures the voltage V dc The average value is calculated. The voltage averaging unit 102 calculates the average value of the detected voltage V dc The voltage V is passed through a low-pass filter. dc Average voltage V DCAVE The voltage averaging unit 102 may also average instantaneous values ​​from the phase estimation unit 101 with respect to the phase of the line voltage, where the phase is between 60n° and 60(nm)° (n and m are integers).

[0020] [Voltage command generation unit] The voltage command generation unit 103 receives the power supply voltage estimation phase θ from the phase estimation unit 101. VDC The voltage average value V is obtained from the voltage averaging unit 102. DCAVE The voltage command generation unit 103 obtains the power supply voltage estimated phase θ and the voltage average value V. DCAVE From this, an ideal three-phase full-wave rectifier is generated, and this ideal three-phase full-wave rectifier is given a voltage command V dc * Let's assume that.

[0021] [Comparison section] The comparison unit 104 processes the voltage command V dc * The voltage detection unit 12 detected the detected value V dc It compares the two and extracts the difference. Specifically, the comparison unit 104 compares the voltage command V dc * From the detected value V dc Subtract it.

[0022] [Gain Correction Section] The gain correction unit 105 controls the resonance suppression gain k FB Corrects i. inv Resonance control is performed by controlling the resonance suppression gain k FB Corrects i. inv Resonance suppression is performed by controlling i inv Since direct control is difficult, resonance suppression is performed by manipulating the output voltage vector of the inverter. In this case, i inv The magnitude of the pulsation (resonance suppression gain k) is assumed to be manipulated. FB Controlling the output voltage vector with ) causes excessive pulsation. Therefore, the effective value of the motor current I rms Depending on the result, the magnitude of the pulsation in the output voltage vector (resonance suppression gain k) FB ) is corrected by. Specifically, the gain correction unit 105 uses the resonance suppression gain k FB Using as the numerator, then the effective value of the motor current I rms Perform division with the denominator being [the value of the denominator].

[0023] [Controllable variable limiter] The manipulated variable limiter 107 sets upper and lower limits for the manipulated variable, ensuring that it does not interfere with the manipulated variable that controls the fundamental wave of the motor current, and that abnormal numerical results are not output.

[0024] [Modulation Rate Command Generation Unit] The modulation rate command generation unit 108 (subtractor) controls the modulation rate command value K that controls the fundamental wave of the motor current. s From ', modulation rate correction value K s By subtracting '', the modulation rate command value K s * Outputs the magnitude of the output voltage vector V. o The modulation rate command value K s * It is proportional to.

[0025] [Control Timing] Figure 5 shows the timing of the control process. The control unit 10 receives the voltage command V dc * It has a voltage command V dc *and detected value V dc The difference is extracted and compensated for. The extracted difference represents the resonant component. The control unit 10 corrects the modulation rate based on the extracted resonant component, thereby reducing the pulsation caused by resonance. inv This is used to press against the DC link section and suppress resonance.

[0026] In Figure 5, the left vertical axis shows the magnitude of the DC voltage [V], and the right vertical axis shows the magnitude of the inverter output voltage vector [V]. The horizontal axis shows time [s]. In Figure 5, the detected value of the DC voltage is V. dc This is shown by a solid line. The command value V generated by the voltage command generation unit 103. dc * These are shown by dotted circles. The instantaneous value of the output vector is shown by a dashed line, and the average value of the output voltage vector is shown by a dashed line. In Figure 5, the vertical dashed lines indicate the phase of the DC voltage command. The phase of the DC voltage command is the phase estimated by the phase estimation unit 101 (see Figure 4). In Figure 5, the vertical double-dotted lines indicate a portion of the timing when the output voltage vector is pulsating.

[0027] The control unit 10 (see Figure 4) uses the phase of one line voltage of the three-phase AC power supply as a reference and controls the output voltage vector so that its magnitude changes from positive to negative or negative to positive, based on the average value of the inverter's output voltage vector between n × (π / 3) and (n+1) × (π / 3) phases, starting at a timing where the phase is n × (π / 3) (where n is an integer).

[0028] Here, a positive magnitude for the output voltage vector relative to the average value of the inverter's output voltage vector means that the magnitude of the output voltage vector is greater than the average value of the output voltage vector. Conversely, a negative magnitude for the output voltage vector relative to the average value of the inverter's output voltage vector means that the magnitude of the average output voltage vector is smaller.

[0029] Furthermore, the control unit 10 may control the inverter's output vector from positive to negative, or from negative to positive, at more timings than (π / 3) × n timings from the reference phase. In this embodiment, the output magnitude of the output voltage vector changes depending on the difference between the DC voltage command and the detected DC voltage, so the output line voltage vector pulsates in synchronization with the LC filter's resonance period. In the example shown in the figure, the output voltage vector pulsates one or more times in synchronization with the LC resonance period within the phase range n × (π / 3) ≤ θ ≤ n × (π / 3) + (π / 6).

[0030] [Details of gain correction] Next, the gain correction unit 105 adjusts the resonance suppression gain k FB Using this as the numerator, then the RMS value of the motor current I rms Let's explain the point about dividing with the denominator. Figure 6 shows an example of a model for controlling the motor control device 1. inv ' represents the current value determined by motor control. The i shown in Figure 6 is... inv '' is the value of the current manipulated by resonance suppression control. Also, i inv And, i inv ' and i inv The relationship with '' is given by equation (1-3).

[0031]

number

[0032] Figure 7 shows the original equivalent control model of the motor control device 1. Figure 7 shows the output voltage V0 applied to the motor load i inv This is determined. Also, the motor load value changes depending on the motor's operating point. In Figure 7, V0' is the voltage value determined by the motor's operation. V0'' is the voltage value manipulated by resonance suppression control. Furthermore, the relationship between V0, V0', and V0'' is given by equation (1-4).

[0033]

number

[0034] Here, considering the motor load, we consider the magnitude of the V0 pulsation. The magnitude of the V0 pulsation is the modulation rate correction value K of the inverter inside the control system. s The resonance suppression gain correction value k is determined by ''. FBVDC By changing this value, the magnitude of the V0 pulsation can be adjusted. Equation (1-5) is derived from the subtractor 203 and resonance suppression gain 272 in Figure 6 and equation (1-3).

[0035]

number

[0036] Furthermore, equation (1-6) can be derived from Figure 4.

[0037]

number

[0038] Furthermore, from the subtractor 203 and gain 271 in Figure 7 and equation (1-4), equation (1-7) can be derived.

number

[0039] Here, assuming an inverter efficiency of 1 and a motor power factor of 1, the RMS value of the motor current I rms and inverter current i inv And the inverter output voltage V0 and voltage V dc The relationship is expressed by equation (1-8) below.

[0040]

number

[0041] By rearranging equations (1-8), we get output voltage V0 and voltage V dc , modulation rate command value K s * Equation (1-9), which is the relationship between the two, is derived.

[0042]

Number

[0043] Applying equations (1-5) and (1-6) to equation (1-8) and dividing both sides by V dc yields equation (1-11).

Number

[0044] Here, the second term on the left side and the second term on the right side of equation (1-11) are terms determined by the pulsation magnitude for resonance suppression. Also, the first term on the left side and the first term on the right side are terms determined by the operating point of the motor. Extracting the pulsation magnitude for resonance suppression from the second term on the left side and the second term on the right side gives the following equation (1-12).

[0045]

Number

[0046] Thus, the correction value of k FBVDC is inversely proportional to I rms .

[0047] Figure 8 is a circuit diagram showing a simplified equivalent circuit of the motor control device 1 in Figure 1. The voltage V in is the maximum value among the absolute values of the three line-to-line voltages of the three-phase AC power supply 20. The current i L is the current flowing through the reactor 5. The voltage V s is the DC voltage output from the rectifier circuit 2. The voltage V dc is the voltage across the capacitor 4. The current i c / is the current flowing through the capacitor 4. The current i inv is the current flowing into the inverter circuit 3.

[0048] FIG. 9 is a diagram showing Comparative Example 1 of a control block that controls the circuit shown in FIG. 8. This Comparative Example 1 is a control model that does not perform resonance suppression control. Comparative Example 1 includes three subtractors 201, 202, 204 and three transfer functions G1, G2, G3. The subtractor 201 feeds back the current i L through the transfer function G1, subtracts it from V in and outputs the voltage V s . The subtractor 202 feeds back the voltage V dc , subtracts the voltage V s from the voltage V dc and outputs the voltage V L . The transfer function G2 inputs the voltage V L and outputs the current i L . The subtractor 204 subtracts the current i L from the current i inv and outputs the current i c . The transfer function G3 inputs the current i c and outputs V dc .

[0049] When this control block is arranged to obtain the transfer function G NoDamp (s)=V dc / V in , Equation (1-13) is obtained.

[0050]

Equation

[0051] FIG. 10 is a diagram showing Comparative Example 2 of a control block that controls the circuit shown in FIG. 8. In this Comparative Example 2, compared with Comparative Example 1, the voltage V L applied across the reactor 5 is multiplied by the resonance suppression gain k FB to obtain the current i invIt differs in that it outputs [something]. Comparative Example 2 comprises three subtractors 201, 202, and 204, three transfer functions G1, G2, and G3, and a resonance suppression gain 272.

[0052] Subtractor 201 processes current i L This is fed back through the transfer function G1, V in Subtract from the voltage V s Outputs. Subtractor 202 is used for voltage V dc The voltage V is fed back. s From voltage V dc Subtracting this gives the voltage V L Outputs. The transfer function G2 is given by the voltage V L Enter the current i L Outputs. Gain 272 corresponds to voltage V L and gain k FB Multiply by the current i inv Outputs. Subtractor 204 processes current i L From, current i inv Subtracting this, the current i c Outputs. The transfer function G3 is given by the current i c Enter V dc Outputs.

[0053] The control block of this Comparative Example 2 is organized to show the transfer function G, which exhibits frequency characteristics. VL (s) = V dc / V in When we find this, we get equation (1-14).

[0054]

number

[0055] Figure 11 shows the control block of this embodiment. The control block of this embodiment comprises four subtractors 201, 202, 203, and 204, three transfer functions G1 to G3, and a gain 273. Subtractor 201 processes current iL This is fed back through the transfer function G1, V in Subtract from the voltage V s Outputs. Subtractor 202 is used for voltage V dc The voltage V is fed back. s From voltage V dc Subtracting this gives the voltage V L Outputs. The subtractor 203 receives the command value V dc * From the detected value V dc Subtract it. Gain 273 corresponds to command value V dc * From the detected value V dc The value obtained by subtracting the gain k FB Multiply by the current i inv Outputs. Subtractor 204 processes current i L From, current i inv Subtracting this, the current i c Outputs. The transfer function G3 is given by the current i c Enter V dc Outputs.

[0056] The control block of this embodiment is organized to show the transfer function G, which exhibits frequency characteristics. VDC (s) = V dc / V in When we find this, we get equation (1-15).

[0057]

number

[0058] Here, we use equations (1-13), (1-14), and (1-15) to show the frequency characteristics of the transfer function. Figure 12 shows the frequency characteristics of the transfer function when there is no power supply Z. The horizontal axis shows frequency [Hz], and the vertical axis shows gain [dB] and phase [deg]. Comparative Example 2 is shown by a solid line, this embodiment by a dashed line, and Comparative Example 1 by a dotted line. In the absence of power supply Z, Comparative Example 1, which does not perform resonance suppression, shows an increase in gain at the resonant frequency. On the other hand, both Comparative Example 2, which performs resonance control, and this embodiment, which also performs resonance control, have a gain of 0 or less at the resonant frequency, thus suppressing the increase in gain.

[0059] Figure 13 shows the frequency characteristics of the transfer function when a power supply Z is present. Figure 13, like Figure 12, shows frequency [Hz] on the horizontal axis and gain [dB] and phase [deg] on the vertical axis. Comparative Example 2 is shown by a solid line, this embodiment by a dashed line, and Comparative Example 1 by a dotted line. In the example shown in the figure, the frequency characteristics of the transfer function are shown when the magnitude of the power supply Z is 1.5 [mH]. When power supply Z is present, in both Comparative Examples 1 and 2, the resonant component at the resonant frequency is not attenuated, and a peak is observed. On the other hand, in the resonance control of this embodiment, the resonant component is attenuated. In the resonance control of this embodiment, since the gain characteristic is 0 [dB] or less, the maximum value of the DC voltage is less than or equal to the maximum absolute value of the voltages between multiple power lines.

[0060] Figure 14 shows the frequency characteristics of the transfer function of a second-order lag element. Figure 14 shows frequency [Hz] on the horizontal axis and gain [dB] and phase [deg] on the vertical axis. The solid line shows the frequency response when the attenuation coefficient is 10. The dashed line shows the frequency response when the attenuation coefficient is 3, and the double dashed line shows the frequency response when the attenuation coefficient is 1. Furthermore, the long dashed line shows the frequency response when the attenuation coefficient is 0.7, the dashed line shows the frequency response when the attenuation coefficient is 0.4, and the dotted line shows the frequency response when the attenuation coefficient is 0.1. As shown in Figure 14, the gain at the resonant frequency increases when the damping coefficient is 0.1 and when it is 0.4. On the other hand, when the damping coefficient is 0.7 or higher, the gain is 0 or less, and the gain at the resonant frequency does not increase.

[0061] Figure 15 shows the relationship between the magnitude of the power supply Z and the magnitude of the damping coefficient ξ. Figure 15 shows the power supply Z [mH] on the horizontal axis and the magnitude of the damping coefficient ξ on the vertical axis. Comparative Example 2 is shown by a solid line, and this embodiment is shown by a dashed line. Here is the damping coefficient ξ of Comparative Example 2. VL This can be expressed by equation (1-16), based on equation (1-14).

[0062]

number

[0063] Furthermore, the damping coefficient ξ of this embodiment VDC This can be expressed by equation (1-17) based on equation (1-15).

[0064]

number

[0065] Figure 15 shows the resonant gain k FB When the power supply Z is uniquely determined, it is shown that in Comparative Example 2 and this embodiment, the damping coefficient ξ changes as the power supply Z increases. In the example shown in the figure, in this embodiment, the damping coefficient ξ is 0.7 or more when the power supply Z is 0, and the value of the damping coefficient ξ increases as the power supply Z increases. The power supply Z is often unknown as its value changes depending on the commercial power supply environment, so if the damping coefficient ξ is designed to be 0.7 when the power supply Z is 0, the damping coefficient ξ will be 0.7 or more for any power supply Z, and the resonance component can be suppressed. On the other hand, if the damping coefficient ξ is designed to be greater than 0.7 when the power supply Z is 0, the gain will be excessive when the power supply Z is 0, and the control will become unstable.

[0066] [DC voltage waveform] Figure 16 shows the waveform of a DC voltage. Figure 16 shows the case without a power supply Z in the upper table and the case with a power supply Z of 0.5 [mH] in the lower table. In the tables, time [s] is shown on the horizontal axis and the magnitude of the DC voltage [V] is shown on the vertical axis. The DC voltage waveform for Comparative Example 2 is shown by a solid line, and the DC voltage waveform for this embodiment is shown by a dotted line.

[0067] As shown in the lower waveform of Figure 16, when the power supply Z is 0.5 [mH], the difference between the maximum and minimum DC voltages in Comparative Example 2 is larger than the difference between the maximum and minimum DC voltages in this embodiment. In the resonance control of this embodiment, as shown in Figure 13, the gain characteristic is 0 [dB] or less even when the power supply Z is 1.5 [mH], so the maximum DC voltage is less than or equal to the maximum absolute value of the multiple power supply line voltages. On the other hand, in Comparative Example 2, as shown in Figure 13, when there is a power supply Z, the gain becomes large at the resonant frequency, and the resonant component cannot be sufficiently attenuated. Therefore, in Comparative Example 2, as shown in the lower waveform of Figure 16, the maximum DC voltage is larger than the maximum absolute value of the multiple power supply line voltages.

[0068] [Harmonic components] Figure 17 shows the harmonic components included in the DC voltage waveform in this embodiment. Figure 17 shows the phase [deg] on the horizontal axis and the ratio of the DC component to the harmonic component of the DC voltage on the vertical axis. In this embodiment, since capacitor 4 (see Figure 1) does not smooth the DC voltage, the DC voltage contains 6 × m-th order harmonic components (m is an integer greater than or equal to 0) with the frequency of the power supply line voltage as the fundamental wave. The DC voltage can be expressed by separating it into the DC component and the harmonic component using the following equation.

[0069]

number

[0070] The first term on the right-hand side of equation (1-18) is the DC component of the DC voltage. The second term on the right-hand side of equation (1-18) is the 6 × m harmonic component. Furthermore, the third term and subsequent terms on the right-hand side of equation (1-18) include carrier components and other elements. Figure 17 shows this DC component V. dc0 and harmonic component V dc6m cos(6mω g t+θ 6m The ratio to ) is shown, and the harmonic components up to m = 3 are shown.

[0071] Figure 17 shows the reference phase with a dotted line. Here, the reference phase is the phase of one line voltage of the AC power supply 20. In Figure 17, the 6th harmonic is shown with a solid line, the 12th harmonic with a dashed line, and the 18th harmonic with a broken line. As shown in Figure 17, the ratio of the 6th harmonic to the DC component of the DC voltage is kept within 6%. Furthermore, the ratio of the 12th harmonic to the DC component of the DC voltage is kept within 2%. Additionally, the ratio of the 18th harmonic to the DC component of the DC voltage is kept within 1%.

[0072] The DC voltage can be controlled within the ranges of equations (1-19) and (1-20) by a command value of a DC voltage having the amplitude and phase of the 6 × m-th harmonic component (where m is a non-negative integer) of an ideal three-phase full-wave rectified waveform.

[0073]

number

[0074] [Effects] The motor control device 1 of this disclosure includes a rectifier circuit 2 that rectifies and outputs AC power output from a three-phase AC power supply 20, an inverter circuit 3 having a plurality of switching elements that converts the DC power output by the rectifier circuit 2 into AC power through the switching operation of the plurality of switching elements and supplies it to the motor, a capacitor 4 connected between the input nodes of the inverter circuit 3, which allows for voltage fluctuations output by the rectifier circuit 2 and has a capacitance value set to suppress ripple voltage caused by the switching operation of the inverter circuit 3, and a switch between the three-phase AC power supply 20 and the capacitor 4. A motor control device comprising a reactor 5 whose capacitance value is set to suppress ripple current caused by pulsation, and a capacitor 4 and a reactor 5 which function as a filter, controls the output voltage vector of the inverter circuit 3 to change from positive to negative or from negative to positive, based on the average value of the output voltage vector of the inverter circuit 3 when the phase Θ is in the range of n×π / 3≦Θ<(n+1)×π / 3 (n is an integer of 0 or more), with the phase Θ starting at π / 3×n (n is an integer of 0 or more) as the reference, and the phase Θ is in the range of n×π / 3≦Θ<(n+1)×π / 3 (n is an integer of 0 or more). The motor control device improves the distortion of the power supply input current caused by the suppression control of LC resonance by the rectifier circuit 2, the capacitor 4 which allows DC voltage pulsation, the reactor 5 which forms an LC filter, and the inverter circuit 3. Here, the motor control device 1 controls the output voltage of the inverter circuit 3 so that the output voltage vector pulsates at least once in synchronization with the period of the LC resonance between the reactor 5 and the capacitor 4, within the range where the phase Θ of one line voltage of the three-phase AC power supply 20 satisfies n×π / 3≦Θ≦n×π / 3+π / 6 (where n is a non-negative integer). Here, the motor control device 1 controls the magnitude of the output voltage vector of the inverter circuit 3 from positive to negative or negative to positive, such that the magnitude of the change in the output voltage vector of the inverter circuit 3 from positive to negative or negative to positive is inversely proportional to the effective value of the current flowing to the motor supplied with AC from the inverter circuit 3. In this case, a decrease in control stability or resonance suppression performance is prevented. Here, the motor control device 1 controls the output voltage of the inverter circuit 3 such that the DC voltage across the capacitors 4 contains 6 × m-th order (m is an integer greater than or equal to 0) harmonic components with the frequency of the line voltage as the fundamental wave, the amplitude of the 6 × m-th order (m is an integer greater than or equal to 0) harmonics is 6% or less of the DC component of the DC voltage, and the phase of the 6 × m-th order (m is an integer greater than or equal to 0) harmonics is π ± π / 16 with respect to the fundamental wave phase of the line voltage. In this case, the power factor of the input current on the power supply side is improved and power supply harmonics are suppressed. The motor control device 1 is characterized by controlling the output voltage of the inverter circuit 3 such that the transfer characteristic consisting of the DC voltage across the capacitors 4 and the maximum of the absolute values ​​of the three line voltages of the three-phase AC power supply 20 forms a second-order lag element, and the damping coefficient of the transfer function of the second-order lag element is greater than 0.7. In this case, the deterioration of resonance suppression performance is suppressed. The motor control device 1 controls the output voltage of the inverter circuit 3 so that the DC voltage across the capacitors 4 is less than or equal to the maximum value among the three absolute values ​​of the line voltages of the three-phase AC power supply 20. In this case, the deterioration of resonance suppression performance is suppressed.

[0075] Although embodiments have been described above, the technical scope of this disclosure is not limited to the embodiments described above. It is clear from the claims that combinations of two or more of the above embodiments, as well as various modifications or improvements to the above embodiments, are also included in the technical scope of this disclosure. [Explanation of Symbols]

[0076] 1...Motor control device, 2...Rectifier circuit, 3...Inverter circuit, 4...Capacitor, 5...Reactor, 10...Control unit, 11...Phase detection unit, 12...Voltage detection unit, 20...AC power supply, 21r, 21s, 21t...Power supply impedance, 65...Motor, 101...Phase estimation unit, 102...Voltage averaging unit, 103...Voltage command generation unit, 104...Comparison unit, 105...Gain correction unit, 106...Multiplier, 107...Operation variable limiter, 108...Modulation rate command generation unit

Claims

1. A rectifier circuit that rectifies and outputs the AC power output from a three-phase AC power supply, An inverter circuit having multiple switching elements, which converts the DC output from the rectifier circuit into AC through the switching operation of the multiple switching elements and supplies it to the motor, A DC link capacitor connected between the input nodes of the inverter circuit, which allows for voltage fluctuations output by the rectifier circuit and has a capacitance value set to suppress ripple voltage caused by the switching operation of the inverter circuit, The system comprises a reactor whose capacitance value is set to suppress ripple current caused by switching operation, between a three-phase AC power supply and the DC link capacitor, A motor control device having a filter function with the DC link capacitor and the reactor, A motor control device characterized by controlling the output voltage vector of the inverter circuit to change from positive to negative, or from negative to positive, based on the average value of the output voltage vector of the inverter circuit when the phase Θ is within the range that satisfies Equation 1, with the phase Θ of one line voltage of a three-phase AC power supply as the reference, starting at a timing where the phase Θ is π / 3 × n (where n is an integer greater than or equal to 0) and the phase Θ is within the range that satisfies Equation 1. (Equation 1) n × π / 3 ≤ Θ < (n + 1) × π / 3 (where n is a non-negative integer)

2. The motor control device according to claim 1, characterized in that, with reference to the phase Θ of one of the line voltages of the three-phase AC power supply, the output voltage of the inverter circuit is controlled such that the output voltage vector pulsates once or more in synchronization with the period of LC resonance by the reactor and the DC link capacitor, within a range where the phase Θ satisfies equation 2. (Equation 2) n × π / 3 ≤ Θ ≤ n × π / 3 + π / 6 (where n is a non-negative integer)

3. When controlling the magnitude of the output voltage vector of the inverter circuit from positive to negative or from negative to positive, The magnitude of the change in the output voltage vector of the inverter circuit from positive to negative or negative to positive is inversely proportional to the effective value of the current flowing to the motor supplied with AC from the inverter circuit. The motor control device according to claim 1, characterized by its ability to control.

4. The DC voltage between the DC link capacitors contains 6 × m (where m is an integer greater than or equal to 0) harmonic components with the frequency of the line voltage as the fundamental frequency, the amplitude of the 6 × m (where m is an integer greater than or equal to 0) harmonic being 6% or less of the DC component of the DC voltage, and the phase of the 6 × m (where m is an integer greater than or equal to 0) harmonic being π ± π / 16 with respect to the fundamental wave phase of the line voltage. The motor control device according to claim 1, characterized in that it controls the output voltage of the inverter circuit.

5. The transfer characteristic consisting of the DC voltage between the DC link capacitors and the maximum of the three absolute values ​​of the line voltages of the three-phase AC power supply forms a second-order lag element, and the damping coefficient of the transfer function of this second-order lag element is greater than 0.

7. The motor control device according to claim 1, characterized in that it controls the output voltage of the inverter circuit.

6. The DC voltage between the DC link capacitors is set to be less than or equal to the maximum value of the three absolute values ​​of the line voltages of the three-phase AC power supply. The motor control device according to claim 1 or 5, characterized in that it controls the output voltage of the inverter circuit.

Images