Optical signal demodulation device and optical signal demodulation method

Abstract

To solve such a problem that the whole interruption occurs if the optical loss exceeds 29 dB in the current high-capacity optical communication represented by the IEEE 802.3av standard.SOLUTION: An optical signal demodulation device includes: an index demodulation unit to which a complex digital signal obtained by converting a transmitted optical signal is converted and which decodes an index decoding code on the basis of the position of the symbol within a time slot within one frame; and a symbol demodulation unit to which the complex digital signal is input and which demodulates the symbol placed in the time slot to a symbol decoding code (DEAP). When receiving the time slot with the symbol, the index demodulation unit transmits a notification signal indicating the presence of the symbol to the symbol demodulation unit. Since the symbol demodulation unit can decode the symbol on the basis of the notification signal and an optical signal demodulator can demodulate the index signal with a low communication rate, communication can be maintained without whole interruption even if the optical loss exceeds 29 dB.SELECTED DRAWING: Figure 8

Description

[Technical field]

[0001] The present invention relates to an optical signal demodulator that is applied to an optical access network, particularly an optical fiber transmission network. The present invention realizes an optical signal demodulator that can receive an optical signal based on a time domain index optical modulation method that can maintain a connection of a communication line even with a low average optical signal power compared to the conventional technology.

[0002] By using the optical signal demodulator of the present invention, it is possible to realize the same operation as the frequency offset compensation circuit, phase compensation circuit, and phase slip reduction circuit built into a conventional digital coherent optical signal receiver, even for optical symbols arriving at irregular timings in accordance with the time domain index modulation rules of a time domain index optical modulation signal. This makes it possible not only to transmit and receive codes in accordance with the time domain index modulation rules, but also to coherent detection of arriving optical symbols. In other words, like a conventional digital coherent optical signal receiver, it is possible to transmit and receive codes in accordance with the M-phase quadrature phase amplitude optical modulation rules. [Background technology]

[0003] Optical access networks have spread to more than 20 million households in Japan and have become an essential social infrastructure system in our highly information-driven society.

[0004] In particular, in recent years, optical access network services that realize a maximum transmission rate of 10 Gbit / s in each household are being developed in accordance with the IEEE 802.3av (10GE-PON) standard (Non-Patent Document 1). In many cases, this system adopts a configuration in which optical fiber lines are branched in a tree shape from an Optical Line Terminal (OLT) installed in a communication station in each local city, and connected to multiple (maximum 32) Optical Network Units (ONUs) installed in each household (Figure 30).

[0005] In this system, the optical loss budget between the OLT and the ONU is 29 dB, and the transmission distance is 20 km. If these conditions are no longer met, the communication line cannot be maintained and the 10 Gbit / s transmission rate will be completely cut off.

[0006] On the other hand, when the time domain index optical modulation method (Non-Patent Document 2) is adopted, it becomes possible to maintain the optical signal-to-noise ratio by increasing the optical signal peak power while keeping the average optical power constant by thinning out optical symbols in accordance with the change in the transmission line conditions between the OLT and the ONU (Fig. 31). At this time, the absolute value of the mutual information of the communication line decreases, but it becomes possible to maintain the normalized mutual information at a value close to 1 bit / symbol, and the connectivity of the communication line can be maintained with the support of the error correction function. [Prior art documents] [Patent documents]

[0007] [Patent Document 1] International Publication No. 2014 / 115840 (Patent No. 5886984) [Non-patent literature]

[0008] [Non-Patent Document 1] IEEE Std 802.3av-2009, IEEE Standard for Ethernet-Physical Layer Specifications and Management Parameters for 10Gb / s Passive Optical Networks. [Non-Patent Document 2] M. Nakao, T. Ishihara, and S. Sugiura, “Dual-Mode Time-Domain Index Modulation for Nyquiest-Criterion and Faster-Than-Nyquist Single-Carrier Transmissions,” IEEE Access, vol. 5, no. 11, pp. 27659-27667, Nov. 2017. [Non-Patent Document 3] K. Kikuchi, “Fundamentals of coherent optical fiber communications,” J.Lightw. Technol., vol.34, no.1, pp.157-179, Aug. 2016. [Non-Patent Document 4] H. Imai and S. HIRAKAWA, “A New Multilevel Coding Method Using Error-Correcting Codes,” IEEE Trans.On Information Theory, vol.IT-23, no.3, pp.371-377, Aug. 1977. [Non-Patent Document 5] Senta Suzuki, Yutaka Miyamoto, Masato Tomizawa, Hisakazu Sakano, Koichi Murata, Shinji Mino, Mitsufumi Shibayama, Masamasa Shibuya, Kiyoshi Fukuchi, Hiroshi Onaka, Takeshi Hoshida, Kosuke Komaki, Takashi Mizuochi, Kazuo Kubo, Yoshikuni Miyata, and Toru Kamio, “Research and Development of Digital Coherent Signal Processing Technology for Increasing the Capacity of Optical Communication Networks,” Journal of the Institute of Electronics, Information and Communication Engineers, vol. 95, no. 12, pp. 1100-1116, December 2012. [Non-Patent Document 6] K.Itoh, “Analysis of the phase unwrapping algorithm,” Applied Optics, vol. 21, no.14, pp.2470, July 1982. [Non-Patent Document 7] A. Alvarado, T. Fehenberger, B. Chen, and FJ Willems, “Achievable Information Rates for Fiber Optics Applications Aand Computations,” J.Lightw. Technol., vol.36, no.2, pp.424-439, Jan. 2018. [Non-Patent Document 8] J. Li, S. Lin, K. Abdel-Ghaffar, WERyan, and D. L. Costello, Jr., “LDPC Code Designs, Constructions, and Unification, Chapter 11” Cambridge University Press, 2017, ISBN978-1-107-17568-6. [Non-Patent Document 9] K. Kikuchi, T. Okoshi, M. Nagamatsu, and N. Henmi, “Degradation of bit-error rate in coherent optical communications due to spectral spread of the transmitter and the local oscillator,” J. Lightw. Technol., vol. 2, no.6, pp. 1024-1033, Dec., 1984. [Non-Patent Document 10] S. Ishimura and K. Kikuchi, “Multi-dimensional permutation modulation format for coherent optical communications,” Opt. Exp., vol. 23, no. 12, pp. 15587-15597, 2015. Summary of the Invention [Problem to be solved by the invention]

[0009] As represented by the IEEE 802.3av (10GE-PON) standard (Non-Patent Document 1), phase diversity coherent optical communication is expected to become the basis for large-capacity optical communication. However, due to the fact that the communication capacity per communication is increasing due to the spread of video content, the social demand for extending the transmission distance is expected, and the number of accesses increases during disasters, etc., there may be cases in the future where the optical loss budget exceeds 29 dB.

[0010] Even in such cases, it is considered necessary to provide a best-effort optical access service according to the communication traffic state and the transmission line conditions. In other words, the object of the present invention is to provide a demodulation device for optical communication that can maintain the communication state even if the optical loss budget exceeds 29 dB, even if the communication rate is reduced.

[0011] In addition, in order to achieve the above object, the present invention combines a coherent optical modulation method and a time-domain index optical modulation method. In this case, there is a problem that it is necessary to perform frequency offset compensation, phase compensation, etc. for intermittently arriving optical symbols in the same way as in a conventional coherent optical signal receiving device. [Means for solving the problem]

[0012] In view of the above problems, the present invention provides a demodulation device and method for optical communications that provides a best-effort optical access service in accordance with the communication traffic state and transmission line conditions even when the optical loss budget exceeds 29 dB.

[0013] More specifically, the optical signal demodulation device according to the present invention comprises: 1 1 An optical signal demodulation device that receives an optical signal transmitting an original signal using a frame composed of a plurality of time slots, a symbol carried on at least one of the time slots, and an index code formed at a time position of the time slot carrying the symbol, and demodulates the original signal, comprising: a receiver (RX) that performs phase diversity detection on the optical signal and converts it into a complex digital signal (CS); The complex digital signal (CS) is input. Notes An index decoding code (DE) is generated based on the position of the symbol in the time slot in the frame. Id an index demodulation process unit (10) for demodulating the The complex digital signal (CS) is input, and the symbol carried on the time slot is decoded into a symbol decoding code (DE AP ) a symbol demodulation process unit (20) for demodulating the When the index demodulation process unit (10) receives the time slot carrying the symbol, it transmits a notification signal (activation pattern signal AP CS ) and The symbol demodulation process unit (20) receives the notification signal (activation pattern signal AP CS ) to obtain and decode the symbol.

[0014] Further, the optical signal demodulation method according to the present invention comprises the steps of: 1. An optical signal demodulation method for receiving an optical signal that transmits an original signal using a symbol carried on at least one of the time slots and an index code formed at a time position of the time slot carrying the symbol, the optical signal demodulating method comprising the steps of: a receiving step (RX) of detecting the optical signal by phase diversity and converting it into a complex digital signal (CS); The complex digital signal (CS) is input, and an index decoding code (DE) is generated based on the position of the symbol in the time slot within the one frame. Id (10) an index demodulation step of demodulating the When the time slot is received, a notification signal (activation pattern signal AP CS ) and The complex digital signal (CS) is input, and the notification signal (activation pattern signal AP CS The symbol carried in the time slot is decoded based on the symbol decoding code (DE AP ) into a symbol demodulation step (20). Effect of the Invention

[0015] According to the optical signal demodulation device and method of the present invention, communication can be maintained by transmitting using the coherent optical modulation method as long as the optical loss does not exceed 29 dB, and by also using the time-domain index modulation method when the optical loss exceeds 29 dB.

[0016] Furthermore, the present invention realizes operations similar to those of the frequency offset compensation circuit, phase compensation circuit, and phase slip reduction circuit built into conventional digital coherent optical signal receivers for optical symbols arriving at irregular timings in accordance with the time-domain index modulation rules of a time-domain index optical modulation signal. This not only enables transmission and reception of codes in accordance with the time-domain index modulation rules, but also enables coherent detection of arriving optical symbols. In other words, like conventional digital coherent optical signal receivers, it is possible to transmit and receive codes in accordance with the M-phase quadrature phase amplitude optical modulation rules.

[0017] As a result, the present invention can realize a plurality of operation modes as described below. Specifically, in a normal operation mode, a system employing the present invention realizes data transmission by M-phase quadrature amplitude modulation (M-QAM) optical modulation method, and ensures communication capacity (normal mode). On the other hand, when customer data traffic is low, a system employing the present invention switches to an index optical modulation operation mode, and reduces the average optical signal power while ensuring the peak power of the optical signal (ECO mode). On the other hand, when the transmission line distance is long or the transmission line conditions are poor, the system switches to the index optical modulation operation mode, and increases the peak power of the optical signal to ensure the connectivity of the communication line (Long Reach / Fault Tolerant mode). In this way, the present invention makes it possible to provide a "best effort" optical access service in accordance with the communication traffic state and the transmission line conditions. [Brief description of the drawings]

[0018] [Figure 1] 1 shows an example of a modulation rule in a time domain index optical modulation method. [Diagram 2] An example of a modulation rule in an M-phase quadrature amplitude modulation system is shown. [Diagram 3] FIG. 1 is a diagram showing the relationship between the transmission path loss (which can be converted into the transmission distance) and the communication capacity in the time-domain index optical modulation method that is to be realized by the present invention. [Figure 4] FIG. 1 shows a logical model diagram of a time domain index optical modulation method to be realized in the present invention. [Diagram 5] FIG. 1 is a diagram for explaining the effects realized by a system to which the present invention is applied. [Figure 6] FIG. 2 is a diagram showing the optical frequency difference between an optical signal and a local light. [Figure 7] FIG. 2 illustrates a transmission mode in which the invention can be implemented. [Figure 8] 1 is a diagram showing a configuration of an optical communication receiving device according to the present invention; [Figure 9] 2 is a diagram showing a configuration of a demodulation unit of the optical communication receiving device of the present invention; [Figure 10]FIG. 2 is a diagram showing the frame and symbol configuration of the index optical modulation method. [Figure 11] FIG. 2 is a diagram illustrating a configuration of a frequency offset compensation unit of a demodulation unit. [Figure 12] FIG. 13 is a diagram illustrating the effect of frequency offset compensation. [Figure 13] FIG. 2 is a diagram illustrating a configuration of a phase compensation unit of a demodulation unit. [Figure 14] 11A and 11B are diagrams illustrating the effect of a phase compensation unit. [Figure 15] FIG. 13 is a diagram illustrating a configuration in which a provisional decision calculation unit is used in the frequency offset compensation unit. [Figure 16] FIG. 2 is a diagram illustrating a configuration of a provisional decision calculation unit. [Figure 17] FIG. 13 is a diagram illustrating a configuration of a phase compensation unit using a provisional decision calculation unit. [Figure 18] FIG. 13 is a diagram illustrating a configuration of a provisional decision calculation unit used in the phase compensation unit. [Figure 19] FIG. 13 is a diagram showing the configuration of another demodulation unit. [Figure 20] FIG. 13 is a diagram illustrating a configuration of another phase compensation unit. [Figure 21] FIG. 13 is a diagram illustrating a configuration of an averaging unit according to the third embodiment. [Figure 22] FIG. 13 is a diagram illustrating a configuration of a demodulation unit according to a fourth embodiment. [Diagram 23] FIG. 13 is a diagram illustrating a configuration of a phase slip guaranteed phase compensation unit used in a fourth embodiment. [Figure 24] FIG. 13 is a diagram illustrating a configuration of a demodulation unit according to a fifth embodiment. [Diagram 25] FIG. 13 is a diagram illustrating a configuration of an IndexBit demapping unit with a soft-decision FEC function used in the fifth embodiment. [Figure 26] 4 is a graph showing simulation results of the error rate (BER) as a function of the signal-to-noise ratio (SNR) for a transmission scheme according to the invention; [Figure 27]A typical constellation diagram when a 2.5 Gbps time-domain (8,1,BPSK) optical signal is polarization-multiplexed. (a) shows the result before frequency correction, and (b) shows the result after frequency and phase correction. [Figure 28] Graphs showing experimental results showing the relationship between received optical power and error rate. (a) shows the BER of a 2.5 GBaud polarization multiplexed time-domain (8,1,BPSK) optical signal as a function of received optical power, and (b) shows the BER of a 2.5 GBaud polarization multiplexed time-domain (8,1,DBPSK) optical signal as a function of received optical power. [Figure 29] The BER of a time-domain (8,1,BPSK) optical signal and a BPSK optical signal are shown as a function of frequency offset normalized to the symbol rate, both measured and simulated. [Diagram 30] 1 is a diagram showing the relationship between an OLT and an ONU. [Diagram 31] FIG. 13 is a diagram showing a signal configuration when an index modulation method is used. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0019] The configuration and operation of the optical signal demodulation device according to the present invention will be described below with reference to the drawings. Note that the following description illustrates one embodiment of the present invention and an example, and the present invention is not limited to the following description. The following description can be modified within the scope of the present invention. Furthermore, the parts referred to as "~ unit" in the following embodiments can be realized by either a hardware configuration or a software configuration. Furthermore, the parts referred to as "~ unit" may be referred to as "processes" having the functions to be performed. Furthermore, the addition unit, multiplication unit, and division unit are elements that perform complex calculations, even if they are not described as "complex."

[0020] The principle of the time-domain index optical modulation method, which is the premise of this invention, is as follows. In the time-domain index optical modulation method, an index optical modulation signal frame (hereinafter simply called a "frame") is composed of multiple time slots. Optical signals are transmitted and received using this frame as a single unit. A time slot has a predetermined time width. There is a time gap between one time slot and the next time slot. However, if the sending side and receiving side are completely synchronized, there is no need for a time cap. The time width of a time slot, including the time gap, is represented by "Ts".

[0021] Under these conditions, we consider the case where the average optical signal power of each frame is operated at a constant value near the upper limit of the transmitter. In this case, by thinning out a fixed number of optical symbols in each frame, the amplitude of the optical symbols, and therefore the signal-to-noise ratio of the optical symbols, increases. In addition, it becomes possible to transmit and receive codes by changing the transmission and reception time slots of the thinned optical symbols in accordance with the time-domain index modulation rules (Figure 1).

[0022] Here, a symbol refers to light (optical symbol carrier: hereafter also simply referred to as "optical signal LS") that carries information from the sender during a time slot. Symbol thinning refers to a state in which there is no light or the intensity is low (dark) during a time slot. Changing the time slot includes changing the time of the time slot or changing the intensity of the symbol during the time slot.

[0023] Figure 1 shows eight frames. Each frame consists of eight time slots. In each time slot, the black time slots carry optical symbols. Information is carried in the optical symbols using the M-QAM method, such as QPSK. For example, the third frame shows that the third time slot carries an optical symbol. The white time slots are time slots that do not carry optical symbols. Figure 1 shows the case where an optical symbol is carried in one of the eight time slots. Eight activation patterns are possible depending on the position of the optical time slot carrying the optical symbol. By assigning an index bit to each activation pattern, three bits of information can be specified.

[0024] Furthermore, by simultaneously performing coherent detection of the optical symbols, it becomes possible to transmit and receive codes using M-QAM optical modulation symbols in the same way as conventional digital coherent optical signal receivers. Figure 2 shows an example of signal points of optical symbols sent in a time slot. Figure 2 (a) shows a QPSK (Quadrature Phase Shift Keying) signal, and (b) shows a 16QAM (16 Quadrature Amplitude Modulation) signal.

[0025] Figure 3 shows an example of the relationship between the normalized mutual information of the QPSK method (thin line) and the (8,1,QPSK) time domain index optical modulation method (thick line) of the present invention, which will be described in detail later. "(8,1,QPSK)" means that one frame is composed of eight time slots, and communication using the index modulation method is performed by placing a symbol in one of the time slots, and that the symbol is modulated by QPSK to carry information.

[0026] Referring to Figure 3, the horizontal axis represents optical loss (dB) and the vertical axis represents normalized mutual information R (bits / symbol). In the existing QPSK optical modulation method, when the optical loss budget of 29 dB is exceeded, bit errors are introduced and the communication line is cut off. Here, the communication line is completely cut off when the normalized mutual information R is 0.81 bits / symbol or less.

[0027] However, by implementing a (forward) error correction function (labeled "FEC: Forward Error Correction" in Figure 3) in the receiver and adopting a time-domain index optical modulation method (labeled "Index modulation"), the margin of the optical loss budget can be increased by a total of about 10 dB. Assuming that the optical fiber loss coefficient of the optical access network is 0.5 dB / km, this value is equivalent to the optical loss in a 20 km section. In other words, this suggests that the transmission distance of the access optical network can be doubled to 40 km.

[0028] The communication path realized by the time-domain index optical modulation method is logically a communication path consisting of a parallel index bit (hereinafter referred to as "INDBit") channel that transmits and receives codes by changing the symbol position within one frame of the optical symbol transmission and reception time slot in accordance with the time-domain index modulation rules, and an amplitude phase modulation bit (hereinafter referred to as "APMBit") channel that transmits and receives codes using M-QAM optical modulation.

[0029] Figure 4 is a conceptual diagram of the overall configuration of a communication system using INDBit and APMBit. On the transmitting side (optical signal transmitting device 1), the original information INF is divided into INDBit and APMBit in the serial / parallel conversion unit (S / P). Each signal has an error correction code added (FEC Enc.), is mapped (MAP), and then is transmitted from the transmitting unit (TX).

[0030] Logically, the INDBit channel (Index ch) and the APMBit channel (APM ch) are transmitted on the transmission path 2. Physically, the optical signal LS with a phase modulation signal superimposed thereon is sent during the time slot shown in Fig. 1.

[0031] On the receiving side (optical signal demodulator 3), the signal received at the receiver (RX) is subjected to index demodulation process 10, where the index modulated signal is decoded (DMAP) and error correction is performed (FEC Dec.). Also, in symbol demodulation process 20, the amplitude phase modulated signal is frequency offset compensated (FC), phase shift compensated (PC), decoded (DMAP), and error correction is performed (FEC Dec.). Also, in the optical signal demodulator 3 of the present invention, an activation pattern signal AP, which will be described later, is transmitted from index demodulation process 10 to symbol demodulation process 20. CS In this way, INDBit and APMBit are demodulated. These signals are converted back to serial signals by the parallel / serial converter 29, and decoded information DE(INF) corresponding to the original information INF is obtained. The overall communication capacity C (bit / sec) of these is given by equation (1).

[0032]

number

[0033] Here, N is the number of symbols per frame of the time-domain index optical modulation signal, K is the number of symbols to insert optical symbols into, [ ] is the Gaussian symbol, and M is the number of signal points defined for the M-QAM modulated optical symbols. The first term on the right-hand side corresponds to the communication capacity of the INDBit channel, and the second term corresponds to the communication capacity of the APMBit channel.

[0034] Figure 5 shows the relationship between the number of optical symbols per frame (K / N) and the transmission data rate B (bits / symbol). The open marks indicate the case where only amplitude-phase modulation bits (APMbits) are used, and the filled marks indicate the case where INDBit is also used. Each APMbit is QPSK. The square marks indicate the case where one frame has 16 time slots, the circle marks indicate the case where one frame has 8 time slots, and the triangle marks indicate the case where one frame has 4 time slots. Here, there are two methods for transmitting codes under a low average optical signal power state in which the average number of optical symbols is narrowed down to 1 / 16 to 1 / 4: a method of allocating optical symbols to fixed time slots (referred to as FIM in Figure 5) and a time-domain (N,K,QPSK) index optical modulation method (referred to as IM in Figure 5).

[0035] For example, in the time-domain (8,1,QPSK) index optical modulation transmission method (in Figure 5, the circle indicates K where K is 1: K / N=0.125), which transmits codes under low average optical signal power conditions with the average number of optical symbols narrowed down to 1 / 8, the communication capacity can be expanded by approximately 2.5 times compared to the 1 / 8 rate QPSK modulation transmission method (0.25 bits / symbol) in which the number of optical symbols is narrowed down to 1 / 8.

[0036] The present invention is an optical signal demodulation device and method for demodulating an optical signal in which one frame is composed of multiple time slots and symbols are carried in at least one of the time slots, as shown in Fig. 1. In other words, the present invention aims to apply an intradyne digital coherent detection method (Non-Patent Document 3) to an optical receiver in an optical access network that employs a time-domain index optical modulation method. In this optical signal demodulation device, in order to correctly decode the received optical symbols, it is necessary to mainly solve the following problems. (1) Under conditions in which optical symbols arrive irregularly in accordance with index optical modulation rules, it is necessary to perform frequency offset compensation and phase compensation. (2) The implementation circuit must be applicable to conventional M-QAM modulation transmission.

[0037] The problem (1) is an issue that inevitably arises when applying the time-domain index optical modulation method. In conventional M-QAM optical modulation symbols, an optical symbol is always present in each time slot, and the frequency offset and phase difference between the received optical symbol and the local laser light are extracted by interfering with this optical symbol and the local laser light built into the receiver. However, in the present invention, it is necessary to extract the frequency offset and phase difference between the received optical symbol and the local laser light from the M-QAM optical modulation symbols that arrive at the optical receiver irregularly.

[0038] 6 shows the relationship between the optical signal LS and the frequency of the local light Lo used in the optical signal demodulator 3. The horizontal axis represents the optical frequency, and the vertical axis conceptually represents the optical intensity. In the optical signal demodulator 3, it is desirable to use light of the same frequency as the optical signal LS as the local light Lo, but it is not easy to finely adjust the frequency of the local light Lo, and the two lights are shifted in frequency by an optical frequency difference Δf (hereinafter also referred to as the "offset frequency Δf").

[0039] The object of (2) is derived from the objective of making the transmission rate variable in the optical access network system that can be realized by the present invention. In the present invention, the transmission rate can be made variable by thinning out optical symbols in accordance with changes in the transmission path conditions between the OLT and the ONU in the optical access network system to be realized.

[0040] Figure 7 shows one frame transmitted in the section from the OLT (see Figure 30) to the ONU. When the transmission line conditions are good, conventional M-QAM optical modulation symbols are transmitted (Figure 7(a): normal mode). On the other hand, when the transmission line conditions are poor and exceed the optical loss budget of 29 dB, operation is performed based on the time domain index optical modulation method, and optical symbols are thinned out (Figure 7(b): ECO mode). In an optical receiving device that realizes such operation, it is necessary to make it possible for the same optical receiver to demodulate both M-QAM optical modulation symbols and time domain index optical modulation symbols, and to increase the common parts of the receiving device to efficiently realize the expected operation.

[0041] Furthermore, when the transmission path distance is long or the transmission path conditions are poor, the operation mode is switched to index optical modulation and the peak power of the optical signal is increased to ensure connectivity of the communication line (Fig. 7(c): LR / FT mode [Long Reach / Fault Tolerant mode]). The optical modulation operation itself is the same as in ECO mode, but the peak power of the optical signal is increased. In this case, the INDbit demodulator determines whether or not there is an optical symbol, and then generates an activation pattern, which is input to the APMbit demodulator. The APMbit demodulator demodulates the M-QAM optical modulation symbol according to this activation pattern only for time slots where an optical symbol exists.

[0042] In the ECO mode of Fig. 7(b), under conditions where optical symbols arrive irregularly in accordance with the index optical modulation rules, it is necessary to decode the transmission code of the INDbit channel according to the index modulation rules while simultaneously performing frequency offset compensation and phase compensation to decode the transmission code of the APMbit channel.

[0043] The present invention provides a means for realizing an intradyne digital coherent detection optical signal decoding circuit that can support a time-domain index optical modulation method, which can solve these problems.The present invention not only provides a means for solving problem (1), but also satisfies the requirements of problem (2) and provides a means for efficiently realizing decoding of conventional M-QAM optical modulation symbols and decoding of time-domain (8,1,M-QAM) index optical modulation symbols while increasing the common use parts of the optical signal demodulation circuit.

[0044] The optical signal LS that can be demodulated by the optical signal demodulation device according to the present invention is a signal in which one frame is composed of N time slots, and M-QAM symbols are carried in K of the slots. The optical signal LS modulated in this way is called a "time domain (N, K, M-QAM) index optical modulated signal."

[0045] (First embodiment) Figure 8 shows a block diagram of the main parts of the optical signal demodulator 3. The optical signal LS transmitted through the optical fiber is received by the phase diversity detector RXa and converted into a digital signal by the analog-to-digital converter RXb. The polarization relationship is then compensated for by the optical polarization compensation unit RXc. The optical polarization compensation unit RXc includes a dispersion compensation unit, a timing phase synchronization unit, a polarization dispersion compensation unit, etc. These constitute the receiver unit RX, which performs phase diversity detection on the optical signal LS and converts it into a complex digital signal CS. The receiver unit RX outputs the complex digital signal CS. The complex digital signal CS still contains the frequency difference Δf (see Figure 6) between the local light Lo and the optical signal LS, and the phase error of the carrier component of the M-QAM signal.

[0046] The complex digital signal CS output from the receiver RX is demodulated by the index demodulation process 10 to convert the INDbit channel signal into a transmission code DE. Id The symbol demodulation process 20 performs frequency offset compensation and phase compensation to convert the APMbit channel signal into the transmission code DE AP The transmission code DE is decoded as follows. Note that error correction may be performed in the index demodulation process 10 or the symbol demodulation process 20. Furthermore, error correction may be performed in the final stage of decoding. Id and transmission code DE AP is converted back to serial data by a parallel / serial converter 29 (referred to as a "P / S converter"), and a decoded signal DS(INF) is obtained by decoded the original signal INF. The optical signal demodulator 3 according to the present invention only needs to have a demodulator 6 located after the receiver RX. The demodulator 6 only needs to have an index demodulation process unit 10 and a symbol demodulation process unit 20. The transmission code DE Id is the index decoded code DE Id This is called the transmission code DE AP The symbol decoded code DE AP It may also be called.

[0047] The optical signal demodulation performed by the demodulator 6 to realize the time domain (N, K, M-QAM) index optical modulation method will be explained with reference to Fig. 9. The complex digital signal CS (a signal consisting of 2CHs of I-axis component and Q-axis component) output from the receiver RX shown in Fig. 8 is input to the demodulator 6. The complex digital signal CS is branched at a branching unit TIN and input to an amplitude calculation unit 11 of the index demodulation process unit 10 and a delay unit 21 of the symbol demodulation process unit 20.

[0048] The amplitude calculation unit 11 of the index demodulation process unit 10 calculates the absolute value (or the square of the absolute value) of the complex digital signal CS, calculates the intensity of the complex digital signal CS corresponding to the received optical symbol, and sets this as an output value. This output is called the symbol intensity P CS Let us assume that.

[0049] The frame timing extraction unit 12 extracts the output value of the amplitude calculation unit 11 (activation pattern signal AP CS ) and the code pattern decoded by the index demodulation process unit 10 (index decoded code DE Id ) to extract the head timing of the index optical modulation frame. Then, a head notification signal S indicating the head timing is generated. TOP is output to the IndexBit demapping unit 14.

[0050] <Frame timing extraction unit 12> In the case of a time domain (N, K, M-QAM) index optical modulation method, the frame timing extraction unit 12 grasps the start timing of a frame consisting of N symbols. Fig. 10 shows a frame consisting of N symbols. With reference to Fig. 10, the horizontal axis represents time. The further to the right it is, the more past the past it is. In Fig. 10, one frame is made up of eight time slots (white rectangles), and a symbol (black rectangle) is placed in one time slot in one frame. Fig. 10 shows a frame consisting of one symbol.

[0051] Referring again to Fig. 9, there are two main methods for implementing the frame timing extraction unit 12: (1) a method for grasping the head timing from a training symbol included in the main signal optical symbol, and (2) a method for extracting timing from a control optical channel signal.

[0052] In this embodiment, as shown in Fig. 10, a method of extracting from a training frame included in a main signal optical symbol (1) is exemplified. Of course, timing may be extracted from a control optical channel signal (2). In this case, a specific frame pattern consisting of N symbols is arranged in succession for J frames, and this is used as a training frame. Here, a symbol is always placed in the first time slot of a training frame. Such training frames are inserted at a rate of about 1 / 100 to 1 / 10 of a transmitted signal in existing digital coherent transmission systems, and a method of outputting a frame timing signal in accordance with the symbol placed at the beginning of the frame is known (Non-Patent Document 5).

[0053] Furthermore, the method of determining the start timing of a frame from a specific bit pattern is a traditional method that is also used in many existing transmission systems. For example, the start position of an Ethernet frame is extracted from a 1010101 pattern while performing timing synchronization. Also, Synchronous Digital Hierarchy (SDH) transmission systems that have been used in 10-gigabit-class transmission systems are designed to be able to specify the start position of a frame by inserting a specific code pattern. The frame timing extraction unit 12 of the present invention also recognizes the start symbol of a frame using such a known method.

[0054] In this embodiment, the head notification signal S is generated while checking that the training symbol is placed at the head symbol of the frame from the bit pattern decoded by the IndexBit demapping unit 14 described later. TOP The start notification signal S, which indicates the start time slot of the frame, is output at the appropriate timing. TOPFor example, when the system is designed according to the time domain index modulation rule as shown in FIG. 1, the frame timing extraction unit 12 outputs the index decoding code DE corresponding to the activation pattern of “000” (or “10000000”) from the IndexBit demapping unit 14. Id ) is output, the head notification signal S TOP is controlled so that

[0055] <Index demodulation process unit 10> Next, the operation of the index demodulation process unit 10 will be described. The delay unit 13 outputs the head notification signal S in accordance with the frame head symbol when the frame timing extraction unit 12 is correctly driven. TOP is sent and input to the IndexBit demapping unit 14 and the timing at which the first symbol of the frame is input to the IndexBit demapping unit 14 are synchronized.

[0056] The IndexBit demapping unit 14 receives the frame timing signal S from the frame timing extraction unit 12. TOP and the output signal of the delay unit 13 (absolute value of the complex digital signal CS) P CS The IndexBit demapping unit 14 receives the head notification signal S TOP Based on this, the frame timing of the Index symbol is identified, and the symbol intensity P CS The IndexBit is demapped from the absolute value (or the square of the absolute value) of the complex digital signal CS calculated by the amplitude calculation unit 11, and the decoded code, that is, the index decoded code DE Id Output.

[0057] The demapping rule may be based on the time domain index modulation rule of FIG. 1. In this case, the input signal from the delay unit 13 (symbol intensity P CS ) is the activation pattern signal AP CSThe activation pattern signal AP is decoded by demapping the activation pattern to index bits. CS From this, it is possible to know the time slot in which a symbol exists in one frame.

[0058] <Register section 16> Referring again to FIG. 9, the decoded index decoded code DE Id The register unit 16 receives the head notification signal S TOP The signal is driven based on the head notification signal S TOP Only when the index bit demapping unit 14 is ON, the index decoded code DE Id The register unit 16 receives and stores the error code and decoding process.

[0059] On the other hand, the IndexBit demapping unit 14 decompresses the input signal (symbol intensity P CS ) is determined, so the activation pattern signal AP CS This activation pattern signal AP CS is transmitted to the frequency offset compensation unit 22 and the APMBit demapping unit 25 of the symbol demodulation process unit 20. The above is the operation of the index demodulation process unit 10.

[0060] <Symbol demodulation process unit 20> Next, the operation of the symbol demodulation process unit 20 will be described. The delay unit 21 delays the activation pattern signal AP CS arrives at the frequency offset compensation unit 22 in synchronization with the timing at which the complex digital signal CS branched by the branch unit TIN arrives at the frequency offset compensation unit 22. In other words, the frequency offset compensation unit 22 can obtain a time slot containing a symbol in the complex digital signal CS.

[0061] The purpose of the frequency offset compensator 22 is to estimate the offset frequency Δf in the complex digital signal CS and to remove the component. The offset frequency Δf in the complex digital signal CS is generated in the phase diversity detector RXa (see FIG. 8) installed in the receiver RX of the present invention (see FIG. 6).

[0062] The phase diversity detector RXa tries to extract the phase information of the received optical symbol by interfering with the local laser light to detect the received optical symbol, but the optical frequencies of the received optical symbol and the local laser light do not necessarily match, and there is an optical frequency difference (Fig. 6). This optical frequency difference becomes the oscillation frequency of the electrical signal, and becomes the offset frequency Δf of the complex digital signal CS.

[0063] <Frequency offset compensator 22> Next, the frequency offset compensator 22 will be described in detail with reference to Fig. 11. The frequency offset compensator 22 of the present invention removes the influence of modulation from the complex digital signal CS by a method called the power method, and extracts the phase rotation of the received optical symbol carrier.

[0064] The complex digital signal CS is input to the frequency offset compensation unit 22 from the delay unit 21. This complex digital signal CS is branched, and one of the branched complex digital signals CS is input to a power calculation unit 221. Here, the data modulation component contained in the complex digital signal CS is removed.

[0065] For example, in the case of a complex digital signal CS generated from QPSK optically modulated optical symbols, the phase angles of the data modulation components are π / 4, 3π / 4, -π / 4, and -3π / 4 (rad), so the complex representation of the complex digital signal CS is given by equation (2).

[0066]

number

[0067] To remove the effect of the data modulation component, a fourth power operation is performed. In other words, by raising both sides of equation (2) to the fourth power, it is expressed as equation (3).

[0068]

number

[0069] The phase angles of the data modulation components, π / 4, 3π / 4, -π / 4, -3π / 4 (rad), are eliminated by raising them to the fourth power. Here, Δωt is the angular frequency expression of the frequency offset Δf. In other words, the influence of the data modulation components is eliminated, and the optical angular frequency offset Δω and optical phase difference Δθ (phase shift) between the carrier component of the optical symbol and the local light Lo ("carrier offset-phase component CC ωθ This carrier offset and phase component CC ωθ to the sample and hold units 222 and 225. ωθ If we write it explicitly, it becomes as equation (4).

[0070]

number

[0071] The sample and hold units 222, 224, and 225 generate a timing signal (or an activation pattern signal AP) generated at the timing when the IndexBit demapping unit 14 determines that an optical symbol exists. CS ) arrives, the carrier offset and phase components CC ωθ This holds the activation pattern signal AP CS The carrier offset and phase component CC corresponding to the optical symbol at the timing when ωθ can be held.

[0072] The output of the sample and hold unit 222 (carrier offset and phase components CC ωθ) is input to a delay unit 223. This delay unit 223 gives a delay equivalent to the time slot time of the optical symbol, and the signal is input to a sample and hold unit 224. CS ) arrives twice in succession at equal intervals with respect to the optical symbol time slot, the complex digital signal held by the sample and hold unit 224 is the carrier offset and phase components CC ωθ In comparison, the past value for one optical symbol time slot is held.

[0073] Also, if the timing signal (or activation pattern signal AP CS ) is held in the sample and hold unit 224, the carrier offset and phase components CC ωθ is the carrier offset and phase component CC held in the sample and hold units 222 and 225. ωθ In comparison, past values ​​for N optical symbol time slots are held.

[0074] The carrier offset and phase components CC held by the sample and hold unit 224 are ωθ "CC ωθ1 " and the carrier offset and phase components CC held by the sample and hold units 222 and 225 are ωθ and "CC ωθ2 " Carrier offset phase component CC ωθ1 is the carrier offset and phase component CC ωθ2 Carrier offset and phase components of the previous symbol CC ωθ It is.

[0075] In either case, the activation pattern signal AP CS The carrier offset and phase component CC corresponding to the optical symbol at the timing when ωθThe carrier offset and phase components CC held in the sample and hold units 224 and 225 are also stored. ωθ can be calculated from the number of optical symbol time slots from when the data is held in the sample & hold unit 224 until when the data is held in the sample & hold unit 225. This makes it possible to calculate the phase difference per unit time even in a situation where the arrival of optical symbols is intermittent due to the introduction of index optical modulation, and therefore to reproduce the optical angular frequency offset Δω (corresponding to the offset frequency Δf) between the optical symbol and the local light Lo.

[0076] The multiplier 226 multiplies the carrier offset and phase components CC ωθ1 Then, the complex conjugate of the carrier offset and phase component CC ωθ2 is multiplied by the multiplier 226 to obtain a signal from which the optical phase angle difference between the carrier component of the optical symbol and the local light (hereinafter, Δθ) has been removed (this is referred to as the carrier offset component CC ω That is, it can be obtained as shown in equation (5).

[0077]

number

[0078] This carrier offset component CC ω (t) is the carrier offset phase component CC, which is the input signal from the sample and hold unit 225. ωθ2 and the carrier offset and phase components CC, which are input signals from the sample and hold unit 224. ωθ1 Here, CC ωθ1 * (t) is CC ωθ1is the complex conjugate of (t), and t2-t1 is the time interval of each symbol. In other words, if the hold values ​​of the sample and hold units 224 and 225 are complex digital signals sampled and held at a time interval of N optical symbol time slots, the carrier offset component CC in equation (5) is ω It can be seen that (t) has information on the amount of phase change of the complex digital signal (corresponding to the carrier component of the regenerated optical symbol) that has changed over the course of time equivalent to N optical symbol time slots.

[0079] The sample and hold unit 222, the delay unit 223, the sample and hold units 224 and 225, and the multiplication unit 226 are connected to the carrier offset and phase components CC ωθ and activation pattern signal AP CS is input and the carrier offset component CC ω 11, the carrier offset component extraction unit 22B1 is written as “22B1: 222, 223, 224, 225, 226” to mean that the carrier offset component extraction unit 22B1 includes the sample & hold unit 222, the delay unit 223, the sample & hold units 224 and 225, and the multiplication unit 226.

[0080] The phase calculation unit 227 outputs Δω(t2-t1), which is the exponent part of the formula (5). That is, it outputs the value of the phase angle. Here, the phase angle value is limited to the range of -π≦Δω(t2-t1)≦π. Therefore, if the phase angle value suddenly jumps from a value near π to a value near -π, an error occurs. Therefore, the unwrap unit 228 performs signal processing so that the jump error of δ1=Δω(t2-t1) is minimized. For example, the algorithm of this operation may adopt Non-Patent Document 6. The phase rotation value φΔf, which is the output of this unwrap unit 228, is input to the selector unit 229. Note that the phase rotation value φΔf is a phase rotation value due to the offset frequency Δf, and is therefore called the phase rotation value φΔf due to the offset.

[0081] The delay unit 22A1 receives the phase rotation value φΔf due to the offset input to the selector unit 229 and the activation pattern signal AP CS In this way, the selector unit 229 adjusts the delay so that the timing of the activation pattern signal AP CS At the timing when this signal is ON, the phase rotation value φΔf due to the offset input from the unwrap unit 228 can be appropriately output.

[0082] On the other hand, the selector unit 229 selects the activation pattern signal AP CS When the activation pattern signal AP is off, the register section 22A0 selects the value 0 stored in the register section 22A0 (shown as "0" in FIG. 11) and outputs it. CS When the phase rotation value φΔf due to the offset is output, the activation pattern signal AP CS When the output of the adder 22A2 is OFF, a value of 0 can be stably output. These output values ​​are then input to the adder 22A2.

[0083] The adder 22A2 calculates the cumulative value of the amount of phase rotation φΔf due to the offset. In order to calculate the cumulative value of the phase rotation value φΔf due to the offset, the output value of the adder 22A2 is branched to a delay unit 22A3 and delayed by a time (Ts) equivalent to the width of one optical symbol time slot. Then, the output value is input to the adder 22A2 together with the phase rotation value φΔf due to the offset output from the selector unit 229. This makes it possible to calculate the cumulative value ΣφΔf of the phase rotation value φΔf due to the offset (also simply referred to as the "offset cumulative value ΣφΔf"). In other words, after n symbol time slots time t n The cumulative offset value ΣφΔf(t n ) is expressed as equation (6), where the amount of phase rotation at time t1 is θ1.

[0084]

number

[0085] This offset cumulative value ΣφΔf(t) is nothing but the sum of the phase rotation amounts (offset-induced phase rotation values ​​φΔf) caused by frequency offsets occurring at the same time. Therefore, this offset cumulative value ΣφΔf(t) is input to the exponent calculation unit 22A4.

[0086] The exponent calculation unit 22A4 calculates the output value of the addition unit 22A2, ΣφΔf(t n ) is input, and exp{-jΣφΔf(t n )} and input it to the multiplication unit 22A6. This complex value is used as the complex offset compensation amount C ΣφΔf It is called.

[0087] The phase calculation unit 227, the unwrap unit 228, the selector unit 229, the register unit 22A0, the adder unit 22A2, the delay unit 22A3, and the exponent calculation unit 22A4 calculate the carrier offset component CC ω to complex offset compensation amount C ΣφΔf 11, the complex offset compensation amount calculation unit 22B2 is described as "22B2: 227, 228, 229, 22A0, 22A2, 22A3, 22A4" to mean that the complex offset compensation amount calculation unit 22B2 includes the phase calculation unit 227, the unwrap unit 228, the selector unit 229, the register unit 22A0, the addition unit 22A2, the delay unit 22A3, and the exponent calculation unit 22A4.

[0088] The multiplication unit 22A6 multiplies the complex digital signal CS input to the frequency offset compensation unit 22 by the complex offset compensation amount C output from the exponential calculation unit 22A4. ΣφΔf This compensates for the frequency offset component Δf of the complex digital signal CS expressed by equation (2) input to the frequency offset compensator 22, and produces an output of a complex digital signal expressed by equation (7). This complex digital signal is a signal whose offset frequency has been compensated for, and the offset compensation signal CS f It is called.

[0089]

number

[0090] Here, the delay unit 22A5 adds a time delay (t1 to t n The delay is then adjusted to the same value as the time it takes for the

[0091] The output of the multiplication unit 22A6 is input to the sample and hold unit 22A8 via the delay unit 22A7. By setting an appropriate delay amount in the delay unit 22A7, the sample and hold unit 22A8 outputs the activation pattern signal AP CS When the offset frequency Δf is compensated for, the complex digital signal CS f The offset compensation signal CS output from the sample and hold unit 22A8 can be output. f is the overall output of the frequency offset compensator 22. Therefore, the frequency offset compensator 22 receives the complex digital signal CS and outputs an offset compensated signal CS in which the offset frequency Δf generated in the receiver RX is compensated for. f Output.

[0092] The effect of the overall operation of the frequency offset compensation unit 22 is shown in Fig. 12. In each photograph in Fig. 12, the horizontal axis corresponds to the real axis, and the vertical axis corresponds to the imaginary axis. This shows the effect of operating the frequency offset compensation unit 22 using a PN7 stage pseudorandom data pattern as a data modulation component. As shown in Fig. 12(a), the value of the complex digital signal CS before being input to the frequency offset compensation unit 22 exhibits a rotating behavior on the complex plane over time. On the other hand, after being subjected to compensation processing by the frequency offset compensation unit 22, the value of the complex digital signal CS becomes stationary at the output of the frequency offset compensation unit 22, as shown in Fig. 12(b). This output value, the offset compensation signal CS f is input to the phase compensation unit 23.

[0093] <Phase compensation unit 23> Next, the phase compensation unit 23 will be described. The phase compensation unit 23, which is arranged immediately after the frequency offset compensation unit 22 as shown in FIG. 9, is configured as shown in FIG. 13. First, the input complex digital signal (offset compensation signal CS f ) is branched into two. One of them is input to the power calculation unit 231. Here, the complex digital signal (offset compensation signal CS f ) and recovers the phase difference information between the carrier phase of the received optical symbol and the local light phase. The operation of this power calculation unit 231 may be the same as that of the power calculation unit 221 of the frequency offset compensation unit 22. The output of the power calculation unit 221 is the carrier phase component CC θ The carrier phase component CC θ If we write it explicitly, it becomes as equation (8).

[0094]

number

[0095] The averaging unit 232 calculates the complex digital signal (carrier phase component CC θ ) for L symbols for each of the real and imaginary parts θ The phase calculation unit 233 calculates the average value Av(CC θ ) to calculate the optical phase difference Δθ. Here, the optical phase difference Δθ is the optical phase difference between the carrier component of the optical symbol and the local light.

[0096] The optical phase difference Δθ is limited within the range of -π≦Δθ(t2-t1)≦π. Therefore, if the optical phase difference Δθ suddenly jumps from a value near π to near -π, an error occurs. Therefore, the unwrap unit 234 performs signal processing so that the jump error of δ1=Δθ(t2-t1) is minimized. The operation of this unwrap unit 234 realizes the same operation as the unwrap unit 228 of the frequency offset compensation unit 22. The optical phase difference Δθ, which is the output of this unwrap unit 234, is multiplied by 1 / M in the divider 235 to become a phase shift rotation value Φθ. Here, M is a value that depends on the modulation method of the optical symbol, and for example, in the case of QPSK, M=4.

[0097] The exponent calculation unit 236 receives the phase shift rotation value Φθ, which is the output value of the division unit 235, and outputs the complex value of exp(-jΦθ), which is input to the multiplication unit 238. This complex value is used as the complex phase shift compensation amount C Φθ The unwrap unit 234, the divider 235, and the exponent calculator 236 receive the optical phase difference Δθ and calculate the complex phase shift compensation amount C Φθ The above calculation may be called a complex phase shift compensation amount calculation unit 23B1.

[0098] The delay unit 237 delays the complex digital signal CS f (Offset compensation signal CS f The delay unit 237 receives the complex digital signal CS f (Offset compensation signal CS f ) is given a delay equal to the time it takes for it to pass through the exponent calculation unit 236 and be input to the multiplication unit 238, and then output to the multiplication unit 238.

[0099] The multiplier 238 multiplies the complex digital signal CS f (Offset compensation signal CS f ) is the complex phase shift compensation amount C output from the exponent calculation unit 236 ΦθMultiply. The complex digital signal output from the multiplication unit 238 is phase-compensated by this multiplication process, so that the phase angle of the data modulation component matches the phase angle of the defined signal point, and the optical phase difference Δθ between the carrier component of the optical symbol and the local light emission is compensated. Since the offset frequency and the phase difference of this signal are compensated, it is called the offset & phase compensation signal CS fθ This is the output from the multiplication unit 238, the offset & phase compensation signal CS fθ is used as the output of the entire phase compensation unit 23 and input to the APM Bit demapping unit 25. The offset & phase compensation signal CS fθ is expressed as in Equation (9).

[0100]

Equation

[0101] The above is the overall configuration of the phase compensation unit 23. As shown in FIG. 14, the value of the complex digital signal CS f (offset compensation signal CS f ) before being input to the phase compensation unit 23 is deviated from the position of the signal point defined on the complex plane. On the other hand, as a result of the compensation process in the phase compensation unit 23, the value of the complex digital signal (offset & phase compensation signal CS fθ ) at the output of the phase compensation unit 23 is near the defined signal point, as shown in FIG. 14. This output value is input to the APM Bit demapping unit 25.

[0102] <APM Bit demapping unit 25> Referring to FIG. 9 again, the Apm Bit demapping unit 25 will be described. The APM Bit demapping unit 25 operates in synchronization with the activation pattern signal AP CS . Here, the activation pattern signal AP CS is the complex digital signal (offset & phase compensation signal CS fθ detected at the timing when the optical symbol is ON) is input to the APMBi demapping unit 25 via a delay unit 24 so as to be synchronized with the activation pattern signal APMBi. CS The complex digital signal (offset & phase compensation signal CS fθ ) to digital code.

[0103] Complex digital signal (offset & phase compensation signal CS fθ ) into a digital code, there is a method of assigning Gray codes to signal points on the complex plane. For example, in the case of QPSK modulation, the codes "11", "01", "10", and "00" are assigned to the signal points 1+j, -1+j, 1-j, and -1-j on the complex plane.

[0104] The APMBit demapping unit 25 converts the input complex digital signal (offset & phase compensation signal CS fθ ), the likelihood is calculated for each signal point (Non-Patent Document 7), and the code of the signal point with the highest likelihood is selected and decoded. The output of the APMBit demapping unit 25 is AP Note that error correction processing may be performed on the downstream side of the APMBit demapping unit 25 up to the register unit 27.

[0105] In addition, the signal point constellation has a rotational symmetry of 90° in phase angle. Therefore, when decoding optical symbols, the interpretation of the I-axis and Q-axis of the complex plane may be reversed. To avoid this, the APM Bit demapping unit 25 has an IQ channel discrimination function.

[0106] To drive this IQ channel discrimination function, the transmitter periodically inserts a training optical symbol (complex value 1+j) to provide a phase reference for the received optical symbol. After discriminating the training optical symbol, the IQ channel discrimination function obtains the phase reference (complex value 1+j) of the optical symbol from this and correctly discriminates the phase rotation of the optical symbol, i.e., discriminates the IQ channel.

[0107] In addition, in the optical transmission system having the functions of the present invention, by adopting a differential encoding method, it is possible to omit the IQ channel discrimination function of the APMBit demapping unit 25. In the APMBit demapping unit 25 corresponding to the differential encoding method, the differentially encoded complex digital signal S N-1 and the differentially encoded complex digital signal S input at the timing of the Nth activation pattern input. N For example, when the phase difference is 0, it is decoded as 00, when it is π / 2, it is decoded as 01, when it is -π / 2, it is decoded as 10, when it is π, it is decoded as 11, etc.

[0108] <Register section 27> The register unit 27 receives the activation pattern signal AP CS With input, activation pattern signal AP CS When the decoded APMBit code string (symbol decoded code DE AP ) and holds it. Here, the delay unit 26 acquires and holds the activation pattern signal AP CS and the corresponding APMBit decoded symbol code DE decoded by the demapping unit 25 AP The delay time is set so that the arrival timings of the AMP Bit data (symbol decoded code DE AP ) is the code string of the IndexBit decoded by the index demodulation process unit 10 and stored in the register unit 16 (index decoded code DE Id ) and the signal is read out while being synchronized with the timing of the symbol demodulation process unit 20.

[0109] <Parallel / serial conversion section 29> The parallel / serial conversion unit 29 converts the index decoded code DE of the IndexBit decoded by the index demodulation process unit 10 and stored in the register unit 16 into Id and the symbol decoded code DE of APMBit decoded by the symbol demodulation process unit 20 and stored in the register unit 27. APand converts these into a serial code string SDAT, and decodes the transmission code string (original information INF) to be transmitted by the optical signal transmitter 1, resulting in a decoded original signal (decoded information DE(INF)). Here, the parallel / serial converter 29 inputs a trigger signal whose timing has been adjusted by the delay unit 28, and reads out the codes stored in the register units 16 and 27 based on this signal.

[0110] The above is the first embodiment of the present invention. This embodiment has the function of receiving optical symbols generated based on a time-domain (N, K, M-QAM) index optical modulation scheme, but is also characterized in that it is possible to receive optical symbols generated based on an M-QAM optical modulation scheme.

[0111] When only M-QAM optical symbols are received, the demapping (conversion from a complex digital signal to a code) in the IndexBit demapping unit 14 is stopped, but the activation pattern signal AP CS is continuously output to the frequency offset compensating unit 22. This allows the frequency offset compensating unit 22 to operate continuously in accordance with the arrival period of the optical symbols.

[0112] As a result, the frequency offset compensation operation can be realized even for optical symbols generated based on the M-QAM optical modulation method. In addition, the phase compensation unit 23 and the APM Bit demapping unit 25 of the present invention are originally designed to operate continuously at the period of the optical symbol time slot regardless of the time domain (N, K, M-QAM) index optical modulation method or the M-QAM optical modulation method, so they operate in common regardless of either method. However, for discrimination of the training optical symbol, the circuit is switched according to the operation mode of the time domain (N, K, M-QAM) index optical modulation method and the M-QAM optical modulation method. Note that this mode switching and the stop of demapping in the Index Bit demapping unit 14 may be received as separate signals.

[0113] The parallel / serial conversion unit 29 changes the readout patterns from the register units 16 and 27 provided in the previous stage according to the operation mode of the time domain (N, K, M-QAM) index optical modulation method and the M-QAM optical modulation method. For example, in the case of the time domain (8, 1, QPSK) index optical modulation method, one frame is composed of a time slot for eight optical symbols, and only one optical symbol is ON during one frame. In this case, there are eight types of symbol ON / OFF patterns, and 3 bits can be transmitted with the index bit.

[0114] On the other hand, APMBit is possible to transmit 2 bits because only one QPSK optically modulated optical symbol arrives. When the parallel / serial conversion unit 29 operates in the time domain (N, K, M-QAM) index optical modulation method, the activation pattern signal AP CS At the timing when this signal is input, a 3-bit code is read out from the register unit 16 and a 2-bit code is read out from the register unit 27.

[0115] On the other hand, when receiving QPSK optically modulated optical symbols, the parallel / serial conversion unit 29 converts the activation pattern signal AP CS At the timing when the signal is input, the code is read out two bits at a time from the register unit 27. By implementing such an operation, the operation mode switching between the time domain (N, K, M-QAM) index optical modulation method and the M-QAM optical modulation method is realized.

[0116] In this way, the optical signal transmitter 1 of the present invention has an advantage in that it can support both the time domain (N, K, M-QAM) index optical modulation method and the M-QAM optical modulation method with approximately the same circuit configuration.

[0117] Second embodiment In the second embodiment, tentative decision calculation units 221a (see FIG. 15) and 231a (see FIG. 17) described later are applied instead of the power calculation unit 221 (see FIG. 11) and the power calculation unit 231 (see FIG. 13) arranged in the frequency offset compensation unit 22 and the phase compensation unit 23 (see FIG. 9) arranged in the symbol demodulation process unit 20 shown in the first embodiment. The tentative decision calculation units 221a and 231a provide the same function as the power calculation unit 221 in the first embodiment in that they are applied as a means for removing the influence of the data modulation component from the complex digital signal CS corresponding to the received optical symbol.

[0118] The frequency offset compensation unit and the phase compensation unit having the tentative decision calculation unit 221a and the tentative decision calculation unit 231a are referred to as the frequency offset compensation unit 22a and the phase compensation unit 23a. The frequency offset compensation unit 22a and the phase compensation unit 23a perform the same operations as the frequency offset compensation unit 22 and the phase compensation unit 23. Therefore, the operation of the symbol demodulation process unit 20 including the frequency offset compensation unit 22a and the phase compensation unit 23a is also the same as in the first embodiment.

[0119] 15 shows the configuration of the frequency offset compensation unit 22a. The difference from the frequency offset compensation unit 22 in FIG. 11 is that the power calculation unit 221 is replaced with a provisional decision calculation unit 221a, and the complex offset compensation amount C ΣφΔf is the provisional decision calculation unit 2 This is the point entered in 21a.

[0120] 16 shows the configuration of the tentative decision calculation unit 221a incorporated in the frequency offset compensation unit 22a. First, a complex digital signal CS of the following equation (7) corresponding to an optical symbol is input to the tentative decision calculation unit 221a. Here, it is assumed that the optical symbol is QPSK optically modulated, and the optical angular frequency offset between the carrier component of the optical symbol and the local light is Δω, and the optical phase difference is Δθ. When the complex digital signal CS expressed by equation (10) is input to this tentative decision calculation unit 221a, the complex digital signal CS is branched into two, one of which is input to the multiplication unit 2211 and the other to the multiplication unit 2213.

[0121]

number

[0122] On the other hand, the exponential calculation unit 22A4 at the output side of the frequency offset compensation unit 22a outputs a signal for compensating for the optical angular frequency offset (complex offset compensation amount C ΣφΔf ) is fed back. However, this feedback signal is n time t earlier than m A complex digital signal CS(t m ), it is expressed as in equation (11).

[0123]

number

[0124] The complex offset compensation amount C in Eq. (11) ΣφΔf is input to the multiplier 2211 and multiplied by the complex digital signal CS of the equation (10). The result of the multiplication is the complex digital signal CS of the equation (12). np It becomes.

[0125]

number

[0126] This signal can be said to be a signal in which the continuous phase rotation of the complex vector caused by the frequency offset Δf is suppressed. This signal is called the suppressed complex digital signal CS np This signal is input to the provisional decision unit 2212. The provisional decision unit 2212 outputs the suppressed complex digital signal CS np Select the most likely signal point for the optical angular frequency offset and optical phase difference component ω(t n -t m ) + Δθ n -Δθ m The reciprocal value TCS of the provisional decision data modulation component expressed by equation (13) with -1 fθ and inputs it to the multiplication unit 2213.

[0127]

number

[0128] The multiplier 2213 multiplies the complex digital signal CS input to the frequency offset compensator 22a and expressed by equation (7) by the complex digital signal expressed by equation (10), thereby removing the data modulation component from the complex digital signal CS input to the frequency offset compensator 22a, and obtaining the carrier offset·phase component CC of the optical symbol expressed by the following equation (14): ωθ This signal is the carrier offset / phase component CC, which is the output of the frequency offset compensator 22 shown in the first embodiment. ωθ is the same as:

[0129]

number

[0130] In this way, the tentative judgment calculation unit 221a of this embodiment realizes the same operation as the power calculation unit 221 of the first embodiment.

[0131] Fig. 17 shows the configuration of the phase compensation unit 23a, and Fig. 18 shows an embodiment of the tentative decision calculation unit 231a newly provided in this embodiment. This embodiment realizes the same operation as the tentative decision calculation unit 221a provided in the frequency offset compensation unit 22a. However, the range in which the same operation is realized is related to the procedure from equation (13) to equation (14).

[0132] 18, the temporary decision unit 2311 receives the offset compensation signal CS f is input. Offset compensation signal CS f was calculated as in (7). Equation (7) is shown below again.

[0133]

number

[0134] The temporary decision unit 2311 outputs the complex digital signal CS f Select the most likely signal point for the optical angular frequency offset and optical phase difference component Δθ n -Δθ m The reciprocal value TCS of the tentative decision data modulation component expressed by equation (15) with -1 θ and inputs it to the multiplication unit 2312.

[0135]

number

[0136] Signal in equation (15) and offset compensation signal CS f By multiplying the carrier phase component CC expressed by equation (16), θ This is the same operation as that of the power calculation unit 231 in the phase compensation unit 23 in the first embodiment.

[0137]

number

[0138] The second embodiment is also compatible with a time-domain (N, K, M-QAM) index optical modulation scheme in which arbitrary signal points are defined, rather than a time-domain (N, K, M-QAM) index optical modulation scheme in which signal points arranged at equal phase intervals are defined.

[0139] The index demodulation process unit 10 and the symbol demodulation process unit 20 in the second embodiment have the same configuration as that shown in Fig. 9 in the first embodiment. Therefore, the same signals are input and output to and from the frequency offset compensation unit 22a and the phase compensation unit 23. Therefore, in Fig. 15 as well, the sample and hold unit 222, the delay unit 223, and the sample and hold units 224 and 225 detect the carrier offset and phase components CC ωθ and activation pattern signal AP CS is input and the carrier offset component CC ω The phase calculation unit 227, the unwrap unit 228, the selector unit 229, the register unit 22A0, the adder unit 22A2, the delay unit 22A3, and the exponent calculation unit 22A4 extract the carrier offset component CC. ω to complex offset compensation amount C ΣφΔf The complex offset compensation amount calculation unit 22B2 calculates the following.

[0140] (Third embodiment) 19 is a block diagram of a demodulator 6b according to a third embodiment of the present invention. This embodiment has almost the same configuration as the first embodiment, but the phase compensation unit 23 also detects the activation pattern signal AP CS The second embodiment differs from the first embodiment in that the phase compensation unit 23 in this embodiment is a phase compensation unit 23b. The phase compensation unit 23b receives an offset compensation signal CS f Here, the activation pattern signal AP CS is generated by the delay unit 30 (see FIG. 9) as the offset compensation signal CS f is adjusted so as to be synchronized with the timing at which it is input to the phase compensation unit 23b.

[0141] The phase compensation unit 23b of this embodiment is configured as shown in FIG. f ) is branched, and one signal passes through a tentative decision unit 231a, an averaging unit 232b, a phase calculation unit 233, an unwrapping unit 234, a division unit 235, and an exponential calculation unit 236 to produce an output signal (complex phase shift compensation amount C Φθ ) is extracted and sent to the multiplication unit 238. Note that although the input and output signals of the averaging unit 232b are the same as those of the averaging unit 232 in the first embodiment, the configuration is different. This will be described later.

[0142] The other signal passes through delay unit 237 and reaches multiplication unit 238. The signal processing during this period is the same as in the first embodiment. Note that power calculation unit 231 used in the first embodiment is replaced with temporary determination unit 231a. However, as explained in the second embodiment, power calculation unit 231 and temporary determination unit 231a can be replaced because the input signal and output signal are exactly the same. Of course, in FIG. 20, temporary determination unit 231a may be replaced with power calculation unit 231.

[0143] In addition to the configuration of the first embodiment, the phase compensation unit 23b in this embodiment further includes an averaging unit 232b that outputs an activation pattern signal AP CS is input. Furthermore, a sample and hold unit 23A0 is provided on the output side, and an activation pattern signal AP CS The phase-compensated offset and phase compensation signal CS arrives at the timing when fθ Here, the activation pattern signal AP CS The delay of the offset and phase compensation signal CS fθ and the corresponding activation pattern signal AP CS The timing of arrival of the activation pattern signal AP CS Therefore, the M-QAM demodulated signal in the time slot where

[0144] The averaging unit 232b applied to the phase compensation unit 23b of this embodiment is configured as shown in FIG. θ ) is the activation pattern signal AP CS Each time it is input, it is sequentially pushed into a vertical configuration of one symbol time delay unit 2321 and sample & hold unit 2322. As a result, complex digital signals of only the symbol time slots in which optical symbol power exists based on the time domain (N, K, QAM) index optical modulation rule are added in complex adder 2323 and divided in complex divider 2324, so that the average value of the complex digital signal can be calculated more accurately, and finally, more accurate carrier phase information can be estimated in phase calculation unit 233 (see FIG. 20) arranged immediately after averaging unit 232b.

[0145] (Fourth embodiment) 22 shows a fourth embodiment of the present invention. This embodiment has almost the same configuration as the third embodiment, but differs from the third embodiment in that the function of the phase compensation unit 23 is enhanced and a phase slip reduction function is added. The phase compensation unit 23 of this embodiment is called a phase slip reduction function-equipped phase compensation unit 23c.

[0146] The phase compensation unit 23c with phase slip reduction function of this embodiment is configured as shown in Fig. 23. The difference from the phase compensation unit 23b of the third embodiment is that the phase compensation unit 23c with phase slip reduction function further includes a phase slip compensation unit 310 in addition to the phase compensation unit 230 having the same configuration as the phase compensation unit 23b of the third embodiment. Here, the phase slip compensation unit 310 is basically the same as the configuration described in Patent Document 1.

[0147] A phase slip is a phenomenon in which the phase of a complex digital signal that has been frequency offset compensated and phase compensated from a received optical symbol transitions to a state rotated by ±90 degrees or 180 degrees from the original phase. In order to detect a phase slip, carrier phase information (phase shift rotation value Φθ) that is regenerated by the phase compensation unit 230 and output from the division unit 235 is used. In this embodiment, the carrier phase information is branched and input to the delay difference unit 312. Here, the delay difference unit 312 divides the activation pattern signal AP CS This activation pattern signal AP CS is adjusted by the delay unit 311 so as to be synchronized with the timing at which carrier phase information (phase shift rotation value Φθ) recovered from a complex digital signal corresponding to an optical symbol is input to the delay difference unit 312 .

[0148] The delay difference unit 312 calculates the difference in carrier phase information (phase shift rotation value Φθ) at a specific time difference. Here, the specific time difference is set to a time related to reception of optical symbols in a number of about 1.0 to 1.5 times the number of received optical symbols averaged in the averaging unit 232. The delay difference unit 312 detects the difference in carrier phase occurring at this set time. If no phase slip has occurred, the difference in carrier phase at the set time will be a small value. On the other hand, if a phase slip has occurred, this carrier phase difference will be large and will be a value of 1 rad or more and less than π / 2 rad. This carrier phase difference is output to the slip determination unit 313.

[0149] The slip determination unit 313 determines the rotation angle of the phase slip from the differential value of the carrier phase output from the delay difference unit 312. That is, it determines whether the phase slip is π / 2, π, or -π / 2 rad. Then, it outputs the determination result Res to the sample & hold unit 314. Moreover, the slip determination unit 313 issues a trigger signal Tr to the sample & hold unit 314 every time it determines that a new phase slip has occurred.

[0150] Here, the sample & hold unit 314 holds the negative value of the determination result Res output from the slip determination unit 313 until the next trigger signal Tr is input, and continues to output the held value to the exponent calculation unit 315. The exponent calculation unit 315 receives the input of the determination result Res and outputs exp(-j·Res) to the multiplication unit 317.

[0151] The multiplier 317 multiplies the offset and phase compensation signal CS fθ This offset and phase compensation signal CS fθ is input via a delay unit 316. This delay unit 316 is given a delay equivalent to the delay times related to the delay difference unit 312, the slip determination unit 313, the sample & hold unit 314, and the exponent calculation unit 315. As a result, the multiplication unit 317 outputs an enhanced offset & phase compensation signal FCS that has been phase slip compensated. fθ will be output.

[0152] In this way, the fourth embodiment of the present invention makes it possible to realize phase slip compensation even for time-domain (N, K, M-QAM) index optical modulated signals. This embodiment also has all the features realized in the first to third embodiments, and makes it possible to receive not only time-domain (M, K, M-QAM) index optical modulated signals but also M-QAM optical modulated signals using the same circuit.

[0153] Fifth embodiment 24 shows a fifth embodiment of the present invention. This embodiment has almost the same configuration as the fourth embodiment, but unlike the fourth embodiment, it realizes a function of correcting demapping errors in the index demodulation process unit 10 and reducing the bit error rate generated in the optical signal demodulation device 3 of the present invention. The correction of demapping errors in the index demodulation process unit 10 is realized by an IndexBit demapping unit 14d with a soft decision FEC (Forward Error Correction) function.

[0154] The IndexBit demapping unit 14d with soft decision FEC function is configured as shown in Fig. 25. The component receives an amplitude calculation value for a complex digital signal, and performs soft decision processing on the amplitude calculation value of a time domain (N, K, M-QAM) index optical modulated signal (soft decision unit 141) and demaps it to an IndexBit. Here, the trigger signal (head notification signal S TOP ) is used to determine the start timing of the N-symbol frame of the index optical modulated signal.

[0155] Then, the likelihood of N symbols is calculated, and the likelihood of whether the optical symbol in each symbol time slot is On or Off is calculated. Then, the N pieces of likelihood information are input to a multi-ary LDPC (Low Density Parity Check) code decoder 142 (Non-Patent Document 8). The multi-ary LDPC code decoder 142 correctly identifies the time slot in which the optical symbol exists, demaps it to an IndexBit, and generates an index decoding code DE Id At the same time, the index decoded code DE Id Based on this, an activation pattern signal AP corresponding to the time slot in which the optical symbol exists is generated. CS Generate.

[0156] In this way, the fifth embodiment can reduce not only the bit error rate of the IndexBit but also the generation error of the activation pattern, compared to the fourth embodiment. For example, by appropriately designing a non-ary LDPC code, the generation error rate of the activation pattern can be reduced to 4×10 -2 About 10 -9 can be significantly reduced to EXAMPLES

[0157] <Simulation> A simulation was performed to evaluate the performance of the demodulator 6 in FIG. 9. The simulator was designed to evaluate the performance of pre-FEC (Forward Error Correction) Bit Error Rate (BER) for K=1,2,3,4 in a subframe with N=8 symbol length, which is less than half the subframe length. This is because the Euclidean distance in the time domain (N,K,BPSK) is half that of BPSK. This means that the optical power of each activated symbol in the time domain (N,K,BPSK) needs to be more than twice that of each symbol in BPSK. For reference, the simulation was also performed for the time domain (16,1,BPSK) and BPSK.

[0158] In the simulation, the white Gaussian noise (AWGN), frequency offset, and phase noise in the simulated signal source and the local oscillator source were expressed as in Eq. (17).

[0159]

number

[0160] where x p (k) is the symbol at the time of transmission, n(k) (in equation (17) there is a caret above “n”) is the AWGN component, Δf is the frequency offset between the signal and the local oscillator, ΔT is the sampling time, and θ(k) (in equation (17) there is a caret above “θ”) is the phase noise component added by the laser light source.

[0161] In addition, in the simulation, it was assumed that the spectral linewidths of the signal light source and the local oscillator were the same, and the phase noise from each light source was zero-mean Gaussian noise with a variance obtained from Non-Patent Document 9. Specifically, this is given by Equation (18).

[0162]

number

[0163] Figure 26 shows the simulation results of BER (vertical axis) as a function of signal-to-noise ratio (SNR: horizontal axis). In order to emulate the experimental conditions as faithfully as possible, a root-raised cosine filter with a roll-off factor of 0.1 was incorporated as a low-pass filter.

[0164] In the simulation, we assume a symbol rate H of 25 GBaud, a frequency offset ΔA of 100 MHz, and a laser spectrum linewidth Δν of 100 kHz. The results show that the time-domain IM method extends the lower limit of the receivable SNR. For modulation parameters of N=8 and K=4, i.e., time-domain (8,4,BPSK), the data rate is 25% higher than BPSK, but the lower limit of the SNR is 0.2 dB worse than BPSK. For time-domain (8,3,BPSK), the lower limit of the SNR is 1.0 dB lower, but the line rate is the same as BPSK. For time-domain (8,2,BPSK), the lower limit of the SNR is significantly lowered by 2.5 dB, and the line rate is reduced from 1 bit / symbol to 6 / 8 bit / symbol.

[0165] This value extends to 6.5 dB in the time domain (8,1,BPSK) which achieves a 4 / 8 bit / symbol line rate, and to 8.5 dB in the time domain (16,1,BPSK) which achieves a 5 / 16 bit / symbol line rate. Compared to the time domain (8,1,BPSK), the extension of the lower SNR limit is only 2.0 dB in the time domain (16,1,BPSK).

[0166] Thus, time-domain (N,K,BPSK) systems can achieve more energy-efficient operation by lowering the line rate from the basic line rate achieved by conventional BPSK systems. These enhancements stem from the higher peak power of the time-domain (N,K,BPSK) optical signals. These higher peak powers also contribute to the reduction of cycle slips that appear in the low SNR region of conventional BPSK optical signals.

[0167] It is also important to note that these enhancements are achieved without applying inner FEC to the Index-ch. Time-domain (N, K, BPSK) systems do not require heavy circuit implementation requirements such as iterative frequency or phase recovery techniques with feed-forward error correction during low-power operation.

[0168] <Transmission experiment> The demodulator 6 shown in FIG. 9 was created. The baud rate here refers to a symbol rate that includes a "null symbol" that forms an activation pattern and does not transmit APM bit data. 7 A data stream (1) was generated and demultiplexed to Index-ch and APM-ch. In this experiment, the data stream was separated into 4-bit data parts. The separated data was demultiplexed into 3-bit data parts for Index-ch and 1-bit data parts for APM-ch.

[0169] The first part of the 3-bit data part of the Index-ch was mapped to an activation pattern of 8 symbol time slot length such as (01000000). Meanwhile, the second part of the 1-bit part of the APM-ch was differentially precoded such as (1) and (-1). The complex symbols were generated simply by multiplying the activation pattern and the precoded complex symbols. The actual electrical signal was generated using a 1CH arbitrary wave generator with positive and negative output ports.

[0170] Polarization multiplexing was emulated by a delay line with a time slot length of two symbols. These two electrical signals were input to a polarization multiplexing LiNbO3-based optical I / Q modulator. Due to the limited analog bandwidth of the ADC, a low symbol rate of 2.5 GBaud was considered. An external cavity laser source was adopted to reduce frequency drift, and its spectral linewidth was on the order of 10-100 kHz.

[0171] The receiver consists of an optical preamplifier with an optical bandpass filter. The received optical signal is amplified to -17 dBm and input to a polarization and phase diversity receiver. To minimize the optical frequency drift, an external cavity laser source is adopted as the local oscillator.

[0172] The frequency offset from the optical signal was controlled to about 15 MHz by controlling the temperature of the laser source. The output power was 5 dBm. The four output signals from the polarization and phase diversity receiver were sampled by an ADC operating at 5 Gsamples / s using low-pass filters with analog bandwidth, steepness, and stopband attenuation of 2.0 GHz, 0.85, and 60 dB, respectively. Thus, the noise bandwidth is almost twice the ideal condition for 2.5 Gbaud symbol detection. Therefore, in this experiment, the implementation of a dispersion compensator and a polarization compensator was omitted. However, it is expected that there will be no major problems if a conventional dispersion compensator is applied.

[0173] Figure 27 shows a typical constellation diagram when a 2.5 Gbps time-domain (8,1,BPSK) optical signal is polarization-multiplexed. As shown in Figure 27(a), the constellation appears near zero, which is the same as that obtained by permutation modulation in Non-Patent Document 10. Unlike permutation modulation, the TDSC-IM method uses multi-stage decoding to decode the optical signal in a set-partitioning manner. As shown in Figure 27(b), we successfully removed the constellation near the zero point and recovered the carrier frequency and phase.

[0174] Figure 28(a) shows the BER of a 2.5 GBaud polarization multiplexed time-domain (8,1,BPSK) optical signal as a function of received optical power. Measurements were performed in both back-to-back and 20 km transmissions using single-mode fiber. We also measured the back-to-back BER of the 2.5 GBaud polarization multiplexed BPSK optical signal. The bit error rate increased sharply when the optical signal power was below -47 dBm.

[0175] This corresponds to a penalty of 2 dB from the theoretical shot noise limit, given the noise bandwidth of the experimental set. -3 The received optical power obtained was -49 dBm when a 2.5 Gbaud BPSK optical signal was polarization-multiplexed, while it was -55 dBm when a 2.5 Gbaud polarization-multiplexed time-domain (8,1,BPSK) optical signal was used.

[0176] In this way, we confirmed that the receiver sensitivity improved by 6 dB while lowering the data transmission rate to half that of the BPSK optical signal. This 6 dB improvement in receiver sensitivity is consistent with the results shown in Figure 26, where the lower limit of the SNR is extended to 6.5 dB in the time domain (8,1,BPSK).

[0177] This is E b This corresponds to a 3 dB improvement in the / N0 ratio. This gain comes from the use of an activation pattern to transmit information data. Next, the performance of the Delayed BPSK receiver was evaluated, since this detection scheme can reduce the complexity of the CFR and CPR modules for correcting the carrier frequency and phase deviation of the receiver's local oscillator. Figure 28(b) shows the BER of a 2.5 GBaud polarization multiplexed TDSC (8,1,DBPSK) IM and DBPSK optical signal as a function of received optical power.

[0178] The sensitivity of the DBPSK optical signal was degraded by nearly 3 dB compared to the BPSK optical signal. On the other hand, no degradation was observed in the time-domain (8,1,DBPSK) optical signal. This indicates that bit errors caused by the index bit of the time-domain (8,1,DBPSK) optical signal are avoided by using the APM bit detection method. This result also indicates that there exists a difference in the minimum value of receiver sensitivity between the Index-ch and the APM-ch.

[0179] <Tolerance to frequency offset> Although the time-domain (8,1,BPSK) scheme achieves good performance in terms of receiver sensitivity, it is inherently vulnerable to frequency offset between the carrier frequency and the local oscillator frequency. This is because the time-domain (N,K,BPSK) receiver must recover the carrier frequency and phase from the existing K time slots for each N time slot long subframe with the help of the activation pattern signal. This increases the time interval and directly affects the tolerance to phase changes between two consecutive activation optical symbols.

[0180] Figure 29 shows the BER of the time-domain (8,1,BPSK) optical signal and the BPSK optical signal as a function of the frequency offset normalized by the symbol rate, both measured and simulated. The horizontal axis is the normalized offset frequency, and the vertical axis is the error rate. It was confirmed that the upper limit of the frequency offset of the 2.5 GBaud time-domain (8,1,BPSK) optical signal with a frequency offset of -50 dBm is only 17 MHz, while the upper limit of the frequency offset of the 2.5 GBaud BPSK optical signal with a frequency offset of -43 dBm reaches 800 MHz. Since the longest time interval of the activated optical signal spans 15 symbol time slots, the upper limit of the time-domain (8,1,BPSK) optical signal was expected to be about 53 MHz, one-fifteenth of that of the BPSK optical signal. However, this result indicates that the tolerance of the frequency offset between the carrier frequency and the receiving local oscillator is small.

[0181] As described above, we have successfully demodulated a 2.5 GBaud polarization-multiplexed time-domain (8,1,BPSK) optical signal over a 20 km single-mode fiber. Although the data rate was reduced to half that of the BPSK optical signal, the receiving sensitivity of the time-domain (8,1,BPSK) optical signal was improved by nearly 6 dB.

[0182] This is E bThis represents a 3 dB improvement in the / N0 ratio, providing connectivity over 20 km or more of transmission distance over existing systems. Conversely, reducing the optical power of the local oscillator reduces the energy consumption of each optical network unit (ONU) in a system without preamplifiers. These benefits are sufficient to address the issue of low tolerance to frequency offsets in the demodulator by adding a frequency discriminator to the CFR module.

[0183] Even with this constraint, time-domain (N,K,M-QAM) schemes can improve the performance of carrier phase recovery without relying on further heavy circuit implementation requirements such as iterative frequency and phase recovery with feed-forward error correction, without enhancing the redundancy of the error correction code or the circuit size for the error correction code.

[0184] We see three main use cases for time-domain (N,K,MQAM) schemes in optical access networks. In the first use case, time-domain (N,K,M-QAM) schemes are applied on links with low traffic, where the links can operate in energy-saving mode. In the second use case, time-domain (N,K,MQAM) schemes are applied to serve remote users who require more economical connections than bandwidth.

[0185] The link is operated in a higher energy mode to obtain higher optical peak power. Time-domain (N,K,M-QAM) schemes could address this requirement by developing a common platform with conventional M-QAM schemes, at a moderate sacrifice of channel capacity. This could solve the digital divide problem faced by remote rural areas. Automatic combination of these two operating modes could also pave the way for the development of elastic link capacity services that adapt to the conditions of the fiber optic access link, like conventional asymmetric digital subscriber line (ADSL)-based copper access services.

[0186] Time-domain (N,K,M-QAM) schemes are also expected to be beneficial in improving receiver sensitivity in higher order M-QAM-based systems that can support dual mode index modulation schemes such as time-domain (N,K,QPSK / 8-PSK). These schemes are expected to improve receiver sensitivity by about 1 dB compared to 8-PSK while maintaining the same channel capacity as 8-PSK. [Industrial Applicability]

[0187] This invention is applied to optical access networks, which are essential social infrastructure systems in an advanced information society. In conventional optical access networks, the optical loss budget between the OLT and the ONU is 29 dB, and the transmission distance is 20 km. If these conditions cannot be met, the communication line cannot be maintained, and the 10 Gbit / s transmission rate is completely cut off.

[0188] The present invention realizes a time domain index optical modulation method, and makes it possible to vary the transmission rate by thinning out optical symbols in accordance with changes in the transmission line conditions between the OLT and the ONU. In other words, even under conditions where the transmission line conditions are poor and exceed the optical loss budget of 29 dB, if the additional optical loss budget is up to about 8 dB, it is possible to avoid a complete disconnection of the communication line by gradually reducing the transmission rate. In other words, even for ONUs deployed in locations farther away from the OLT, the connectivity of the communication line is ensured, although the transmission rate is reduced, and this contributes to the expansion of the area where optical access network services are provided, and ultimately to the elimination of the digital divide. [Explanation of symbols]

[0189] 1 Optical signal transmitter 2 Transmission Path 3. Optical signal demodulator 6 Demodulation section 10 Index demodulation process 11 Amplitude calculation section 12 Frame timing extraction section 13 Delay section 14 IndexBit Demapping 16 Register section 20 Symbol demodulation process section 21 Delay section 22 Frequency offset compensation section 221 Power calculation unit 222 Sample and hold section 224 Sample and hold section 225 Sample and Hold Section 223 Delay Section 226 Multiplication section 227 Phase calculation section 228 Unwrap 229 Selector Section 22A0 Register section 22A2 Addition section 22A3 Delay section 22A4 Exponential calculation section 22A6 Multiplication section 22A5 Delay section 22A7 Delay section 22A8 Sample and hold section 23 Phase compensation section 231 Power calculation unit 232 Averaging section 233 Phase calculation section 234 Unwrap Section 235 Division part 236 Exponential calculation section 238 Multiplication section 237 Delay Section 22B2 Complex offset compensation amount calculation unit 24 Delay section 25 APMBit Demap Section 26 Delay section 27 Register section 28 Delay section 29 Parallel / serial conversion section 221a Provisional decision calculation unit 231a Provisional decision calculation unit 22a Frequency offset compensation section 23a Phase compensation section 2211 Multiplication section 2213 Multiplication section 2212 Provisional Judgment Section 2311 Provisional Judgment Department 2312 Multiplication section 23b Phase compensation section 30 Delay section 231a Provisional Judgment Section 232b Averaging section 234 Unwrap Section 23A0 Sample and hold section 239 Delay Section 2321 Delay Department 2322 Sample and hold section 2323 Complex Addition Unit 2324 Complex division unit 23B1 Complex phase shift compensation amount calculation unit 23c Phase compensation unit with phase slip reduction function 230 Phase compensation section 310 Phase slip compensation section 312 Delay difference part 311 Delay Division 313 Slip Judgment Section 314 Sample and hold section 315 Exponential calculation section 317 Multiplication section 316 Delay Division 14d IndexBit Demapping Unit with Soft Decision FEC (Forward Error Correction) Function 14d IndexBit Demapping Unit with Soft Decision FEC Function 141 Soft decision section 142 Multidimensional LDPC (Low Density Parity Check) code / decoder RX Receiver RXa Phase Diversity Detector LS optical signal Lo Local light AP CS Activation Pattern Signal DE(INF) Original information INF Decryption information Δf (offset frequency Δf) Optical frequency difference CS Complex digital signal DEId Index Decoding Code DE AP Symbol Decoding Code TIN Branch P CS Symbol Intensity S TOP Leading signal Δω Optical angular frequency offset Δθ Optical phase difference CC ωθ Carrier offset and phase components CC ω Carrier offset component 22B1 Carrier offset component extraction unit φΔf Phase rotation value φΔf Phase rotation due to offset ΣφΔf offset cumulative value C ΣφΔf Complex offset compensation amount CS f Offset Compensation Signal CC θ Carrier Phase Component Av(CC θ ) Average Φθ Phase shift rotation value C Φθ Complex phase shift compensation amount CS fθ Offset & Phase Compensation Signal SDAT Serial code string CS np Suppression Complex Digital Signal TCS -1 fθ Reciprocal value of provisional decision data modulation component FCS fθ Enhanced offset & phase compensated signal

Claims

1. An optical signal demodulation device that receives an optical signal transmitting an original signal using a symbol carried on at least one of the time slots and an index code formed at a time position of the time slot carrying the symbol, the optical signal demodulation device comprising: a receiving unit (RX) that performs phase diversity detection on the optical signal and converts it into a complex digital signal (CS); The complex digital signal (CS) is input, and an index decoding code (DE) is generated based on the position of the symbol in the time slot within the frame. Ｉｄ an index demodulation process unit (10) for demodulating the The complex digital signal (CS) is input, and the symbols carried in the time slots are converted into a symbol decoding code (DE ＡＰ ) a symbol demodulation process unit (20) for demodulating the When the index demodulation process unit (10) receives the time slot carrying the symbol, it transmits a notification signal (activation pattern signal AP) indicating the presence of the symbol to the symbol demodulation process unit (20). ＣＳ ) and The symbol demodulation process unit (20) receives the notification signal (activation pattern signal AP ＣＳ ) and obtains and decodes the symbols.

2. The index demodulation process unit (10) The symbol intensity (P ＣＳ ) an amplitude calculation unit (11) for calculating The symbol intensity (P ＣＳ ) is input, and a head notification signal (S ＴＯＰ A frame timing extraction unit (12) that outputs The symbol intensity (P ＣＳ ) and the head notification signal (S ＴＯＰ ) is input, and the notification signal (activation pattern signal AP ＣＳ ), and the index decoded code (DE Ｉｄ 2. The optical signal demodulation device according to claim 1, further comprising an IndexBit demapping unit (14) for outputting an IndexBit signal.

3. The IndexBit demapping unit (14) decodes the index decoded code (DE Ｉｄ 3. The optical signal demodulation device according to claim 2, wherein error correction is performed when decoding the optical signal.

4. The symbol demodulation process unit (20) The complex digital signal (CS) is input and a frequency offset generated in the receiving unit (RX) is compensated for to generate an offset compensated signal (CS ｆ A frequency offset compensator (22) that outputs a The offset compensation signal (CS ｆ ) is input and the phase shift generated in the receiving section (RX) is compensated for to produce an offset & phase compensation signal (CS ｆθ ), and a phase compensation unit (23) that outputs The offset & phase compensation signal (CS ｆθ ) is input, and a symbol decoded code (DE ＡＰ 2. The optical signal demodulation device according to claim 1, further comprising an APMBit demapping unit (25) for outputting an APMBit signal.

5. The frequency offset compensator (22) The complex digital signal (CS) is input, the data modulation component is removed, and the carrier offset and phase components (CC ωθ a power calculation unit (221) that outputs The consecutive symbols and the notification signal (activation pattern signal AP ＣＳ ) is the carrier offset component (CC ω A carrier offset component extraction unit (22B1) for extracting The carrier offset component (CC ω ) to the complex offset compensation amount (C ΣφΔｆ a complex offset compensation amount calculation unit (22B2) for calculating The complex digital signal (CS) and the complex offset compensation amount (C ΣφΔｆ ) and multiplying the offset compensation signal (CS ｆ 5. The optical signal demodulation device according to claim 4, further comprising a multiplication section (22A6) for calculating the signal.

6. 6. An optical signal demodulation device according to claim 5, further comprising a tentative decision calculation section (221a) instead of said power calculation section (221).

7. The phase compensation unit (23) The offset compensation signal (CS ｆ ) is input, the data modulation component is removed, and the carrier phase component (CC θ a power calculation unit (231) that outputs The carrier phase component (CC θ ) is averaged to obtain the average value of the carrier phase component (AV(CC θ ) an averaging unit (232) for calculating The average value of the carrier phase component (AV(CC θ a phase calculation unit (233) for extracting an optical phase difference (Δθ) from the The amount of complex phase shift compensation (C Φθ A complex phase shift compensation amount calculation unit (23B1) that calculates The offset compensation signal (CS ｆ ) and the complex phase shift compensation amount (C Φθ ) and multiply it by The offset & phase compensation signal (CS ｆθ 5. The optical signal demodulation device according to claim 4, further comprising a multiplication unit (238) for calculating the signal.

8. 8. An optical signal demodulation device according to claim 7, further comprising a tentative decision calculation section (231a) instead of said power calculation section (231).

9. 8. The optical signal demodulation device according to claim 7, further comprising a phase slip compensation section (310) for compensating for a phase slip provided immediately after the phase compensation section (23).

10. 1. An optical signal demodulation method, comprising: receiving an optical signal transmitting an original signal using a symbol carried on at least one of the time slots and an index code formed at a time position of the time slot carrying the symbol, the optical signal demodulating the original signal comprising: a receiving step (RX) for detecting the optical signal by phase diversity and converting it into a complex digital signal (CS); The complex digital signal (CS) is input, and an index decoding code (DE) is generated based on the position of the symbol in the time slot within the one frame. Ｉｄ (10) an index demodulation step of demodulating the When the time slot is received, a notification signal (activation pattern signal AP ＣＳ ) and The complex digital signal (CS) is input, and the notification signal (activation pattern signal AP ＣＳ The symbol carried in the time slot is decoded based on a symbol decoding code (DE ＡＰ ) into a symbol demodulation step (20).

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