Radar signal processing circuit and radar signal processing method

Abstract

To provide a radar signal processing circuit capable of suppressing degradation of radar detection performance caused by circuit errors.SOLUTION: The radar signal processing circuit includes: a transmit phase rotation unit adds a phase rotation amount that is obtained by dividing an integer multiple of 2π by Ne (Ne is an integer greater than 1) multiple times for each transmission period to each of the multiple radar signals; a radar-transmitting radio unit that multiplexes and transmits multiple radar signals with phase rotation amounts added to each transmission period via multiple transmitting antennas; and a radar radio receiving section that receives one or more echo signals reflected at the target, which are at least one radar signal among the multiple radar signals multiplexed and transmitted via multiple receiving antennas. In the phase rotation amount, the sign polarity of the phase rotation amount is controlled multiple times for each period corresponding to the transmission period of Ne.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present disclosure relates to a radar signal processing circuit and a radar signal processing method. [Background technology]

[0002] In recent years, radar devices using short-wavelength radar transmission signals, including microwaves and millimeter waves, which provide high resolution, have been under study. In addition, to improve outdoor safety, there is a demand for the development of radar devices (wide-angle radar devices) that can detect objects (targets), including pedestrians, in a wide-angle range in addition to vehicles.

[0003] For example, a pulse compression radar device is known as a radar device. When detecting a vehicle / pedestrian in a pulse compression radar device, the reflected wave from the pedestrian is weaker than the reflected wave from the vehicle. For this reason, the radar transmitter is required to have a transmission configuration that transmits a pulse compression wave with a low range side lobe, and the radar receiver is required to have a reception configuration with a wide reception dynamic range.

[0004] As a pulse compression code for obtaining low range side lobe characteristics, for example, a Barker code, a PN sequence code, a complementary code, or the like is known to be used. In the following, as an example, a case where a complementary code is used will be described. A complementary code is a code that constitutes a pair (hereinafter, a complementary code a n , b n Here, n=1,...,L, where L is the code length. The autocorrelation calculations of the two codes are expressed by the following equations (1) and (2).

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[0005] In equations (1) and (2), when n>L and n<1, n =0, b n= 0. Complementary codes have the property that by adding the autocorrelation calculation results of two codes with the same shift time τ, the correlation value becomes zero except when τ = 0, as shown in the following equation (3), and the range side lobes become zero.

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[0006] Such a complementary code a n , b n A pulse compression radar is known that transmits the above-mentioned signals in a time-division manner at a predetermined radar transmission period.

[0007] A method for generating complementary codes is disclosed in Non-Patent Document 1. According to Non-Patent Document 1, for example, a code A=[a 1, a2]=

[0011] , B=[b 1, b2]=[1 -1], code lengths L=4, 8, 16, 32, …, 2 P The pulse compression radar can increase the received signal level of the radar reflection wave by repeatedly transmitting the above-mentioned pulse compression code a predetermined number of times for each radar transmission period. [Prior art documents] [Patent documents]

[0008] [Patent Document 1] Japanese Patent Application Publication No. 2-243022 [Patent Document 2] JP 2000-338226 A [Non-patent literature]

[0009] [Non-Patent Document 1] Budisin, SZ, "New complementary pairs of sequences," Electron. Lett., 1990, 26, (13), pp.881-883 [Non-Patent Document 2] Egashira et al., "IQ Imbalance Compensation Method Using Pilot Signals in OFDM Systems," IEICE Transactions on Information and Communication Engineers Vol. J91-B No. 5 pp. 558-565, 2008 [Non-Patent Document 3] E. Spano and O. Ghebrebrhan, "Sequences of complementary codes for the optimum decoding of truncated ranges and high sidelobe suppression factors for ST / MST radar systems," IEEE Transactions on Geoscience and Remote Sensing, Vol.34, No.2, pp.330-345,1996 Summary of the Invention [Problem to be solved by the invention]

[0010] When attempting to realize the above-mentioned pulse compression radar using radio frequency (RF) bands such as millimeter waves, circuit errors are introduced into the RF circuits and analog baseband circuits in the radar transmitter or radar receiver of the radar device, resulting in a deterioration of radar detection performance (or radar ranging performance) compared to ideal characteristics without circuit errors.

[0011] One aspect of the present disclosure provides a radar signal processing circuit and a radar signal processing method that can suppress deterioration of radar detection performance due to circuit errors. [Means for solving the problem]

[0012] A radar signal processing circuit according to one embodiment of the present disclosure includes a transmission phase rotation unit that imparts to each of a plurality of radar signals a phase rotation amount calculated by dividing an integer multiple of 2π by Ne (Ne is an integer greater than 1) multiple times for each transmission period, a radar radio transmission unit that multiplexes the plurality of radar signals to which the phase rotation amount has been imparted for each transmission period via multiple transmission antennas, and a radar radio reception unit that receives, via multiple receiving antennas, one or more reflected wave signals of at least one radar signal among the multiplexed radar signals that has been reflected by a target, and a sign polarity of the phase rotation amount is controlled for each period equivalent to the multiple transmission periods of Ne.

[0013] A radar signal processing method according to one embodiment of the present disclosure includes the steps of: imparting a phase rotation to each of a plurality of radar signals, the phase rotation being calculated by dividing an integer multiple of 2π by Ne multiple times; multiplexing the plurality of radar signals to which the phase rotation has been imparted for each transmission period via a plurality of transmitting antennas; and receiving, via a plurality of receiving antennas, one or more reflected wave signals of at least one of the multiplexed radar signals that has been reflected by a target, wherein a sign polarity of the amount of phase rotation is controlled for each period equivalent to Ne multiple transmission periods.

[0014] These comprehensive or specific aspects may be realized as a system, a method, an integrated circuit, a computer program, or a recording medium, or may be realized as any combination of a system, an apparatus, a method, an integrated circuit, a computer program, and a recording medium. Effect of the Invention

[0015] According to one aspect of the present disclosure, it is possible to suppress deterioration of radar detection performance due to circuit errors.

[0016] Further advantages and benefits of certain aspects of the present disclosure will become apparent from the specification and drawings, in which such advantages and / or benefits are provided by some of the embodiments and features described in the specification and drawings, respectively, but not necessarily all of which are provided to obtain one or more identical features. [Brief description of the drawings]

[0017] [Figure 1] FIG. 1 is a block diagram showing a configuration example of a radar device according to a first embodiment of the present disclosure. [Diagram 2] FIG. 1 is a diagram showing an example of a radar transmission signal according to a first embodiment of the present disclosure. [Diagram 3] FIG. 11 is a block diagram showing another configuration of the radar transmission signal generating unit according to the first embodiment of the present disclosure. [Figure 4A] FIG. 13 is a diagram showing an example of a computer simulation result of the output of the Doppler analysis unit when phase inversion is not performed. [Figure 4B] FIG. 1 is a diagram showing an example of a computer simulation result of an output of a Doppler analysis unit according to the first embodiment of the present disclosure. [Diagram 5] FIG. 1 is a diagram showing another configuration example of a radar device according to a first embodiment of the present disclosure. [Figure 6] FIG. 11 is a block diagram showing a configuration example of a radar device according to a second embodiment of the present disclosure. [Figure 7] FIG. 13 is a diagram showing an example of a computer simulation result according to the second embodiment of the present disclosure. [Figure 8] FIG. 11 is a block diagram showing a configuration example of a radar device according to a third embodiment of the present disclosure. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0018] In a radar device, circuit errors include IQ mismatch and DC offset in the quadrature modulation circuit / quadrature demodulation circuit, phase noise in the frequency conversion section, and quantization noise in the AD converter and DA converter.

[0019] When a DC offset is included as a circuit error, the noise level increases compared to when there is no error. Therefore, in a radar device, if the reflected wave from a target is lower than the noise level, the target goes undetected, leading to a decrease in the detection rate and a deterioration in radar detection performance.

[0020] In response to this problem, a method has been proposed for removing DC offset by using a circuit configuration in a radar transmitter or a radar receiver. Specifically, Patent Document 1 discloses a configuration for removing DC offset by arranging a high-pass filter for discrete data output from an AD converter. Patent Document 2 discloses a configuration for removing DC offset by arranging a band-pass filter in the input stage of an AD converter. Non-Patent Document 2 discloses a circuit configuration for correcting IQ imbalance circuit errors.

[0021] However, in the conventional technology, it is necessary to provide a DC offset removal circuit or an IQ imbalance circuit error correction circuit, which complicates the circuit configuration. In addition, when a high-pass filter or a band-pass filter is provided to remove the DC offset, the desired radar reflected wave components other than the DC offset components may be weakened, and amplitude or phase distortion may occur due to the filter response, degrading the radar detection performance.

[0022] Furthermore, even in a configuration with a DC offset removal circuit, if the DC offset component or IQ imbalance component cannot be completely removed and circuit error components remain, the circuit error components are also accumulated by the coherent integration process in the radar reception process, resulting in a deterioration of radar detection performance. For example, if even a small residual component of the circuit error is included, the residual component increases to about 30 to 40 dB due to the accumulation effect, requiring a high-precision error detection mechanism, which complicates the hardware configuration of the radar device.

[0023] On the other hand, Non-Patent Document 3 discloses a radar device that transmits a code that combines phase modulation between multiple pulse transmission periods and cancels the received DC offset component by performing coherent integration processing on the reflected wave.

[0024] As an example, a code A=[a1, a2, … , a L ] and the code A is inverted by 180 degrees in phase with the code -A=[-a1, -a2, … , -a L The principle by which the received DC offset component is cancelled by transmitting codes A and -A in two transmission periods using [A, -A], performing correlation processing on the receiving side, and coherent integration is performed is shown below.

[0025] In the following, a case will be described in which there is no noise component and the sum of each element of code A is not zero as shown in the following equation (4).

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[0026] <Example 1: When the received signal of the radar reflection wave contains a received DC offset component> (1-1) When transmitting code A, the received DC offset component α Rx The received signal of code A containing (γA+α Rx ) and code A, the autocorrelation value obtained by the autocorrelation calculation is expressed by the following equation (5).

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[0027] In equation (5), γ represents the complex reception response of the radar reflection wave, and the asterisk (*) is the complex conjugate operator.

[0028] (1-2) When transmitting code-A, the received DC offset component α Rx The received signal of code-A containing (-γA+α Rx The autocorrelation value obtained by calculating the autocorrelation between the code -A and the code -B is expressed by the following equation (6).

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[0029] On the receiving side, when the autocorrelation value (equation (5)) obtained only by (1-1) above is subjected to coherent integration, the received DC offset component α Rx This causes the noise level (floor level) to rise over the entire distance range, degrading radar detection performance.

[0030] On the other hand, when the autocorrelation values ​​(Equations (5) and (6)) obtained in (1-1) and (1-2) above are subjected to coherent integration on the receiving side, the received DC offset component α Rx can be cancelled.

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[0031] This prevents an increase in the noise level and suppresses the deterioration of the radar detection performance of the radar device. In addition, the received DC offset component α Rx This makes it possible to prevent an increase in the noise level (floor level) and suppress deterioration of radar detection performance.

[0032] Example 2: The received signal of the radar reflection wave does not include Doppler fluctuation, and the transmitted DC offset component α Tx If it contains >

[0033] (2-1) When transmitting code A, the transmission DC offset component α Tx The received signal (γA+α Tx ) and code A, the autocorrelation value obtained by the autocorrelation calculation is expressed by the following equation (8).

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[0034] (2-2) When transmitting code-A, the transmission DC offset component α Tx The received signal of code-A containing (-γA+α Tx The autocorrelation value obtained by calculating the autocorrelation between the code -A and the code -B is expressed by the following equation (9).

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[0035] At the receiving side, the autocorrelation values ​​(equations (8) and (9)) obtained in (2-1) and (2-2) above are subjected to coherent integration to obtain the received DC offset component α Rx Similarly, when the transmit DC offset component α Tx is cancelled, preventing an increase in the noise level (floor level) and suppressing deterioration of radar detection performance.

[0036] As explained in <Example 1> and <Example 2>, by canceling the transmission DC offset component or the reception DC offset component, it is possible to suppress the deterioration of radar detection performance.

[0037] However, even with the above method, if a transmission DC offset component (including a carrier leak component) exists in the radar transmitter and there is Doppler fluctuation in the radar reflected wave, the transmission DC offset component remains. This is because the transmission DC offset component is subject to Doppler fluctuation, and a cancellation error occurs in the above method. This causes an issue that the noise level of the Doppler component included in the remaining transmission DC offset component increases, degrading radar detection performance.

[0038] The following describes the case where the radar reflection wave contains Doppler fluctuation.

[0039] Example 3: A transmitted DC offset component α is added to the received radar reflected wave signal containing Doppler fluctuations. Tx If it contains >

[0040] In this case, the Doppler fluctuation contained in the radar reflection is expressed as "exp(j2πf d ×T r )=exp(jΨ d )" (f d : Doppler frequency, T r : Radar transmission period. However, this is under the condition that the Doppler fluctuation within the code can be considered constant.

[0041] (3-1) When transmitting code A, the transmission DC offset component α Tx The received signal γ(A+α Tx The autocorrelation value obtained by calculating the autocorrelation between the code A and the code .DELTA..times ...

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[0042] (3-2) When transmitting code-A, the transmission DC offset component α Tx The received signal γ(-A+α Tx )exp(jΨ d The autocorrelation value obtained by calculating the autocorrelation between the code -A and the code -B is expressed by the following equation (11).

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[0043] At the receiving side, when the autocorrelation values ​​(equations (10) and (11)) obtained in (3-1) and (3-2) above are subjected to coherent integration processing, the transmission DC offset component α Tx Two items including the above will be accumulated.

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[0044] In general, coherent integration processing using Doppler frequency analysis is applied to the reception processing of radar reflected waves containing Doppler fluctuations, so the noise level (floor level) of the Doppler frequency components corresponding to the two items in equation (12) increases, degrading radar detection performance. In addition, the transmission DC offset component α Tx The received power of |γα Tx | 2 Therefore, the higher the received power of the radar reflection wave, the greater the increase in the noise level (floor level), resulting in a greater deterioration of the radar detection performance.

[0045] Next, we will explain a method for preventing an increase in the noise level (floor level) and suppressing deterioration of radar detection performance in a pulse compression radar, even when a transmission DC offset (carrier leak) and a reception DC offset are included, without adding a high-precision correction circuit for circuit error correction.

[0046] In the above-mentioned method of transmitting a pulse compression code to which phase modulation has been applied at a plurality of pulse transmission periods, if a transmission DC offset component (including a carrier leak component) is present in the radar transmitter and there is Doppler fluctuation in the radar reflected wave, the transmission DC offset component remains and the noise level of a specific Doppler component increases. In response to this problem, the transmission DC offset component can be reduced by the following method.

[0047] Specifically, the radar device uses a code A=[a1, a2, … , a L ] and the phase-inverted code -A=[-a1, -a2, … , -a L ] is used to transmit the codes A, -A, -A, and A in four radar transmission cycles, respectively, and correlation processing and coherent integration processing are performed on the receiving side.

[0048] This method will be specifically described below. Note that, as in the above, the following describes the case where there is no noise component and the sum of the elements of code A is not zero (see equation (4)).

[0049] Example 4: The received signal of a radar reflected wave containing Doppler fluctuations is added to the transmitted DC offset component α Tx If it contains >

[0050] In this case, the Doppler fluctuation contained in the radar reflection is expressed as "exp(j2πf d ×T r )=exp(jΨ d )" (f d : Doppler frequency, T r : Radar transmission period. However, this is under the condition that the Doppler fluctuation within the code can be considered constant.

[0051] (4-1) When transmitting code A, the transmission DC offset component α Tx The received signal γ(A+α Tx ) and code A, the autocorrelation value obtained by the autocorrelation calculation is expressed by the following equation (13).

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[0052] (4-2) When transmitting code-A, the transmission DC offset component α Tx The received signal γ(-A+α Tx ) exp(jΨ d The autocorrelation value obtained by calculating the autocorrelation between the code -A and the code -B is expressed by the following equation (14).

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[0053] (4-3) When transmitting code-A, the transmission DC offset component α Tx The received signal contains γ(-A+α Tx ) exp(j2Ψ d The autocorrelation value obtained by calculating the autocorrelation between the code -A and the code -B is expressed by the following equation (15).

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[0054] (4-4) When transmitting code A, the transmission DC offset component α Tx The received signal γ(A+α Tx ) exp(j3Ψ d ) and code A, the autocorrelation value obtained by the autocorrelation calculation is expressed by the following equation (16).

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[0055] At the receiving side, when the autocorrelation values ​​(equations (13) to (16)) obtained in (4-1) to (4-4) above are subjected to coherent integration processing, the transmission DC offset component α Tx Two items including the above will be accumulated.

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[0056] In equation (17), the Doppler phase shift Ψ d If is less than π / 6, then |1-exp(j2Ψ d )|<1", the residual component of the transmit DC offset can be reduced more than in the case shown in <Example 3> (see equation (12)). However, even in <Example 4>, the residual component of the transmit DC offset cannot be completely canceled.

[0057] As described above, coherent integration processing using Doppler frequency analysis is applied to the reception processing of the radar reflected wave containing Doppler fluctuation. Therefore, the noise level (floor level) of the Doppler frequency components containing the residual transmission DC offset components (the two items in Equation (17)) rises, and the radar detection performance deteriorates. For example, as described in <Example 4>, the transmitting side transmits codes A, -A, -A, and A every four radar transmission periods, and the receiving side performs correlation reception processing using the transmission codes on the radar reflected wave, and performs Doppler frequency analysis on the output obtained by coherent integration processing every two radar transmission periods with the transmission code (A, -A) that cancels the reception DC offset as one unit. In this case, the noise level (floor level) of the Doppler frequency components contained in the residual transmission DC offset components rises, and the radar detection performance deteriorates.

[0058] In the Doppler frequency analysis, the noise level (floor level) of a particular frequency component increases due to the residual transmission DC offset component because the output obtained by the coherent integration process has a fixed phase fluctuation.

[0059] Therefore, in one aspect of the present disclosure, in order to prevent the output obtained by coherent integration processing on the receiving side from having a fixed phase fluctuation, a random phase fluctuation is applied to the output obtained by coherent integration processing, with a transmission code (A, -A) that cancels the reception DC offset being treated as one unit.

[0060] Specifically, when a radar device according to one embodiment of the present disclosure performs coherent integration processing using Doppler frequency analysis on the receiving side, the radar device randomly switches between whether to make each unit of transmission codes into a code with a phase inversion added (a code with a phase change of π added) or not, with multiple transmission codes that cancel the reception DC offset as one unit (corresponding to A, -A transmitted in two radar transmission periods in the above cases of <Example 1> to <Example 4>) so that the Doppler phase fluctuation does not become a stationary phase shift amount (phase change amount).

[0061] This causes variation in the phase fluctuation of the output obtained by coherent integration of the radar reflected wave for each transmission code unit, and the remaining transmission DC offset component can be whitened in the Doppler frequency domain. This makes it possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component, and suppress deterioration of radar detection performance.

[0062] Hereinafter, an embodiment according to one aspect of the present disclosure will be described in detail with reference to the drawings. In the embodiments, the same components are denoted by the same reference numerals, and the description thereof will be omitted to avoid duplication.

[0063] [Embodiment 1] [Radar device configuration] FIG. 1 is a block diagram showing a configuration of a radar device 10 according to this embodiment.

[0064] The radar device 10 includes a radar transmitter 100 and a radar receiver 200 .

[0065] The radar transmitter 100 generates a high-frequency (radio frequency) radar signal (radar transmission signal) and transmits the radar transmission signal at a predetermined transmission period.

[0066] The radar receiver 200 receives a reflected wave signal, which is a radar transmission signal reflected by a measurement target. The radar receiver 200 performs processing synchronized with the radar transmitter, for example, using a reference signal (not shown). The radar receiver 200 may also process the received reflected wave signal, for example, to detect the presence or absence of a target, estimate the direction, etc. The measurement target is an object to be detected by the radar device 10, and includes, for example, a vehicle (including four-wheeled and two-wheeled vehicles) or a person.

[0067] [Configuration of radar transmitter 100] The radar transmitter 100 includes a radar transmission signal generator 101 , a radio transmission unit 106 , and a transmission antenna 107 .

[0068] The radar transmission signal generator 101 generates a radar transmission period (T r ), a baseband radar transmission signal (pulse compressed signal) is generated by modulating a code of code length L.

[0069] The radar transmission signal generator 101 operates based on a transmission reference clock obtained by multiplying a reference signal (not shown) by a predetermined factor. Hereinafter, the transmission reference clock frequency is referred to as f TxBB Here, the radar transmission period (T r ) is the transmission reference clock frequency (f TxBB ) at discrete time intervals (1 / f TxBB ) integer N r times(N r ×(1 / f TxBB )).

[0070] The radar transmission signal generating unit 101 includes a code generating unit 102 , a phase rotation control unit 103 , a transmission phase rotating unit 104 , and a modulating unit 105 .

[0071] Specifically, the code generator 102 generates a radar transmission period (T r ), the code generator 102 generates a transmission code of code length L. Specifically, the code generator 102 generates a transmission code Code(m) of code length L in the m-th radar transmission period.

[0072] Hereinafter, each element of the transmission code Code(m) is C n In other words, the transmission code Code(m) is written as L elements {C1(m), C2(m), …, C L (m)}. Also, the element C n (m) is a binary value such as {-1, 1}, or a quadratic value such as {1, -1, -j, j}, where j is an imaginary unit. Also, n=1, 2, ..., L, and m=1, 2, ..., Q. Here, Q represents the number of radar transmission periods used when the radar device 10 measures the distance, Doppler, and direction of arrival, etc.

[0073] As the transmission code, for example, a Barker code, a complementary code, an M-sequence code, a Gold code, or the like, which can obtain low range side lobe characteristics, is preferably applied. The transmission code in each radar transmission cycle may be the same code or different codes. Alternatively, a plurality of transmission codes may be switched in each radar transmission cycle.

[0074] The phase rotation control unit 103 has a plurality of N e Radar transmission period (=N e ×T r ) as a unit, multiple (N e Within the radar transmission period, an integer multiple of 2π (2πN s ) to the transmission code, to the transmission phase rotation unit 104 and the radar reception unit 200 (reception phase rotation unit 206). e is an integer greater than 1, and N s is an integer greater than or equal to 1.

[0075] The phase rotation control unit 103 is e The radar transmission period (=N e ×T r ), the phase rotation control unit 103 outputs a phase rotation amount signal that has a constant phase shift amount (phase change amount) over each radar transmission period. r ), the phase rotation amount φ×0, φ×1, φ×2, …, φ(N e -1) is periodically applied to the phase rotation amount signal. Here, φ=2πN s / N e For example, N e =4, N s If =1, the radar transmission period (T r ) the phase shift amount φ is π / 2. In this case, 4(=N e ) radar transmission periods (T r ) the phase rotation amounts output are, for example, 0, π / 2, π, and 3π / 2.

[0076] When complementary codes are used as transmission codes (including cases where multiple complementary codes are combined and transmitted, such as Spano codes), the amount of phase shift for the code pair that constitutes the complementary codes is set to zero. In other words, the same phase rotation is applied to the code pair that constitutes the complementary codes. This has the effect of maintaining the high side lobe suppression characteristics of the complementary codes.

[0077] For example, when using complementary codes, the phase rotation control unit 103 selects an even number N e radar transmission period (T r ), the phase rotation amount is φ×0, φ×0, φ×1, φ×1, φ×2, φ×2, …, φ(N e -1), φ(N e -1) is periodically applied to the phase rotation amount signal. Here, φ=2πN s / (N e / 2) = 4πN s / N e For example, N e =8, N s = 1, the phase shift amount φ is π / 2. In this case, 8(=N e ) radar transmission periods (T r ) are, for example, 0, 0, π / 2, π / 2, π, π, 3π / 2, and 3π / 2.

[0078] Furthermore, when the phase shift amount φ is set to π / 2, phase rotation can be achieved by swapping the I signal component and the Q signal component (accompanying conversion of positive and negative signs), making it possible to eliminate the need for a multiplier in the transmission phase rotation unit 104.

[0079] Also, N e / N s >2 (N for complementary codes) e / N s By setting the filter to 4), it is possible to obtain an effect of canceling even when the DC offset component has an I signal and a Q component.

[0080] Furthermore, the phase rotation control unit 103 is e Radar transmission period (=Ne ×T r ) in units of a plurality of N e The period corresponding to the radar transmission cycle (=N e ×T r The phase rotation pattern for the radar transmission signal during the period (N e ×T r ) for each

[0081] Here, a random pattern that varies the phase rotation may be used as the phase variable pattern. For example, the phase rotation control unit 103 uses a pseudorandom code (PN code), an M-sequence code, or a Gold code as the phase variable pattern, and controls the amount of phase rotation according to the code polarity of each code of the phase variable pattern.

[0082] As an example, the phase variable pattern is N PP Each element has two values, {-1, 1}. In the following, each element of the phase variable pattern is denoted as PP(q), where q=1, 2, ..., N PP The phase rotation control unit 103 sequentially reads out each element PP(q) of the phase variable pattern and e Radar transmission period (=N e ×T r ), the value of the same element is repeatedly output over the m-th radar transmission period. That is, the phase rotation amount variable signal PC(m) output from the phase rotation control unit 103 in the m-th radar transmission period is expressed by the following equation (18).

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[0083] where m=1,…,N e ×N d Note that N d is a parameter defined in the Doppler analysis unit 208 described later. Note that the number of elements of the phase variable pattern N PPThe number of radar transmission periods Q (=N e ×N d ), the phase rotation control unit 103 cyclically reads out the phase variable pattern in Q radar transmission periods.

[0084] That is, the phase variable pattern is a plurality of N e radar transmission period (T r ), each element PP(q) is set to change randomly.

[0085] The transmission phase rotation unit 104 imparts phase rotation to the transmission code output from the code generation unit 102, based on the phase rotation amount signal and the phase rotation amount variable signal instructed by the phase rotation control unit 103. The transmission phase rotation unit 104 outputs the transmission code with the phase rotation imparted to the modulation unit 105. For example, the transmission phase rotation unit 104 outputs a signal GP(m) obtained by imparting phase rotation to the transmission code Code(m) output from the code generation unit 102 in the m-th radar transmission period, as shown in the following equation (19).

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[0086] In this way, the transmission phase rotation unit 104 imparts phase rotation to the transmission code (radar transmission signal) in accordance with the phase variable pattern (PC(m)).

[0087] As described above, when the transmission codes are complementary codes, the effect of maintaining the characteristic of canceling the distance side lobe between the codes constituting the complementary code pair (high side lobe suppression characteristic) can be obtained by not shifting the phase between the codes constituting the complementary code pair (applying the same phase rotation). That is, when the transmission codes are complementary codes, the transmission phase rotation unit 104 outputs a signal GP(m) obtained by applying phase rotation to the transmission code Code(m) so that the phase shift within two transmission periods in which the codes constituting the complementary code pair are transmitted becomes zero, as shown in the following equation (20).

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[0088] The modulation section 105 performs pulse modulation (amplitude modulation (ASK: Amplitude Shift Keying) or phase modulation (PSK: Phase Shift Keying)) on the transmission code output from the transmission phase rotation section 104, and outputs a modulated signal (radar transmission signal) to the transmission radio section .

[0089] For example, when the modulation unit 105 uses phase-shift keying (PSK), phase modulation in which the transmission code is two-valued, such as {-1, 1}, becomes BPSK, and phase modulation in which the transmission code is four-valued, such as {1, -1, -j, j}, becomes QPSK or 4-phase PSK, and a specified modulation symbol on the IQ phase plane is assigned.

[0090] Moreover, the modulation unit 105 passes the modulated signal, which is obtained by modulating the transmission code, through a band-limiting filter (not shown) to output the modulated signal restricted within a predetermined band.

[0091] Here, the in-phase component of the modulating signal is I(n s ), and the quadrature components are Q(n s ), the modulation signal G(n s ) can be expressed as follows:

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[0092] where n s is a natural number and represents discrete time. Also, the discrete time interval is (1 / f TxBB ), and f TxBB is a transmission reference clock frequency obtained by multiplying the reference signal by a predetermined factor.

[0093] Furthermore, the modulation unit 105 performs modulation using No samples of the transmission reference clock for each code output from the transmission phase rotation unit 104. As a result, for a transmission code with a code length of L, Nw=No×L samples are included in the radar signal section Tw. r ) in the no-signal section (T r -Tw) is the number of transmission reference clocks Nu (=N r −Nw) samples are included (see, for example, FIG. 2). Therefore, the modulated signal in the m-th radar transmission period can be expressed as the following equation (22).

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[0094] Radio transmission unit 106 performs quadrature modulation on the signal output from modulation unit 105, performs frequency conversion to generate a radar transmission signal in a carrier frequency (Radio Frequency: RF) band, amplifies the signal to a predetermined transmission power by a transmission amplifier, and outputs the signal to transmission antenna 107. Transmission antenna 107 radiates the radar transmission signal output from radio transmission unit 106 into space.

[0095] A common reference signal is applied to the local oscillators of the radio transmission unit 106 and the radio reception unit 202 described later. This makes it possible to synchronize the local oscillators of the radio transmission unit 106 and the radio reception unit 202.

[0096] Moreover, the radar transmitter 100 may include a radar transmission signal generator 101a shown in Fig. 3 instead of the radar transmission signal generator 101. The radar transmission signal generator 101a does not include the code generator 102, phase rotation controller 103, transmission phase rotator 104, and modulator 105 shown in Fig. 1, but instead includes a code storage unit 111 and a DA converter 112. The code storage unit 111 stores in advance the code sequence generated in the code generator 102 (Fig. 1) and cyclically reads out the stored code sequences in sequence. The DA converter 112 converts the code sequence (digital signal) output from the code storage unit 111 into an analog signal.

[0097] [Configuration of radar receiver 200] In FIG. 1, the radar receiver 200 includes a receiving antenna 201 , a radio receiving unit 202 , and a signal processing unit 203 .

[0098] The receiving antenna 201 receives a signal (reflected wave signal) that is the RF band radar transmission signal transmitted from the radar transmitting unit 100 reflected by a reflecting object including a measurement target, and outputs the received reflected wave signal to the receiving radio unit 202 as a received signal.

[0099] The receiving radio unit 202 amplifies the received signal output from the receiving antenna 201 to a predetermined level, frequency-converts the high-frequency band received signal to a baseband band, and converts the baseband band received signal to a baseband band received signal including an I signal (In-Phase signal component) and a Q signal (Quadrature signal component).

[0100] The signal processing unit 203 has an AD conversion unit 204 , a correlation calculation unit 205 , a reception phase rotation unit 206 , a coherent integration unit 207 , and a Doppler analysis unit 208 .

[0101] Each unit in the signal processing unit 203 operates based on a reception reference clock obtained by multiplying a reference signal (not shown) by a predetermined number. In the following, the reception reference clock frequency is defined as f RxBB Here, the radar transmission period (T r ) is the reception reference clock frequency (f RxBB ) at discrete time intervals (1 / f RxBB ) integer N v times(N v ×(1 / f RxBB In the following, the transmission reference clock frequency f TxBB is the reception reference clock frequency f RxBB Integer multiple of N TR Relationship between f TxBB =f RxBB ×N TR It shall be deemed to be in.

[0102] The AD conversion unit 204 converts the baseband signal including the I signal and the Q signal output from the radio reception unit 202 into a received reference clock frequency f RxBB Based on discrete time (1 / f RxBB ) to convert the I and Q signals into digital data.

[0103] In the following description, a baseband received signal including I and Q signals at discrete time k is treated as a complex signal x(k)=I r (k)+j Q r In the following, the discrete time k is the mth radar transmission period (T r ) is used as a reference (k=1), and the signal processor 203 calculates the radar transmission period T r The measurement is performed periodically until the end of k=1,…,N. v Here, j is the imaginary unit.

[0104] Therefore, the output signal of AD conversion unit 204 in the m-th radar transmission period can be expressed as in the following equation (23): Hereinafter, X(k) will be referred to as a complex baseband signal.

number

[0105] The correlation calculation unit 205 calculates the radar transmission period T r The complex baseband signal X(N v (m-1)+k) and the transmission code C transmitted by the radar transmitter 100 n (m), where n = 1, ..., L. For example, a correlation calculation value AC(k, m) of the sliding correlation calculation at discrete time k in the m-th radar transmission period is calculated based on the following equation (24).

number

[0106] In equation (24), the asterisk (*) represents the complex conjugate operator, and k = 1, ..., N v It is.

[0107] The correlation calculation unit 205 calculates the correlation coefficient k=1, . . . , N v The measurement range (i.e., the range of k) may be limited according to the range of the target to be measured by the radar device 10. This allows the radar device 10 to reduce the amount of calculation processing by the correlation calculation unit 205. For example, the correlation calculation unit 205 may perform the correlation calculation for k=Nw / N TR +1,…, (Nu - Nw) / N TR In this case, the radar device 10 does not perform measurements in a time period corresponding to the code transmission period Tw.

[0108] As a result, even if the radar transmission signal directly penetrates the radar receiver 200, the radar device 10 does not perform processing by the correlation calculator 205 during the period in which the radar transmission signal penetrates, and therefore can perform measurements without the effects of the interference. When limiting the measurement range (range of k), similar processing that limits the measurement range (range of k) can be applied to the processing of the reception phase rotation unit 206, the coherent integration unit 207, and the Doppler analysis unit 208 described below. This can reduce the amount of processing in each component, and can reduce power consumption in the radar receiver 200.

[0109] The reception phase rotation unit 206 rotates the radar transmission period (T r ), a phase rotation PC(m)exp[-j{(m-1)modN e}φ] is added to the signal (correlation calculation value) output from the correlation calculation unit 205. That is, r ), reception phase rotation section 206 outputs signal ACP(k,m) obtained by applying phase rotation to output AC(k,m) from correlation calculation section 205, as shown in the following equation (25).

number

[0110] That is, the reception phase rotation unit 206 imparts phase rotation in the opposite direction to the phase rotation of the transmission phase rotation unit 104 to the output AC(k,m) (reflected wave signal) of the correlation calculation unit 205 in accordance with the phase variable pattern (PC(m)).

[0111] For example, N e =4, N s = 1, PC(m) = 1, the radar transmission period (T r ) the phase shift amount -φ is -π / 2. In this case, 4(=N e ) radar transmission periods (T r The phase rotation amount output in each of N e =4, N s = 1, PC(m) = -1, the radar transmission period (T r ) the phase shift amount -φ is -π / 2. However, when PC(m)=-1, exp(jπ)=-1, so 4(=N e ) radar transmission periods (T r The phase rotation amounts output in each of the phase variable patterns PC(m) are, for example, π, π / 2, 0, -π / 2. That is, the difference between the corresponding phase rotation amounts -φ (here, {0, -π / 2, -π, -3π / 2} and {π, π / 2, 0, -π / 2}) between different phase variable patterns (PC(m)=1, -1) is π.

[0112] In addition, when complementary codes are used as transmission codes, in order to make the pair of codes constituting the complementary codes have the same phase, the reception phase rotation unit 206 outputs the signal ACP(k, m) obtained by applying phase rotation to the output AC(k, m) of the correlation calculation unit 205 so that the phase shift within the transmission period of the two codes constituting the complementary codes is zero, as shown in the following equation (26).

number

[0113] For example, N e =8, N s = 1, PC(m) = 1, the phase shift amount -φ is -π / 2. e ) radar transmission periods (T r The phase rotation amounts output in each of N e =8, N s = 1, PC(m) = -1, the phase shift amount -φ is -π / 2. However, when PC(m) = -1, due to the relationship of exp(jπ) = -1, 8(=N e ) radar transmission periods (T r ) are, for example, π, π, π / 2, π / 2, 0, 0, -π / 2, -π / 2.

[0114] The coherent integrator 207 calculates the correlation coefficient ACP(k,m) a predetermined number of times N using the correlation calculation value ACP(k,m) output from the reception phase rotation unit 206 at each discrete time k in the m-th radar transmission period. e Over the radar transmission period, the correlation calculation value ACP(k,m) is summed (coherent integrated) at each discrete time k to calculate the coherent integration value ACC(k,v) at each discrete time k, where k=1,...,N v It is.

[0115] Specifically, the v-th coherent integral value ACC(k,v) is calculated as shown in the following equation (27).

number

[0116] Here, the period of the radar transmission cycle during which the correlation calculation value ACP(k,v) is added at each discrete time k in the coherent integrator 207 (i.e., the integration period of the coherent integrator 207) is set to a predetermined number of times N eBy setting it to, even if the correlation calculation value ACP(k,v) contains a reception DC offset, the reception DC offset component can be cancelled based on the following principle, and the noise component can be suppressed while preventing the deterioration of radar detection performance due to the reception DC offset.

[0117] That is, the received DC offset component α Rx , and Doppler frequency variation f dTx The transmit DC offset component α Tx exp(j2πf dTx ×T r ), the output AC(k,m) of correlation calculation section 205 contains the component shown in the following equation (28), regardless of k and m.

number

[0118] Moreover, the output ACP(k,m) of the reception phase rotation unit 206 includes a component shown in the following equation (29), regardless of k.

number

[0119] Therefore, the output ACC(k,v) of the coherent integrator 207 contains the component shown in the following equation (30), regardless of k.

number

[0120] As shown in equation (30), in the integration interval of the coherent integrator 207,

number

number

[0121] On the other hand, the Doppler frequency fluctuation f dTx When is not zero, that is, when the reflected wave signal contains Doppler frequency fluctuations, the output ACC(k,v) of the coherent integrator 207 contains a transmission DC offset component shown in the following equation (31), regardless of k.

number

[0122] The phase rotation control unit 103 repeats the predetermined number of times N e (i.e., for each integration interval of the coherent integration unit 207), a phase variable pattern PC(N e Control is performed to randomly change (k, v) to 1 or -1. As a result, in the output of the Doppler analysis unit 208, the residual transmission DC offset component contained in the output ACC(k, v) of the coherent integration unit 207 shown in equation (31) is whitened in the Doppler frequency domain. Therefore, it is possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component, and to suppress deterioration of the radar detection performance.

[0123] The Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integrator 207. Specifically, the Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integrator 207 obtained at each discrete time k. d Outputs ACC(k, 1) to ACC(k,N d ) as one unit, and the timing of the discrete time k is aligned to perform Doppler frequency analysis. d Outputs ACC(k, 1) to ACC(k,N d) is whitened in the Doppler frequency domain, so that it is possible to suppress an increase in the noise level (floor level) of a specific frequency component due to the transmission DC offset component in the Doppler frequency analysis by the Doppler analysis unit 208.

[0124] Specifically, the Doppler analysis unit 208 calculates the Doppler vector 2N f Different Doppler frequencies f s Phase variation Ψ(f s )=2πf s (T r ×N e ) ΔΨ is corrected before coherent integration is performed.

number

[0125] Here, FT_CI(k, f s ) is the Doppler frequency f at discrete time k of the Doppler analysis unit 208 s The coherent integration result of ΔΨ is shown below. Note that f s =-N f +1,…,0,…,N f where k=1,…, (N r +N u )N s / N o where ΔΨ is the phase rotation unit.

[0126] As a result, the signal processing unit 203 calculates 2N f The coherent integration results for the Doppler frequency components FT_CI(k, -N f +1), …, FT_CI(k, N f -1) for the radar transmission period T r Multiple times of N e ×N d Period (T r ×N e ×N d =T r × Q), where j is the imaginary unit.

[0127] ΔΨ=1 / N d In this case, the processing of the Doppler analysis unit 208 is performed at a sampling interval (T r ×N e ), sampling frequency 1 / (T r ×N e ) is equivalent to subjecting the output of coherent integrator 207 to discrete Fourier transform (DFT).

[0128] Also, N f By setting N to a power of 2, the Doppler analysis unit 208 can apply Fast Fourier Transform (FFT) processing, thereby reducing the amount of calculation processing. f >N d So, q>N d In the domain where ACC(k,N d By performing zero padding such that (w-1)+q+1)=0, FFT processing can be similarly applied, thereby reducing the amount of calculation processing.

[0129] The radar device 10 receives the output FT_CI(k, f s ) to estimate the range / Doppler frequency (relative velocity) of the radar measured target.

[0130] That is, the output FT_CI(k, f s ) squared absolute value |FT_CI(k, f s )| 2 is the Doppler frequency f s This corresponds to the reflected wave reception level from the target for each time. Based on the discrete time k at which the noise level reaches a peak power value equal to or greater than a predetermined value and the Doppler f, the radar device 10 calculates the distance R(k) to the target and the relative velocity v based on the Doppler frequency. d (f s ) can be estimated as follows: s =-N f +1,..,0,...,N f and k=1,…, (N r +N u )Ns / N o It is.

[0131] The following equation (33) is used to convert the time information k into distance information R(k): where Tw is the code transmission section, L is the pulse code length, and C0 is the speed of light.

number

[0132] In addition, the Doppler frequency information f s The relative velocity component v d (f s ), the following equation (34) is used: where λ is the wavelength of the carrier frequency of the radar transmission signal in the RF band output from the transmission radio unit 106.

number

[0133] The effects of the above-described operations of phase rotation control section 103, transmission phase rotation section 104 and reception phase rotation section 206 were confirmed using a computer simulation, and the results are shown below.

[0134] 4A and 4B show the output (reception level) of the Doppler analysis unit 208 under the condition that one measurement target is moving away from the radar device 10 at a distance of 5 m at a speed of 20 km / h.

[0135] 4A and 4B, it is assumed that a transmission DC offset (carrier leak) exists in the transmission radio unit 106. Also, in FIG. 4A and FIG. 4B, a complementary code (code length L=64) is used as the radar transmission code, and the number of additions in the coherent integrator 207 is N e = 32, the number of samples in the Doppler analysis unit 208 N d The results of a computer simulation using =512 are shown below.

[0136] 4A shows the result when the phase inversion (PC(m)) is not randomly switched as in the conventional radar device 10, unlike the operation of the radar device 10 according to this embodiment. On the other hand, FIG. 4B shows the result when the phase inversion is randomly switched by the phase variable pattern PC(m) in the phase rotation control unit 103 as described above.

[0137] In both FIG. 4A and FIG. 4B, a sharp peak appears at the coordinates where the distance is 5 m and the speed is 20 km / h, and it can be seen that the desired measurement target has been detected.

[0138] However, in Fig. 4A, in addition to the peak of the desired measurement target at a distance of 5 m and a speed of 20 km / h, weak peaks of the reception level appear uniformly in the entire distance range (0 to 200 m) in the Doppler frequency component at a speed of 20 km / h. These weak peaks of the reception level are frequency components that do not actually exist, and they can cause false detection.

[0139] On the other hand, in Fig. 4B showing the results based on the operation of this embodiment, apart from the peak of the desired measurement target at a distance of 5 m and a speed of 20 km / h, there is no phenomenon detected in Fig. 4A in which weak peaks of reception levels appear uniformly over all distance ranges for a specific Doppler frequency component. In other words, it can be confirmed that according to this embodiment, even under conditions in which a transmission DC offset (carrier leak) exists in the transmission radio unit 106, there is no degradation of radar performance that could cause erroneous detection.

[0140] As described above, the radar device 10 has a plurality of N e radar transmission period (T r ) period (N e ×T r ) for the radar transmission signal (transmission code) in the period (N e ×T r), a transmission phase rotation unit 104 that imparts a phase rotation (first phase rotation) to the radar transmission signal in accordance with the phase variable pattern, and a reception phase rotation unit 206 that imparts a phase rotation in the opposite direction to the first phase rotation to the reflected wave signal (correlation calculation value ACC(k,m) in FIG. 1 ) in accordance with the phase variable pattern.

[0141] That is, by the operations of the phase rotation control unit 103, the transmission phase rotation unit 104, and the reception phase rotation unit 206, the radar device 10 e ×T r ) are treated as one unit of transmission codes that cancel the received DC offset in the BERT. For each unit of transmission codes, a random switch is made between whether or not to use a code with phase inversion added (a code with a phase change of π added).

[0142] As a result, the output of the coherent integration processing in the coherent integrator 207 of the radar device 10 does not have a fixed phase fluctuation, and the remaining transmission DC offset component is whitened in the Doppler frequency domain. Therefore, in the output from the Doppler analyzer 208, it is possible to prevent the noise level (floor level) of a specific Doppler frequency component from increasing, and to suppress deterioration of radar detection performance.

[0143] Therefore, according to this embodiment, it is possible to suppress the deterioration of radar detection performance caused by circuit errors.

[0144] Furthermore, according to the present embodiment, the radar device 10 can prevent the deterioration of radar detection performance caused by a transmission DC offset without having a circuit configuration for correcting the transmission DC offset (carrier leak) with high accuracy, and therefore the configuration of the radar device 10 can be simplified.

[0145] Although FIG. 1 shows a case where the reception phase rotation unit 206 of the radar device 10 is arranged after the correlation calculation unit 205, as shown in FIG. 5, the reception phase rotation unit 206 may be arranged before the correlation calculation unit 205, and the same results as those of the above embodiment can be obtained.

[0146] [Embodiment 2] The radar device according to this embodiment has a basic configuration in common with the radar device 10 according to the first embodiment, and therefore will be described with reference to FIG.

[0147] In this embodiment, a case where a plurality of radar devices 10 (FIG. 1) according to the first embodiment are provided will be described as shown in Fig. 6. In the following, as an example, a case where two radar devices A and B shown in Fig. 6 are provided will be described.

[0148] In this embodiment, the phase variable patterns controlled by the phase rotation control units 103 of the multiple radar devices 10 are made different from one another. For example, in the radar device A shown in FIG. 6, the phase variable pattern PP (1) (q) (where q=1,2,…,N pp ) is set, and in radar device B, PP (1) Phase variable pattern PP different from (q) (2) (q) (where q=1,2,…,N pp ) is set.

[0149] As an example, the phase rotation control unit 103 of each radar device 10 may set different phase variable patterns by shifting the transmission timing of the same M sequence code. For example, the phase variable pattern PP (1) (q) is a pattern using an M-sequence code with a code length of 511 (where q=1, . . . , 511). In this case, the phase variable pattern PP (2) (q) is the M sequence code used in radar device A. shift In other words, the timing of sending the PP (2) (q)=PP (1) (q+N shift ).

[0150] For example, FIG. 6 shows an example in which the radar receiver 200 of radar device A receives a received signal (desired signal) that is a reflected wave of a radar transmission signal transmitted by radar device A, and a reflected wave (interference signal) of a radar transmission signal transmitted by radar device B.

[0151] Even in this case, as described above, by making the phase variable patterns different between the radar devices A and B, the transmission DC offset components remaining in the received signals from the radar devices A and B are whitened in the Doppler frequency domain, as in the first embodiment. This makes it possible to prevent the noise level (floor level) of a specific Doppler frequency component from increasing, and suppress deterioration of the radar detection performance of the radar device A. In other words, it is possible to obtain the effect of reducing mutual interference between multiple radar devices 10 that use the same frequency band or partially overlapping frequency bands.

[0152] FIG. 7 shows the results of a computer simulation evaluation of the amount of mutual interference when radar device A and radar device B transmit radar transmission signals using different phase variable patterns.

[0153] In FIG. 7, radar device A and radar device B use complementary codes as transmission codes and adjust the phase shift in phase rotation control unit 103 to N e =16, N s In addition, in the Doppler analysis unit 208 of the radar device A and the radar device B, N d =512 was used.

[0154] The horizontal axis of Fig. 7 is the radar transmission period (T r The vertical axis shows the amount of time shift in the transmission timing of the phase-variable pattern in units of 1 / s, and the vertical axis shows the received power of radar device A (Desired Power [plotted with black circles]), the received power of radar device B (Undesired Power [plotted with crosses]), and the signal-to-interference ratio (SIR [plotted with white circles]) of radar device A.

[0155] From the simulation results shown in FIG. 7, the phase variable patterns of radar device A and radar device B are 16 (=N e It can be confirmed that if there is a time shift in the transmission timing of at least one radar transmission period, the SIR of radar device A is improved by about 20 dB.

[0156] The phase rotation control unit 103 sequentially reads out each element PP(q) of the phase variable pattern and e Radar transmission period (=N e ×T r ), the same element value is output repeatedly. shift ≧1, 16(=N e ) radar transmission period, the effect of suppressing interference can be improved. The improvement in SIR is proportional to the code length N pp and the parameter N used in the Doppler analysis unit 208 d (For example, N d For this reason, we recommend that we set the N p and N d It is more preferable to use the above in terms of improving SIR.

[0157] As described above, in this embodiment, the phase variable patterns are different among the multiple radar devices 10. For example, the transmission timing of the same M sequence code used in the phase variable patterns of each of the multiple radar devices 10 is shifted. In this way, mutual interference between the multiple radar devices 10 can be reduced.

[0158] In the above case, the transmission timings of the multiple radar devices 10 may coincide by chance, so there is a possibility that the mutual interference cannot be suppressed probabilistically. In response to this, the mutual interference can be reduced probabilistically by the following method.

[0159] Specifically, in the multiple radar devices 10, the code length N ppAmong the M-sequence codes, preferred pairs with low cross-correlation are used in each phase variable pattern.

[0160] Or, code length N pp Among the M-sequence codes, N are selected from the preferred pairs with low cross-correlation. pp It is known that different Gold codes can be generated from the preferred pair having low cross-correlation. Therefore, in the multiple radar devices 10, the Gold codes generated from the preferred pair having low cross-correlation may be used for each phase variable pattern.

[0161] Alternatively, in a plurality of radar devices 10, the phase variable pattern of each may be varied randomly for each radar measurement (or for each predetermined number of measurements), thereby reducing mutual interference probabilistically.

[0162] [Embodiment 3] In the second embodiment, a method for reducing mutual interference between multiple radar devices 10 has been described by making the phase variable patterns different among the multiple radar devices 10. In contrast, in the present embodiment, a configuration of a MIMO (Multiple Input Multiple Output) radar using multiple transmission and reception antennas instead of multiple radar devices 10 will be described.

[0163] That is, in this embodiment, the phase variable patterns are different between a plurality of transmitting antennas (that is, between MIMO streams).

[0164] Fig. 8 is a block diagram showing a configuration example of a radar device 20 according to this embodiment. In Fig. 8, the same components as those in the first embodiment (Fig. 1) are given the same reference numerals, and the description thereof will be omitted.

[0165] The radar device 20 shown in Fig. 8 shows a configuration of a time-division MIMO radar that switches between multiple transmission antennas in a time-division manner to transmit different radar transmission signals that are time-division multiplexed, and separates and processes the radar transmission signals. Note that the configuration of the radar device is not limited to this, and the radar device 20 may also be configured to send out different transmission signals that are frequency-division multiplexed or code-division multiplexed from multiple transmission antennas, and separate and process the transmission signals.

[0166] [Configuration of radar transmitter 300] The radar transmitter 300 of the radar device 20 includes radar transmission signal generators 101-1 to 101-Nt, a switching controller 301, a transmission switching unit 302, and a transmitting array antenna unit 303.

[0167] The transmitting array antenna unit 303 is made up of Nt transmitting antennas (Tx#1 to Tx#Nt).

[0168] Radar transmission signal generators 101-1 to 101-Nt are provided corresponding to the Nt transmitting antennas (Tx#1 to Tx#Nt), respectively. Each radar transmission signal generator 101 operates in the same manner as in the first embodiment (FIG. 1). However, the phase variable patterns set in each radar transmission signal generator 101 are different from each other.

[0169] For example, the phase variable pattern in each of the Nt radar transmission signal generators 101 is set by shifting the transmission timing of the same M-sequence code by one or more code elements.

[0170] Specifically, the phase variable pattern in the radar transmission signal generating unit 101-1 is set to PP (1) (q) (where q = 1,…, N pp In this case, the phase variable pattern in the radar transmission signal generating unit 101-2 is set as PP (2) (q+N shift ), and the phase variable pattern in the radar transmission signal generating unit 101-3 is set as PP (3) (q+2N shift)), and similarly, the phase variable pattern in the radar transmission signal generator 101-Nt is set to PP (Nt) (q+(Nt-1)N shift ) is also acceptable. However, N shift ≧1.

[0171] In addition, as the phase variable pattern in each of the Nt radar transmission signal generators 101, pp Alternatively, a preferred pair with low cross-correlation may be used among the M-sequence codes of the code length N pp Among the M-sequence codes, N are selected from the preferred pairs with low cross-correlation. pp It is known that different Gold codes can be generated from a preferred pair having low cross-correlation. Therefore, a Gold code generated from a preferred pair having low cross-correlation may be used for each phase variable pattern in the Nt radar transmission signal generators 101. Alternatively, the phase variable patterns in the Nt radar transmission signal generators 101 may be randomly varied for each radar measurement (or for each predetermined number of measurements), thereby reducing mutual interference probabilistically.

[0172] The switching control unit 301 outputs a control signal (hereinafter referred to as a switching control signal) that instructs the switching timing of the transmitting antennas (Tx#1 to Tx#Nt) of the transmitting array antenna unit 303 (i.e., output switching of the radar transmission signal), to the transmission switching unit 302 and the radar receiving unit 400 (the correlation calculation unit 205 and the output switching unit 403).

[0173] The transmission switching unit 302 selects one of the Nt transmitting antennas of the transmitting array antenna unit 303 based on a switching control signal from the switching control unit 301, and inputs to the selected transmitting antenna an output signal of the radar transmission signal generation unit 101 corresponding to the selected transmitting antenna among outputs of the Nt radar transmission signal generation units 101. The transmission switching unit 302 frequency-converts the output signal of the selected radar transmission signal generation unit 101 (radar transmission signal in baseband) to a predetermined radio frequency band, and outputs the converted signal to the selected (connected) transmitting antenna.

[0174] The transmitting array antenna unit 303 radiates the radar transmission signal output from the transmission switching unit 302 into space from the transmitting antenna selected (connected) by the transmission switching unit 302 .

[0175] The following describes the control operation of the switching control unit 301 over the transmission switching unit 302. Note that the control operation of the switching control unit 301 over the radar receiver 400 will be described later in the description of the operation of the radar receiver 400.

[0176] The switching control unit 301 is e The radar transmission signal generator 101 outputs a switching control signal to the transmission switching unit 302 for each radar transmission period, which switches between the radar transmission signal generator 101 and the transmission antenna in sequence.

[0177] For example, the switching control unit 301 e Radar transmission period (N e ×T r ), the output signal of the radar transmission signal generation unit 101-1 is input to the transmission switching unit 302, which converts the signal into a high-frequency signal and outputs it to the transmission antenna (Tx#1) of the transmission array antenna unit 303.

[0178] The switching control unit 301 selects the next N e Radar transmission period (N e ×T r ), the output signal of the radar transmission signal generation unit 101-2 is input to the transmission switching unit 302, which converts the signal into a high-frequency signal and outputs it to the transmission antenna (Tx#2) of the transmission array antenna unit 303.

[0179] The switching control unit 301 repeats the same operation, inputting the output signal of the radar transmission signal generation unit 101-Nt to the transmission switching unit 302, and outputs the signal converted into a high-frequency signal to the transmitting antenna (Tx#Nt) of the transmitting array antenna unit 303.

[0180] In addition, the switching control unit 301 selects the next N e Radar transmission period (N e ×T r), the output signal of the radar transmission signal generation unit 101-1 is again input to the transmission switching unit 302, which converts the signal into a high-frequency signal and outputs it to the transmission antenna (Tx#1) of the transmission array antenna unit 303.

[0181] The switching control unit 301 repeats the above operation a predetermined number of times (N d ×N t Repeat 1 time.

[0182] In the above operation, the radar transmitter 300 operates to sequentially read out GP(m) (see, for example, equation (19) or equation (20)) described in the first embodiment as the signal generated by each radar transmission signal generator 101. Here, m=1,...,N e ×N d It is.

[0183] [Configuration of radar receiver 400] The radar receiving unit 400 of the radar device 20 includes a receiving array antenna unit 401 , an antenna system processing unit 402 , and a direction estimating unit 404 .

[0184] The receiving array antenna unit 401 is composed of Na receiving antennas (Rx#1 to Rx#Na). The Na receiving antennas each receive a signal (reflected wave signal) that is a radar transmission signal transmitted from the radar transmitting unit 300 and reflected by a reflecting object including a radar measurement target. Each signal received by the Na receiving antennas is input as a received signal to an antenna system processing unit 402 corresponding to each receiving antenna (Rx#1 to Rx#Na).

[0185] Each antenna system processing unit 402 includes a receiving radio unit 202 and a signal processing unit 203 .

[0186] The receiving radio unit 202 of the z-th antenna system processor 402-z amplifies the received signal from the z-th receiving antenna (Rx#z) to a predetermined level, frequency-converts the high-frequency band received signal to a baseband band, and converts the baseband band received signal to a baseband band received signal including an I signal and a Q signal, where z=1,...,Na.

[0187] The signal processing unit 203 of the z-th antenna system processing unit 402-z is composed of an A / D conversion unit 204, a correlation calculation unit 205, a reception phase rotation unit 206, an output switching unit 403, a coherent integrator 207, and a Doppler analyzer 208. The signal processing unit 203 includes Nt coherent integrators 207 and Doppler analyzers 208 corresponding to the transmitting antennas (Tx#1 to Tx#Nt), respectively.

[0188] Hereinafter, the operations of each component of signal processing unit 203 in z-th antenna system processing unit 402-z that differ from those in the first embodiment will be mainly described.

[0189] The correlation calculation unit 205 calculates the complex baseband signal X(N v (m-1)+k) (see, for example, equation (23)) and N e Radar transmission period (N e ×T r ) and a correlation calculation is performed with the transmission code generated in the radar transmission signal generator 101 selected for each of the signals.

[0190] The reception phase rotation unit 206 has N e Radar transmission period (N e ×T r ), a phase rotation in the opposite direction (cancelling direction) to the phase rotation imparted by the transmission phase rotation unit 103 of the radar transmission signal generation unit 101 selected by the switching control unit 301 is imparted to the signal (correlation calculation value) output from the correlation calculation unit 205.

[0191] The output switching unit 403 is a switching control unit 301. e Radar transmission period (N e ×T r The outputs are switched to coherent integrators 207-1 to 207-Nt corresponding to the transmitting antenna numbers (#1 to #Nt) selected for each of the coherent integrators.

[0192] For example, when the switching control unit 301 selects the transmitting antenna (Tx#1), ​​the output switching unit 403 switches the signal from the reception phase rotation unit 206 to the coherent integrator 207-1 corresponding to the transmitting antenna (Tx#1) and outputs it.

[0193] Furthermore, when the switching control unit 301 selects the transmitting antenna (Tx#2), the output switching unit 403 switches the signal from the reception phase rotation unit 206 to the coherent integrator 207-2 corresponding to the transmitting antenna (Tx#2) and outputs it.

[0194] The output switching unit 403 repeats the same operation, and when a transmitting antenna (Tx#Nt) is selected by the switching control unit 301, it switches the signal from the receiving phase rotation unit 206 to the coherent integrator unit 207-Nt corresponding to the transmitting antenna (Tx#Nt) and outputs it.

[0195] No. N D 207-Nth coherent integrator D In the switching control unit 301, e Radar transmission period (N e ×T r ) for the output of the reception phase rotation unit 206 selected for each N e Period (T r ×N e ) as a unit of coherent integration. Here, N D = 1,…,Nt.

[0196] The Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integration unit 207. That is, the Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integration unit 207 obtained at each discrete time k. d Using these outputs, the timing of discrete time k is aligned and Doppler frequency analysis is performed.

[0197] As described above, in the radar device 20, the predetermined number of times N eThe switching control unit 301 switches the transmitting antennas for each k, v (i.e., integration interval of the coherent integrator 207), and different phase variable patterns PC(m) are set between the transmitting antennas. As a result, in the output of the Doppler analysis unit 208 corresponding to each transmitting antenna, the remaining transmission DC offset component contained in the output ACC(k, v) (see equation (31)) of the coherent integrator 207 for the radar transmission signal transmitted from each transmitting antenna is whitened in the Doppler frequency domain, as in the first embodiment. This makes it possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component, and suppress deterioration of radar detection performance. Furthermore, by setting different phase variable patterns between the transmitting antennas, it is possible to reduce mutual interference between the radar transmission signals transmitted by each transmitting antenna.

[0198] In the following description, the wth output FT_CI from the Doppler analysis unit 208 at each discrete time k is obtained by performing similar processing in each of the antenna system processing units 402-1 to 402-Na. (z) (1) (k,fs,w) ,…, FT_CI (z) (Na) (k, fs, w) are collectively expressed as a virtual receiving array correlation vector h(k, fs, w) as shown in the following equations (35) and (36). The virtual receiving array correlation vector h(k, fs, w) includes Nt×Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na. The virtual receiving array correlation vector h(k, fs, w) is used in the description of the process of estimating the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas, which will be described later. Here, z=1,...,Nt,N D = 1,..,, Na. Note that f s =-N f +1,..,0,...,N f It is.

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[0199] The above has described the processing in each component of the signal processing unit 203 of the antenna system processing unit 402-z.

[0200] The direction estimation unit 404 calculates an array correction value h_cal for correcting the phase deviation and amplitude deviation between the transmitting antennas of the transmitting array antenna unit 303 and between the receiving antennas of the receiving array antenna unit 401 for the virtual array correlation vector h(k, fs, w) from the w-th Doppler analysis unit 208 at every discrete time k output from the antenna system processing units 402-1 to 402-Na. [b] By multiplying by , the virtual receiving array correlation vector h _after_cal (k, fs, w) is calculated. The virtual receiving array correlation vector h _after_cal (k, fs, w) are expressed by the following equation (37), where b = 1, .., (Nt × Na).

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[0201] Virtual receiving array correlation vector h with correction for deviation between antennas _after_cal (k, fs, w) is Na×N r In the following, the virtual receiving array correlation vector h _after_cal Let each element of (k, fs, w) be h1(k, fs, w),…,h Na×Nr The notation (k, fs, w) is used to explain the direction estimation process.

[0202] Then, the direction estimator 404 calculates the virtual receiving array correlation vector h _after_cal Using (k, fs, w), direction estimation processing is performed based on the phase difference of the reflected wave signals between the receiving antennas.

[0203] The direction estimation unit 404 calculates a direction estimation evaluation function value P HThe azimuth direction θ in (θ, k, fs, w) is varied within a specified angle range to calculate a spatial profile, and a specified number of maximum peaks in the calculated spatial profile are extracted in descending order, and the azimuth direction of the maximum peak is used as the estimated direction of arrival.

[0204] In addition, the evaluation function value P H There are various types of (θ, k, fs, w) depending on the arrival direction estimation algorithm. For example, the estimation method using an array antenna disclosed in Reference Non-Patent Document 1 may be used.

[0205] (Reference Non-Patent Document 1) Direction-of-arrival estimation using signal subspace modeling Cadzow, JA; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1 Publication Year: 1992, Page(s): 64-79

[0206] For example, the beamformer method can be expressed as the following equations (38) and (39). Other methods such as Capon and MUSIC can also be applied in a similar manner.

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[0207] Here, the superscript H is the Hermitian transpose operator. H (θ u ) is the azimuth direction θ u The direction vector of the virtual receiving array for the incoming wave of θ u is obtained by changing the azimuth range for estimating the direction of arrival at a predetermined azimuth interval β1. For example, θ u is set as follows: θ u =θmin + uβ1, u=0,…, NU NU = floor [(θmax-θmin) / β1] + 1

[0208] Here, floor(x) is a function that returns the maximum integer value that does not exceed the real number x.

[0209] The configuration of the radar receiving unit 400 has been described above.

[0210] As described above, in this embodiment, in the radar device 20, which is a MIMO radar, by varying the phase variable pattern for the radar transmission signal transmitted from each transmitting antenna, it is possible to reduce mutual interference between the radar transmission signals transmitted by switching between multiple transmitting antennas.

[0211] Furthermore, in this embodiment, the mutual interference between radar transmission signals transmitted by switching between multiple transmission antennas can be reduced, and the time interval for switching between the multiple transmission antennas can be narrowed, thereby shortening the detection time.

[0212] An embodiment according to one aspect of the present disclosure has been described above.

[0213] The operations according to the above-described embodiment and the variations may be combined as appropriate.

[0214] [Other embodiments] (1) In the above-described embodiments, the radar devices 10 and 20 may include a circuit configuration for simply correcting the transmit DC offset. In this way, the radar devices 10 and 20 can further suppress an increase in the noise level caused by the transmit DC offset by using both the configuration for controlling the phase variable pattern described above and the configuration for correcting the transmit DC offset component.

[0215] (2) In the above embodiment, a case has been described in which a coded pulse radar is used. However, the present disclosure is also applicable to radar systems that use frequency-modulated pulse waves, such as chirp pulse radar.

[0216] (3) In the radar devices 10 and 20 shown in FIGS. 1, 5, 6 and 8, the radar transmitters 100 and 300 and the radar receivers 200 and 400 may be individually disposed in physically separate locations.

[0217] (4) Although not shown, the radar devices 10 and 20 each have, for example, a CPU (Central Processing Unit), a storage medium such as a ROM (Read Only Memory) that stores a control program, and a working memory such as a RAM (Random Access Memory). In this case, the functions of the above-mentioned components are realized by the CPU executing the control program.

[0218] Although various embodiments have been described above with reference to the drawings, it goes without saying that the present disclosure is not limited to such examples. It is clear that a person skilled in the art can come up with various modified or amended examples within the scope of the claims, and it is understood that these also naturally belong to the technical scope of the present disclosure. In addition, the components in the above embodiments may be arbitrarily combined within the scope of the disclosure.

[0219] In each of the above embodiments, the present disclosure has been described as an example configured using hardware, but the present disclosure can also be realized by software in cooperation with hardware.

[0220] Moreover, each functional block used in the description of each of the above embodiments is typically realized as an LSI, which is an integrated circuit. The integrated circuit may control each functional block used in the description of the above embodiments and may have input terminals and output terminals. These may be individually integrated into one chip, or may be integrated into one chip that includes some or all of them. Here, the term LSI is used, but depending on the degree of integration, it may also be called an IC, a system LSI, a super LSI, or an ultra LSI.

[0221] The method of integration is not limited to LSI, but may be realized using a dedicated circuit or a general-purpose processor. A field programmable gate array (FPGA) that can be programmed after LSI manufacturing, or a reconfigurable processor that can reconfigure the connections or settings of circuit cells inside the LSI may also be used.

[0222] Furthermore, if a new integrated circuit technology that can replace LSI appears due to the progress of semiconductor technology or a derivative technology, it is possible to integrate the functional blocks using that technology. The application of biotechnology is also a possibility.

[0223] <Summary of this disclosure> A radar device according to the present disclosure includes a radar transmitter that transmits a radar signal at a predetermined transmission period, and a radar receiver that receives a reflected wave signal obtained by reflecting the radar signal off a target, wherein the radar transmitter includes a phase rotation control unit that changes a pattern of a phase rotation amount to be imparted to the radar signal within a period corresponding to the transmission period multiple times Ne (Ne is an integer greater than 1) for each of the periods, the amount of phase rotation being calculated by dividing an integer multiple of 2π by Ne multiple times, and the pattern of the phase rotation amount is obtained by multiplying the amount of phase rotation by a code polarity of a predetermined code sequence, and a transmission phase rotation unit that imparts a first phase rotation to the radar signal in accordance with the pattern of the phase rotation amount, and the radar receiver includes a reception phase rotation unit that imparts a second phase rotation to the reflected wave signal in a direction opposite to the first phase rotation in accordance with the pattern of the phase rotation amount.

[0224] In the radar device disclosed herein, when the code polarity of the specified code sequence is +1, the phase change of the phase rotation amount is 0, and when the code polarity of the specified code sequence is -1, the phase change of the phase rotation amount is π.

[0225] In the radar device of the present disclosure, the predetermined code sequence is based on a pseudorandom code, an M-sequence code, or a Gold code.

[0226] In the radar device disclosed herein, when the radar signal is generated using complementary codes, the transmission phase rotation unit imparts the same amount of phase rotation within two transmission periods in which a pair of codes constituting the complementary code is transmitted, and imparts a different amount of phase rotation every two transmission periods.

[0227] In the radar devices of the present disclosure, the predetermined code sequence differs among the multiple radar devices.

[0228] In the radar device of the present disclosure, the radar transmitter includes a plurality of transmitting antennas, and the predetermined code sequence differs between the transmitting antennas.

[0229] A radar method disclosed herein changes, for each period, a pattern of a phase rotation amount to be imparted to a radar signal within a period equivalent to a transmission cycle of multiple Ne (Ne is an integer greater than 1), imparts a first phase rotation to the radar signal in accordance with the pattern of phase rotation amount, transmits the radar signal to which the first phase rotation has been imparted in the transmission cycle, receives a reflected wave signal resulting from the radar signal to which the first phase rotation has been imparted being reflected by a target, and imparts a second phase rotation in a direction opposite to the first phase rotation to the reflected wave signal in accordance with the pattern of phase rotation amount, wherein the amount of phase rotation is obtained by dividing an integer multiple of 2π by Ne multiple times, and the pattern of the amount of phase rotation is obtained by multiplying the amount of phase rotation by a code polarity of a predetermined code sequence. [Industrial Applicability]

[0230] The present disclosure is suitable for a radar device that detects a wide angle range. [Explanation of symbols]

[0231] 10,20 Radar equipment 100,300 Radar transmitter 101 Radar transmission signal generator 102 Code generator 103 Phase rotation control section 104 Transmission phase rotation unit 105 Modulation section 106 Transmitting radio section 107 Transmitting Antenna 111 Code storage unit 112 DA conversion section 200,400 Radar receiver 201 Receiving antenna 202 Receiving radio section 203 Signal Processing Section 204 AD conversion section 205 Correlation calculation unit 206 Receiving phase rotation unit 207 Coherent Integration Section 208 Doppler Analysis Unit 301 Switching control unit 302 Transmission Switching Unit 303 Transmitting array antenna section 401 Receiving array antenna section 402 Antenna system processing unit 403 Output switching unit 404 Direction estimation part

Claims

1. a transmission phase rotation unit that imparts to each of the plurality of radar signals a phase rotation amount calculated by dividing an integer multiple of 2π by Ne (Ne is an integer greater than 1) multiple times for each transmission period; a radar radio transmission unit that multiplexes and transmits the plurality of radar signals, to which the phase rotation amount has been imparted for each transmission period, via a plurality of transmission antennas; a radar reception radio unit that receives, via a plurality of receiving antennas, one or more reflected wave signals that are at least one of the multiplexed radar signals and that are reflected by a target; Equipped with The phase rotation amount is controlled such that the code polarity of the phase rotation amount is controlled for each period corresponding to the transmission cycle of the plurality of times Ne. Radar signal processing circuit.

2. When the code polarity that changes every period is controlled by +1, a phase of 0 is added to the phase rotation amount, When the code polarity that changes for each period is controlled by −1, a phase π is added to the phase rotation amount.

2. The radar signal processing circuit according to claim 1.

3. The code polarity controlled for each period is based on a predetermined code sequence.

2. The radar signal processing circuit according to claim 1.

4. The code polarity controlled for each period differs among the plurality of transmitting antennas.

2. The radar signal processing circuit according to claim 1.

5. The phase rotation amount differs among the plurality of transmitting antennas.

2. The radar signal processing circuit according to claim 1.

6. imparting a phase rotation amount to each of the plurality of radar signals, the phase rotation amount being calculated by dividing an integer multiple of 2π by Ne (Ne is an integer greater than 1) a plurality of times for each transmission period; multiplexing the plurality of radar signals to which the phase rotation amount has been imparted for each transmission period via a plurality of transmission antennas; receiving, via a plurality of receiving antennas, one or more reflected wave signals that are at least one of the multiplexed radar signals and that are reflected by a target; 1. A radar signal processing method, comprising: The phase rotation amount is controlled such that the code polarity of the phase rotation amount is controlled for each period corresponding to the transmission cycle of the plurality of times Ne. Radar signal processing method.

7. When the code polarity that changes every period is controlled by +1, a phase of 0 is added to the phase rotation amount, When the code polarity that changes for each period is controlled by −1, a phase π is added to the phase rotation amount.

7. A radar signal processing method according to claim 6.

8. The code polarity controlled for each period is based on a predetermined code sequence.

7. A radar signal processing method according to claim 6.

9. The code polarity controlled for each period differs among the plurality of transmitting antennas.

7. A radar signal processing method according to claim 6.

10. The phase rotation amount differs among the plurality of transmitting antennas.

7. A radar signal processing method according to claim 6.

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