Motor

Abstract

To provide a motor for improving motor torque.SOLUTION: A motor rotates and drives a rotor by arranging N-pole stator poles 12, 14, 15 and 16 and S-pole stator magnetic poles 11, 13, and 15 excited by unidirectional current alternately in the circumferential direction, and driving the stator magnetic poles capable of torque generation in the required direction, as a configuration in which magnets 1H, 1P, 1Q, and 1R are arranged, and each rotor magnetic pole utilizes the magnetic flux passing through the rotor magnetic poles adjacent in the circumferential direction to supply more magnetic flux by alternately arranging the N and S poles rotor magnetic poles in the circumferential direction and arranging permanent magnets between the N-pole rotor magnetic poles 1H, 1K, and 1M and the S-pole rotor magnetic poles 1G, 1J, and 1L in their polar direction.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] Due to global environmental issues, there is a movement to replace fossil fuels with natural energy. Gasoline engine drives are being replaced by motor drives, and the importance of motors and their drive devices is increasing. The present invention relates to main motors for electric vehicles (EVs), motors for home appliances, motors for industrial machinery, and their drive technologies. The present invention relates to motors with high torque, high efficiency, small size, light weight, low cost, etc.

[0002] In addition, since conventional motors and drive devices often have in common the conventional motor technology, conventional power elements, and conventional control technology, such as three-phase AC and sinusoidal voltage and current, the motor and drive device have sometimes been discussed separately. However, when pursuing new possibilities, they can sometimes be realized by closely combining a new motor, a new drive circuit, and a new control technology. The motor of the present invention cannot be driven by a commercially available three-phase inverter, but by combining the motor, drive circuit, and control, it achieves high torque, high efficiency, compact size, light weight, and low cost. [Background technology]

[0003] FIG. 63 shows an example of a cross-sectional view of a conventional three-phase switched reluctance motor. 639 is a stator, which is a three-phase stator and has six salient stator poles. 63B is a rotor shaft. 63A, 63F, etc. are rotor salient poles, which have a circumferential width of 30° and are arranged at four equal intervals around the circumference. 631 is an A-phase stator pole, which has an A-phase concentrated winding 637 wound around it as shown by the double line at the coil end. The current in each winding of this motor is unidirectional, and each winding is shown by a current symbol to indicate the direction of current flow. A mark with an X-shaped circle encircled carries A-phase current Ia flowing from the front side to the back side of the paper, and a mark with a black circle encircled carries A-phase current Ia flowing from the back side to the front side of the paper. Therefore, when current is flowing, the A-phase stator pole 631 becomes an S pole. Reference numeral 632 denotes an A / phase stator pole that is in a reverse phase relationship with the A / phase, and an A / phase concentrated winding 638 is wound as shown by the double line at the coil end. A phase current Ia is also applied to the A / phase winding, and the A / phase stator pole 632 becomes an N pole. By simultaneously exciting 631 and 632, the A phase magnetic flux component φa shown by arrow 63E passes from the bottom to the top of the page through stator pole 632, rotor pole 63F, rotor pole 63A, and stator pole 631, and φa makes a circuit through the back yoke of the stator. In the state of FIG. 63, a torque in the counterclockwise direction CCW is generated in the rotor.

[0004] Similarly, 633 is a B-phase stator pole, around which concentrated winding 63C is wound and through which B-phase current Ib is passed. 634 is a B / -phase stator pole, around which concentrated winding 63D is wound and through which B-phase current Ib is passed. The B-phase magnetic flux passing from stator pole 634 to 633 is φb. 635 is a C-phase stator pole, around which concentrated winding 63G is wound and through which C-phase current Ic is passed. 636 is a C / -phase stator pole, around which concentrated winding 63G is wound and through which C-phase current Ic is passed. The C-phase magnetic flux passing from stator pole 636 to 635 is φc. The circumferential width of each stator pole is 30°, and they are arranged at six equal intervals around the circumference. For ease of understanding, the names of the A phase and the like of each stator pole are indicated in parentheses, such as (A), on the outside of stator 639.

[0005] Next, the operation of the switched reluctance motor in Fig. 63 will be described. Regarding the rotational position of the rotor, the rotational position of the clockwise end of the A-phase stator pole 631 is defined as the starting point of the rotor. The rotor rotation angle θr is the rotational angle from this starting point to the CCW end of the rotor pole 63A, as shown in the figure.

[0006] Next, the operation of rotating the switched reluctance motor in Figure 63 in the CCW direction will be described. When the rotor rotation angle θr is between 0° and 30°, current Ia is passed through phases A and A / to generate CCW torque. When θr is between 30° and 60°, current Ib is passed through phases B and B / to generate CCW torque. When θr is between 60° and 90°, current Ic is passed through phases C and C / to generate CCW torque. These operations of phases A, B, and C are repeated four times to make one rotation of the rotor.

[0007] Next, an example of torque generated when a constant A-phase current Ia is applied to the A-phase and A / -phase of the switched reluctance motor of FIG. 63 is shown and described in FIG. 64. The horizontal axis is the rotor rotation angle θr, which ranges from -5° to 30°. The vertical axis is the relative value of the torque T. For example, in FIG. 63, when the A-phase current Ia is the continuous rated current, torque is generated as shown by the solid line in FIG. 64 from around θr=-5° before the rotor pole 63A faces the A-phase stator pole 631, a large torque is generated near θr=0°, and the torque gradually decreases after θr=15°. When the A-phase current Ia is twice the continuous rated current, as shown by the dashed line in FIG. 64, and when it is three times the continuous rated current, as shown by the dashed line in FIG. 64, the angle width in which the torque is generated is relatively decreased. The cause of this reduction in torque width is related to the distribution of leakage magnetic flux in the air gap between the stator poles and rotor poles and in the vicinity before and after it, as well as magnetic saturation of the stator teeth and rotor teeth.

[0008] The advantages of the conventional switched reluctance motor in Figure 63 include the fact that the rotor has a simple structure and is robust, making it easy to rotate at high speeds. In addition, it can be driven without using permanent magnets. Torque is generated by the attractive force, which is a reluctance force, and the drive algorithm is relatively simple. The stator windings are also concentrated on the salient poles, making them simple and easy to manufacture. And because expensive rare earth permanent magnets are not required, the motor system can be constructed at low cost.

[0009] Next, the problems of the conventional switched reluctance motor in FIG. 63 will be described. The first problem is that when a large torque is generated, magnetic saturation occurs in the stator teeth and rotor teeth, causing a decrease in the torque constant. The second problem is that since torque is generated sequentially by using 1 / 3 of the total windings, the utilization rate of the windings is low at 33%, which results in a relatively large winding resistance and large copper loss. The excitation burden for exciting the magnetic flux of each phase is also large. The motor tends to be large compared to other permanent magnet motors. The third problem is that when a large torque is generated, torque saturation occurs partially, resulting in a large inverter. The fourth problem is that, compared to other permanent magnet motors, the torque ripple tends to be large, and vibration and noise caused by fluctuations in the attractive force between the stator and rotor are large. [Prior art documents] [Patent documents]

[0010] [Patent Document 1] Patent No. 3157162 [Patent Document 2] Patent Publication No. 2020-025377 [Non-patent literature]

[0011] [Non-Patent Document 1] The Journal of the Industrial Applications Division of the Institute of Electrical Engineers of Japan, 2016, 3_36 (Formula 1) Summary of the Invention [Problem to be solved by the invention]

[0012] In this invention, a permanent magnet is used in the rotor to greatly increase the magnetic flux that can pass in the forward direction. In addition We propose a motor that is configured with rotor poles that can pass forward and at the same time have little magnetic flux passing in the reverse direction. As a result, the torque is increased, and the motor's efficiency is improved and it is made smaller. In addition, by using permanent magnets in the stator, we realize stator poles that greatly increase the magnetic flux that can pass in the forward direction, and at the same time, we realize stator poles that have little magnetic flux passing in the reverse direction. By increasing the passing magnetic flux on both the rotor side and the stator side, it is possible to achieve a large magnetic flux density in the air gap that exceeds 2 [T], thereby increasing the torque and reducing the copper loss of the stator winding. As a result, it is possible to make the motor smaller, lighter, and less costly. In addition, as a method of configuring the motor, we optimize the combination of the number of stator poles and the number of rotor poles, improve the utilization rate of the windings and drive transistors, and realize the miniaturization and cost reduction of the motor and inverter. In addition, we also propose technology to optimize the shape of each part and technology to combine magnetic materials. [Means for solving the problem]

[0013] The invention described in claim 1 includes Nps stator poles Ps arranged in the circumferential direction of a stator, slots SLs between the stator poles Ps, stator windings Ws arranged in the slots SLs and exciting the stator poles Ps, unidirectional drive circuits Dhv capable of driving a unidirectional current to each of the stator windings Ws, a plurality of N-pole rotor poles Prn arranged in the circumferential direction of a rotor, and a plurality of S-pole rotor poles Prs arranged alternately with the N-pole rotor poles Prn in the circumferential direction of the rotor, A common back yoke for the rotor, Rotor common The above The magnetic path MPrn of the soft magnetic material that is magnetically connected from the back yoke to each of the N-pole rotor magnetic poles Prn and the magnetic path MPrn of the soft magnetic material that is common to the rotor The abovea magnetic path MPrs of a soft magnetic material magnetically connected from a back yoke to each of the S-pole rotor poles Prs; and a permanent magnet PMrbi disposed between the magnetic paths MPrn and MPrs arranged in the circumferential direction and at the circumferential boundary between the N-pole rotor poles Prn and the S-pole rotor poles Prs such that the polarity and magnetic pole orientation of the N-pole rotor poles Prn and the S-pole rotor poles Prs are the same; A magnetic flux can smoothly pass from the back yoke common to the rotor to each of the N-pole rotor magnetic poles Prn and each of the S-pole rotor magnetic poles Prs, The sum of the number of the N-pole rotor poles Prn and the number of the S-pole rotor poles Prs, Npr, is greater than the number of the stator poles Ps, Nps, and each of the stator windings Ws The unidirectional drive circuit Dhv is One-way current is passed , or the current value is set to 0 The configuration of the driving motor. According to this configuration, a large magnetic flux can be applied to the rotor poles that are excited and acted upon, so that a large torque can be generated.

[0014] The invention described in claim 2 is a motor configuration in which, in claim 1, the stator poles Ps are arranged so that north poles and south poles alternate in the circumferential direction, with north-pole stator poles Psn acting as north poles, south-pole stator poles Pss arranged alternately with the north-pole stator poles Psn in the circumferential direction and acting as south poles, and permanent magnets PMsbi arranged between the north-pole stator poles Psn and the south-pole stator poles Pss aligned in the circumferential direction such that the polarity and magnetic pole orientation of both stator poles Psn and Pss coincide. According to this configuration, a large magnetic flux can be applied to the stator poles and rotor poles Prn, Prs that are excited and acting, so that a large torque can be generated.

[0015] The invention described in claim 3 is the same as claim 1, Tamaki The wire Ws is a motor configuration of concentrated windings Wscp that excite each of the stator poles Ps. According to this configuration, each stator pole winding Ws is little affected by the control state of the other stator poles, and can be freely excited to drive the rotor.

[0016] The invention described in claim 4 is the same as claim 1, Tamaki The wire Ws is a motor configuration of a full-pitch stator winding Wsfp with a winding pitch that is approximately 1 / 2 the stator pole pair period. With this configuration, if the excitation current component of the acting stator pole and the excitation current component of the circumferentially adjacent stator pole are controlled so that they become current components in the same direction and do not overlap, the copper loss in the slot can be reduced to approximately half.

[0017] The invention described in claim 5 is the motor configuration of claim 1, including Nps number of the stator poles Ps, where Nps=2+4×Ns, and a total of Npr number of the N-pole rotor poles Prn and S-pole rotor poles Prs, where Npr=2+4×Nr, where Ns and Nr are integers equal to or greater than 1. With this configuration, the north stator poles Psn, south stator poles Pss, each stator winding, north rotor poles Prn, and south rotor poles Prs can be evenly arranged in the circumferential direction, resulting in high torque generation efficiency and easy production of the motor.

[0018] The invention described in claim 6 is the same as that described in claim 1, wherein the number of phases of the plurality of stator poles Ps is Nph, and the N poles are arranged alternately in the circumferential direction of the rotor. Rotor Magnetic pole Prn and The above S pole Rotor The rotor pole pitch of the magnetic poles Prs is θppr, and the motor is configured with Nph stator poles that are out of phase with the rotor poles by (2×θppr) / Nph, partially arranged in the circumferential direction of the stator, where Nph is an integer of 2 or greater. This configuration allows for two-phase, three-phase, etc., motor configurations that do not require uniform circumferential arrangement to be realized, and allows for the desired specific characteristics to be obtained.

[0019] The invention described in claim 7 is the same as that described in claim 1, wherein the circumferential length of the magnetic poles of the stator poles Ps facing the air gap portion is Lsg, and the teeth of the stator poles Ps are Outermost diameter The circumferential width of The above This is a motor configuration with a value 20% or more greater than Lsg. According to this configuration, the restriction of the magnetic flux passing through the stator poles can be reduced, and therefore the motor torque can be increased.

[0020] The invention described in claim 8 is the same as that described in claim 1, wherein the N pole of the stator pole Ps Stator Magnetic poles Psn and S pole Stator This motor configuration includes a permanent magnet PMssur that is arranged near the air gap of the magnetic pole Pss so that the polarity of the stator pole matches the polarity of the magnetic pole Pss. According to this configuration, the burden of exciting the magnetic flux can be reduced, i.e., the reactive current that excites the magnetic flux can be reduced, and the adverse effects of voltage caused by the flow of magnetic energy between the power supply and the motor can be reduced.

[0021] The invention described in claim 9 is the same as claim 4, and further comprises: stator poles Ps1, Ps2, Ps3, Ps4, and Ps5 arranged in the circumferential direction, a slot SLs1 located between the stator poles Ps1 and Ps2, a slot SLs2 located between the stator poles Ps2 and Ps3, a slot SLs3 located between the stator poles Ps3 and Ps4, and a slot SLs4 located between the stator poles Ps4 and Ps5; and a full-pitch winding wound between two slots spaced apart by approximately 1 / 2 the pole pair period of the stator, the full-pitch winding Wsfp1 being arranged in the slot SLs1, a full-pitch winding Wsfp2 being arranged in the slot SLs2, and a full-pitch winding Wsfp3 being arranged in the slot SLs3. Similarly, a full-pitch winding Wsfp4 is arranged in the slot SLs4, a rotor having Nkb×N2 or more rotor poles of N poles and S poles arranged alternately on the circumference, and a A part of the unidirectional driving circuit Dhv A transistor TR1 and a full-pitch winding Wsfp2 are connected in series. A part of the unidirectional driving circuit Dhv A transistor TR2 is connected in series with the full-pitch winding Wsfp3. A part of the unidirectional driving circuit Dhv A transistor TR3 is connected in series with the full-pitch winding Wsfp4. A part of the unidirectional driving circuit Dhvand a transistor TR4, the transistor TR1 being connected to the full-pitch winding Wsfp1. One-way current The full-pitch winding Wsfp1, the full-pitch winding Wsfp2 and the transistor TR2 are connected in series, and the transistor TR2 is connected to the full-pitch winding Wsfp2. One-way current The full-pitch winding Wsfp2, the full-pitch winding Wsfp3 and the transistor TR3 are connected in series, and the transistor TR3 is connected to the full-pitch winding Wsfp3. One-way current The full-pitch winding Wsfp3 is energized, the full-pitch winding Wsfp4 is connected in series with the transistor TR4, and the transistor TR4 is connected to the full-pitch winding Wsfp4. One-way current When the full-pitch winding Wsfp3 and the full-pitch winding Wsfp1 are connected in parallel and connected to the transistor TR4, the full-pitch winding Wsfp4 is energized, and an excitation current is applied to each of the full-pitch windings and each of the transistors TR1, TR2, TR3, and TR4 connected in series to excite each of the stator poles Ps1, Ps2, Ps3, Ps4, and Ps5. When the number of full-pitch windings of the motor is three, the full-pitch winding Wsfp1 and the full-pitch winding Wsfp4 are the same winding, and the full-pitch winding Wsfp3 and the full-pitch winding Wsfp1 are arranged in parallel and connected to the transistor TR4. One-way current The motor is configured to pass current through the Here, Nkb is the number of pole pairs of the stator and is an integer of 1 or more, N1 is an integer of 6 or more, and N2 is an integer of 6 or more. With this configuration, it is possible to accurately excite a specific stator pole with its excitation current component, without that excitation current component affecting the stator poles of other phases. In addition, the interlinkage magnetic flux of other phases is offset by two windings connected in series, minimizing the effect of the magnetic flux of other phases, and the current can be passed and controlled.

[0022] The invention described in claim 10 is the same as that described in claim 1, wherein a component of a magnetic flux excitation current corresponding to an operating state is continuously applied to each phase winding of the stator winding Ws, or each slot of the stator is continuously applied to each phase winding of the stator winding Ws. In addition to the stator winding Ws Flux excitation winding Add Roll it up , the flux excitation winding of each slot is This is a motor configuration in which the rotors are connected in series and a magnetic flux excitation current is passed through them. With this configuration, the flow of magnetic energy between the power supply and the motor is automatic by continuously passing a constant current, and in particular, large magnetic flux changes, i.e., overvoltage, that occur when magnetic energy is regenerated can be reduced, thereby reducing the adverse effects on current control of other phases.

[0023] The invention described in claim 11 is the same as the invention described in claim 1, further comprising a DC power source POS2, a DC power source POS3 arranged in series with the DC power source POS2, an intermediate potential portion TYV between the DC power sources POS2 and POS3, and a power supply connected to the DC power source POS2. A part of the unidirectional driving circuit Dhv A transistor TR7 and a transistor arranged between the transistor TR7 and the intermediate potential portion TYV. Stator The winding Ws2 is connected to the DC power supply POS3. A part of the unidirectional driving circuit Dhv A transistor TR8 and a transistor arranged between the transistor TR8 and the intermediate potential portion TYV. Stator Equipped with winding Ws3 Similarly to the stator windings Ws2 and Ws3, before Note The theta winding Ws is supplied with the DC power supply POS2 and the DC power supply POS3. The one-way This is the configuration of the motor through which current flows. According to this configuration, one direct current can be driven by one transistor, which is effective in terms of space and cost, particularly when the number of currents to be controlled is large.

[0024] In addition, other technologies includeThe motor is configured to include a reverse drive circuit Drhv, and a negative unidirectional current component is added by the reverse drive circuit Drhv to the positive unidirectional current component of the stator winding Ws caused by the unidirectional drive circuit Dhv, and the circumferential width of each of the stator poles Ps facing the air gap surface is 1 / 2 the circumferential pitch of each of the stator poles Ps, and the back yokes of the N-pole rotor poles Prn and the S-pole rotor poles Prs are magnetically connected, and when the current of the stator is not flowing, the magnetic flux generated by the permanent magnet PMrbi passes through the N-pole and S-pole of the permanent magnet PMrbi and the soft magnetic material part of the back yoke of the rotor and circulates within the rotor, and magnetic flux required for generating torque is generated in the N-pole rotor poles Prn and the S-pole rotor poles Prs facing each other via the air gap surface in accordance with the current of the stator. This configuration allows current to flow in both positive and negative directions, doubling the opportunities for torque generation, increasing motor torque and improving motor efficiency.

[0025] The invention described in claim 12 is the same as claim 1, When exciting multiple circumferentially adjacent stator poles Ps with a full-pitch stator winding Wsfp, a current component Isfpv1 is passed through a full-pitch winding Wsfpv1 arranged in a slot Slsv adjacent to one stator pole Psv1 in the circumferential direction, and a part or all of a current component of a negative value (-Isfpv1) of the current component Isfpv1 is passed through one or more full-pitch windings WsfpvN arranged in slots two or more away in the opposite direction from the slot Slsv relative to the stator pole Psv1. According to this configuration, it operates as a vernier motor, reducing copper loss in the motor and increasing efficiency.

[0026] Claim 13 The invention described in claim 12In this motor configuration, during low speed rotation, the stator pole PsvN is excited by a current component Isfpv1 that excites the stator pole PsvN and the negative value (-Isfpv1) of the current component Isfpv1 of one or more full-pitch windings WsfpvN arranged in slots two or more apart in the circumferential direction, and during high speed rotation, the full-pitch windings WsfpvF and WsfpvR on either side of the stator pole PsvN in the circumferential direction are connected in series to pass the current component IsvN that excites the stator pole PsvN. With this configuration, at low speeds, the motor rotates and drives as a vernier motor with high torque and high efficiency characteristics, and at high speeds, each stator pole is excited and driven individually, with little effect from the magnetic flux of other phases and large torque being output.

[0027] Claim 14 The invention described in claim 1 is a motor configuration in which the main magnetic circuit of the rotor is constructed from a soft magnetic material MagA, and a soft magnetic material MagB having a higher saturation magnetic flux density than the soft magnetic material MagA is used near the air gap portion of the north pole rotor magnetic pole Prn and the south pole rotor magnetic pole Prs of the rotor. With this configuration, a motor is constructed by combining the features of multiple types of soft magnetic materials and utilizing them more effectively.For example, by using an amorphous magnetic steel sheet with low iron loss but low saturation magnetic flux density as the soft magnetic material MagA, and a permendur magnetic steel sheet with high saturation magnetic flux density but high cost and high iron loss as the soft magnetic material MagB, a motor can be realized that can operate efficiently up to high speed rotation and has a large maximum torque. Effect of the Invention

[0028] The present invention proposes new rotor poles that utilize permanent magnets, and new stator poles that utilize permanent magnets, which increases the magnetic flux density in the air gap, increases motor torque, and reduces copper loss in the stator, thereby enabling the motor to be made smaller, lighter, and less costly. [Brief description of the drawings]

[0029] [Figure 1] Cross-sectional view of the motor of the present invention [Diagram 2] Cross-sectional view showing magnetic flux components [Diagram 3] Cross-sectional view showing magnetic flux components [Figure 4] Cross-sectional view showing magnetic flux components [Diagram 5] Cross-sectional view showing magnetic flux components [Figure 6] Excitation current and magnetic flux density of soft magnetic material [Figure 7] Linear development of cross section [Figure 8] Linear development of cross section [Figure 9] Linear development of cross section [Figure 10] Linear development of cross section [Figure 11] Linear development of cross section [Figure 12] Linear development diagram showing operation [Figure 13] Current Waveform Example [Figure 14] Cross-sectional view of the motor of the present invention [Figure 15] Linear development of cross section [Figure 16] Linear development of cross section [Figure 17] Linear development of cross section [Figure 18] Linear development of cross section [Figure 19] Linear development of cross section [Figure 20] Enlarged cross-section [Figure 21] Cross-section of a rotor with 40 rotor poles [Figure 22] Example of an enlarged view of a rotor cross section [Figure 23] Cross-sectional view of the motor of the present invention [Figure 24] Current and voltage waveforms [Diagram 25] Example of a unidirectional current drive circuit [Figure 26] Cross-section of a full-pitch winding motor [Figure 27] Cross section of a full-pitch winding, two-pole pair motor [Figure 28] Examples of current and voltage waveforms [Figure 29] Example of a drive circuit for a three-phase full-pitch motor [Diagram 30] Compound motor with double inner and outer diameters [Diagram 31] Linear development diagram showing operation [Diagram 32] Cross-sectional view of the motor of the present invention [Diagram 33] Linear expansion diagram showing 7-phase operation [Diagram 34] Cross-sectional view of a two-pole motor according to the present invention [Diagram 35] Example of a drive circuit for a 7-phase full-pitch motor [Diagram 36] Example of current waveforms for a 7-phase full-pitch motor [Figure 37] Linear development diagram showing operation [Figure 38] Cross-sectional view of a 5-phase motor according to the present invention [Figure 39] Linear development diagram showing operation [Diagram 40] Example of current waveforms for a five-phase full-pitch motor [Diagram 41] Linear development diagram showing operation [Diagram 42] Cross-sectional view of a two-phase motor according to the present invention [Diagram 43] Linear development diagram showing operation [Diagram 44] Enlarged view of the shapes of the stator poles and rotor poles facing the air gap [Diagram 45] Current waveform of a two-phase concentrated winding motor [Figure 46] Example of a drive circuit for a two-phase concentrated winding motor [Figure 47] Enlarged view of the stator tooth shape [Figure 48] Example of stator tooth tip shape [Figure 49] Example of magnetomotive force H and magnetic flux density B of a permanent magnet [Figure 50] Example of a drive circuit for a 7-phase full-pitch motor [Figure 51] Example of a drive circuit for a three-phase full-pitch motor [Figure 52] Example of three-phase current and voltage waveforms [Diagram 53]Example of a unidirectional current driver with two power supplies [Figure 54] Flywheel current drive circuit [Figure 55] Example of a drive circuit with a reverse drive circuit added [Figure 56] Example of a 5-phase motor drive circuit [Figure 57] Cross-sectional view of a 5-phase motor according to the present invention [Figure 58] Linear expansion diagram showing 5-phase operation [Figure 59] Linear expansion diagram showing 5-phase operation [Figure 60] A drive circuit that supplies bidirectional and unidirectional current to a five-phase full-pitch motor [Figure 61] An example of placing soft magnetic materials with different properties on the tips of the stator and rotor poles [Figure 62] Example of a vertical cross section showing a motor structure when using extremely thin electromagnetic steel sheets [Figure 63] Cross-section of a conventional switched reluctance motor [Figure 64] Torque characteristic example DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS EXAMPLES

[0030] FIG. 1 shows an embodiment of claim 1 of the present invention. It is a cross-sectional view of a motor. 17 is a stator, and its outer circular peripheral part is the back yoke of the stator. 11 is an A-phase stator pole, and 1A is an A-phase winding. This A-phase winding 1A is a concentrated winding, and its coil end part is symbolically shown with double lines. The current in each winding of this motor is a unidirectional current, and each winding is shown with a current symbol to show the direction of current flow. A mark with an X-shaped circle surrounding it carries the A-phase current Ia flowing from the front side to the back side of the paper, and a mark with a black circle surrounded by a circle carries the A-phase current Ia flowing from the back side to the front side of the paper. Therefore, the A-phase stator pole 11 becomes an S pole when the A-phase current Ia is carried. 14 is an A / phase stator pole, and 1D of the concentrated A / phase winding, whose coil end part is shown with double lines, is wound around it. The A-phase current Ia, which is a unidirectional current, is passed through this A / phase winding 1D, and the A / phase stator pole 14 becomes an N pole. Normally, the same A-phase current Ia is passed through the A / phase winding 1A and the A / phase winding 1D, and an A-phase magnetic flux φa is generated between the A-phase stator pole 11 and the A / phase stator pole 14. This A-phase magnetic flux φa circulates through the back yoke of the stator.

[0031] Similarly, 13 is a B-phase stator pole, around which concentrated winding winding 1C is wound and which carries B-phase current Ib, which is a unidirectional current. B-phase stator pole 13 becomes an S pole when B-phase current Ib is carried. 16 is a B / -phase stator pole, around which concentrated winding B-phase winding 1F is wound. B-phase current Ib is carried to this B / -phase winding 1F and B / -phase stator pole 16 becomes an N pole. The same B-phase current Ib is carried to B-phase winding 1C and B / -phase winding 1F, generating B-phase magnetic flux φb between B-phase stator pole 13 and B / -phase stator pole 16. This B-phase magnetic flux φb circulates through the back yoke of the stator.

[0032] Similarly, 15 is the stator pole of phase C, around which the concentrated winding phase C winding 1E is wound, and the phase C current Ic, which is a unidirectional current, is passed through. When the phase C current Ic is passed through the stator pole 15 of phase C, it becomes the S pole. 12 is the stator pole of phase C / , around which the concentrated winding phase C / winding 1B is wound. The phase C current Ic is passed through this phase C / winding 1B, and the stator pole 12 of phase C / becomes the N pole. The same phase C current Ic is passed through the phase C winding 1E and the phase C / winding 1B, and a phase C magnetic flux φc is generated between the stator pole 15 of phase C and the stator pole 12 of phase B / . This phase C magnetic flux φc circulates through the back yoke of the stator. For the sake of clarity, (A), (A / ), (B), (B / ), (C), (C / ) are appended near the outer periphery of the motor in Fig. 1 to indicate the positions of the respective stator poles.

[0033] 1S is the rotor shaft. 1H is the N - pole magnet of the rotor made of a soft magnetic material. 1L is the S - pole magnet located on the opposite side, 180° away from the rotor N - pole magnet 1H. 1J is the S - pole magnet of the rotor made of a soft magnetic material. 1M is the N - pole magnet located on the opposite side, 180° away from the rotor S - pole magnet 1J. The two rotor magnets 180° apart have opposite polarities. However, in terms of shape, they are symmetric with respect to the center of the rotor. Then, the rotor N - pole magnets and the rotor S - pole magnets are alternately arranged in the circumferential direction, and a total of 10 rotor magnets are arranged. Also, the circumferential width of the opening of the stator pole and the slot on the air - gap surface is 30° as an example. The circumferential width of the soft magnetic materials such as the rotor magnets 1G, 1H, 1J is 30° as an example. The circumferential width of the air - gap surface of the part where the permanent magnets 1N, 1P, etc. are arranged between the rotor magnets is 6° as an example.

[0034] A permanent magnet 1N with a polarity facing the rotor magnetic pole is placed between the rotor south pole 1G and the rotor north pole 1H. When no magnetomotive force from the stator side is applied, the magnetic flux of the permanent magnet 1N is generated as shown by the dashed line with an arrow. A permanent magnet 1P with a polarity facing the rotor magnetic pole is placed between the rotor north pole 1H and the rotor south pole 1J. When no magnetomotive force is applied from the stator side, the magnetic flux of the permanent magnet 1N is generated as shown by the dashed line with an arrow. The area surrounded by the rectangular line 1T and its vicinity is the soft magnetic material part of the rotor north pole 1H, and as described above, in the state of FIG. 1, the magnetic flux of the permanent magnet shown by the dashed line with an arrow passes through it. As will be explained later, the magnetic flux of this soft magnetic material part changes in various ways depending on the rotor rotation position θr and the value of each current of the stator. A permanent magnet 1Q with a polarity facing the rotor magnetic pole is placed between the rotor north pole 1K and the rotor south pole 1L. When no magnetomotive force is applied from the stator side, the magnetic flux of the permanent magnet 1Q is generated as shown by the dashed line with an arrow. The permanent magnet 1R, whose polarity faces the rotor magnetic pole, is placed between the rotor south magnetic pole 1L and the rotor north magnetic pole 1M. When no magnetomotive force is applied from the stator side, the magnetic flux of the permanent magnet 1R is generated as shown by the dashed line with an arrow. The other six permanent magnets are also placed at the boundary between the rotor north magnetic pole and the rotor south magnetic pole, and have the same characteristics. The polarity direction of each permanent magnet is indicated by a small arrow in the center of the permanent magnets 1N, 1P, 1Q, and 1R. In addition, when the magnetic flux of each of these permanent magnets is large, a part of the magnetic flux of each permanent magnet passes through the air gap side and the stator side outside the rotor, but this is omitted here and is not shown.

[0035] Here, the description rules in the specification of the present invention are defined and described. When the number of stator poles Ps is Nps, and the total number of the number of rotor north poles Prn and the number of rotor south poles Prs is Npr, the motor model is called (Nps)S(Npr)R. For example, the motor in FIG. 1 is a 6S10R. Regarding the definition of the motor's electrical angle, in conventional IPMSM and SPMSM, the circumferential width of the rotor poles, consisting of north and south poles, is defined as an electrical angle of 360°. However, as in the conventional switched reluctance motor in FIG. 63, the stator poles are configured to be separated in the circumferential direction, and the rotor poles are also configured to be separated in the circumferential direction, so that the conventional electrical angle may not be sufficient to explain the electromagnetic relationship. Similarly, in the motor of the present invention such as FIG. 1, the definition may not be sufficient, so that in the specification of the present invention, the period of one pole pair of the stator is defined as an electrical angle of 360° and described. For example, in the motor of FIG. 1, when the number of stator poles Ps is 6, and the total number of rotor poles Prn and rotor poles Prs is 10, the total number of rotor poles is 10, the total electrical angle is 360° and the mechanical angle is 360°. Six stator poles are called one stator pole pair, and the electrical angle of the stator is determined based on the stator. Also, for example, when two pairs of the motor configuration of FIG. 1 are arranged in the circumferential direction, when the number of stator poles Ps is 12 and the total number of rotor poles is 20, the total electrical angle is 720° and the mechanical angle is 360°. Note that the stator pole pitch θpps, rotor pole pitch θppr, etc. will be explained later by expressing the electrical angle [°].

[0036] Regarding the definition of the magnetic pole pairs of the motor, in the conventional IPMSM and SPMSM, the circumferential width of two rotor poles, an N pole and an S pole, is set to an electrical angle of 360°, and one magnetic pole pair is defined between the width. However, in the motor of the present invention shown in FIG. 1 and the like, there is a problem that the basic motor configuration does not fit within the circumferential width of two rotor poles, an N pole and an S pole of the rotor. To address this problem, the motor configuration is defined and shown by the "number of stator magnetic pole pairs Nkb" so that all stator poles are included. For example, in the motor shown in FIG. 1, in the case of a configuration in which the number of stator magnetic poles Ps is Nps=6 and the number of rotor magnetic poles is Npr=10, the number of stator magnetic pole pairs Nkb=1. In addition, in the case of a configuration in which two motor configurations shown in FIG. 1 are arranged in the circumferential direction, in the case of a configuration in which the number of stator magnetic poles Ps is Nps=12 and the number of rotor magnetic poles is Npr=20, the number of stator magnetic pole pairs Nkb=2. In addition, other patents and documents may use the rotor pole pitch as a standard, so we will confirm this here to avoid confusion in the definition of the motor configuration.

[0037] Regarding the number of phases of a motor, in the case of a point-symmetrical configuration with respect to the rotor center point, as in the configuration of Figure 1, the stator poles of a certain phase and the stator poles 180° opposite to it are counted as one phase. In the case of the 6S10R motor configuration of Figure 1, it will also be called a three-phase motor. However, as will be explained later, the motor in Figure 1 is a motor that passes a unidirectional current and is different from a conventional three-phase AC motor. There are also motors with an odd number of stator poles and a configuration that is not point-symmetrical with respect to the rotor center point, in which case they will be called by the motor model of (Nps)R(Npr)S mentioned above. In addition, a toroidal winding that is wound from the stator slot through the outside of the back yoke is almost equivalent to a full-pitch winding when two toroidal windings separated by 180° in electrical angle are connected in series, so the explanation of the full-pitch winding will also apply to the toroidal winding.

[0038] In addition, the characters used in the description of the present invention are treated as the same characters, full-width characters and half-width characters, and are not distinguished. Uppercase and lowercase characters are treated as different characters. The operation of the motor is mainly described using a development diagram in which the stator magnetic pole shape and rotor magnetic pole shape facing the air gap part of the motor are linearly developed. For the sake of simplicity, the resistance value of the stator winding is ignored and described as 0 [Ω]. The magnitude of the magnetic flux is described assuming that the magnetic flux passes between the stator magnetic pole and the rotor magnetic pole only through the part where the stator magnetic pole and the rotor magnetic pole face each other. The magnetic characteristics of the soft magnetic material are treated as simplified characteristics as shown in FIG. 6, which will be shown later. When the electromagnetic steel sheet is a normal silicon steel sheet, the maximum magnetic flux density is assumed to be 2 [T], and a simplified calculation is performed and shown. The magnetic resistance of the soft magnetic material such as the back yoke part is ignored and a fundamental model is described. However, when the magnetic flux density of the air gap exceeds 2.0 [T], the state of a part near the air gap will be specially explained separately from the characteristics of FIG. 6. The magnetic resistance [A / Wb] of the air gap is also included in the characteristics of FIG. 6, but the regenerative operation of magnetic energy, which is one of the important issues of the motor of the present invention, will be explained separately. In this way, when roughly evaluating a motor using a model, problems can be confirmed with a simplified model, countermeasures for the problems can be proposed, and the motor can be qualitatively evaluated and explained. However, of course, these simplifications cannot be ignored when evaluating, examining, and designing a motor accurately. And, to evaluate the magnetic flux density, torque, voltage, etc. of each part of the motor accurately, electromagnetic field analysis using the finite element method (FEM) using a personal computer is necessary.

[0039] Next, an example of operation when current is passed through the windings of the motor in Fig. 1 will be described with reference to Fig. 2 and Fig. 3. Fig. 2 shows a state in which the A-phase current Ia is passed through the A-phase winding 1A and the A / -phase winding 1D with the A-phase stator pole 11 facing the rotor N-pole pole 1H and the A / -phase stator pole 14 facing the rotor S-pole pole 1L. The A-phase current Ia excites 21 and 22 indicating the A-phase magnetic flux component φa. In the area enclosed by square lines 23 and 24, the A-phase magnetic flux component φa 21 and 22 are shown overlapping with the magnetic flux components of the permanent magnets 1N, 1P, 1Q, and 1R shown by dashed lines. At this time, the magnetic flux direction of the A-phase magnetic flux component φa is opposite to that of the permanent magnets 1N, 1P, 1Q, and 1R, and the two magnetic fluxes cancel each other out in the regions enclosed by the rectangular lines 23 and 24. Therefore, in the soft magnetic material in the regions 23 and 24, the magnetic flux density component from the bottom to the top of the paper has a small value.

[0040] FIG. 3 is a diagram in which the magnetic flux distribution shown in FIG. 2 as two magnetic flux components overlap is rewritten to an actual magnetic flux distribution. Therefore, FIG. 2 and FIG. 3 show the same magnetic flux distribution state. Specifically, the overlap of two magnetic flux components in the area surrounded by the square lines 35 and 36 in FIG. 3 is eliminated. The magnetic flux components shown by 31, 32, 33, and 34 in FIG. 3 pass from the A / phase stator north pole magnetic pole 14 to the rotor south pole magnetic pole 1L, and from the rotor north pole magnetic pole 1H to the A-phase stator south pole magnetic pole 11. These magnetic fluxes make a circuit through the back yoke of the stator. Inside the rotor, magnetic flux can pass through any position without difficulty. However, near the tips of the rotor south pole magnetic pole 1L and the rotor north pole magnetic pole 1H, i.e., near the air gap, magnetic flux density increases because magnetic flux from three directions is concentrated.

[0041] Here, let us consider the distribution of magnetic flux inside the rotor in Fig. 3 and the way in which the magnetic flux passes. Of the magnetic flux components 31, 32, 33, and 34 that pass through the rotor north pole 1H, the magnetic flux components 31 and 32 pass through the soft magnetic magnetic path of the rotor south pole 1L and the soft magnetic magnetic path of the rotor north pole 1H. However, the magnetic flux 33 uses and passes through the soft magnetic magnetic path of the rotor south pole 1J, which is not used in the state of Fig. 3, due to the action of the permanent magnet 1P, and is led to the air gap surface of the rotor north pole 1H. The same is true for the magnetic flux 34, which uses and passes through the soft magnetic magnetic path of the rotor south pole 1G, which is not used in the state of Fig. 3, due to the action of the permanent magnet 1N, and is led to the air gap surface of the rotor north pole 1H.

[0042] Therefore, in the rotor configuration of FIG. 3, the magnetic flux of the rotor pole that generates torque by excitation also passes through the soft magnetic magnetic path of the adjacent rotor pole in the circumferential direction, so it can be said that a larger magnetic flux can pass through the rotor N pole 1H. However, if the air gap vicinity exceeds 2 [T], the relative permeability decreases even in the soft magnetic material part, so the burden of exciting the air gap and the soft magnetic material part in the vicinity increases. Since a large magnetic flux can pass through the inside of the rotor and the back yoke part of the rotor, the excitation burden does not become excessive. Compared to the conventional motor of FIG. 63, the magnetic resistance of the rotor is small, so the torque can be increased. Note that the size of the magnetic flux that can pass through the stator poles 11 and 14 in FIG. 1, 2, and 3 is limited by their shape and configuration, so some further improvements that will be explained later are possible.

[0043] As a result, in the rotor configuration of FIG. 3, the magnetic flux density on the rotor pole surface may be increased to a value exceeding 2 [T]. As will be explained later with reference to equation (19) and the like, the motor output torque can be increased by increasing the magnetic flux density in the air gap. The configuration and method for increasing the magnetic flux density in the air gap to exceed 2 [T] will be explained later. The rotor S pole 1L in FIG. 3 also acts in the same way as the magnetic flux passing through the rotor N pole 1H. In addition, in the rotor rotation positions of FIGS. 1, 2, and 3, the A-phase stator pole 11 and the A / phase stator pole 14 cannot generate torque. In this rotor rotation position, the stator poles 11 and 14 can pass the largest magnetic flux. As will be explained later, the magnitude of the magnetic flux that can pass is approximately proportional to the magnitude of the output torque.

[0044] Next, Fig. 4 shows and explains the characteristics when the stator poles Ps and the rotor poles Pr have the same polarity and face each other through an air gap. In this specification, the rotor rotation angle θr=0° is defined as the rotor rotation position θr immediately before the A-phase stator S-pole 11 electromagnetically acts on the rotor N-pole 1H to generate CCW torque. Specifically, in the paper of Fig. 4, the rotor rotation position where the CCW corner of the rotor N-pole 1H approaches the lower right corner of the A-phase stator S-pole 11 is defined as θr=0°. Fig. 5 illustrates an example where the rotor rotation position θr is 12°. The rotor rotation position in Fig. 4 is θr=-6°, which is shifted from the position of θr=0° by the circumferential width of 6° of the permanent magnet 1P. The rotor rotation angle in Figures 1, 2 and 3 is a position rotated in the CCW direction by the circumferential width of the stator pole, 30°, from the rotor rotation position of θr=0°, and is the rotor rotation angle θr=30°.

[0045] In Fig. 4, when the A-phase current Ia is applied to the A-phase winding 1A and the A / -phase winding 1D, a magnetic flux component indicated by a dashed line 45 is generated. However, in the region enclosed by a square line 46, the magnetic flux component of the permanent magnet indicated by a dashed line with an arrow is directed from the bottom to the top of the page, and if the magnetic flux generated by the permanent magnets 43 and 1P is sufficiently large, the direction of the magnetic flux component 45 is the same, so the magnetic resistance through which the magnetic flux passes is large. The same is true in the region enclosed by a square line 47, where the magnetic resistance through which the magnetic flux 45 passes is large due to the action of the permanent magnets 1R and 44. Therefore, the magnetic flux component 45 in Fig. 4 is suppressed to a relatively small value.

[0046] The extent to which the magnetic resistance increases with respect to the magnetic flux moving from the bottom to the top on the paper surface of Fig. 4 varies depending on the characteristics of the permanent magnet and the shape of each part. Also, the above-mentioned action of increasing magnetic resistance in Fig. 4 is the opposite action to the action of decreasing magnetic resistance in the area surrounded by the square lines 23 and 24 in Fig. 2. Also, Fig. 4 shows the rotor rotation position θr where the torque that the A-phase stator poles 11 and A / phase stator poles 14 can generate is just 0 [Nm], and the following explanation will be given assuming that the rotor rotation position in the state of Fig. 4 is θr = -6 [°]. In Fig. 1, the rotor rotation position θr = 30 [°].

[0047] Next, an example of a state in which a torque T in a counterclockwise direction CCW is generated is shown and described in FIG. 5. The rotor rotation position θr is indicated by an arrow in FIG. 5. θr=12[°], which is a position where the rotor of θr=-6[°] in FIG. 4 is rotated 16° in the counterclockwise direction CCW. In FIG. 5, when the A-phase current Ia is applied to the A-phase winding 1A and the A / -phase winding 1D, a magnetic flux component indicated by a thick solid line 53 is generated. The magnetic flux component 53 is the magnetic flux component shown in FIG. 2 and FIG. 3, and has a large magnetic flux density at the air gap surface because the magnetic resistance in the rotor is small. At this time, a magnetic flux component indicated by a thin broken line 54 is also generated in parallel. The magnetic flux component 54 is the magnetic flux component shown in FIG. 4, and has a relatively small magnetic flux density at the air gap surface because the magnetic resistance in the rotor is large. As a result, the magnetic flux density at the air gap surface created by the magnetic flux component 53 becomes superior, and a torque T [T] in the counterclockwise direction CCW is generated.

[0048] Here, the soft magnetic materials of the stator and rotor are described in this specification with reference to an example of the magnetic properties of the soft magnetic materials shown in FIG. 6. The magnetic properties of electromagnetic steel sheets such as silicon steel sheets are nonlinear, as shown by the dashed line 61 in FIG. 6. The horizontal axis of FIG. 6 is the excitation current Iexe [A], and the vertical axis is the magnetic flux density B [T]. If the motor is magnetically nonlinear, the explanation of the general characteristics of the motor becomes complicated, so in order to explain the basic motor model characteristics, the characteristics shown by the thick solid line 62 in FIG. 6 are assumed to be those of the motor. In addition, in a situation where a large torque is generated, the large current region on the left or right of FIG. 6 is used.

[0049] In particular, the motor of the present invention is driven by a current in one direction, but magnetically utilizes both positive and negative magnetic flux density regions in FIG. 6 by utilizing permanent magnets. For example, in the absence of stator current in FIG. 1, for example, in the region surrounded by square line 1T, which is the soft magnetic material portion of the rotor N pole magnetic pole 1H, and in its vicinity, there is magnetic flux from the top to the bottom of the paper of the permanent magnets 1P and 1N. The magnetic operating point of the part with the largest magnetic flux density in that region is, for example, a value from 63 to 64 in FIG. 6. Next, the magnetic flux passing through the region surrounded by square line 35 in FIG. 3 is from the bottom to the top of the paper, and its magnetic flux density is, for example, a value from 67 to 66 in FIG. 6. In this way, the magnetic flux density B of the soft magnetic material portion of the rotor N pole magnetic pole 1H changes from -2[T] to +2[T] in FIG. 6.

[0050] As shown in Figure 5, when this motor generates torque, magnetic flux passes from the bottom to the top of the paper and acts as an N pole, so the soft magnetic material portion indicated by 1T at the rotor N pole 1H in Figure 1 can be seen as a state in which the magnetic flux is reverse biased by permanent magnets 1P and 1N. By using the negative magnetic flux density region in Figure 6, it becomes possible to utilize magnetic flux density twice as high as conventionally. More details will be explained later. In the case of the conventional switched reluctance motor shown in Figure 63, the drive current is one-way, and the magnetic flux density of each part of the motor is also driven using only one-way magnetic characteristics.

[0051] Next, an example of expressing the motors in Figs. 1 and 2 in the linear development diagram of Fig. 7 will be shown and explained. It is easier to understand the operation of the entire motor when a circular motor is developed linearly. In addition, since Figs. 1 and 2 show a cross section of one pole pair of the stator, the soft magnetic material part of each rotor pole is fan-shaped. Each slot of the stator is also fan-shaped. The soft magnetic material part of each rotor pole and each slot of the stator shown in Fig. 7 are deformed linearly, so they become rectangular, and their shapes change. However, for example, motors with a large output capacity of 3k [kW] to 100 [kW] often have a large motor shape and multi-polarity. In that case, the soft magnetic material part of the rotor pole and each slot of the stator are closer to a rectangular fan-shaped shape. In addition, the motors in Figs. 1 and 2 are three-phase motors, but in 5-phase, 7-phase, 9-phase, and 11-phase motors described later, the number of stator poles and rotor poles increases, so the fan-shaped shape approaches a rectangular shape. Therefore, although the cross-sectional shape of one stator pole pair in Fig. 1 and the linear development in Fig. 7 differ in particular in the shape of each sector, this does not pose a major problem in a rough evaluation of a motor model with a large number of poles. However, when evaluating a motor more accurately, attention must be paid to this difference in shape.

[0052] The linear development diagram of FIG. 7 linearly develops the motor configuration of FIG. 2. As in FIG. 2, A-phase current Ia is applied, and A-phase magnetic flux φa is generated as shown at 79 and 7A. 71 is the stator, and 73 is the rotor. Each stator pole Ps of the stator 71 and each rotor pole Pr of the rotor 73 face each other through an air gap. The air gap length is 7F, and is greatly enlarged for easy visual understanding. In a motor of about 10 kW, the air gap length is usually about 0.5 mm to 1 mm. 72 is the back yoke of the stator, and 74 is the back yoke of the rotor. 7D is the stator back yoke, and 7E is the length of the stator teeth and the depth of the slot. The left and right ends of FIG. 7 are represented by wavy lines, indicating that the left and right ends of FIG. 7 are connected by going around one circle. The drawing is slightly wider than the electrical angle of 360° of one stator pole pair.

[0053] Here, the direction of the linear development in FIG. 7 is the right direction on the paper, which is the counterclockwise rotation direction CCW of the rotor in FIG. 1 and FIG. 2, which may visually feel strange, so care should be taken. In the case of rotational motion as in FIG. 1 and FIG. 2, the CCW direction from the first quadrant to the second quadrant is used as the reference. On the other hand, in the case of linear motion, the movement from the left side to the right side is used as the reference. Therefore, the arrangement direction of the stator poles is reversed in FIG. 1, FIG. 2, and FIG. 7. Also, the leftward movement of the rotor around the A-phase S-pole stator pole 11 in FIG. 1 and FIG. 2 becomes the rightward movement of the rotor around the A-phase S-pole stator pole 7L in FIG. 7, which is visually reversed. Note that if you consider linear development from the back side of the paper of FIG. 1 and FIG. 2, the visual left-right movement will match that in FIG. 7.

[0054] Figure 7 7J is the B / phase north stator pole and corresponds to 16 in Figure 2. 7L is the A / phase south stator pole and corresponds to 11 in Figure 2. 75 is the C / phase north stator pole and corresponds to 12 in Figure 2. 7M is the A / phase north stator pole and corresponds to 14 in Figure 2. 7N is the rotor north pole and corresponds to 1H in Figure 2. 7P is the A / phase concentrated winding that excites the A / phase south stator pole 7L and corresponds to 1A in Figure 2. 7Q is the A / phase concentrated winding that excites the A / phase north stator pole 7M and corresponds to 14 in Figure 2.

[0055] In the rotor 73 in FIG. 7, the N and S rotor poles are arranged alternately in the circumferential direction, for example, 7N is the rotor N pole and 7K is the rotor S pole. At the boundary between the rotor poles, permanent magnets such as 76 and 77 are arranged in the direction of the rotor poles. When no stator current is flowing, the magnetic flux generated by each permanent magnet circulates mainly in the soft magnetic material in the rotor, forming a closed circuit. The magnetic flux is generated as shown by the broken lines 78 and 7R of the permanent magnet 76, and most of the magnetic flux circulates in the soft magnetic material, forming a closed circuit. The rotor rotation angle in FIG. 7 is θr=30°.

[0056] If the magnetic performance of the permanent magnets 76, 77, etc. is improved in order to improve the motor torque, the magnetic flux components shown by the dashed lines 78, 7R, etc. will increase in leakage magnetic flux components to the air gap side even when no stator current is flowing. Although not shown in FIG. 7, part of the leakage magnetic flux components will pass through the stator poles and circulate. The leakage magnetic flux components to the air gap side are shown in FIG. 10, etc. Naturally, when the stator current is flowing, the magnetic flux components 78, 7R, etc. will follow complex magnetic flux paths as shown in FIG. 8, FIG. 9, FIG. 10, and FIG. 11.

[0057] 7, A-phase magnetic flux component φa of 79 and 7A is generated, and the A-phase magnetic flux component φa circulates through the stator back yoke 72 and the rotor back yoke 74. The areas surrounded by rectangular lines 7B and 7C show a state in which the A-phase magnetic flux component φa of 79 and 7A is superimposed on the magnetic flux component of the permanent magnet shown by the dashed line with an arrow. For reference, the names of the stator poles, such as A phase, B phase, and C phase, are shown in parentheses at the top of the page in FIG.

[0058] Next, to show the magnetic flux distribution in more detail, a part of FIG. 7 is enlarged and shown in FIG. 8. FIG. 8(a) is an enlarged view of the periphery of the stator S-pole magnetic pole 7L in FIG. 7. The dashed lines with arrows 81, 82, 83, 84, etc. are magnetic flux components of permanent magnets. The A-phase magnetic flux φa of 79 and the magnetic flux components of the permanent magnets shown by the dashed lines are drawn overlapping in the area enclosed by the rectangular line 7B.

[0059] FIG. 8B is a diagram in which the overlapping magnetic flux components in FIG. 8A are rewritten into an actual magnetic flux distribution. This is a state in which the A-phase current Ia is flowing through the A-phase winding 7P and the A / phase winding 7Q. For example, in a state in which the A-phase current Ia is not flowing, the area surrounded by the square line 89 is assumed to be at the magnetic operating point 64 in FIG. 6, and the magnetic flux density is set to 68. In this case, if the magnetic flux density from the bottom to the top on the paper surface of FIG. 8 is assumed to be a positive value, the positive / negative sign of the magnetic flux density at 68 is negative. That is, in FIG. 8A, when the A-phase current Ia is 0 [A], the area surrounded by the square line 7B is in a magnetic reverse bias state, which is a preparation state for utilizing a large change in magnetic flux density such as 69 or 6A in FIG. 6, and the reverse bias is important. 8(b) shows a state when the A-phase current Ia is passed such that the magnetic flux density in the region surrounded by the rectangular line 89 becomes 0 [T]. Magnetic fluxes 85, 86, 87, and 88 passing through the S-pole stator pole 7L do not pass through the rectangular region 89 of the rotor, but pass through the permanent magnets located on the left and right sides of the N-pole rotor pole 7N on the page, and supply magnetic flux to the S-pole stator pole 7L.

[0060] In the state of the magnetic flux distribution in FIG. 8(b), the magnetic fluxes 85, 86, 87, and 88 passing through the S-pole stator magnetic pole 7L utilize the magnetic paths of the rotor magnetic poles 7K and 7S adjacent to the rotor magnetic pole 7N on the paper surface, and are supplied to the S-pole stator magnetic pole 7L. Therefore, in the area surrounded by the rectangular line 89 of the rotor magnetic pole 7N, the magnetic flux density is 0 [T], so there is still sufficient magnetic flux supply capacity to the stator. In other words, it can be said that the rotor with this structure "can utilize the soft magnetic magnetic path of the rotor magnetic pole adjacent in the circumferential direction to supply magnetic flux to the vicinity of the air gap of the rotor magnetic pole and to the stator magnetic pole." In addition, the stator teeth shown in FIG. 1 and the like have a large space in the slot between the teeth, and it is possible to increase the passing magnetic flux by widening the circumferential width of the teeth, which will be described in detail later. It is also possible to add permanent magnets between the stator teeth to increase the magnetic flux acting on the stator poles, as will be described in more detail below.

[0061] Next, FIG. 9 shows the magnetic flux distribution when the A-phase current Ia of 7P is increased from the state of FIG. 8(b). In FIG. 9, magnetic fluxes 91 and 92 increase compared to the state of FIG. 8(b). The magnetic flux passing through the region surrounded by the square line 89 increases. Expressed in terms of the magnetic operating points of FIG. 6, if the A-phase current Ia is 0 [A] in the region 89 and the operating point is 64, it can be said that the magnetic flux density in the region 89 has increased to 69 in FIG. 6, assuming that the magnetic flux density becomes 2 [T] in the state of FIG. 9. That is, the A-phase current Ia is a one-way current excitation, but the magnetic flux density of the rotor magnetic pole is effectively utilized by changing it from a negative value to a positive value as shown in FIG. 6. It is also a technology that utilizes the soft magnetic magnetic path of the rotor magnetic pole that is adjacent in the circumferential direction and is not used.

[0062] Next, FIG. 10(a) shows a linear development diagram in a state where the A-phase S-pole stator pole 7L of the stator and the S-pole 7S of the rotor face each other, and the A-phase current Ia of the A-phase winding 7P is not energized. The rotor rotation position is the same as FIG. 4, and θr=0°. FIG. 10 shows an example in which the rotor's permanent magnets 77, 10A, 10B, etc. have slightly higher performance and a larger magnetic flux density. For example, the magnetic flux components 101, 102, 103, and 104 of the rotor's permanent magnets 10A and 10B circulate through the back yoke side of the rotor, but when the magnetic flux density passing through the square area 105 approaches 2 [T], the magnetic resistance increases, and the magnetic flux components 107 and 108 on the air gap side become large enough to be ignored. The same applies to the other magnetic flux components 106 and 109. 8 and 9, the magnetic flux components on the air gap side are ignored, but in FIG. 10, the magnetic flux components 106, 107, 108, and 109 are added.

[0063] In Fig. 10(b), A-phase current Ia is passed through A-phase winding 7P and A / -phase winding 7Q in Fig. 10(a) to excite the magnetic flux component 10C. The magnetic flux components shown by dashed lines of each permanent magnet of the rotor and the magnetic flux component 10C are shown overlapping each other. As mentioned above, the magnetic flux density in square region 105 is already large, so the magnetic resistance and magnetic flux component 10C passing through square region 10D cannot be a large value.

[0064] Figure 11(a) is a qualitatively rewritten view of the magnetic flux distribution combining the two types of magnetic flux components overlaid in Figure 10(b). For example, the magnetic flux component 107 in Figure 10(b) becomes like the magnetic flux component 111 in Figure 11(a), and similarly, 108 becomes like the magnetic flux component 112, and there is also magnetic flux component 113 that passes directly from the south pole 7S of the rotor to the south pole 7L of the stator. Since the magnetic flux density in the square region 118 is already large, the magnetic resistance becomes large to allow the passing magnetic flux to increase further, and the magnetic flux components 111, 112, and 113 are small values.

[0065] FIG. 11B shows an example of magnetic flux distribution when the A-phase current Ia is applied to the A-phase winding 7P and the A / phase winding 7Q at a position where the torque T can be generated with the rotor rotation angle θr=12°. The rotor rotation angle θr=12° is a rotor position where the rotor N-pole 7N and the adjacent S-pole 7S on the right side face the stator S-pole 7L to the same extent. The magnetic flux 114 passes through the magnetic path of the S-pole 7K on the left side of the rotor N-pole 7N that generates force with the A-phase current Ia, passes through the permanent magnet 77, and passes through the A-phase stator S-pole 7L of the stator. The magnetic fluxes 115 and 116 pass through the magnetic path of the S-pole 7S on the right side of the rotor N-pole 7N, passes through the permanent magnet 10A, and passes through the A-phase stator S-pole 7L of the stator. The magnetic flux 117 passes through the rotor south pole 7S and directly through the A-phase stator south pole 7L of the stator, and since the magnetic flux density in the square area 119 is already large, the magnetic resistance is large and the magnetic flux 117 does not have a large value. Note that the magnetic flux density in the square area 11A is a negative magnetic flux density as the magnetic flux density from the bottom to the top of the paper, since the magnetic flux of the permanent magnet 77 and the magnetic flux 102 pass from the top to the bottom of the paper. Therefore, the magnetic flux passes from the rotor north pole 7N to the stator south pole 7L, and there is sufficient surplus power to supply.

[0066] In Fig. 1 to Fig. 11, the passage and interruption of the magnetic flux φ [Wb] between the stator and the rotor have been described. In Fig. 11(b), the generation of the torque T [Nm] due to the passage and interruption of the magnetic flux φ [Wb] has also been described. The relationship between the magnetic flux φ [Wb] and the torque T [Nm] is shown in the following equations (1) to (7). In a motor such as Fig. 1, in order to obtain a larger torque T [Nm], and in order to make the rotor rotation angle that can be generated by one rotor pole a large value close to the rotation angle width of the rotor pole, it is important to make the magnetic flux component 79 in Fig. 8(a) a large value and to suppress the magnetic flux component 10C in Fig. 10(b) to a small value. The maximum value of the torque T [Nm] is proportional to the difference between the maximum value of 79 and the maximum value of 10C.

[0067] In particular, the methods of blocking or reducing the magnetic flux shown in Fig. 10(b) and Fig. 11(a) are important. That is, for example, as shown in Fig. 11(a), in a state where the stator poles and rotor poles of the same polarity face each other, the magnetic flux density B in the square region 118 is close to the maximum value 2 [T] of the soft magnetic material. Since the magnetic flux density is high in the region 118, the relative permeability is low and the magnetic resistance is high, so that the magnetic flux in 113 can be suppressed to a small value. As for the magnetic flux components that wrap around 111 and 112, the magnitude of the magnetic flux is suppressed by the action of the magnetic resistance in the region 118 becoming large.

[0068] There are several methods for suppressing the magnetic flux 113 to a small value. One of the suppression methods is to increase the magnitude of the magnetic flux generated by the permanent magnet of the rotor as shown in FIG. 10(b). However, in that case, the magnetic fluxes 106, 107, 108, 109, etc. shown in FIG. 10(b) will increase, so care must be taken to avoid adverse effects such as increased torque ripple. Another suppression method is to place a permanent magnet near the rotor surface facing the air gap, facing the direction of the rotor magnetic poles. Another suppression method is to add a field winding inside the rotor and pass a field current through it. It is also possible to control the magnitude of the field current.

[0069] Here, we will confirm the relationship between the interlinkage magnetic flux φ [Wb] of the windings and the torque T [Nm]. One way to observe and evaluate the torque T [Nm] of the motor is to observe the magnetic flux φ [Wb] interlinked with the windings. In other words, it can be evaluated by the magnitude of the magnetic flux change Δφ [Wb], which is the change in the magnetic flux φ [Wb] interlinked with the windings as the rotor rotates. Now, the power Pe [W], voltage V [V], and current I [A] supplied from the power supply to the motor side become the mechanical output Pm [W], torque T [Nm], and rotational angular frequency ω [rad / sec]. Assuming that there is no internal loss, the following relationship is obtained from Faraday's law of electromagnetic induction. The sum of the number of turns of the A-phase and A / phase windings is Nw [turn], and the rotor rotation position is θ [rad]. ω = dθ / dt. Pe = V × I (1) = Nw × dφ / dt × I (2) = Nw × dφ / dθ × dθ / dt × I (3) = Nw × dφ / dθ × ω × I Pm = T × ω (4) =T×dθ / dt (5) 1, the A-phase winding 1A and the A / phase winding 1D are connected in series to pass the A-phase current Ia, and the sum of the number of turns of both windings is Nw [turns]. The number of turns of the A-phase winding 1A is Nw / 2.

[0070] If the value of the supplied power Pe [W] in equation (3) and the value of the motor's mechanical output Pm [W] in equation (5) are equal, the torque T [Nm] can be approximately expressed by equation (7). T = Nw × dφ / dθ × I (6) =Nw×I×Δφ / Δθ (7) Here, what can be said from equation (7) is that when the rotor rotation angle is Δθ [rad], a torque proportional to the change in the interlinkage magnetic flux of the windings, Δφ [Wb], can be obtained. Therefore, the torque T [Nm] is obtained almost proportional to the change Δφ [Wb] in the A-phase winding 7P's flux linkage, which changes from θr = 0°, which is moved 6° from the position (a) in Figure 11, to the state of θr = 30° in Figure 9. From this perspective, the torque becomes 0 in the state of θr = 30° in Figure 9, but the flux linkage and magnetic flux distribution are shown, which are maximum at θr = 30°. It was also shown that the rotor's ability to supply magnetic flux to the stator is sufficiently large.

[0071] Also, as shown in equations (6) and (7), the torque T [Nm] is proportional to the rate of change dφ / dθ or Δφ / Δθ of the interlinkage magnetic flux φ [Wb], not the magnitude of the interlinkage magnetic flux φ [Wb]. As shown in equation (2), power Pe [W] is supplied, and is electromagnetically converted to mechanical power Pm [W] as shown in equations (4) and (5). That is, the conventional switched reluctance motor shown in FIG. 63 utilizes the change in magnetic flux density [T] in one direction shown in 6B of FIG. 6, but the rotor of the motor of the present invention shown in FIG. 1 and the stator shown later generate the torque T [Nm] by utilizing the change in magnetic flux density [T] in both directions shown in 69 and 6A of FIG. 6. Note that, for simplicity, the explanation is given here ignoring the motor internal loss and magnetic energy.

[0072] Next, FIG. 12 shows a linear development diagram showing the operation of the 6S10R motor in FIG. 1. The development diagram in FIG. 12 shows the shape of the stator magnetic poles facing the air gap surface and the shape of the rotor magnetic poles, and allows the analysis of the mutual passing magnetic flux and electromagnetic action. Specifically, it is a linear development diagram for the purpose of drawing the CCW torque generation section. The CCW direction in FIG. 1 is the forward rotation direction, and the right direction in FIG. 12 is the CCW direction. As mentioned above, it is necessary to be careful because it may visually feel the opposite direction. The horizontal axis in FIG. 12 is the rotor rotation angle θr, and the rotor rotation position where the upper left corner of the rotor's N pole magnetic pole 1H approaches the lower right corner of the S pole stator magnetic pole 11 of the A phase, which is the first phase, in FIG. 1 is θr=0°. FIG. 12 shows the rotor rotation angle θr from -30° to 360°. Although it is a little confusing, in FIG. 12, the horizontal axis θr indicates the position of each part of the rotor and also the position of each part of the stator in the rotation direction. The rotational position of the rotor is shown on the left side of each row in Fig. 12. Note that the rotational position of the rotor in Fig. 1 is θr = 30°.

[0073] In this example, the circumferential width of each stator pole is 30°, the circumferential width of each slot is 30°, and the stator pole pitch θpps is 60°. The pitch θppr of each rotor pole is 36°, and a total of 10 rotor north poles and rotor south poles are alternately arranged around the entire circumference. Figure 12 illustrates the case where the circumferential width of each rotor pole is 30°. Figure 12 (a) shows the shape of each stator pole facing the air gap surface. The A-phase south stator pole is shown between θr 0° and 30°, and corresponds to 11 in Figure 1, so it is indicated with the same reference symbol. To the right of the A-phase, the stator poles of each phase, C / phase, B / phase, A / phase, C / phase, and B / phase, are shown arranged in the same manner. Note that the circumferential width of the stator poles and the circumferential width of the rotor poles can be modified and designed by reducing or expanding them within the space permitted in terms of motor design.

[0074] In a linear development diagram like Figure 12, the position of the stator pole in Figure 12(a) is fixed, and the rotor pole position is moved left and right on the paper, and written as in Figure 12(b) onwards, to investigate the section in which CCW torque can be generated. At the top of each row, the section in which CCW torque can be generated is shown by a thick line above the rotor pole shape. At this time, the position and width of the thick line correspond to the position and width of the corresponding stator pole.

[0075] To repeat, Figure 12(b) shows the shape of the rotor poles facing the air gap surface. The rotor's north and south poles are alternately arranged, for a total of 10 rotor poles, at a pitch of 36°. Each stator pole in Figure 12(a) faces each rotor pole in Figure 12(b) via an air gap. The rotor rotation position θr=0° in Figure 12(b) corresponds to the rotor rotation position θr in Figure 4. The value of the rotor rotation position θr is also written at the left end of Figure 12(b).

[0076] Next, the generated torque and the torque generation section in FIG. 12(b) will be described. The torque in the CCW direction in FIG. 1 is the torque in the right direction on the paper of FIG. 12. 121 is the N-pole magnetic pole of the rotor, which is attracted to the A-phase stator S-pole magnetic pole of the stator 11, and generates an attractive force in the right direction on the paper between the rotor rotation angle θr = 0° to 30°. The section where this attractive force is generated is shown by a thick line on the upper right of the N-pole magnetic pole 121. Similarly, 122 is the S-pole magnetic pole of the rotor which is 180° away from the N-pole magnetic pole 121 at an electrical angle which is 1 / 2 of the electrical angle of 360° of the stator 1 magnetic pole pair, and is attracted to the A / phase stator N-pole magnetic pole of the stator 14, and generates an attractive force in the right direction on the paper between the rotor rotation angle θr = 0° to 30°. The section where this attractive force is generated is shown by a thick line on the upper right of the S-pole magnetic pole 122. Also, 123 is a rotor N-pole magnetic pole, which is attracted to the B-phase stator S-pole magnetic pole of the stator 13, and generates an attractive force in the right direction of the paper between rotor rotation angles θr = -24° to 6°. The section where this attractive force is generated is shown by a thick line almost above the N-pole magnetic pole 123. Similarly, 124 is a rotor S-pole magnetic pole, which is attracted to the B / phase stator N-pole magnetic pole of the stator 16, and generates an attractive force in the right direction of the paper between rotor rotation angles θr = -24° to 6°. The section where this attractive force is generated is shown by a thick line almost above the N-pole magnetic pole 124. As described above, at the position of rotor rotation angle θr = 0° in FIG. 12(b), CCW torque can be generated at four points. Note that the rotor N-pole magnetic pole 121 is the same as the N-pole magnetic pole 125 shown by a broken line at a position where the phase difference is 360°.

[0077] Next, in FIG. 12(c), each rotor pole moves to the right, and the rotor rotation angle is θr=6°. In this position, stator poles 13 and 16 can no longer generate an attractive force to the right of the page. The A-phase stator south pole 11 and the A / phase stator north pole generate an attractive force to the right. In this way, at the rotor rotation angle θr=6° position in FIG. 12(c), CCW torque can be generated in two places. In other words, because the north and south polarities of both the stator and rotor poles are fixed, the other four stator poles cannot generate CCW torque.

[0078] Next, in FIG. 12(d), the stator poles 11, 12, 14, and 15 can generate torque at the rotor rotation angle θr=24°, as shown in the figure. In FIG. 12(e), the stator poles 12 and 15 can generate torque at the rotor rotation angle θr=30°, as shown in the figure. In FIG. 12(f), the stator poles 12, 13, 15, and 16 can generate torque at the rotor rotation angle θr=48°, as shown in the figure. In FIG. 12(g), the stator poles 13 and 16 can generate torque at the rotor rotation angle θr=54°, as shown in the figure. In FIG. 12(h), the state returns to the same state as in FIG. 12(b). Then, the motor in FIG. 1 repeats the same operation five times in a 72° cycle, and the rotor makes one rotation.

[0079] Next, Fig. 13 shows examples of currents flowing through each phase winding, and several current flow methods will be described. Fig. 13 (a), (b), and (c) show examples of A-phase current Ia, B-phase current Ib, and C-phase current Ic flowing during the operation of the motor in Fig. 1 and Fig. 12. The circumferential width of the stator pole and the circumferential width of the rotor pole are both 30°, and this is an example of generating CCW torque by rotating in the CCW direction. In the case of the rectangular wave-shaped thick solid line current in Fig. 13 (a), (b), and (c), the A-phase current Ia flows from 3° to 27°, the C-phase current Ic flows from 27° to 51°, and the B-phase current Ib flows from 51° to 75°, and these operations are repeated in a 72° cycle. The A-phase, C-phase, and B-phase generate torque sequentially at 24° each, and in terms of the motor model, it is possible to continuously output a nearly uniform torque.

[0080] Also, it is possible to pass trapezoidal A-phase current Ia, B-phase current Ib, and C-phase current Ic, as shown by the dashed lines in Fig. 13(a), (b), and (c). The A-phase current Ia increases from 0° to 6°, is constant from 6° to 24°, and decreases to 0 [A] from 24° to 30°, passing a trapezoidal current. Similarly, the C-phase current Ic passes a trapezoidal current between 24° and 54°. Similarly, the B-phase current Ib passes a trapezoidal current between 48° and 78°. These operations are repeated for each of them in a 72° cycle. When these Ia, Ib, and Ic are added together, the current is always constant, and logically, a uniform torque can be expected. Since the current increases and decreases gradually, the voltage burden on the drive circuit at high speed rotation can be reduced, and reductions in torque ripple, vibration, and noise can also be expected. It is also possible to increase the average torque by making each of the current waveforms Ia, Ib, and Ic a rectangular waveform with a width of 30 degrees. In that case, however, it is necessary to take measures to reduce torque pulsation. Various current waveforms can be used as necessary.

[0081] Next, an example of a motor model in which the circumferential width θsg of the stator pole facing the air gap in FIG. 1 and FIG. 12 and the circumferential width θrg of the rotor pole are both increased to 36° and the pole width is widened will be described in (d), (e), and (f) of FIG. 13. One method is to drive with a rectangular wave current shown by the thick solid line in the rectangular wave shape in (d), (e), and (f) of FIG. 13. The A-phase current Ia in (d) of FIG. 13 flows from 0° to 36°, the C-phase current Ic in (f) flows from 24° to 60°, and the B-phase current Ib in (e) flows from 48° to 84°, and these operations are repeated in a 72° cycle. The average torque increases, but torque pulsation is expected, and measures to reduce torque pulsation must also be considered. Various methods, such as amplitude correction, can be applied.

[0082] Next, as shown by the dashed lines in (d), (e), and (f) of FIG. 13, Ia, Ib, and Ic can also be driven with trapezoidal currents. The A-phase current Ia in FIG. 13(d) increases from 0° to 12°, is constant from 12° to 24°, and decreases to 0 [A] from 24° to 36°, passing a trapezoidal current. Similarly, the C-phase current Ic passes a trapezoidal current between 24° and 60°. Similarly, the B-phase current Ib passes a trapezoidal current between 48° and 84°. These operations are repeated for each with a cycle of 72°. When these Ia, Ib, and Ic are added together, the current is always constant, and a logically uniform torque can be expected. Because the trapezoidal current increase and decrease is even more gradual than that shown by the dashed lines in Figures 13(a), (b), and (c), the voltage burden on the drive circuit at high speed rotation can be reduced, and reductions in torque ripple, vibration, and noise can also be expected.For example, one specific drive method that can be considered is to generate large torque by driving with a trapezoidal current waveform that is close to a square wave at low to medium speed rotation, and to drive with a trapezoidal current waveform with a gradual increase and decrease at high speed rotation.

[0083] Figures 12, 13 (a), (b), and (c) show examples where the stator pole width and rotor pole width are 30°. In this case, if the torque generation width of each phase is 24°, the motor can output continuous torque by generating each phase torque in sequence. In the case of the conventional switched reluctance motor shown in Figure 63, in order for the motor to output continuous torque, each phase torque needs to be generated at 30°. This characteristic is different from the motor configuration in Figure 1. In the 6S10R configuration with the rotor pole characteristics in Figure 1, the torque width of each phase is sufficient at 24°, so it is possible to add a time for increasing and decreasing the current, as in the current example shown by the dashed lines in Figures 13 (a), (b), and (c). This freedom in the current flow time is one of the features of the motor in Figure 1.

[0084] Also, as shown by some examples in Fig. 13, even though the basic parts of the motor structure in Fig. 1 and the operation in Fig. 12 are the same, various modifications are possible for the shape of the stator poles, the shape of the rotor poles, and the current waveform. Measures for improving torque by driving in a region where torque can be generated more effectively, measures for reducing torque ripple, measures for reducing vibration and noise, etc. can be applied. Also, for example, the pole shape can be adjusted not only by the circumferential width but also by skewing, the shape of the poles, radial unevenness, the air gap inside the poles, etc., and addition of permanent magnets is also possible. Regarding the current waveform applied to the stator winding, a rectangular wave current, a trapezoidal current, a sine wave current, a quadratic function increase or decrease of the current, or correction of the current amplitude is possible. For example, as a specific driving method, at low speed rotation, drive with a current waveform close to a rectangular wave to increase the average torque, and at high speed rotation, drive with a trapezoidal current waveform to gently increase and decrease the current and reduce the burden of current drive, and at the same time, perform driving to reduce torque ripple and vibration and noise. Note that regarding the increase and decrease of the current in Fig. 13 of the motor in Fig. 1, there are problems such as the problem of the time for regenerating the magnetic energy of each phase to the inverter side, the problem of torque reduction at high speed rotation, and the problem of vibration and noise. More specific problems and solutions, such as the method of constantly energizing a current component that excites magnetic flux in each phase current, will be described later.

[0085] Let's briefly summarize the embodiments from Fig. 1 to Fig. 12. Conventional reluctance motors such as Fig. 63 driven by unidirectional current have rotor poles made of soft magnetic materials and no polarities such as N poles and S poles. In the configuration of Fig. 1 of the present invention, the rotor poles have a configuration with fixed polarities of N poles and S poles. That is, it is a motor configuration in which both the stator and rotor poles have polarities of N poles and S poles. By fixing the stator poles to N poles and S poles, it can be driven by unidirectional current, so the drive circuit can be simplified and the cost can be reduced. However, of course, an attractive force acts between unlike poles of N poles and S poles, but no attractive force acts between like poles. Therefore, it is necessary to find a special and convenient relative relationship between the stator poles and the rotor poles to construct the motor.

[0086] As shown in FIG. 1 and the like, the stator is similar to that of the conventional reluctance motor in FIG. 63, but by fixing the rotor magnetic poles to N and S poles, a permanent magnet can be used in the rotor. By utilizing the permanent magnets in the rotor in a unique arrangement as shown in FIG. 1, the soft magnetic magnetic paths of the rotor magnetic poles that are adjacent in the circumferential direction and are not used can be utilized as shown in 85, 86, 87, and 88 in FIG. 8(b) and FIG. 9(a). As a result, the magnetic flux supply capacity on the rotor side can be significantly improved and the torque can be increased. On the other hand, for the magnetic flux components that try to pass in the reverse direction, as shown in 101, 102, 103, and 104 in FIG. 11, the square area 118 becomes magnetically saturated and the magnetic resistance becomes large, suppressing and limiting the passage of the magnetic flux. In addition, as shown in the example of the current waveform in FIG. 13, there is a degree of freedom in the current waveform, and by making the current waveform trapezoidal or the like, the increase time and decrease time of the current can be secured, so that the current control becomes easy and the vibration noise of the motor can be reduced. This is significantly different from the conventional reluctance motor shown in Fig. 63. As mentioned above, in this specification, the stator and rotor are combined as shown in Fig. 1, and the mechanical angle of 360° is treated as the electrical angle of one stator pole pair of 360°. In the case of the motor in Fig. 1, the circumferential range of the ten rotor poles is treated as an electrical angle of 360°.

[0087] Furthermore, it is possible to increase the magnetic flux acting on the stator poles by adding permanent magnets between the teeth of the stator in FIG. 1, as described in the embodiment of claim 2. Also, the stator winding in FIG. 1 is shown as an example of concentrated winding, but it can also be full-pitch winding. This is a current supply method in which the locations of each part of the rotor are selectively excited with full-pitch winding of unidirectional current, and the direction of excitation is also specified, as described in the embodiment of claim 4. Also, the combination of the stator poles and the rotor poles has been described as an example of three phases in FIG. 1, but it can be expanded to five, seven, nine, eleven phases, etc., and there are specific combinations that have a high utilization rate inside the motor, are very practical, and enable high efficiency, miniaturization, and weight reduction. The embodiment of claim 5 and other embodiments will be described later. Also, the teeth of the stator shown in FIG. 1 have a margin in the slot space between the teeth as shown, and it is possible to expand and deform the circumferential width of the teeth to increase the passing magnetic flux, as described in the embodiment of claim 7. In the present specification, the term "utilization rate of winding" is used to refer to the utilization rate of the motor winding. Additionally, the term utilization rate of transistor TR is used to refer to the percentage of current used to drive the transistor TR. The percentage of current that can be passed through the windings to generate torque ultimately affects the motor's winding resistance. For example, compared to a motor that can generate torque by passing current continuously through all windings for 100% of the time, a motor in which 50% of the windings generate torque passes twice as much current through half the windings, doubling the total copper loss. When the utilization rate of transistor TR is 50%, the total current capacity of all transistors TR in the inverter increases by two times, simply compared to when the utilization rate of transistor TR is 100%. Generally speaking, a high utilization rate allows for smaller size, lighter weight, and lower costs.

[0088] Next, an embodiment of claim 2 will be described. FIG. 14 shows a configuration in which permanent magnets 145, 146, 147, 148, 149, and 14A are added to the configuration of FIG. 1 and FIG. 2. The arrow marks above each permanent magnet indicate its polarity direction. The dashed lines of 14B and 14C show the magnetic flux components of the permanent magnets. A-phase current Ia is passed through the A-phase winding 1A and the A / phase winding 1D of the stator to excite the A-phase magnetic flux components 141 and 142. The square line 143 shows the soft magnetic material area of ​​the N-pole magnetic pole 1N of the rotor. The square line 144 shows the soft magnetic material area of ​​the S-pole magnetic pole 1R of the rotor. The A-phase magnetic flux components 141 and 142 are drawn superimposed on the magnetic flux shown by the dashed lines of each permanent magnet of the rotor and the magnetic flux shown by the dashed lines of each permanent magnet of the stator. It circulates through the back yoke of the stator. The other components in FIG. 14 are denoted by the same reference numerals as those in FIGS.

[0089] In Figure 14, permanent magnets 145, 146, 147, 148, 149, and 14A are shown as long in the circumferential direction, but this is done to show the motor configuration as a model. At the motor design stage, if the motor in Figure 14, which has two stator pole pairs, is increased to eight stator pole pairs, for example, if the circumferential length of each permanent magnet is reduced to about 1 / 4, the cross-sectional shape of the permanent magnet will approach a parallelogram. In addition, the shape of the soft magnetic material in contact with each permanent magnet can be freely modified to match the shape of the permanent magnet.

[0090] FIG. 15 is a linear development of the cross-sectional view of the motor in FIG. 14. 151 is a stator, and 152 is a back yoke of the stator. 15L is an A-phase stator south pole, which corresponds to 11 in FIG. 14, 15M is an A-phase stator north pole, which corresponds to 14 in FIG. 14, 15J is a B-phase stator north pole, which corresponds to 16 in FIG. 14, and 155 is a C-phase stator north pole, which corresponds to 12 in FIG. 14. 15P is an A-phase winding, and 15Q is an A-phase winding. Both windings are connected in series and A-phase current Ia is passed through them to excite magnetic fluxes 159 and 7A that circulate through the back yoke. Permanent magnets such as 15H and 15S are arranged between the stator poles, facing the polarity of the stator poles. The dashed line of 15G is the magnetic flux component of the permanent magnet 15H, and the dashed line of 15T is the magnetic flux component of the permanent magnet 15S. The same applies to the magnetic flux components shown by dashed lines for the other permanent magnets of the stator.

[0091] The reference symbols for the various rotor components at the bottom of the paper in Fig. 15 are the same as those in Fig. 7. However, the distribution of magnetic flux density and its effects change significantly with an increase in magnetic flux density on the stator side, which will be explained later. The air gap length 7F is enlarged for ease of viewing. The rotor rotation angle in Fig. 15 and Fig. 14 is θr = 30°. The CCW rotation in Fig. 14 corresponds to the rightward movement of rotor 73 in Fig. 15.

[0092] Next, in order to show the magnetic flux distribution in more detail, a part of FIG. 15 is enlarged and shown in FIGS. 16, 17, 18, and 19. FIG. 16(a) is an enlarged view of the periphery of the stator S - pole magnet 15L in FIG. 15. In FIG. 16(a), the stator current is not energized, and the distribution states of the magnetic - flux components 15G, 15T of each permanent magnet of the stator and the magnetic - flux components 78, 7R of each permanent magnet of the rotor are shown. In the teeth of the A - phase stator S - pole magnet 15L, the magnetic - flux components 15G, 15T pass from above the paper surface to below, which is in the opposite direction to the magnetic - flux direction excited by the A - phase winding 15P and the A / - phase winding 15Q, and the magnetic - flux density is negatively biased. The magnetic flux in the region surrounded by the square line 161 of the rotor N - pole magnet 7N of the rotor is from above the paper surface to below, which is in the opposite direction to the magnetic - flux direction in which the rotor N - pole magnet 7N acts on the stator to generate torque, and the magnetic - flux density is negatively biased. In the magnetic characteristics of the soft magnetic material in FIG. 6, it is the corresponding operating point of 63 or 64, and the change in magnetic - flux density of 6A or 69 is possible. When the rotor rotation angle θr = 30°, the A - phase stator S - pole magnet 15L and the rotor N - pole magnet 7N of the rotor face each other through the air gap. Therefore, since the S - pole and the N - pole are facing each other, when the A - phase current Ia is passed through the A - phase winding 15P and the A / - phase winding 15Q, it is the rotor rotation position θr at which the A - phase magnetic flux φa most easily passes. And the maximum magnetic flux and the maximum magnetic - flux density that can pass between the stator and the rotor can be analyzed and evaluated. Note that at this rotation position of θr = 30°, the A - phase stator S - pole magnet 15L cannot generate torque.

[0093] FIG. 16(b) shows the state in FIG. 16(a) where the A - phase current Ia is passed through the A - phase winding 15P and the A / - phase winding 15Q to excite the A - phase magnetic flux φa of 159. The magnetic - flux components generated by the permanent magnets of the stator and the rotor and the 159 of the A - phase magnetic flux φa are superimposed and shown. In the region surrounded by the square line 161 of the rotor N - pole magnet 7N, the magnetic - flux component created by the permanent magnet of the rotor and the 159 of the A - phase magnetic flux φa are in opposite directions and cancel each other out. In the teeth of the stator S - pole magnet 15L, the magnetic - flux component 15G created by the permanent magnet 15H of the stator and the magnetic - flux component 15T created by the permanent magnet 15S of the stator and the 159 of the A - phase magnetic flux φa are in opposite directions and cancel each other out.

[0094] Figures 17(a) and (b) show the superimposed magnetic flux in Figure 16(b) converted into its distribution state and replaced. Figure 17(a) shows the case where the A-phase current Ia of A-phase winding 15P is not very large, while Figure 17(b) shows an example of magnetic flux distribution when the A-phase current Ia is large. Magnetic fluxes 171 and 172 in Figure 17(a) pass through the soft magnetic material magnetic path of rotor south pole 7K, through permanent magnet 77, through rotor north pole 7N, through the air gap, through stator south pole 15L, through permanent magnet 15H, through the teeth of stator north pole 15J, and through the stator back yoke. Magnetic flux 173 and 174 pass through the soft magnetic magnetic path of rotor south pole 7S, through permanent magnet 10A, through rotor north pole 7N, through the air gap, through stator south pole 15L, through permanent magnet 15S, through the teeth of stator north pole 155, and through the stator back yoke.

[0095] In FIG. 17(a), A-phase winding 15P and A / phase winding 15Q are excited by A-phase current Ia, and magnetic fluxes 171, 172, 173, and 174 pass from rotor N-pole 7N through the air gap to stator S-pole 15L. However, these magnetic fluxes have not yet passed through the area surrounded by square line 161, which is the soft magnetic material part of rotor N-pole 7N, and the teeth of stator S-pole 15L. Therefore, by increasing A-phase current Ia, the magnetic flux passing between rotor N-pole 7N and stator S-pole 15L can be increased. The magnetic flux passing near the air gap, indicated by elliptical thick dashed line 179, has the value of magnetic flux component 159 in FIG. 16(b). As described above, even if the area 177 which is the soft magnetic material portion of the tooth of the stator south pole 15L and the rotor north pole 7N is still negatively biased by the permanent magnet, regardless of their state, the magnetic flux passing through the air gap portion near 179 is a positive value, which is the value of the magnetic flux component 159.

[0096] Fig. 17(b) shows an example of magnetic flux distribution when the A-phase current Ia of Fig. 17(a) is increased. As shown in the figure, magnetic fluxes 171, 172, 173, 174, 175, and 176 pass from the rotor north pole 7N through the vicinity of the air gap in the region indicated by the elliptical thick dashed line 17A to the stator south pole 15L. The magnetic fluxes 175 and 176 have increased compared to Fig. 17(a). These magnetic fluxes 175 and 176 pass through the region enclosed by the square line 161, which is the soft magnetic material portion of the rotor north pole 7N, pass near the air gap indicated by the elliptical thick dashed line 17A, and pass through the teeth of the stator south pole 15L.

[0097] In FIG. 17B, if the A-phase current Ia is sufficiently large, and the magnetic flux density of the region surrounded by the square line 161, which is the soft magnetic material portion of the rotor N-pole magnetic pole 7N, and the tooth of the A-phase stator S-pole magnetic pole 15L is a magnetic flux φa1 of 2.0 [T] due to the magnetic fluxes 175 and 176, and the sum of the magnetic fluxes 171, 172, 173, and 174 is a magnetic flux φa2 of the same magnitude. In this case, the magnetic flux density Bagap of the air gap in the region indicated by the elliptical thick broken line 17A is 4.0 [T] when considered as a simple magnetic model, since the magnetic fluxes φa1 and φa2 overlap. In the region indicated by the elliptical thick broken line 17A, the magnetic resistance of not only the air gap portion but also the soft magnetic material portion near the air gap increases significantly. If the magnetic characteristics of a soft magnetic material are as shown in Figure 6, the relative permeability of the soft magnetic material for magnetic flux components exceeding 2.0 [T] approaches 1.0.

[0098] However, except for the region indicated by the thick elliptical broken line 17A, the magnetic flux density does not exceed 2.0 [T]. Therefore, if the A-phase current Ia is increased, a large amount of magnetomotive force [A] is applied to the region indicated by the thick elliptical broken line 17A, resulting in a large magnetic field strength [A / m] and an increase in magnetic flux density up to 4.0 [T]. The magnetic flux density of the soft magnetic material outside this 17A region is 2.0 [T] or less, and the relative permeability is large, such as a value of 100 or more, so the required consumed magnetomotive force, i.e., the excitation load, is relatively small. The thickness of the back yoke of the stator can be designed to be sufficiently large to keep the magnetic resistance small. However, in order to achieve a magnetic flux density of 4.0 [T] in the air gap with the A-phase current Ia, the magnetic path of the adjacent rotor poles 7K and 7S and the teeth of the adjacent stator poles 15J and 155 are used, so electromagnetic operation is not complicated if the stator poles 15J and 155 are not excited simultaneously with the excitation of the S stator pole 15L. Note that a driving method in which, when exciting a certain stator pole, both adjacent stator poles in the circumferential direction are not used at the same time, a driving method in which the size is reduced and limited and used simultaneously, or a driving method in which both adjacent stator poles are used simultaneously will be described later.

[0099] Next, the case where the stator S-pole magnetic pole 15L and the rotor S-pole magnetic pole 7S face each other through an air gap will be described with reference to (a) and (b) of FIG. 18. The rotor rotation angle is θr=-6°. The stator S-pole and the rotor S-pole face each other exactly, and the rotor rotation position is θr=-6° where magnetic flux is difficult to pass. (a) of FIG. 18 shows a state where A-phase current Ia is applied to the A-phase winding 15P and the A / phase winding 15Q, and A-phase magnetic flux φa shown by the thick dashed line with an arrow is excited. The A-phase magnetic flux φa shown by the 181 and each magnetic flux component shown by the slightly thinner dashed line of each permanent magnet of the stator and rotor are overlapped. The A-phase magnetic flux φa shown by the 181 can easily pass through because the teeth of the A-phase stator S-pole magnetic pole 15L on the stator side are in a reverse bias state of magnetic flux. However, in the region enclosed by square line 182 of rotor S pole 7S, the magnetic flux of permanent magnets 10A, 10B has the same direction, so the magnetic flux density is high and the magnetic resistance is large, making it difficult for the A-phase magnetic flux φa of 181 to pass through. As a result, the value of the A-phase magnetic flux φa of 181 becomes small.

[0100] FIG. 18(b) shows the magnetic flux distribution, which is drawn by qualitatively converting the superposed magnetic flux components in FIG. 18(a) into a distribution state. In FIG. 18(b), the A-phase magnetic flux φa component 184 passes from the rotor back yoke through the area surrounded by the rectangular line 183, which is the soft magnetic material part of the rotor S-pole magnetic pole 7S, through the tip of the A-phase stator S-pole magnetic pole 15L on the stator side, through the permanent magnet 15H, through the teeth of the stator N-pole magnetic pole 15J, and through the stator back yoke. The A-phase magnetic flux φa component 185 passes from the rotor back yoke through the area surrounded by the rectangular line 183, which is the soft magnetic material part of the rotor S-pole magnetic pole 7S, through the tip of the A-phase stator S-pole magnetic pole 15L on the stator side, through the permanent magnet 15S, through the teeth of the stator N-pole magnetic pole 155, and through the stator back yoke. The magnetic flux components 184 and 185 are small values ​​because the magnetic flux density in the region enclosed by the rectangular line 183 is already high and the magnetic resistance is large. Therefore, the magnetic flux component 187 of the permanent magnet 15H and the magnetic flux component 188 of the permanent magnet 15S remain, although they are slightly reduced. And, the tooth of the A-phase stator S-pole magnetic pole 15L is still magnetically reverse biased by the magnetic flux components 187 and 188.

[0101] Next, the state in which a torque in the CCW direction is generated will be described with reference to (a) and (b) of FIG. 19. The rotor rotation angle is θr=12°. Half of the rotor N-pole 7N and half of the rotor S-pole 7S face the A-phase stator S-pole 15L through an air gap. In (a) of FIG. 19, an A-phase current Ia is applied to the A-phase winding 15P and the A / phase winding 15Q, and the A-phase magnetic flux φa component 191 shown by a thick line with an arrow and passing through the rotor N-pole 7N, and the A-phase magnetic flux φa component 192 shown by a thin dashed line and passing through the rotor S-pole 7S pass through the A-phase stator S-pole 15L. These magnetic flux components 191 and 192 are illustrated overlapping with the magnetic flux components of each permanent magnet shown by dashed lines. The thick magnetic flux component 191 passes easily because the magnetic flux component of the permanent magnet is generated in the opposite direction in the area surrounded by a square line 196 of the rotor north pole 7N and is reverse biased. The thin dashed total magnetic flux component 192 is magnetic flux in the same direction as the magnetic flux component of the permanent magnet in the area surrounded by a square line 197 of the rotor south pole 7S, so the magnetic resistance is large and the amount of magnetic flux that passes through is small.

[0102] Figure 19(b) shows the magnetic flux distribution, which is drawn by qualitatively converting the superposed magnetic flux components in Figure 19(a) into a distribution state. The magnetic flux component 193 passes from the rotor back yoke through the area enclosed by square line 198 of rotor south pole 7K, through permanent magnet 77, through rotor north pole 7N, through the air gap, through the tip of A-phase stator south pole 15L on the stator side, through permanent magnet 15H, through the teeth of stator north pole 15J, and then through the stator back yoke. The magnetic flux component 194 passes from the rotor back yoke through the area surrounded by the square line 19A of the rotor south pole 7S, through the permanent magnet 7A, through the rotor north pole 7N, through the air gap, through the A-phase stator south pole 15L on the stator side, through the permanent magnet 15S, through the teeth of the stator north pole 155, and through the stator back yoke. The magnetic flux component 195 passes from the rotor back yoke through the area surrounded by the square line 19A, through the air gap from the rotor south pole 7S, through the A-phase stator south pole 15L on the stator side, through the permanent magnet 15S, through the teeth of the stator north pole 155, and through the stator back yoke. In this distribution state, there is no magnetic flux passing through the teeth of the A-phase stator south pole 15L, and the magnetic flux of the permanent magnets 15H and 15S is reverse biased. Therefore, there remains sufficient capacity to increase the A-phase current Ia and pass it through the stator S pole 15L.

[0103] Also, at this time, the magnetic flux component 193 is already a large value, and the magnetic flux component 19C of the permanent magnet 15H is a small value. Furthermore, when the A-phase current Ia increases or the rotor rotation angle θr increases, the magnetic flux component 193 increases, and the initial magnetic flux component 19C of the permanent magnet 15H disappears. The magnetic flux component 194 is also already a large value, and the magnetic flux component 19D of the permanent magnet 15S is a small value. Furthermore, when the A-phase current Ia increases or the rotor rotation angle θr increases, the magnetic flux component 194 increases, and the initial magnetic flux component 19D of the permanent magnet 15S disappears. Then, when the A-phase current Ia increases or the rotor rotation angle θr increases, a magnetic flux passing through the teeth of the A-phase stator S-pole magnetic pole 15L from the bottom to the top of the paper is generated and increases. At this time, it means that the magnetic flux passing through the teeth of the stator S-pole magnetic pole 15L becomes a positive value from a negative bias state value and increases. Moreover, since the magnetic flux density in the region enclosed by the rectangular line 19A is already high, the magnetic resistance is large and the magnetic flux component 195 has a small value.

[0104] At this time, if the A-phase current Ia is further increased, the magnetic flux is concentrated in the soft magnetic material of the stator south pole 15L and the rotor north pole 7N, and the air gap between them, as shown by the thick dashed circle in 19B, resulting in a large magnetic flux density. Except for the thick dashed circle area 19B, the margin for the magnetic flux of other magnetic paths is large and the magnetic resistance is small. Therefore, the magnetomotive force [A·turn] of the increase in the A-phase current Ia can be applied to the thick dashed circle area 19B, so a large magnetic field strength [A / m] can be given and excitation can be performed. Here, the thick dashed circle area 19B refers to the air gap, the soft magnetic material part near the tip of the stator magnetic pole where the magnetic flux density is 2.0 [T] or more and the magnetic permeability is significantly reduced, and the soft magnetic material part near the tip of the rotor magnetic pole. In the magnetic characteristics of Figure 6, when the magnetic flux density is 2.0 [T] or more, the relative magnetic permeability of the soft magnetic material decreases to a small value close to 1. However, if it is a very limited narrow area, it is possible to increase the A-phase current Ia [A·turn] to give a large magnetic field strength [A / m] to that narrow area, and for example, the magnetic flux density in that area can be made large, even 4.0 [T] or more. As will be explained later with equation (19) and other expressions, it is possible to obtain the force and torque that are the square of the magnetic flux density. For example, the magnetomotive force of an excitation current that increases a soft magnetic material part with a relative permeability of 1 and a length of 5 [mm] from a state of 2 [T] to 4 [T] is (4-2) / μo×0.005=7958 [A·turn]. μo is the vacuum magnetic permeability. The current at which a motor of 10 [kW] or more outputs its maximum torque is a realistic value as the magnetomotive force obtained by multiplying the winding current and the number of turns. At 4.0 [T], the force and torque are four times that of 2.0 [T]. Of course, magnetic flux densities between 2.0 and 4.0T can also be used; for example, at 3.0T the force and torque can be calculated as 2.25 times that of 2.0T.

[0105] Next, the change in magnetic flux and torque T when the motor in FIG. 14 rotates from rotor rotation angle θr=0° to 30° will be described. Assume that the motor rotates in a state where A-phase current Ia=Ia1 is applied to the A-phase winding 15P and the A / phase winding 15Q. The circumferential width of the air gap surface between the stator poles and the rotor poles is 30°. The rotor pole pitch is 36°. As in FIG. 1, the rotor rotation angle θr=0° is defined as the rotor rotation position immediately before the A-phase stator S pole 11 electromagnetically acts on the rotor N pole 1H to generate CCW torque. FIG. 5 shows an example where the rotor rotation position θr is 12°. Therefore, the rotor rotation angle in FIG. 18(b) is θr=-6°. The state of CCW rotation and θr=12° is shown in FIG. 19(b). The state of θr=30° when the rotor rotates further CCW is shown in either (a) or (b) of Figure 17, but considering the case where the current Ia1 is large enough to excite the magnetic flux, we use (b) of Figure 17. In this way, with the A-phase current Ia=Ia1 flowing, the rotor rotates from (b) of Figure 18 to (b) of Figure 19 and then to (b) of Figure 17, and the distribution of the magnetic flux changes.

[0106] Next, the magnetic flux density Ba of the teeth of the stator S-pole magnetic pole 15L and the interlinking magnetic flux φa of the A-phase winding 15P in each state are obtained. The rotor position θr is considered and compared for θr=0°, which is 6° from the position in FIG. 18(b), θr=15°, which is 3° from the position in FIG. 19(b), and θr=30° in FIG. 17(b). At θr=0°, which is 6° from -6° in FIG. 18(b), it is assumed that the magnetic flux density Ba1 in the area surrounded by the square line 183 in the initial state is reverse biased by the permanent magnet and is Ba1=-2.0[T]. Since the magnetic flux is saturated, the magnetic resistance is large, and the magnetic fluxes 184 and 185 are assumed to be 0[Wb] as an approximate value for a general explanation. The rotor radius is Rr and the axial length is Lr. The circumferential angle between the stator poles and rotor poles on the air gap surface is 30°, and the circumferential width, i.e., the circumferential length Lpcir [m], is given by the following equation. Lpcir = 30° × (2π / 180) × Rr (8) At this time, the teeth of the stator S pole 15L are reverse biased by the permanent magnets 15H and 15S, resulting in a negative magnetic flux and negative magnetic flux density Ba2. Now, assume that the negative bias value of Ba1 is exactly the magnetic flux density Ba2=-2.0[T]. At this time, the interlinkage magnetic flux φa1[Wb] of the A-phase winding 15P is given by the following equation. Also, the magnetic flux density Bgap1 and magnetic flux φgap1 in the air gap at this time are 0[Wb]. Bgap1=0 φgap1=0 (9) Ba1=-2.0 φa1=Ba1×30°×(2π / 180)×Rr×Lr =-2.0×30°×(2π / 180)×Rr×Lr (10)

[0107] 19(b), it is assumed that the current Ia1 is sufficiently large at θr=15°, which is 3° from (b) in the model, and the magnetic flux density Bgap2=+4.0[T] in the air gap with a circumferential width of 15° where the stator south pole 15L and the rotor north pole 7N face each other. The magnetic flux φgap2 passing through the air gap is expressed by the following equation. Bgap2=+4.0 φgap2=Bgap2×15°×(2π / 180)×Rr×Lr =4.0×15°×(2π / 180)×Rr×Lr (11) The magnetic flux φa2 of the teeth of the stator S pole 15L is the sum of the reverse bias magnetic flux components 19C and 19D by the permanent magnets 15H and 15S, φa1, and the above φgap2, which cancel each other out to become 0 [Wb]. The magnetic flux density Ba2 is also 0 [T]. Ba2=0 φa2=φa1+φgap2 =-2.0×30°×(2π / 180)×Rr×Lr+4.0×15°×(2π / 180)×Rr×Lr =0 (12)

[0108] 17(b), where θr=30°, it is assumed as a model that the magnetic flux density Bgap3=+4.0[T] in the air gap with a circumferential width of 30° where the stator south pole 15L and the rotor north pole 7N face each other. The magnetic flux φgap3 passing through the air gap is expressed by the following equation. Bgap3=+4.0 φgap3=Bgap3×30°×(2π / 180)×Rr×Lr =4.0×30°×(2π / 180)×Rr×Lr (13) The magnetic flux φa3 of the tooth of the stator S pole 15L is the sum of the reverse bias magnetic flux component φa1 and the φgap3, and is expressed by the following formula. φa3 = φa1 + φgap3 =-2.0×30°×(2π / 180)×Rr×Lr+4.0×30°×(2π / 180)×Rr×Lr =2.0×30°×(2π / 180)×Rr×Lr (14) At this time, the magnetic flux density Ba3 of the tooth is divided by the area Ss3 of the opposing stator pole to obtain the following equation. Ss3=Lpcir×Lr=30°×(2π / 180)×Rr×Lr Ba3 = φa3 / Ss3 =2.0×30°×(2π / 180)×Rr×Lr / (30°×(2π / 180)×Rr×Lr) =2.0 (15)

[0109] As described above, the magnetic flux density and magnetic flux of the air gap portion and the magnetic flux density and magnetic flux of the teeth of the stator S-pole magnetic pole 15L, i.e., the interlinkage magnetic flux φ of the A-phase winding 15P, when the rotor rotates with the rotor rotation position θr of 0°, 15°, and 30° with the A-phase current Ia=Ia1 being applied, are shown in equations (9) to (15). The interlinkage magnetic flux φ of the A-phase winding 15P changes from a negative value in equation (10) to 0 in equation (12) and a positive value in equation (14). On the other hand, it has been shown that the power supply to the motor when the motor in FIG. 14 rotates at a constant rotation speed ω is expressed by equation (3). Equation (3) shows that the power supply is proportional to the rate of change of the magnetic flux φ, not the magnitude of the magnetic flux φ. That is, for example, when the interlinkage magnetic flux changes to 0, 2, and 4, the power supply of equation (3) is the same as when the interlinkage magnetic flux changes to -2, 0, and 2. Therefore, it was shown that each magnetic path of the stator and rotor can be biased to negative magnetic flux using each permanent magnet as shown in Fig. 16 etc. Also, the magnetic characteristics of the soft magnetic material shown in Fig. 6 showed that torque can be generated by utilizing the change in magnetic flux density of 69 or 6A, which shows the change in magnetic flux density from negative to positive values, when driven by a unidirectional current.

[0110] Moreover, the value of the torque T [Nm] of a motor such as Figure 14 is given by equation (7). Δφ / Δθ is the angle change rate of the magnetic flux linking the windings, and is therefore the angle change rate of the values ​​of equations (10), (12), and (14) that increases with the rotor rotation angle θr. Although the magnetic flux in equation (12) is 0, as shown in Figure 11(b) where torque is generated at the rotor rotation angle θr = 12°, the torque generated is proportional to the angle change rate of the magnetic flux linking the windings, not the magnitude of the magnetic flux linking the windings. The torque formula (7) is a formula that indirectly estimates and calculates the torque from the power supplied to the motor, as shown in formulas (2) and (3). The internal loss in the motor is ignored.

[0111] The interlinkage magnetic flux φa of the A-phase winding 15P shown in equations (10), (12), and (14) is linked to the magnetic flux φgap of the air gap shown in equations (9), (11), and (13). As shown in Figure 18(a), Figure 16(b), and Figure 19(a), the interlinkage magnetic flux φa of the A-phase winding 15P is a value obtained by subtracting the reverse bias magnetic flux φbias of the permanent magnet of the stator from the magnetic flux φgap of the air gap. φa=φgap-φbias (16) If we assume that this reverse bias magnetic flux φbias is a constant value, then by substituting the magnetic flux φ in equation (7), we obtain the following equation: The torque T is proportional to the rate of change of the magnetic flux φgap in the air gap. T=Nw×I×Δ(φgap-φbias) / Δθ (17) =Nw×I×Δφgap / Δθ (18)

[0112] There is also a known method to directly calculate the electromagnetic force and torque using a method different from equations (7) and (17). This method involves modifying Maxwell's stress, and the circumferential force Fmaxwell [N] generated in the air gap of the motor is expressed by the following equation. Fmaxwell=Brad×Bcir / μo (19) Here, Brad [T] is the radial magnetic flux density component, Bcir [T] is the circumferential magnetic flux density component, and μo is the vacuum permeability. An example of Brad and Bcir is illustrated in Figure 20, which will be explained later. The circumferential force Fmaxwell [N] in equation (19) is obtained by observing the magnetic flux distribution and magnetic flux density in the air gap as a result of excitation with current, so equation (19) does not include the current value. The force Fmaxwell [N] can be considered and designed only from the magnetic flux distribution and magnetic flux density. This circumferential force Fmaxwell [N] is also a force density, and the motor torque T [Nm] is obtained by integrating the circumferential force Fmaxwell [N] generated in the air gap over one revolution and multiplying it by the rotor radius Rr [m] and the rotor axial length Lr [m].

[0113] It is also known that the attractive force generated between opposing N and S poles is proportional to the square of the magnetic flux density. Equation (19) also shows that the force Fmaxwell [N] is proportional to the square of the magnetic flux density. For example, if the magnetic flux density can be increased from 2.0 [T] to 4.0 [T], the force and torque will be four times as large. In principle, if it can be increased to 6.0 [T], the force and torque can be increased nine times. Therefore, if a motor configuration that can increase the magnetic flux density can be realized, a significant increase in torque can be expected. Also, the supplied power in equations (1) and (2), the torque equation in equation (7), and the Lorentz force in equation (20) show that if the magnetic flux density can be doubled as a result of doubling the excitation current, the supplied power, torque, and force will be four times as large, respectively. Although conventional motors may exceed 2.0 [T] in some places, the entire magnetic path of the motor is designed taking into account the magnetic saturation of the soft magnetic material, so even if a large current [A·turn] is passed through it, the magnetomotive force [A·turn] is consumed throughout the entire magnetic path, and the magnetic flux density in the air gap does not increase significantly. Therefore, the maximum torque of the motor is often limited by the magnetic saturation of the entire motor magnetic path.

[0114] In addition, when the magnetic flux density exceeds 2.0 [T], the magnetic flux density may not be increased in proportion to the excitation current due to the configuration of the magnetic circuit. For example, even if the magnetic flux density is increased 2.5 times from the magnetic flux density of 2.0 [T] to finally reach 4.0 [T], the torque will only be 4 times according to formula (19). This is the case when, for example, the magnetomotive force [A] of 0.5 times the excitation current is consumed in some magnetic resistance part through which the magnetic flux passes. In the motor of the present invention in FIG. 14, it is assumed that a very small part of the soft magnetic material magnetic path near the air gap exceeds the magnetic flux density of 2.0 [T] and becomes magnetically saturated. In addition, such magnetic energy is not converted into heat but is regenerated into electric energy.

[0115] FIG. 20 shows an example in which a portion of the torque generating portion of the motor in FIG. 14 is extracted and simplified. FIG. 20 shows a portion enlarged and the surrounding configuration is omitted. 201 in FIG. 20 is the stator south pole magnetic pole 15L in FIG. 19(b). 202 in FIG. 20 is the rotor north pole magnetic pole 7N in FIG. 19(b). In particular, the air gap portion is extremely enlarged for the purpose of explanation. 203 and 204 in FIG. 20 are the A-phase winding 15P in FIG. 19(b). Permanent magnets 15H, 15S, 77, 10A and their outer periphery are omitted. The magnetic flux distribution other than the air gap portion is shown simplified and in principle. Numerals 207, 208, and 209 indicate magnetic flux, 205 is the radial magnetic flux density component Brad [T] in equation (19), and 206 is the circumferential magnetic flux density component Bcir [T]. The direction of torque T is indicated by an arrow. As shown in equation (19) and Figure 20, the motor torque is indicated by the magnetic flux distribution in the air gap, and it can be confirmed that a torque proportional to the magnitude of the magnetic flux density can be obtained.

[0116] The value of equation (19) can be calculated by determining the magnetic flux density distribution in the air gap using finite element analysis (FEM) on a PC. However, this involves a large amount of calculations, making it difficult to determine by hand at a desk. Furthermore, the torque T calculated for the entire circumference of the motor using equation (7) and the torque T calculated using equation (19) are nearly identical. It can also be inferred from equation (19) that if the magnetic flux density in the air gap can be increased, an increase in torque T [T] can be expected. This can be said to be an effective torque evaluation method.

[0117] The force F [N], also known as the Lorentz force or Fleming's left-hand and right-hand rules, is expressed as the following equation for a winding with length Lr [m], number of turns Nw, and its current I [A], placed in a uniform magnetic flux density B [T]. The torque T is obtained by multiplying it by the rotor radius Rr. F = B × (Nw × I) × Lr (20) T = F × Rr =B×(Nw×I)×Lr×Rr (21) In addition, Δφ in equation (7) is expressed by equation (22) and substituted into equation (7). Δφ=B×Δθ×Rr×Lr (22) T = Nw × I × (Δφ / Δθ) =Nw×I×(B×Δθ×Rr×Lr) / Δθ (23) =B×(Nw×I)×Lr×Rr (24) Here, equation (21) and equation (24), which is a transformation of equation (7), are the same. In other words, the torque caused by a current of uniform magnetic flux density B [T] and the torque generated by the attractive force between salient poles as shown in Figure 20 are the same value. From the perspective of the rate of change of magnetic flux linkage, both conditions can be said to be the same. In addition, the torque calculated from Fmaxwell, which is the force density in equation (19), also becomes the same as equation (21) when transformed. Because these are expressions that observe the same physical phenomenon from different perspectives, they can be used to understand and evaluate conditions.

[0118] The motor of the present invention in Fig. 14 will be compared with the conventional switched reluctance motor in Fig. 63 from the viewpoints of torque T and magnetic flux density B. Assuming that the soft magnetic material has the magnetic characteristics shown in Fig. 6, the magnetic flux density Bgap of the air gap of the motor of the present invention in Fig. 14 can be increased up to 4.0 [T] in terms of the motor model, as shown in equations (11) and (13). In that case, the maximum torque of the motor of the present invention in Fig. 14 is four times that of the conventional switched reluctance motor in Fig. 63.

[0119] In addition, in the explanations of Figures 16 to 19, the amount of magnetic flux that reverse-biases the soft magnetic magnetic path with a permanent magnet was set assuming that the maximum value of the magnetic flux density in the area enclosed by the square lines of the stator south pole 15L teeth and rotor north pole 7N teeth in Figure 16(a) is -2.0 [T]. This is consistent with the assumed soft magnetic material magnetic characteristics in Figure 6. Equations (9) to (15) were explained based on this assumption.

[0120] Also, it is possible to further increase the magnetic flux amount of the permanent magnets of the stator and rotor in FIG. 14 and configure it so that magnetic flux overflows to the air gap side as in 106, 107, 108, and 109 in FIGS. 10 and 11. In that case, as can be understood from (b) in FIGS. 15 and 17, by passing a large current through the exciting winding 15P of the A-phase stator S-pole magnet 15L and the exciting winding 15Q in the A / -phase, the stator magnets 15J and 155 adjacent to both sides of the stator S-pole magnet 15L can be utilized. Therefore, the magnetic flux passing through these soft magnetic bodies becomes up to three times the maximum, and the air gap magnetic flux density Bgap [T] becomes 6.0 [T] in a simple theoretical calculation. In that case, the maximum torque of the motor of the present invention in FIG. 14, from equation (19), compared to the conventional switched reluctance motor in FIG. 63, can obtain a torque nine times as large in a simple calculation. Note that the size and cost of the motor are often designed under the most severe driving conditions rather than the motor efficiency under light load. Therefore, for example, in the case of the main motor of an electric vehicle, climbing a steep slope is severe, so a region of high torque at low speed is required. There are many such applications, and for miniaturization, weight reduction, and cost reduction of the motor, the maximum torque, the power factor at that time, the torque constant, losses such as copper loss, and efficiency are important. The motor in FIG. 14 has an advantage in maximum torque as described above.

[0121] As can be seen from the configurations of Figures 14, 15, and 17(b), when two stator poles of one phase are excited, each uses two adjacent teeth on both sides, totaling six teeth, to pass magnetic flux and generate torque. Figure 17(b) shows the distribution of magnetic flux at the rotor rotation position where the magnetic flux acting is maximum, and is a diagram for analyzing the magnitude of magnetic flux in each magnetic path. Here, the maximum torque of the motor does not refer to the maximum torque at a specific partial angle of the rotor rotation angle θr as in the example of Figure 64, but refers to the maximum value of the average torque when the rotor makes one rotation. In order to increase the maximum torque, that is, the average torque per rotation, it is necessary to configure the magnetic circuit of the motor to be effectively utilized so that the magnetic circuit of the motor, except for the area around the air gap, is not magnetically saturated, as shown in Figure 17(b). Furthermore, in the conventionally widely used built-in magnet synchronous motor IPMSM, in the operating range of maximum torque or in the operating range of high speed rotation where field weakening control is performed for constant output control, there is a problem that a phase difference occurs between the voltage phase and current phase of each winding due to the effect of armature reaction, resulting in a decrease in power factor. As a result, there is a problem that the motor copper loss increases and the current capacity of the transistors in the drive circuit increases, which causes the motor to become larger and costlier. All conventional motors have restrictions and limits on maximum torque.

[0122] Also, when the required motor torque is small, the current value is naturally small, and the magnetic flux density of the stator pole tips and rotor pole tips near the air gap does not exceed the saturation magnetic flux density of 2.0 [T]. When a large torque is required, the current is increased, and the magnetic flux density of the area 19B indicated by the dashed circle in FIG. 19(b) increases. As mentioned above, if the reverse bias permanent magnet has sufficient performance, the magnetic flux density of the area 19B can be as large as 4.0 [T] to 6.0 [T]. In other words, until the area 19B reaches 6.0 [T], the magnetic flux density of the magnetic paths of the soft magnetic material other than the area 19B is 2.0 [T] or less, and the relative permeability is large in the magnetic characteristics assumed in FIG. 6, so that the magnetic flux can pass through without difficulty.

[0123] Above we have shown the possibility of generating large torque by increasing the magnetic flux density in the air gap. However, magnetic circuits that exceed the saturation magnetic flux density of soft magnetic materials pose problems such as leakage flux, demagnetization of permanent magnets, and magnetic energy (B·H / 2) in areas with high magnetic flux density, and they also pose challenges. For example, in the case of main motors for electric vehicles, forward torque is mainly used, so a structure that can effectively generate forward torque can be used. Various permanent magnets can be used. Low iron loss amorphous steel sheets can also be used, and permendur steel sheets with high magnetic flux density can also be used in parts. Higher speeds can also be achieved by using power MOSFETs, SiC, GaN, and other power elements.

[0124] Next, examples of the shape of the rotor permanent magnet and its vicinity are shown and explained in Fig. 21 and Fig. 22. The motor in Fig. 14 has a 6S10R configuration with six stator poles and ten rotor poles, and a configuration of one stator pole pair is shown to make the basic configuration and operation of the motor easier to understand. However, in the design of a motor with a diameter of over 200 mm, the number of stator pole pairs can be increased to reduce the size. However, if the number of rotor poles increases, there will be a limit to high-speed rotation due to increased iron loss and the frequency limit of the current control of the drive circuit. Here, we assume that the motor in Fig. 14 is a 24S40R motor with four pole pairs, and show an example of a rotor shape with 40 rotor poles in Fig. 21. 211 is the rotor shaft, 212 is a permanent magnet, 213 is a north pole, and 214 is a south pole. The magnetic poles of each permanent magnet are oriented in the polarity direction of each rotor pole.

[0125] Next, various examples of rotor magnetic poles are shown and explained in Fig. 22(a) to (f). These are partial enlarged views of the dashed circle portion shown in Fig. 215. 221 in Fig. 22(a) is a permanent magnet. The direction of magnetic flux of each permanent magnet is shown by the direction of the arrow drawn on the magnet. 222 is a south magnetic pole, and 22H is a north magnetic pole. The shape of the south magnetic pole 223 in Fig. 22(b) is not symmetrical in the front and rear in the circumferential direction, and the characteristics of CCW torque and CW torque are different. In a motor for which the torque in one direction is mainly important, it is possible to emphasize the torque in one direction and sacrifice the torque characteristics in the opposite direction to some extent. The shape of the permanent magnet 224 in Fig. 22(c) is thicker in the circumferential direction than the permanent magnet 225. In the motor in Fig. 14, a large current is passed through to generate a large torque, so a large magnetomotive force H [A / m] acts on the permanent magnets, especially those near the air gap. By increasing the circumferential thickness like the shape of permanent magnet 224, it is possible to improve the demagnetization resistance performance. The shape of permanent magnet 226 in Fig. 22(d) is a shape in which the circumferential thickness is large on the air gap side and decreases toward the inner diameter side. Like permanent magnet 224, the air gap side of permanent magnet 226 has a shape that makes it difficult to demagnetize. The shape of permanent magnet 226 can be modified, such as a shape intermediate between permanent magnets 224 and 225.

[0126] Also, 227 in FIG. 22(d) is an example in which the shape of the S pole magnetic pole is a convex shape, and it can be a rectangular shape or a trapezoidal shape. The shape of the S pole magnetic pole 229 in FIG. 22(e) is an arc shape. The magnetic pole shape of the rotor can be changed. The permanent magnet 228 in FIG. 22(e) and the permanent magnet 22A have different thicknesses in the circumferential direction and are arranged separately in the radial direction. 22B and 22C are spaces and may be non-magnetic materials such as resin. 22D, 22E, 22F, and 22G in FIG. 22(f) show magnets of different types and characteristics. Some of these magnets may be spaces or non-magnetic materials such as resin. The S pole magnetic pole 22K has a slit 22J, which is a long and narrow space. The torque characteristics can be changed by changing the magnetic flux distribution inside the S pole magnetic pole 22K. The CCW torque and CW torque characteristics can also be changed by changing the direction of the slit 22J and arranging it diagonally. The number and shape of the slits 22J may be changed, and the slits 22J may be replaced with permanent magnets.

[0127] In the motor of the present invention, the space on the rotor side may be relatively insufficient. From this viewpoint, the design freedom of the rotor can be improved by arranging the rotor on the outer diameter side and the stator on the inner diameter side, a so-called outer rotor structure. In this case, since the motor structure has a relatively large stator slot cross-sectional area, the slot cross-sectional area, i.e., the winding space, can be secured even if the stator is arranged on the inner diameter side. Also, in a so-called axial gap type motor configuration in which the stator and rotor are arranged opposite each other in the rotor axial direction, the stator side and the rotor side can be arranged magnetically equivalently.

[0128] It has been shown that the magnetic flux density of the air gap of the motor of the present invention shown in FIG. 14 can be increased up to 4.0 [T] and a large torque can be generated. Furthermore, from the magnetic flux limit of the magnetic circuit that goes around, a theoretically simple calculation shows that a magnetic flux density of 6.0 [T] is possible. Although FIG. 14 shows an example of concentrated winding, different characteristics can be achieved by using full-pitch winding, and this technology will be explained later. Although an example of three phases is shown, five, seven, nine, 11, 13, etc. are possible, and in that case, a specific number of rotor poles will show particularly excellent characteristics. When the number of phases is a prime number and is large, there is an effect of canceling out the harmonic components of the generated force, making it easier to reduce noise.

[0129] Next, an embodiment of claim 3 will be described. Claim 3 is a motor with a so-called concentrated winding configuration, which is wound around the teeth of the stator poles 11 and 12 in Fig. 1 and Fig. 14. One of the features of Fig. 1 and Fig. 14 is that, as mentioned above, the passing magnetic flux of the rotor poles can be increased, so that the torque can be increased by reducing the magnetic resistance of the rotor poles. In addition, as shown in Fig. 13, there is also a feature that the degree of freedom of the current waveform during rotation is increased compared to the conventional motor in Fig. 63, such as making the current waveform trapezoidal. In addition, compared to full-pitch winding, in which the winding pitch of concentrated winding spans multiple stator poles, winding production is easier, the space factor of the winding can be easily improved, and the motor can be made smaller. In addition, compared to full-pitch winding, the axial protrusion length of the coil end can be made smaller. Since the motor length can be reduced, it is excellent in terms of miniaturization. On the other hand, there are also problems, and issues to be solved will be explained one by one.

[0130] Next, to explain the operation of the motor in Fig. 14, Fig. 23 shows the rotor rotational position θr = 0°, with the A-phase magnetic flux component φa, B-phase magnetic flux component φb, and C-phase magnetic flux component φc superimposed. As shown in Fig. 23, the rotor starting point, rotational position θr = 0°, is the position where the lower right corner of the A-phase stator S-pole 11 faces the upper left corner of the rotor N-pole 1H across an air gap. This rotor rotational position θr is the rotational position θr where the A-phase stator N-pole 11 starts to generate torque.

[0131] As described above, when the voltage of the A-phase concentrated winding 1A and the A / -phase concentrated winding 1D in Fig. 23 are connected in series, the sum of these voltages, the A-phase voltage Va, is given by the following equation. For example, the flux linkage of the A-phase concentrated winding 1A is (φa-φbiasa), and the flux linkage of the A / -phase concentrated winding 1D is (φa-φbiasa / ). Also, as explained in Fig. 18, the passing flux in the portion where the S pole 11 of the stator faces the S pole 1J of the rotor is small because they are the same S pole, so here we will simplify by assuming that there is no passing flux. Va=Nw / 2×d(φa-φbiasa) / dt+Nw / 2×d(φa-φbiasa / ) / dt (31) Similarly, the B-phase voltage Vb and the C-phase voltage Vc are expressed by the following equations. Vb=Nw / 2×d(φb-φbiasb) / dt+Nw / 2×d(φb-φbiasb / ) / dt (32) Va=Nw / 2×d(φc-φbiasc) / dt+Nw / 2×d(φc-φbiasc / ) / dt (33) Here, the number of turns of each concentrated winding is Nw / 2 [turns]. φbiasa is the bias flux of the tooth of the A-phase stator S-pole 11 due to the permanent magnets 145, 146 arranged on the side of the A-phase stator S-pole 11. φbiasa / is the bias flux of the tooth of the A / phase stator N-pole 14 due to the permanent magnets 148, 149 arranged on the side of the A / phase stator S-pole 14. Similarly, φbiasb is the bias flux of the tooth of the B-phase stator S-pole 13. φbiasb / is the bias flux of the tooth of the B / phase stator N-pole 16. Similarly, φbiasc is the bias flux of the tooth of the C-phase stator S-pole 15. φbiasc / is the bias flux of the tooth of the C / phase stator N-pole 12.

[0132] Now, if we assume that the values ​​of the bias magnetic fluxes φbiasa, φbiasa / , φbiasb, φbiasb / , φbiasc, and φbiasc / do not change and are constant, equations (31), (32), and (33) can be simplified using assumed values ​​Vak, Vbk, and Vck, resulting in the following equations. Vak = Nw × dφa / dt (34) Vbk = Nw × dφb / dt (35) Vck = Nw × dφc / dt (36) Incidentally, as described above, the interlinked magnetic flux of the concentrated winding 1A of phase A is (φa - φbiasa), the interlinked magnetic flux of the concentrated winding 1C of phase B is (φc - φbiasc), and the interlinked magnetic flux of the concentrated winding 1E of phase C is (φc - φbiasc). In this specification, equations (34), (35), and (36) are used for relationships with the voltages of the full-pitch windings described later, etc. This is the relationship between FIG. 23 of the concentrated winding and FIGS. 26 and 27 of the full-pitch winding. Also, in the case of the motor with the concentrated winding in FIG. 1, since the stator does not have a permanent magnet for reverse bias, there is no such bias magnetic flux, and equations (34), (35), and (36) hold true.

[0133] For the magnetic flux φa [Wb] of phase A in FIG. 23, when the rotor rotation angle θr rotates CCW from 0° to 30°, the current flowing through the phase A winding 1A and the phase A / winding 1D is a constant value Io, and the magnetic flux density in the air gap portion is a constant value Bo, the following equation holds in the range of θr from 0° to 30°. φa = Bo · θr · Rr · Lr (37) The circumferential angle width of the rotor poles such as 1H in FIG. 23 is set to 30°, and the circumferential width of the permanent magnet such as 231 is set to 6°. Let the total number of turns of both windings 1A and 1D be Nw, and the voltage Vak [V] across both windings connected in series is given by the following equation from equation (34). ω is the rotational angular velocity [rad / sec]. Vak = Nw × dφa / dt (38) = Nw × d(Bo · θr · Rr · Lr) / dt (39) = Nw × d(Bo · θr · Rr · Lr) / dθr · dθr / dt = Nw × (Bo · Rr · Lr) · ω (40) Thus, under the said driving conditions, the phase A voltage Vak in equation (40) is a voltage proportional to the magnetic flux density Bo [T] and the rotational angular velocity ω [rad / sec].

[0134] Here, an example of the current and voltage when the currents Ia, Ib, and Ic of each phase are set to the values ​​shown by the dashed lines in (a), (b), and (c) of FIG. 13 is shown in FIG. 24 and will be described. It is assumed that the magnetic flux density Bo in Equation (40) is 2.0 [T] or less, and that the soft magnetic material is not magnetically saturated according to the characteristics in FIG. 6. (a) of FIG. 24 shows the A-phase current Ia, which is a current that flows in synchronization with the CCW rotation from the state in FIG. 23, and the current amplitude is normalized to 1.0. The A-phase current Ia increases when the rotor rotation angle θr is between 0° and 6°, is a constant value of 1.0 between 6° and 24°, decreases to 0 between 24° and 30°, and is a constant value of 0 from 36° to 72°, repeating these values ​​in a 72° cycle. The B-phase current Ib in FIG. 24(b) has the same current waveform with a phase delay of 48° relative to the A-phase current. The C-phase current Ic in FIG. 24(c) has the same current waveform as the A-phase current but with a phase delay of 24°.

[0135] FIG. 24(d) shows the A-phase voltage Vak, which corresponds to the values ​​of equations (34) and (40). The A-phase voltage Vak rotates at a constant rotational speed in the CCW direction from the state shown in FIG. 23, and the voltage waveform is synchronized with the rotation. When the rotor rotation angle θr is between 0° and 6°, the area where the stator's S pole 11 and the rotor's N pole 1H face each other increases, and at the same time, the A-phase current Ia in FIG. 24(a) also increases linearly, so that the A-phase magnetic flux φa that links with the A-phase concentrated winding 1A and the A / phase concentrated winding 1D in FIG. 23 increases as a square function. The A-phase voltage Vak increases linearly as a linear function from equation (34). When θr is between 6° and 24°, the A-phase current Ia is a constant value, so it becomes the voltage of equation (40). When θr is between 24° and 30°, the area of ​​the stator's S pole 11 and the rotor's N pole 1H facing each other increases, and at the same time, the A-phase current Ia in FIG. 24(a) decreases linearly, resulting in a large negative voltage and waveform shape as shown in the figure. This is a phenomenon in which torque generation and regeneration of magnetic energy to the power supply overlap. From 36° to 72°, the value is constant at 0, and these values ​​are repeated in a 72° cycle. The B-phase voltage Vbk in FIG. 24(e) has the same voltage waveform with a phase delay of 48° relative to the A-phase voltage Vak. The C-phase voltage Vck in FIG. 24(f) has the same voltage waveform with a phase delay of 24° relative to the A-phase voltage Vak.

[0136] When the negative regenerative voltages in (d), (e), and (f) of FIG. 24 become large, a problem occurs in which the current control is restricted. When this negative voltage exceeds the power supply voltage of the drive circuit, the regenerative time becomes longer because it is restricted by the power supply voltage. In that case, negative torque may occur, and the average torque decreases. Alternatively, if the timing of the current decrease is advanced, the positive torque generation time is shortened, and the average torque decreases. Note that the regenerative voltage increases when the magnetic flux of each phase increases. If the current decrease time in FIG. 24 is shortened, the regenerative voltage increases. If the rotor rotation speed increases, the regenerative voltage increases. One method of reducing this regenerative voltage problem is to constantly excite the front stator pole, which will be explained in detail later. Also, in the case of full-pitch windings, the negative regenerative voltage in FIG. 24 is also generated in the windings of other phases, which becomes a bigger problem. The phenomenon and the solution will be explained later.

[0137] Moreover, the A-phase torque Ta [Nm] generated by the A-phase and A / -phase when the rotor rotation angle θr in FIG. 24 is between 0° and 30° is expressed by the following equation from equations (1) and (4). Ta = Vak Io / ω =Nw×(Bo・Rr・Lr)・ω・Io / ω (41) = Nw × (Bo Rr Lr) Io (42) Thus, under the above driving conditions, it can be confirmed that the A-phase torque Ta [Nm] in equation (42) is proportional to the magnetic flux density Bo [T]. In particular, in the motor of the present invention, a large magnetic flux density such as 4.0 [T] is used near the maximum torque of the motor.

[0138] Furthermore, the magnetic flux density Bo in equation (42) is also related to the current Io, i.e., the A-phase current Ia, and in the region where the magnetic flux density of the magnetic characteristics in Figure 6 is proportional to the current, the torque value in equation (42) is proportional to the square of the current. As the magnetic flux density approaches 2.0 [T] and increases, the magnetic flux density near the air gap increases and the magnetic resistance changes significantly, resulting in a nonlinear torque characteristic with respect to the current Io. In other words, the relationship between the phase currents Ia, Ib, and Ic and the magnetic flux density of each phase is nonlinear, so it cannot be expressed by a simple formula. In any case, the torque T follows the value of equation (19) in the air gap section.

[0139] Here, equations (1), (4), and (42) are hypothetical equations that ignore not only iron loss and copper loss, but also magnetic energy in the motor. Although these are hypothetical, the problem can be solved by clarifying the general qualitative relationship. As mentioned above, the problem is how to transfer magnetic energy in the motor between the inverter side and the motor side, particularly in the motor of the present invention. The magnetic energy and the method of transferring it will be explained later. Note that the B-phase magnetic flux φb and the B-phase torque Tb have the same relationship as in FIG. 24, and have a characteristic in which the phase lags 48° with respect to the A-phase. The C-phase magnetic flux φc and the C-phase torque Tc have the same relationship as in FIG. 24, and have a characteristic in which the phase lags 24° with respect to the A-phase.

[0140] Next, an example of a drive circuit for supplying each current to each winding in FIG. 23 is shown in FIG. 25 and will be described. 25A is a motor control circuit. 25B is a DC power supply that outputs a positive voltage Vp and a negative voltage Vn. 257 is a winding in which the A-phase concentrated winding 1A and the A / phase concentrated winding 1D are connected in series. 251 is a transistor, the collector of which is connected to a positive voltage Vp and the emitter of which is connected to one end of the winding 257. A regenerative diode is connected between the emitter of the transistor 251 and the negative voltage Vn. 252 is a transistor, the collector of which is connected to the other end of the winding 257 and the emitter of which is connected to a negative voltage Vn. A regenerative diode is connected between the collector of the transistor 252 and the positive voltage Vp. 258 is a winding in which the B-phase concentrated winding 1C and the B / phase concentrated winding 1F are connected in series. Reference numeral 253 denotes a transistor, the collector of which is connected to a positive voltage Vp, and the emitter of which is connected to one end of the winding 258. A regenerative diode is connected between the emitter of the transistor 253 and the negative voltage Vn. Reference numeral 254 denotes a transistor, the collector of which is connected to the other end of the winding 258, and the emitter of which is connected to the negative voltage Vn. A regenerative diode is connected between the collector of the transistor 254 and the positive voltage Vp. Reference numeral 259 denotes a winding in which the C-phase concentrated winding 1E and the C / phase concentrated winding 1B are connected in series. Reference numeral 255 denotes a transistor, the collector of which is connected to a positive voltage Vp, and the emitter of which is connected to one end of the winding 259. A regenerative diode is connected between the emitter of the transistor 255 and the negative voltage Vn. Reference numeral 256 denotes a transistor, the collector of which is connected to the other end of the winding 259, and the emitter of which is connected to the negative voltage Vn. A regenerative diode is connected between the collector of the transistor 256 and the positive voltage Vp.

[0141] For example, the trapezoidal A-phase current Ia shown in Fig. 24(a) is supplied by controlling the transistors 251 and 252 by PWM control or the like. For example, the trapezoidal B-phase current Ib shown in Fig. 24(b) is supplied by controlling the transistors 253 and 254 by PWM control or the like. For example, the trapezoidal C-phase current Ic shown in Fig. 24(c) is supplied by controlling the transistors 255 and 256 by PWM control or the like. The motor control is generally performed as follows: a torque command Tc is calculated from a speed error calculated by speed control; a current amplitude Io corresponding to the torque command Tc is calculated; and the phase currents Ia, Ib, and Ic shown in Fig. 13 are supplied and controlled according to the current amplitude Io.

[0142] Next, we will explain the magnetic energy inside the motor in Figure 23. Generally, the density of magnetic energy Em in space [J / m 3 ] is expressed as follows, where B [T] is the magnetic flux density and H [A / m] is the magnetic field strength. Em=B·H / 2 (43) For example, even with the same magnetic flux density, if the relative permeability of the soft magnetic material is large, the required magnetic field strength H [A / m] is small, and the magnetic energy Em of that part is small. The relative permeability of the air gap part and the like is small at 1, so the magnetic energy Em of that part is large. In the motor of the present invention, the air gap part and its vicinity are made to have a large magnetic flux density, even though it is partial, so a large magnetic energy Em is accumulated and is repeatedly exchanged between the motor and the inverter. As mentioned above, when the rotation speed increases, there is a problem related to the regeneration of magnetic energy. Also, if the attractive force between the stator and rotor changes suddenly, there are problems with vibration and noise. Examples of solutions will be explained later.

[0143] Next, the utilization rate of concentrated windings such as those in FIGS. 1, 14, and 23 will be described. As shown in FIG. 24, the concentrated winding 1A of phase A and the concentrated winding 1D of phase A / are energized with the phase A current Ia to drive, and then similarly, phases B and C are energized to drive and rotate the rotor. At this time, the energization period for each winding is about 1 / 3 of the entire period, and it can be said that the utilization rate of the concentrated winding is about 1 / 3. Since the winding space of the motor is limited, the thickness and number of turns of the winding are limited. And when the winding utilization rate is low, for example, 1 / 3, it is necessary to pass a current three times that in the case where the winding utilization rate is 3 / 3. The copper loss is 3 2 / 3 = 3, resulting in a value three times larger. Thus, improving the utilization rate of the winding is important for reducing copper loss, improving efficiency, miniaturizing, lightening, and reducing the cost of the motor. The method for improving the utilization rate of the winding will be described later.

[0144] When the utilization rate of the winding is low, the utilization rate of the driving transistor also decreases. And as described above, since the current value increases, it is necessary to increase the current capacity of the transistor, resulting in problems such as the enlargement of the inverter and cost issues. Examples of solutions to this will be described later. It includes the motor configuration of the number of stator poles and rotor poles, the configuration of the drive circuit, the current energization method, etc.

[0145] Next, an embodiment of claim 4 will be described with reference to FIG. 26. In the motor of FIG. 26, the concentrated windings of each phase of FIG. 14 are replaced with full-pitch windings. The rest of the motor configuration is the same. 261 and 262 are AB-phase windings, which are full-pitch windings with a winding pitch of 180° electrical angle, which is half the electrical angle of 360° of the stator 1 pole pair, and their coil ends are indicated by 267 shown by a thick dashed line. AB-phase current Iab is passed through the AB-phase windings. The positive winding portion of the concentrated winding of the A-phase winding 1A in FIG. 14 and the positive winding portion of the concentrated winding of the B / phase winding 1F are arranged in the same slot, but the positive winding portion of the full-pitch winding of the AB-phase winding in FIG. 26, which is the positive winding portion 261, integrates the two windings. The negative winding portion of the concentrated winding of the A / phase winding 1D in Fig. 14 and the negative winding portion of the concentrated winding of the B / phase winding 1C in Fig. 26 are arranged in the same slot, but the negative winding portion 262 of the full-pitch winding of the AB-phase winding in Fig. 26 integrates both windings. The AB-phase windings 261 and 262 in Fig. 26 each occupy one slot, and the cross-sectional area of ​​the winding can be doubled, so that the winding resistance in the slot can be reduced to half compared to the winding resistance of the concentrated winding in Fig. 14. The operation of the AB-phase windings 261, 267, and 262 will be described later, but they are referred to as AB-phase windings because they are involved in the operation of both the A-phase stator south pole magnetic pole 11 and the A / phase stator north pole magnetic pole 14 and the B-phase stator south pole magnetic pole 13 and the B / phase stator north pole magnetic pole 16, that is, they are involved in the electromagnetic operation of both the A-phase and the B-phase.

[0146] Similarly, 263 and 264 are BC-phase windings, which are full-pitch windings with a winding pitch of 180° electrical angle, which is half the electrical angle of 360° of the stator 1 pole pair, and their coil ends are indicated by 268, which is indicated by a thick dashed line. A BC-phase current Ibc is applied to the BC-phase winding. The positive winding portion of the concentrated winding of the B-phase winding 1C in FIG. 14 and the positive winding portion of the concentrated winding of the C / phase winding 1B are arranged in the same slot, but the positive winding portion of the full-pitch winding of the AB-phase winding in FIG. 26, which is the positive winding portion, is integrated together. The negative winding portion of the concentrated winding of the B / phase winding 1F in FIG. 14 and the negative winding portion of the concentrated winding of the C / phase winding 1E in FIG. 26, which is the positive winding portion, is integrated together. The winding resistance of the BC-phase windings 263 and 264 in Fig. 26 can be reduced to 1 / 2 compared to the winding resistance of the concentrated winding in Fig. 14. Since the BC-phase windings 263, 268, 264 are involved in the electromagnetic operations of both the B phase and the C phase, these full-pitch windings are referred to as BC-phase windings.

[0147] Similarly, 265 and 266 are CA-phase windings, which are full-pitch windings with a winding pitch of 180° electrical angle, which is half the electrical angle of 360° of the stator 1 pole pair, and their coil ends are indicated by 269, which is indicated by a thick dashed line. A CA-phase current Ica is applied to the CA-phase winding. The positive winding portion of the concentrated winding of the C-phase winding 1E in FIG. 14 and the positive winding portion of the concentrated winding of the A / phase winding 1D are arranged in the same slot, but the positive winding portion of the full-pitch winding of the CA-phase winding in FIG. 26, which is the positive winding portion, is integrated together. The negative winding portion of the concentrated winding of the C / phase winding 1B in FIG. 14 and the negative winding portion of the concentrated winding of the A / phase winding 1A are arranged in the same slot, but the positive winding portion of the full-pitch winding of the BC-phase winding in FIG. 26, which is the positive winding portion, is integrated together. The winding resistance of the CA-phase windings 265 and 266 in Fig. 26 can be reduced to 1 / 2 compared to the winding resistance of the concentrated winding in Fig. 14. Note that the BC-phase windings 265, 269, 266 are involved in the electromagnetic operations of both the C-phase and A-phase, and therefore these full-pitch windings are referred to as CA-phase windings.

[0148] Next, the relationship between the currents Iab, Ibc, and Ica of the full-pitch winding of the motor in FIG. 26 and the currents Ia, Ib, and Ic of the concentrated winding of the motor in FIG. 14 will be shown. Iab = Ia + Ib (44) Ibc=Ib+Ic (45) Ica=Ic+Ia (46) When each phase current is applied based on this relationship, the same magnetomotive force acts on each part of the stator and rotor, generating the same torque in the rotor, in the motor with full-pitch windings in Fig. 26 and the motor with concentrated windings in Fig. 23. The circular path and location on which the magnetomotive force acts can be determined for each current based on Ampere's law of circular integration. In other words, the magnetomotive force of each current acts on the part of the circular path with the largest magnetic resistance, and a force is generated in the direction that reduces the magnetic resistance.

[0149] Next, the relationship between the voltage and magnetic flux of the full-pitch winding in Fig. 26 will be explained. The number of turns of the full-pitch windings such as 261 is Nw / 2 [turns], the same as the A-phase concentrated winding 1A in Fig. 23, and the number of turns is under the same conditions. As shown in Fig. 26, there is a magnetic flux component of a permanent magnet, and each full-pitch winding is linked with magnetic fluxes φa, φb, and φc of each phase, making the voltage complex. The voltage Vab of the AB-phase windings 261, 267, and 262 is expressed by the following equation from Faraday's law of electromagnetic induction. Vab=Nw / 2×d(φa-φbiasab-φbiasab / +φb-φc) / dt (47) 26, and is the magnetic flux component generated by permanent magnet 145. φbiasab / is the magnetic flux component 26A that links with the winding 262 in FIG. 26, and is the magnetic flux component generated by permanent magnet 148. If we assume that the bias magnetic fluxes φbiasab and φbiasab / do not change and are constant values, then similar to Vak in equation (34), equation (47) can be simplified as an assumed value Vabk, resulting in the following equation. Vabk=Nw / 2×d(φa+φb-φc) / dt (48)

[0150] Similarly, the simplified assumed voltage Vbck of the BC phase windings 263, 268, and 264 and the simplified assumed voltage Vcak of the CA phase windings 265, 269, and 266 are expressed by the following equations. Vbck=Nw / 2×d(-φa+φb+φc) / dt (49) Vcak=Nw / 2×d(φa-φb+φc) / dt (50) Moreover, expressing it in terms of Vak in equation (34), Vbk in equation (35), and Vck in equation (36) gives the following equation: The voltages of these full-pitch windings are complex voltages that include the phase voltages Vak, Vbk, and Vck in the case of concentrated windings. Vabk = (Vak + Vbk - Vck) / 2 (51) Vbck = (-Vak + Vbk + Vck) / 2 (52) Vcak=(Vak-Vbk+Vck) / 2 (53)

[0151] Next, examples of currents and voltages for driving Iab in equation (44), Ibc in equation (45), and Ica in equation (46) using drive circuit diagram 25 are shown in FIG. 28, and problems associated with these will be described. When driving the motor in FIG. 26 using drive circuit diagram 25, winding 257 is the A-phase full-pitch winding 261, 267, and 262 in FIG. 26. Similarly, winding 258 is the B-phase full-pitch winding 263, and winding 259 is the C-phase full-pitch winding 265. Now, currents the same as currents Ia, Ib, and Ic of the concentrated winding windings shown in FIG. 24 are passed as Iab, Ibc, and Ica of full-pitch windings 261, 263, and 265 in accordance with the relationships of equations (44), (45), and (46). At this time, the same current [A·turn] is passed through each slot of the concentrated winding motor in FIG. 23 and the full-pitch winding motor in FIG. 26. Therefore, the torque at rest when the rotor is not rotating is the same.

[0152] FIG. 28 shows the currents Iab, Ibc, and Ica of the full-pitch windings of FIG. 26, which are related by equations (44), (45), and (46). This is a specific example of current flow, and can be drawn from (a), (b), and (c) of FIG. 24. Also, the voltages Vabk, Vbck, and Vcak of the full-pitch windings of FIG. 26 are related by equations (51), (52), and (53), so for this example, they can be drawn from (d), (e), and (f) of FIG. 24, resulting in (d), (e), and (f) of FIG. 28. As mentioned above, the magnetic fluxes of all phases are linked in each full-pitch winding, and voltages related to the magnetic fluxes of other phases are also generated. Conversely, the current of each full-pitch winding applies a magnetomotive force [A] to the magnetic circuits of the stator poles of all phases. In addition, conventional motors with full-pitch windings that use three-phase sinusoidal AC often have motor characteristics that are expressed by relatively simple equations based on the so-called three-phase AC theory, using linear theory. However, motors such as those shown in Figures 23 and 26 have stator poles that are arranged discretely in the circumferential direction and are driven sequentially for each drive step, so it is necessary to consider the generated voltage and the magnetomotive force that acts on each operation.

[0153] Here, consider the AB phase voltage Vabk in FIG. 28(d) when the rotor rotation angle θr is from 48° to 54°, which is the current increase section of the AB phase current Iab in FIG. 28(a). In this section, a large voltage is generated in the AB phase voltage Vabk in FIG. 28(d), and it is difficult to increase the AB phase current Iab. As can be seen from FIG. 24(b), this increasing current component is the B phase current Ib component of the AB phase current Iab. On the other hand, as can be seen from FIG. 24(f), this is the time when the C phase current Ic is decreasing, and this is the timing to regenerate the magnetic energy stored in the C phase magnetic circuit to the power supply side. This C phase voltage component is superimposed on the AB phase voltage Vabk in FIG. 28(d). It is the third term on the right side of Equation (51). The voltages of the other phases in FIG. 28 are similar.

[0154] There is a big problem here. When the motor with full-pitch windings in FIG. 26 is driven by the drive circuit in FIG. 25 as in FIG. 28, it becomes difficult to increase the current once a certain current value and rotation speed are exceeded. This is a problem related to the magnitude of the power supply voltage, and is a problem when the voltage shown in FIG. 28 cannot be generated or supplied. For example, the CA-phase current Ica in FIG. 28(c) flows the current of Equation (46), and the C-phase current Ic is reduced in the section where the rotor rotation angle θr is from 48° to 54°. At this time, if the regenerative voltage reaches the power supply voltage Vsour, Vck of the CA-phase voltage Vcak in Equation (53) becomes (-Vsour), and at the same time, (-Vck) of the AB-phase voltage Vabk in Equation (51) becomes Vsour. As a result, the AB-phase winding generates Vsour, which is the same as the power supply voltage, as an induced voltage, and it becomes impossible to increase the B-phase current Ib component of the AB-phase current Iab in Equation (44) from the point when θr is 48°. At this time, the BC phase current Ibc in equation (45) flows, which causes the current balance in equations (44), (45), and (46) to be lost. As a result, it becomes difficult to drive the full-pitch winding in FIG. 26 with the drive circuit in FIG. 25 as shown in FIG. 28.

[0155] There are several factors that cause the large voltage and overvoltage in Figure 28. One factor is that the voltage of the full-pitch winding has the relationship shown in equations (51), (52), and (53), which also includes the magnetic flux changes of other phases. The second factor is that motors such as those in Figures 23 and 26 use the attractive force of reluctance torque, and therefore the supply and regeneration of magnetic energy is repeated on the power supply side and the motor side. The third factor is that in the motor in Figure 26, the timing at which the torque generation of the C-phase current ends coincides with the timing at which the other phases start to generate torque, and so the timing overlaps. Note that the motors in Figures 23 and 26 are three-phase motors with the characteristics shown in Figure 28, but when they become multi-phase, such as five-phase motors or seven-phase motors, the magnetic flux changes, voltage, and magnetic energy supply and regeneration become even more complicated.

[0156] Next, consider the interlinkage flux and voltage of the full-pitch winding in Fig. 26. From equations (51), (52), and (53), the following equation can be obtained. Vcak+Vabk=Vak (54) Vabk+Vbck=Vbk (55) Vbck+Vcak=Vck (56) The AB-phase winding 261 and the BC-phase winding 265 in FIG. 26 are both linked to the A-phase magnetic flux φa, and conversely, are in the opposite directions to the B-phase magnetic flux φb and the C-phase magnetic flux φc. When the AB-phase winding 261 and the CA-phase winding 265 are connected in series, only the A-phase magnetic flux φa component remains. Voltages accompanying the flux changes of the B-phase magnetic flux φb and the C-phase magnetic flux φc are induced in the AB-phase winding 261 and the CA-phase winding 265, but these voltages are offset. This is the logic behind the sum of equations (51) and (53) being equation (54). It is also the sum of equations (48) and (50). The same is true for the other phases. It is a very important relationship that the voltage of the complex and large full-pitch winding shown in FIG. 28 can be turned into the single-phase voltages Vak, Vbk, and Vck shown in FIG. 24 by connecting two windings in series. This shows that the voltage is simplified and that it is not affected by the magnetic flux of other phases.

[0157] Moreover, the current component passing through the AB-phase winding 261 and the CA-phase winding 265 in series is the A-phase current component Ia, according to equations (44) and (46). When this A-phase current component Ia is passed through the AB-phase winding 261 and the CA-phase winding 265 in series, it does not affect the B-phase magnetic flux φb and the C-phase magnetic flux φc. That is, it can be seen from FIG. 26 that the magnetomotive force of the A-phase current component Ia does not act on the B-phase south pole 13 and the B / phase north pole 16, and on the C-phase south pole 15 and the C / phase north pole 12, according to Ampere's law of circular integral. As a result, the series connection of the two full-pitch windings is not affected by the magnetic flux of the other phase, and the series current component does not generate a magnetomotive force on the other phase. Although the case of three phases is explained in Fig. 26, the same relationship applies to multi-phases such as five, seven, nine, and eleven phases, which will be described later, and this is an effective method. In particular, in the case of multi-phases, the voltage of the full-pitch winding becomes complicated, so this is an effective method for accurately controlling the magnetic flux components of each phase.

[0158] Next, an example of a drive circuit that is not affected by changes in magnetic flux of other phases will be described with reference to FIG. The drive circuit in FIG. 29 can be controlled while maintaining the relationships of equations (44), (45), and (46) and equations (54), (55), and (56), and is a drive circuit with high drive efficiency and utilization rate. The drive circuit in FIG. 29 is configured by arranging two full-pitch windings of the same phase due to the symmetry of the circuit configuration. FIG. 27 shows an example of the motor in FIG. 26 configured with two stator pole pairs so that two full-pitch windings of the same phase can be obtained. 271 and 274 are AB-phase windings, 272 and 275 are BC-phase windings, and 273 and 276 are CA-phase windings. The thick dashed lines show the coil ends of each winding, indicating the connection relationship and the winding destination. 277 and 27D are A-phase stator south poles, 278 and 27E are A / phase stator north poles, 279 and 27F are B-phase stator south poles, 27A and 27G are B / phase stator north poles, 27B and 27H are C-phase stator south poles, and 27C and 27J are C / phase stator north poles. A permanent magnet facing the respective polarity is placed between each of the stator poles. 27K is the north pole of the rotor, and indicates the position of the rotor rotation angle start point θr=0, as in Figure 26. For ease of understanding, the names of the phases and the names of the energizing currents are shown in parentheses.

[0159] In FIG. 29, 29R is a DC voltage source. 291 is a transistor that drives an AB-phase current Iab1 to an AB-phase winding 297. 294 is a transistor that drives an AB-phase current Iab2 to an AB-phase winding 29A. The AB-phase windings 297 and 29A correspond to the AB-phase windings 271 and 274 in FIG. 27. 292 is a transistor that drives a BC-phase current Ibc1 to a BC-phase winding 298. 295 is a transistor that drives a BC-phase current Ibc2 to a BC-phase winding 29B. The BC-phase windings 298 and 29B correspond to the BC-phase windings 272 and 275 in FIG. 27. 293 is a transistor that drives a CA-phase current Ica1 to a CA-phase winding 299. 296 is a transistor that drives a CA-phase current Ica2 to a CA-phase winding 29C. CA-phase windings 299 and 29C correspond to CA-phase windings 273 and 276 in Fig. 27. Diodes 29D, 29E, 29F, 29G, 29H, and 29J regenerate the energy of each winding to DC voltage source 29R. Diodes 29K, 29L, 29M, 29N, 29P, and 29Q are used to reduce interference between voltages and currents in the left and right directions on the paper surface of Fig. 29. Also, to make the circuit operation of Fig. 29 easier to understand, arrows indicating the direction of the flowing current and the names of the currents are provided.

[0160] The circuit operation of FIG. 29 passes currents of each phase as shown in (a), (b), and (c) of FIG. 28, for example, and the breakdown is as shown in the relationships of formulas (44), (45), and (46). Therefore, the components Ia, Ib, and Ic shown in (a), (b), and (c) of FIG. 24 are passed. The AB-phase current Iab1 passing through the AB-phase winding 297 is the sum of the A-phase current Ia and the B-phase current Ib as shown in formula (44). Of these, the B-phase current Ib passes through the diode 29K and is passed to the BC-phase winding 298. Similarly, the AB-phase current Iab2 passing through the AB-phase winding 29A is the sum of the A-phase current Ia and the B-phase current Ib as shown in formula (44). Of these, the B-phase current Ib passes through the diode 29N and is passed from the BC-phase winding 29B.

[0161] Moreover, the BC phase current Ibc1 passing through the BC phase winding 298 is the sum of the B phase current Ib and the C phase current Ic as shown in formula (45). Of these, the C phase current Ic passes through the diode 29L and is conducted from the CA phase winding 299. Similarly, the BC phase current Ibc2 passing through the BC phase winding 29B is the sum of the B phase current Ib and the C phase current Ic as shown in formula (45). Of these, the C phase current Ic passes through the diode 29P and is conducted to the CA phase winding 29C. Moreover, the CA phase current Ica1 passing through the CA phase winding 299 is the sum of the C phase current Ic and the A phase current Ia as shown in formula (46). Of these, the A phase current Ia passes through the diode 29M and is conducted to the AB phase winding 29A. Similarly, the CA phase current Ica2 passing through the CA phase winding 29C is the sum of the C phase current Ic and the A phase current Ia as shown in formula (46). Of these, the A-phase current Ia flows from the AB-phase winding 297 through the diode 29Q.

[0162] At this time, the voltages of each full-pitch winding in FIG. 29 are the voltages shown in (d), (e), and (f) of FIG. 28. These voltages are large and complicated, following equations (51), (52), and (53) and including the voltages shown in (d), (e), and (f) of FIG. 24. However, on the paper surface of FIG. 29, the voltage from the upper end of the AB-phase winding 297 to the lower end of the BC-phase winding 298 is the B-phase voltage Vbk shown in (e) of FIG. 24, which is a relatively simple voltage. The voltages across the other two windings connected in series are similarly the A-phase voltage Vak and the C-phase voltage Vck.

[0163] For example, when the B-phase current Ib increases between the rotor rotation angle θr of 48° and 54°, the B-phase voltage Vbk is only Vbk in the formula (35) due to the increase in the B-phase magnetic flux φb. Therefore, when the B-phase current Ib between the transistors 291 and 292 increases, for example, by turning on the transistor 291 and turning on the transistor 292, the B-phase current Ib can be increased within the range allowed by the power supply voltage. Also, when the rotor rotation angle θr is between 72° and 78°, as shown in (a) and (b) of FIG. 24, the B-phase current Ib decreases from 1 to 0 and the A-phase current Ia increases from 0 to 1. In this case, for example, the transistor 292 is turned off to regenerate the B-phase current Ib to the DC voltage source 29R through the diode 29E, and at the same time, the transistor 296 is turned on. Assuming that PWM control is performed by repeatedly turning transistor 291 on and off, when 291 is on, A-phase current Ia passing through diode 29Q increases, and B-phase current Ib is regenerated to the power supply and decreases. When transistor 291 is off, B-phase current Ib is regenerated through diodes 29D and 29E and decreases, while A-phase current Ia continues to increase during this period.

[0164] In reality, transistors 291, 292, and 296 can each perform PWM control, so if the corresponding current is smaller than the command value, the on state is increased, and if it is larger than the command value, the off state is increased, and PWM control is performed to control precisely. The voltages across the two full-pitch windings arranged in series in Figure 29 are related by equations (54), (55), and (56), so each phase current component Ia, Ib, and Ic can be controlled. Note that each current in Figure 29 is direct current, so one transistor has the ability to PWM control one direct current that flows, and if the value of the flowing current can be detected, it is possible to increase or decrease it to an appropriate value. Compared to PWM control of AC current, PWM control of DC current can be performed easily with a simple circuit configuration.

[0165] As mentioned above, diodes 29K, 29L, 29M, 29N, 29P, and 29Q reduce the interference between the voltage and current on the left and right sides of the paper in Fig. 29, but each current can be PWM controlled by the corresponding transistor, so this is not necessarily required. Also, as shown in Fig. 29, in DC current control of each full-pitch winding, the corresponding current value can be relatively easily controlled by PWM control of each transistor, and the direction of current flow can be easily selected or branched depending on the design of the drive circuit. These actions of a DC drive circuit are much simpler than AC control, and are a major feature.

[0166] Next, the utilization rate of each full-pitch winding in the case where the motor of FIG. 27, which has 6S10R full-pitch windings and two stator pole pairs, is driven by the drive circuit of FIG. 29 will be described. As described above, an example has been shown in which the currents of each phase shown in (a), (b), and (c) of FIG. 24 and (a), (b), and (c) of FIG. 28 are passed according to equations (54), (55), and (56). For example, the component of the A-phase current Ia passes through the transistor 291, the AB-phase full-pitch winding 297, the CA-phase full-pitch winding 29C, and the transistor 296 of FIG. 29. On the other hand, the component of the A-phase current Ia passes through the transistor 293, the CA-phase full-pitch winding 299, the AB-phase full-pitch winding 29A, and the transistor 294 of FIG. 29. At this time, four of the six full-pitch windings of the motor of FIG. 27 are passed through to generate torque. The utilization rate of these full-pitch windings is 2 / 3. The same applies when passing other components of the B-phase current Ib and C-phase current Ic. In the concentrated winding motor of Figure 23, with the drive circuit of Figure 25, and when current is passed as in Figure 24(a), (b), and (c), the winding utilization rate is 1 / 3. In comparison, the full-pitch winding motors of Figures 27, 29, and 28 have improved the winding utilization rate by two times. Note that the doubling of the winding utilization rate and the reduction in the resistance value in the slots of the full-pitch winding are two sides of the same coinciding factor. As a result, the copper loss of the motor can be reduced by half.

[0167] Next, the utilization rate of each transistor in the drive circuit of FIG. 29 will be described. As shown in the figure, each transistor such as 291, 292 is connected to each full-pitch winding, and each current is PWM controlled. Therefore, the utilization rate of each transistor is 2 / 3, which is the same as each winding. Compared to the drive of concentrated windings in FIG. 23, FIG. 25, and FIG. 24, the utilization rate of each transistor is doubled. When the utilization rate of each transistor is doubled, the current capacity of the transistor can be reduced to 1 / 2, so that the drive circuit can be made smaller and less expensive. Note that the conventional surface permanent magnet synchronous motor SPMSM and the built-in magnet synchronous motor IPMSM are driven by three-phase sinusoidal AC. The utilization rate of these transistors is 1 / 3 according to the above-mentioned calculation method of utilization rate. Therefore, in the motors with full-pitch windings in FIG. 27, FIG. 29, and FIG. 28, the utilization rate of each transistor can be improved to 2 times that of the drive circuits of the conventional SPMSM and IPMSM, so that the drive circuit can be made smaller and less expensive than the conventional ones. 26 and 27 show examples of a three-phase 6S10R and a 12S20R with two stator pole pairs, but the winding utilization rate and transistor utilization rate can be further improved by increasing the number of phases. Examples such as a five-phase 10S18R and a seven-phase 14S26R will be explained later.

[0168] As shown above, the effects of the drives in Figures 27, 29, and 28 are that it is difficult to pass current due to excessive voltages in other phases generated in the full-pitch winding, that the motor with concentrated windings in Figure 23 has large copper loss, and that the utilization rate of the drive circuit in Figure 25 is low and the current capacity of the drive circuit increases, resulting in a large size and high cost. Specifically, it is possible to pass current through the full-pitch winding, halving the motor copper loss and halving the current capacity of the drive circuit, resulting in a smaller size and lower costs.

[0169] Conversely, in the motors with full-pitch windings in Figures 26 and 27, as shown in the figures, the coil end portion of the winding is long, which causes problems with increased wire material, copper loss, and cost. Other problems include poor manufacturability of the winding, which can lead to a decrease in the winding space factor, and the rotor axial length of the coil end portion tends to increase, which can lead to large motors, but these can be improved with production technology. One method of shortening the coil end length of the winding is to increase the number of pole pairs, and in Figure 27, the coil end length has been shortened by half compared to Figure 26 by using two stator pole pairs. It is also possible to shorten the length by using three or four pole pairs.

[0170] In addition, a composite motor is configured by incorporating two motors on the inner diameter side and the outer diameter side, and the full-pitch winding of each phase is wound in a toroidal shape as an annular winding, thereby minimizing the length of the coil end portion. The motor in FIG. 30 is a composite motor in which another motor is arranged on the outer diameter side in addition to 1 / 4 of the first quadrant of FIG. 27. 301 is the first rotor on the inner diameter side, and 302 is the first stator on the inner diameter side, which has the same configuration as FIG. 27. 30C is the rotor shaft of the first rotor 301. 303 is the second stator on the outer diameter side, and 304 is the second rotor on the outer diameter side, which is a so-called outer rotor motor configuration. The first rotor 301 and the second rotor 304 are mechanically connected. 305 is the A-phase stator south pole, 306 is the B / phase stator north pole, 307 is the C-phase stator south pole, and 308 is the A / phase stator north pole. 309 is an AB-phase toroidal winding. It has the shortest coil end length and can be wound in an aligned manner while applying tension, which increases the winding space factor and improves productivity. 30A is a BC-phase toroidal winding, which is a similar toroidal winding. 30B is a CA-phase toroidal winding, which is a similar toroidal winding.

[0171] The motor in FIG. 30 is a diagram for showing the annular windings 309, 30A, and 30B, and the shape of each part, including the number of pole pairs, needs to be optimized. In addition, the combination of two motors can be a so-called axial gap type motor configuration in which the motor in FIG. 30 is arranged in the rotor axial direction. In that case, the annular winding configuration can be used, and the length of the coil end part can be shortened. In addition, if the motor is combined in the rotor axial direction, the circumferential length does not change like the motor in FIG. 30, so it is easy to make a motor with a well-balanced shape. In addition, in the motor form in FIG. 27, if the stack thickness of the stator core in the rotor axial direction is smaller than the coil end length and the motor core is shaped like a flat motor, the annular winding structure shortens the total wire length. Conversely, if the motor is elongated, the burden of the coil end length is relatively small. Furthermore, if two annular windings located at an electrical angle of 180°, which is half the electrical angle of 360° of one stator pole pair, are connected in series, the flux linkage will be the same as that of a full-pitch winding, and electromagnetically equivalent windings can be constructed.

[0172] In addition, the use of each stator pole in the motor with full-pitch windings in Figures 26 and 27 is concentrated on two of the six stator poles. Since the current flowing through 2 / 3 of the windings is configured to concentrate the magnetomotive force on 1 / 3 of the stator poles, a large magnetomotive force is applied to a specific part. In order to obtain a magnetic flux of 2.0 [T] or more, the relative permeability of that specific part drops to nearly 1, making this a motor configuration that is convenient for concentrating the magnetomotive force. In addition, as explained in Figures 16, 17, 18, and 19, the magnetic flux is passed through six teeth of the six stator poles, and the magnetic flux is concentrated on two stator poles in the air gap to generate torque. Therefore, it can be said that most of the motor is used to generate torque.

[0173] In addition, an example of using the drive circuit of FIG. 29 in a motor with various stator pole pairs will be described. In the case of a motor with full-pitch windings of one pole pair as shown in FIG. 26, in order to obtain two windings for each phase, two windings for each phase are arranged in each slot to form six windings, which can be driven by the drive circuit of FIG. 29. In a motor with full-pitch windings of three pole pairs, the winding of one pole pair is divided into two, and the windings of each phase are connected to 1.5 windings each, forming a total of six windings for each phase, and the drive circuit of FIG. 29 can be used. In a motor with full-pitch windings of four pole pairs, there are four windings for each phase, so two of them are connected in series, forming a total of six windings, and the drive circuit of FIG. 29 can be used. Similarly, even if the number of pole pairs is changed, the drive circuit of FIG. 29 can be used. Conversely, the drive circuit of FIG. 29 can be modified according to the number of pole pairs. Examples of drive circuits for motors with full-pitch windings as shown in Figs. 26 and 27, which are different from those shown in Fig. 29, and drive circuits for multi-phase motors such as 5-phase and 7-phase motors will be described later. Multi-phase motors such as 5-phase and 7-phase motors in which the stator poles are evenly arranged in the circumferential direction and motors in which the stator poles are not evenly arranged in the circumferential direction will be described later. In addition, in relation to a method of regenerating magnetic energy, a method of constantly passing a field current component, particularly at high speed rotation, to reduce the winding voltage will be described later. In addition, as the number of pole pairs increases, the cross section of the permanent magnet of the stator becomes shorter in the circumferential direction and approaches a parallelogram, which is closer to a practical shape. This makes it easier to design, manufacture, and fix and install the permanent magnets.

[0174] Next, an embodiment of claim 5 will be described. The present invention does not limit the number Nps of stator poles Ps and the number Npr of rotor poles Pr, but good characteristics can be obtained with a specific relationship. Claim 5 is a motor in which multiple rotor poles are arranged at equal intervals in the circumferential direction, and the stator poles are also arranged at equal intervals in the circumferential direction. If Ns and Nr are integers equal to or greater than 1, the number Nps of stator poles Ps and the number Npr of rotor poles Pr have the following relationship: Nps = 2 + 4 × Ns (57) Npr = 2 + 4 × Nr (58) One example is the 6S10R motor and three-phase motor shown in Figures 1, 14, 26, and 27. Other excellent configurations such as 14S26R and 10S18R will be explained later. The number of stator pole pairs can also be increased to 2, 3, or 4. All of these configurations are magnetically point-symmetric with respect to the rotor center point. The stator windings can be used as full-pitch windings, which are convenient in terms of the utilization rate of each winding, the utilization rate of the transistors in the drive circuit, and maximum torque. Concentrated windings can also be configured.

[0175] In addition, when there are multiple stator pole pairs, some of the stator poles may be shifted in the circumferential direction from the equally spaced arrangement described above in order to cancel out some of the harmonic components of the torque, i.e., torque ripple. In addition, in the case of a motor configuration with concentrated windings, spaces may be provided in the circumferential direction of the stator poles. If the total width of the circumferential spaces is set to twice the pitch θppr of the rotor poles Pr or an integer multiple thereof, it is possible to secure space within the stator without significantly changing the basic characteristics of the motor. For example, it is possible to detect the rotor position, observe the rotor state, and operate it.

[0176] As an embodiment of claim 5, there is a 6S10R full-pitch winding motor and a three-phase motor shown in Figures 26 and 27. It satisfies the conditions of equations (57) and (58). It has been described in detail as an example of a relatively simple motor of the present invention. As already explained, it has been shown that the problem of the winding induced voltage becoming excessive and being unable to energize can be solved by driving the motor as shown in Figures 12, 13, and 28 with the drive circuit of Figure 29, the utilization rate of the windings and the utilization rate of each transistor can be reduced to about 2 / 3, and the maximum torque can be increased.

[0177] As another embodiment, a linear development diagram showing the operation of a 6S14R motor is shown in FIG. 31. This is an example in which the number of rotor poles of the 6S10R in FIG. 1, FIG. 14, FIG. 26, and FIG. 27 is increased from 10 to 14. The stator is the same as FIG. 26. Since there are 14 rotor poles, the rotor pole pitch is 25.7°, and the operating period is twice as long, 51.4°, as shown in (a) to (h) of FIG. 31. By showing the shape of the stator poles and the rotor poles facing the air gap surface in a similar manner to the development diagram in FIG. 12, the mutual passing magnetic flux and electromagnetic action can be analyzed. Specifically, this is a linear development diagram for the purpose of drawing the generation section of CCW torque. The horizontal axis of FIG. 31 is the rotor rotation angle θr, and the right direction is the CCW direction. FIG. 31 shows the rotor rotation angle from -30° to 360°. At the top of each row, the section where CCW torque can be generated is shown by a thick line above the rotor pole shape. At this time, the position and width of the thick line correspond to the position and width of the corresponding stator pole.

[0178] FIG. 31(a) shows the shape of each stator pole facing the air gap surface. FIG. 31(b) shows the rotor rotation position θr=0°, which is the starting point of rotor rotation. On the paper surface of FIG. 31, the left side position of the A-phase stator south pole 11 coincides with the right side position of the rotor north pole 311. At this position, the A-phase stator south pole 11, the A / phase stator north pole 14, the C / phase stator north pole 12, and the C-phase stator south pole 15 can generate an attractive force in the CCW direction, as shown by the thick line at the top of FIG. 31(b). FIG. 31(c) shows the rotor rotation position θr=8.6°, where the C / phase stator north pole 12 and the C-phase stator south pole 15 can no longer generate an attractive force in the CCW direction. FIG. 31(d) shows θr=17.1°, where the B-phase stator south pole 13 and the B / phase stator north pole 16 start to generate an attractive force in the CCW direction. FIG. 31(e) shows θr=25.7°, where the A-phase stator south pole 11 and the A / phase stator north pole 14 can no longer generate an attractive force in the CCW direction. FIG. 31(f) shows θr=34.3°, where the C / phase stator north pole 12 and the C-phase stator south pole 15 start to generate an attractive force in the CCW direction. FIG. 31(g) shows θr=42.9°, where the B-phase stator south pole 13 and the B / phase stator north pole 16 can no longer generate an attractive force in the CCW direction. FIG. 31(h) shows θr=51.4°, where this state returns to the same state as FIG. 31(b). The motor in Fig. 31 repeats the same operation seven times in a 51.4° cycle to make one rotation of the rotor. In this way, the 6S14R motor in Fig. 31 can generate torque from two or more stator poles at the same time, just like the 6S10R motor in Fig. 26. The drive circuit for the 6S14R motor in Fig. 31 can be made into a motor with two stator pole pairs, and it can be driven by the drive circuit in Fig. 29, just like the motor in Fig. 27.

[0179] As another embodiment, a cross-sectional view of a motor configuration with 14S26R full-pitch winding is shown in Fig. 32. In this example, Ns in equation (57) is 3 and Nr in equation (58) is 6. The basic electromagnetic operation and the like have been explained so far using three-phase motors such as Figs. 1, 14, 26, and 27, but the motor of the present invention can be expanded and developed to a multi-phase motor by applying the same technology. Claim 5 can be expanded to a multi-phase motor such as a three-phase motor, a five-phase motor, a seven ...

[0180] In Fig. 32, 328 is an A-phase stator south pole, and 32A is an A / -phase stator north pole, and the A-phase magnetic flux φa shown passes from the A / -phase stator north pole 32A through the rotor to the A-phase stator south pole 328. As with the relationship between the A-phase, A / -phase, and A-phase magnetic flux φa, the stator poles of each phase are arranged in the CCW direction, and the phase of each stator pole is indicated in parentheses on the outer periphery of the stator, and the magnetic flux components of that phase are indicated. B-phase, B / -phase, and B-phase magnetic flux φb, C-phase, C / -phase, and B-phase magnetic flux φb, D-phase, D / phase, and D-phase magnetic flux φd, E-phase, E / phase, and E-phase magnetic flux φe, F-phase, F / phase, and F-phase magnetic flux φf, and G-phase, G / phase, and G-phase magnetic flux φg are indicated.

[0181] In Fig. 32, 321 is an AD-phase winding with a full-pitch winding wound in slots spaced 180° apart, and passes an AD-phase current Iad. Both slots are connected at the coil end, and the connection is indicated by a dashed line in Fig. 32. Similarly, 322 is a BE-phase winding that passes a BE-phase current Ibe, 323 is a CF-phase winding that passes a CF-phase current Icf, 324 is a DG-phase winding that passes a DG-phase current Idg, 325 is an EA-phase winding that passes an EA-phase current Iea, 326 is an FB-phase winding that passes an FB-phase current Ifb, and 327 is a GC-phase winding that passes a GC-phase current Igc.

[0182] The currents Iad, Iea, Ibe, Ifb, Icf, Igc, and Idg of the seven-phase full-pitch windings are expressed by the following equations, where Ia is the A-phase current, Ib is the B-phase current, Ic is the C-phase current, Id is the D-phase current, Ie is the E-phase current, If is the F-phase current, and Ig is the G-phase current. Iad=Ia+Id (59) Ibe = Ib + Ie (60) Icf=Ic+If (61) Idg = Id + Ig (62) Iea = Ie + Ia (63) Ifb = If + Ib (64) Igc=Ig+Ic (65) Both currents are positive and DC. The currents in the full-pitch windings can be calculated from the phase currents. Conversely, the phase currents can be calculated from the currents in the full-pitch windings. The values ​​on both sides can be converted to each other.

[0183] For example, if a component of the A-phase current Ia is supplied as the AD-phase current Iab in equation (59) to the AD-phase winding 321, which is a full-pitch winding in Fig. 32, and at the same time a component of the A-phase current Ia is supplied as the EA-phase current Iea in equation (63) to the EA-phase winding 325, then the A-phase magnetic flux φa in Fig. 32 is excited. This is the same as when the full-pitch winding motor in Fig. 32 is changed to a concentrated winding motor by winding concentrated windings around A-phase stator pole 328 and A / -phase stator pole 32A, and the A-phase current Ia is supplied to these concentrated windings. In this case as well, a magnetic flux the same as the A-phase magnetic flux φa of the full-pitch winding in Fig. 32 is excited. The same applies to the other phases.

[0184] Also, although it is a little repetitive, for example, when AD-phase current Iab is applied to AD-phase winding 321 in FIG. 32, the magnetomotive force of Iab is applied to all stator poles and all rotors in FIG. 32 according to Ampere's law of circular integral. Each phase current affects the entire motor. As described above, if a component of A-phase current Ia is applied as AD-phase current Iab and at the same time a component of A-phase current Ia is applied as EA-phase current Iea, the magnetomotive forces on the magnetic flux components of other phases are cancelled out, so that the magnetic flux components of other phases other than A-phase magnetic flux φa are not affected. Therefore, in order to control the magnetic flux of each phase shown in FIG. 32, it is necessary to apply the currents of equations (59) to (65) to each full-pitch winding.

[0185] 32 are linked to all full-pitch windings, and are induced in the voltages of the full-pitch windings according to Faraday's law of electromagnetic induction, resulting in the following relationship: The number of turns of the full-pitch winding is Nw / 2, and is expressed in the same way as the three-phase equations (48), (49), and (50). Vadk=Nw / 2×d(φa+φb+φc+φd-φe-φf-φg) / dt =Vak+Vbk+Vck+Vdk-Vek-Vfk-Vgk (66) Vbek=Nw / 2×d(-φa+φb+φc+φd+φe-φf-φg) / dt =-Vak+Vbk+Vck+Vdk+Vek-Vfk-Vgk (67) Vcfk=Nw / 2×d(-φa-φb+φc+φd+φe+φf-φg) / dt =-Vak-Vbk+Vck+Vdk+Vek+Vfk-Vgk (68) Vdgk=Nw / 2×d(-φa-φb-φc+φd+φe+φf+φg) / dt =-Vak-Vbk-Vck+Vdk+Vek+Vfk+Vgk (69) Veak=Nw / 2×d(φa-φb-φc-φd+φe+φf+φg) / dt =Vak-Vbk-Vck-Vdk+Vek+Vfk+Vgk (70) Vfbk=Nw / 2×d(φa+φb-φc-φd-φe+φf+φg) / dt =Vak+Vbk-Vck-Vdk-Vek+Vfk+Vgk (71) Vgck=Nw / 2×d(φa+φb+φc-φd-φe-φf+φg) / dt =Vak+Vbk+Vck-Vdk-Vek-Vfk+Vgk (72)

[0186] In this way, each full-pitch winding is affected by the multi-phase magnetic flux, resulting in complex voltages. However, the voltages of the full-pitch windings have the following relationship, and the voltage relationship can be simplified. Vadk+Veak=Nw / 2×d(2×φa)=Vak (73) Veak+Vbek=Nw / 2×d(2×φe)=Vek (74) Vbek+Vfbk=Nw / 2×d(2×φb)=Vbk (75) Vfbk+Vcfk=Nw / 2×d(2×φf)=Vfk (76) Vcfk+Vgck=Nw / 2×d(2×φc)=Vck (77) Vgck+Vdgk=Nw / 2×d(2×φg)=Vgk (78) Vdgk+Vadk=Nw / 2×d(2×φd)=Vdk (79)

[0187] As described above, according to Ampere's law of circular integral, if the component of the A-phase current Ia is supplied as the AD-phase current Iab in equation (59) to the AD-phase winding 321 in FIG. 32, and at the same time, the component of the A-phase current Ia is supplied as the EA-phase current Iea in equation (63) to the EA-phase winding 325, the A-phase magnetic flux φa in FIG. 32 is excited, and at the same time, the component of the A-phase current Ia in both windings does not affect the magnetic flux components of other phases. Equation (73) is the flip side of this, and according to Faraday's law of electromagnetic induction, the sum of the AD-phase voltage Vadk and the EA-phase voltage Veak is related only to the A-phase magnetic flux φa and the A-phase voltage Vak, and is not affected by the magnetic flux of other phases. Equations (74) to (79) have a similar relationship. Note that these relationships are also related to the motor configuration that is point-symmetric with respect to the rotor center. In addition, there is a control method that applies the simplification method for these voltages and is less susceptible to the effects of many other phase voltages, which will be explained later in Figure 35 etc.

[0188] Next, Fig. 33 shows a linear development diagram showing the operation of the 14S26R motor in Fig. 32 to generate CCW torque. This development diagram is similar to Fig. 12 and Fig. 31, and shows the shape of the stator pole facing the air gap and the shape of the rotor pole, allowing the mutual passing magnetic flux and electromagnetic action to be analyzed. The rotor rotation angle in Fig. 32 is θr=0°, and the CCW direction of the motor is the right direction in Fig. 33. In each row of Fig. 33, the section where CCW torque can be generated is shown by a thick line above the rotor pole shape. At this time, the position and width of the thick line are the position and width of the corresponding stator pole. In Fig. 33, the stator pole width is θsg=360° / 28=12.857°. The rotor pole width θrg can be less than 360° / 26=13.846°, but θrg=θsg=12.857° is used. The stator pole width θsg and rotor pole width θrg can be increased or decreased to suit the required motor specifications, and the pole shape can also be changed.

[0189] FIG. 33(a) shows the shape of each stator pole facing the air gap surface. 331 corresponds to the A-phase stator south pole 328 in FIG. 32. The stator poles in the CCW direction in FIG. 32 are arranged in order to the right in FIG. 33(a). The horizontal axis θr in FIG. 33 is a little confusing, but θr on the horizontal axis shows the electrical angle position of one stator pole pair, which is 360°, and is also the electrical angle position in the rotation direction of each part of the stator. The rotational position of the rotor is shown on the left side of each row in FIG. 33. On the paper surface of FIG. 33, each part of the rotor is moved to the right as shown in (b), (c), (d), and (e). FIG. 33(b) shows the rotor rotational position θr=0°, which is the starting point of rotor rotation. 332 in FIG. 33(b) is the rotor north pole, which corresponds to the rotor north pole 329 in FIG. 32. On the paper surface of FIG. 33, the left side position of the A-phase stator south pole magnetic pole 328 coincides with the right side position of the rotor north pole magnetic pole 329. In this position, a total of six stator poles, including the A-phase stator south pole magnetic pole 328, the A / phase stator north pole magnetic pole 32A, the B-phase stator south pole magnetic pole, the B / phase stator north pole magnetic pole, the C-phase stator south pole magnetic pole, and the C / phase stator north pole magnetic pole, can generate an attractive force in the CCW direction, and are shown by thick lines in six places at the top of FIG. 33(b). A-phase magnetic flux φa, B-phase magnetic flux φb, and C-phase magnetic flux φc shown in FIG. 32 are used to generate torque in the CCW direction. FIG. 33(c) shows the rotor rotation position θr=4.0°, at which the C-phase stator south pole magnetic pole and the C / phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The G-phase stator south pole magnetic pole and the G / -phase stator north pole magnetic pole start to generate an attractive force in the CCW direction. In FIG. 33(d), θr=7.9°, and the B-phase stator south pole magnetic pole and the B / -phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The F-phase stator south pole magnetic pole and the F / -phase stator north pole magnetic pole start to generate an attractive force in the CCW direction. In FIG. 33(e), θr=11.9°, and the A-phase stator south pole magnetic pole 328 and the A / phase stator north pole magnetic pole 32A can no longer generate an attractive force in the CCW direction. The E-phase stator south pole magnetic pole and the E / -phase stator north pole magnetic pole start to generate an attractive force in the CCW direction. In FIG. 33(f), θr=15.8°, and the G-phase stator south pole magnetic pole and the G / -phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The D-phase stator south pole and the D / phase stator north pole begin to generate an attractive force in the CCW direction.In Fig. 33(g), at θr = 19.8°, the attraction force cannot be generated in the CCW direction between the F-phase stator S-pole and the F / -phase stator N-pole. The attraction force begins to be generated in the CCW direction between the C-phase stator S-pole and the C / -phase stator N-pole. In Fig. 33(h), at θr = 23.7°, the attraction force cannot be generated in the CCW direction between the E-phase stator S-pole and the E / -phase stator N-pole. The attraction force begins to be generated in the CCW direction between the B-phase stator S-pole and the B / -phase stator N-pole. In Fig. 33(i), at θr = 27.7°, this state returns to the same state as that in Fig. 33(b). Then, the motors in Figs. 32 and 33 repeat the same operation 13 times at a period of 27.7°, and the rotor makes one rotation.

[0190] As shown in Fig. 33, the 14S26R motor can generate torque with six stator poles while changing the stator poles that operate with the rotation of the rotor. In Fig. 33(c), the C-phase stator S-pole and the C / -phase stator N-pole can, in theory, still generate a CCW torque for the remaining approximately 1°, but it is very small and confusing, so the thick-line mark described above the rotor pole shape is omitted. The same applies to each row after Fig. 33(d). Also, by modifying the circumferential width θsg of the stator poles and the circumferential width θrg of the rotor poles, the torque characteristics can be improved. In addition, the air-gap surface shape of the stator poles and the rotor poles can be deformed into a convex shape, an arc shape, etc. Also, one or both of the stator poles and the rotor poles can be skewed or stepped-skewed. Skewing has the effect of widening the torque generation range. Also, skewing has the effect of smoothing the changes in the attraction forces in the circumferential and radial directions generated during rotation, and also has the effect of reducing vibration and noise.

[0191] FIG. 34 shows a cross-sectional view of the motor configuration of the 14S26R motor in FIG. 32 with two stator pole pairs. This motor is a seven-phase motor, and the phases of each stator pole are shown in parentheses on the outside of the stator 34F. The seven-phase stator pole configuration is A-phase, A / phase, B-phase, B / phase, C-phase, C / phase, D-phase, D / phase, E-phase, E / phase, F-phase, F / phase, G-phase, and G / phase. Since FIG. 34 shows the 14S26R with two stator pole pairs, there are 28 stator poles and 14 full-pitch windings. The stator pole pitch θpps is 12.9°, and the circumferential width θsg of the stator pole Ps is 6.4° in this example. As shown in the figure, the polarity of each stator pole Ps is the N-pole stator pole Psn and the S-pole stator pole Pss, and they are arranged alternately in the circumferential direction. Stator permanent magnets PMsbi are arranged between each stator pole in a direction that matches the polarity of the stator pole. A direct current is applied to each full-pitch winding in the direction indicated by the current symbol. 34G is the rotor shaft. In Figure 34, there are two stator pole pairs, so 52 rotor poles Pr are arranged, and the north rotor poles Prn and south rotor poles Prs are arranged alternately in the circumferential direction. A rotor permanent magnet PMrbi is arranged between each rotor pole in a direction that matches the polarity of the rotor pole Pr. Each stator winding is a full-pitch winding, and a direct current is applied in the direction indicated by the current symbol. The winding pitch is 180° electrical angle, which is 1 / 2 of the electrical angle of 360° for one stator pole pair, and 90° mechanical angle, and the coil end is indicated by a thick dashed line. One of the advantages of the motor in Fig. 34 is that the number of full-pitch windings is an even number, 14, and the drive circuit in Fig. 35 below has a symmetrical structure with little waste. Also, increasing the number of pole pairs in the stator makes it possible to reduce the thickness of the back yoke, which has the effect of making the motor more compact.

[0192] In FIG. 34, 341 and 342 are AD-phase windings, which carry AD-phase current Iad. The slots to which they are connected, separated by an electrical angle of 180°, are indicated by the coil ends of dashed lines. Similarly, 343 and 344 are BE-phase windings, which carry BE-phase current Ibe. 345 and 346 are CF-phase windings, which carry CF-phase current Icf. 347 and 348 are DG-phase windings, which carry DG-phase current Idg. 349 and 34A are EA-phase windings, which carry EA-phase current Iea. 34B and 34C are FB-phase windings, which carry FB-phase current Ifb. 34D and 34E are GC-phase windings, which carry GC-phase current Igc.

[0193] In addition, since the number of stator pole pairs in FIG. 34 is two, there are two sets of windings of the same phase. There are two slots for positive current and two slots for negative current of the same phase, and two ways can be selected for wiring the full-pitch winding from one slot to another. Two ways can also be selected for the other phases, so there are a total of 2 to the power of 7, or 128 ways of connection and winding. The electromagnetic action at this time is basically the same, except for the leakage flux in the space near the coil end, assuming that the current of each winding is accurately controlled. FIG. 34 shows one example of the connection method. Therefore, it is not specified how the coil ends in FIG. 34 should be connected for each winding in the drive circuit in FIG. 35 shown below. In addition, in electromagnetic field analysis using the finite element method (FEM), the leakage flux in the coil end is usually ignored, and is relatively small compared to the flux in the iron core. There are also other winding methods, such as a toroidal ring winding.

[0194] Next, an example of a drive circuit that supplies voltage and current to each full-pitch winding of 7 phases such as FIG. 34 will be described with reference to FIG. 35. For example, when generating a CCW torque in synchronization with a CCW rotation, the drive circuit of FIG. 35 energizes in the section shown by the bold line in FIG. 33, and excites the magnetic flux component of the corresponding phase in FIG. 32 to generate a CCW torque. Each full-pitch winding has a current shown in Equations (59) to (65), and the relationship between the magnetic flux and voltage of each phase shown in Equations (66) to (72). Each full-pitch winding on the drive circuit of FIG. 35 has a drive circuit configuration that can drive two windings in series so that the full-pitch winding of FIG. 34 can be energized while satisfying the conditions of Equations (59) to (65). This is just as well a configuration in which the windings are arranged in the order of Equations (73) to (79), and the voltage across the two windings connected in series can be simplified to the right side of Equations (73) to (79). Furthermore, the control paths for each phase current, such as the Ia component and Ib component, are clear, and each phase magnetic flux, such as φa and φb shown in FIG. 32, can be easily and individually controlled.

[0195] In Figure 35, 35F and 35N are AD-phase windings, and they carry the AD-phase currents Iad1 and Iad2 of equation (59). Note that Iad1 and Iad2 are theoretically the same value. Similarly, 35G and 35P are EA-phase windings, and they carry the EA-phase currents Iea1 and Iea2 of equation (63). 35H and 35Q are BE-phase windings, and they carry the BE-phase currents Ibe1 and Ibe2 of equation (60). 35J and 35R are FB-phase windings, and they carry the FB-phase currents Ifb1 and Ifb2 of equation (64). 35K and 35S are CF-phase windings, and they carry the CF-phase currents Icf1 and Icf2 of equation (61). 35L and 35T are GC-phase windings, and they carry the GC-phase currents Igc1 and Igc2 of equation (65). 35M and 35U are DG phase windings that conduct the DG phase currents Idg1 and Idg2 of equation (62).

[0196] In Fig. 35, 29R is a DC power supply. 351, 352, 353, 354, 355, 356, 357, 358, 359, 35A, 35B, 35C, 35D, and 35E are driving transistors that pass the current of each phase to the windings of each phase. 35V, 35W, 35X, 35Y, 35Z, 281, 282, 283, 284, 285, 286, 287, 288, and 289 are diodes that regenerate the magnetic energy of each phase winding to the DC power supply 29R by changing the current passing state of each transistor connected in series from the on state to the off state. The diodes 28A, 28B, 28C, 28D, 28E, 28F, 28G, 28H, 28J, 28K, 28L, 28M, and 28N have the effect of suppressing and blocking the influence and interference of the voltage and current of each phase. Since each transistor has the ability to control the current passing through it, these diodes are not necessarily required, and some or all of them can be omitted.

[0197] With the configuration and operation of Fig. 35, the currents Iad, Ibe, Icf, Idg, Iea, Ifb, and Igc of each phase shown in formulas (59) to (65) can be supplied with the simple voltage relationships shown in formulas (73) to (79). At this time, the voltages of each full-pitch winding are complex voltages as shown in formulas (66) to (72), but supplying power to two windings connected in series arranged above and below on the paper surface of Fig. 35 realizes supplying power with the simple voltage relationships shown in formulas (73) to (79). In particular, while a winding of one phase regenerates magnetic energy to the voltage Vsour of the DC power supply 29R, the voltage Vsour is generated as an induced voltage in the other winding, so it is necessary to drive the winding by canceling the induced voltage components of the other phases as shown in formulas (59) to (65). The voltages across the two full-pitch windings connected in series above and below on the paper surface of FIG. 35 are any of the values ​​on the right-hand sides of equations (73) to (79). The arrangement order of the windings on the paper surface of the drive circuit in FIG. 35 is the same as the arrangement order on the paper surface of the motor in FIG. 34, except for the connection of the coil ends. If the left half of the windings from the AD-phase winding 341 on the paper surface of FIG. 34 were connected to the right half of the windings with coil ends, the arrangement order of the windings would be that of FIG. 35, but this would be undesirable since the coil ends would be concentrated on the top and bottom of the paper surface of FIG. 34. In the case of a toroidal annular winding, the arrangement order of the windings on the motor can be made to match the arrangement order of the windings in the drive circuit in FIG. 35 without any problems.

[0198] In the drive circuit of Fig. 35, the phase currents Ia, Ib, Ic, Id, Ie, If, and Ig shown on the right-hand sides of equations (59) to (65) are currents that pass through the positions between the two windings where the diodes are located. The A-phase current Ia is a current that passes through diodes 28B and 28J, and the two components of the A-phase current Ia excite the A-phase magnetic flux φa. The E-phase current Ie is a current that passes through diodes 28C and 28K, and the two components of the E-phase current Ie excite the E-phase magnetic flux φe. The B-phase current Ib is a current that passes through diodes 28D and 28L, and the two components of the B-phase current Ib excite the B-phase magnetic flux φb. The F-phase current If is a current that passes through diodes 28E and 28M, and the two components of the F-phase current If excite the F-phase magnetic flux φf. The C-phase current Ic is a current passing through the diodes 28F and 28N, and the two components of the C-phase current Ic excite the C-phase magnetic flux φc. The G-phase current Ig is a current passing through the diodes 28G and 28P, and the two components of the G-phase current Ig excite the G-phase magnetic flux φg. The D-phase current Id is a current passing through the diodes 28H and 28A, and the two components of the D-phase current Id excite the D-phase magnetic flux φd. In this way, the magnetic fluxes φa, φe, φb, φf, φc, φg, and φd of each phase can be controlled individually. Note that there is also a method in which the number of stator pole pairs in FIG. 32 is one, and each winding is wound in parallel to make the number of windings 14. There is also a method in which the number of transistors in the drive circuit is reduced. In that case, the drive can be performed with seven windings. This will be explained later.

[0199] Next, Fig. 36 shows examples of the waveforms of the currents of each phase when the full-pitch windings of the seven phases in Fig. 34 are energized by the drive circuit in Fig. 35 using the operation in Fig. 33. This is an example of generating a CCW torque. (h) to (n) in Fig. 36 are the full-pitch winding currents Iad, Ibe, Icf, Idg, Iea, Ifb, and Igc energized to each full-pitch winding. These currents are related by equations (59) to (65), and the current components Ia, Ib, Ic, Id, Ie, If, and Ig on the right-hand side are shown in (a) to (g) in Fig. 36. The horizontal axis shows the rotor rotation angle θr, but when rotating at a constant speed, the shape of the current waveforms can also be expressed as time on the horizontal axis. As described above, the motors in Fig. 32 and Fig. 34 pass current through each phase in a cycle of 27.692° electrical angle, so Fig. 36 shows a current waveform over a range of 55.4° for two cycles. Note that this electrical angle also assumes that one stator pole pair has an electrical angle of 360°. Naturally, the current is controlled by changing the magnitude of the current according to the magnitude of the motor load, so the current amplitude is increased or decreased as shown in Fig. 36(f) to (j). In the case of negative torque, i.e., CW torque, the phase of current flow changes.

[0200] Each current in FIG. 36 shows good characteristics. Each full-pitch winding current flows in a 6 / 7 section, each contributing to torque generation, and the utilization rate of the winding is large at 6 / 7. In addition, the two current components on the right side of equations (59) to (65) for each full-pitch winding current are not flowing at the same time, so copper loss does not increase exponentially. These are indicators of reduced copper loss and high efficiency of the motor. The utilization rate of the drive transistor is also 6 / 7, and the two current components are not flowing at the same time, so the current capacity of the drive circuit can be reduced. These effects will be explained later. Note that each current waveform in FIG. 36 is shown in a rectangular shape, but as shown by the dashed line in FIG. 13, the current may be increased or decreased with a slope to provide an increase or decrease time.

[0201] Next, an example of a good operating state of a motor with full-pitch windings in Figs. 32 and 34 will be described. The contents are shown in the linear development diagram showing the operation in Fig. 33, the drive circuit in Fig. 35, and the waveforms of the currents of each phase in Fig. 36. Then, the relationship with low loss and high efficiency of the motor and an increase in maximum torque in a short time, which are the objectives of the present invention, will be described. Also, the relationship with the reduction in the current capacity of the transistors in the drive circuit will be described. Note that these are reflected in smaller size, lighter weight, and lower costs.

[0202] As explained in Figures 33 and 36, the motor with full-pitch winding in Figure 32 generates CCW torque with six stator poles acting at all times. The motor in Figure 34 has two stator pole pairs, so the number of stator poles is doubled, but the operation is basically the same as that of the motor with one stator pole pair in Figure 32, so Figure 32 will be used to explain each current component and each magnetic flux component of each winding. At rotor rotation angle θr=0° shown in Figure 33(b) above, six stator poles, indicated by thick lines in the figure, A-phase and A / phase, B-phase and B / phase, and C-phase and C / phase, can generate CCW torque. Figure 36 shows the current waveforms in the range of θr from 0° to 4°. At this time, in the motor of FIG. 32, A-phase stator pole 328 and A / -phase stator pole 32A apply a magnetomotive force with the two A-phase current Ia components of AD-phase current Iad=Ia+Id of AD-phase winding 321 located at the front and rear of the circumferential direction and EA-phase current Iea=Ie+Ia of EA-phase winding 325, thereby exciting the A-phase magnetic flux φa component, and generating a magnetic attraction force on the rotor pole to generate torque in the CCW direction.

[0203] This torque generation section depends on the circumferential width θsg of the stator poles on the air gap surface and the circumferential width θrg of the rotor poles. In FIG. 33, the stator pole width is θsg=360° / 28=12.857°. The rotor pole width θrg can be less than 360° / 26=13.846°, but θrg=θsg=12.857° is used. Therefore, the poles of A-phase and A / phase can theoretically generate CCW torque when θr is between 0° and 12.857°. Similarly, the sections in which torque can be generated for other phases can be found from the geometric configuration. The torque generation width can be changed by modifying the circumferential width θrg of the rotor poles. The torque generation width can also be changed by skewing the stator and rotor, or by changing the pole shape of the air gap surface from a parallelogram to an irregular shape with projections and recesses.

[0204] 33B, at rotor rotation angle θr=0°, similarly to the A-phase, the B-phase stator poles and the B / -phase stator poles apply magnetomotive force with two B-phase current Ib components of the BE-phase current Ibe=Ib+Ie of the BE-phase winding 322 and the FB-phase current Ifb=If+Ib of the FB-phase winding 326 located in front and behind in the circumferential direction, to excite the B-phase magnetic flux φb component, and generate a torque in the CCW direction by applying a magnetic attractive force to the rotor poles. The C-phase stator poles and the C / -phase stator poles apply magnetomotive force with two C-phase current Ic components of the CF-phase current Icf=Ic+If of the CF-phase winding 323 and the GC-phase current Igc=Ig+Ic of the GC-phase winding 327 located in front and behind in the circumferential direction, to excite the C-phase magnetic flux φc component, and generate a torque in the CCW direction by applying a magnetic attractive force to the rotor poles.

[0205] Therefore, in the range of rotor rotation angles θr=0° to θr=4° in FIG. 33(b), the A-phase current Ia component is passed through the AD-phase winding 321 and the EA-phase winding 325, the B-phase current Ib component is passed through the BE-phase winding 322 and the FB-phase winding 326, and the C-phase current Ic component is passed through the CF-phase winding 323 and the GC-phase winding 327 to generate CCW torque. At this time, the magnetic fluxes to be utilized are three of the seven magnetic fluxes, φa, φb, and φc, and the CCW torque is generated by utilizing six of the seven full-pitch windings. The utilization rate of the windings is as high as 6 / 7, and it can be said that most of the windings are utilized to generate effective torque. Moreover, the currents of each winding are passed so that two components do not overlap in the currents of equations (59) to (65), so the copper loss of the motor can be reduced. Here, the condition for preventing the two current components in equations (59) to (65) from overlapping is that the stator poles used to generate torque must be spaced apart by at least two in the circumferential direction. Compared to the 1 / 3 winding utilization rate of the conventional three-phase switched reluctance motor in Figure 63, this can be improved to (6 / 7) / (1 / 3) = 2.57 times.

[0206] Also, from the perspective of utilizing the magnetic circuit of the soft magnetic material in the stator, it is preferable to generate CCW torque by utilizing one or more stator poles separated in the circumferential direction. As shown in FIG. 17, the magnetic flux of the stator pole can be passed through by using the teeth on both circumferential sides of the stator pole that generates torque. This has the effect of reducing the magnetic resistance of the magnetic circuit in the stator and effectively generating a larger torque. In particular, in order to obtain a large magnetic flux density of 2.0 [T] or more in the vicinity of the air gap portion of the stator pole that generates torque and generate a large torque, it is effective to use the adjacent teeth on both sides. Note that in order to use the teeth on both circumferential sides, it is necessary to use the bypass permanent magnet PMsbi described above.

[0207] Also, it is important that the stator poles used for torque generation are separated from each other by one or more in the circumferential direction, and two full-pitch windings are utilized to generate a large magnetomotive force in the vicinity of the air gap portion. Among the seven magnetic flux components, the magnetomotive force can be concentrated in the vicinity of the air gap of three magnetic flux components by using six out of the seven full-pitch windings. Similar to the above, in order to obtain a large magnetic flux density of 2.0 [T] or more in the vicinity of the air gap portion of the stator pole that generates torque and generate a large torque, this is an effective method. Also, in the operating region where the magnetic flux density is relatively small at light load, the effect of reducing the excitation burden of the magnetic flux can also be expected.

[0208] As shown in the motor cross section of FIG. 32 and the linear development of FIG. 33, the rotor poles directly involved in torque generation are four or more apart in the circumferential direction, and generate torque. As shown in FIG. 8, FIG. 9, FIG. 10, FIG. 11, FIG. 16, FIG. 17, FIG. 18, and FIG. 19, since torque is generated by the unique magnetic action of the present invention, it is not preferable to use adjacent rotor poles on both sides in the circumferential direction at the same time. If necessary, it may be necessary to limit the current control of each phase. As with the stator magnetic flux, the soft magnetic material magnetic paths on both sides in the circumferential direction of the rotor pole that generates torque are utilized to pass the magnetic flux of the rotor pole. In that sense, as shown in FIG. 33, it is preferable to have a configuration and action in which the stator poles used for torque generation are two or more apart in the circumferential direction. Furthermore, the circumferential width θsg of the stator poles of the motor of the present invention can be reduced or enlarged, and the circumferential width θrg of the rotor poles can also be reduced or enlarged. To make the increase and decrease of the magnetic flux passing through smoother, it is possible to reduce or expand θsg and θrg, and it is also possible to devise skew and magnetic pole shapes. However, even in such a case, if the two rotor poles used to generate torque are close in the circumferential direction, there is a possibility that magnetic influence and interference may occur. It is necessary to configure or control the rotor poles adjacent in the circumferential direction so that they are not used at the same time.

[0209] As described above, as an example of the operation of the motor with the stator 1 pole pair in Fig. 32 and the motor with the stator 2 pole pair in Fig. 34, the state in which the rotor rotation angle θr in Fig. 33(b) is between 0° and 4° and phases A, B, and C generate torque. Similarly, in Fig. 33(c), (d), (e), (f), (g), and (h), the stator poles of three phases generate torque in parallel, and the stator poles of the three phases are separated by two or more poles in the circumferential direction. In (c) of Figure 33, phases A, B, and G operate between 4° and 7.9° of θr, in (d) of Figure 33, phases A, F, and G operate between 7.9° and 11.9° of θr, in (e) of Figure 33, phases E, F, and G operate between 11.9° and 15.8° of θr, in (f) of Figure 33, phases E, F, and D operate between 15.8° and 19.8° of θr, in (g) of Figure 33, phases E, C, and D operate between 19.8° and 23.7° of θr, and in (h) of Figure 33, phases B, C, and D operate between 23.7° and 27.7° of θr. It goes around once in a period of 27.7° electrical angle, and in the case of a motor with one stator pole pair in Fig. 32, this operation is repeated 13 times to make one rotation of the rotor. Note that this electrical angle is also 360° for one stator pole pair. In each of these states, the stator poles that generate torque are separated by two or more poles in the circumferential direction. For example, if phase A and phase E generate torque at the same time, adjacent stator poles in the circumferential direction will operate at the same time, but such a state does not exist in each of the operations shown in Fig. 33. The 14S26R stator poles and rotor poles in Figs. 32 and 34 are an excellent combination.

[0210] The above-described operation and function are reflected in the current waveforms of each full-pitch winding in FIG. 36. Current is passed through most of the 6 / 7 section, and all of the current provides a magnetomotive force to generate torque, effectively generating torque. Since the two current components on the right side of Equation (65) from Equation (59) are not passed simultaneously, the current capacity of the transistors can be small. As can be seen from the current waveforms in FIG. 36 and the drive circuit in FIG. 35, 12 of the 14 transistors shown in FIG. 35 are used to pass current through the 12 full-pitch windings to supply power and generate torque. The drive circuit in FIG. 35, i.e., the utilization rate of each transistor is as high as 6 / 7, and the total current capacity of the drive circuit can be reduced, making it possible to reduce the size and cost of the drive circuit. For example, a three-phase AC surface permanent magnet synchronous motor SPMSM or an internal magnet synchronous motor IPMSM is widely used, and the utilization rate of the drive circuit and transistors is 1 / 3. On average, two of six transistors are used to supply power to the motor, so the utilization rate is 1 / 3. When the motors in Figs. 32 and 34 are driven by the drive circuit in Fig. 35, the utilization rate of the SPMSM is (6 / 7) / (1 / 3) = 2.57 times that of the IPMSM when driven by a conventional three-phase drive circuit. Therefore, although the drive circuit in Fig. 35 has a large number of elements, the total current capacity of the drive circuit can be significantly reduced, making it possible to reduce the size and cost. Note that in drive circuits for motors exceeding 10 kW, IGBTs and the like are often used in parallel, so it can be assumed that the number of actual power elements will not increase significantly.

[0211] As described above, the full-pitch windings in Figs. 32 and 34 have a serious problem of superimposing voltage components of other phases as shown in equations (66) to (66). However, the drive circuit in Fig. 35 can be configured such that the voltage components of other phases are not induced in the voltage across the two windings due to the offsetting effect as shown in equations (73) to (79). There is also a method of constantly passing a current in a range where the magnetic flux density of each phase increases, that is, a current component equivalent to the field current, to reduce the voltage burden on the drive circuit, which will be described later. One of the factors that makes it possible to realize the drive circuit in Fig. 35 is that the driving current is a direct current. In the case of a direct current, it is relatively easy to pass two or more current components together and to branch them. A drive circuit for an alternating current is complicated in order to supply positive and negative currents.

[0212] Next, another embodiment of claim 5 will be described with reference to (a), (b), (c), (d), and (e) in Fig. 37. It is a linear development showing the operation of a motor with full-pitch winding of 14S18R. This is an example where Ns in equation (57) is 3 and Nr in equation (58) is 4, and it is a type of 7-phase motor. Although a cross-sectional view of the 14S18R motor is not shown, its stator is the same as that of the 14S26R in Fig. 32, and the rotor has a similar structure, but the number of rotor poles is 18. Fig. 37(a) shows the shape of each stator pole facing the air gap, and the circumferential stator pole width is θsg=360° / 28=12.857°. Fig. 37 is expressed in the same way as the linear development of 14S26R in Fig. 33. Fig. 37(b) shows the shape of each rotor pole facing the air gap when the rotor rotation angle θr=0°. The rotor pole pitch is θppr=360 / 18=20°, and the rotor pole width θrg is 12.857°, the same as the stator pole width θsg. The distance between the rotor poles is a relatively large 7.143°. As mentioned above, these motors are constructed with point symmetry, including the full-pitch windings, with respect to the rotor center point, so for example, the A-phase stator pole and the A / phase stator operate in the same way. However, because they are driven by DC current, the direction of the current and the direction of the magnetic flux are not symmetrical but are opposites.

[0213] At θr=0°, as shown by the thick line in FIG. 37(b), it is possible to generate CCW torque at the stator poles of phases A, D, and G. Similarly, at θr=5.7° in FIG. 37(c), it is possible to generate CCW torque at phases A, D, and E. At θr=11.4° in FIG. 37(d), it is possible to generate CCW torque at phases A, B, and E. At θr=17.1° in FIG. 37(e), it is possible to generate CCW torque at phases B, E, and F. By repeating this operation, the stator poles that generate CCW torque change with the rotor rotation angle θr, and the torque generation pattern completes one cycle at 40°, and 360° in 9 cycles, completing one rotation of the rotor. At any rotor rotation angle θr, it is possible to generate CCW torque at six stator poles.

[0214] However, as shown in the figure, in either case, a CCW torque is generated by three stator poles arranged consecutively in the circumferential direction, which may cause an undesirable state. One problem is that the two current components on the right side of Equations (59) to (65) are simultaneously applied to the full-pitch windings arranged between the three stator poles, and copper loss increases by two times because copper loss is proportional to the square of the current value. In addition, when the stator pole width θsg and the rotor pole width θrg are made larger, there is a timing when two stator poles adjacent in the circumferential direction excite the same rotor pole, and the two stator poles are magnetically connected through the soft magnetic material part at the tip of the rotor pole, and magnetic flux passes through. In addition, a double reverse magnetomotive force acts on the bypass permanent magnet PMsbi between two adjacent stator poles. In addition, the leakage magnetic flux between both stator poles also increases. For these reasons, when using a motor with 14S18R full-pitch windings, more effective control can be achieved by, for example, taking care to prevent an increase in copper loss and devising current control for each phase. On the other hand, while a motor with 14S18R full-pitch windings is shown in FIG. 37(b) for a rotor pole width θrg = 12.857°, the rotor pole width θrg can be expanded up to a maximum of 20°, providing a high degree of freedom. Furthermore, the torque characteristics can be improved.

[0215] Next, another embodiment of claim 5 will be described with reference to (f), (g), (h) and (i) in Fig. 37. It is a linear development showing the operation of a motor with full-pitch winding of 14S22R. This is an example where Ns in equation (57) is 3 and Nr in equation (58) is 5, and it is a type of 7-phase motor. Although a cross-sectional view of the 14S22R motor is not shown, its stator is the same as that of the 14S26R in Fig. 32, and the rotor has a similar structure, but the number of rotor poles is 22. Fig. 37(f) shows the shape of each rotor pole facing the air gap when the rotor rotation angle θr=0°. The pitch of the rotor poles θppr=360 / 22=16.364°, and the rotor pole width θrg is 12.857°, the same as the stator pole width θsg.

[0216] At θr=0°, as shown by the thick line in FIG. 37(f), it is possible to generate CCW torque with the stator poles of phases A, D, and F. Similarly, at θr=3.5° in FIG. 37(g), it is possible to generate CCW torque with phases A and F. At θr=4.7° in FIG. 37(h), it is possible to generate CCW torque with phases A, C, and F. At θr=8.2° in FIG. 37(i), it is possible to generate CCW torque with phases A and C. By repeating this operation, the stator poles generating CCW torque change with the rotor rotation angle θr, and the torque generation pattern completes one cycle at 32.7°, and the rotor completes one rotation at 360° in 11 cycles. As the rotor rotates, it is possible to generate CCW torque with approximately four stator poles. Compared to the 14S26R motor in FIG. 32, it generates 2 / 3 of the torque. The utilization rate of the windings and the transistors is reduced to 2 / 3 of that of the 14S26R.

[0217] Next, other embodiments of claim 5 are shown and explained in Fig. 37(j), (k), (l) and (m). It is a linear development showing the operation of a motor with full-pitch winding of 14S30R. This is an example where Ns in equation (57) is 3 and Nr in equation (58) is 7, and it is a type of 7-phase motor. Although a cross-sectional view of the 14S30R motor is not shown, its stator is the same as that of the 14S26R in Fig. 32, and the rotor has a similar structure, but the number of rotor poles is 30. Fig. 37(j) shows the shape of each rotor pole facing the air gap when the rotor rotation angle θr=0°. The rotor pole pitch θppr=360 / 30=12°, and the rotor pole width θrg is 12.0°.

[0218] At θr=0°, as shown by the thick line in FIG. 37(j), it is possible to generate CCW torque at the stator poles of phases A, G, and F. Similarly, at θr=3.4° in FIG. 37(k), it is possible to generate CCW torque at phases A, B, and G. At θr=6.7° in FIG. 37(l), it is possible to generate CCW torque at phases A, B, and C. At θr=10.3° in FIG. 37(m), it is possible to generate CCW torque at phases B, C, and D. By repeating this operation, the stator poles that generate CCW torque change with the rotor rotation angle θr, and the torque generation pattern completes one cycle at 24°, and 360° at 15 cycles, completing one rotation of the rotor. As the rotor rotates, it is possible to generate CCW torque at six stator poles. The motor has the same six stator poles as the 14S26R motor in Figure 32 above, and in this respect it is the same. The utilization rates of the windings and transistors are also the same. However, because there are a large number of rotor poles (30), it is necessary to devise a layout for the rotor's bypass permanent magnets PMrbi and a soft magnetic magnetic path for the rotor poles. For example, if the rotor is positioned on the outer periphery as an outer rotor structure, the space for the rotor poles will be wider, allowing for greater freedom in design.

[0219] Next, as another embodiment of claim 5, an example of a 10S18R motor configuration will be described with reference to Figs. 38, 39, and 40. Fig. 38 is an example of a cross-sectional view of a motor with a 10S18R full-pitch winding. This is an example of a 5-phase motor in which Ns in equation (57) is 2 and Nr in equation (58) is 4. 387 is an A-phase stator S-pole magnetic pole, and 388 is an A / phase stator N-pole magnetic pole, which pass through the component of the A-phase magnetic flux φa shown in the figure. The configuration is point-symmetrical with respect to the rotor center. Similarly, 389 and 38A are a B-phase stator S-pole magnetic pole and a B / phase stator N-pole magnetic pole, which pass through the component of the B-phase magnetic flux φb shown in the figure. 38B and 38C are a C-phase stator S-pole magnetic pole and a C / phase stator N-pole magnetic pole, which pass through the component of the C-phase magnetic flux φc shown in the figure. Reference numerals 38D and 38E denote a D-phase stator south pole and a D / phase stator north pole, through which the component of the D-phase magnetic flux φd shown in the figure passes. Reference numerals 38F and 38G denote an E-phase stator south pole and an E / phase stator north pole, through which the component of the E-phase magnetic flux φe shown in the figure passes. The phase of each stator pole is indicated in parentheses on the outer periphery of the stator.

[0220] Reference numeral 381 denotes an AC-phase full-pitch winding, which is wound around slots spaced 180° apart at an electrical angle that is 1 / 2 of the electrical angle of 360° for one stator pole pair, with the coil ends shown by dashed lines connected to each other, and which carries an AC-phase current Iac. Similarly, reference numeral 382 denotes a BD-phase full-pitch winding, which carries a BD-phase current Ibd. Reference numeral 383 denotes a CE-phase full-pitch winding, which carries a CEBD-phase current Ice. Reference numeral 384 denotes a DA-phase full-pitch winding, which carries a DA-phase current Ida. Reference numeral 385 denotes an EB-phase full-pitch winding, which carries an EB-phase current Ieb. These currents are related by the following equation for the five-phase currents, similar to the currents in equations (59) to (65) for the seven-phase motor in Figure 32. Iac=Ia+Ic (80) Ibd=Ib+Id (81) Ice=Ic+Ie (82) Ida=Id+Ia (83) Ieb = Ie + Ib (84)

[0221] On the other hand, the magnetic flux components φa, φb, φc, φd, and φe of all five phases shown in Figure 38 are linked to all full-pitch windings, and are induced in the voltages of each full-pitch winding according to Faraday's law of electromagnetic induction, resulting in the following relationship. Note that the number of turns of a full-pitch winding is Nw / 2. The absolute value of the flux linkage of each winding also includes the magnetic flux component, but assuming that the magnetic flux is constant and does not appear in the time rate of change, the following five-phase expression is used, similar to the above equations (48), (49), and (50) for the three-phase case. Vack=Nw / 2×d(φa+φb+φc-φd-φe) / dt =Vak+Vbk+Vck-Vdk-Vek (85) Vbdk=Nw / 2×d(-φa+φb+φc+φd-φe) / dt =-Vak+Vbk+Vck+Vdk+Vek (86) Vcek=Nw / 2×d(-φa-φb+φc+φd+φe) / dt =-Vak-Vbk+Vck+Vdk+Vek (87) Vdak=Nw / 2×d(φa-φb-φc+φd+φe) / dt =Vak-Vbk-Vck+Vdk+Vek (88) Vebk=Nw / 2×d(φa+φb-φc-φd+φe) / dt =Vak+Vbk-Vck-Vdk+Vek (89)

[0222] The complex voltages of each of these five-phase full-pitch windings can be simplified in the same way as the seven-phase equations (73) to (79), resulting in the following five-phase voltage relationship: Vack+Vdak=Nw / 2×d(2×φa)=Vak (90) Vdak+Vbdk=Nw / 2×d(2×φd)=Vdk (91) Vbdk+Vebk=Nw / 2×d(2×φb)=Vbk (92) Vebk+Vcek=Nw / 2×d(2×φe)=Vek (93) Vcek+Vack=Nw / 2×d(2×φc)=Vck (94)

[0223] As in the case of the seven-phase motor described above, according to Ampere's law of circular integral, if the component of the A-phase current Ia is supplied as the AC-phase current Iac of equation (80) to the AC-phase winding 381 in FIG. 38, and at the same time, the component of the A-phase current Ia is supplied as the DA-phase current Ida of equation (83) to the DA-phase winding 384, the A-phase magnetic flux φa in FIG. 38 is excited, and at the same time, the component of the A-phase current Ia in both windings does not affect the magnetic flux components of the other phases. Equation (90) is the flip side of this, and according to Faraday's law of electromagnetic induction, the sum of the AC-phase voltage Vack and the DA-phase voltage Vdak is related only to the A-phase magnetic flux φa and the A-phase voltage Vak, and is not affected by the magnetic flux of the other phases. Equations (91) to (94) have a similar relationship. These relationships are also related to the motor configuration that is point-symmetric with respect to the rotor center. In addition, there is a control method that is less susceptible to the effects of many other phase voltages by applying the simplification method of these voltages, which will be explained later.

[0224] Next, FIG. 39 shows a linear development diagram showing the operation of the 10S18R motor in FIG. 38 to generate CCW torque. This is a development diagram similar to FIG. 12 and FIG. 33. The rotor rotation angle in FIG. 38 is θr=0°, and the CCW direction of the motor is the right direction in FIG. 39. In each row of FIG. 39, the section where CCW torque can be generated is shown by a thick line above the rotor pole shape. In FIG. 39, the stator pole width is θsg=360° / 20=18°. The rotor pole width θrg can be 360° / 18=20° or less, but in FIG. 33, it is θrg=18°. Note that the stator pole width θsg and the rotor pole width θrg can be increased or decreased, and can be optimized according to the motor requirements, and the pole shape can also be changed.

[0225] FIG. 39(a) shows the shape of each stator pole facing the air gap surface. 391 corresponds to the A-phase stator south pole 387 in FIG. 38. Similarly, 392, 393, 394, and 395 in FIG. 39 correspond to 389, 38B, 38D, and 38F in FIG. 38. The horizontal axis θr in FIG. 39 is a little confusing, but θr on the horizontal axis shows the electrical angle position of one stator pole pair at an electrical angle of 360°, and is also the electrical angle position in the rotation direction of each part of the stator. The rotational position of the rotor is shown on the left side of each row in FIG. 39. On the paper surface of FIG. 39, each part of the rotor is moved to the right as shown in (b), (c), (d), and (e). FIG. 39(b) shows the rotor rotational position θr=0°, which is the starting point of rotor rotation. 396 in (b) of FIG. 39 is a rotor north pole magnetic pole, which corresponds to the rotor north pole magnetic pole 386 in FIG. 38. On the paper surface of FIG. 39, the left position of the A-phase stator south pole magnetic pole 391 coincides with the right position of the rotor north pole magnetic pole 396. In this position, a total of four stator poles, the A-phase stator south pole magnetic pole 391, the A / phase stator north pole magnetic pole, the B-phase stator south pole magnetic pole 392, and the B / phase stator north pole magnetic pole, can generate an attractive force in the CCW direction, and are shown by thick lines in four places at the top of (b) of FIG. 39. A torque in the CCW direction is generated using the A-phase magnetic flux φa and the B-phase magnetic flux φb shown in FIG. 38. In (c) of FIG. 39, the rotor rotation position θr=8°, and the B-phase stator south pole magnetic pole and the B / phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The E-phase stator south pole magnetic pole 395 and the E / phase stator north pole magnetic pole start to generate an attractive force in the CCW direction. In (d) of FIG. 39, θr=16°, and the A-phase stator south pole magnetic pole and the A / phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The D-phase stator south pole magnetic pole 394 and the D / phase stator north pole magnetic pole start to generate an attractive force in the CCW direction. In (e) of FIG. 39, θr=24°, and the E-phase stator south pole magnetic pole 395 and the E / phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The C-phase stator south pole magnetic pole 393 and the C / phase stator north pole magnetic pole start to generate an attractive force in the CCW direction. In (f) of FIG. 39, θr=32°, and the D-phase stator south pole magnetic pole and the D / phase stator north pole magnetic pole can no longer generate an attractive force in the CCW direction. The B-phase stator south pole 392 and the B / phase stator north pole start to generate an attractive force in the CCW direction. Figure 39(g) shows θr=40°, and this state returns to the same state as Figure 39(b).The motors in Figures 38 and 39 repeat the same operation nine times in a 40° cycle, making one rotation of the rotor.

[0226] Next, an example of driving a 10S18R motor will be described. A 5-phase 10S18R motor can be driven in the same way as the 7-phase motor of FIG. 34 with 2 stator pole pairs is driven by energizing it with the drive circuit of FIG. 35. Although not shown, the motor of FIG. 38 can be modified to have 2 stator pole pairs to form a motor with 10 full-pitch windings. The drive circuit can be a 5-phase drive circuit by removing the drive circuits for 2 phases such as transistors 35B, 35C, 35D, and 35F from the 7-phase drive circuit of FIG. 35. Here, the 7-phase AD, EA, BE, FB, and CF phases are replaced with the 5-phase AC, BD, CE, DA, and EB phases, respectively, and are controlled as the currents of Equations (80) to (84). In this case, in a five-phase drive circuit modified from the drive circuit in Fig. 35, the voltages corresponding to both ends of the two windings arranged above and below on the paper surface of Fig. 35 are the voltages given by equations (90) to (94), and are therefore not affected by the magnetic fluxes of other phases or the voltages of other phases. As a result, the currents given by equations (80) to (84) can be more easily controlled.

[0227] Next, Fig. 40 shows an example of current waveforms when a motor with two stator pole pairs in the configuration of Fig. 38 is energized in the operating sequence of Fig. 39 to generate torque in the CCW direction and is controlled by the above-mentioned five-phase drive circuit. In the operating sequence of Fig. 39, the phase magnetic flux components of φa, φb, φc, φd, and φe in Fig. 38 are excited with the current components Ia in Fig. 40(a), Ib in Fig. 40(b), Ic in Fig. 40(c), Id in Fig. 40(d), and Ie in Fig. 40(e). The currents energized to the full-pitch windings in Fig. 38 are Iac, Ibd, Ice, Ida, and Ieb in Fig. 40(f) to (j), and can be calculated from the current components Ia, Ib, Ic, Id, and Ie in Fig. 40(a) to (e) according to the current relationships of Equations (80) to (84). The operation in FIG. 39 has a cycle of 40°, and FIG. 40 shows a range of 80°, which is two cycles.

[0228] The full-pitch winding currents Iac, Ibd, Ice, Ida, and Ieb in (f) to (j) of FIG. 40 show good characteristics. Each current flows in the 4 / 5 section, and each current contributes to torque generation. The winding utilization factor is large at 4 / 5=0.8, which is slightly lower than the 7-phase winding utilization factor of 6 / 7=0.857 shown in FIG. 36, but the difference is small. In addition, the two current components on the right side of equations (80) to (84) for each full-pitch winding current are not flowing simultaneously, so copper loss does not increase exponentially. These are indicators of reduced copper loss and high efficiency of the motor. The utilization factor of the drive transistor is also 4 / 5, and the two current components are not flowing simultaneously, so the current capacity of the drive circuit can be reduced. Note that each current waveform in FIG. 40 shows an example of a rectangular shape, but of course various increasing and decreasing waveforms can be used, such as a trapezoidal waveform with a slope when the current increases and decreases. As will be explained later, a current sufficient to excite the magnetic flux can be constantly applied, or a field winding can be provided on the rotor and applied with current.

[0229] Also, as shown in Figures 38, 39, and 40, a 10S18R 5-phase full-pitch winding motor generates CCW torque by utilizing stator poles that are one or more away from each other in the circumferential direction. Therefore, as shown in Figure 17, the magnetic flux of the stator pole that generates torque can be passed through the teeth on both sides of the circumferential direction of the stator pole. This has the effect of reducing the magnetic resistance of the magnetic circuit in the stator and effectively generating a larger torque. In particular, it is effective to use the teeth on both sides to obtain a large magnetic flux density of 2.0 [T] or more near the air gap of the stator pole that generates torque and generate a large torque. In order to use the teeth on both sides of the circumferential direction, it is necessary to use the bypass permanent magnet PMsbi described above.

[0230] In addition, since the stator poles that generate torque are spaced apart in the circumferential direction by one or more, the rotor poles that generate torque are also spaced apart in the circumferential direction, and magnetic interference between the rotor poles is small. In addition, the utilization rate of each transistor in the drive circuit of the motor with the configuration of Figure 38 that has two stator pole pairs is 4 / 5 = 0.8, which is a large value. There is also the advantage of simplifying the DC current drive. The total current capacity of the drive circuit can be reduced, making it possible to reduce the size and cost. In addition, the 10S18R five-phase full-pitch winding motor in Figure 38 has excellent torque continuity in terms of torque ripple as well. The circumferential width θsg of the stator poles and the circumferential width θrg of the rotor poles can be increased or decreased to optimize the motor.

[0231] Next, another embodiment of claim 5 will be described with reference to (a), (b), (c), (d), and (e) in Fig. 41. It is a linear development showing the operation of a motor with full-pitch winding of 10S14R. This is an example where Ns in equation (57) is 2 and Nr in equation (58) is 3, and it is a type of 5-phase motor. Although a cross-sectional view of the 10S14R motor is not shown, its stator is the same as that of the 10S18R in Fig. 38, and the rotor has a similar structure, but the number of rotor poles is 14. (a) in Fig. 41 is the same as that in Fig. 39, and the stator pole width is θsg = 360° / 20 = 18°. 396 in (b) in Fig. 41 is the rotor N pole, and shows the shape of each rotor pole facing the air gap at the rotor rotation angle θr = 0°. The rotor pole pitch is θppr=360 / 14=25.714°, and the rotor pole width θrg is 18°, the same as the stator pole width θsg. The distance between the rotor poles is a relatively large 7.714°. As mentioned above, these motors are constructed with point symmetry, including the full-pitch winding, with respect to the rotor center point, so for example, the A-phase stator pole and the A / -phase stator operate in the same way. However, because they are driven by DC current, the current direction and magnetic flux direction of the A-phase and A / -phase are not symmetrical, but are opposite to each other with respect to the rotor center point.

[0232] At θr=0°, as shown by the thick line in FIG. 41(b), a CCW torque can be generated by the four stator poles of A and C phases. Similarly, at θr=7.7° in FIG. 41(c), a CCW torque can be generated by the A phase. At θr=10.3° in FIG. 41(d), a CCW torque can be generated by the A and D phases. At θr=18.0° in FIG. 41(e), a CCW torque can be generated by the D phase. By repeating this operation, the torque generation pattern completes one cycle at 51.43°, and seven cycles make 360°, and the rotor rotates once. With these characteristics, only two stator poles can continuously generate torque. However, the rotor pole width θrg of the 10S14R can be expanded from 18° in FIG. 41 to a maximum of 25.714°, and the stator pole width θsg can also be expanded, so torque can be generated by four stator poles. However, because there are two stator poles lined up in the circumferential direction, copper loss increases in one of the three full-pitch windings that are energized. The current capacity of the transistors used also doubles. However, there is no problem with the current capacity of the transistors as long as it does not exceed their maximum current value. When the motor outputs maximum torque, there is a problem with the current capacity of the transistors.

[0233] Next, another embodiment of claim 5 will be described with reference to (f), (g), (h), and (i) in FIG. 41. It is a linear development diagram showing the operation of a motor with full-pitch winding of 10S22R. This is an example where Ns in equation (57) is 2 and Nr in equation (58) is 5, and this is a type of 5-phase motor. The stator is the same as the 10S18R stator in FIG. 38, and the number of rotor poles is 22. The pitch of the rotor poles in FIG. 41(f) is θppr=360 / 22=16.364°. At θr=0°, as shown by the thick line in FIG. 41(f), a CCW torque can be generated by the four stator poles of phases A and E. Similarly, at θr=6.5° in FIG. 41(g), a CCW torque can be generated by phases A and B. At θr=13.1° in FIG. 41(h), CCW torque can be generated by phases B and C. At θr=19.6° in FIG. 41(i), CCW torque can be generated by phases C and D. By repeating this operation, the torque generation pattern completes one cycle at 32.73°, and the rotor completes one rotation at 360° in 11 cycles. With these characteristics, torque can be continuously generated by four stator poles. Moreover, since the stator poles that generate torque are separated by two or more in the circumferential direction, the two currents shown on the right side of equations (80) to (84) do not overlap, which reduces copper loss and the current capacity of each transistor that conducts current. It can also be said that the utilization rate of the windings is 4 / 5 and the utilization rate of the transistors is 4 / 5.

[0234] As for claim 5, the embodiments of 3-phase, 7-phase, and 5-phase have been described. Furthermore, configurations of other phases such as 9-phase, 11-phase, and 13-phase can be realized. As the number of phases increases, the motor becomes more complicated, but in principle, the motor does not become larger, and the burden on each part of the motor, such as the permanent magnets, is reduced. The drive circuit also becomes more complicated, but in principle, the total current capacity of the drive circuit does not increase. As the number of phases increases, the control becomes more complicated, but the recent increase in speed, high integration, and cost reduction of microprocessors and the like have reduced the burden on the calculation capacity of the control device. In addition, the motor of claim 5 can be partially deleted or added. Various modifications are also possible. For example, the number of stator pole pairs can be configured to be 4, and the stator poles for two pole pairs can be deleted. The stator of another type of motor can be added to the deleted space. In other words, partial motor configurations can be combined.

[0235] Next, a cross-sectional view of a motor according to an embodiment of claim 6 is shown in FIG. 42. This is a two-phase 4S10R motor configuration, with a small number of stator poles (4) and a rotor pole number (10). In claim 5 shown in formulas (57) and (58), the stator and rotor are arranged equally in the circumferential direction. In contrast, the circumferential arrangement of the stator poles in FIG. 42 is not equal. It is an uneven arrangement. Also, it is not easy to obtain continuous rotational torque with stator poles excited by two-phase DC current, so this motor is limited to unidirectional rotation and has unidirectional magnetic properties that are uneven, so that unidirectional torque can be obtained continuously. This is an asymmetric motor in CCW and CW.

[0236] The shape of the air gap surface of the motor in FIG. 42 is shown in a linear development in FIG. 43. FIG. 43 shows the circumferential positional relationship between the stator poles and the rotor poles, and is a linear development diagram showing the torque generation operation. In FIG. 42 and FIG. 43, the same parts are denoted by the same reference numerals. FIG. 43(a) shows the air gap surface shape of the stator poles, and FIG. 43(b) shows the air gap surface shape of each rotor pole at a rotor rotation angle θr=-4°. In FIG. 42 and FIG. 43(a) and (b), 431 is the A-phase stator S-pole magnetic pole, 421 is the A-phase winding, 432 is the A / phase stator N-pole magnetic pole, and 422 is the A / phase winding. 433 is the B-phase stator S-pole magnetic pole, 423 is the B-phase winding, 434 is the B / phase stator N-pole magnetic pole, and 424 is the B / phase winding. Each winding is a concentrated winding. Each stator pole is arranged with S poles and N poles alternately in the circumferential direction. Permanent magnets such as 425 are arranged between the stator poles in the direction of the poles, with the polarity indicated by an arrow. 426 indicates the bias magnetic flux of the permanent magnet when no current is flowing through the stator. Note that in Fig. 42, an example with one stator pole pair is shown to show the basic shape, and the shape of the permanent magnet such as 425 is strangely long in an arc shape, but in actual motor design, three pole pairs, four pole pairs, or more pole pairs are assumed, and the shape of the stator permanent magnet can also be designed to be a short flat shape in the circumferential direction. In addition, when the number of pole pairs is two or more, the imbalance of the attractive forces toward the center of the A and B phases can also be eliminated.

[0237] The rotor in Fig. 42 has 10 magnetic poles, and is the same rotor example as Fig. 1, Fig. 14, etc. The rotor rotation position in Fig. 42 is θr = 0°, which corresponds to position (c) in the linear development diagram in Fig. 43. 435 and 437 are the north magnetic poles of the rotor, and 436 is the south magnetic pole of the rotor. Permanent magnets such as 427 are arranged between each rotor pole, with the polarity indicated by an arrow in the direction of the magnetic poles. 426 indicates the bias magnetic flux of the permanent magnet when no current is flowing through the stator.

[0238] As described above, (a) of FIG. 43 shows the air gap surface shape of the stator pole in FIG. 42. (a) to (h) of FIG. 43 show the air gap surface shape of each rotor pole at each rotor rotation position. In order to show the shape and operation of each part of the motor in FIG. 42, FIG. 44 shows partial enlarged views of (a) and (c) of FIG. 43. In FIG. 44, 431 is an A-phase stator S-pole, 435 and 437 are rotor N-pole, and 436 is rotor S-pole, and these symbols are the same as those in FIG. 42 and FIG. 43. The rotor axial length of the front part of the stator pole 431 in the CW direction is small as Lr1, and the rotor axial length in the CCW direction is Lr2, and here, an example in which Lr1 is 1 / 2 of Lr2 is shown. The circumferential length of the part Lr1 is θsb, and the circumferential length of the part Lr2 is θsc.

[0239] The rotor pole pitch θppr is 36°, and in order to obtain continuous torque by alternately driving the A and B phases, the circumferential length θsa of the stator poles must be greater than the rotor pole pitch θppr, as determined by the following equation: θsa > θppr (95) The rotor rotates in the CCW direction, and the rotor axial length θra is greater than θsb in order to continue generating torque in the portion Lr1, and the following equation is a necessary condition. θra > θsb (96) Furthermore, in order for the rotor to rotate in the CCW direction and move a total distance greater than the magnetic pole pitch θppr of the rotor, the sum of θra and θsb is greater than θppr, and the following condition is satisfied: θra+θsb>θppr (97) When the rotor rotates CCW and generates torque for the first time, in order to prevent the torques generated by rotor poles 435 and 437 from interfering with each other, the necessary condition is that the sum of the circumferential length θra of the rotor pole and the above θsa is smaller than twice the above θppr, as shown in the following equation. θra+θsa < 2×θppr (98)

[0240] 43 and 44 show examples of θppr=36°, θsa=40°, θsb=15°, θsc=25°, and θra=27°. Although FIG. 44 shows a two-stage shape of the stator pole 431, other shapes such as a trapezoidal shape may be used as long as the axial length of the right side of the stator pole 431 is greater than that of the left side of the stator pole 431 on the paper surface of FIG. 44. In addition, in the case of a motor having two or more stator pole pairs, if the average value of the magnetic resistance of multiple A-phase stator S poles is distributed like the magnetic resistance of the stator pole 431 and the magnetic resistance is smaller on the right side on the paper surface, a CCW torque can be generated. In addition, the magnetic resistance distribution may be the same not only in the stator pole air gap surface shape but also inside the stator pole. Rather, creating holes or slits in an electromagnetic steel sheet inside the stator pole to create an equivalent magnetic resistance distribution like 431 is easier to process since it only requires devising the internal shape of the electromagnetic steel sheet, and therefore results in superior motor productivity.

[0241] Next, FIG. 43, which is a development diagram of the operation of the motor in FIG. 42, will be described. FIG. 43(b) shows the rotor rotation angle θr at which the rotor N-pole magnetic pole 435 approaches the A-phase stator S-pole magnetic pole 431 when the rotor rotates CCW at the rotor rotation angle θr=-4°, making it possible to generate CCW torque. At the same time, the A / phase stator N-pole magnetic pole 432 can also generate CCW torque with the rotor S-pole magnetic pole. Also, at this rotation angle, the B-phase stator S-pole magnetic pole 433 and the B / phase stator N-pole magnetic pole 434 can also generate CCW torque. FIG. 43(c) shows the rotor rotation angle θr=0°, making it impossible for the B-phase stator S-pole magnetic pole 433 and the B / phase stator N-pole magnetic pole 434 to generate CCW torque. FIG. 43(d) shows that at rotor rotation angle θr=11°, rotor north pole 435 approaches the portion of stator south pole 431 with rotor axial length Lr2 on the right side of the page.

[0242] FIG. 43(e) shows the rotor rotation angle θr at which the rotor N-pole magnetic pole approaches the B-phase stator S-pole magnetic pole 433 when the rotor rotates CCW at the rotor rotation angle θr=-32°, making it possible to generate CCW torque. At this time, the B / phase stator N-pole magnetic pole 434 can also generate CCW torque with the rotor S-pole magnetic pole. The rotor is at a rotor rotation angle θr that is advanced by a rotor pole pitch θppr=36° from the rotor rotation angle of FIG. 43(b), and is also in a position where the rotor N-pole magnetic pole and the rotor S-pole magnetic pole are swapped compared to FIG. 43(b). FIG. 43(f) shows the rotor rotation angle θr=36°, making it impossible for the A-phase stator S-pole magnetic pole 431 and the A / phase stator N-pole magnetic pole 432 to generate CCW torque. Figure 43(g) shows that the rotor north pole is at a position where the rotor rotation angle θr=47°, and the rotor axial length Lr2 approaches the right side of the stator south pole 433 on the page. Figure 43(h) shows that the rotor is at a position where the rotor rotation angle θr=68°, and is in the same state as Figure 43(b), and the rotor has rotated CCW by 72°, twice the rotor pole pitch θppr. These operations are repeated five times, and the rotor rotation angle θr reaches 360°, making one rotation.

[0243] Next, FIG. 45 shows examples of current waveforms that drive the motors shown in FIG. 42, FIG. 43, and FIG. 44. The horizontal axis of FIG. 45 is the rotor rotation angle θr. When the rotor rotates at a constant rotation speed, the waveform will be the same as that of FIG. 45 on the time axis. FIG. 45(a) shows the A-phase current Ia, which flows from the rotor rotation angle θr=-4° to θr=36°, and further from θr=68° to θr=108°. FIG. 45(b) shows the B-phase current Ib, which is a current with a phase lag of 36° relative to the A-phase current Ia, and has the same current waveform. With the two currents, the A-phase current Ia and the B-phase current Ib, it is possible to generate torque continuously, although only in one direction.

[0244] The drive circuits for the A-phase current Ia and the B-phase current Ib can be energized using two of the circuits in FIG. 25. In this case, 257 is a winding in which the A-phase winding 421 and the A / phase winding 422 are connected in series. 258 is a winding in which the B-phase winding 423 and the B / phase winding 424 are connected in series. It can be driven simply with four transistors. It can also be energized using the drive circuit in FIG. 46. Since it can be driven with two transistors, it is a simpler drive circuit. 462 and 463 are capacitors, and point 461 is the neutral point of the circuit. 464 is a winding in which the A-phase winding 421 and the A / phase winding 422 are connected in series, and a transistor 466 is used to energize the A-phase current Ia. 465 is a winding in which the B-phase winding 423 and the B / phase winding 424 are connected in series, and a transistor 467 is used to energize the B-phase current Ib. Reference numerals 468 and 469 denote diodes for the circuit. Also, capacitors 462 and 463 may be replaced with two DC power sources, one positive and one negative. In this way, unidirectional rotation can be realized with simple configurations such as those shown in Figures 42, 43, 44, 45, and 46. There are many applications for unidirectional rotation, and it is desirable to realize it with a simpler configuration, and there are many applications with strict cost requirements.

[0245] In the 4S10R two-phase motor of FIG. 42, the number of phases of the stator poles Ps is 2, and the phase difference between the two stator poles and the rotor poles is 1 / 2 the sum of the N-pole rotor pole pitch θppr and the S-pole rotor pole pitch θppr. That is, it can be calculated as (2×θppr) / Nph=θppr. The number of rotor poles of the 4S10R two-phase motor of FIG. 42 may be increased, and other motor components may be present on the circumference of the stator poles. In a three-phase motor, the number of phases is Nph=3, and the three-phase stator poles may be arranged so that the relative phase difference between the rotor poles is (2×θppr) / Nph=2 / 3×θppr. In a four-phase motor, the number of phases is Nph=4, and the four-phase stator poles may be arranged so that the relative phase difference between the rotor poles is (2×θppr) / Nph=1 / 4×θppr. In other words, the relative phase difference with respect to the rotor poles may be set to 0, 1 / 4×θppr, 2 / 4×θppr, or 3 / 4×θppr.

[0246] In addition, when the number of stator poles Nps is Nps=2+4×Ns in equation (57) and the number of rotor poles Npr is Npr=2+4×Nr in equation (58), and they are uniformly arranged in the circumferential direction, the full-pitch windings can be arranged equally as shown in FIG. 26, FIG. 32, FIG. 38, etc. However, since the electromagnetic characteristics of the motor of the present invention are generated by the relative relationship between the multiple stator poles and the multiple rotor poles that face each other through the air gap, there are cases where the number of rotor poles different from that in equation (58) is effective for the multiple stator poles. For example, there is a case where the number of stator poles Nps=14 with Ns of equation (57) being 3, and the number of rotor poles Npr=24 that does not follow equation (58). This motor has two fewer rotor poles than the configuration in FIG. 32. In this case, the polarity of the rotor poles on the 180° opposite side of the rotor is the same, which is a problem in that driving with full-pitch windings is inconvenient. As an example of a solution to this problem, 14 stator poles arranged in the circumferential direction are divided into two groups of seven poles each, and two spaces for one rotor pole are provided between the two groups. Then, two rotor poles are added, so that Npr = 24 + 2 = 26. As a result, 12 rotor poles face the seven stator poles of one group via an air gap, and 12 rotor poles face the seven stator poles of the other group via an air gap. For example, the polarity of the rotor poles facing the stator poles of phase A is opposite to the polarity of the rotor poles facing the stator poles of phase A. The same is true for the stator poles of the other phases. In other words, it becomes possible to drive with full-pitch windings, and the electromagnetic characteristics and torque characteristics of a configuration in which 12 rotor poles face the seven stator poles via an air gap can be obtained. In addition, the ratio of the rotor pole width to the stator pole width is also related to the shape and characteristics of the permanent magnet PMrbi disposed in the rotor.

[0247] Next, an embodiment of claim 7 is shown in Fig. 47. Fig. 47 is an enlarged view of a part of the stator of Fig. 34 shown at 471, and the circumferential width of the teeth is enlarged. The left and right parts of Fig. 47 are omitted as shown by the wavy lines, and the windings, rotor, etc. are also omitted. The teeth of the three stator poles of Fig. 34 are shown by dashed lines as shown at 474. Lsg is the circumferential width of the air gap surface of the stator pole. 473indicates a shape in which the circumferential width of the teeth on the outer diameter side of the stator pole Ps is expanded. The circumferential width of the teeth is expanded from Lsg to Lsge. Also, as in 475, the tooth shape can be various shapes, such as a tapered shape.

[0248] In the case of the motor configuration of the present invention in which a permanent magnet is not arranged in the stator, such as that shown in FIG. 1, FIG. 9, and FIG. 11(b), a magnetic path for passing the magnetic flux on the rotor side can be sufficiently secured. In contrast, the tooth width in the circumferential direction of the stator pole is narrow, and there is a problem that the magnetic flux passing capacity is low. As shown in FIG. 47, 473, the circumferential width of the tooth of the stator pole Ps is set to be 20% or more larger than the circumferential length Lsg of the magnetic pole facing the air gap part of the stator pole Ps. For example, if the tooth width is expanded by 20% and the magnetic flux density in the air gap part of the stator pole can be increased by 20%, there is a possibility that the torque can be increased by 1.44 times, or 44%, as force is the square of the magnetic flux density, according to formula (19).

[0249] Next, an embodiment of claim 8 is shown in FIG. 48. FIG. 48 shows an enlarged view of the upper right part of the 14S26R motor shown in FIG. 34, which corresponds to the first quadrant. Various examples of shapes including permanent magnets are shown at the tip of each tooth. In the motor of the present invention shown in FIG. 1, FIG. 14, FIG. 34, etc., the stator poles and rotor poles are made of soft magnetic material, so basically, it is necessary to excite the magnetic flux when generating torque. The drive circuit needs to give magnetic energy to the motor when generating torque, and then recover and regenerate that magnetic energy, and since the size is not negligible, the burden is a problem. As a method of reducing the burden of this magnetic energy, permanent magnets such as 483 and 484 can be attached to the stator 481 in FIG. 48 near the air gap surface of the stator pole. The orientation of the permanent magnets 483 and 484 is the direction of the polarity of each stator pole, and is indicated by an arrow. The excitation burden of the motor and the drive circuit can be reduced by the permanent magnets 483 and 484, etc., so that it is possible to reduce the size. Furthermore, reducing the excitation load also leads to a reduction in the time required for excitation, improving the control performance of the motor.

[0250] The thickness of the permanent magnets 483, 484, etc. can be thin and limited to the extent that it assists the excitation of each stator pole. For example, in the case of the magnetic characteristics of a permanent magnet as shown in FIG. 49, the horizontal axis is the magnetic field strength H [A / M] and the vertical axis is the magnetic flux density B [T]. If the residual magnetic flux density of 491 is 1.5 [T], the area of ​​high magnetic flux density of 494 can be used even if the relative permeability is a small value close to 1, by passing a large excitation current through the motor for excitation. In particular, the motor of the present invention assumes a large magnetic flux density exceeding 2.0 [T] near the air gap of each stator pole. Also, even if the permanent magnet is demagnetized by operating at the coercive force point of 492 or in the area of ​​493 due to the control conditions of the motor, it can be easily remagnetized by the motor current and can be used continuously. Also, the shape of the permanent magnets 483, 484 can be reduced to a part of the stator pole tip as in 487. Alternatively, the permanent magnets 482, 485 between the stator poles may be formed into a shape that increases the magnetic flux, such as 486. Also, the permanent magnets 482, 485 between the stator poles may be combined with the permanent magnets 483, 484, 486, 487, etc., and manufactured as a single unit. The configurations and functions of the permanent magnets 483, 484, 486, 487, etc., may also be applied to the rotor poles on the rotor side.

[0251] Next, claim 9 will be explained. Claim 9 is a technology related to a drive circuit for driving a motor with full-pitch windings. Specific examples of the drive circuit have already been shown and explained in Fig. 29 for a three-phase drive circuit and in Fig. 35 for a seven-phase drive circuit. In the case of a three-phase motor, as shown in equations (48), (49), and (50), the full-pitch winding is linked with the magnetic fluxes of all phases, so the voltage becomes complicated. In particular, the voltage of the full-pitch winding of the phase during regeneration becomes the negative value of the power supply voltage, and the full-pitch winding to be driven is induced with the same value as the power supply voltage, resulting in a problem of overvoltage that makes it impossible to increase the current. In the three-phase drive circuit of Fig. 29, equations (54), (55), and (56) are used to connect two windings in series, solving the problem of overvoltage, and a method is shown for passing current through each full-pitch winding of each phase of the three-phase motor shown in Fig. 27.

[0252] In the case of a seven-phase motor, similarly, as shown in equations (66) to (72), the magnetic flux of all phases is linked to the full-pitch winding, making the voltage complex. In the seven-phase drive circuit of Fig. 35, equations (73) to (79) are used to connect two windings in series, solving the problem of overvoltage, and a method is shown for passing current through each full-pitch winding of each phase of the seven-phase motor shown in Fig. 36. In the case of a five-phase motor, two phases are removed compared to the seven-phase motor, and similarly, current can be passed through each full-pitch winding of each phase.

[0253] The seven-phase drive circuit in FIG. 35 shows an example of driving a 28S52R motor in FIG. 34 with two stator pole pairs compared to the 14S26R motor in FIG. 32. Here, the 28S52R motor in FIG. 34 has a large number of rotor poles to rotate at high speed. In addition, the seven-phase drive circuit in FIG. 35 has a large number of transistors and a large number of currents to control. The reason for the large number of transistors is that the number of windings of the full-pitch windings related to the three, five, and seven phases is an odd number, such as 3, 5, and 7, when the number of stator pole pairs is one. That is, if two windings are arranged in series at the top and bottom of the paper as in FIG. 35 and lined up from left to right, both the left and right end windings are on the upper side, and the left and right ends cannot be connected as in FIG. 35 with the 28Q. If the number of stator pole pairs is 2 and the number is 6, 10, or 14, the number becomes an even number. The entire circuit can be configured with a symmetrical structure, such as the three-phase drive circuit in Fig. 29 and the seven-phase drive circuit in Fig. 35. Although the number of elements in the drive circuit increases, each phase can be controlled in a well-balanced manner.

[0254] Next, a method of reducing the number of elements in the seven-phase drive circuit of FIG. 35, that is, a drive circuit, is shown in FIG. 50. A seven-phase motor such as that of FIG. 32 with one stator pole pair can be driven. Alternatively, in a motor with two or more stator pole pairs, windings of the same phase can be connected in series to drive the motor. In the seven-phase drive circuit of FIG. 50, the drive circuit on the right half of the paper of FIG. 35 is deleted. The rest are indicated by the same reference numerals. The cathode of the diode 28H is connected to 504, that is, to the location of 504 by the connection of 503. In FIG. 50, a transistor 501 and a diode 502 are added, and a current of (Id×2) is passed, which is the D-phase current component Id of (Ia+Id) passed through the AD-phase winding 35F and the D-phase current component Id of (Id+Ig) passed through the DG-phase winding 35M.

[0255] In FIG. 50, the problem of an odd number of seven phases is solved by passing two D-phase current components Id through transistor 501. The voltage of AD-phase winding 35F related to this D-phase current component Id is the value of equation (66), and the voltage of DG-phase winding 35M is the value of equation (69), and thus a large voltage is generated. However, since it is directly driven by transistor 501 from DC power supply 29R, there is twice the voltage margin, and D-phase current component Id can be passed. Note that only D-phase current component Id is under different conditions from the current components of the other phases, so care must be taken in controlling it. Note that diodes 28A, 28B, 28C, 28D, 28E, 28F, 28G, and 28H in FIG. 35 and FIG. 50 can be removed in part or in whole depending on the circuit conditions.

[0256] Also, comparing the drive circuits of FIG. 35 and FIG. 50, the seven transistors 351 to 357 in FIG. 50 need to pass twice as much current as the 14 transistors 351 to 35E in FIG. 35, assuming the same motor power. In terms of total current capacity, the two have the same current capacity. However, the drive circuit of FIG. 50 adds transistor 501 and diode 502, so the drive circuit is larger in that respect. That is, the drive circuit of FIG. 50 has fewer elements, but if counted purely in terms of the total current capacity of the transistors, the drive circuit of FIG. 50 is larger than that of FIG. 35. Both have their own characteristics and can be used.

[0257] Next, FIG. 51 shows an example of a drive circuit in which the number of elements is reduced by using the same method as the drive circuit in FIG. 50 for the three-phase drive circuit in FIG. 29. In the three-phase drive circuit in FIG. 51, the right half of the drive circuit in FIG. 29 is deleted. The other parts are indicated by the same reference numerals. The cathode of diode 29M is connected to 294, that is, to location 514 by connection 513. In FIG. 51, transistor 511 and diode 512 are added, and a current of (Ia×2) is passed, which is the A-phase current component Ia of (Ia+Ib) passed through AB-phase winding 297 and the A-phase current component Ia of (Ia+Ic) passed through CA-phase winding 299. Note that diodes 29Q, 29K, 29M, and 29L can be partially or completely removed depending on the circuit conditions.

[0258] Also, comparing the drive circuits of FIG. 29 and FIG. 51, the three transistors 291 to 293 in FIG. 51 need to pass twice as much current as the six transistors 291 to 296 in FIG. 29, assuming the same motor power. The six transistors in FIG. 29 and the three transistors in FIG. 51 have the same total current capacity. However, the drive circuit in FIG. 51 has an additional transistor 511 and diode 512, so the total current capacity is larger in the drive circuit in FIG. 51. That is, the number of elements in the drive circuit in FIG. 51 is smaller, but if counted purely by the total current capacity of the transistors, the drive circuit in FIG. 51 is larger than that in FIG. 29. Both have their own characteristics and can be used. The same configuration can be used for drive circuits of 5 phases, 9 phases, 11 phases, etc.

[0259] Next, claim 10 will be described. Claim 10 relates to a motor in which a component of a current Ifk sufficient to excite a magnetic flux is always applied to each stator winding, and a component of a current It corresponding to a torque is applied by superimposing it on each phase current, and an example of this is shown in FIG. 52. Previously, an example of applying each phase current of FIG. 28 to a motor with a full-pitch winding of 6S10R in FIG. 26 was described. FIG. 28 is an example of applying each phase current Iab, Ibc, and Ica according to the rotor rotation position θr. For example, an example of applying a constant value Ifk [A] indicated by a dashed line to each phase winding as Iab in FIG. 52(a), Ibc in FIG. 52(b), and Ica in FIG. 52(c) will be described. As can be seen from the positional relationship in Figure 26, a magnetomotive force Ifk [A] acts as the sum of the phase currents on the path of the A-phase magnetic flux φa of the A-phase stator south pole 11 and the A-phase stator north pole 14. On the other hand, as shown in Figures 16, 17, 18, and 19, the magnetic characteristics of the rotor are such that magnetic flux can be easily generated by excitation in the magnetic forward direction of the rotor poles, but the rotor magnetic flux generated in the magnetic reverse direction is small. Here, we will simplify the motor model by assuming that rotor magnetic flux is not generated in the magnetic reverse direction.

[0260] In this case, the A-phase magnetic flux φa passes through the position where the A-phase stator south pole 11 faces the rotor north pole via an air gap. As described above, the circumferential width θsg of the stator poles and the circumferential width θrg of the rotor poles in Fig. 26 are set to 30°. The magnetic flux φa starts to be generated when the rotor rotation position θr = 0°, the magnetic flux φa is maximum when θr = 30°, and gradually decreases until the magnetic flux φa becomes 0 when θr = 60°. At this time, the value of the A-phase voltage Vak component is as shown in Fig. 52 (d), and equation (34) can be transformed into the following equation. Vak = Nw × dφa / dt = Nw × dφa / dθr × dθr / dt (99) Vbk=Nw×dφb / dθr×dθr / dt (100) Vck=Nw×dφc / dθr×dθr / dt (101) The increase and decrease in the A-phase magnetic flux φa is due to the rotor rotation, and is not a sudden decrease in magnetic flux due to a sudden decrease in the A-phase current Ia, so it is not a large reverse voltage that reaches the power supply voltage during regeneration. The value of the A-phase voltage Vak at this time is generated in proportion to the rotor rotation speed dθr / dt, like the induced voltage of a surface permanent magnet synchronous motor SPMSM. The magnetic energy stored in the magnetic flux path of the A-phase magnetic flux φa when the rotor rotation position θr = 30° is regenerated to the DC power supply at the product of the negative voltage in Figure 52(d) and the constant value Ifk [A] indicated by the dashed line of Iab in Figure 52(a) between θr = 30° and 60°.

[0261] Similarly, a magnetomotive force of a constant value Ifk [A] shown by a dashed line acts on the path of the B-phase magnetic flux φb of the B-phase stator south pole 13 and the B / phase stator north pole 16. The value of the B-phase voltage Vbk calculated by the formula (100) is shown in FIG. 52(e). A magnetomotive force of a constant value Ifk [A] shown by a dashed line acts on the path of the C-phase magnetic flux φc of the C-phase stator south pole 15 and the C / phase stator north pole 12. The value of the C-phase voltage Vck calculated by the formula (101) is shown in FIG. 52(f). The phase voltages Vab, Vbc, and Vca of the full-pitch winding are expressed by the formulas (51), (52), (53), (54), (55), and (56), and are therefore expressed by the formulas (d), (e), and (f) of FIG.

[0262] When generating torque with the motor of FIG. 26, it is sufficient to subtract the current components for magnetic flux excitation from Iab, Ibc, and Ica in (a), (b), and (c) of FIG. 28 and add them to the current of the dashed line in FIG. 52, and the current values ​​are as shown by the solid lines in FIG. 52 (a), (b), and (c). The voltage of each winding at this time will not change in (d), (e), and (f) of FIG. 52 if the current value of the dashed line in FIG. 52 is sufficiently large, the magnetic characteristics of the soft magnetic material are ideal characteristics shown by the solid line in FIG. 6, and it is assumed that the rotor magnetic flux is not generated in the magnetic reverse direction as described above, and the magnetic flux of the permanent magnet does not change. However, in reality, all of the magnetic characteristics described above are nonlinear and complex characteristics and are not the assumed characteristics, so the voltages in FIG. 52 (d), (e), and (f) are voltages that are a mixture of the voltage components in FIG. 24 (d), (e), and (f). In any case, by constantly passing a current of a constant value Ifk [A] shown by the dashed line in Fig. 52 (a), (b), and (c), the voltage to apply magnetic energy in the motor and the magnitude of the voltage regenerated to the power supply can be reduced. In addition, as an example of a drawback of the voltage accompanying the application and regeneration of magnetic energy, there is a problem of the restriction of the passage of the torque current component, but this problem can be alleviated. Note that the technology of continuously passing a DC excitation current component to each phase winding to excite each stator pole is a method that can be realized because the polarity of the N pole and S pole of each stator pole is fixed and the current passing through each winding is a unidirectional DC current. It is difficult to achieve this in a motor driven by an AC current. The same is true in the case of the concentrated winding winding in Fig. 23 instead of the full-pitch winding in Fig. 26. In addition, the magnitude of the continuously passing current component is variable, and for example, the magnitude of the continuously passing excitation current component can be reduced to reduce the induced voltage during high-speed rotation.

[0263] Alternatively, a field winding for passing a field current component may be added to the stator poles and arranged to pass the field...

Claims

1. Nps number of stator poles Ps arranged in the circumferential direction of the stator; Each slot SLs between each stator pole Ps; A stator winding Ws is disposed in the slot SLs and excites each of the stator poles Ps; a unidirectional drive circuit Dhv capable of driving a current in one direction to each of the stator windings Ws; A plurality of N-pole rotor magnetic poles Prn arranged in the circumferential direction of the rotor; a plurality of S-pole rotor poles Prs arranged alternately with the N-pole rotor poles Prn in the circumferential direction of the rotor; A common back yoke for the rotor, a magnetic path MPrn made of a soft magnetic material that is magnetically connected from the back yoke common to the rotor to each of the N-pole rotor magnetic poles Prn; A magnetic path MPrs made of a soft magnetic material that is magnetically connected from the back yoke common to the rotor to each of the S-pole rotor magnetic poles Prs; a permanent magnet PMrbi disposed between the magnetic paths MPrn and MPrs arranged in the circumferential direction and at a boundary portion in the circumferential direction between the N-pole rotor magnetic pole Prn and the S-pole rotor magnetic pole Prs such that the polarity and magnetic pole orientation of the N-pole rotor magnetic pole Prn and the S-pole rotor magnetic pole Prs are the same; A magnetic flux can smoothly pass from the back yoke common to the rotor to each of the N-pole rotor magnetic poles Prn and each of the S-pole rotor magnetic poles Prs, The sum of the number of the N-pole rotor poles Prn and the number of the S-pole rotor poles Prs, Npr, is greater than the number Nps of the stator poles Ps, Each of the stator windings Ws is driven by passing a unidirectional current generated by the unidirectional drive circuit Dhv or by setting the current value to 0. A motor characterized by:

2. In claim 1, The stator poles Ps are arranged such that N poles and S poles are alternately arranged in the circumferential direction, and N-pole stator poles Psn function as N poles, S-pole stator poles Pss are arranged alternately with the N-pole stator poles Psn in the circumferential direction and act as S poles; A permanent magnet PMsbi is disposed between the N-pole stator pole Psn and the S-pole stator pole Pss arranged in the circumferential direction such that the polarity and magnetic pole orientation of both the stator poles Psn and Pss coincides. Equipped A motor characterized by:

3. In claim 1, The stator winding Ws is a concentrated winding Wscp that excites each of the stator poles Ps. A motor characterized by:

4. In claim 1, The stator winding Ws is a full-pitch winding Wsfp with a winding pitch that is approximately half the pole pair period of the stator. A motor characterized by:

5. In claim 1, Nps number of the stator poles Ps, where Nps=2+4×Ns; Npr = 2 + 4 × Nr, and a total of Npr of the N-pole rotor magnetic poles Prn and the S-pole rotor magnetic poles Prs are provided. A motor characterized by: Here, Ns and Nr are integers of 1 or more.

6. In claim 1, The number of phases of the plurality of stator poles Ps is Nph, The rotor pole pitch of the N-pole rotor poles Prn and the S-pole rotor poles Prs, which are alternately arranged in the circumferential direction of the rotor, is θppr, and Nph stator poles, each having a phase difference with respect to the rotor poles, are partially provided in the circumferential direction of the stator. A motor characterized by: Here, Nph is an integer of 2 or more.

7. In claim 1, The circumferential length of the magnetic pole of the stator pole Ps facing the air gap portion is Lsg, and the circumferential width of the tooth of the stator pole Ps on the outermost radial side is 20% or more larger than the Lsg. A motor characterized by:

8. In claim 1, A permanent magnet PMssur is provided in the vicinity of the air gap between the N-pole stator pole Psn and the S-pole stator pole Pss of the stator pole Ps, and is arranged so that the polarities of the stator poles are aligned. A motor characterized by:

9. In claim 4, The number of the stator poles Ps is Nkb × N1, and among them, the stator poles Ps1, Ps2, Ps3, Ps4, and Ps5 arranged in the circumferential direction are A slot SLs1 located between the stator poles Ps1 and Ps2; A slot SLs2 located between the stator poles Ps2 and Ps3; A slot SLs3 located between the stator poles Ps3 and Ps4; A slot SLs4 located between the stator poles Ps4 and Ps5; A full-pitch winding Wsfp1 is wound between two slots spaced apart by approximately half the pole pair period of the stator, and is disposed in the slot SLs1; Similarly, a full-pitch winding Wsfp2 disposed in the slot SLs2, Similarly, a full-pitch winding Wsfp3 disposed in the slot SLs3, Similarly, a full-pitch winding Wsfp4 disposed in the slot SLs4, A rotor having Nkb×N2 or more rotor poles, which are N poles and S poles arranged alternately on a circumference; A transistor TR1 that is part of the unidirectional drive circuit Dhv and is connected in series with the full-pitch winding Wsfp1; A transistor TR2 that is part of the unidirectional drive circuit Dhv and is connected in series with the full-pitch winding Wsfp2; A transistor TR3 that is part of the unidirectional drive circuit Dhv and is connected in series with the full-pitch winding Wsfp3; a transistor TR4 that is part of the unidirectional drive circuit Dhv and is connected in series with the full-pitch winding Wsfp4; The transistor TR1 supplies a unidirectional current I sfp1 to the full-pitch winding W sfp1, and connects the full-pitch winding W sfp1, the full-pitch winding W sfp2, and the transistor TR2 in series; The transistor TR2 passes a unidirectional current I sfp2 through the full-pitch winding W sfp2, and connects the full-pitch winding W sfp2, the full-pitch winding W sfp3, and the transistor TR3 in series; The transistor TR3 supplies a unidirectional current I sfp3 to the full-pitch winding W sfp3, and connects the full-pitch winding W sfp3, the full-pitch winding W sfp4, and the transistor TR4 in series; The transistor TR4 passes a unidirectional current I sfp4 through the full-pitch winding W sfp4 , The full-pitch windings and transistors TR1, TR2, TR3, and TR4 connected in series supply excitation currents to excite the stator poles Ps1, Ps2, Ps3, Ps4, and Ps5, When the number of full-pitch windings of the motor is three, the full-pitch winding Wsfp1 and the full-pitch winding Wsfp4 are the same winding, and the full-pitch winding Wsfp3 and the full-pitch winding Wsfp1 are arranged in parallel and connected to the transistor TR4 to pass a unidirectional current. A motor characterized by: Here, Nkb is the number of pole pairs of the stator and is an integer of 1 or more, N1 is an integer of 6 or more, and N2 is an integer of 6 or more.

10. In claim 1, A component of the magnetic flux excitation current corresponding to the operating condition is continuously applied to each phase winding of the stator winding Ws, or a magnetic flux excitation winding is additionally wound in each slot of the stator in addition to the stator winding Ws, and the magnetic flux excitation windings of each slot are connected in series to apply the magnetic flux excitation current. A motor characterized by:

11. In claim 1, DC power supply POS2, a DC power supply POS3 arranged in series with the DC power supply POS2; an intermediate potential portion TYV between the DC power source POS2 and the DC power source POS3; a transistor TR7 that is part of the unidirectional drive circuit Dhv connected to the DC power source POS2; a stator winding Ws2 disposed between the transistor TR7 and the intermediate potential portion TYV; a transistor TR8 that is part of the unidirectional drive circuit Dhv connected to the DC power source POS3; a stator winding Ws3 disposed between the transistor TR8 and the intermediate potential portion TYV, Similarly to the stator windings Ws2 and Ws3, the unidirectional current is applied to the stator winding Ws using the DC power supplies POS2 and POS3. A motor characterized by:

12. In claim 1, When exciting a plurality of stator poles Ps adjacent in the circumferential direction with a full-pitch stator winding Wsfp, a current component Isfpv1 is passed through the full-pitch winding Wsfpv1 arranged in the slot Slsv adjacent in the circumferential direction of one stator pole Psv1, A part or all of the current component of the negative value (−Isfpv1) of the current component Isfpv1 is applied to one or more full-pitch windings WsfpvN arranged in slots two or more away from the slot Slsv in the opposite direction to the stator pole Psv1. A motor characterized by:

13. In claim 12, At low speeds, the stator pole PsvN is excited by a current component Isfpv1 and one or more full-pitch windings WsfpvN arranged in slots spaced apart from each other in the circumferential direction are excited by the negative value (-Isfpv1) of the current component Isfpv1. At the time of high speed rotation, the full-pitch windings WsfpvF and WsfpvR on both sides of the stator pole PsvN in the circumferential direction are connected in series, and a current component IsvN that excites the stator pole PsvN is passed through. A motor characterized by:

14. In claim 1, The main magnetic circuit of the rotor is made of soft magnetic material MagA. A soft magnetic material MagB having a higher saturation magnetic flux density than the soft magnetic material MagA is used near the air gap between the N-pole rotor magnetic pole Prn and the S-pole rotor magnetic pole Prs of the rotor. A motor characterized by:

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