Radar signal processing circuit and radar signal processing method

Abstract

To suppress deterioration of radar detection performance caused by circuit errors.SOLUTION: A radar signal processing circuit includes: a transmission phase rotation unit that imparts phase rotation based on a plurality of different phase rotation amounts controlled by a series of a plurality of mutually different phase variable patterns to a transmission signal for each transmission period; a radar transmission radio unit that outputs a plurality of different radar transmission signals to which multiplexing processing has been applied to a plurality of transmission antennas; and a radar reception radio unit that inputs one or more reflected wave signals resulting from at least one of the plurality of different radar transmission signals being reflected by a target via a plurality of reception antennas. The plurality of different phase rotation amounts are a series in which a shift amount that is an integer multiple of two or greater of 2π is sequentially added for each of a plurality of transmission periods for each transmission period, and the plurality of mutually different phase variable patterns are patterns that have elements of each of the plurality of transmission periods and vary the amount of phase rotation for each of the plurality of transmission periods.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] This disclosure relates to a radar signal processing circuit and a radar signal processing method. [Background technology]

[0002] In recent years, research has been progressing on radar systems that use short-wavelength radar transmission signals, including microwaves or millimeter waves, which can achieve high resolution. Furthermore, in order to improve safety outdoors, there is a need for the development of radar systems (wide-angle radar systems) that can detect objects (targets), including pedestrians, in addition to vehicles, over a wide-angle range.

[0003] For example, pulse compression radar systems are known as radar devices. When detecting vehicles / pedestrians in a pulse compression radar system, the reflected waves from pedestrians are weaker than the reflected waves from vehicles. Therefore, the radar transmitter requires a transmission configuration that transmits pulse compression waves with low range sidelobes, and the radar receiver requires a reception configuration that has a wide reception dynamic range.

[0004] To obtain low-range sidelobe characteristics, it is known that pulse compression codes such as Barker codes, PN sequence codes, or complementary codes can be used. Below, as an example, the case of using complementary codes will be described. Complementary codes consist of codes that form a pair (hereinafter referred to as complementary code a n , b n Let n = 1, ..., L, where L is the code length. The autocorrelation operation of each of the two codes is expressed by equations (1) and (2).

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[0005] In equations (1) and (2), for n > L and n < 1, a n =0, b nIt is equal to 0. The complementary code has the property that the correlation values other than τ = 0 become zero and the range side lobe becomes zero as shown in the following equation (3) by adding the autocorrelation operation results of each of the two codes with the shift time τ made the same.

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[0006] Such complementary codes a n , b n are known for a pulse compression radar that time-division transmits them every predetermined radar transmission period.

[0007] Regarding the method for generating the complementary code, it is disclosed in Non-Patent Document 1. According to Non-Patent Document 1, for example, based on the codes A = [a 1, a2] =

[0011] , B = [b 1, b2] = [1 -1] having complementarity consisting of elements '1' or '-1', complementary codes with code lengths L = 4, 8, 16, 32,..., 2 P can be sequentially generated. The pulse compression radar can increase the received signal level of the radar reflected wave by repeatedly transmitting the above-described pulse compression code a predetermined number of times for each radar transmission period.

Prior Art Documents

Patent Documents

[0008]

Patent Document 1

Patent Document 2

Non-Patent Documents

[0009]

Non-Patent Document 1

[0010] When attempting to implement the aforementioned pulse-compressed radar using radio frequency bands (RF) such as millimeter waves, circuit errors are introduced into the RF circuit and analog baseband circuit in the radar transmitter or radar receiver of the radar device. As a result, the radar detection performance (or radar ranging performance) deteriorates compared to the ideal characteristics when there are no circuit errors.

[0011] One aspect of this disclosure provides a radar signal processing circuit and a radar signal processing method that can suppress the degradation of radar detection performance due to circuit errors. [Means for solving the problem]

[0012] A radar signal processing circuit according to one aspect of the present disclosure comprises: a signal generation unit that generates a transmission signal; a transmission phase rotation unit that applies a phase rotation to the transmission signal at each transmission period based on a plurality of different phase rotation amounts controlled by a sequence of a plurality of mutually different phase variable patterns; a multiplexing unit that applies multiplexing to the plurality of different radar transmission signals to which the phase rotation has been applied; a radar transmission radio unit that outputs the plurality of different radar transmission signals to which the multiplexing has been applied to a plurality of transmitting antennas; and a radar receiving radio unit that receives one or more reflected wave signals in which at least one of the plurality of different radar transmission signals has been reflected at a target via a plurality of receiving antennas, wherein the plurality of different phase rotation amounts is a sequence obtained by sequentially adding shift amounts that are integer multiples of 2 or more of 2π at each transmission period, and the plurality of mutually different phase variable patterns have elements for each of the plurality of transmission periods and are patterns that vary the phase rotation amount at each of the plurality of transmission periods.

[0013] A radar signal processing method according to one aspect of the present disclosure involves generating a transmission signal, applying a phase rotation to the transmission signal at each transmission period based on a plurality of different phase rotation amounts controlled by a sequence of a plurality of mutually different phase variable patterns, applying multiplexing to the plurality of different radar transmission signals to which the phase rotation has been applied, outputting the plurality of different radar transmission signals to which the multiplexing has been applied to a plurality of transmitting antennas, and inputting one or more reflected wave signals in which at least one of the plurality of different radar transmission signals has been reflected at a target via a plurality of receiving antennas, wherein the plurality of different phase rotation amounts is a sequence obtained by sequentially adding shift amounts that are integer multiples of 2 or more of 2π at each transmission period, and the plurality of mutually different phase variable patterns have elements for each of the plurality of transmission periods and are patterns that vary the phase rotation amount at each of the plurality of transmission periods.

[0014] These comprehensive or specific embodiments may be implemented as a system, method, integrated circuit, computer program, or recording medium, or as any combination of a system, device, method, integrated circuit, computer program, and recording medium. [Effects of the Invention]

[0015] According to one aspect of this disclosure, it is possible to suppress the deterioration of radar detection performance due to circuit errors.

[0016] Further advantages and effects of one aspect of this disclosure will be made apparent from the specification and drawings. Such advantages and / or effects are provided by several embodiments and features described in the specification and drawings, but not all of them are necessarily provided in order to obtain one or more identical features. [Brief explanation of the drawing]

[0017] [Figure 1] Block diagram showing an example configuration of a radar device according to Embodiment 1 of this disclosure. [Figure 2] A diagram showing an example of a radar transmission signal according to Embodiment 1 of this disclosure. [Figure 3] Block diagram showing other configurations of the radar transmission signal generation unit according to Embodiment 1 of this disclosure. [Figure 4A] This figure shows an example of the computer simulation results of the output of the Doppler analysis unit when phase inversion is not performed. [Figure 4B] This figure shows an example of the computer simulation results of the output of the Doppler analysis unit according to Embodiment 1 of this disclosure. [Figure 5] This figure shows another example of the configuration of the radar device according to Embodiment 1 of this disclosure. [Figure 6] Block diagram showing an example configuration of a radar device according to Embodiment 2 of this disclosure. [Figure 7] This figure shows an example of a computer simulation result according to Embodiment 2 of this disclosure. [Figure 8]Block diagram showing an example configuration of a radar device according to Embodiment 3 of this disclosure. [Modes for carrying out the invention]

[0018] In radar systems, circuit errors such as IQ mismatch and DC offset occur in the quadrature modulation / demodulation circuits, phase noise occurs in the frequency conversion section, and quantization noise occurs in the AD converter and DA converter.

[0019] When a DC offset is included as a circuit error, the noise level increases compared to when there is no error. Therefore, in a radar system, if the reflected wave from the target is lower than the noise level, the target will not be detected, leading to a decrease in the detection rate and a deterioration in radar detection performance.

[0020] Conventionally, methods have been proposed to remove DC offset by the circuit configuration of the radar transmitter or radar receiver. Specifically, Patent Document 1 discloses a configuration in which DC offset is removed by placing a high-pass filter on discrete data output from an AD converter. Patent Document 2 discloses a configuration in which DC offset is removed by placing a band-pass filter in the input stage of an AD converter. Non-Patent Document 2 discloses a circuit configuration for correcting IQ imbalance circuit errors.

[0021] However, conventional technologies require the inclusion of a DC offset removal circuit or an IQ imbalance circuit error correction circuit, which complicates the circuit configuration. Furthermore, when a high-pass filter or band-pass filter is used to remove the DC offset, it may weaken desired radar reflected wave components other than the DC offset component, or amplitude distortion or phase distortion may occur due to the filter response, degrading radar detection performance.

[0022] Furthermore, even with a configuration that includes a DC offset removal circuit, if the DC offset component or IQ imbalance component cannot be completely removed and circuit error components remain, the circuit error components will also be accumulated by the coherent integration process in radar reception processing, resulting in a deterioration of radar detection performance. For example, even if only a small amount of residual circuit error is included, the accumulation effect will increase the residual component to about 30-40 dB, requiring a highly accurate error detection mechanism, which complicates the hardware configuration of the radar device.

[0023] On the other hand, Non-Patent Document 3 discloses a radar device that transmits a code combining phase modulation between multiple pulse transmission periods and cancels the received DC offset component by coherent integration processing of the reflected wave.

[0024] As an example, the code A = [a1, a2, … , a] used for pulse compression on the transmitting side of a radar device. L ] and the sign -A = [-a1, -a2, … , -a L The principle is shown below that by using ] and transmitting codes A and -A in two transmission periods, performing correlation processing on the receiving side, and performing coherent integration, the received DC offset component is canceled.

[0025] In the following section, we will explain the case where there are no noise components and the sum of each element of code A is not zero, as shown in equation (4).

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[0026] <Example 1: When the received signal of a radar reflected wave contains a received DC offset component> (1-1) When transmitting code A, the received DC offset component α Rx The received signal of code A that includes (γA+α Rx The autocorrelation value obtained by the autocorrelation operation between ( ) and sign A is expressed by the following equation (5).

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[0027] In equation (5), γ represents the complex received response of the radar reflected wave, and the asterisk (*) is the complex conjugate operator.

[0028] (1-2) When transmitting code A, the received DC offset component α Rx The received signal containing the code -A (-γA+α Rx The autocorrelation value obtained by the autocorrelation operation between ) and the sign -A is expressed by the following equation (6).

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[0029] At the receiving end, when the autocorrelation value obtained solely from (1-1) above (equation (5)) is subjected to coherent integration, the received DC offset component α is obtained even if γ is zero (i.e., even if no radar reflection exists). Rx Two items, including this one, are added together. As a result, the noise level (floor level) increases across the entire distance range, degrading radar detection performance.

[0030] On the other hand, when the autocorrelation values ​​obtained in (1-1) and (1-2) above (equations (5) and (6)) are subjected to coherent integration on the receiving side, the received DC offset component α is obtained as shown in equation (7). Rx It can be canceled.

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[0031] This prevents an increase in noise levels and suppresses the degradation of the radar detection performance of the radar equipment. Similarly, for received signals of radar reflected waves that include Doppler fluctuations, the received DC offset component α is also suppressed. Rx This allows for cancellation of noise, preventing an increase in noise levels (floor level) and minimizing degradation of radar detection performance.

[0032] <Example 2: The received signal of a radar reflected wave that does not contain Doppler fluctuations has a transmitted DC offset component α Tx If it includes >

[0033] (2-1) When transmitting code A, the transmitted DC offset component α Tx The received signal (γA+α) contains Tx The autocorrelation value obtained by the autocorrelation operation between ) and sign A is expressed by the following equation (8).

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[0034] (2-2) When transmitting code A, the transmitted DC offset component α Tx The received signal containing the code -A (-γA+α Tx The autocorrelation value obtained by the autocorrelation operation between ) and the sign -A is expressed by the following equation (9).

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[0035] At the receiving end, when the autocorrelation values ​​obtained in (2-1) and (2-2) above (equations (8) and (9)) are subjected to coherent integration, the received DC offset component α shown in <Example 1> is obtained. Rx Similar to the case where the transmitted DC offset component α is included, Tx This cancels out noise, preventing an increase in the noise level (floor level) and minimizing the degradation of radar detection performance.

[0036] As explained in Example 1 and Example 2, the degradation of radar detection performance can be suppressed by canceling the transmitted DC offset component or the received DC offset component.

[0037] However, even with the above method, if a transmitted DC offset component (including carrier leakage components) exists in the radar transmitter and there is Doppler variation in the radar reflected wave, the transmitted DC offset component will remain. This is because the transmitted DC offset component is affected by Doppler variation, causing a cancellation error in the above method. As a result, the noise level of the Doppler component contained in the remaining transmitted DC offset component increases, leading to a problem of degraded radar detection performance.

[0038] The following explains the case where Doppler fluctuations are included in the radar reflection.

[0039] <Example 3: The received signal of a radar reflected wave containing Doppler fluctuations includes a transmitted DC offset component α Tx If it includes >

[0040] Here, the Doppler variation contained in the radar reflected wave is expressed as "exp(j2πf d ×T r )=exp(jΨ d )" (f d : Doppler frequency, T r : Radar transmission period. (However, this is under the condition that Doppler variation within the code can be considered constant).

[0041] (3-1) When transmitting code A, the transmitted DC offset component α Tx The received signal γ(A+α) contains Tx The autocorrelation value obtained by the autocorrelation operation between ) and sign A is expressed by the following equation (10).

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[0042] (3-2) When transmitting code-A, the transmitted DC offset component α Tx The received signal γ(-A+α) contains the code -A. Tx )exp(jΨ d The autocorrelation value obtained by the autocorrelation operation between ) and sign -A is expressed by the following equation (11).

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[0043] At the receiving end, when the autocorrelation values ​​obtained in (3-1) and (3-2) above (equations (10) and (11)) are subjected to coherent integration, the transmitted DC offset component α depends on γ, as shown in equation (12). Tx The two items that include this will be added together.

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[0044] Generally, when receiving radar reflected waves that contain Doppler fluctuations, coherent integration processing using Doppler frequency analysis is applied. This results in an increase in the noise level (floor level) of the Doppler frequency component corresponding to the second term of equation (12), degrading radar detection performance. In addition, the transmitted DC offset component α Tx The received power is |γα Tx | 2 Because it is proportional to the received power of the radar reflected wave, the greater the increase in noise level (floor level), and the greater the deterioration of radar detection performance.

[0045] Next, we will explain a method for preventing an increase in the noise level (floor level) and suppressing the deterioration of radar detection performance in a pulse-compressed radar, even when it includes transmitted DC offset (carrier leak) and received DC offset, without adding a high-precision correction circuit for circuit error correction.

[0046] In the method of transmitting pulse-compressed codes with phase modulation applied over multiple pulse transmission periods as described above, if a transmitted DC offset component (including a carrier leakage component) exists in the radar transmitter, and there is a Doppler fluctuation in the radar reflected wave, the transmitted DC offset component remains, and the noise level of a specific Doppler component increases. To address this problem, the transmitted DC offset component can be reduced by the following method.

[0047] Specifically, the radar device uses the code A=[a1, a2, … , a] for pulse compression on the transmitting side. L ] and the phase-inverted signs -A=[-a1, -a2, … , -a L This method involves transmitting codes A, -A, -A, and A in four radar transmission cycles using [ ], performing correlation processing on the receiving end, and then performing coherent integration processing.

[0048] The following will explain this method in detail. Note that, as above, the following explanation will cover the case where there are no noise components and the sum of the elements of code A is not zero (see equation (4)).

[0049] <Example 4: The received signal of a radar reflected wave containing Doppler fluctuations includes a transmitted DC offset component α Tx If it includes >

[0050] Here, the Doppler variation contained in the radar reflected wave is expressed as "exp(j2πf d ×T r )=exp(jΨ d )" (f d : Doppler frequency, T r : Radar transmission period. (However, this is under the condition that Doppler variation within the code can be considered constant).

[0051] (4-1) When transmitting code A, the transmitted DC offset component α Tx The received signal γ(A+α) contains Tx The autocorrelation value obtained by the autocorrelation operation between ) and sign A is expressed by the following equation (13).

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[0052] (4-2) When transmitting code-A, the transmitted DC offset component α Tx The received signal γ(-A+α) contains the code -A. Tx ) exp(jΨ d The autocorrelation value obtained by the autocorrelation operation between ) and sign -A is expressed by the following equation (14).

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[0053] (4-3) When transmitting code-A, the transmitted DC offset component α Tx The received signal γ(-A+α) contains Tx ) exp(j2Ψ d The autocorrelation value obtained by the autocorrelation operation between ) and sign -A is expressed by the following equation (15).

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[0054] (4-4) When transmitting code A, the transmitted DC offset component α Tx The received signal γ(A+α) contains code A. Tx ) exp(j3Ψ d The autocorrelation value obtained by the autocorrelation operation between ) and sign A is expressed by the following equation (16).

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[0055] At the receiving end, when the autocorrelation values ​​obtained in (4-1) to (4-4) above (equations (13) to (16)) are subjected to coherent integration, the transmitted DC offset component α depends on γ, as shown in equation (17). Tx The two items that include this will be added together.

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[0056] In equation (17), the Doppler phase variation Ψ d If it is in the range less than π / 6, then |1-exp(j2Ψ d Since )|<1”, the residual component of the transmitted DC offset can be reduced compared to the case shown in <Example 3> (see equation (12)). However, even in <Example 4>, the residual component of the transmitted DC offset cannot be completely canceled.

[0057] As described above, coherent integration processing using Doppler frequency analysis is applied to the reception processing of radar reflected waves that contain Doppler fluctuations. As a result, the noise level (floor level) of the Doppler frequency component, which includes the residual transmitted DC offset component (the second term of equation (17)), increases, and the radar detection performance deteriorates. For example, as explained in Example 4, the transmitting side transmits the codes A, -A, -A, A every four radar transmission cycles, and the receiving side performs correlation reception processing on the radar reflected wave using the transmitted codes, and performs Doppler frequency analysis on the output obtained by coherent integration processing every two radar transmission cycles, with the transmitted code (A, -A) that cancels the received DC offset as one unit. In this case, the noise level (floor level) of the Doppler frequency component included in the residual transmitted DC offset component increases, and the radar detection performance deteriorates.

[0058] In Doppler frequency analysis, the reason why the noise level (floor level) of a specific frequency component increases due to the residual transmitted DC offset component is that the output after coherent integration becomes a fixed phase fluctuation.

[0059] Therefore, in one aspect of this disclosure, in order to prevent the output processed by coherent integration at the receiving end from having a fixed phase fluctuation, a random phase fluctuation is applied to the coherent integrated output, with a transmission code (A, -A) that cancels the received DC offset as one unit.

[0060] Specifically, in one aspect of the present disclosure, when the receiving side performs coherent integration processing using Doppler frequency analysis, multiple transmission codes that cancel the received DC offset are treated as one unit (corresponding to A and -A transmitted in two radar transmission cycles in the above Examples 1 to 4), and the system randomly switches whether or not to add a phase inversion to each unit of transmission code (a code with a phase change of π) so that the Doppler phase fluctuation does not become a steady phase shift amount (phase change amount).

[0061] This results in variations in the phase fluctuation of the output obtained by coherent integration of the radar reflected wave for each of the above-mentioned units of transmitted code, allowing the residual transmitted DC offset component to be whitened in the Doppler frequency domain. This prevents an increase in the noise level (floor level) of specific Doppler frequency components, thereby suppressing the degradation of radar detection performance.

[0062] Hereinafter, an embodiment relating to one aspect of this disclosure will be described in detail with reference to the drawings. In the embodiment, the same reference numerals are used for the same components, and their descriptions will be omitted as they would be redundant.

[0063] [Embodiment 1] [Radar equipment configuration] Figure 1 is a block diagram showing the configuration of the radar device 10 according to this embodiment.

[0064] The radar device 10 includes a radar transmitting unit 100 and a radar receiving unit 200.

[0065] The radar transmitter 100 generates a high-frequency (radio frequency) radar signal (radar transmission signal). The radar transmitter 100 then transmits the radar transmission signal at a predetermined transmission cycle.

[0066] The radar receiver 200 receives a reflected wave signal, which is a radar transmission signal reflected from the measurement target. The radar receiver 200 performs processing synchronized with the radar transmitter, for example, using a reference signal (not shown). The radar receiver 200 may also perform signal processing on the received reflected wave signal to perform processing such as detecting the presence or absence of a target and estimating its direction. The measurement target is an object that the radar device 10 detects, and includes, for example, vehicles (including four-wheeled and two-wheeled vehicles) or people.

[0067] [Configuration of radar transmitter 100] The radar transmission unit 100 includes a radar transmission signal generation unit 101, a transmitting radio unit 106, and a transmitting antenna 107.

[0068] The radar transmission signal generation unit 101 generates the radar transmission period (T r For each step, a baseband radar transmission signal (pulse-compressed signal) is generated by modulating a code of code length L.

[0069] The radar transmission signal generation unit 101 operates based on a transmission reference clock obtained by multiplying a reference signal (not shown) by a predetermined number. Hereinafter, the transmission reference clock frequency is set to f TxBB Let's assume that the radar transmission period (T r ) is the transmission reference clock frequency (f) obtained by multiplying the reference signal by a predetermined factor. TxBB The discrete time interval (1 / f) is determined by ) TxBB The integer N of ) r times(N r ×(1 / f TxBB ))

[0070] The radar transmission signal generation unit 101 includes a code generation unit 102, a phase rotation control unit 103, a transmission phase rotation unit 104, and a modulation unit 105.

[0071] Specifically, the code generation unit 102 generates the radar transmission period (T r Each time, a transmission code with code length L is generated. Specifically, the code generation unit 102 generates a transmission code Code(m) with code length L in the mth radar transmission cycle.

[0072] Below, each element of the transmission code Code(m) is C n It is denoted as (m). That is, the transmission code Code(m) has L elements { C1(m), C2(m), ..., C L It consists of (m)}. Also, the transmission code element C n (m) consists of binary values ​​such as {-1,1} or tetrametric values ​​such as {1, -1, -j, j}, where j is the imaginary unit. Also, n = 1, 2, ..., L and m = 1, 2, ..., Q. Here, Q represents the number of radar transmission cycles used by the radar device 10 when measuring distance, Doppler, and direction of arrival.

[0073] As the transmission code, for example, it is preferable to apply a Barker code, a complementary code, an M-sequence code, a Gold code, etc., which can obtain low-range side-lobe characteristics. Also, the transmission code in each radar transmission cycle may be the same code or different codes. Alternatively, a plurality of transmission codes may be switched in each radar transmission cycle.

[0074] The phase rotation control unit 103 has a plurality of N e radar transmission cycles (= N e ×T r ) as a unit, and within a plurality of (N e ) radar transmission cycles, it outputs a phase rotation amount signal for applying a phase rotation that is an integer multiple of 2π (2πN s ) to the transmission code to the transmission phase rotation unit 104 and the radar reception unit 200 (reception phase rotation unit 206). Here, N e is an integer greater than 1, and N s is an integer of 1 or more.

[0075] The phase rotation control unit 103 outputs a phase rotation amount signal that has a constant phase shift amount (phase change amount) over each radar transmission cycle within a period of a plurality of (N e ) radar transmission cycles (= N e ×T r ). For example, the phase rotation control unit 103 outputs a phase rotation amount signal that periodically applies phase rotation amounts of φ×0, φ×1, φ×2, …, φ(N r -1) for each radar transmission cycle (T e ). Here, φ = 2πN s / N e . For example, when N e = 4 and N s = 1, the phase shift amount φ for each radar transmission cycle (T r ) is π / 2. In this case, the phase rotation amounts output in each of the 4 (= N e ) radar transmission cycles (T r ) are, for example, 0, π / 2, π, 3π / 2.

[0076] When using complementary codes as transmission codes (including cases where multiple complementary codes are combined for transmission, such as in Spano codes), the phase shift amount for pairs of codes constituting the complementary codes is set to zero. That is, the same phase rotation is applied to pairs of codes constituting the complementary codes. Thereby, an effect of maintaining the high side lobe suppression characteristics of the complementary codes can be obtained.

[0077] For example, when using complementary codes, the phase rotation control unit 103, for even N e cycles of the radar transmission period (T r ), outputs a phase rotation amount signal that periodically applies phase rotation amounts of φ×0, φ×0, φ×1, φ×1, φ×2, φ×2, …, φ(N e -1), φ(N e -1). Here, φ = 2πN s / (N e / 2)= 4πN s / N e . For example, when N e = 8 and N s = 1, the phase shift amount φ is π / 2. In this case, the phase rotation amounts output in each of the 8(=N e ) cycles of the radar transmission period (T r ) are, for example, 0, 0, π / 2, π / 2, π, π, 3π / 2, 3π / 2.

[0078] Also, when the phase shift amount φ is π / 2, the phase rotation can be realized by swapping the I signal component and the Q signal component (accompanied by sign conversion of positive and negative), so a multiplier can be made unnecessary in the transmission phase rotation unit 104.

[0079] Also, by setting N e / N s > 2 (in the case of complementary codes, N e / N s > 4), an effect can be obtained that the DC offset component can also be canceled even when it has I signal and Q components.

[0080] Furthermore, the phase rotation control unit 103, for a plurality of N e cycles of the radar transmission period (= Ne ×T r The phase rotation amount variable signal is output to the transmitting phase rotation unit 104 and the radar receiving unit 200 (receiving phase rotation unit 206) to control the amount of phase rotation according to a predetermined phase variable pattern, using ) as the unit. In other words, the phase rotation control unit 103 controls multiple N e The period corresponding to one radar transmission cycle (=N) e ×T r The pattern of the phase rotation amount for the radar transmission signal within the period (N) e ×T r Change it every time.

[0081] Here, a random pattern that varies the phase rotation may be used as the phase-variable pattern. For example, the phase rotation control unit 103 uses a pseudo-random code (PN code), an M-sequence code, or a Gold code as the phase-variable pattern and controls the amount of phase rotation according to the code polarity of each code in the phase-variable pattern.

[0082] As an example, the phase-variable pattern is N PP It consists of n elements, each element having the binary values ​​{-1,1}. Below, each element of the phase-variable pattern will be denoted as PP(q), where q = 1, 2, ..., N PP The phase rotation control unit 103 reads out each element PP(q) of the phase variable pattern in order, N e The radar transmission cycle (=N) e ×T r The same element's value is repeatedly output over the mth radar transmission period. That is, the variable phase rotation signal PC(m) output from the phase rotation control unit 103 in the mth radar transmission period is expressed by the following equation (18).

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[0083] Here, m=1,…,N e ×N d That is the case. Furthermore, N d This parameter is defined in the Doppler analysis unit 208, which will be described later. The number of elements in the phase-variable pattern is N. PPHowever, the number of radar transmission cycles Q(=N) e ×N d If the number is less than Q, the phase rotation control unit 103 cycles through the radar transmission period and reads out the phase variable pattern.

[0084] In other words, the phase-variable pattern is multiple N e The radar transmission cycle (T) r Each time this happens, the element PP(q) is set to change randomly.

[0085] The transmitting phase rotation unit 104 applies a phase rotation to the transmitted code output from the code generation unit 102 based on the phase rotation amount signal and the variable phase rotation amount signal instructed by the phase rotation control unit 103. The transmitting phase rotation unit 104 outputs the transmitted code with the applied phase rotation to the modulation unit 105. For example, as shown in equation (19) below, the transmitting phase rotation unit 104 outputs a signal GP(m) obtained by applying a phase rotation to the transmitted code Code(m) output from the code generation unit 102 during the mth radar transmission cycle.

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[0086] In this manner, the transmission phase rotation unit 104 applies a phase rotation to the transmission code (radar transmission signal) according to the phase variable pattern (PC(m)).

[0087] As mentioned above, when the transmitted code is a complementary code, by not applying a phase shift (applying the same phase rotation) between the codes constituting the complementary code pair, the effect of maintaining the characteristic of canceling distance sidelobes between the codes constituting the complementary code pair (high sidelobe suppression characteristic) can be obtained. That is, as shown in equation (20) below, when the transmitted code is a complementary code, the transmitting phase rotation unit 104 outputs a signal GP(m) obtained by applying a phase rotation to the transmitted code Code(m) such that the phase shift within the two transmission cycles in which the codes constituting the complementary code pair are transmitted becomes zero.

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[0088] The modulation unit 105 performs pulse modulation (amplitude modulation (ASK: Amplitude Shift Keying) or phase modulation (PSK: Phase Shift Keying)) on the transmission code output from the transmission phase rotation unit 104, and outputs the modulated signal (radar transmission signal) to the transmission radio unit 106.

[0089] For example, when the modulation unit 105 uses phase modulation (PSK), phase modulation with a binary transmission code such as {-1, 1} becomes BPSK, and phase modulation with a quaternary transmission code such as {1,-1, -j, j} becomes QPSK or 4-phase PSK, and a predetermined modulation symbol is assigned on the IQ phase plane.

[0090] Furthermore, the modulation unit 105 outputs a modulated signal that is limited within a predetermined bandwidth by passing the modulated signal, which is obtained by modulating the transmission code, through a bandwidth-limiting filter (not shown).

[0091] Here, the in-phase component of the modulated signal is defined as I(n s ) is expressed as Q(n s If expressed as ), the modulated signal G(n s ) can be expressed as shown in equation (21).

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[0092] Here, n s is a natural number and represents discrete time. Also, the discrete time interval is (1 / f TxBB ) and f TxBB This is the transmission reference clock frequency obtained by multiplying the reference signal by a predetermined amount.

[0093] Furthermore, the modulation unit 105 modulates the transmitted code output from the transmitted phase rotation unit 104 using No samples of the transmitted reference clock for each code. As a result, for a transmitted code of code length L, the radar signal section Tw contains Nw = No × L samples. Also, the radar transmission period (T r ) No signal section (T r -Tw) is the number of Nu (=N) transmission reference clocks. r The sample is comprised of -Nw (see, for example, Figure 2). Therefore, the modulated signal in the mth radar transmission period can be expressed as shown in equation (22).

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[0094] The transmitting radio unit 106 quadrature modulates the signal output from the modulation unit 105, performs frequency conversion to generate a radar transmission signal in the carrier frequency (Radio Frequency: RF) band, amplifies it to a predetermined transmission power using a transmitting amplifier, and outputs it to the transmitting antenna 107. The transmitting antenna 107 radiates the radar transmission signal output from the transmitting radio unit 106 into space.

[0095] A common reference signal is applied to the local oscillators of the transmitting radio unit 106 and the receiving radio unit 202, which will be described later. This allows for synchronization between the local oscillators of the transmitting radio unit 106 and the receiving radio unit 202.

[0096] Furthermore, the radar transmission unit 100 may include a radar transmission signal generation unit 101a, as shown in Figure 3, instead of the radar transmission signal generation unit 101. The radar transmission signal generation unit 101a does not have the code generation unit 102, phase rotation control unit 103, transmission phase rotation unit 104, and modulation unit 105 shown in Figure 1, but instead includes a code storage unit 111 and a DA conversion unit 112. The code storage unit 111 stores the code sequences generated by the code generation unit 102 (Figure 1) in advance and reads out the stored code sequences sequentially in a cyclical manner. The DA conversion unit 112 converts the code sequences (digital signals) output from the code storage unit 111 into analog signals.

[0097] [Configuration of radar receiver 200] In Figure 1, the radar receiver 200 includes a receiving antenna 201, a receiving radio unit 202, and a signal processing unit 203.

[0098] The receiving antenna 201 receives the RF band radar transmission signal transmitted from the radar transmission unit 100, which has been reflected by a reflective object including the measurement target (reflected wave signal), and outputs the received reflected wave signal as a received signal to the receiving radio unit 202.

[0099] The receiving radio unit 202 amplifies the received signal output from the receiving antenna 201 to a predetermined level, frequency-converts the high-frequency band received signal to the baseband band, and converts the baseband band received signal into a baseband band received signal that includes the I signal (In-Phase signal component) and the Q signal (Quadrature signal component).

[0100] The signal processing unit 203 includes an AD conversion unit 204, a correlation calculation unit 205, a reception phase rotation unit 206, a coherent integration unit 207, and a Doppler analysis unit 208.

[0101] Furthermore, each part within the signal processing unit 203 operates based on a received reference clock obtained by multiplying a reference signal (not shown) by a predetermined factor. In the following, the received reference clock frequency is set to f RxBB Let's assume that the radar transmission period (T r ) is the received reference clock frequency (f) obtained by multiplying the reference signal by a predetermined factor. RxBB The discrete time interval (1 / f) is determined by ) RxBB The integer N of ) v times(N v ×(1 / f RxBB )) Also, in the following, the transmission reference clock frequency f TxBB The received reference clock frequency f RxBB N is an integer multiple of TR Relationship f TxBB =f RxBB ×N TR It is assumed to be located in [location].

[0102] The AD conversion unit 204 applies the received reference clock frequency f to the baseband signal, which includes the I signal and Q signal output from the receiving wireless unit 202. RxBB Based on discrete time (1 / f RxBB By sampling at ), the I and Q signals are converted into digital data.

[0103] In the following explanation, the baseband received signal containing the I and Q signals at discrete time k is a complex number signal x(k) = I r (k) + j Q r (k) is denoted as (k). In the following, the discrete time k is the mth radar transmission period (T r Using the start timing of (k=1) as a reference, the signal processing unit 203 determines the radar transmission period T r Measurements are taken periodically until the process is completed. That is, k=1,…,N v This is the result. Here, j is the imaginary unit.

[0104] Therefore, the output signal of the AD converter 204 in the mth radar transmission cycle can be expressed as shown in equation (23). Hereafter, X(k) will be referred to as the complex baseband signal.

number

[0105] The correlation calculation unit 205 calculates the radar transmission period T r Each time, the complex baseband signal X(N) is output from the AD conversion unit 204. v (m-1)+k) and the transmission code C transmitted in the radar transmission unit 100 n A correlation operation is performed with (m). Here, n=1,...,L. For example, the correlation value AC(k, m) of the sliding correlation operation of discrete time k in the m-th radar transmission period is calculated based on the following equation (24).

number

[0106] In equation (24), the asterisk (*) represents the complex conjugate operator. Also, k=1,…,N v That is the case.

[0107] The correlation calculation unit 205 is configured for k=1,…,N v The correlation calculation is not limited to the case where the correlation calculation is performed on k; the measurement range (i.e., the range of k) may be limited according to the range of the target being measured by the radar device 10. This makes it possible to reduce the amount of computation processing performed by the correlation calculation unit 205 in the radar device 10. For example, the correlation calculation unit 205 calculates k = Nw / N TR +1, ..., (Nu - Nw) / N TR The measurement range may be limited to a specific range. In this case, the radar device 10 does not perform measurements in the time interval corresponding to the code transmission section Tw.

[0108] As a result, even when the radar transmission signal directly feeds back into the radar receiver 200, the radar device 10 does not perform processing by the correlation calculation unit 205 during the period when the radar transmission signal feeds back, thus enabling measurement that eliminates the effects of feedback. Furthermore, when limiting the measurement range (range of k), the same method of limiting the measurement range (range of k) can be applied to the processing of the receiver phase rotation unit 206, coherent integration unit 207, and Doppler analysis unit 208, which will be described below. This reduces the processing load in each component and lowers the power consumption of the radar receiver 200.

[0109] The receiving phase rotation unit 206 controls the radar transmission period (T r ) For each instance, the phase rotation applied by the transmission phase rotation unit 104 of the radar transmission unit 100 is in the opposite direction (cancellation direction) to the phase rotation PC(m)exp[-j{(m-1)modN e The φ] is added to the signal (correlation calculation value) output from the correlation calculation unit 205. That is, the mth radar transmission period (T r In this configuration, the receiving phase rotation unit 206 outputs a signal ACP(k, m) obtained by applying a phase rotation to the output AC(k, m) of the correlation calculation unit 205, as shown in equation (25).

number

[0110] In other words, the receiving phase rotation unit 206 applies a phase rotation to the output AC(k, m) (reflected wave signal) of the correlation calculation unit 205 in the opposite direction to the phase rotation of the transmitting phase rotation unit 104, according to the phase variable pattern (PC(m)).

[0111] For example, N e =4, N s When =1 and PC(m)=1, the radar transmission period (T r The phase shift amount -φ for each ) is -π / 2. In this case, 4(=N e ) radar transmission cycle (T r The phase rotation amounts output in each of the following are, for example, 0, -π / 2, -π, and -3π / 2. Also, N e =4, N s When =1 and PC(m)=-1, the radar transmission period (T r The phase shift amount -φ for each ) is -π / 2. However, when PC(m)=-1, from the relationship exp(jπ)=-1, 4(=N e ) radar transmission cycle (T r The phase rotation amounts output in each of these are, for example, π, π / 2, 0, and -π / 2. In other words, the difference between the corresponding phase rotation amounts -φ (here, {0, -π / 2, -π, -3π / 2} and {π, π / 2, 0, -π / 2}) between different phase variable patterns (PC(m)=1,-1) is π.

[0112] When complementary codes are used as transmission codes, the receiving phase rotation unit 206 outputs a signal ACP(k, m) obtained by applying a phase rotation to the output AC(k, m) of the correlation calculation unit 205, so that the phase shift of the two codes constituting the complementary code within the transmission period becomes zero, as shown in equation (26) below.

number

[0113] For example, N e = 8, N s = 1, when PC(m) = 1, the phase shift amount -φ becomes -π / 2. For 8 (= N e ) radar transmission periods (T r ), the amount of phase rotation output in each of them is, for example, 0, 0, -π / 2, -π / 2, -π, -π, -3π / 2, -3π / 2. Also, N e = 8, N s = 1, when PC(m) = -1, the phase shift amount -φ becomes -π / 2. However, when PC(m) = -1, due to the relationship exp(jπ) = -1, for 8 (= N e ) radar transmission periods (T r ), the amount of phase rotation output in each of them is, for example, π, π, π / 2, π / 2, 0, 0, -π / 2, -π / 2.

[0114] The coherent integrator 207 uses the correlation operation value ACP(k, m) output from the reception phase rotation unit 206 for each discrete time k of the m-th radar transmission period, and adds (coherent integration) the correlation operation value ACP(k, m) for each discrete time k over a period of a predetermined number N e of radar transmission periods, and calculates the coherent integration value ACC(k, v) for each discrete time k. Here, k = 1, …, N v is true.

[0115] Specifically, the v-th coherent integration value ACC(k, v) is calculated as shown in the following equation (27).

Equation

[0116] Here, the period of the radar transmission period (that is, the integration section of the coherent integrator 207) in which the correlation operation value ACP(k, v) is added for each discrete time k in the coherent integrator 207 is a predetermined number N eBy setting it as such, even when the received DC offset is included in the correlation operation value ACP(k, v), the received DC offset component can be canceled out based on the following principle. Along with the suppression of the noise component, the deterioration of the radar detection performance due to the received DC offset can be prevented.

[0117] That is, the received DC offset component α Rx , and the transmitted DC offset component α dTx including the Doppler frequency fluctuation f Tx exp(j2πf dTx ×T r ) are included. The output AC(k, m) of the correlation operation unit 205 includes the components shown in the following equation (28) regardless of k and m.

Equation

[0118] Also, the output ACP(k, m) of the received phase rotation unit 206 includes the components shown in the following equation (29) regardless of k.

Equation

[0119] Therefore, the output ACC(k, v) of the coherent integration unit 207 includes the components shown in the following equation (30) regardless of k.

Equation

[0120] As shown in equation (30), in the integration interval of the coherent integration unit 207,

Equation

Equation

[0121] On the other hand, Doppler frequency variation f dTx If k is not zero, that is, if the reflected wave signal contains Doppler frequency fluctuations, the output ACC(k,v) of the coherent integrator 207 will contain the transmitted DC offset component shown in equation (31) below, regardless of k.

number

[0122] The phase rotation control unit 103 performs a predetermined number of rotations N e (That is, for each integration interval of the coherent integrator 207) the phase rotation phase variable pattern PC(N e The control is performed to randomly change (v-1)+m) to 1 or -1. As a result, in the output of the Doppler analysis unit 208, the residual transmitted DC offset component included in the output ACC(k,v) of the coherent integral unit 207 shown in equation (31) becomes white in the Doppler frequency domain. Therefore, it is possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component and suppress the deterioration of radar detection performance.

[0123] The Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integration unit 207. Specifically, the Doppler analysis unit 208 analyzes the N of the coherent integration unit 207 obtained at discrete time k intervals. d The number of outputs ACC(k, 1) ~ ACC(k, N d Using ) as one unit, the Doppler frequency analysis is performed by aligning the timing of the discrete time k. Note that, as described above, the N of the coherent integral unit 207 d The number of outputs ACC(k, 1) ~ ACC(k, N dSince the remaining transmitted DC offset components included in each of ) are whitened in the Doppler frequency domain, it is possible to suppress an increase in the noise level (floor level) of a specific frequency component due to the transmitted DC offset component in the Doppler frequency analysis by the Doppler analysis unit 208.

[0124] Specifically, as shown in the following equation (32), the Doppler analysis unit 208 performs coherent integration after correcting the phase fluctuation Ψ(f f ΔΨ) corresponding to 2N s different Doppler frequencies f s ) = 2πf s (T r ×N e ).

Equation

[0125] Here, FT_CI(k, f s ) represents the coherent integration result of the Doppler frequency f s ΔΨ at the discrete time k of the Doppler analysis unit 208. Note that f s = -N f + 1,..., 0,..., N f , k = 1,..., (N r + N u )N s / N o , and ΔΨ is the phase rotation unit.

[0126] As a result, the signal processing unit 203 obtains, for each discrete time k, the coherent integration results FT_CI(k, -N f + 1),..., FT_CI(k, N f - 1) corresponding to 2N f Doppler frequency components during each period of a plurality of times N r ×N e ×N d of the radar transmission period T r ×N e ×N d = T r ×Q). Note that j is the imaginary unit.

[0127] ΔΨ = 1 / N d In this case, the processing of the Doppler analysis unit 208 is performed at the sampling interval (T r ×N e ), sampling frequency 1 / (T r ×N e This is equivalent to performing a Discrete Fourier Transform (DFT) on the output of the coherent integral unit 207.

[0128] Also, N f By setting N to a power of 2, the Doppler analysis unit 208 can apply Fast Fourier Transform (FFT) processing, thereby reducing the amount of computation required. f >N d So, q>N d In the region where ACC(k, N d By performing zero-padding (w-1)+q+1)=0, the same FFT process can be applied, reducing the amount of computation required.

[0129] The radar device 10 outputs FT_CI(k, f) from the Doppler analysis unit 208. s Based on this, the distance / Doppler frequency (relative velocity) of the radar measurement target is estimated.

[0130] In other words, the output of the Doppler analysis unit 208 is FT_CI(k, f s The square of the absolute value of |FT_CI(k, f s )| 2 This is a discrete time interval k, with a Doppler frequency f. s This corresponds to the reflected wave reception level from each target. From this, the radar device 10 determines the distance R(k) to the target based on the Doppler f and the discrete time k at which the noise level reaches a peak power value above a predetermined level, and the relative velocity v based on the Doppler frequency. d (f s ) can be estimated as follows. Note that f s =-N f +1,..,0,...,N f k=1,…, (N r +N u )Ns / N o That is the case.

[0131] When converting time information k to distance information R(k), the following equation (33) is used. Here, Tw represents the code transmission interval, L represents the pulse code length, and C0 represents the speed of light.

number

[0132] Also, Doppler frequency information f s The relative velocity component v d (f s When converting to ), use the following equation (34). Here, λ is the wavelength of the carrier frequency of the RF band radar transmission signal output from the transmitting radio unit 106.

number

[0133] The effects of the operation of the phase rotation control unit 103, the transmission phase rotation unit 104, and the reception phase rotation unit 206, as described above, were confirmed using computer simulations, and the results are shown below.

[0134] Figures 4A and 4B show the output (received level) of the Doppler analysis unit 208 under the condition that one measurement target is moving away from the radar device 10 at a speed of 20 km / h to a distance of 5 m.

[0135] In Figures 4A and 4B, the condition is assumed that a transmission DC offset (carrier leak) exists in the transmitting radio unit 106. Also, in Figures 4A and 4B, the radar transmission code is a complementary code (code length L=64), and the number of additions in the coherent integral unit 207 is N. e =32, Number of samples N in the Doppler analysis unit 208 d The results of a computer simulation using =512 are shown below.

[0136] Furthermore, Figure 4A shows the results when the phase inversion (PC(m)) is not randomly switched, unlike the operation of the radar device 10 according to this embodiment. On the other hand, Figure 4B shows the results when the phase rotation control unit 103 randomly switches the phase inversion using the phase variable pattern PC(m) as described above.

[0137] In both Figure 4A and Figure 4B, a sharp peak appears at a coordinate where the speed is 20 km / h at a distance of 5 m, indicating that the desired measurement target has been detected.

[0138] However, in Figure 4A, in addition to the peak of the desired measurement target at a distance of 5m and a speed of 20km / h, weak reception level peaks uniformly appear across the entire distance range (0 to 200m) in the Doppler frequency component at 20km / h. These weak reception level peaks are frequency components that should not exist and can cause false detections.

[0139] On the other hand, Figure 4B, which shows the results based on the operation according to this embodiment, does not show the phenomenon, as detected in Figure 4A, where weak reception level peaks uniformly appear across all distance ranges for a specific Doppler frequency component, in addition to the peak of the desired measurement target at a distance of 5m and a speed of 20km / h. In other words, according to this embodiment, it can be confirmed that even under conditions where a transmission DC offset (carrier leak) exists in the transmitting radio unit 106, there is no degradation of radar performance that could cause false detection.

[0140] As described above, the radar device 10 has multiple N e The radar transmission cycle (T) r ) the period equivalent to (N e ×T r The phase rotation amount of the phase variable pattern PC(m) for the radar transmission signal (transmission code) within ) is set for a period (N e ×T rThe system includes a phase rotation control unit 103 that changes the phase rotation for each step, a transmission phase rotation unit 104 that applies a phase rotation (first phase rotation) to the radar transmission signal according to a phase variable pattern, and a reception phase rotation unit 206 that applies a phase rotation in the opposite direction to the first phase rotation to the reflected wave signal (correlation calculation value ACC(k,m) in Figure 1) according to a phase variable pattern.

[0141] In other words, through the operation of the phase rotation control unit 103, the transmitting phase rotation unit 104, and the receiving phase rotation unit 206, the radar device 10 operates during a period (N e ×T r Multiple transmission codes that cancel the received DC offset within the given field are treated as one unit, and for each unit of transmission codes, it is randomly switched whether or not to apply a phase inversion (a code with a phase change of π).

[0142] As a result, in the coherent integration unit 207 of the radar device 10, the output after coherent integration processing does not become a fixed phase fluctuation, and the residual transmitted DC offset component is whitened in the Doppler frequency domain. Therefore, it is possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component in the output of the Doppler analysis unit 208, thereby suppressing the deterioration of radar detection performance.

[0143] Therefore, according to this embodiment, it is possible to suppress the deterioration of radar detection performance due to circuit errors.

[0144] Furthermore, according to this embodiment, the radar device 10 can prevent deterioration of radar detection performance caused by the transmitted DC offset without requiring a circuit configuration to correct the transmitted DC offset (carrier leak) with high precision, thus simplifying the configuration of the radar device 10.

[0145] Although Figure 1 shows the case where the receiving phase rotation unit 206 of the radar device 10 is located after the correlation calculation unit 205, as shown in Figure 5, the same results as in the above embodiment can be obtained even if the receiving phase rotation unit 206 is located before the correlation calculation unit 205.

[0146] [Embodiment 2] Since the radar device according to this embodiment has the same basic configuration as the radar device 10 according to Embodiment 1, it will be explained with reference to Figure 1.

[0147] In this embodiment, we will describe a case where multiple radar devices 10 (Figure 1) according to Embodiment 1 are provided, as shown in Figure 6. Below, as an example, we will describe a case where two radar devices A and B shown in Figure 6 are provided.

[0148] In this embodiment, the phase-variable patterns controlled by the phase rotation control unit 103 of each radar device 10 are made different from each other. For example, in radar device A shown in Figure 6, the phase-variable pattern PP (1) (q) (where q = 1, 2, ..., N) pp ) is set, and in radar device B, PP (1) (q) A different phase-variable pattern PP (2) (q) (where q = 1, 2, ..., N) pp ) will be set.

[0149] As an example, the phase rotation control unit 103 of each radar device 10 may set different phase variable patterns by shifting the transmission timing of the same M-sequence code. For example, the phase variable pattern PP of radar device A. (1) (q) is a pattern using an M-sequence code with code length 511 (where q=1,..., 511). In this case, the phase-variable pattern PP of radar device B. (2) (q) is the M-sequence code used in radar device A, N shift It is also acceptable to use a pattern where only the transmission timing is shifted. That is, PP (2) (q) = PP (1) (q+N shift )

[0150] For example, Figure 6 shows an example in which the radar receiver 200 of radar device A receives a received signal (desired signal), which is a reflected wave of the radar transmission signal transmitted by radar device A, and a reflected wave (interference signal) of the radar transmission signal transmitted by radar device B.

[0151] Even in this case, as described above, by making the phase-variable patterns between radar devices A and B different, the transmitted DC offset component remaining in the received signals from each radar device A and B is whitened in the Doppler frequency region, similar to Embodiment 1. Therefore, it is possible to prevent an increase in the noise level (floor level) of a specific Doppler frequency component, and to suppress the deterioration of the radar detection performance of radar device A. In other words, the effect of reducing mutual interference between multiple radar devices 10 that overlap in the same frequency band or some frequency bands can be obtained.

[0152] Figure 7 shows the results of a computer simulation evaluating the amount of mutual interference when radar devices A and B transmit radar transmission signals using different phase-variable patterns.

[0153] In Figure 7, radar devices A and B use complementary codes as transmission codes to control the phase shift in the phase rotation control unit 103, N e =16, N s =2. Also, in the Doppler analysis unit 208 of radar device A and radar device B, N d I used =512.

[0154] The horizontal axis in Figure 7 represents the radar transmission period (T). r The graph shows the time shift amount of the transmission timing of the phase-variable pattern, with units of ). The vertical axis shows the received power of radar device A (Desired Power [plotted with black circles]), the received power of radar device B (Undesired Power [plotted with × marks]), and the signal-to-interference power ratio (SIR [plotted with white circles]) of radar device A.

[0155] From the simulation results shown in Figure 7, the phase variable pattern for radar device A and radar device B is 16(=N) at the transmission timing of the M-sequence code. e It can be confirmed that if there is a time shift in the transmission timing that is greater than the number of radar transmission cycles, the SIR of radar device A improves by about 20 dB.

[0156] The phase rotation control unit 103 sequentially reads out each element PP(q) of the phase variable pattern, N e The radar transmission cycle (=N) e ×T r It repeatedly outputs the same element's value over ) . From this, N shift If ≥ 1, then 16 (=N) between different phase-variable patterns. e This results in a time shift in the transmission timing that is greater than the radar transmission cycle, indicating that the interference suppression effect can be enhanced. Note that the amount of SIR improvement is equal to the code length N of the code used in the phase-variable pattern. pp And the parameter N used in the Doppler analysis unit 208 d (For example, N d It depends on (=512). Therefore, within the allowable measurement time, the longest possible N p and N d Using this method is more preferable from the perspective of improving SIR.

[0157] As described above, in this embodiment, the phase-variable patterns differ among the multiple radar devices 10. For example, the transmission timing of the same M-sequence code used in the phase-variable patterns of each of the multiple radar devices 10 is shifted. This reduces mutual interference among the multiple radar devices 10.

[0158] In the above case, the transmission timings of multiple radar devices 10 may coincide by chance, which may result in cases where mutual interference cannot be suppressed probabilistically. In response to this, mutual interference can be reduced probabilistically by the following method.

[0159] Specifically, in multiple radar devices 10, code length N ppAmong the M-sequence codes, the Preferred pair with low cross-correlation is used for each phase-variable pattern.

[0160] Alternatively, code length N pp From the M-sequence codes, starting with the preferred pair with low cross-correlation, N pp It is known that multiple different Gold codes can be generated. Therefore, in multiple radar devices 10, Gold codes generated from preferred pairs with low cross-correlation may be used for each phase-variable pattern.

[0161] Alternatively, in multiple radar devices 10, mutual interference may be reduced probabilistically by randomly varying the phase variation pattern of each device for each radar measurement (or every predetermined number of measurements).

[0162] [Embodiment 3] Embodiment 2 described a method for reducing mutual interference between multiple radar devices 10 by making the phase-variable patterns different among the multiple radar devices 10. In contrast, this embodiment describes a MIMO (Multiple Input Multiple Output) radar configuration that uses multiple transmitting and receiving antennas instead of multiple radar devices 10.

[0163] In other words, in this embodiment, the phase-variable patterns differ from one another between multiple transmitting antennas (i.e., between MIMO streams).

[0164] Figure 8 is a block diagram showing an example of the configuration of the radar device 20 according to this embodiment. In Figure 8, components similar to those in Embodiment 1 (Figure 1) are denoted by the same reference numerals, and their descriptions are omitted.

[0165] The radar device 20 shown in Figure 8 is a time-division MIMO radar configuration that switches between multiple transmitting antennas in a time-division manner to transmit different time-division multiplexed radar transmission signals, and separates each radar transmission signal for reception processing. However, the configuration of the radar device is not limited to this, and the radar device 20 may also be configured to send out different transmission signals that are frequency-division multiplexed or code-division multiplexed from multiple transmitting antennas, and separates each transmission signal for reception processing.

[0166] [Configuration of radar transmitter 300] The radar transmission unit 300 of the radar device 20 includes radar transmission signal generation units 101-1 to 101-Nt, a switching control unit 301, a transmission switching unit 302, and a transmission array antenna unit 303.

[0167] The transmitting array antenna section 303 is composed of Nt transmitting antennas (Tx#1 to Tx#Nt).

[0168] The radar transmission signal generation units 101-1 to 101-Nt are provided, each corresponding to Nt transmitting antennas (Tx#1 to Tx#Nt). Each radar transmission signal generation unit 101 operates in the same manner as in Embodiment 1 (Figure 1). However, the phase variable patterns set for each radar transmission signal generation unit 101 are different from each other.

[0169] For example, the phase-variable pattern in each of the Nt radar transmission signal generation units 101 is set by shifting the transmission timing of the same M-sequence code by one or more code elements.

[0170] Specifically, the phase-variable pattern in the radar transmission signal generation unit 101-1 is PP (1) (q) (where q=1,..., N) pp ) In this case, the phase variable pattern in the radar transmission signal generation unit 101-2 is set to PP (2) (q+N shift ) and the phase variable pattern in the radar transmission signal generation unit 101-3 is set to PP (3) (q+2N shift)) and similarly, the phase variable pattern in the radar transmission signal generation unit 101-Nt is set to PP (Nt) (q+(Nt-1)N shift ) may also be used. However, N shift Assume ≥ 1.

[0171] Furthermore, the code length N is set as the phase-variable pattern for each of the Nt radar transmission signal generation units 101. pp Among the M-sequence codes, a Preferred pair with low cross-correlation may be used. Alternatively, code length N pp From the M-sequence codes, starting with the preferred pair with low cross-correlation, N pp It is known that Nt different Gold codes can be generated. Therefore, Gold codes generated from preferred pairs with low cross-correlation may be used for each of the Nt radar transmission signal generation units 101's phase-variable patterns. Alternatively, mutual interference may be probabilistically reduced by randomly varying each of the Nt radar transmission signal generation units 101's phase-variable patterns for each radar measurement (or every predetermined number of measurements).

[0172] The switching control unit 301 outputs a control signal (hereinafter referred to as the switching control signal) that instructs the switching timing of the transmitting antennas (Tx#1 to Tx#Nt) of the transmitting array antenna unit 303 (i.e., the output switching of the radar transmission signal) to the transmitting switching unit 302 and the radar receiving unit 400 (correlation calculation unit 205 and output switching unit 403).

[0173] The transmission switching unit 302 selects one of the Nt transmitting antennas of the transmission array antenna unit 303 based on the switching control signal from the switching control unit 301, and uses the output signal of the radar transmission signal generation unit 101 corresponding to the selected transmitting antenna as the input to the selected transmitting antenna. The transmission switching unit 302 then frequency-converts the output signal of the selected radar transmission signal generation unit 101 (baseband radar transmission signal) to a predetermined radio frequency band and outputs it to the selected (connected) transmitting antenna.

[0174] The transmitting array antenna unit 303 radiates the radar transmission signal output from the transmitting switching unit 302 into space from the transmitting antenna selected (connected) to the transmitting switching unit 302.

[0175] The following describes the control operation of the switching control unit 301 over the transmission switching unit 302. The control operation of the switching control unit 301 over the radar receiving unit 400 will be described later in the explanation of the operation of the radar receiving unit 400.

[0176] The switching control unit 301 is N e For each radar transmission cycle, a switching control signal is output to the transmission switching unit 302 to sequentially switch between the radar transmission signal generation unit 101 and the transmitting antenna.

[0177] For example, the switching control unit 301 first N e The radar transmission cycle (N) e ×T r In this configuration, the output signal from the radar transmission signal generation unit 101-1 is used as the input to the transmission switching unit 302, and the signal converted to a high-frequency signal is output to the transmission antenna (Tx#1) of the transmission array antenna unit 303.

[0178] The switching control unit 301 then controls the next N e The radar transmission cycle (N) e ×T r In this configuration, the output signal from the radar transmission signal generation unit 101-2 is used as the input to the transmission switching unit 302, and the signal converted to a high-frequency signal is output to the transmission antenna (Tx#2) of the transmission array antenna unit 303.

[0179] The switching control unit 301 repeats the same operation, taking the output signal from the radar transmission signal generation unit 101-Nt as the input to the transmission switching unit 302, and outputting the signal converted to a high-frequency signal to the transmission antenna (Tx#Nt) of the transmission array antenna unit 303.

[0180] Furthermore, the switching control unit 301 will then perform the following N e The radar transmission cycle (N) e ×T r) Then, the output signal of the radar transmission signal generation unit 101-1 is again used as the input to the transmission switching unit 302, and the signal converted to a high-frequency signal is output to the transmission antenna (Tx#1) of the transmission array antenna unit 303.

[0181] The switching control unit 301 performs the above operations a predetermined number of times (N d ×N t Repeat (times).

[0182] In the above operation, the radar transmission unit 300 operates to sequentially read out the GP(m) (see, for example, equation (19) or equation (20)) described in Embodiment 1 as signals generated by each radar transmission signal generation unit 101. Here, m = 1, ..., N e ×N d That is the case.

[0183] [Configuration of radar receiver 400] The radar receiving unit 400 of the radar device 20 includes a receiving array antenna unit 401, an antenna system processing unit 402, and a direction estimation unit 404.

[0184] The receiving array antenna section 401 consists of Na receiving antennas (Rx#1 to Rx#Na). Each of the Na receiving antennas receives the signal (reflected wave signal) that is reflected off a reflective object, including a radar measurement target, from the radar transmission signal transmitted from the radar transmission section 300. Each signal received by the Na receiving antennas is input as a received signal to the antenna system processing unit 402 corresponding to each receiving antenna (Rx#1 to Rx#Na).

[0185] Each antenna system processing unit 402 includes a receiving radio unit 202 and a signal processing unit 203.

[0186] The receiving radio unit 202 of the z-th antenna system processing unit 402-z amplifies the received signal from the z-th receiving antenna (Rx#z) to a predetermined level, frequency-converts the high-frequency band received signal to the baseband band, and converts the baseband band received signal to a baseband band received signal including the I signal and the Q signal. Here, z=1,...,Na.

[0187] The signal processing unit 203 of the z-th antenna system processing unit 402-z consists of an A / D conversion unit 204, a correlation calculation unit 205, a reception phase rotation unit 206, an output switching unit 403, a coherent integration unit 207, and a Doppler analysis unit 208. The signal processing unit 203 includes Nt coherent integration units 207 and Doppler analysis units 208, each corresponding to a transmitting antenna (Tx#1 to Tx#Nt).

[0188] The following describes the operation of each component of the signal processing unit 203 of the z-th antenna system processing unit 402-z, mainly focusing on the differences in operation compared to Embodiment 1.

[0189] The correlation calculation unit 205 processes the complex baseband signal X(N) output from the AD conversion unit 204. v (m-1)+k) (see, for example, equation (23)) and N by the switching control unit 301 e The radar transmission cycle (N) e ×T r A correlation calculation is performed with the transmission code generated in the radar transmission signal generation unit 101 selected for each of the following.

[0190] The receiving phase rotation unit 206 is N e The radar transmission cycle (N) e ×T r For each instance, the switching control unit 301 applies a phase rotation in the opposite direction (cancel direction) to the phase rotation applied by the transmission phase rotation unit 103 of the radar transmission signal generation unit 101 selected by the switching control unit 301 to the signal (correlation calculation value) output from the correlation calculation unit 205.

[0191] The output switching unit 403 is configured in the switching control unit 301 to N e The radar transmission cycle (N) e ×T r The output is switched to the coherent integrator 207-1 to 207-Nt corresponding to the selected transmitting antenna number (#1 to #Nt) for each step.

[0192] For example, when the transmitting antenna (Tx#1) is selected by the switching control unit 301, the output switching unit 403 switches the signal from the receiving phase rotation unit 206 to the coherent integration unit 207-1 corresponding to the transmitting antenna (Tx#1) and outputs it.

[0193] Furthermore, when the transmitting antenna (Tx#2) is selected by the switching control unit 301, the output switching unit 403 switches the signal from the receiving phase rotation unit 206 to the coherent integration unit 207-2 corresponding to the transmitting antenna (Tx#2) and outputs it.

[0194] The output switching unit 403 repeats the same operation, and when the transmitting antenna (Tx#Nt) is selected by the switching control unit 301, it switches the signal from the receiving phase rotation unit 206 to the coherent integration unit 207-Nt corresponding to the transmitting antenna (Tx#Nt) and outputs it.

[0195] No. N D The second coherent integral section 207-N D In the switching control unit 301, N e The radar transmission cycle (N) e ×T r For each selected output of the receiving phase rotation unit 206, multiple N e Period of time (T r ×N e Coherent integration is performed using ) as the unit. Here, N D =1, ..., Nt.

[0196] The Doppler analysis unit 208 performs Doppler frequency analysis on the output signal of the coherent integration unit 207. That is, the Doppler analysis unit 208 analyzes the N obtained from the coherent integration unit 207 at discrete time k intervals. d Using these outputs, Doppler frequency analysis is performed by aligning the timing of discrete time points k.

[0197] As described above, the radar device 20 performs a predetermined number of operations N eThe transmitting antenna is switched by the switching control unit 301 for each integration interval of the coherent integration unit 207, and different phase-variable patterns PC(m) are set between the transmitting antennas. As a result, in the output of the Doppler analysis unit 208 corresponding to each transmitting antenna, similar to Embodiment 1, the residual transmitted DC offset component included in the output ACC(k,v) of the coherent integration unit 207 (see equation (31)) for the radar transmission signal transmitted from each transmitting antenna becomes white in the Doppler frequency domain. Therefore, an increase in the noise level (floor level) of a specific Doppler frequency component can be prevented, and the deterioration of radar detection performance can be suppressed. Furthermore, by setting different phase-variable patterns between the transmitting antennas, mutual interference between radar transmission signals transmitted by each transmitting antenna can be reduced.

[0198] In the following description, the w-th output FT_CI at each discrete time k from the Doppler analysis unit 208 is obtained by performing the same processing in each of the antenna system processing units 402-1 to 402-Na. (z) (1) (k, fs, w) ,…, FT_CI (z) (Na) The elements (k, fs, w) are collectively expressed as the virtual receive array correlation vector h(k, fs, w) as shown in equations (35) and (36). The virtual receive array correlation vector h(k, fs, w) contains Nt × Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na. The virtual receive array correlation vector h(k, fs, w) will be used later to explain the process of estimating the direction of the reflected wave signal from the target based on the phase difference between the receiving antennas. Here, z = 1, ..., Nt, N D =1,..,, Na. Note that f s =-N f +1,..,0,...,N f That is the case.

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[0199] The processing in each component of the signal processing unit 203 of the antenna system processing unit 402-z has been described above.

[0200] The direction estimation unit 404 corrects the phase shift deviation and amplitude deviation between the transmitting antennas of the transmitting array antenna unit 303 and the receiving antennas of the receiving array antenna unit 401 for the virtual array correlation vector h(k, fs, w) from the w-th Doppler analysis unit 208 at discrete time k, which is output from the antenna system processing units 402-1 to 402-Na, by an array correction value h_cal. [b] By multiplying by this, the virtual received array correlation vector h corrected for inter-antenna deviation is obtained. _after_cal Calculate (k, fs, w). Virtual receive array correlation vector h _after_cal (k, fs, w) are expressed by the following equation (37). Note that b = 1, ..., (Nt × Na).

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[0201] virtual receive array correlation vector h corrected for inter-antenna deviation _after_cal (k, fs, w) is Na × N r It is a column vector consisting of n elements. Below, we define the virtual receive array correlation vector h _after_cal Each element (k, fs, w) is h1(k, fs, w), ..., h Na×Nr The notation (k, fs, w) is used to explain the direction estimation process.

[0202] Then, the direction estimation unit 404 generates a virtual received array correlation vector h _after_cal Using (k, fs, w), direction estimation is performed based on the phase difference of the reflected wave signals between the receiving antennas.

[0203] The direction estimation unit 404 calculates the direction estimation evaluation function value P. HA spatial profile is calculated by varying the azimuth direction θ within a predetermined angular range at (θ, k, fs, w). A predetermined number of maximum peaks from the calculated spatial profile are extracted in descending order, and the azimuth direction of the maximum peaks is used as the estimated direction of arrival.

[0204] Note that the evaluation function value P H (θ, k, fs, w) can be set in various ways depending on the direction of arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Reference Non-Patent Document 1 may be used.

[0205] (Reference Non-Patent Document 1) Direction-of-arrival estimation using signal subspace modeling. Cadzow, JA; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1, Publication Year: 1992, Page(s): 64 - 79

[0206] For example, the beamformer method can be expressed as shown in equations (38) and (39). Other methods such as Capon and MUSIC can also be applied in a similar manner.

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[0207] Here, the superscript H is the Hermitian transpose operator. Also, a H (θ u ) is the azimuth direction θ u This shows the direction vector of the virtual receiving array for the incoming wave, and also θ u This is the azimuth range used for estimating the direction of arrival, which is changed by a predetermined azimuth interval β1. For example, θ u It will be set as follows: θ u =θmin + uβ1, u=0,…, NU NU = floor[(θmax-θmin) / β1]+1

[0208] Here, floor(x) is a function that returns the largest integer value not exceeding the real number x.

[0209] The configuration of the radar receiver unit 400 has been described above.

[0210] As described above, in this embodiment, in the radar device 20 which is a MIMO radar, by making the phase variable pattern for the radar transmission signal transmitted by each transmitting antenna different, it is possible to reduce mutual interference between radar transmission signals transmitted by switching between multiple transmitting antennas.

[0211] Furthermore, in this embodiment, mutual interference between radar transmission signals transmitted by switching between multiple transmitting antennas can be reduced, allowing for a shorter time interval between switching between multiple transmitting antennas and thus shortening the detection time.

[0212] The embodiments relating to one aspect of this disclosure have been described above.

[0213] Furthermore, the above embodiments and the operations related to each variation may be implemented in appropriate combinations.

[0214] [Other embodiments] (1) In the above embodiment, the radar devices 10 and 20 may be equipped with a circuit configuration for easily correcting the transmitted DC offset. By using in combination the configuration for controlling the phase variable pattern described above and the configuration for correcting the transmitted DC offset component, the radar devices 10 and 20 can further suppress the increase in noise level caused by the transmitted DC offset.

[0215] (2) In the above embodiments, the case using coded pulse radar was described, but this disclosure is also applicable to radar systems using frequency-modulated pulse waves, such as chirp pulse radar.

[0216] (3) In the radar devices 10 and 20 shown in Figures 1, 5, 6, and 8, the radar transmitters 100 and 300 and the radar receivers 200 and 400 may be individually arranged in physically separate locations.

[0217] (4) The radar devices 10 and 20, although not shown in the figures, include, for example, a CPU (Central Processing Unit), a storage medium such as ROM (Read Only Memory) storing a control program, and working memory such as RAM (Random Access Memory). In this case, the functions of each of the above-mentioned parts are realized by the CPU executing the control program.

[0218] Although various embodiments have been described above with reference to the drawings, it goes without saying that this disclosure is not limited to such examples. It is clear to those skilled in the art that various modifications or alterations can be conceived within the scope of the claims, and these will naturally also fall within the technical scope of this disclosure. Furthermore, the components of the above embodiments may be combined in any way without departing from the spirit of the disclosure.

[0219] In the embodiments described above, the disclosure has been explained using examples configured with hardware, but the disclosure can also be implemented with software in conjunction with hardware.

[0220] Furthermore, each functional block used in the description of the above embodiments is typically implemented as an integrated circuit (LSI). The integrated circuit controls each functional block used in the description of the above embodiments and may have input and output terminals. These may be individually integrated into a single chip, or some or all of them may be integrated into a single chip. Here, we refer to it as an LSI, but depending on the degree of integration, it may also be called an IC, system LSI, super LSI, or ultra LSI.

[0221] Furthermore, the method of integrated circuit implementation is not limited to LSIs; it may also be implemented using dedicated circuits or general-purpose processors. After LSI manufacturing, FPGAs (Field Programmable Gate Arrays) that can be programmed, or reconfigurable processors that allow for the reconfiguration of the connections or settings of circuit cells inside the LSI, may also be used.

[0222] Furthermore, if advances in semiconductor technology or other derived technologies lead to the emergence of integrated circuit technologies that can replace LSIs, then naturally, functional blocks can be integrated using those technologies. The application of biotechnology, for example, is a possibility.

[0223] <Summary of this disclosure> The radar apparatus of the present disclosure comprises a radar transmitting unit that transmits a radar signal at a predetermined transmission period, and a radar receiving unit that receives a reflected wave signal in which the radar signal has been reflected at a target, wherein the radar transmitting unit comprises a phase rotation control unit that changes the pattern of the phase rotation amount to be applied to the radar signal for each period corresponding to the transmission period of multiple times Ne (Ne is an integer greater than 1), the phase rotation amount is obtained by dividing an integer multiple of 2π by Ne multiple times, and the pattern of the phase rotation amount is obtained by multiplying the phase rotation amount by the sign polarity of a predetermined code sequence, and a transmitting phase rotation unit that applies a first phase rotation to the radar signal according to the pattern of the phase rotation amount, and the radar receiving unit comprises a receiving phase rotation unit that applies a second phase rotation to the reflected wave signal in the opposite direction to the first phase rotation according to the pattern of the phase rotation amount.

[0224] In the radar device of this disclosure, when the sign polarity of the predetermined code sequence is +1, the phase change of the phase rotation amount is 0, and when the sign polarity of the predetermined code sequence is -1, the phase change of the phase rotation amount is π.

[0225] In the radar device of this disclosure, the predetermined code sequence is based on a pseudo-random code, an M-sequence code, or a Gold code.

[0226] In the radar device of this disclosure, when generating the radar signal using complementary codes, the transmission phase rotation unit applies the same phase rotation amount within two transmission cycles in which the pairs of codes constituting the complementary codes are transmitted, and applies a phase rotation amount that changes with each of the two transmission cycles.

[0227] In the radar device of this disclosure, a predetermined code sequence is different from one another among multiple radar devices.

[0228] In the radar device of this disclosure, the radar transmitting unit comprises a plurality of transmitting antennas, and the predetermined code sequence is different from one another among the transmitting antennas.

[0229] The radar method of this disclosure is a radar method which involves changing the pattern of the amount of phase rotation applied to a radar signal over a period corresponding to multiple transmission cycles of Ne (where Ne is an integer greater than 1) for each period, applying a first phase rotation to the radar signal according to the pattern of the amount of phase rotation, transmitting the radar signal to which the first phase rotation has been applied in the transmission cycle, receiving a reflected wave signal to which the radar signal to which the first phase rotation has been applied has been reflected at a target, and applying a second phase rotation in the opposite direction to the first phase rotation to the reflected wave signal according to the pattern of the amount of phase rotation, wherein the amount of phase rotation is obtained by dividing an integer multiple of 2π by Ne multiple times, and the pattern of the amount of phase rotation is obtained by multiplying the amount of phase rotation by the sign polarity of a predetermined code sequence. [Industrial applicability]

[0230] This disclosure is suitable as a radar device for detecting a wide-angle range. [Explanation of symbols]

[0231] 10,20 Radar equipment 100,300 Radar Transmitter 101 Radar transmission signal generation unit 102 Code generator 103 Phase rotation control unit 104 Transmitting Phase Rotation Section 105 Modulation section 106 Transmitting Radio Unit 107 Transmitting Antenna 111 Code storage unit 112 DA conversion section 200,400 Radar Receiver 201 Receiving Antenna 202 Receiving Radio Unit 203 Signal Processing Unit 204 AD Conversion Unit 205 Correlation Calculation Unit 206 Receiving Phase Rotation Section 207 Coherent Integrator 208 Doppler Analysis Department 301 Switching Control Unit 302 Transmission switching section 303 Transmitting array antenna section 401 Receiving array antenna section 402 Antenna System Processing Unit 403 Output switching section 404 Direction estimation section

Claims

1. A signal generation unit that generates a transmission signal, A transmission phase rotation unit that applies a phase rotation based on multiple different phase rotation amounts controlled by a series of multiple mutually different phase variable patterns to the transmission signal at each transmission period, A multiplexing unit that applies multiplexing to a plurality of different radar transmission signals, which are the transmission signals to which the phase rotation has been applied, A radar transmission radio unit that outputs the multiple different radar transmission signals to which the above multiplexing process has been applied to multiple transmitting antennas, A radar receiving radio unit that receives one or more reflected wave signals in which at least one radar transmission signal from among the multiple different radar transmission signals is reflected at a target via multiple receiving antennas, Equipped with, The aforementioned multiple different phase rotation amounts are a series obtained by sequentially adding shift amounts that are integer multiples of 2π or more for each transmission period. The plurality of distinct phase-variable patterns have elements for each of the plurality of transmission cycles, and are patterns that vary the amount of phase rotation for each of the plurality of transmission cycles. Radar signal processing circuit.

2. The signal generation unit outputs a signal sequence that constitutes the transmission signal. The plurality of different phase rotation amounts include a phase rotation amount that sets the shift amount to zero for one signal composed of the signal sequence. The radar signal processing circuit according to claim 1.

3. The multiplexer outputs one radar transmission signal of the multiple different radar transmission signals to one antenna selected from the multiple transmitting antennas for each of the multiple transmission cycles. The radar signal processing circuit according to claim 1.

4. The multiplexing unit outputs the multiple different radar transmission signals to the multiple transmitting antennas simultaneously for each of the multiple transmission cycles. The radar signal processing circuit according to claim 1.

5. Generate a transmission signal, A phase rotation based on multiple different phase rotation amounts controlled by a series of multiple mutually different phase variable patterns is applied to the transmitted signal for each transmission period. Multiplexing is applied to a plurality of different radar transmission signals, which are the transmission signals to which the phase rotation has been applied. The multiple different radar transmission signals to which the above multiplexing process has been applied are output to multiple transmitting antennas. The system receives one or more reflected wave signals, in which at least one of the multiple different radar transmission signals is reflected at the target, via multiple receiving antennas. The aforementioned multiple different phase rotation amounts are a series obtained by sequentially adding shift amounts that are integer multiples of 2π or more for each transmission period. The plurality of distinct phase-variable patterns have elements for each of the plurality of transmission cycles, and are patterns that vary the amount of phase rotation for each of the plurality of transmission cycles. Radar signal processing method.

6. The aforementioned transmission signal has a signal sequence, The plurality of different phase rotation amounts include a phase rotation amount for one signal composed of the signal sequence in which the shift amount becomes zero. The radar signal processing method according to claim 5.

7. The multiplexing process outputs one radar transmission signal from the multiple different radar transmission signals to one antenna selected from the multiple transmitting antennas for each of the multiple transmission cycles. The radar signal processing method according to claim 5.

8. The multiplexing process outputs the multiple different radar transmission signals to the multiple transmitting antennas simultaneously for each of the multiple transmission cycles. The radar signal processing method according to claim 5.

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