Signal transmission link, signal calibration link, signal compensation link, signal transceiving link, integrated circuit, electromagnetic wave sensor, and device

Abstract

The present disclosure relates to the technical field of electromagnetic wave sensors, and in particular to a signal transmission link, a calibration signal link, a signal compensation link, a signal transceiving link, an integrated circuit, an electromagnetic wave sensor, and a device. The transmission link comprises an analog signal source and a digital phase shifter, wherein the analog signal source may be configured to provide an initial analog signal, and the digital phase shifter may be configured to generate a phase shift signal in a digital domain, and perform phase shifting on the initial analog signal on the basis of the phase shift signal, so as to perform a preset phase shift operation on the initial analog signal, such that the phase modulation resolution and phase modulation accuracy are effectively improved, and an off-line calibration operation on a link and a device such as a phase shifter in the transmission link can also be avoided, thus reducing the complexity and difficulty of engineering implementation. In addition, the area and loss of a transmission link having a phase shifter architecture can also be effectively reduced, thereby improving the stability of a system, and reducing a channel coupling degree.

Description

Signal transmission, calibration, compensation and transceiver links, integrated circuits, electromagnetic wave sensors and equipment

[0001] This application claims priority to the Chinese patent application filed with the Patent Office of China on June 14, 2023, with application number 202310702586.5, and invention name “Signal transmission, calibration, compensation and transceiver link, IQ mixer, integrated circuit, sensor and equipment”, and the Chinese patent application filed with the Patent Office of China on May 14, 2024, with application number 202410599441.1, and invention name “Signal transmission link and method, transceiver link, integrated circuit and electromagnetic wave device”, the contents of which should be understood as incorporated into this disclosure by reference. Technical Field

[0002] The embodiments of the present disclosure relate to, but are not limited to, the field of electromagnetic wave sensor technology, and specifically relate to a signal transmission, calibration, compensation and transceiver link, an integrated circuit, an electromagnetic wave sensor and equipment. Background Art

[0003] Since the signal transmission link adopts an analog phase shifter architecture, it has problems such as low phase modulation precision (resolution) and phase modulation accuracy (accuracy), which makes it unable to meet various requirements of high precision and accuracy.

[0004] Summary of the Invention

[0005] To address the aforementioned technical issues, the disclosed embodiments provide a signal transmission, calibration, compensation, and transceiver link, IQ mixers, integrated circuits, electromagnetic wave sensors, and devices, all based on a digital phase shifter architecture to form a signal transceiver link, along with corresponding calibration and compensation link technologies. These technologies effectively improve phase modulation resolution and accuracy while avoiding offline calibration of phase shifters and other links and components in the transmission link, thereby reducing the complexity and difficulty of engineering implementation. Furthermore, they effectively reduce the area and loss of the phase shifter architecture's transmission link, improving system stability and reducing channel coupling.

[0006] An embodiment of the present disclosure provides a signal transmission link for use in an electromagnetic wave sensor. The transmission link includes an analog signal source and a digital phase shifter. The analog signal source can be configured to provide an initial analog signal. The digital phase shifter can be configured to generate a phase-shifted signal in the digital domain and perform phase shifting on the initial analog signal based on the phase-shifted signal to perform a preset phase shift operation on the initial analog signal.

[0007] Exemplarily, the signal transmission link further includes a transmitting antenna, which can be configured to radiate the phase-shifted initial analog signal to a preset spatial area.

[0008] Exemplarily, the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter and a mixer. The digital phase shifter can be configured to generate a digital phase shift signal. The digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal. The mixer can be configured to use the received analog phase shift signal to perform a mixing operation on the received initial analog signal so as to perform a preset phase shift operation on the initial analog signal.

[0009] Exemplarily, the digital phase-shifted signal source includes a direct digital frequency synthesizer, the digital-to-analog converter is an IQ digital-to-analog converter, and the mixer is an IQ mixer.

[0010] Exemplarily, the digital phase-shifted signal is a single-tone signal, and the initial analog signal is a swept-frequency signal; or, the digital phase-shifted signal is a swept-frequency signal, and the initial analog signal is a single-tone signal.

[0011] Exemplarily, the signal transmission link transmits a frequency modulated continuous wave signal.

[0012] An embodiment of the present disclosure also provides a signal transmission link, comprising a main signal transmission path and a signal calibration link integrated in the same integrated circuit; wherein: the signal calibration link is configured to calibrate the main signal transmission path to obtain compensation information; and the main signal transmission path is configured to generate a radio frequency transmission signal after performing a compensation operation based on the compensation information, so as to achieve target detection and / or communication.

[0013] Exemplarily, the compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.

[0014] Exemplarily, the main signal transmission path includes a first signal source and a phase shifter; wherein, the first signal source is configured to generate a first analog signal; and the phase shifter is configured to frequency shift and / or phase shift the first analog signal to form a radio frequency transmission signal.

[0015] Exemplarily, when the phase shifter adopts a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting mixer is configured to mix the first analog signal and the second analog signal to form the radio frequency transmit signal;

[0016] When the phase shifter is an orthogonal architecture, the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting-end mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting-end mixer is configured to perform frequency shifting and / or phase shifting on the first analog signal based on the second analog signal to form the RF transmit signal.

[0017] Exemplarily, the main transmitting path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter, and the compensation circuit is used to merge the compensation signal and the signal output by the second signal source and then output it.

[0018] Exemplarily, when the signal output by the main signal transmission path is two orthogonal signals, the compensation signal used by the compensation circuit is two orthogonal signals; when the signal output by the main signal transmission path is not two orthogonal signals, the compensation signal used by the compensation circuit is a signal of the same type as the signal output by the second signal source, where the signal type is a digital signal or an analog signal.

[0019] Exemplarily, the compensation circuit includes a compensation signal generator and an adder, wherein: the compensation signal generator is configured to generate the compensation signal; the adder is connected to the compensation signal generator and the second signal source, and is used to perform a signal superposition operation on the signal output by the second signal source and the compensation signal output by the compensation signal generator.

[0020] Exemplarily, when the signal output by the main signal transmission path is a two-way orthogonal signal, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal imbalance compensation signal unit, and a local oscillator leakage compensation signal unit; when the signal output by the main signal transmission path is not a two-way orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leakage compensation signal unit; wherein, the compensation signal generated by the harmonic compensation signal unit is used to eliminate the harmonic signal of the main frequency signal in the main signal transmission path; the compensation signal of the local oscillator leakage compensation signal unit is used to compensate for the leakage signal generated by the transmitting end local oscillator signal in the main signal transmission path; and the compensation signal generated by the orthogonal imbalance compensation signal unit is used to compensate for the mirror signal of the main frequency signal in the main signal transmission path.

[0021] Exemplarily, the compensation signal generated by the harmonic compensation signal unit has the same frequency and amplitude as the harmonic signal in the main signal transmission path, and has an opposite phase.

[0022] Exemplarily, the harmonic compensation signal unit includes an n-th power module or an n-fold frequency signal generator; an n-th power module, wherein the signal input end of the n-th power module is connected to the signal output end of the second signal source, and is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source to obtain the compensation signal; an n-fold frequency signal generator, which is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source to obtain the compensation signal, wherein the value of n is a positive integer. Exemplarily, the value of n is an odd number. Exemplarily, the value of n is 3.

[0023] Exemplarily, the compensation signal generated by the local oscillator leakage compensation signal unit is generated based on a leakage signal corresponding to the first analog signal used by the phase shifter.

[0024] Exemplarily, the compensation signal generated by the local oscillator leakage compensation signal unit has the same frequency and amplitude as the leakage signal but an opposite phase.

[0025] Exemplarily, the compensation signal used by the orthogonal imbalance compensation circuit has the same frequency, the same amplitude, and the opposite phase as the image signal corresponding to the main signal transmission path and the image signal corresponding to the desired signal generated by the main signal transmission path.

[0026] Exemplarily, the compensation signal used by the quadrature imbalance compensation circuit is determined according to a signal output by the second signal source and a frequency-inverted complex conjugate signal of the signal output by the second signal source.

[0027] Exemplarily, the method for obtaining the compensation signal generated by the orthogonal imbalance compensation circuit includes: obtaining the product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal; calculating the difference between the signal output by the second signal source and the adjustment signal to obtain the compensation signal generated by the orthogonal imbalance compensation circuit.

[0028] Exemplarily, the pre-compensation coefficient is determined based on a ratio of an amplitude of a desired signal to an amplitude of an image signal corresponding to the desired signal.

[0029] Exemplarily, the signal calibration link includes a signal receiving link corresponding to the main signal transmission path, wherein the signal receiving link is used to receive and process the echo signal corresponding to the FMCW RF transmission signal; wherein: the signal receiving link is configured to obtain a sampling signal from the main signal transmission path and obtain configuration information of the compensation signal based on the sampling signal; wherein the frequency of the transmitting end local oscillator signal used in the main signal transmission path and the receiving end local oscillator signal used in the signal receiving link are different.

[0030] Exemplarily, the signal calibration link further includes: a frequency adjustment circuit configured to adjust the frequency of at least one of the sampling signal and the receiving end local oscillator signal.

[0031] Exemplarily, the signal input end of the frequency adjustment circuit is connected between the main signal transmission path and the transmitting antenna, and the signal output end of the frequency adjustment circuit is connected between the signal receiving link and the receiving antenna, wherein the frequency adjustment circuit is configured to adjust the frequency of the RF transmission signal output by the main signal transmission path; correspondingly, the signal receiving link is configured to obtain a sampling signal from the frequency adjustment circuit and obtain configuration information of the compensation signal based on the sampling signal; or, the frequency adjustment circuit is connected between a receiving end mixer and a receiving end local oscillator in the signal receiving link, and is configured to adjust the frequency of the receiving end local oscillator signal, wherein the receiving end local oscillator is used to generate a receiving end local oscillator signal, and the receiving end mixer is used to demodulate the received signal using the receiving end local oscillator signal; the signal receiving link is connected between the main signal transmission path and the transmitting antenna, and is configured to obtain a sampling signal from the main signal transmission path, and process the sampling signal using the signal output by the frequency adjustment circuit to obtain configuration information of the compensation signal.

[0032] Exemplarily, when the radio frequency transmission signal transmitted by the signal transmission main path is two orthogonal signals, the signal calibration link is a signal receiving link that supports processing echo signals including the two orthogonal signals.

[0033] Exemplarily, when the RF transmission signal transmitted by the main signal transmission path is two orthogonal signals, the signal calibration link is provided with an orthogonal processing circuit and two signal receiving links, wherein each signal receiving link does not support processing of orthogonal echo signals; wherein the orthogonal processing circuit is configured to perform RF processing on the received orthogonal signal to obtain two signals, and send the two signals to the two signal receiving links respectively.

[0034] Exemplarily, the signal calibration link, the first input end of the signal calibration link is connected between the voltage-current converter and the current switch in the transmitting end mixer in the signal transmission main path, the second input end is between the signal transmission main path and the transmitting antenna, and the signal output end is connected to the compensation circuit in the signal transmission main path, and is used to obtain the signal in the signal transmission main path from at least one of the first input end and the second input end, and determine the compensation information based on the obtained signal.

[0035] Exemplarily, the signal calibration link includes a calibration demodulator, a multiplexer and a calibration module; wherein: the calibration demodulator is configured to obtain the signal in the signal transmission main path from the second input terminal and perform demodulation processing; the multiplexer has two signal input terminals and one signal input terminal, one of which is connected between the voltage-current converter and the current switch in the transmitting mixer, and the other signal input terminal is connected to the signal input terminal of the calibration demodulator, for outputting a signal corresponding to one of the two signal input terminals; the calibration module is configured to determine the compensation information based on the signal output by the multiplexer.

[0036] Exemplarily, when the radio frequency transmission signals transmitted by the signal transmission main path are two orthogonal signals, the acquisition signals transmitted by the signal calibration link are two orthogonal signals.

[0037] Exemplarily, the acquisition signals transmitted by the signal calibration link are two orthogonal signals, and the signal calibration link also includes: an analog-to-digital converter, connected between the signal acquisition circuit and the calibration module, and configured to perform analog-to-digital conversion on the acquisition signals output by the signal acquisition circuit.

[0038] An embodiment of the present disclosure also provides a signal transceiver link, comprising a signal transmission link as described in any embodiment of the present disclosure, and a signal receiving link; the signal receiving link comprises a receiving end mixer, an analog-to-digital converter and a digital signal processing module; wherein the receiving end mixer can be configured to down-convert the received echo signal based on the received receiving end local oscillator signal to obtain an analog intermediate frequency signal, the analog-to-digital converter can be configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal, and the digital signal processing module can be configured to process the digital intermediate frequency signal to obtain a target parameter; the echo signal is a signal formed when the signal transmitted by the signal transmission link is reflected and / or scattered by a target object.

[0039] Exemplarily, the receiving mixer is a real mixer, and the analog-to-digital converter is a real analog-to-digital converter; or, the receiving mixer is an orthogonal mixer, and the analog-to-digital converter is an orthogonal analog-to-digital converter.

[0040] Exemplarily, the receiving end local oscillator signal is a swept frequency signal; or, the receiving end local oscillator signal is a single tone signal.

[0041] An embodiment of the present disclosure also provides a signal calibration link, comprising a signal transceiver link as described in any embodiment of the present disclosure; the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link, and the signal receiving link can be configured to calibrate the signal transmitting link.

[0042] Exemplarily, there is a preset frequency difference between the local oscillator signal of the signal receiving link and the local oscillator signal of the signal transmitting link.

[0043] Exemplarily, the signal calibration link also includes a BIST module, which is arranged between the local oscillator signal source and the receiving end mixer; wherein, the BIST template can be configured to mix the received local oscillator signal based on a preset frequency deviation signal so that there is a preset difference frequency between the local oscillator signal received by the receiving end mixer and the local oscillator signal of the signal transmission link.

[0044] An embodiment of the present disclosure also provides a signal calibration link, comprising a signal transceiver link as described in any embodiment of the present disclosure, and a BIST module; the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link through the BIST module, and the signal receiving link can be configured to calibrate the signal transmitting link.

[0045] An embodiment of the present disclosure also provides a signal calibration link, comprising two signal receiving links, a BIST module, an auxiliary circuit unit and a signal transmitting link as described in any embodiment of the present disclosure, and; any of the signal receiving links comprises a real mixer, a real analog-to-digital converter and a digital signal processing module; the real mixer can be configured to down-convert the received echo signal based on the received local oscillator signal to obtain an analog intermediate frequency signal, the real analog-to-digital converter can be configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal, and the digital signal processing module can be configured to process the digital intermediate frequency signal to obtain a target parameter; the echo signal is a signal formed by the signal transmitted by the signal transmitting link being reflected and / or scattered by the target object; the receiving antenna connection ports of the two signal receiving links are respectively connected to the transmitting antenna connection port of the signal transmitting link through the auxiliary circuit unit and the BIST module in sequence, and the signal receiving link can be configured to calibrate the intermediate frequency part of the signal transmitting link.

[0046] An embodiment of the present disclosure also provides a signal calibration link for a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to a compensation coefficient, so as to achieve target detection and / or communication, wherein: the signal calibration link is configured to obtain current observation information of the main signal transmission path under the current compensation coefficient; and when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition.

[0047] Exemplarily, the compensation coefficient includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.

[0048] Exemplarily, the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.

[0049] Exemplarily, the integrated circuit is a millimeter wave radar chip, and / or the radio frequency transmission signal is an FMCW signal.

[0050] Exemplarily, the current compensation coefficient includes the initial compensation coefficient h(0), the first compensation coefficient h(1),…, the k-1th compensation coefficient h(k-1) and the k-1th compensation coefficient h(k), in sequence, where the k-1th compensation coefficient h(k) is determined based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1); where k is an integer greater than or equal to 2.

[0051] Exemplarily, when the difference value between the initial observation information O(0) and the first observation information O(1) is greater than a preset difference threshold, the kth compensation coefficient h(k) is determined based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1); when the difference value between the initial observation information O(0) and the first observation information O(1) is less than the preset difference threshold, the initial observation information O(0) is adjusted according to a preset phase adjustment amount to obtain the adjusted first compensation coefficient h(1), and then the new first observation information O(1) is obtained again based on the adjusted first compensation coefficient h(1), and the kth compensation coefficient h(k) is determined based on the new first observation information O(1); wherein, the difference value between the initial observation information O(0) and the new first observation information O(1) is greater than the said difference threshold.

[0052] Exemplarily, the method of determining the kth compensation coefficient h(k) based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1) includes: when the difference between the absolute value of the initial observation information O(0) and the absolute value of the first observation information O(1) is greater than a preset difference threshold, iteratively calculating the sum of the k-1th compensation coefficient h(k-1) and the k-1 observation information O(k-1) based on the first compensation coefficient h(1) being updated to the initial observation value O(0) to determine the kth compensation coefficient h(k); when the difference between the absolute value of the first observation information O(1) and the absolute value of the initial observation information O(0) is greater than a preset difference threshold, iteratively calculating the difference between the k-1th compensation coefficient h(k-1) and the k-1 observation information O(k-1) based on the first compensation coefficient h(1) being updated to the inverse of the initial observation value O(0) to determine the kth compensation coefficient h(k).

[0053] Exemplarily, the kth compensation coefficient h(k) is determined based on the new first observation information O(1), including: when the difference between the absolute value of the initial observation information O(0) and the absolute value of the new first observation information O(1) is greater than the difference threshold, iteratively calculating the sum of the k-1th observation information O(k-1) and the adjusted k-1th observation information O(k-1) obtained after the k-1th observation information O(k-1) is processed by the phase adjustment amount, to determine the kth compensation coefficient h(k); when the difference between the absolute value of the new first observation information O(1) and the absolute value of the initial observation information O(0) is greater than the difference threshold, updating the first compensation coefficient h(1) to the inverse of the adjusted first compensation coefficient h(1), iteratively calculating the difference between the k-1th compensation coefficient h(k-1) and the adjusted k-1th observation information O(k-1) obtained after the k-1th observation information O(k-1) is processed by the phase adjustment amount, to determine the kth compensation coefficient h(k).

[0054] Exemplarily, the first compensation coefficient h(1) is determined based on initial observation information O(0).

[0055] Exemplarily, the value of the initial compensation coefficient h(0) is 0.

[0056] Exemplarily, the main signal transmission path includes a baseband module and a radio frequency module, wherein the baseband module is used to generate a baseband signal; the radio frequency module is used to frequency shift and / or phase shift the baseband signal using a transmitting end local oscillator signal to form a radio frequency transmission signal; wherein the observation information is obtained from the radio frequency module.

[0057] Exemplarily, the RF module includes a transmitting mixer and a power amplifier; wherein the transmitting mixer uses a transmitting local oscillator signal to frequency shift and / or phase shift the baseband signal; the power amplifier is used to perform power amplification processing on the signal output by the transmitting mixer; wherein the observation information is obtained from at least one of the signal output end of the transmitting mixer, the signal output end of the power amplifier, and the signal output end of the voltage-current converter in the transmitting mixer.

[0058] The embodiment of the present disclosure also provides a signal calibration link for a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to a compensation coefficient, so as to achieve target detection and / or communication, wherein: the signal calibration link is configured to determine the initial observation information O(0), the first observation information O(1) and the second observation information O(2) corresponding to the main signal transmission path under the conditions of different values ​​of the initial compensation coefficient h(0), the first compensation coefficient h(1) and the second compensation coefficient h(2); and to determine the third compensation coefficient h(3) using the initial observation information O(0), the first observation information O(1) and the second observation information O(2) as the compensation coefficient used in the compensation operation of the signal transmission link.

[0059] Exemplarily, the compensation coefficient includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.

[0060] Exemplarily, the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.

[0061] Exemplarily, the integrated circuit is a millimeter wave radar chip, and / or the radio frequency transmission signal is an FMCW signal.

[0062] Exemplarily, the method of determining the third compensation coefficient h(3) by using the initial observation information O(0), the first observation information O(1) and the second observation information O(2) includes: normalizing the first observation information O(1) and the second observation information O(2) respectively using the initial observation information O(0); and determining the third compensation coefficient h(3) using the result of the normalization processing on the condition that the third observation information O(3) is 0.

[0063] Exemplarily, the method of determining the third compensation coefficient h(3) based on the result of normalization processing, conditional on the third observation information O(3) being 0, includes: determining the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2, conditional on the third observation information O(3) being 0; wherein the first ratio d1 is the ratio between the first difference and the initial observation information O(0), and the first difference is the difference between the initial observation information O(0) and the first observation information O(1); the second ratio d2 is the ratio between the second difference and the initial observation information O(0), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2); calculating the product between the first coefficient x1 and the first compensation coefficient h(1) to obtain a first multiplication result; and calculating the product between the second coefficient x2 and the second compensation coefficient h(2) to obtain a second multiplication result; and determining the third compensation coefficient h(3) based on the first multiplication result and the second multiplication result.

[0064] Exemplarily, the method of determining the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2 includes: constructing an observation equation to determine the first coefficient x1 and the second coefficient x2 in the observation equation; wherein the calculation expression of the observation equation is the product of the inverse matrix of the 2*2 matrix and the first matrix of 1 row and 2 columns to obtain a second matrix of 1 row and 2 columns; wherein the first row of the 2*2 matrix records the numerical value of the real part and the numerical value of the imaginary part of the first ratio d1, and the second row records the numerical value of the real part and the numerical value of the imaginary part of the second ratio d2; the value of the first row of the first matrix is ​​1, and the value of the second row is 0; the first row of the second matrix is ​​the value of the first coefficient x1, and the second row is the value of the second coefficient x2.

[0065] Exemplarily, determining the third compensation coefficient h(3) based on the first multiplication result and the second multiplication result includes: calculating the sum of the first multiplication result and the second multiplication result to obtain the third compensation coefficient h(3).

[0066] Exemplarily, the first compensation coefficient h(1) and the second compensation coefficient h(2) are determined according to initial observation information O(0). Exemplarily, the value of the initial compensation coefficient h(0) is 0.

[0067] Exemplarily, the main signal transmission path includes a baseband module and a radio frequency module, wherein the baseband module is used to generate a baseband signal; the radio frequency module is used to frequency shift and / or phase shift the baseband signal using a transmitting end local oscillator signal to form a radio frequency transmission signal; wherein the observation information is obtained from the radio frequency module.

[0068] Exemplarily, the RF module includes a transmitting mixer and a power amplifier; wherein the transmitting mixer uses a transmitting local oscillator signal to frequency shift and / or phase shift the baseband signal; the power amplifier is used to perform power amplification processing on the signal output by the transmitting mixer; wherein the observation information is obtained from at least one of the signal output end of the transmitting mixer, the signal output end of the power amplifier, and the signal output end of the voltage-current converter in the transmitting mixer.

[0069] An embodiment of the present disclosure also provides a signal compensation link, comprising a signal transmission link as described in any embodiment of the present disclosure, and a compensation unit, wherein the compensation unit can be configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion defects of the signal transmission link.

[0070] Exemplarily, the compensation unit may be configured to compensate the signal transmission link based on calibration data obtained from the signal calibration link as described in any embodiment of the present disclosure.

[0071] Exemplarily, the signal receiving link in the signal calibration link is integrated as an auxiliary calibration module in an area near the transmitting link to be calibrated, so as to perform real-time calibration operations during intervals when the transmitting link is operating.

[0072] An embodiment of the present disclosure also provides a method for compensating for unequal feeder lengths, which is applied to an antenna array of an electromagnetic wave sensor having at least two signal links. The method includes: taking the one with the shortest feeder line among the at least two signal links as a reference link, obtaining the delay difference of the remaining transmission links relative to the reference link; and compensating for the unequal feeder lengths of the antenna array in the digital domain based on the delay difference.

[0073] Exemplarily, the unequal feeder lengths are between different transmitting links or between different receiving links, the feeder lengths between the local oscillator LO and the mixers in each transmitting channel are unequal, and / or the feeder lengths between the PA of each transmitting channel and its transmitting antenna are unequal.

[0074] An embodiment of the present disclosure also provides a signal calibration system for use in an electromagnetic wave sensor. The signal calibration system includes a signal transmission link and an auxiliary link. The auxiliary link is integrated into the electromagnetic wave sensor adjacent to the signal transmission link. The auxiliary link can be configured to perform real-time calibration on the signal transmission link.

[0075] Exemplarily, the signal transmission link includes a signal transmission link and / or a signal reception link; the auxiliary link corresponding to the signal transmission link includes an auxiliary reception link and / or an auxiliary signal transmission link; wherein the auxiliary reception link can be configured to calibrate the transmission link, and the auxiliary transmission link can be configured to calibrate the reception link.

[0076] Exemplarily, the signal receiving link includes a radio frequency part and a baseband part, and the auxiliary transmission link corresponding to the signal receiving link includes a radio frequency auxiliary transmission unit and a baseband auxiliary transmission unit; and the radio frequency auxiliary transmission unit can be configured to calibrate the radio frequency part, and the baseband auxiliary transmission unit can be configured to calibrate the baseband part.

[0077] Exemplarily, the RF auxiliary transmitting unit includes an IQ device; and wherein, when using the auxiliary transmitting link to calibrate the signal receiving link, the baseband part is first calibrated using the baseband auxiliary transmitting unit, and then the calibrated baseband part is used to calibrate the RF auxiliary transmitting unit, and then the calibrated RF auxiliary transmitting unit is used to calibrate the RF part.

[0078] Exemplarily, the auxiliary receiving chain includes an auxiliary receiving unit and a calibration receiving unit, and the auxiliary receiving unit includes an IQ device; wherein the calibration receiving unit can be configured to calibrate the auxiliary receiving unit, and the calibrated auxiliary receiving unit can be configured to calibrate the transmitting chain.

[0079] An embodiment of the present disclosure also provides an IQ mixer, which includes an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit, wherein the I-branch mixing unit can be configured to output an I-branch signal, the Q-branch mixing unit can be configured to output a Q-branch signal, and the transformer unit can be configured to magnetically couple the I-branch signal and the Q-branch signal to synthesize the IQ mixing output signal.

[0080] Exemplarily, the transformer unit is a three-turn transformer structure, which includes two branch inductors and one magnetic coupling inductor; wherein, a branch inductor is connected in series between the output ends of the I branch mixing unit and the output ends of the Q branch mixing unit respectively, and the magnetic coupling inductor is arranged between the two branch inductors.

[0081] Exemplarily, the IQ mixer is applied to a digital phase shifter in a signal transmission link described in any embodiment of the present disclosure, and / or, to a signal calibration link described in any embodiment of the present disclosure, and / or, to a signal calibration system described in any embodiment of the present disclosure.

[0082] An embodiment of the present disclosure also provides an integrated circuit, comprising a radio frequency module, an analog signal processing module and a digital signal processing module connected in sequence; the radio frequency module is used to generate a radio frequency transmission signal and receive a radio frequency reception signal; the analog signal processing module is used to down-convert the radio frequency reception signal to obtain an intermediate frequency signal; the digital signal processing module is used to perform analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal; and the radio frequency module includes the signal transmission link described in any embodiment of the present disclosure, the signal transceiver link described in any embodiment of the present disclosure, the signal calibration link described in any embodiment of the present disclosure, the signal compensation link described in any embodiment of the present disclosure, the signal calibration system described in any embodiment of the present disclosure, and / or the IQ mixer described in any embodiment of the present disclosure; and / or the digital signal processing module performs compensation in the digital domain based on the feeder unequal length compensation method described in any embodiment of the present disclosure.

[0083] Exemplarily, the integrated circuit further includes a data processing module, which is used to process the digital signal to achieve target detection and / or wireless communication. Exemplarily, the integrated circuit is a millimeter wave or UWB chip.

[0084] Exemplarily, the radio frequency receiving signal is an echo signal formed when the radio frequency transmitting signal is transmitted and / or scattered by a target, and the integrated circuit is a sensor chip.

[0085] An embodiment of the present disclosure further provides an electromagnetic wave sensor, comprising: a carrier; the integrated circuit described in any embodiment of the present disclosure, disposed on the carrier; an antenna, disposed on the carrier, or the antenna and the integrated circuit are integrated into a single device and disposed on the carrier; wherein the integrated circuit is connected to the antenna for transmitting the radio frequency transmit signal and / or receiving the radio frequency receive signal.

[0086] An embodiment of the present disclosure further provides a device, comprising: a device body; and an electromagnetic wave sensor as described in any embodiment of the present disclosure, which is arranged on the device body; wherein the electromagnetic wave sensor is used for target detection and / or communication to provide reference information for the operation of the device body.

[0087] An embodiment of the present disclosure further provides a non-transitory computer-readable storage medium having computer-readable instructions stored thereon. When the instructions are executed by a processor, the processor executes the feeder unequal length compensation method described in any embodiment of the present disclosure.

[0088] Summary of the Figures

[0089] The above and other objects, features and advantages of the present disclosure will become more apparent through the following description of the embodiments of the present disclosure with reference to the accompanying drawings.

[0090] FIG1A is a simplified schematic diagram of a signal transmission link of an analog phase shifter architecture;

[0091] FIG1B is a simplified schematic diagram of an analog phase shifter in the signal transmission link shown in FIG1A ;

[0092] FIG2A is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture according to an embodiment of the present disclosure;

[0093] FIG2B is a simplified schematic diagram of another signal transmission link of the digital phase shifter architecture according to an embodiment of the present disclosure;

[0094] FIG2C is a waveform diagram of an FMCW transmission signal and an echo signal using sawtooth wave modulation;

[0095] FIG3A is a schematic diagram of a signal model of a transmitter quadrature modulation when quadrature imbalance occurs;

[0096] FIG3B is a schematic diagram of a signal generated by a real mixer;

[0097] FIG3C is a schematic diagram of third-order harmonic distortion;

[0098] FIG4A is a schematic structural diagram of a signal transmission link provided by an embodiment of the present disclosure;

[0099] FIG4B is a schematic diagram of quadrature imbalance compensation provided by an embodiment of the present disclosure;

[0100] FIG4C is a first schematic diagram of harmonic distortion compensation provided by an embodiment of the present disclosure;

[0101] FIG4D is a second schematic diagram of harmonic distortion compensation provided by an embodiment of the present disclosure;

[0102] FIG5A is a schematic diagram of a digital phase shifter architecture in a signal transmission link according to an embodiment of the present disclosure;

[0103] FIG5B is a schematic diagram of a transmission link including a specific compensation unit according to an embodiment of the present disclosure;

[0104] FIG6A is a schematic structural diagram of a signal transceiver link provided by an embodiment of the present disclosure;

[0105] FIG6B is a schematic structural diagram of another signal transceiver link provided by an exemplary embodiment of the present disclosure;

[0106] FIG7A is a schematic diagram of a transceiver link including a TX IQ Mod, a BIST IQ Mod, and an RX Real Mixer according to an embodiment of the present disclosure;

[0107] FIG7B is a schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod, and LO Freq Diff according to an embodiment of the present disclosure;

[0108] FIG8 is a schematic diagram of a transceiver link based on the structure shown in FIG7B in combination with BIST according to an embodiment of the present disclosure;

[0109] FIG9 is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod, and RX IQ De-Mod according to an embodiment of the present disclosure;

[0110] FIG10 is a schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod according to an embodiment of the present disclosure;

[0111] FIG11 is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod according to an embodiment of the present disclosure;

[0112] FIG12A is a schematic structural diagram of a mixer according to an embodiment of the present disclosure;

[0113] FIG12B is a schematic diagram of the structure of a compensation unit in a transmitter according to an embodiment of the present disclosure;

[0114] FIG13A is a schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module according to an embodiment of the present disclosure;

[0115] FIG13B is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency-doubling waveform generator module according to an embodiment of the present disclosure;

[0116] FIG13C is a schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture according to an embodiment of the present disclosure;

[0117] FIG14A is a schematic diagram of a flow chart of a signal transmission method provided in an embodiment of the present disclosure;

[0118] FIG14B is a schematic diagram illustrating the principle of determining a compensation coefficient according to an embodiment of the present disclosure;

[0119] FIG15 is a schematic diagram of a transmission link having at least two transmission channels according to an embodiment of the present disclosure;

[0120] FIG16 is a schematic structural diagram of a digital LO signal generator according to an embodiment of the present disclosure;

[0121] FIG17 is a schematic structural diagram of a transmission link including a feeder unequal length compensation module according to an embodiment of the present disclosure;

[0122] FIG18 is a schematic diagram of calibrating and compensating a transceiver link using an auxiliary circuit in an embodiment of the present disclosure;

[0123] FIG19 is a schematic diagram of calibrating and compensating a receiving link using an auxiliary transmitting circuit in an embodiment of the present disclosure;

[0124] FIG20 is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present disclosure;

[0125] FIG21 is a schematic structural diagram of an auxiliary circuit according to an embodiment of the present disclosure;

[0126] FIG22 is a schematic structural diagram of another auxiliary circuit according to an embodiment of the present disclosure;

[0127] FIG23 is a schematic diagram of a circuit module of an IQ Mixer according to an embodiment of the present disclosure;

[0128] FIG24 is a schematic diagram of the structure of an IQ Mixer according to an embodiment of the present disclosure;

[0129] FIG25 is a schematic diagram corresponding to the structure shown in FIG24;

[0130] FIG26 is a schematic diagram of the physical structure of another IQ Mixer according to an embodiment of the present disclosure;

[0131] FIG27 is a schematic diagram of a flow chart of another signal transmission method provided in an embodiment of the present disclosure;

[0132] FIG28 is a schematic diagram of a flow chart of another signal transmission method provided in an embodiment of the present disclosure;

[0133] FIG29 is a flow chart of another signal transmission method provided in an embodiment of the present disclosure.

[0134] Details

[0135] To facilitate understanding of the present disclosure, a more comprehensive description of the present disclosure will be provided below with reference to the accompanying drawings. The accompanying drawings illustrate preferred embodiments of the present disclosure. However, the present disclosure may be implemented in various forms and is not limited to the embodiments described herein. Rather, these embodiments are provided to facilitate a more thorough and comprehensive understanding of the present disclosure.

[0136] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as those commonly understood by those skilled in the art in the art of the present disclosure. The terms used in the specification of the present disclosure are only for the purpose of describing specific embodiments and are not intended to limit the present disclosure.

[0137] Hereinafter, the technical solution of the present disclosure will be described in detail with reference to the accompanying drawings.

[0138] FIG1A is a simplified schematic diagram of a signal transmission link of an analog phase shifter architecture, and FIG1B is a simplified schematic diagram of an analog phase shifter in the signal transmission link shown in FIG1A .

[0139] As shown in Figure 1A, when a sensor transmits a signal, for one transmission link, a signal generator 11, such as one comprised of a phase-locked loop (PLL), generates an LO signal (e.g., a swept-frequency signal in the 77 GHz band), which can be, for example, an FMCW signal. An analog phase shifter (Analog PS) 12 performs a phase shift on the received LO signal, which is then radiated via a transmitting antenna 13 into a predetermined spatial region for operations such as target detection and measurement.

[0140] Optionally, in the transmission chain structure shown in FIG1A , the corresponding analog phase shifter architecture may be as shown in FIG1B , and the specific phase shifting principle may be expressed as follows:

[0141] in

[0142] That is, by receiving the LO signal (ie LO IN ) are phase shifted by 0° and 90° respectively to obtain two signals (i.e. LO I and LO Q ), and is amplified by the power amplifier (PA) to form an output signal LO OUT , that is, the output signal LO OUT Compared with LO IN The input signal is phase-shifted by θ, where the value of the phase shift θ is determined by the two signals LO I The amplitude of A and LO Q Determined by the amplitude B.

[0143] In addition, the above analog phase shifter architecture can also be implemented using a delay line unit. That is, by using the narrow-band assumption of the signal, phase shifting is performed through time delay. The principle is shown in the following equation:

[0144] Where τ is the delay time of the delay line.

[0145] Because the aforementioned analog phase shifters have low resolution and accuracy, they cannot meet the requirements of current sensors. Furthermore, although calibration can be used to improve resolution and accuracy, offline calibration of the analog phase shifters is required, significantly increasing the difficulty and complexity of engineering implementation and product mass production. Offline calibration results may not be accurate when operating in different states (temperatures). Furthermore, analog phase shifters also have serious issues such as large area, high loss, stability, and channel coupling. When multiple antenna phase shifters work together, the performance of the phase shifters will affect each other, further deteriorating system performance.

[0146] Based on this, the inventors of this disclosure have creatively proposed a signal transmission link using a digital phase shifter architecture. This design effectively improves phase modulation precision and accuracy while also avoiding the need for offline calibration of the phase shifter and other components in the transmission link, thereby reducing the complexity and difficulty of engineering implementation. Furthermore, this design effectively reduces the area and loss of the phase shifter architecture's transmission link, improving system stability and lowering channel coupling.

[0147] Figure 2A is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture in an embodiment of the present disclosure, Figure 2B is a simplified schematic diagram of another signal transmission link of a digital phase shifter architecture in an embodiment of the present disclosure, and Figure 2C is a waveform diagram of an FMCW transmission signal and an echo signal using sawtooth wave modulation.

[0148] An embodiment of the present disclosure provides a signal transmission link that can be applied to an electromagnetic wave sensor. The transmission link may include an analog signal source and a digital phase shifter. The analog signal source can be configured to provide an initial analog signal (such as an LO signal), and the digital phase shifter can be configured to generate a phase-shifted signal in the digital domain. The digital phase shifter can also phase-shift the initial analog signal based on the generated phase-shifted signal to perform a preset phase shift operation on the initial analog signal.

[0149] As shown in FIG2A , in some optional embodiments, a signal transmission link with a digital phase shifter architecture may include an analog signal source 21, a digital phase shifter (Digital PS) 22, and a transmitting antenna 23. Specifically, the analog signal source 21 may be configured to provide an LO signal, and the digital phase shifter 22 may be configured to perform a predetermined phase shift on the received LO signal, so that the phase-shifted LO signal is radiated into a predetermined spatial region via the transmitting antenna 23. The analog signal source 21 may also include a phase-locked loop (PLL) architecture that provides electromagnetic wave (e.g., laser, microwave, etc.) signals. The analog signal source 21, digital phase shifter 22, and transmitting antenna 23 may be integrated into a single device or may be separate components. For example, the analog signal source 21 and digital phase shifter 22 may be integrated into a package such as a SoC chip, while the transmitting antenna 23 may be connected via a peripheral port of the chip and formed on a substrate such as a PCB. At the same time, in some optional embodiments, the transmitting antenna 23 can also be integrated on the chip package to form AiP or AoP, with a chip structure of a packaged antenna.

[0150] As shown in Figure 2A, the digital phase shifter 22 in the embodiment of the present disclosure may include a mixer (Mixer) 221, a digital-to-analog converter (i.e., DAC) 222 and a phase-shifted signal source (such as a digital baseband signal source Baseband) 223, etc., that is, the phase-shifted signal source 223 can be configured to provide a digital phase-shifted signal; the digital-to-analog converter 222 can be configured to perform analog-to-digital conversion on the received digital phase-shifted signal to convert the digital phase-shifted signal into an analog phase-shifted signal; the mixer 221 can be configured to perform a mixing operation on the received analog phase-shifted signal with the received electromagnetic wave signal from the analog signal source 21 to achieve a phase-shifting operation of setting the above-mentioned electromagnetic wave signal using the digital phase-shifted signal. Optionally, when the above-mentioned signal transmission link provides a swept frequency signal, such as an FMCW laser signal or an FMCW microwave signal, a swept frequency electromagnetic wave signal can be provided based on the analog signal source 21, and / or a swept frequency digital phase-shifted signal can be provided based on the phase-shifted signal source 223, so that after mixing by the mixer 221, a swept frequency continuous wave signal is output.

[0151] In some optional embodiments, based on the structure shown in Figure 2A, the analog signal source 21 can be configured to provide an FMCW signal in the centimeter wave band or millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in the microwave, and the phase-shifted signal source 223 can be configured to provide a digital phase-shifted signal at the MHz level (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog converter 222 performs digital-to-analog conversion on the MHz-level digital phase-shifted signal to obtain an analog phase-shifted signal in the corresponding frequency range, and the mixer 221 can be configured to perform up-mixing or down-mixing operations on the received FMCW signal in the millimeter wave band based on the received analog phase-shifted signal of the fixed frequency band, so as to realize the operation of preset phase shifting of the FMCW signal.

[0152] In some optional embodiments, the centimeter wave signal in the 3.1GHz frequency band may include 3.1GHz to 10.6GHz, such as 3.1GHz, 5GHz, 5GHz, 6GHz, 8GHz, 10.6GHz, etc.; the millimeter wave signal in the 77GHz frequency band may include 76GHz to 81GHz signals, such as 76GHz to 77GHz, 77GHz to 79GHz, 79GHz to 81GHz and other swept frequency signals, or 76GHz, 77GHz, 78GHz, 79GHz, 80GHz, 81GHz and other fixed frequency band signals.

[0153] Based on the structure shown in Figure 2A, since phase-shift signal source 223 provides a digital signal, to further adapt the signal characteristics, mixer 221 can be configured as an IQ mixer and digital-to-analog converter 222 as an IQ DAC. Furthermore, phase-shift signal source 223 can be configured to provide a digital baseband signal source (DDFS) for phase shifting and / or to provide a corresponding source signal as a waveform controller.

[0154] As shown in FIG2B , the embodiment of the present disclosure provides a signal transmission link, which is applied to a radar system. The signal transmission link includes: a baseband digital module 201 (including the aforementioned phase-shift signal source 223), a digital-to-analog converter (DAC), and a baseband digital module 202 (including the aforementioned phase-shift signal source 223). The digital-to-analog converter module 202 comprises a baseband digital module 201 configured to generate two orthogonal digital baseband signals (i.e., digital phase-shifted signals) and send the two orthogonal digital baseband signals to a respective digital-to-analog converter module 202. The digital-to-analog converter module 202 is configured to convert the two orthogonal digital baseband signals into two analog baseband signals (i.e., analog phase-shifted signals). The local oscillator 203 is configured to provide a local oscillator signal TX_LO (i.e., an initial analog signal). The quadrature modulator 204 is configured to perform frequency shifting and phase shifting on the local oscillator signal TX_LO based on the two analog baseband signals to form a predetermined phase-shifted FMCW RF transmit signal.

[0155] In the disclosed embodiment, the transmitting digital baseband signal provided by the transmitting baseband digital module 201 includes preset phase information; the digital-to-analog conversion module 202 performs digital-to-analog conversion on the received transmitting digital baseband signal to convert the transmitting digital baseband signal into a transmitting analog baseband signal (e.g., without changing the phase information); the transmitting orthogonal modulator 204 mixes the received transmitting analog baseband signal with the transmitting local oscillator signal TX_LO generated by the transmitting local oscillator 203 to achieve frequency shifting of the transmitting local oscillator signal based on the transmitting analog baseband signal while performing a preset phase shift operation to form an FMCW RF transmission signal after a predetermined phase shift.

[0156] The signal transmission link of the disclosed embodiment utilizes a transmitting baseband digital module 201, a digital-to-analog conversion module 202, and a transmitting quadrature modulator 204 to form a digital phase shifter architecture. Because the baseband signal of this architecture is generated in the digital domain, it has better orthogonality and lower sidelobes. Therefore, its phase shift can be generated very accurately, resulting in higher phase modulation accuracy. This implements a vehicle-mounted radar system with a high-precision digital phase shifting function, reduces the isolation requirements between antennas, and has the advantages of low link loss, low cost, and no need for offline calibration. It can also support more flexible transmission schemes, such as high-performance Doppler multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0157] In the embodiment of the present disclosure, since the transmitting baseband digital module 201 provides a digital signal, in order to further adapt the signal characteristics, the transmitting modulator is set as an orthogonal modulator (IQ Modulator) and the digital-to-analog conversion module 202 is set as an orthogonal digital-to-analog converter (IQ DAC).

[0158] In the embodiment of the present disclosure, the transmitting local oscillator 203 may include a phase-locked loop (PLL) structure, and may provide electromagnetic wave (eg, laser, microwave, etc.) signals.

[0159] In some exemplary embodiments, the signal transmission link further includes a power amplifier (PA) 205 , wherein the power amplifier 205 is configured to amplify the power of the phase-shifted RF signal and output the amplified signal to the transmitting antenna.

[0160] In some exemplary embodiments, the signal transmission link further includes a transmitting antenna 206 , wherein the transmitting antenna 206 is configured to radiate the amplified signal to a preset spatial region.

[0161] In the disclosed embodiment, the signal amplified by the power amplifier 205 can be radiated into a predetermined spatial region via a transmit antenna 206, which may be integrated into the package or externally mounted. Specifically, the transmitting local oscillator 203, digital phase shifter, and transmit antenna 206 can be integrated into a single device or can be separate components. For example, the transmitting local oscillator 203 and digital phase shifter can be integrated into a package, such as a SoC chip, while the transmit antenna 206 can be connected via a peripheral port of the chip and mounted on a substrate, such as a PCB. Furthermore, in some optional embodiments, the transmit antenna 206 can be integrated into the chip package to form an AiP or AoP, resulting in a chip-on-package antenna structure.

[0162] In some exemplary embodiments, the bandwidth of the frequency sweep signal is above 2 GHz. For example, the electromagnetic wave of the transmission signal emitted by the transmitting antenna of the frequency modulated continuous wave radar system is a high-frequency frequency modulated continuous wave, and the echo signal received by the receiving antenna of the frequency modulated continuous wave radar system is the electromagnetic wave reflected / scattered back by the target object. Figure 2C shows a waveform diagram of an exemplary FMCW transmission signal and echo signal. As shown in Figure 2C, the frequencies of the transmission signal and the echo signal change regularly over time. The frequency modulated continuous wave is generally sawtooth-shaped, triangular, etc. The present disclosure uses the sawtooth shape as an example for explanation. The electromagnetic wave within each frequency modulation period T is called a chirp, and the frequency of the signal of each chirp increases linearly with time. In the embodiment of the present disclosure, the bandwidth range B of a chirp is greater than or equal to 2 GHz.

[0163] In some exemplary embodiments, the transmitting digital baseband signal is a single-tone signal, and the transmitting local oscillator signal is a swept-frequency signal. In the disclosed embodiment, the transmitting local oscillator 203 can be configured to provide an FMCW signal in a centimeter-wave frequency band or a millimeter-wave frequency band (e.g., 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in the microwave frequency band. The transmitting baseband digital module 201 can be configured to provide a MHz-level (e.g., 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.) single-tone transmitting digital baseband signal. That is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz-level single-tone transmitting digital baseband signal to obtain a single-tone transmitting analog baseband signal in a corresponding frequency range. The transmitting quadrature modulator 204 can be configured to perform an up-mixing or down-mixing operation on the received millimeter-wave frequency band FMCW signal based on the received single-tone transmitting analog baseband signal to achieve a preset phase shift operation on the FMCW signal.

[0164] For example, the FMCW signal in the 3.1 GHz frequency band may include a swept frequency signal between 3.1 GHz and 10.6 GHz, such as 7.163-8.812 GHz; the FMCW signal in the 77 GHz frequency band may include a swept frequency signal between 76 GHz and 81 GHz, or swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, or 79 GHz to 81 GHz.

[0165] In other exemplary embodiments, the digital baseband signal at the transmitting end may be a swept frequency signal, and the local oscillator signal at the transmitting end may be a single-tone signal.

[0166] In the embodiment of the present disclosure, the transmitting local oscillator 203 can be configured to provide a single-tone transmitting local oscillator signal in a centimeter-wave frequency band or a millimeter-wave frequency band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the transmitting baseband digital module 201 can be configured to provide a MHz (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.) level transmitting digital baseband FMCW signal, that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz-level transmitting digital baseband FMCW signal to obtain a transmitting analog baseband FMCW signal in a corresponding frequency range, and the transmitting orthogonal modulator 204 can be configured to perform up-mixing or down-mixing operations on the received single-tone transmitting local oscillator signal in the centimeter-wave frequency band or the millimeter-wave frequency band based on the received transmitting analog baseband FMCW signal, so as to realize preset phase shifting and frequency sweeping operations on the single-tone transmitting local oscillator signal.

[0167] For example, the single-tone transmitting local oscillator signal in the 3.1 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 3.1 GHz, 5 GHz, 6 GHz, 8 GHz, or 10.6 GHz; the single-tone transmitting local oscillator signal in the 77 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, or 81 GHz.

[0168] In some exemplary embodiments, the signal transmission link further includes: a low-pass filter (LPF) 207, which is disposed between the digital-to-analog conversion module 202 and the transmitting-end quadrature modulator 204 and is configured to perform low-pass filtering on the transmitting-end analog baseband signal output by the digital-to-analog conversion module 202 and output the signal to the transmitting-end quadrature modulator 204.

[0169] As shown in Figure 2B , the transmitting baseband digital module 201 generates two orthogonal digital baseband signals, namely, an I-channel digital baseband signal and a Q-channel digital baseband signal. These digital baseband signals are then fed into the digital-to-analog conversion module 202 (which includes two identical DACs, namely, IQ DACs) to produce two analog baseband signals. These two analog baseband signals are then fed into a low-pass filter 207 to remove out-of-band noise. These signals are then orthogonally modulated by the transmitting quadrature modulator 204 to produce modulated RF signals. These modulated RF signals are then radiated through a power amplifier 205 and a transmitting antenna 206.

[0170] In some exemplary embodiments, the signal transmission link may further include a direct digital frequency synthesizer (DDFS) (not shown in FIG2B ), which is disposed between the transmitting baseband digital module 201 and the digital-to-analog conversion module 202. The direct digital frequency synthesizer may be configured to implement at least one of a variety of signal waveforms and transmission modes, such as CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), and Digital IF (Digital Intermediate Frequency), based on the received source signal, so as to achieve flexible configuration of the signal transmission form and the transmission waveform.

[0171] Figure 3A is a schematic diagram of the signal model of the transmitter's quadrature modulation when the quadrature is unbalanced. As shown in Figure 3A, DSP is a digital signal processor, DAC is a digital-to-analog converter, LPF is a low-pass filter, and LO is a local oscillator signal source. I (f) is the frequency domain expression of the I-channel LPF output signal, E I (f) is the baseband frequency response of channel I, A′ I (f) is the frequency domain expression of the baseband output signal of channel I; S I (f) is the frequency domain expression of the output signal of the I channel modulator, G I (f) is the RF frequency response of channel I, S′ I (f) is the frequency domain expression of the RF output signal of channel I; where α I is the amplitude gain of the I channel mixer, is the phase response of the I channel mixer; A Q (f) is the frequency domain expression of the Q-channel LPF output signal, E Q (f) is the Q channel baseband frequency response, A′ Q (f) is the frequency domain expression of the Q channel baseband output signal; S Q (f) is the frequency domain expression of the Q-channel modulator output signal, G Q (f) is the Q channel RF frequency response, S′ Q (f) is the frequency domain expression of the Q channel RF output signal; where α Q is the amplitude gain of the Q-channel mixer, is the phase response of the Q-channel mixer; RF (f) is the frequency domain expression of the transmitted modulated signal. In the embodiment of the present disclosure, the RF includes all circuits from the mixer to the RF signal, and the baseband includes all circuits from the DAC to the mixer.

[0172] According to the signal model shown in FIG3A , the output U of the quadrature demodulator is RF (f) can be expressed as:

[0173] 2U RF (f) = A(ff m )V1(ff m )+A * (-ff m )V1 * (-ff m )+A * (-f+f m )V2(ff m )+A(f+f m )V2 * (-ff m )

[0174] Among them, A(ff m )V1(ff m ) is the frequency f m nearby desired signals;

[0175] A * (-ff m )V1 * (-ff m ) is the frequency -f m nearby desired signals;

[0176] A * (-f+f m )V2(ff m ) is the frequency f m nearby mirror signals;

[0177] A(f+f m )V2 * (-ff m ) is the frequency -f m nearby mirror signals;

[0178] use Indicates U RF (f) The complex baseband form of the signal, i.e. Among them, when f<0, u(f)=0; when f≥0, u(f)=1.

[0179] Therefore, the baseband equivalent model of the quadrature regulator can be: Where A(f)V1(f) is the frequency f m Nearby desired signal; A * (-f)V2(f) is the frequency at -f m The relative image ratio (RIR) is used to measure the ability of the quadrature mixer to suppress unwanted image frequency components. Therefore, the definition of RIR is the ratio of the power of the desired signal and the image signal components, so it can be written as The above equations indicate that the cascaded gain and phase response of the I and Q channels need to be equal (the cascaded gain and phase response of each channel should be considered), otherwise the channels will be unbalanced and the image components will not be fully suppressed.

[0180] Figure 3B is a schematic diagram of the signal generated by a real mixer. As shown in Figure 3B, F IN represents the mixer input signal, F OUT Represents the output signal of the mixer, F LORepresents the local oscillator signal used by the mixer, where the input signal of the mixer is: frequency F BB The baseband signal used by the mixer is: the frequency is F LO The output signal of the mixer is: the frequency is F LO ±F BB signal.

[0181] An ideal mixer will produce an output that is the product of its two inputs. In terms of frequency, the output should be F LO +F BB and F LO -F BB , without other terms. If any input is not driven, there will be no output. However, in F BB and F LO , a real mixer will also generate some energy. BB The energy generated at F is negligible because it is far away from the desired output and will be filtered out by the RF components after the mixer output. BB How is the energy generated at F? LO Energy generated anywhere in the LO can be a problem. It's very close to or within the desired output signal and difficult or impossible to remove through filtering, which would also remove the desired signal. LO can also leak to the system output via other pathways, such as through the power supply or across the silicon itself. Regardless of how the LO leaks, it's referred to as LO leakage (LOL).

[0182] Although analog (RF) circuits can be approximated by linear models representing small signals, the nonlinearity of devices often produces some interesting and important phenomena that are unpredictable. The nonlinear input and output characteristics of memoryless systems can be approximated as: y(t) = α1x(t) + α2x 2 (t)+α3x 3 (t)+...≈α1x(t)+α2x 2 (t)+α3x 3 (t)

[0183] Many of the nonlinear effects of these devices are caused by third-order terms. and So

[0184] in:

[0185] If the system has odd symmetry, that is, the system is completely differential, then the even harmonic terms can cancel each other out, and what remains are the odd harmonic terms, which is also called harmonic distortion.

[0186] Figure 3C is a schematic diagram of third-order harmonic distortion. As shown in Figure 3C, the baseband processor generates a baseband signal at frequency BB. After processing by the IQ modulator, a desired signal Wanted, an image signal IMG, and a third-order harmonic HD3 are obtained. The power amplifier PA outputs the desired signal Wanted, the image signal IMG, and the third-order harmonic HD3.

[0187] The primary source of HD3 is third-order nonlinearity in baseband analog components. As shown in the above derivation, the third-order harmonic component HD3 of the baseband BB signal is located at -3BB. After passing through the mixer, the baseband HD3 is upconverted to LO-3BB, or C-IM3. After PA amplification, the resulting modulated signal is difficult to distinguish at high frequencies.

[0188] Based on the above description, it can be seen that the signal transmission link for digital quadrature modulation needs to deal with problems such as IQ imbalance, harmonic distortion, and LO leakage. After the ideal single-tone baseband signal passes through the orthogonal up-mixer, it will generate multiple non-ideal single-tone signals with different frequencies at high frequencies. Therefore, from a system architecture perspective, the present disclosure utilizes a digital pre-compensation module at the transmitter to address the problems introduced by the non-ideal characteristics of analog devices in the digital quadrature modulation radar TX. When designing the system architecture, the present disclosure not only includes a pre-compensation module but also a calibration loop to achieve real-time calibration and compensation.

[0189] Figure 4A is a schematic diagram of the structure of a signal transmission link according to an embodiment of the present disclosure. As shown in Figure 4A, the signal transmission link includes a main signal transmission path and a signal calibration path integrated into the same integrated circuit. The signal calibration path is configured to calibrate the main signal transmission path to obtain compensation information, and the main signal transmission path is configured to generate a radio frequency transmission signal after performing a compensation operation based on the compensation information, thereby enabling target detection and / or communication.

[0190] Since the signal calibration link is integrated into the integrated circuit including the main transmission path, the signal calibration link can perform calibration operations on the main transmission path in real time, and the calibration operation of the calibration link can be unaffected by changes in the operating environment of the main transmission path, so that the main transmission path can obtain more accurate calibration information, thereby improving the signal processing performance of the main transmission path.

[0191] Furthermore, the compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and a quadrature imbalance compensation parameter. Because the compensation information generated by the signal calibration link can address quadrature imbalance, LO leakage, and harmonic distortion issues, it effectively improves signal quality in the main transmit path of the digital phase shifter architecture.

[0192] In one implementation, the integrated circuit is a millimeter wave radar chip or an ultra-wideband (UWB) chip, and / or the radio frequency transmission signal is an FMCW signal.

[0193] The main signal transmission path may include a first signal source and a phase shifter; wherein the first signal source is configured to generate a first analog signal; and the phase shifter is configured to frequency shift and / or phase shift the first analog signal to form a radio frequency transmission signal.

[0194] When the phase shifter is a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting-end mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting-end mixer is configured to mix the first analog signal and the second analog signal to form the RF transmit signal; when the phase shifter is an orthogonal architecture, the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting-end mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting-end mixer is configured to perform frequency shifting and / or phase shifting on the first analog signal based on the second analog signal to form the RF transmit signal.

[0195] Among them, the main transmission path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter. The compensation circuit is used to merge the compensation signal and the signal output by the second signal source and then output it.

[0196] When the signal output by the main signal transmission path is two orthogonal signals, the compensation signal used by the compensation circuit is two orthogonal signals; when the signal output by the main signal transmission path is not two orthogonal signals, the compensation signal used by the compensation circuit is a signal of the same type as the signal output by the second signal source, where the signal type is a digital signal or an analog signal.

[0197] In which, the compensation circuit includes a compensation signal generator and an adder, the compensation signal generator can be configured to generate the compensation signal; the adder is connected to the compensation signal generator and the second signal source, and is used to perform a signal superposition operation on the signal output by the second signal source and the compensation signal output by the compensation signal generator.

[0198] Specifically, when the signal output by the main signal transmission path is a two-way orthogonal signal, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal imbalance compensation signal unit, and a local oscillator leakage compensation signal unit; when the signal output by the main signal transmission path is not a two-way orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leakage compensation signal unit; wherein, the compensation signal generated by the harmonic compensation signal unit is used to eliminate the harmonic signal of the main frequency signal in the main signal transmission path; the compensation signal of the local oscillator leakage compensation signal unit is used to compensate for the leakage signal generated by the transmitting end local oscillator signal in the main signal transmission path; and the compensation signal generated by the orthogonal imbalance compensation signal unit is used to compensate for the mirror signal of the main frequency signal in the main signal transmission path.

[0199] The compensation signal used by the quadrature imbalance compensation circuit has the same frequency, the same amplitude, and the opposite phase as the image signal corresponding to the main signal transmission path and the image signal corresponding to the desired signal generated by the main signal transmission path.

[0200] In an exemplary embodiment, the compensation signal used by the quadrature imbalance compensation circuit is determined based on a signal output by a second signal source and a frequency-inverted complex conjugate signal of the signal output by the second signal source. The compensation signal generated by the quadrature imbalance compensation circuit is obtained by: multiplying a preset pre-compensation coefficient by the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal; and calculating the difference between the signal output by the second signal source and the adjustment signal to obtain the compensation signal generated by the quadrature imbalance compensation circuit. The pre-compensation coefficient is determined based on the ratio of the amplitude of the desired signal to the amplitude of the mirror signal corresponding to the desired signal.

[0201] FIG4B is a schematic diagram of quadrature imbalance compensation provided by an embodiment of the present disclosure. As shown in FIG4B , in order to achieve compensation of the TX quadrature modulator, that is, to suppress the presence of unwanted image signals, a pre-compensation method can be used. Let h IQ is the pre-compensation coefficient, acting on the complex conjugate signal A of the baseband signal to be transmitted * (-f). The frequency domain expression of the pre-compensated baseband signal can be written as A pc (f)=A(f)-h IQ ·A * (-f)→apc (n) = α(n) - h IQ ·a * (n)

[0202] When this pre-compensated signal is now used as the input to the quadrature modulator, the frequency domain form of the modulator output can be expressed as:

[0203] From the above formula, we can see that we only need to ensure that V2(-f t )-h IQ V1(f t )=0, the image component can be suppressed by pre-compensation. Therefore, the pre-compensation coefficient can be expressed as:

[0204] Here, the precompensation coefficients represent the frequency response of the optimal precompensation filter, which will be used in the digital domain of the modulator to predistort (precompensate) the I and Q signals before transmission. t ) and V1(f t ) is the ratio between them.

[0205] In an exemplary embodiment, the compensation signal generated by the local oscillator leakage compensation signal unit is generated based on a leakage signal corresponding to the first analog signal used by the phase shifter. The compensation signal generated by the local oscillator leakage compensation signal unit has the same frequency and amplitude as the leakage signal but an opposite phase.

[0206] Specifically, LO leakage correction (LOC) is achieved by generating a signal with equal amplitude but opposite phase to the LOL, thereby canceling it. After determining the exact amplitude and phase of the LOL, the cancellation signal can be generated by applying a DC offset to the transmitter input. The quadrature mixer structure enables the generation of cancellation signals. Because the mixer contains orthogonal signals to the LO, a signal of any phase and amplitude can be generated at the LO frequency. This can be interpreted geometrically in the complex plane. The first line shows the result of time-domain IQ two-way synthesis, while the second line provides a geometric interpretation from the perspective of the IQ two-way signal spectrum. After applying a DC offset to the transmit signal, the mixer output will contain the desired transmit signal and the desired LOL cancellation signal. The intentionally generated LOC cancellation signal cancels the unwanted LOL, leaving only the desired transmit signal.

[0207] Ideally, Tx LOC =h DC，I ×LO I -h DC，Q ×LO Q =-Tx LOLTherefore, the compensation method of LO leakage can be expressed as: pc (n) = a(n) - h DC

[0208] In practice, LOL can be modeled as follows:

[0209] The ideal IQ is α I cos(w m t) and α Q sin(w m t); the actual IQ is α I cos(w m t+φ I ) and α Q sin(w m t+φ Q ); LO leakage is Therefore, LO Leakage needs to be eliminated in the following ways:

[0210] Therefore, LOL can be calibrated by adding different DC offsets to the I and Q signals.

[0211] In an exemplary embodiment, the compensation signal generated by the harmonic compensation signal unit has the same frequency and amplitude as the harmonic signal in the signal transmission main path, and has an opposite phase.

[0212] Specifically, the harmonic compensation signal unit includes an n-th power module or an n-fold frequency signal generator; wherein: the n-th power module, wherein the signal input end of the n-th power module is connected to the signal output end of the second signal source, is used to use the signal output by the second signal source to generate a signal with a frequency n times the frequency of the signal output by the second signal source to obtain the compensation signal; the n-fold frequency signal generator is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source to obtain the compensation signal, wherein the value of n is a positive integer. The value of n is an odd number. Further, the value of n is 3.

[0213] Figures 4C and 4D are schematic diagrams of harmonic distortion compensation provided by embodiments of the present disclosure. As shown in Figures 4C and 4D, similar to the principle of eliminating LOL, there are two compensation methods for third-order harmonic distortion: a compensation architecture based on an n-th power module and a compensation architecture based on an n-times frequency waveform generator. pc (n) = a(n) - h HD3 ·(α * (n)) 3 a pc (n) = a(n) - h HD3 a3(n)

[0214] FIG5A is a schematic diagram of a digital phase shifter architecture in a signal transmission link shown in FIG2A ; FIG5B is a schematic diagram of a transmission link including a specific compensation unit according to an embodiment of the present disclosure. As shown in FIG5A , the signal transmission link (TX digital phase shifter architecture) of the digital phase shifter architecture may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (DDFS), an IQ digital to analog converter (DAC), a low-pass filter (LPF), an IQ modulator (IQ modulator / IQ mixer), a power amplifier (PA), etc., that is, the baseband signal source is configured to provide a digital phase-shifted source signal, and the direct digital frequency synthesizer may be configured to implement CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital Intermediate IF (Digital Intermediate IF) based on the received source signal. Frequency (DIF), digital intermediate frequency (DIF), and other signal waveforms and transmission modes to achieve flexible configuration of signal transmission form and transmission waveform. The signal amplified by the power amplifier can be radiated to a predetermined spatial area via an integrated or external transmitting antenna.

[0215] Regarding the signal transmission link of the digital phase shifter architecture in the embodiment of the present disclosure, since the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain and can generate an analog baseband signal through a DAC, and then modulate the transmission signal to a high frequency through an orthogonal mixer, that is, because the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower sidelobes, so its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy.

[0216] In some optional embodiments, when using RF LO frequency sweeping to achieve an FMCW signal transmission link, a compensation unit can be added to the signal transmission link to address issues such as TX IQ imbalance, signal leakage (e.g., TX LO leakage), and harmonic distortion (HD) caused by IQ mismatch. As shown in Figure 5A , a compensation unit (TX compensation) can be provided between the TX DDFS and the IQ DAC to perform calibration and compensation operations on the signal transmission link of the digital phase shifter architecture, thereby addressing at least one of the aforementioned issues. The HD caused by baseband third-order nonlinearity can be referred to as HD3.

[0217] In some optional embodiments, as shown in FIG5A and FIG5B , the TX compensation unit may include at least one of a TX LO leakage compensation unit, a TX IQ imbalance compensation unit, and a TX HD3 compensation unit. The LO leakage compensation unit may be configured to compensate for signal leakage, the IQ imbalance compensation unit may be configured to compensate for IQ imbalance, and the HD3 compensation unit may be configured to compensate for the aforementioned HD3. The IQ imbalance compensation unit may be configured to compensate for at least one of an IQ modulator imbalance and an IQ channel imbalance. In addition, when the TX compensation unit includes at least two of the LO leakage compensation unit, the IQ imbalance compensation unit, and the HD3 compensation unit, compensation can be performed simultaneously (e.g., in parallel) or sequentially (e.g., in series) based on actual needs and signal characteristics. As shown in FIG5B , IQ imbalance compensation can be performed first, followed by LO leakage compensation, and finally HD3 compensation.

[0218] In some optional embodiments, the signal transmission link of the digital phase shifter architecture may also include an error correction module for the DAC (TX DAC Board Error Correction) and an AWGN (additive white gaussian noise) module for Gaussian white noise, etc., which are not shown in the figure and can be added or deleted according to actual needs. Among them, the IQ mentioned in the embodiments of the present disclosure can be represented by the abbreviation of In-Phase (i.e., in phase), Q can be represented by the abbreviation of Quadrature (i.e., orthogonal), and RF can be represented by the abbreviation of Radio Frequency (i.e., radio frequency).

[0219] The embodiment of the present disclosure also provides a signal transceiver link, including a signal transmission link and a signal reception link, as shown in Figure 6A or Figure 6B. The signal transmission link may include: a transmitting baseband digital module 201, a digital-to-analog conversion module 202, a transmitting local oscillator 203 and a transmitting orthogonal modulator 204, wherein: the transmitting baseband digital module 201 is configured to generate two orthogonal transmitting digital baseband signals and send the generated transmitting digital baseband signals to the digital-to-analog conversion module 202; the digital-to-analog conversion module 202 is configured to convert the transmitting digital baseband signal into a transmitting analog baseband signal; the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO; and the transmitting orthogonal modulator 204 is configured to perform a phase shift operation on the transmitting local oscillator signal TX_LO based on the transmitting analog baseband signal to obtain a phase-shifted RF signal.

[0220] The signal receiving link may include a receiving end local oscillator 302, a receiving end mixer 303, an analog-to-digital converter (ADC) 304, and a receiving end baseband digital module 305; wherein the receiving end local oscillator 302 is configured to provide a receiving end local oscillator signal; the receiving end mixer 303 is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the analog-to-digital converter 304 is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; and the receiving end baseband digital module 305 is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication, for example, to obtain parameter information of the target such as distance, speed, angle, altitude, and micro-motion characteristics.

[0221] In the disclosed embodiment, the two ideal I-channel digital baseband signals and Q-channel digital baseband signals generated by the transmitting baseband digital module 201 can be converted into highly ideal complex signals after passing through the digital-to-analog conversion module 202. The phase of the complex signals can be precisely controlled by the transmitting baseband digital module 201. The receiver structure in the signal transceiver link shown in FIG6A or FIG6B can effectively obtain the phase information of the radio frequency signal in the signal transmission link, thereby realizing phase modulation of multiple antennas.

[0222] In some exemplary embodiments, the signal transmission link may further include a power amplifier 205 , wherein the power amplifier 205 is configured to amplify the power of the phase-shifted radio frequency signal and output the amplified signal to the transmitting antenna.

[0223] In some exemplary embodiments, the signal transmission link may further include a transmitting antenna 206 , wherein the transmitting antenna 206 is configured to radiate the amplified signal to a preset spatial region.

[0224] In some exemplary embodiments, the signal receiving link may further include a receiving antenna 301, wherein the receiving antenna 301 is configured to receive an echo signal, which is a signal formed when the signal transmitted by the signal transmitting link is reflected and / or scattered by a target object.

[0225] In some exemplary embodiments, the local oscillator signal at the receiving end may be a frequency sweep signal, or the local oscillator signal at the receiving end may be a single-tone signal.

[0226] In the disclosed embodiment of the signal transceiver link, the TX-LO signal received by the transmitting-end quadrature modulator 204 in the signal transmission link and the RX-LO signal received by the receiving-end mixer 303 in the signal reception link can have the same frequency. For example, assuming that the signal output by the transmitting-end baseband digital module 201 is a sine wave of x MHz, the TX-LO signal and the RX-LO signal can both be sine waves of z GHz, where x and z are both positive numbers, generally ranging from 0 to 1000.

[0227] The principles of the present disclosure are described below using the signal transceiver link shown in Figure 6A. As previously mentioned, for FMCW radar systems, the signal transmission link can have two transmission schemes: 1) the transmitting local oscillator signal is frequency-swept, and the transmitting digital baseband signal is single-tone; 2) the transmitting local oscillator signal is single-tone, and the transmitting digital baseband signal is frequency-swept. Assuming that the transmitting local oscillator signal TX_LO, the transmitting digital baseband signal, and the modulated transmit signal are represented by TLO(t), BB(t), and TX(t), respectively, and the subscripts I and q are used to represent the I-path signal and the Q-path signal, and the superscript a is used to represent their complex signal form, then under the two transmission schemes, the signals at each stage of the signal transmission link can be represented as follows:

[0228] 1) The local oscillator signal at the transmitting end is swept, and the digital baseband signal at the transmitting end is a single tone:

[0229] 2) Single tone local oscillator signal at the transmitting end, frequency sweep of digital baseband signal at the transmitting end:

[0230] in, is the sweep slope of the FMCW signal, f bb is the starting frequency of the transmitting digital baseband signal, f tlo is the starting frequency of the local oscillator signal at the transmitting end, is the initial phase of the transmitting digital baseband signal, is the initial phase of the local oscillator signal at the transmitting end.

[0231] Assume that the receiving end local oscillator signal RX_LO in the signal receiving link shown in FIG6A is represented by RLO(t). In this embodiment, assuming that the receiving end local oscillator signal RX_LO is a swept frequency signal, RLO(t) can be expressed as:

[0232] Among them, f rlo is the starting frequency of the local oscillator signal at the receiving end, and Φ0 is the initial phase of the local oscillator signal at the receiving end. Then, the RF signal transmitted by the signal transmission link generates an echo signal after being reflected / scattered by the target. After passing through the receiving antenna 301, the receiving mixer 303, and the receiving low-pass filter 307, the following is obtained: Where τ represents the time delay of the RF signal transmitted by the signal transmission link and then returned to the signal receiving link after being reflected / scattered by the target.

[0233] The above formula shows that both of the above two frequency sweeping methods can be used to control the initial phase of the digital baseband signal at the transmitting end. To perform precise phase modulation, thereby achieving high-precision and high-accuracy phase shifting, avoiding direct phase shifting at high frequencies.

[0234] In some optional embodiments, the receiving antenna 301 can be connected via a peripheral port of the chip and formed on a carrier such as a PCB. Meanwhile, in other optional embodiments, the receiving antenna can also be integrated into the chip package to form an AiP or AoP, that is, a chip structure with a packaged antenna.

[0235] In some exemplary embodiments, the signal receiving link may further include a low noise amplifier (LNA) 306 disposed between the receiving antenna 301 and the receiving mixer 303 , for performing low noise amplification on the echo signal received by the receiving antenna 301 before sending the signal to the receiving mixer 303 .

[0236] In some exemplary embodiments, the signal receiving link may further include a low-pass filter (LPF) 307 and a high-pass filter (HPF) 308 connected in series, disposed between the receiving mixer 303 and the analog-to-digital converter 304. The low-pass filter 307 and the high-pass filter 308 constitute a bandpass filter for filtering out-of-band noise.

[0237] In some exemplary embodiments, as shown in FIG6A , in the signal receiving link, the receiving mixer 303 may be a real mixer, and the analog-to-digital converter 304 may be a real analog-to-digital converter.

[0238] In the embodiment of the present disclosure, although the signal transmission link adopts a digital phase-shifting architecture, the signal receiving link may include a receiver with an orthogonal receiving architecture or a non-orthogonal receiving architecture. Therefore, it can effectively be compatible with sensors of receiving links of various architectures, effectively reducing the development cost of the entire transceiver link system.

[0239] In some other exemplary embodiments, as shown in FIG6B , in the signal receiving link, the receiving mixer 303 may be an orthogonal mixer, and the analog-to-digital converter 304 may be an orthogonal analog-to-digital converter.

[0240] To match the signal transmission link of the digital phase shifter architecture, the embodiment of the present disclosure adjusts the receiving mixer 303 in the signal receiving link to an IQ demodulator (IQ Demodulator) and simultaneously adjusts the analog-to-digital converter 304 to an IQ ADC. The echo signal received by the receiving antenna is sequentially processed by the aforementioned low-noise amplifier 306, receiving mixer 303, low-pass filter 307, high-pass filter 308, and analog-to-digital converter 304 before being converted into an IQ digital baseband signal. The subsequent receiving baseband digital module 305 processes this IQ digital baseband signal to obtain target parameter information such as range, speed, angle, altitude, and micro-motion characteristics (i.e., micro-Doppler).

[0241] When the receiving-end mixer 303 is an orthogonal mixer and the receiving-end local oscillator signal is a swept-frequency signal, the receiving-end local oscillator signal RX_LO can be expressed as:

[0242] In other exemplary embodiments, the local oscillator signal at the receiving end of the signal receiving link may be a single tone signal as shown in formula (12) or formula (13) in addition to being a frequency sweep signal as shown in formula (9) or formula (11).

[0243] The receiving mixer 303 is a real mixer: RLO(t)=cos(2πf rlo t+Φ0)(12).

[0244] The receiving mixer 303 is an orthogonal mixer:

[0245] When the local oscillator signal at the receiving end is a single-tone signal, it is only necessary to add a digital domain signal processing flow (including a digital domain mixing operation) to the receiving end baseband digital module 305 to obtain the required digital baseband signal.

[0246] In summary, the embodiments of the present disclosure can expand various system-level technical solutions according to different combinations of transmission schemes and receiving schemes (for example, whether the transmitting end uses a digital baseband signal single tone, a local oscillator signal sweep frequency or a digital baseband signal sweep frequency, a local oscillator signal single tone; whether the receiving end uses a real mixer, a real analog-to-digital converter or an orthogonal mixer, an orthogonal analog-to-digital converter; whether the receiving end uses a single-tone local oscillator signal or a swept frequency local oscillator signal).

[0247] Figure 7A is a schematic diagram of a transceiver link including TX IQ modulation (Mod) and RX real demodulation (Real De-Mod) in an embodiment of the present disclosure, Figure 7B is a schematic diagram of a transceiver link including TX IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure, Figure 8 is a schematic diagram of a transceiver link based on the structure shown in Figure 7B combined with BIST in an embodiment of the present disclosure, and Figure 9 is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure.

[0248] The following describes the transmit and receive links formed by the transmit link structure described in the embodiments of the present disclosure:

[0249] As shown in Figure 7A, a transceiver link may include a transmitting link and a receiving link. The transmitting link may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (TX DDFS), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), a power amplifier (PA), etc., which are connected in sequence. At the same time, the signal amplified by the power amplifier is radiated to a predetermined spatial area through a transmitting antenna. The receiving chain may include a low-noise amplifier (LNA), a real mixer, a trans-impedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), and a real digital-to-analog converter (ADC), connected in sequence. Specifically, the echo signal received by the receiving antenna is processed by the LNA, real mixer, TIA, LPF, HPF, and real ADC, and then converted into a real digital baseband signal. Subsequent digital signal processing by the digital signal processing module can obtain target parameter information such as range, speed, angle, altitude, and micro-motion characteristics. In the transceiver chain, the TX-LO signal received by the IQ modulator in the transmitting chain and the RX-LO signal received by the real mixer in the receiving chain may have the same frequency. For example, as shown in FIG7A , if the signal output by the Baseband is a sine wave of x MHz, then the TX-LO signal and the RX-LO signal may both be sine waves of z GHz.

[0250] In the embodiment shown in FIG7A , the transmitting link adopts a digital phase-shifting architecture, while the receiving link can adopt analog architecture components, i.e., IQ components are not required. Therefore, it is effectively compatible with sensors of the receiving link of the analog architecture, effectively reducing the development cost of the entire transceiver link system.

[0251] Optionally, in embodiments of the present disclosure, the receiving link may include a receiving antenna, i.e., the receiving antenna may be connected via a peripheral port of the chip and formed on a carrier such as a PCB. Furthermore, in some optional embodiments, the receiving antenna may also be integrated into the chip package to form an AiP or AoP, i.e., a chip structure with a packaged antenna.

[0252] In some optional embodiments, in order to match the transmitting chain of the digital phase shifter architecture, corresponding adjustments may be made to the receiving chain. For example, the transceiver chain shown in FIG7B may include a transmitting chain architecture and a receiving chain similar to that in FIG7A (to avoid redundancy, the same parts will not be described in detail here), that is, the Real Mixer in the receiving chain in FIG7A is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC. In this case, the receiving chain may include a low noise amplifier (LNA), an IQ demodulator (IQ Demodulator), a transimpedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), an IQ digital-to-analog converter (IQ ADC), etc., which are connected in sequence. That is, the echo signal received by the receiving antenna is sequentially passed through the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ After ADC processing, it is converted into an IQ digital baseband signal. The subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information such as distance, speed, angle, height and micro-motion characteristics (i.e. micro-Doppler) of the target.

[0253] At the same time, when performing self-calibration based on the transceiver link shown in FIG7B , as long as the signal output port of the transmitting link is directly connected to the signal input port of the receiving link via a transmission line, that is, the transmitting link directly sends the transmitting signal to the receiving link via the transmission line, so as to achieve self-calibration operation of the receiving and / or transmitting link without passing through the transmitting antenna and the receiving antenna. In this case, there is a certain frequency offset between the TX-LO signal received by the IQ modulator in the transmitting link and the RX-LO signal received by the IQ Demodulator in the receiving link. For example, as shown in FIG7B , if the signal output by the Baseband is a sine wave of x MHz, then the TX-LO signal can be a sine wave of z GHz, and the RX-LO signal can be a sine wave of (z GHz-y MHz), that is, the frequency offset is y MHz.

[0254] In some optional embodiments, as shown in FIG7B , a transmit link (a transmitter (TX) as shown in the figure) can be calibrated by adding a receive link (a receiver (RX) as shown in the figure), and a TX IQ imbalance compensation unit in the transmit link performs compensation based on the calibrated data. Simultaneously, a transmit link (a transmitter (TX) as shown in the figure) can also be calibrated by reusing a receive link (a receiver (RX) as shown in the figure) actually used for signal transmission and reception, and a TX IQ imbalance compensation unit in the transmit link and / or receive link performs compensation based on the calibrated data. Similar implementations can also be performed in other embodiments, which will not be described in detail for simplicity.

[0255] In some optional embodiments, based on the structure shown in FIG7B , to achieve precise calibration of the transceiver link, a built-in self-test (BIST) module may be provided at the RX-LO port of the IQ demodulator of the receive link shown in FIG7B . Specifically, as shown in FIG8 , based on the transceiver link structure shown in FIG7B , an IQ BIST architecture is provided at the RX-LO port of the IQ demodulator of the receive link to input an LO signal with a preset frequency offset to the RX-LO port of the IQ demodulator of the receive link. For example, the IQ BIST, which is composed of a phase angle converter and an IQ modulator, utilizes a received TX-LO signal, such as a TX-LO signal, to pass through the phase angle converter and the IQ modulator. A frequency-offset signal is generated based on the frequency offset signal of another input signal, the BIST-LO, of the IQ modulator and is then input to the RX-LO port of the IQ demodulator. For example, if the TX-LO signal is a z GHz sine wave and the BIST-LO signal is a y MHz sine wave, the frequency-shifted signal input to the RX-LO port of the IQ Demodulator is (z GHz - y MHz). It should be noted that x, y, and z are schematic values ​​in different embodiments and their specific values ​​may be the same or different.

[0256] In some optional embodiments, based on the IQ BIST architecture of the transceiver link structure shown in FIG8 , the transmit link of the digital phase shifter architecture can also be calibrated by multiplexing the receive link in the transceiver link. In other embodiments, the calibration operation of the transmit link using the receive link, and the calibration operation of the receive link using the transmit link, can be implemented by multiplexing the corresponding receive link or transmit link in the link that actually performs signal transmission and reception, or the calibration operation of the corresponding transmit link or receive link in the link that actually performs signal transmission and reception can be implemented by adding a corresponding calibration receive link or calibration transmit link.

[0257] Optionally, the IQ BIST may include a phase angle converter and an IQ modulator. The phase angle converter is used to implement separate calibration of the I and Q paths in the digital architecture's transmit link, while the other input signal BIST-LO of the IQ modulator may be a y MHz sine wave, which is used to simulate the characteristics of the echo signal formed by the transmit signal being reflected by the target. In Figure 8, x, y, and z are all positive numbers, and x≠y≠z, and can generally be between 0 and 1000.

[0258] Optionally, in the transceiver link shown in FIG8 , a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be provided in the transmitting link (e.g., between the TX DDFS and the IQ DAC), and / or a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be provided in the receiving link (e.g., after the IQ ADC). This may be to supplement the transmitted and / or received signals based on the calibration parameters (or coefficients) obtained by the self-calibration operation to resolve issues such as IQ imbalance.

[0259] In some optional embodiments, based on the structure shown in FIG8 , as shown in FIG9 , the above-mentioned IQ BIST module can be set between the signal output port of the transmitting link and the signal input port of the receiving link. That is, the transmitting link directly sends the transmitting signal to the receiving link through the IQ BIST module, so as to achieve self-calibration operation of the receiving link and / or the transmitting link without passing through the transmitting antenna and the receiving antenna.

[0260] It should be noted that the transmit chain structures shown in FIG7A and FIG8 to FIG9 only illustrate the TX IQ imbalance compensation unit. In actual applications, based on actual needs, an LO leakage compensation unit (TX LO leakage compensation) and an HD3 compensation unit (TX HD3 compensation) may be added to the transmit chain to form a compensation unit (TX compensation) that includes the TX LO leakage compensation unit, the TX IQ imbalance compensation unit, and / or the TX HD3 compensation unit.

[0261] FIG10 is a schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present disclosure, and FIG11 is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present disclosure.

[0262] As shown in FIG10 , a transceiver link, combined with the structures and related descriptions shown in FIG5B and FIG9 , may include a transmitting link, a receiving link, and a calibration link. The transmitting link may include a TX digital baseband signal source (TX Baseband), a direct digital frequency synthesizer (TX DDFS), a compensation unit (Compensation), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), and a power amplifier (PA), etc., which are connected in sequence. At the same time, the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving chain may include a low-noise amplifier (LNA), a real mixer, a trans-impedance amplifier (TIA), a high-pass filter (HPF), a variable gain amplifier (VGA), a real digital-to-analog converter (ADC), and an RX baseband for TX RF calibration, all connected in sequence. Specifically, the echo signal received by the receiving antenna is processed by the LNA, real mixer, TIA, HPF, VGA, and ADC before being converted into a real digital baseband signal. Subsequent digital signal processing by the digital signal processing module can obtain target parameter information such as range, speed, angle, altitude, and micro-motion characteristics. For the transmit chain, the compensation unit (TX compensation) arranged between the TX DDFS and the IQ DAC may include a TX LO leakage compensation unit, a TX IQ imbalance compensation unit, and / or a TX HD3 compensation unit, to implement corresponding compensation operations such as LO leakage, IQ imbalance, and HD3 in the transmit chain of the digital phase shifter architecture.

[0263] In some optional embodiments, a calibration module may be provided between the transmit link and the receive link. This calibration module may be configured to multiplex the receive link to perform operations such as calibration on the transmit link of the aforementioned digital phase shifter architecture. Simultaneously, the compensation module (i.e., the various compensation units in the aforementioned embodiments) may compensate for the transmitted signal at the transmit link end based on the parameters or coefficients obtained by the calibration operation of the calibration module. In other embodiments, a corresponding receive compensation module may also be provided simultaneously or separately in the receive link. In this case, the receive compensation module may compensate for the echo signal at the receive link end based on the parameters or coefficients obtained by the calibration operation.

[0264] As shown in FIG10 , the calibration module may include a BIST unit and an auxiliary circuit unit. Specifically, the output port of the transmit link is connected to any node between the real mixer and the real ADC in the receive link via the BIST unit and the auxiliary circuit unit. For example, the IQ Modulator in the transmit link generates a (z GHz±x MHz) RF signal based on an x ​​MHz digital phase-shifted baseband signal and a z GHz LO signal. The RF signal is then output to the BIST unit via the output port. The BIST unit performs a y MHz frequency offset operation on the received RF signal to obtain an analog echo signal of (z GHz±x MHz±y MHz). The BIST unit then down-converts the signal using an IQ De-Modulator in the auxiliary unit to obtain a preset intermediate frequency (IF) signal (z GHz±x MHz±y MHz-z GHz=±x MHz±y MHz). This IF signal is then input to a preset node in the receive link to implement calibration operations in the transmit link.

[0265] Optionally, the auxiliary circuit unit can be a quadrature demodulator circuit, and the output of the auxiliary circuit unit can be connected to any of the nodes between the TIA and HPF, between the HPF and VGA, or between the VGA and the real ADC in the receive chain. Furthermore, to maximize the reuse of the receive chain structure, after the output port of a transmit chain passes through the BIST unit and the auxiliary circuit unit, the I and Q branches can be connected to different transmit chains, respectively. This is shown in Figure 10 , calibrating one transmit chain by multiplexing two receive chains. After calibration of the transmit chain, the aforementioned compensation units (TX compensation), such as the LO leakage compensation unit, the IQ imbalance compensation unit, and / or the HD3 compensation unit, can be used to compensate for LO leakage, IQ imbalance, and HD3 issues in the transmit chain of the digital phase shifter architecture based on the parameters obtained from the calibration.

[0266] In some optional embodiments, the BIST unit may include a phase angle converter and an IQ modulator (IQ Modulator) connected in sequence, and the auxiliary circuit unit may include an LNA, an IQ De-Modulator, and a TIA connected in sequence, wherein the phase angle converter receives the RF signal output from the transmit link, one input of the IQ Modulator is connected to the output of the phase angle converter, and the other input receives a y MHz BIST-LO signal to generate a preset echo signal. The LNA amplifies the received echo signal and sends it to one input of the IQ De-Modulator. The other input of the IQ De-Modulator is used to receive a z GHz RX-LO signal. The two output branches of the IQ De-Modulator (i.e., the I branch and the Q branch) are respectively connected to corresponding nodes in their respective receive links after passing through the TIA, so as to output the generated preset intermediate frequency signal to the two receive links, thereby achieving a more efficient multiplexing of the receive link design while implementing the calibration operation.

[0267] It should be noted that for the calibration operation in the embodiments of the present disclosure, if the transmit link transmits a swept-frequency signal, during the actual calibration operation, the TX LO signal can be used as a single-tone signal for point-by-point calibration. At the same time, the TX LO signal can also be used as a swept-frequency signal for large-bandwidth calibration operations, and even swept-frequency bandwidth calibration can be used to implement calibration operations for swept-frequency signals of the entire frequency band in one go.

[0268] Based on the structure shown in Figure 10, in order to further suppress the preset levels of HD3, LO leakage, and IQ imbalance in the transmit link of the digital phase shifter architecture, at least two BIST units can be cascaded. As shown in Figure 11, by using two BIST units in series, the noise caused by these defects can be suppressed to a level of -50dB, effectively reducing the difficulty of developing and designing related link analog components.

[0269] In some optional embodiments, based on the transmission link of the digital phase shifter architecture described in the embodiments of the present disclosure, when calibrating and compensating for IQ Imbalance, the IQ Imbalance compensation coefficient can be obtained in the time domain (Time-Domain) based on spectrum analysis, and the IQ Imbalance compensation coefficient can also be obtained in the frequency domain (Frequency-Domain) based on the spectrum peak ratio.

[0270] In some optional embodiments, in order to further improve the accuracy of the IQ Imbalance compensation coefficient, the ideal compensation coefficient can be approximated by iterative calibration and compensation, or the ideal compensation coefficient can be obtained by multi-observation calibration and compensation.

[0271] For example, for the iterative calibration and compensation method, it can be determined whether to stop the iterative operation based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iterative conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations. For the multi-observation calibration and compensation method, after multiple (for example, three) calibration and compensation operations, the measurement data obtained from each operation can be subjected to FFT (Fast Fourier Transform) and the corresponding amplitude and phase information can be obtained. The measurement can be subtracted and normalized to obtain relevant data, and an observation matrix can be constructed; the corresponding compensation coefficient can be reversely solved based on the data obtained by inverting the observation matrix.

[0272] In some optional embodiments, based on a similar concept to the above-mentioned IQ Imbalance compensation coefficient, the LO leakage and / or HD3 compensation coefficients may be obtained by adopting an iterative calibration and compensation method or a multiple observation calibration and compensation method.

[0273] The above examples of transmitters with compensation units mentioned the problem of solving harmonic distortion in transmitters. Research has found that some harmonic distortion may be caused by components with nonlinear characteristics, such as mixers in the transmitter.

[0274] Taking Figure 12A as an example, Figure 12A is a schematic diagram of the structure of the mixer in an embodiment of the present disclosure. As shown in Figure 12A, the mixer includes a voltage-current converter (V / I Converter), a current switch (Current Switch), and a current-voltage converter (I / V Converter). The voltage-current converter converts the received voltage signal into a current signal; the current switch is connected to the voltage-current converter and the second signal generator, and is used to process the current signal output by the voltage-current converter using the local oscillator signal; the current-voltage converter is connected to the current switch, and is used to convert the current signal output by the current switch into a voltage signal.

[0275] In the above structure, because the voltage-to-current converter is equipped with a transistor amplifier, due to the nonlinear characteristics of the transistor amplifier and the low frequency of the baseband signal, the current signal output by the voltage-to-current converter contains harmonic signals corresponding to the baseband signal. For example, the HD caused by the third-order nonlinearity of the baseband can be simply referred to as HD3. Similarly, the harmonic caused by the fifth-order nonlinearity is called HD5. When the current switch processes the current signal output by the voltage-to-current converter, the frequency of the harmonic is up-converted to the radio frequency band. Due to the high complexity of suppressing harmonic signals in the radio frequency band, the hardware cost is high. If the harmonic signals in the radio frequency band are not removed, the signal quality of the radar transmission and reception will be affected, and thus the accuracy of the radar measurement.

[0276] The compensation unit is used to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the radio frequency signal. The compensation unit is independent of the first signal generator.

[0277] To this end, the compensation unit may include a cancellation signal generator. The cancellation signal output by the compensation unit can suppress harmonic signals in the radio frequency signal, reduce the harmonic components in the radio frequency signal, and thus improve the signal quality of the radio frequency signal output by the transmitter.

[0278] In the disclosed embodiment, a compensation unit generates a cancellation signal for harmonic signals in a signal transmission link, either through feedback or based on the characteristics of the transmitted wave, and then inputs it into the signal transmission link to cancel the harmonic signals in the RF signal output by the signal transmission link. This cancellation signal has characteristics such as an opposite phase and similar amplitude to the harmonic signal transmitted by the RF transmission circuit, thereby achieving the purpose of suppressing the harmonic signal.

[0279] In some examples, the compensation unit generates a compensation signal including a cancellation effect based on parameters such as the phase, frequency, or amplitude of the baseband signal generated by the first signal generator, or even the path length of the baseband signal combined with the LO signal.

[0280] For example, Figure 5A illustrates a transmitter that incorporates a compensation unit into a signal transmission link. In the structure shown in Figure 5A, the compensation unit is a TX compensation unit. The TX compensation unit includes a generator (not shown) that generates a cancellation signal based on the characteristics of the transmitted wave. This cancellation signal generator can be, for example, the TX HD3 compensation unit shown in Figure 5B.

[0281] Among them, the baseband processor (abbreviated as the baseband frame in Figure 5A) controls the orthogonal digital baseband signal generated by the TX DDFS, and the TX compensation unit generates an orthogonal compensation signal based on the parameters of the orthogonal digital signal, merges the orthogonal compensation signal and the orthogonal digital signal and sends it to the IQ DAC to be converted into an analog baseband signal. After LPF filtering, it is mixed by the mixer (i.e., the IQ modulator in Figure 5A) to obtain a radio frequency signal based on the mixing of the TX LO signal and the analog baseband signal. The PA amplifies the mixed signal and outputs it through the transmitting antenna. Among them, the compensation signal cancels out at least part of the harmonic signals in the radio frequency transmission circuit, such as the HD3 harmonic signal. Therefore, the interference in the transmitted radio frequency signal will be greatly reduced. The radio frequency signal can be an FMCW signal.

[0282] In other examples, the compensation unit generates a compensation signal based on harmonic information fed back from the RF transmission circuit. See Figure 12B, which is a schematic diagram of the compensation unit in the transmitter shown in Figure 5A. As shown in Figure 12B, the compensation unit includes an acquisition circuit and a cancellation signal generator.

[0283] The acquisition circuit is coupled to the RF transmission circuit and is configured to acquire signals from the RF transmission circuit to obtain an acquisition signal. The acquisition signal (or sampling signal) can reflect waveform information (also known as harmonic parameters) in the harmonic signal, such as the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the power of the harmonic signal, and the power of the main frequency signal.

[0284] It should be noted that the harmonic parameters reflected in the acquisition signal are related to the information carried by the signal that can be acquired by the acquisition circuit. For example, if the acquisition circuit is a power acquisition circuit, the corresponding acquisition signal includes the power of the main frequency. In another example, if the acquisition circuit utilizes at least part of the circuitry of a receiver, the acquisition signal reflects the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.

[0285] At least one of the aforementioned harmonic parameters can be extracted using analog circuits. For example, a coupler and power detector can be used to output the power of the main frequency signal. Alternatively, the digital circuitry within the radar chip can be leveraged to perform frequency domain calculations to extract harmonic parameters. For example, by coupling with the RF transmitter circuit, a signal identical to the signal transmitted at the coupling point is obtained as an acquisition signal. This acquisition signal contains both the main frequency signal and harmonic signals. The acquisition signal is converted to a digital signal using an ADC and then passed to the digital circuit for frequency domain calculations to obtain more harmonic parameters.

[0286] In one implementation, the input terminal of the acquisition circuit is connected to the output terminal or the signal detection terminal of the mixer. This approach can detect harmonic signals generated by the voltage-to-current converter and has a simplified acquisition circuit.

[0287] In an exemplary embodiment, the signal calibration link has a first input connected between a voltage-to-current converter and a current switch within a transmitting mixer in the main signal transmission path, a second input connected between the main signal transmission path and a transmitting antenna, and a signal output connected to a compensation circuit in the main signal transmission path. The link is configured to acquire a signal in the main signal transmission path from at least one of the first input and the second input, and determine compensation information based on the acquired signal. For example, in one connection configuration shown in Figures 13A and 13B , the input of the acquisition circuit is connected to a detection terminal between the voltage-to-current converter and the current switch, and is coupled to an ADC.

[0288] As shown in Figure 13C, the signal calibration link includes a calibration demodulator (IQ Demodulator), a multiplexer (MUX) and a calibration module (TX Calbration); wherein: the calibration demodulator is configured to obtain the signal in the signal transmission main path from the second input terminal and perform demodulation processing; the multiplexer has two signal input terminals and one signal input terminal, one of which is connected between the voltage-current converter and the current switch in the transmitting mixer, and the other signal input terminal is connected to the signal output terminal of the calibration demodulator, for outputting a signal corresponding to one of the two signal input terminals; the calibration module is configured to determine the compensation information according to the signal output by the multiplexer.

[0289] In another embodiment, the input of the acquisition circuit is connected to the RF output or RF detection terminal of the RF transmission circuit. The RF output terminal is, for example, the output terminal of the RF transmission circuit. The RF detection terminal is, for example, the input or output terminal of at least one PA stage in the RF transmission circuit. This embodiment can acquire more accurate harmonic parameters from the RF transmission circuit, but has a more complex circuit structure.

[0290] In some chips that include a BIST module, the acquisition circuit can acquire the acquisition signal using some or all of the circuitry in the BIST module. For example, as shown in Figure 13B , the acquisition circuit's input is coupled to the RF output terminal, which in turn includes a downconverter, a filter, and other components, and is connected to an IQ ADC to output the digital acquisition signal. The downconverter, filter, and other components can be reused in the BIST module or receiver.

[0291] The collected signal is input to a cancellation signal generator. The cancellation signal generator is at least one circuit in the compensation unit. The cancellation signal generator is connected to the first signal generator so that the signal received by the RF transmission circuit includes both the baseband signal and the cancellation signal.

[0292] For example, the cancellation signal generator includes the aforementioned cancellation signal generator and a digital circuit for extracting harmonic information, wherein the digital circuit for extracting harmonic information can be configured independently or at least partially shared with the digital circuit in the radar chip.

[0293] The digital circuit for extracting harmonic information utilizes, for example, the digital circuit in the radar chip used to process the difference frequency baseband signal to extract harmonic information such as harmonic frequency, main frequency, and main frequency power, and provides this information to the cancellation signal generator. The cancellation signal generator generates a cancellation signal based on the received parameters.

[0294] For example, the digital circuit for extracting harmonic information extracts the amplitude of the main frequency in the acquired signal and calculates the harmonic amplitude based on the difference between the preset main frequency amplitude and the harmonic amplitude. The cancellation signal generator generates a harmonic signal compensation signal based on the calculated harmonic amplitude and other preconfigured harmonic parameters. These preconfigured harmonic parameters can be calculated based on the sweep frequency range and phase of the main frequency signal to be transmitted by the radar chip.

[0295] The cancellation signal generator in the cancellation signal generator can be configured independently of the first signal generator, or at least partially shared with it. For example, the cancellation signal generated by the cancellation signal generator is input into the first signal generator, so that the baseband signal output by the first signal generator includes the cancellation signal. The cancellation signal generator can include a third harmonic generator and a fifth harmonic generator.

[0296] For another example, the compensation unit further includes an adder coupled to the cancellation signal generator and the first signal generator to combine the baseband signal generated by the first signal generator with the cancellation signal generated by the cancellation signal generator. In this embodiment, the cancellation signals include a cancellation signal Signal_HD3 generated by the third harmonic generator to cancel the third harmonic, and a cancellation signal Signal_HD5 generated by the fifth harmonic generator to cancel the fifth harmonic. Cancellation signals Signal_HD3 and Signal_HD5, along with the baseband signal generated by the first signal generator, are combined by the adder and output to the RF transmission circuit.

[0297] In summary, the transmitter circuit examples provided in the present disclosure that pre-input a cancellation signal into the RF transmission circuit in a feedback manner can ensure that the RF signal transmitted by the chip contains sufficiently low harmonic signals under different environments.

[0298] To effectively suppress harmonic signals during chip use based on the chip's actual operating environment, for example, by taking into account the impact of ambient temperature on semiconductor devices, the present disclosure further provides a method for canceling harmonic signals in the transmitter using a feedback mechanism, comprising: Step 10: Acquiring a signal from a signal transmission link to obtain an acquired signal; wherein the signal transmission link is used to generate a radio frequency signal for radar detection, wherein the radio frequency signal includes a harmonic signal; Step 20: Detecting the acquired signal, generating a cancellation signal for canceling the harmonic signal, and outputting the cancellation signal to the signal transmission link.

[0299] The method provided by the embodiments of the present disclosure performs an acquisition operation on the signal in the signal transmission link to obtain an acquisition signal, and uses the acquisition signal to generate a cancellation signal, and outputs it to the signal transmission link, thereby using the cancellation signal to suppress the harmonic signal in the radio frequency signal and reduce the harmonic component in the radio frequency signal, thereby improving the signal quality of the radio frequency signal output by the transmitter, and further improving the reception performance of the receiver for the radio frequency signal.

[0300] The following is an example of a transmitter and its working process with reference to Figures 7B to 13C:

[0301] For example, Figure 7B illustrates an example of using a feedback mechanism to extract harmonic information from a transmitter, enabling the compensation unit to generate a corresponding cancellation signal. In the structure shown in Figure 7B, the compensation unit includes a TX HD3 compensation unit. The TX HD3 compensation unit generates a compensation signal based on the waveform characteristics of the received signal. This feedback mechanism can be implemented in the radar chip's calibration mode to prevent weakening of the radar chip's signal transmission power during normal detection.

[0302] The baseband processor in the transmitter (such as the baseband block in Figure 7B ) controls the orthogonal digital baseband signals generated by the TX DDFS. These are combined with the orthogonal digital cancellation signals generated by the TX compensation unit and then sent to the IQ DAC for conversion into an analog baseband signal. This analog baseband signal is mixed with an analog cancellation signal used to cancel the harmonic signals in the transmit chain. This analog baseband signal is filtered through the LPF and then enters the first mixer (i.e., the IQ modulator in Figure 7B ). The first mixer uses the TX LO to mix the received filtered signal to obtain an RF signal. This RF signal is coupled through the receiver and output to the TX HD3 calibration circuit in the compensation unit (the TX HD3 calibration block in Figure 7B ). The TX HD3 calibration circuit can be considered a digital circuit for extracting harmonic information.

[0303] In the receiver, the LNA amplifies the transmitter's output signal and then outputs it to the second mixer (i.e., the IQ demodulator in Figure 7B ) to generate a demodulated signal. This demodulated signal is then sent to a transimpedance amplifier for amplification. After passing through an LPF and a HPF, it undergoes analog-to-digital conversion via an IQ ADC before being sent to the TX HD3 calibration circuit. The TX HD3 calibration circuit extracts harmonic information from the transmitter's feedback signal and, through a higher-level controller, converts it into the parameters required to generate a cancellation signal, which is then provided to the TX HD3 compensation unit. Examples of harmonic information obtained by the TX HD3 calibration circuit include one or more of the following parameters: the initial phase, starting frequency, cutoff frequency, frequency variation duration, and center frequency of the harmonic signal (or main frequency signal). Based on the harmonic information, the TX HD3 compensation unit or the higher-level controller determines the parameters used to generate the cancellation signal within the compensation unit, such as the initial phase, frequency, and delay of the cancellation signal.

[0304] It should be noted that the cancellation operations of the third harmonic and / or fifth harmonic in the above examples may be determined according to the requirements of the transmitter.

[0305] FIG13A is a schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module according to an embodiment of the present disclosure, and FIG13B is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module according to an embodiment of the present disclosure.

[0306] In some optional embodiments, based on the transmission chain of the digital phase shifter architecture described in the embodiments of the present disclosure, when calibrating and compensating for HD3, since the main source of HD3 generation in the active mixer is the third harmonic in the nonlinearity of V / IConverter, it can be achieved through a compensation architecture based on a cubic module as shown in Figure 13A, or a compensation structure based on a tripled frequency waveform generator as shown in Figure 13B.

[0307] Figure 13C is a schematic diagram of calibration and compensation for a transmit chain based on a digital phase shifter architecture according to an embodiment of the present disclosure. As shown in Figure 13C , based on the technical details of the calibration and compensation operations for IQ Imbalance, LO leakage, and HD3 according to the embodiment of the present disclosure, IQ Imbalance compensation is achieved by compensating the conjugate signal of the BB (baseband) signal to inversely cancel the image component, and this compensation operation is unaffected by the IQ Imbalance calibration method. LO Leakage compensation is achieved by adjusting the DC component (i.e., DC offset) of both the IQ and Q channels, and similarly, this LO Leakage calibration method has no impact on the compensation scheme. Regarding HD3, since the third-order harmonic distortion of the quadrature mixer V / I converter is the primary source of HD3, and harmonic distortion is affected by DC offset, when both LO Leakage and HD3 calibration are required in the transmit chain, HD3 calibration should be performed after LO Leakage calibration to ensure accurate HD3 calibration.

[0308] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency multiplication waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process.

[0309] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO leakage can be calibrated and compensated first, and then the root cause of the HD3 problem, namely the HD3 compensation coefficient, can be calibrated under a stable DC bias. Subsequently, IQ imbalance is calibrated and further compensated for, and then the third harmonic distortion is compensated for both the IQ and Q channels based on the results of IQ imbalance pre-compensation.

[0310] In an optional embodiment, for a digital pre-compensation architecture based on a frequency-doubling waveform generator module, after calibrating and compensating for LO leakage, the HD3 compensation coefficient can be calibrated to obtain the HD3 compensation coefficient, and the IQ imbalance can be calibrated and compensated under a stable DC bias. Subsequently, based on the compensation results, the actual waveforms of the IQ signals and the HD3 compensation coefficient are calculated, and the waveform information of the 3x and 5x frequencies that require pre-compensation is calculated in reverse.

[0311] In another optional embodiment, for a digital precompensation architecture based on a frequency-doubling waveform generator module, LO leakage can be calibrated and compensated first. Then, multiple (e.g., three) observations are performed to simultaneously calibrate the precompensation coefficients for HD3 and IQ Imbalance. Then, further observations are performed (e.g., two) to calibrate the precompensation coefficients at the HD3 mirror position. Finally, the 3rd and 5th frequency coefficients requiring precompensation are calculated from the precompensation coefficients at the HD3 and HD3 mirror positions. It should be noted that the observations in this disclosed embodiment represent operations such as testing and comparative analysis of different test results.

[0312] Referring to the structure shown in FIG13C , it can be seen that the acquisition circuit has two acquisition branches, and the two acquisition branches can be switched dynamically; wherein, the input end of one acquisition branch is connected between the voltage-current converter and the current switch; the input end of the other acquisition branch is connected to the output end of the power amplifier, and the acquisition branch is provided with an IQ demodulator; in addition, in the structure shown in FIG13C , the acquisition circuit is further provided with a multiplexer, wherein the input end of the multiplexer is respectively connected to the output end of the two acquisition branches, and the output end is used to output the acquired signal.

[0313] In the structures shown in FIG. 13A to FIG. 13C , the acquisition signal may be an analog signal, that is, the output end of the acquisition circuit is connected to the IQ ADC; or, the acquisition signal may be a digital signal, and the acquisition circuit at least includes the IQ ADC.

[0314] It should be noted that when the RF signal sent by the signal transmission link is not an orthogonal signal, the acquisition circuits in Figures 13A to 13C can be implemented using non-orthogonal components. For example, a single-ended down-conversion mixer can be used instead of an IQ decoder, and a single-ended analog-to-digital converter can be used instead of an IQ ADC.

[0315] As shown in Figure 14A, an embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission chain. The signal transmission method includes: step 1401, determining the phase of the radio frequency transmission signal of each signal transmission chain; step 1402, determining the initial phase of the transmitting end digital baseband signal in each signal transmission chain according to the phase of the radio frequency transmission signal; step 1403, generating a transmitting end digital baseband signal according to the determined initial phase; step 1404, converting the transmitting end digital baseband signal into a transmitting end analog baseband signal, and performing a phase shift operation on the transmitting end local oscillator signal based on the transmitting end analog baseband signal.

[0316] The signal transmission method of the disclosed embodiment generates a digital baseband signal in the digital domain by using a digital phase shifter architecture, which has better orthogonality and lower sidelobes. Its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of low link loss, low cost, and no need for offline calibration. It can also support more flexible wave transmission schemes, such as high-performance Doppler multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0317] In some exemplary embodiments, the digital baseband signal at the transmitting end is a single-tone signal, and the local oscillator signal at the transmitting end is a swept-frequency signal; or, the digital baseband signal at the transmitting end is a swept-frequency signal, and the local oscillator signal at the transmitting end is a single-tone signal.

[0318] In some exemplary embodiments, the frequency bandwidth of the swept frequency signal is above 2 GHz.

[0319] In some optional embodiments, to further improve the accuracy of the IQ Imbalance compensation coefficient, iterative calibration and compensation can be used to approximate the ideal compensation coefficient. Alternatively, multiple observations of calibration and compensation can be used to obtain the compensation coefficient used in actual operating conditions. The determination of the compensation coefficient requires additional consideration of the impact of link characteristics on the accuracy of the compensation coefficient calculation.

[0320] Taking quadrature imbalance as an example, the compensation coefficient h under actual working conditions is The compensation coefficient H under ideal conditions is Due to the obvious difference between the two expressions, the compensation coefficient under actual working conditions is different from that under ideal conditions.

[0321] Because f t Compared with the carrier frequency, it is a narrowband signal, so it can be approximately considered

[0322] It is necessary to ensure W1(Δ f -f t )≈W1(Δ f +f t ), or the ratio of these two coefficients can be better estimated through calibration;

[0323] V1(f t ) and V1(-f t )

[0324] *E I (f t )(E Q (ft )) and E I (-f t )(E Q (-f t )) have the same amplitude but opposite phase;

[0325] * and Because f t =5MHz is a narrowband signal compared to the carrier frequency of 76-81GHz, so it can be approximately considered equal;

[0326] * and Not affected.

[0327] If the frequency response characteristics of the link are not considered and the compensation is performed directly using the observation, the image component cannot be cancelled. Therefore, in an actual system, the impact of the frequency response of the link on the calibration operation must be considered.

[0328] Similarly, the compensation coefficients for harmonic distortion and LO leakage also differ between the actual operating state of the signal transmission link and the ideal state.

[0329] Figure 14B is a schematic diagram illustrating the principle of determining the compensation coefficient provided by an embodiment of the present disclosure. The contents of each coordinate system in Figure 14B are explained one by one below: Figure 14B (a) represents the phase and amplitude of the compensation coefficient H in the ideal state of the signal transmission link; Figure 14B (b) represents the phase and amplitude of the compensation coefficient H in the ideal state of the signal transmission link, the phase and amplitude of the compensation coefficient h in the actual working state, and the difference D between the compensation coefficient h in the actual working state and the compensation coefficient H in the ideal state; Figure 14B (c) represents that the three vectors in Figure 14B (b) do not change with changes in amplitude; therefore, it is only necessary to control the compensation coefficient h in phase to be infinitely close to the compensation coefficient H, see Figure 14B (d) for details. (e) in Figure 14B indicates that area 1 indicates that the compensation coefficient h is close to the compensation coefficient H, and area 2 indicates that the compensation coefficient h is not close to the compensation coefficient H; (f) in Figure 14B optimizes the area in (e) in Figure 14B, including reducing the size of area 1, thereby enlarging area 2, and further dividing the enlarged area 2 to obtain area 3 and a new area 2, wherein the compensation coefficient h in area 3 is not close to the compensation coefficient H, and the new area 2 indicates that the compensation coefficient with a phase opposite to that of the compensation coefficient h is close to the compensation coefficient H; for example, the compensation coefficient h in Figure 14B(g) and Figure 14B(h) are not in area 1, but the compensation coefficient h in Figure 14B(h) is closer to the compensation coefficient H than the compensation coefficient h in Figure 14B(g), which can be clearly seen from the length of the difference D in Figure 14B(g) and Figure 14B(h).

[0330] Based on the above analysis, the embodiments of the present disclosure propose an iterative calibration and compensation method.

[0331] Specifically, whether to stop the iterative operation can be determined based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iteration conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations.

[0332] In some optional embodiments, based on a similar concept to the above-mentioned IQ Imbalance compensation coefficient, the LO leakage and / or HD3 compensation coefficients may be obtained by adopting an iterative calibration and compensation method or a multiple observation calibration and compensation method.

[0333] Specifically, an embodiment of the present disclosure provides a signal calibration link of a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to a compensation coefficient, so as to achieve target detection and / or communication, wherein: the signal calibration link is configured to obtain current observation information of the main signal transmission path under the current compensation coefficient; and when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition.

[0334] The signal calibration link is described below using an embodiment:

[0335] In some optional embodiments, a signal calibration method may include the following steps: Step A1: With an initial compensation coefficient h(0) of 0, the signal state of the signal transmission link is obtained to obtain initial observation information O(0). Step A2: The signal transmission link is compensated for the first time using the first compensation coefficient h(1) determined based on the initial observation information O(0), the signal state of the signal transmission link is obtained, and the first observation information O(1) is obtained. Step A3: If the difference between the initial observation information O(0) and the first observation information O(1) is greater than a preset difference threshold, the kth compensation coefficient h(k) is determined based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1), where k is an integer greater than or equal to 2. If the difference between the absolute value of the initial observation information O(0) and the absolute value of the first observation information O(1) is greater than a preset difference threshold, then based on the first compensation coefficient h(1) being updated to the initial observation value O(0), the sum of the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1) is iteratively calculated to determine the kth compensation coefficient h(k). If the difference between the absolute value of the first observation information O(1) and the absolute value of the initial observation information O(0) is greater than a preset difference threshold, based on the first compensation coefficient h(1) being updated to the inverse of the initial observation value O(0), the difference between the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1) is iteratively calculated to determine the kth compensation coefficient h(k). Step A4: If the difference between the initial observation information O(0) and the first observation information O(1) is less than a preset difference threshold, the initial observation information O(0) is adjusted according to the preset phase adjustment amount to obtain the adjusted first compensation coefficient h(1), and then the new first observation information O(1) is obtained again based on the adjusted first compensation coefficient h(1), and the kth compensation coefficient h(k) is determined based on the new first observation information O(1); wherein the difference between the initial observation information O(0) and the new first observation information O(1) is greater than the difference threshold.

[0336] Optionally, if the difference between the absolute value of the initial observation information O(0) and the absolute value of the new first observation information O(1) is greater than the difference threshold, the sum of the k-1th observation information O(k-1) and the adjusted k-1th observation information O(k-1) obtained after processing the k-1th observation information O(k-1) by the phase adjustment amount is iteratively calculated to determine the kth compensation coefficient h(k).

[0337] Optionally, if the difference between the absolute value of the new first observation information O(1) and the absolute value of the initial observation information O(0) is greater than the difference threshold, the first compensation coefficient h(1) is updated to the inverse of the adjusted first compensation coefficient h(1), and the difference between the k-1th compensation coefficient h(k-1) and the adjusted k-1th observation information O(k-1) obtained by processing the k-1th observation information O(k-1) with the phase adjustment amount is iteratively calculated to determine the kth compensation coefficient h(k). In each iterative operation in step A3 and step A4, it is determined whether the observation information corresponding to each compensation coefficient is less than a preset iteration threshold; if the observation information is less than the iteration threshold, the iterative operation is stopped, and the compensation coefficient used for the observation information less than the iteration threshold is used as the compensation coefficient.

[0338] In some other optional embodiments, the compensation operation of quadrature imbalance is taken as an example for description, wherein the symbols mentioned in the above embodiment are distinguished by the subscript of IQ. Specifically: Step B1: When h IQ When (0)=0, it is estimated that O IQ (0). Step B2: When h IQ (1) = O IQ (0), we get O IQ (1). Step B3: Compare O IQ (0) and O IQ (1) determines the size relationship of subsequent iterations; if |O IQ (0)|>|O IQ (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: h IQ (1) = O IQ (0);h IQ (k) = h IQ (k-1)+O IQ (k-1), where k=2,3,…….. If |O IQ (0)|<|O IQ (1)|-ε, where ε represents the threshold, then the subsequent iterative expression is as follows: h IQ (1)=-O IQ (0);h IQ (k) = h IQ (k-1)-O IQ (k-1), where k = 2, 3, .... Step B4: If the relationship in step B3 is not satisfied, let Get O back IQ (1): If |O IQ (0)|>|O IQ (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: Where k=2,3,…….. If |O IQ (0)|<|O IQ (1)|-ε, where ε represents the threshold, and the subsequent iterative expression is as follows: Where k=2,3,…….. Step B5: When O IQ (k) is less than a certain threshold (where the threshold value is less than or equal to -50db), the iteration is stopped.

[0339] In some other optional embodiments, the LO leakage compensation operation is taken as an example for description, wherein the symbols mentioned in the above embodiments are distinguished by the subscript DC. Specifically: Step C1: When h DC,I (0) = h DC,Q When (0)=0, it is estimated that O DC (0); where h DC,I Indicates the initial compensation coefficient of the I-channel signal, h DC,Q Indicates the initial compensation coefficient of the Q channel signal. Step C2: When h DC,I (1)=Re{O DC (0)}、h DC,Q (1)=Im{O DCQ (0)}, we get O DC (1); where Re{O DC (0)} means O DC (0) real part; Im{O DCQ (0)} indicates that DC (0) is the imaginary part. Step C3: Compare O IQ (0) and O IQ (1) determines the size relationship of subsequent iterations; if |O IQ (0)|>|O IQ (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: h DC,I (1)=Re{O DC (0)};h DC,Q (1)=Im{O DCQ (0)};h DC,I (k) = h DC,I (k-1)+Re{O DC (k-1)}, where k = 2, 3, ....; h DC,Q (k) = h DC,Q (k-1)+Im{O DC (k-1)}, where k = 2, 3, ....; if |O IQ (0)|<|O IQ(1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: h DC,I (1)=-Re{O DC (0)};h DC,Q (1)=-Im{O DCQ (0)};h DC,I (k) = h DC,I (k-1)-Re{O DC (k-1)}, where k = 2, 3, ....; h DC,Q (k) = h DC,Q (k-1)-Im{O DC (k-1)}, where k = 2, 3, .... Step C4: If the relationship in step C3 is not satisfied, let Get O back DC (1): If |O IQ (0)|>|O IQ (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: Where k = 2, 3, ....; Where k=2,3,…….; if |O IQ (0)|<|O IQ (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: Where k = 2, 3, ....; Where k=2,3,…….. Step C5: When O DC (k) is less than a certain threshold (where the threshold value is less than or equal to -50db), the iteration is stopped.

[0340] In some other optional embodiments, the compensation operation of harmonic distortion is taken as an example for description, wherein the symbols mentioned in the above embodiments are distinguished by the subscript HD3.

[0341] If the initial phase of the baseband (BB) signal is x degrees, then the initial phase of the subsequent 3BB signal must be 3x degrees, and O(n) represents the amplitude and phase value of the frequency point corresponding to the 3BB signal ÷ the amplitude and phase value of the frequency point corresponding to the BB signal.

[0342] Specifically, step D1: when h HD3 When (0)=0, it is estimated that O HD3 (0). Step D2: When h HD3 (1) = O HD3 (0), we get O HD3 (1). Step D3: Compare O HD3 (0) and OHD3 (1) determines the size relationship of subsequent iterations; if |O HD3 (0)|>|O HD3 (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: h HD3 (1) = O HD3 (0);h HD3 (k) = h HD3 (k-1)+O HD3 (k-1), where k=2,3,…….; if |O HD3 (0)|<|O HD3 (1)|-ε, where ε represents the threshold, then the subsequent iterative expression is as follows: h HD3 (1)=-O HD3 (0);h HD3 (k) = h HD3 (k-1)-O HD3 (k-1), where k = 2, 3, ....; Step D4: If the relationship in step D3 is not satisfied, then let Get O back IQ (1): If |O HD3 (0)|>|O HD3 (1)|+ε, where ε represents the threshold, then the subsequent iterative expression is as follows: Where k=2,3,…….; if |O HD3 (0)|<|O HD3 (1)|-ε, where ε represents the threshold, and the subsequent iterative expression is as follows: Where k=2,3,…….. Step D5: When O HD3 (k) is less than a certain threshold (where the threshold value is less than or equal to -50db), the iteration is stopped.

[0343] In addition, based on the above analysis, the embodiments of the present disclosure also propose a multi-observation calibration and compensation method.

[0344] Specifically, after multiple (for example, three) calibration and compensation operations, the measurement data obtained from each operation can be subjected to FFT (Fast Fourier Transform) and the corresponding amplitude and phase information can be obtained. The measurements can be subtracted and normalized to obtain relevant data, and an observation matrix can be constructed. Subsequently, the corresponding compensation coefficients can be solved in reverse based on the data obtained by inverting the observation matrix.

[0345] In some optional embodiments, based on a similar concept to the above-mentioned IQ Imbalance compensation coefficient, the LO leakage and / or HD3 compensation coefficients may be obtained by adopting an iterative calibration and compensation method or a multiple observation calibration and compensation method.

[0346] Specifically, a signal calibration link of a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on a generated signal according to a compensation coefficient, so as to achieve target detection and / or communication, wherein: the signal calibration link is configured to determine initial observation information O(0), first observation information O(1) and second observation information O(2) corresponding to the main signal transmission path under conditions of different values ​​of an initial compensation coefficient h(0), a first compensation coefficient h(1) and a second compensation coefficient h(2); and to determine a third compensation coefficient h(3) using the initial observation information O(0), the first observation information O(1) and the second observation information O(2) as the compensation coefficient used in the compensation operation of the signal transmission link.

[0347] In another optional embodiment, a method for compensating for quadrature imbalance may include: step E1: obtaining a signal state of a signal transmission link with an initial compensation coefficient h(0) of 0 to obtain initial observation information O(0). step E2: performing a first compensation on the signal transmission link using a first compensation coefficient h(1) determined based on the initial observation information O(0) to obtain the signal state of the signal transmission link to obtain first observation information O(1); and, performing a second compensation on the signal transmission link using a second compensation coefficient h(2) determined based on the initial observation information O(0) to obtain the signal state of the signal transmission link to obtain second observation information O(2); wherein the second compensation coefficient h(2) and the first compensation coefficient h(1) have different values, and the second observation information O(2) and the first observation information O(1) have different values. Step E3: Obtain the ratio between the first difference and the initial observation information O(0) to obtain a first ratio d1; and obtain the ratio between the second difference and the initial observation information O(0) to obtain a second ratio d2; wherein the first difference is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2). Step E4: With the third observation information O(3) being 0 as a condition, determine the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2; specifically, construct a 2*2 matrix, wherein the first row of the 2*2 matrix records the values ​​of the real part and the imaginary part of the first ratio, and the second row records the values ​​of the real part and the imaginary part of the second ratio; calculate the product between the inverse matrix of the 2*2 matrix and the first matrix with 1 row and 2 columns to obtain a second matrix with 1 row and 2 columns, wherein the first row of the first matrix has a value of 1 and the second row has a value of 0, wherein the first row of the second matrix is ​​the value of the first coefficient x1 and the second row is the value of the second coefficient x2. Step E5: Calculate the product between the first coefficient x1 and the first compensation coefficient h(1) to obtain a first multiplication result; and calculate the product between the second coefficient x2 and the second compensation coefficient h(2) to obtain a second multiplication result; use the first multiplication result and the second multiplication result as the third compensation coefficient h(3).

[0348] For example, step 1: when h IQ When (0) = 0, we can estimate Step 2: When h IQ (1) = b = | O IQ (0)|, Step 3: When hour, Step 4: Subtract the measurements and normalize them to get:

[0349] Step 5: The Nature of Compensation:

[0350] In another optional embodiment, the method for LO leakage may include: Step F1: With the initial compensation coefficients of the I-path signal and the Q-path signal both being 0, the signal state of the signal transmission link is obtained to obtain initial observation information O(0). Step F2: With the first compensation coefficient of the I-path signal being a first value and the first compensation coefficient of the Q-path signal being 0, the signal transmission link is compensated for the first time to obtain the signal state of the signal transmission link to obtain first observation information O(1); and, with the second compensation coefficient of the I-path signal being 0 and the second compensation coefficient of the Q-path signal being a second value, the signal transmission link is compensated for the second time to obtain the signal state of the signal transmission link to obtain second observation information O(2). Step F3: Obtaining the ratio between the first difference and the initial observation information O(0) to obtain an in-line ratio CI; and obtaining the ratio between the second difference and the initial observation information O(0) to obtain a quadrature ratio CQ; wherein the first difference is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2). Step F4: Under the condition that the third observation information O(3) is 0, determine the third compensation coefficient h(3) of the in-phase signal and the third compensation coefficient h(3) of the quadrature signal according to the in-phase ratio CI and the quadrature ratio CQ;

[0351] Specifically, a 2*2 matrix is ​​constructed, wherein the first row of the 2*2 matrix records the values ​​of the real part of the inverse ratio CI and the orthogonal ratio CQ, and the second row records the values ​​of the imaginary part of the inverse ratio CI and the orthogonal ratio CQ;

[0352] Calculate the product between the inverse matrix of the 2*2 matrix and the first matrix with 1 row and 2 columns to obtain a second matrix with 1 row and 2 columns, wherein the value of the first row of the first matrix is ​​the value of the real part of the initial observation information O(0), and the value of the second row is the value of the imaginary part of the initial observation information O(0), wherein the first row of the second matrix is ​​the value of the inverse ratio CI, and the second row is the value of the orthogonal signal CQ.

[0353] For example, a multi-observation calibration-compensation algorithm for LO leakage has the following specific steps (it is necessary to perform FFT transformation on each measurement data and obtain Δ f Amplitude and phase information at h): Step 1: When h DC,I =h DC,Q = 0, it is estimated that O DC (0) = Tx LOL TF; where h DC,I It represents the ideal compensation coefficient of I-channel signal, hDC,Q represents the ideal compensation coefficient of Q-channel signal. Step 2: When h DC,I =DC I 、h DC,Q =0, O DC (1)=[Tx LOL +DC I α I cos(w m t+φ I )]·TF. Step 3: When h DC,I =0h DC,Q =DC Q When O DC (2)=[Tx LOL -DC Q α Q cos(w m t+φ Q )]·TF. Step 4: Subtract the measurements and normalize to obtain:

[0354] Step 5: The Nature of Compensation:

[0355] In another optional embodiment, for the HD3 compensation method, if the initial phase of the baseband (BB) signal is x degrees, then the initial phase of the subsequent 3BB signal needs to be 3x degrees, and O(n) represents the amplitude phase value of the frequency point corresponding to the 3BB signal divided by the amplitude phase value of the frequency point corresponding to the BB signal. Specifically: Step G1: With the initial compensation coefficient h(0) being 0, the signal state of the signal transmission link is obtained to obtain the initial observation information O(0). Step G2: The signal transmission link is compensated for the first time using the first compensation coefficient h(1) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and the first observation information O(1); and, the signal transmission link is compensated for the second time using the second compensation coefficient h(2) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and the second observation information O(2); wherein the second compensation coefficient h(2) and the first compensation coefficient h(1) have different values, and the second observation information O(2) and the first observation information O(1) have different values. Step G3: Obtain the ratio of the first difference value to the initial observation information O(0) to obtain a first ratio d1; and obtain the ratio of the second difference value to the initial observation information O(0) to obtain a second ratio d2; wherein the first difference value is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference value is the difference between the initial observation information O(0) and the second observation information O(2). Step G4: Under the condition that the third observation information O(3) is 0, determine the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) based on the first ratio d1 and the second ratio d2. Specifically, a 2*2 matrix is ​​constructed, wherein the first row of the 2*2 matrix records the values ​​of the real part and the imaginary part of the first ratio, and the second row records the values ​​of the real part and the imaginary part of the second ratio; the product between the inverse matrix of the 2*2 matrix and the first matrix with 1 row and 2 columns is calculated to obtain a second matrix with 1 row and 2 columns, wherein the value of the first row of the first matrix is ​​1 and the value of the second row is 0, wherein the first row of the second matrix is ​​the value of the first coefficient x1, and the second row is the value of the second coefficient x2. Step G5: Calculate the product between the first coefficient x1 and the first compensation coefficient h(1) to obtain a first multiplication result; and, Calculate the product between the second coefficient x2 and the second compensation coefficient h(2) to obtain a second multiplication result; and use the first multiplication result and the second multiplication result as the third compensation coefficient h(3).

[0356] For example, step 1: when h HD3 When (0) = 0, we can estimate

[0357] Step 2: When h HD3 (1) = b = | O HD3 (0)|,

[0358] Step 3: When hour,

[0359] Step 4: Subtract the measurements and normalize them to get:

[0360] Step 5: The Nature of Compensation

[0361] The architecture of the transmitting link and / or receiving link based on the digital phase shifter architecture described in the embodiments of the present disclosure, in addition to being used to transmit and / or receive signals in electromagnetic wave sensors, can also address the problems of unequal antenna feed line lengths between different transmitting / receiving links, unequal TXLO feed line lengths (i.e., between the LO and the mixers in each transmitting channel), and use the architecture of the transmitting link and / or receiving link of the digital phase shifter architecture described in the embodiments of the present disclosure as an auxiliary link to calibrate the receiving link and / or transmitting link of the analog phase shifter architecture.

[0362] It should be noted that the electromagnetic waves in the present disclosure include radio waves and light waves. Radio waves include short waves, medium waves, long waves and microwaves, etc. Microwaves include centimeter waves (i.e. electromagnetic waves in the range of 3GHz to 30GHz, such as electromagnetic waves in the range of 3.1GHz to 10.6GHz, electromagnetic waves in the 24GHz frequency band, etc.) and millimeter waves (i.e. electromagnetic waves in the range of 30GHz to 300GHz, such as electromagnetic waves in the 60GHz frequency band, electromagnetic waves in the 77GHz frequency band (such as 77GHz to 81GHz, etc.)). Light waves may include ultraviolet rays, visible light, infrared rays, and lasers, etc., among which the electromagnetic wave frequency band of lasers is (3.846 to 7.895)*10^5GHz, that is, lasers are included in part of the frequency bands of ultraviolet rays and visible light.

[0363] Figure 15 is a schematic diagram of a transmission link with at least two transmission channels in an embodiment of the present disclosure, Figure 16 is a structural schematic diagram of a digital LO signal generator in an embodiment of the present disclosure, and Figure 17 is a structural schematic diagram of a transmission link including a feeder unequal length compensation module in an embodiment of the present disclosure.

[0364] As shown in Figure 15, for a transmit link using at least two transmit channels, since each transmit channel (or transmit channel) is connected to the same local oscillator (LO) source, the feeder length (referred to as TXLO) between the LO signal and the mixer in each transmit channel is different, which in turn causes relative delay between different transmit channels. In addition, since the use of unequal transmit antenna feeders in an antenna array can achieve a better link budget, simpler antenna design and routing, lower inter-antenna coupling, better target angle resolution performance, smaller modules and costs, antenna designs using unequal transmit antenna feeders are becoming increasingly widely used, especially for applications such as close-range measurements, relatively enclosed spaces (such as indoor or cabin lights), or various scenarios using packaged antenna chips (AiP), such as those with wide frequency sweeps. However, due to the different transmit antenna feeder lengths between different transmit channels, it will cause relative delay issues for the RF signal passing through different transmit antennas.

[0365] In response to the above-mentioned problem of relative signal delay in the transceiver link caused by the different feeder lengths between different transmitting channels or between receiving channels, the embodiment of the present disclosure provides a compensation solution for the relative delay caused by the unequal feeder lengths based on the above-mentioned digital phase shift architecture, so as to effectively improve the quality of the transmitting and receiving signals.

[0366] Specifically, as shown in Figure 15, a digital phase shifter-based transmit antenna array includes four transmit channels, each sharing a waveform control module and a local oscillator (LO). Each transmit channel includes a transmit direct digital frequency synthesizer (TX DDFS), an IQ digital-to-analog converter (DAC), an analog low-pass filter (Analog LPF), an IQ mixer (Mixer), a power amplifier (PA), and a transmit antenna, all connected in sequence. Specifically, the waveform control module is connected to the TX DDFS in each transmit channel, while the LO is connected to an input of the mixer in each transmit channel to up-mix the phase-shifted signal output by the LPF in each transmit channel to form a radio frequency signal for transmission via each transmit antenna. To address the issue of unequal feeder lengths between the LO and the mixers in each transmit channel, and / or between the PAs in each transmit channel and their transmit antennas, pre-set compensation can be implemented in the transmit direct digital frequency synthesizer (TX DDFS) to reduce the transmit signal delay caused by these unequal feeder lengths.

[0367] For example, to address the issue of unequal feeder lengths between the local oscillator (LO) and the mixers in each transmit channel, the shortest feeder from the local oscillator (LO) to the mixer in one transmit channel can be used as a benchmark. The length difference between the mixers in the remaining transmit channels and the local oscillator (LO) compared to this benchmark can then be calculated. Because the propagation speed c of the transmitted signal is relatively fixed, the corresponding delay value can be obtained by dividing the length difference between the aforementioned transmit channels by the speed c. Subsequently, when transmitting the signal, the corresponding compensation can be directly performed in the TX DDFS to obtain the delay of each transmit channel compared to the benchmark. At the same time, a similar approach can be used to address the issue of unequal feeder lengths between the PAs of each transmit channel and their transmit antennas to obtain and compensate for the delay caused by feeder differences between the transmit channels.

[0368] When specifically compensating for the transmission link of the FMCW wave, a digital LO generator can be used to compensate for the delay μ caused by the feeder length compared to the reference, the sweep period T, the sweep bandwidth β, and the sweep start frequency f. c As shown in Figure 16, the DDFS uses a digital LO generator for compensation, i.e. one input is △θ=μ*β / T, and the other input is θ0=f c μ-μ 2 *β / 2T; where μ is the delay difference between the feeder corresponding to each transmitting channel and the reference feeder; at the same time, as shown in Figures 16-17, the output LO-I (i.e., cos(ωn+ψ)) of the digital LO generator as the signal compensator is used as the input of the I path in the DAC, and the output LO-Q (i.e., sin(ωn+ψ)) of the digital LO generator is used as the input of the Q path in the DAC.

[0369] It should be noted that Figures 16 and 17 illustrate the structural schematic diagrams of a transmission channel, while Figure 15 shows a transmission antenna array with four transmission channels. In actual applications, the transmission antenna array only needs to include at least two (such as two, three or five, etc.) transmission channels, and compensation operations can also be performed during the reception of signals and / or signal processing. At the same time, for a structure with one transmission channel and / or one reception channel, a similar concept can also be used to directly perform compensation in the TX DFFS or in the reception of signals and / or signal processing.

[0370] Regarding the related technical solutions for using the architecture of the transmitting link and / or receiving link of the digital phase shifter architecture described in the embodiments of the present disclosure as an auxiliary link to perform operations such as calibration on the receiving link and / or transmitting link of the analog phase shifter architecture and the receiving link and / or transmitting link of the digital phase shifter architecture, please refer to Figures 18 to 21.

[0371] FIG18 is a schematic diagram of calibrating and compensating a transceiver link using an auxiliary circuit in an embodiment of the present disclosure, and FIG19 is a schematic diagram of calibrating and compensating a receive link using an auxiliary transmit circuit in an embodiment of the present disclosure.

[0372] Specifically, the signal transmission and reception link may generally include antennas, RF analog devices, baseband analog devices and baseband digital processors, etc. RF analog devices may include phase-locked loops (PLLs), power amplifiers (PAs), phase shifters (PSs), low-noise amplifiers (LNAs), mixer local oscillators (LOs) and power detectors (PDs), etc., while baseband analog devices may include low-pass filters (LPFs), high-pass filters (HPFs), analog-to-digital converters (ADCs), etc.

[0373] Analog components and circuits can exhibit a variety of non-ideal conditions, which can also vary with temperature. For example, the frequency responses of the RF analog components and circuits in the transmit chain (TX), the baseband analog components and circuits in the receive chain (RX), and the RF analog components and circuits in the receive chain (RX) may all be unbalanced. PD accuracy issues may exist at both the PA output and LNA input ports, and local oscillator (LO) leakage may also exist in the receive chain (RX). Furthermore, when analog components operate at different frequencies, their amplitude-frequency responses vary, effectively adding an unexpected window function to the received signal. This can adversely affect subsequent operations such as target range and velocity estimation. This can even lead to false target detection or missed targets in multi-target scenarios. Furthermore, in scenarios with at least two links, the frequency responses of different receive chains can vary, introducing additional noise, such as phase and amplitude variations. This introduced phase, in particular, can directly affect the subsequent estimation of the target's echo direction, reducing the accuracy of target angle estimation.

[0374] In some embodiments, the errors of analog devices and circuits can be measured by external equipment, and then certain compensation operations can be performed on them, such as bench calibration or ATE calibration, to calibrate the performance of analog devices and circuits of the receiving link.

[0375] The disclosed embodiments also provide a solution for calibrating the transmit link and / or receive link based on an auxiliary link method, so as to achieve real-time calibration of analog devices and circuits while achieving precise calibration without the need for external equipment, thereby effectively reducing the impact of changes in RF device parameters due to environmental changes.

[0376] As shown in Figure 18, the electromagnetic wave signal transmission and reception link can be calibrated in real time by setting up an auxiliary calibration circuit (Auxiliary). The auxiliary calibration circuit (i.e., auxiliary calibration link) can be integrated into the electromagnetic wave sensor, thereby achieving real-time and accurate calibration without the need for external equipment, and effectively reducing the impact of changes in RF device parameters due to environmental changes.

[0377] For example, for the transmit path, an auxiliary receiver (ARX) can be positioned adjacent to the transmit path in the sensor to enable real-time calibration of the transmit path. Similarly, for the receive path, an auxiliary transmitter (ATX) can be positioned adjacent to the receive path (i.e., the signal receiving link) in the sensor to enable real-time calibration of the receive path. Furthermore, an auxiliary path can be positioned between two paths to be calibrated, allowing one auxiliary path to be reused for calibration of at least two different paths. As shown in FIG18 , a shared ARX is positioned between the two transmit paths, while a shared ATX is positioned between the two receive paths. This further avoids issues such as excessive wiring or excessively long lines. In an optional embodiment, for the four-transmit, four-receive antenna array shown in FIG18 , two ARXs and two ATXs can be positioned. Specifically, each ARX is positioned between the two transmit paths, while each ATX is positioned between the two receive paths. This effectively reduces wiring complexity and improves the real-time nature of calibration.

[0378] As shown in Figure 18, since the receive path can be divided into the RF portion (i.e., RF Rx) and the baseband portion (i.e., Rx BB), auxiliary calibration paths can be set up for RF Rx and Rx BB to further improve calibration accuracy. For example, an RF auxiliary transmitter unit (RF ATX) can be set up for RF Rx, and an IF ATX can be set up for Rx BB. Furthermore, for the ARX path, an RF tone generator can be set up to calibrate the ARX, and then the calibrated ARX can be used to calibrate the transmit path (TX).

[0379] In some optional embodiments, as shown in FIG18 , for the RF Rx and Rx BB in the receiving path (Receiver), the Rx BB may be calibrated using the IF ATX first, then the RF ATX may be calibrated using the calibrated Rx BB, and finally the Rx BB may be calibrated using the calibrated RF ATX, thereby achieving calibration of the entire receiving path.

[0380] Based on the structure shown in Figure 18 and in conjunction with the content shown in Figure 19, the receiver calibration scheme is described. Specifically, the receiver path may include sequentially connected components such as a receive antenna (or receive port), LNA, mixer, TIA, LPF, HPF, real ADC, and a PD connected to the LNA input port. Based on the mixer, the receiver path can be divided into RF Rx and Rx BB. In other words, along the direction of signal transmission, the mixer and the portion preceding it are defined as RF Rx, and the remaining portion can be defined as Rx BB. As shown in Figure 19, the RF Rx may include a receive antenna (or receive port), LNA, real mixer, and PD.

[0381] The IF ATX corresponding to the Rx BB of the aforementioned receive path may include a frequency divider (e.g., Freq Divider 1 / N) and a Real DAC (e.g., a 1-bit Real DAC) connected in sequence, with the output of the Real DAC connected to the output of the Real Mixer. As shown in Figure 19, the IF ATX may include a frequency divider (e.g., Freq Divider 1 / N) and a 1-bit digital-to-analog converter (e.g., a 1-bit Real DAC) connected in sequence. This allows the IF ATX to stably generate a single-tone signal (single tone) in a variety of environments, and this single-tone signal can be freely configured to a variety of frequencies.

[0382] In some optional embodiments, to improve calibration accuracy, the IF ATX may be calibrated before calibrating the Rx BB. For example, this may involve calibrating the intermediate frequency (IF) frequency response and DC calibration within the IF ATX. Furthermore, after the IF ATX is calibrated, the calibrated IF ATX can be used to calibrate the Rx BB of the receive path. For example, the calibrated IF ATX can be used to transmit single-tone signals of different frequencies to perform calibration and compensation operations on the Rx BB. Alternatively, the frequency (e.g., tens of MHz) of the single-tone signal transmitted by the IF ATX can be determined based on the sampling frequency of the real ADC in the receive path.

[0383] The RF ATX configured for the RF Rx of the receive channel may include a TX DDFS, an IQ imbalance compensation unit (TX IQ Comp), an adder, an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an amplifier (e.g., a PA), a multiplier, a local oscillator (LO, not shown), and a squarer (x^2), connected in sequence. The output of the multiplier is connected to the output of the mixer in the receive path via the squarer to form a first calibration branch. The output of the multiplier is also connected to the link between the PD and the transmit antenna in the receive path to form a second calibration branch. The IQ imbalance compensation unit may be configured to compensate for IQ imbalance in the RF ATX, the adder may be configured to compensate for local oscillator leakage (LO leakage) in the RF ATX, and the squarer may be configured to compensate for residual sideband effects caused by IQ imbalance in the RF ATX.

[0384] In some optional embodiments, to improve calibration accuracy, the RF ATX may be calibrated before the RF ATX calibrates the Rx ATX. For example, calibration operations may be performed to address issues such as local oscillator leakage, RF frequency response, and IQ imbalance within the RF ATX. Furthermore, after the RF ATX is calibrated, the calibrated RF ATX may be used to calibrate the RF Rx in the receive path. For example, single-tone signals of different frequencies may be transmitted through the calibrated RF ATX. The baseband signal processing module in the receive chain then calibrates and compensates for the power detector (PD) at the input of the LNA in the RF portion of the receive path, the total gain from the LNA to the ADC in the receive path, and auxiliary calibration of the frequency response.

[0385] In some optional embodiments, when calibrating a receiver path using an IF ATX and an RF ATX, the IF ATX may be calibrated first, and the calibrated IF ATX may be used to calibrate the Rx BB in the receiver path. Then, the calibrated Rx BB may be used to calibrate the RF ATX via a first calibration branch of the RF ATX (i.e., through a squarer). Finally, the second calibration branch of the calibrated RF ATX may be used to calibrate the RF Rx in the receiver path, thereby achieving calibration and compensation operations for the entire receiver path.

[0386] Specifically, as shown in Figure 19, the Rx BB and Real DAC can be calibrated using a 1-bit Real DAC. The LO leakage and IQ imbalance in the RF ATX can then be calibrated using the calibrated RxBB. Finally, the calibrated RF ATX transmit signal can be input to the LNA in the receive path to correct the RF Rx's RF LO leakage and RF frequency response.

[0387] FIG20 is a schematic diagram of calibrating and compensating the transmission link using an auxiliary receiving circuit in an embodiment of the present disclosure. As shown in FIG20 , the transmission path (Transmitter) may include a phase shift module PS, an amplifier PA, a power detector PD, etc. For example, the transmission path may adopt the transmission link of the digital phase shifter architecture (Digital Phase Shifter) described in any embodiment of the present disclosure. For details, please refer to the relevant figures and text descriptions, which will not be described in detail here. Because the transmission path adopts a digital phase shifter architecture, while achieving more accurate phase shifting operations, the transmission channel can simultaneously support multiple modes such as DDM and FDM (Frequency Division Multiplexing) of multiple antennas, and can also eliminate the calibration operation of the RF phase shifter (Phase Shifter), reduce the isolation and coupling in the phase shift system, and reduce link loss and production costs. In addition, to address possible TX IQ mismatch and LO leakage issues, the transmit path of this digital phase shifter architecture can also support RF frequency response compensation, IQ imbalance, and LO leakage calibration in the digital domain.

[0388] To address issues such as TX IQ mismatch, LO leakage, and frequency response in the transmit path, an auxiliary receiver (ARX) can be configured to perform relevant calibration and compensation operations. As shown in Figure 20, the ARX can include a mixer, TIA, LPF, HPF, IQ ADC, adder, and RF calibration module (RF Calib) connected in sequence. One input of the mixer receives the ARX IQ LO signal, while the other input is connected along the signal transmission direction (i.e., the direction of the arrow in the figure) to a node before the transmit path PD or to any node after the phase shifter (module). For example, it can be connected to the output of the PA (synchronously calibrating the PA) or the input of the PA, thereby calibrating the transmit path through the ARX. The LO signal frequency in the transmit path and the ARX IQ LO signal frequency have a set difference frequency, so that there is a frequency offset between the two signals, simulating a real transmit and receive signal loop.

[0389] In an optional embodiment, to further improve calibration accuracy, a corresponding calibration circuit (i.e., a calibration receiving unit) may be provided for the ARX (i.e., the auxiliary receiving unit). For example, the RF tone signal generation circuit (RF Tone Generator) shown in FIG20 may include a TX DDFS, an adder, a real DAC, an LPF, an amplifier, a multiplier, and a bandpass filter (BPF) connected in sequence. The adder can be configured to calibrate and compensate for TX LO leakage, the multiplier can be configured to compensate for RF tone generator LO leakage, and the BPF can be configured to filter out DC signals generated by LO leakage from the RF tone generator. In other words, the RF tone generator can be configured to generate multiple stable tone signals of different frequencies to implement calibration operations for the ARX.

[0390] In some optional embodiments, as shown in FIG20 , the ARX may be calibrated using an RF Tone Generator first, and then the calibrated ARX may be used to calibrate a transmitter channel including a PA, for example, calibrating the PD at the PA output, the phase shifter in the transmit channel, the total gain from the DAC to the PA output, and the frequency response and other devices and circuits.

[0391] Specifically, as shown in FIG20 , the RF Tone Generator can be used to generate multiple stable single-tone signals of different frequencies to assist in calibrating the ARX. Then, based on the calibrated ARX, the IQ imbalance, local oscillator leakage, inconsistent frequency response, and other issues of the transmit path TX can be calibrated.

[0392] Figure 21 is a schematic diagram of the structure of an auxiliary circuit in an embodiment of the present disclosure, Figure 22 is a schematic diagram of the structure of another auxiliary circuit in an embodiment of the present disclosure, and Figure 23 is a schematic diagram of the circuit module of the IQ Mixer in an embodiment of the present disclosure; Figure 24 is a schematic diagram of the structure of an IQ Mixer in an embodiment of the present disclosure, and Figure 25 is a schematic diagram corresponding to the structure shown in Figure 24.

[0393] In some optional embodiments, based on the structures and related descriptions shown in Figures 18 and 19, the ATX can also be implemented using a squarer combined with a multi-bit DAC. As shown in Figure 21, the receive path includes an LNA, a mixer, a TIA, a HPF, an ADC, a BB processor, and a local oscillator (LO) connected in sequence. Specifically, the mixer uses a first LO signal provided by the LO to downconvert the echo signal provided by the LNA to produce an intermediate frequency (IF) signal (i.e., an analog baseband signal). The ATX may include a first DAC (i.e., DAC1), a mixer, and a squarer (x^2) connected in sequence. The first DAC provides an analog signal to the mixer's receiving end in the ATX. The mixer's other receiving end can be connected to a local oscillator in the receive path, where it mixes the analog echo signal output by the first DAC with a second LO signal provided by the local oscillator, thereby outputting an analog IF signal to the input of the squarer. The squarer then connects the processed analog IF signal to the input of the TIA to calibrate circuits and components in the receive path, such as the TIA, HPF, and ADC. In some other optional embodiments, the ATX can also output the analog echo signal directly to the receiving end of the squarer through the DAC, and the squarer sends the processed analog echo signal to the input end of the LNA to calibrate the LNA, mixer, TIA, HPF, ADC and other circuits and components in the receiving path.

[0394] In some optional embodiments, as shown in FIG22 , based on the structure shown in FIG21 , a second DAC may be provided for a receive path including a VGA, an SDM unit, and an ADC with a decimation filter. The output of the second DAC may be connected to any node in the receive path between the mixer and the TIA, between the TIA and the HPF, between the HPF and the VGA, or between the VGA and the ADC to calibrate the corresponding circuits and components. Optionally, the first DAC may be a multi-bit (e.g., 10-bit) DAC, and the second DAC may be a 1-bit DAC.

[0395] The above-mentioned ATX composed of DAC and squarer can be reasonably modified by those skilled in the art based on the above-mentioned contents. The present disclosure does not limit the specific implementation method, as long as it can achieve the same or similar functions as the ATX in the embodiment of the present disclosure.

[0396] In some optional embodiments, when setting up an ATX near a receiving path, the ATX can be set up by utilizing the gaps between the receiving paths, and at the same time, two or more receiving paths can share one ATX; similarly, when setting up an ARX near a transmitting path, the ARX can be set up by utilizing the gaps between the transmitting paths, and at the same time, two or more transmitting paths can share one ARX. Optionally, the above-mentioned ATX and / or ARX can be operated intermittently. For example, the ATX can transmit a calibration tone signal in the gaps between the working of the receiving path (such as between frames or chirps) to effectively calibrate and compensate the receiver in real time. Similarly, the ARX can also use a tone generator to generate a calibration tone signal in the gaps between the working of the receiver (such as between frames or chirps) to effectively calibrate the ARX first. The transmitter is then effectively calibrated using the calibrated ARX to ensure that calibration and compensation operations can be performed in a preset manner.

[0397] With respect to the scheme for implementing transmit path and / or receive path calibration based on an auxiliary link as shown in Figures 18 to 22, based on the adoption of an auxiliary link, auxiliary circuits are added to integrated circuits such as chips (chips or dies) to assist in calibrating the main path circuits, thereby effectively improving the performance of analog circuits and modules. At the same time, at least some RF circuits and devices can be calibrated in real time online (on the fly), thereby effectively improving the calibration performance, thereby effectively improving the RF performance while reducing the difficulty of RF implementation.

[0398] Because mixers are crucial components for frequency conversion in transceiver links, they are widely used in radio devices such as communications and radar. Examples include single-sideband mixers, which effectively suppress image signals, and IQ mixers in transceiver links in the disclosed embodiments. An IQ mixer (such as a single-sideband mixer) transmits signals through two branches: an I branch and a Q branch, each with a 90° phase difference.

[0399] For application scenarios with relatively compact layout areas, such as high-frequency sensor applications in the millimeter wave band, the physical distance between the I branch and the Q branch, and between the input branch and the output branch before mixing are short. As a result, electromagnetic wave signals may leak between branches, between input and output ports, and between the IQ matching network and the mixer output through methods such as magnetic coupling, substrate coupling, and electrical coupling, thereby causing serious deterioration in the mixer's image rejection ratio and local oscillator leakage.

[0400] In an optional embodiment, based on the mixer output network ensuring power synthesis of the IQ branches while also meeting impedance matching requirements, the present disclosure provides a new mixer structure, that is, by improving the mixer output passive network, the isolation between the IQ branches is effectively improved while effectively reducing the leakage of the local oscillator signal.

[0401] FIG23 is a schematic diagram of the circuit module of the IQ mixer in an embodiment of the present disclosure. As shown in FIG23, an IQ mixer may include an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit, wherein the I-branch mixing unit may be configured to output an I-branch signal, the Q-branch mixing unit may be configured to output a Q-branch signal, and the transformer unit may be configured to magnetically couple the I-branch signal and the Q-branch signal to synthesize an IQ mixing output signal, and transmit the IQ mixing output signal to a subsequent circuit (Next Block). The transformer unit may be arranged between the I-branch mixing unit and the Q-branch mixing unit to increase the physical distance between the I-branch mixing unit and the Q-branch mixing unit relative to a short-circuited IQ mixer, thereby effectively reducing the coupling between the I-branch mixing unit and the Q-branch mixing unit, thereby achieving the purpose of improving the IQ mixer rejection ratio, while also facilitating the layout design of the circuit board.

[0402] A short-circuit IQ mixer is constructed by short-circuiting the outputs of the I and Q branches of the mixer, and then matching them to the subsequent circuit through a matching network, that is, synthesizing the signals through electrical coupling. As shown in FIG23 , in an IQ mixer according to an optional embodiment of the present disclosure, a branch inductor is connected in series between the output terminals of the I branch mixer and the output terminals of the Q branch mixer, and a magnetic coupling inductor is subsequently provided between the two branch inductors, thereby forming a three-turn transformer structure. The output of the I branch mixer and the output of the Q branch mixer are magnetically coupled and synthesized through magnetic coupling to obtain the aforementioned IQ mixing output signal. The common modes of the I and Q branch mixers are not directly connected, so they can be adjusted separately, making the device more flexible in application scenarios. The signals are then directly transmitted to the subsequent circuit (next block) through the two ends of the magnetic coupling inductor as output terminals. This three-turn transformer structure has four input terminals and two output terminals.

[0403] Based on the three-turn transformer structure shown in Figure 23, if the LO signal and the signal to be mixed (such as an echo signal) enter the P and N terminals of the IQ branches of the three-turn transformer, the current and magnetic field directions formed on the three-turn transformer are shown in Figure 24. At this time, the Isb magnetic field is superimposed and the USB magnetic field is offset, thereby outputting a high-quality single-sideband signal. This three-turn transformer structure can achieve down-mixing or up-mixing of the signal to be mixed.

[0404] In some optional embodiments, even in a scenario where there is a phase difference between the LO signal and the signal to be mixed, since the signals are canceled in the same manner on the three-turn transformer, when there is no imbalance between the LO signal and the signal to be mixed, the output of the three-turn transformer is still a single-sideband signal.

[0405] The magnetically coupled IQ mixer provided in the embodiment of the present disclosure performs power synthesis through magnetic coupling, compared to the short-circuited IQ mixer that uses electrical coupling to synthesize power. At the same time, for the I and Q branches, the magnetically coupled IQ mixer can be completely consistent and can be axially symmetrical, without introducing additional phase errors due to the inconsistent wiring lengths of the I and Q branches, thereby making the related circuits (such as millimeter wave and other high-frequency circuits) more robust in subsequent manufacturing processes and more resistant to process instabilities. For example, the magnetically coupled IQ mixer in the embodiment of the present disclosure can be set to the midline between the P and N ports as shown in Figure 25, so that the structure of the mixer can be axially symmetrical based on the midline, thereby making the common-mode path lengths of the P port and the N port equal, that is, the PN symmetry in the magnetically coupled IQ mixer is better than the PN symmetry in the short-circuited IQ mixer, thereby making the leakage of the local oscillator signal smaller.

[0406] In some optional embodiments, the magnetically coupled IQ mixers provided in the embodiments of the present disclosure can be applied to various circuits, such as transmit channels and on-chip self-test (OST), as upconversion mixers in various electromagnetic wave circuits. For example, the IQ mixers in various transmit channels (or transmit links) and BIST in the embodiments of the present disclosure can employ the magnetically coupled IQ mixers in the embodiments of the present disclosure.

[0407] FIG26 is a schematic diagram of the physical structure of another IQ mixer according to an embodiment of the present disclosure. As shown in FIG26 , an embodiment of the present disclosure also provides another structure of a magnetically coupled IQ mixer, in which the tap feed lines of the I and Q branches partially overlap. For example, for the I branch, the tap of the I branch can bypass the Q branch (e.g., through a via) and extend to the I branch before being connected to the common-mode bias voltage. Similarly, the tap of the Q branch can bypass the I branch (e.g., through a via) and extend to the Q branch before being connected to the common-mode bias voltage. The above-mentioned routing method can make the path of the common-mode path more certain and less susceptible to interference from other circuits. At the same time, the three-turn transformer in the IQ mixer according to an embodiment of the present disclosure can be square, octagonal, regular octagonal, etc., that is, as long as it can be axially symmetrically distributed along the center between the NPs in a top view or bottom view.

[0408] As shown in Figure 27, an embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission chain. The signal transmission method includes: step 2701, determining the phase of the radio frequency transmission signal of each signal transmission chain; step 2702, determining the initial phase of the baseband signal in each of the signal transmission chains according to the phase of the radio frequency transmission signal; step 2703, generating an initial baseband signal according to the determined initial phase; step 2704, compensating the initial baseband signal using pre-acquired compensation information to obtain the baseband signal; step 2705, performing a phase shift operation on the transmitting end local oscillator signal based on the baseband signal to obtain the radio frequency transmission signal.

[0409] The signal transmission method of the disclosed embodiment generates a digital baseband signal in the digital domain by using a digital phase shifter architecture, which has better orthogonality and lower sidelobes. Its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of low link loss, low cost, and no need for offline calibration. It can also support more flexible wave transmission schemes, such as high-performance Doppler multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0410] As shown in Figure 28, an embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link. The signal transmission method includes: step 2801, obtaining observation information of the main signal transmission path under the current compensation coefficient, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to the compensation coefficient, so as to achieve target detection and / or communication. Step 2802, when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition. Step 2803, using the compensation coefficient to compensate the baseband signal. Step 2804, performing a phase shift operation on the transmitting end local oscillator signal based on the compensated baseband signal to obtain the radio frequency transmission signal.

[0411] The signal transmission method, signal calibration link, transmission link and method, transceiver link, and integrated circuit of the disclosed embodiments are capable of compensating the generated signal with compensation information generated by the signal calibration link, addressing quadrature imbalance, LO leakage, and harmonic distortion, thereby effectively improving signal quality in the main signal transmission path. The required compensation coefficient for the main signal transmission path is determined iteratively, effectively ensuring the accuracy of the functional compensation coefficient.

[0412] As shown in FIG29, an embodiment of the present disclosure further provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link. The signal transmission method includes: step 2901, determining the initial observation information O(0), first observation information O(1), and second observation information O(2) corresponding to the signal transmission main path under the conditions of different initial compensation coefficients h(0), first compensation coefficients h(1), and second compensation coefficients h(2), wherein the signal transmission main path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to the compensation coefficients, so as to achieve target detection and / or communication. Step 2902, using the initial observation information O(0), first observation information O(1), and second observation information O(2), determine a third compensation coefficient h(3). Step 2903, using the compensation coefficient to compensate the baseband signal. Step 2904, performing a phase shift operation on the transmitting end local oscillator signal based on the compensated baseband signal to obtain the radio frequency transmission signal.

[0413] The signal transmission method, signal calibration link, transmission link and method, transceiver link, and integrated circuit of the disclosed embodiments are capable of compensating the generated signal with compensation information generated by the signal calibration link, addressing issues such as quadrature imbalance, LO leakage, and harmonic distortion, thereby effectively improving signal quality in the main signal transmission path. Multiple calibration operations generate observation information to determine the compensation coefficient required for the main signal transmission path, effectively ensuring the accuracy of the compensation coefficient.

[0414] In some exemplary embodiments, the transmitting digital baseband signal is a single-tone signal and the transmitting local oscillator signal is a swept-frequency signal; or, the transmitting digital baseband signal is a swept-frequency signal and the transmitting local oscillator signal is a single-tone signal. In some exemplary embodiments, the swept-frequency signal has a bandwidth greater than 2 GHz.

[0415] The present disclosure also provides an integrated circuit, which may include a radio frequency module, an analog signal processing module, and a digital signal processing module connected in sequence; the radio frequency module is used to generate a radio frequency transmission signal and receive a radio frequency reception signal; the analog signal processing module is used to down-convert the radio frequency reception signal to obtain an intermediate frequency signal; the digital signal processing module is used to perform analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal; and the radio frequency module includes any of the above-mentioned signal transmission links, the above-mentioned signal transceiver links, any of the above-mentioned signal calibration links, any of the above-mentioned signal compensation links, any of the above-mentioned signal calibration systems, and / or the above-mentioned IQ mixers; and / or the digital signal processing module performs compensation in the digital domain based on the above-mentioned feeder unequal length compensation method; wherein the radio frequency reception signal is an echo signal formed by the radio frequency transmission signal being emitted and / or scattered by the target. Optionally, the integrated circuit may also include a data processing module, which is used to process the digital signal to achieve target detection and / or wireless communication. For example, the integrated circuit may be a millimeter wave radar chip (chip or die).

[0416] In some optional embodiments, the integrated circuit may be an AiP (Antenna-In-Package) chip structure, an AoP (Antenna-On-Package) chip structure, an AoC (Antenna-On-Chip) chip structure, or a RoP (Radiator on Package, i.e., by setting a radiating structure (Radiator) on the package (Package), and surrounding the Radiator with a sphere to form a waveguide structure, the RF signal transitions to the waveguide structure through the radiating structure, and then is converted from the waveguide structure to an external antenna) structure.

[0417] According to other embodiments of the present disclosure, an electromagnetic wave sensor is also proposed. The electromagnetic wave sensor may include an antenna, and an integrated circuit as described above. The integrated circuit is electrically connected to the antenna for transmitting and receiving electromagnetic wave signals. For example, the electromagnetic wave sensor may include: a carrier, an integrated circuit and an antenna as described in any of the above embodiments, etc. The integrated circuit may be provided on the carrier; the antenna may be provided on the carrier, or integrated with the integrated circuit as an integral device provided on the carrier (that is, the antenna may be an antenna provided in an AiP, AoP or AoC structure in this case); the integrated circuit is connected to the antenna (that is, the sensor chip or integrated circuit is not integrated with an antenna in this case, such as a conventional SoC, etc.) for transmitting and receiving electromagnetic wave signals. The carrier may be a printed circuit board PCB, and the first transmission line may be a PCB trace.

[0418] An embodiment of the present disclosure provides a device, which may include: a device body; and an electromagnetic wave sensor as described above, which is arranged on the device body; wherein the electromagnetic wave sensor is used for target detection and / or communication to provide reference information for the operation of the device body.

[0419] The present disclosure also provides an electronic device that can be expressed in the form of a general-purpose computing device. The components of the electronic device may include, but are not limited to, at least one processing unit, at least one storage unit, a bus connecting different system components (including a storage unit and a processing unit), a display unit, and the like. The storage unit stores program code, and the program code can be executed by the processing unit so that the processing unit executes the method described in this specification according to various exemplary embodiments of the present disclosure. The storage unit may include a readable medium in the form of a volatile storage unit, such as a random access memory unit (RAM) and / or a cache memory unit, and may further include a read-only memory unit (ROM).

[0420] The storage unit may also include a program / utility having a set (at least one) of program modules, such program modules including but not limited to: an operating system, one or more application programs, other program modules, and program data, each of which or some combination may include an implementation of a network environment.

[0421] The bus can represent one or more of several types of bus structures, including a memory unit bus or memory unit controller, a peripheral bus, an accelerated graphics port, a processing unit, or a local bus using any of a variety of bus architectures.

[0422] The electronic device may also communicate with one or more external devices (e.g., keyboards, pointing devices, Bluetooth devices, etc.), one or more devices that enable a user to interact with the electronic device, and / or any device that enables the electronic device to communicate with one or more other computing devices (e.g., routers, modems, etc.). Such communication may be performed via an input / output (I / O) interface. Furthermore, the electronic device may also communicate with one or more networks (e.g., local area networks (LANs), wide area networks (WANs), and / or public networks, such as the Internet) via a network adapter. The network adapter may communicate with other modules of the electronic device via a bus. It should be understood that, although not shown in the figures, other hardware and / or software modules may be used in conjunction with the electronic device, including but not limited to: microcode, device drivers, redundant processing units, external disk drive arrays, RAID systems, tape drives, and data backup storage systems.

[0423] For example, the electronic device in the embodiment of the present disclosure may further include: a device body; and an electromagnetic wave sensor as described in any of the above embodiments, which is arranged on the device body; wherein the electromagnetic wave sensor can be used to realize functions such as target detection and / or wireless communication.

[0424] Specifically, based on the above embodiment, in an optional embodiment of the present disclosure, the electromagnetic wave sensor can be disposed outside the device body or inside the device body. In other optional embodiments of the present disclosure, the electromagnetic wave sensor can also be disposed partially inside the device body and partially outside the device body. This embodiment of the present disclosure is not limited to this, and the specific configuration may vary depending on the circumstances.

[0425] In an optional embodiment, the above-mentioned device body can be a component and product used in fields such as smart cities, smart homes, transportation, smart homes, consumer electronics, security monitoring, industrial automation, cabin detection (such as smart cockpits), medical equipment and health care. For example, the device body can be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), liquid level / flow rate detection equipment, smart wearable devices (such as bracelets, glasses, etc.), smart home devices (such as sweeping robots, door locks, televisions, air conditioners, smart lights, etc.), various communication devices (such as mobile phones, tablets, etc.), as well as gates, smart traffic lights, smart signs, traffic cameras and various industrial robotic arms (or robots), etc. It can also be various instruments for detecting life characteristic parameters and various devices equipped with the instruments, such as life characteristic detection in car cabins, indoor personnel monitoring, smart medical equipment, consumer electronic devices, etc.

[0426] The embodiment of the present disclosure further provides a non-transitory computer-readable storage medium having computer-readable instructions stored thereon. When the instructions are executed by a processor, the processor executes the feeder unequal length compensation method as described above.

[0427] Through the description of the above embodiments, it is easy for those skilled in the art to understand that the example embodiments described here can be implemented by software or by combining software with necessary hardware. The technical solution according to the embodiments of the present disclosure can be embodied in the form of a software product, which can be stored in a non-volatile storage medium (which can be a CD-ROM, a USB flash drive, a mobile hard disk, etc.) or on a network, and includes several instructions to enable a computing device (which can be a personal computer, a server, or a network device, etc.) to execute the above method according to the embodiments of the present disclosure.

[0428] The software product can adopt any combination of one or more readable media. The readable medium can be a readable signal medium or a readable storage medium. The readable storage medium can be, for example, but not limited to, a system, device or component of electricity, magnetism, light, electromagnetic, infrared, or semiconductor, or any combination thereof. More specific examples (non-exhaustive list) of readable storage media include: an electrical connection with one or more wires, a portable disk, a hard disk, a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or flash memory), an optical fiber, a portable compact disk read-only memory (CD-ROM), an optical storage device, a magnetic storage device, or any suitable combination thereof.

[0429] Computer-readable storage media may include a data signal propagated in baseband or as part of a carrier wave, which carries readable program code. This propagated data signal may take a variety of forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination thereof. The readable storage medium may also be any readable medium other than a readable storage medium, which may transmit, propagate, or transfer a program for use by or in conjunction with an instruction execution system, apparatus, or device. The program code contained on the readable storage medium may be transmitted using any suitable medium, including but not limited to wireless, wired, optical cable, RF, etc., or any suitable combination thereof.

[0430] The program code for performing the operations of the present disclosure may be written in any combination of one or more programming languages, including object-oriented programming languages ​​such as Java, C++, and the like, as well as conventional procedural programming languages ​​such as "C" or similar programming languages. The program code may be executed entirely on the user computing device, partially on the user device, as a stand-alone software package, partially on the user computing device and partially on a remote computing device, or entirely on a remote computing device or server. In cases involving a remote computing device, the remote computing device may be connected to the user computing device via any type of network, including a local area network (LAN) or a wide area network (WAN), or may be connected to an external computing device (e.g., via the Internet using an Internet service provider).

[0431] The computer-readable medium carries one or more programs. When the one or more programs are executed by the device, the computer-readable medium implements the aforementioned functions.

[0432] Those skilled in the art will appreciate that the modules described above can be distributed in the device according to the description of the embodiment, or can be modified accordingly to be used in one or more devices that are different from the embodiment. The modules of the above embodiment can be combined into one module or further divided into multiple submodules.

[0433] According to an embodiment of the present disclosure, a computer program is proposed, including a computer program or an instruction, which, when executed by a processor, can execute the method described above. In an optional embodiment, the above-mentioned integrated circuit can be a millimeter-wave radar chip. The type of digital function module in the integrated circuit can be determined according to actual needs. For example, in a millimeter-wave radar chip, the data processing module can be used for functions such as range-dimensional Doppler transform, velocity-dimensional Doppler transform, constant false alarm detection, direction of arrival detection, point cloud processing, etc., to obtain information such as the distance, horizontal angle, pitch angle, speed, altitude, micro-Doppler motion characteristics, shape, size, surface roughness and dielectric properties of the target.

[0434] It should be noted that radio devices can achieve functions such as target detection and / or communication by transmitting and receiving radio signals to provide detection target information and / or communication information to the device body, thereby assisting or even controlling the operation of the device body.

[0435] For example, when the above-mentioned device body is applied to an advanced driver assistance system (i.e., ADAS), the radio device (such as millimeter-wave radar) as an on-board sensor can assist the ADAS system to realize application scenarios such as adaptive cruise control, automatic braking assistance (i.e., AEB), blind spot detection warning (i.e., BSD), assisted lane change warning (i.e., LCA), reversing assistance warning (i.e., RCTA), parking assistance, rear vehicle warning, collision avoidance, pedestrian detection, etc.

[0436] The technical features of the above-mentioned embodiments can be combined arbitrarily. In order to make the description concise, not all possible combinations of the technical features in the above-mentioned embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0437] The above-described embodiments merely express preferred embodiments of the present invention and the technical principles employed. While the descriptions are relatively specific and detailed, they should not be construed as limiting the scope of the invention. It is apparent to those skilled in the art that various changes, readjustments, and substitutions can be made without departing from the scope of protection of the present invention. Therefore, although the present invention has been described in detail through the above embodiments, the present invention is not limited to the above embodiments. Without departing from the concept of the present invention, the present invention may also include more other equivalent embodiments. The scope of protection of the present invention patent is determined by the scope of the appended claims.

Claims

1. A signal transmission link, characterized in that: Applied in an electromagnetic wave sensor, the transmission link includes an analog signal source and a digital phase shifter. The analog signal source can be configured to provide an initial analog signal, and the digital phase shifter can be configured to provide a phase-shifted signal generated in a digital domain, and to phase-shift the initial analog signal based on the phase-shifted signal, so as to perform a preset phase shift operation on the initial analog signal.

2. The signal transmission link according to claim 1, characterized in that: Also included is a transmitting antenna, which can be configured to radiate the phase-shifted initial analog signal to a preset spatial area.

3. The signal transmission link according to claim 1, characterized in that: The digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter and a mixer. The digital phase shifter can be configured to generate a digital phase shift signal. The digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal. The mixer can be configured to use the received analog phase shift signal to perform a mixing operation on the received initial analog signal so as to perform a preset phase shift operation on the initial analog signal.

4. The signal transmission link according to claim 3, characterized in that: The digital phase-shifted signal source comprises a direct digital frequency synthesizer, the digital-to-analog converter is an IQ digital-to-analog converter, and the mixer is an IQ mixer.

5. The signal transmission link according to claim 1, characterized in that: The digital phase-shifted signal is a single-tone signal, and the initial analog signal is a swept-frequency signal; or, The digital phase-shifted signal is a frequency-sweep signal, and the initial analog signal is a single-tone signal.

6. The signal transmission link according to any one of claims 1 to 5, characterized in that: The signal transmission link transmits a frequency modulated continuous wave signal.

7. A signal transmission link, characterized in that: It includes a signal transmission main path and a signal calibration link integrated in the same integrated circuit; wherein: The signal calibration link is configured to calibrate the signal transmission main path to obtain compensation information; and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation according to the compensation information, so as to achieve target detection and / or communication.

8. The signal transmission link according to claim 7, characterized in that: The compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.

9. The signal transmission link according to claim 7, characterized in that: The main signal transmission path includes a first signal source and a phase shifter; wherein the first signal source is configured to generate a first analog signal; and the phase shifter is configured to perform frequency shifting and / or phase shifting on the first analog signal to form a radio frequency transmission signal.

10. The signal transmission link according to claim 9, characterized in that: When the phase shifter is a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting mixer is configured to perform mixing processing on the first analog signal and the second analog signal to form the radio frequency transmission signal; When the phase shifter is an orthogonal architecture, the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting-end mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting-end mixer is configured to perform frequency shifting and / or phase shifting on the first analog signal based on the second analog signal to form the RF transmission signal.

11. The signal transmission link according to claim 10, characterized in that: The main transmission path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter, and the compensation circuit is used to combine the compensation signal and the signal output by the second signal source before outputting it.

12. A signal transceiver link, characterized in that: A signal transmission link as claimed in any one of claims 1 to 11, and a signal receiving link; The signal receiving link includes a receiving end mixer, an analog-to-digital converter and a digital signal processing module; Wherein, the receiving end mixer may be configured to down-convert the received echo signal based on the received receiving end local oscillator signal to obtain an analog intermediate frequency signal, the analog-to-digital converter may be configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal, and the digital signal processing module may be configured to process the digital intermediate frequency signal to obtain a target parameter; The echo signal is a signal formed when the signal transmitted by the signal transmission link is reflected and / or scattered by a target object.

13. The signal transceiver link according to claim 12, characterized in that: The receiving mixer is a real mixer, and the analog-to-digital converter is a real analog-to-digital converter; or, The receiving-end mixer is an orthogonal mixer, and the analog-to-digital converter is an orthogonal analog-to-digital converter.

14. The signal transceiver link according to claim 12, characterized in that: The receiving end local oscillator signal is a swept frequency signal; or, the receiving end local oscillator signal is a single tone signal.

15. A signal calibration link, characterized in that: comprising the signal transceiver link as claimed in claim 12; The receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link, and the signal receiving link can be configured to calibrate the signal transmitting link.

16. The signal calibration link of claim 15, wherein: There is a preset difference frequency between the local oscillator signal of the signal receiving link and the local oscillator signal of the signal transmitting link.

17. The signal calibration link of claim 16, wherein: It also includes a BIST module, which is arranged between the local oscillator signal source and the receiving end mixer; The BIST template may be configured to mix the received local oscillator signal based on a preset frequency deviation signal, so that the local oscillator signal received by the receiving mixer has a preset difference frequency with the local oscillator signal of the signal transmission link.

18. A signal calibration link, characterized in that: Comprising the signal transceiver link as claimed in claim 12, and a BIST module; The receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link through the BIST module, and the signal receiving link can be configured to calibrate the signal transmitting link.

19. A signal calibration link, characterized in that: It comprises two signal receiving links, a BIST module, an auxiliary circuit unit and a signal transmitting link as claimed in any one of claims 1 to 11, and; Any of the signal receiving links includes a real mixer, a real analog-to-digital converter and a digital signal processing module; the real mixer can be configured to downconvert the received echo signal based on the received local oscillator signal to obtain an analog intermediate frequency signal, the real analog-to-digital converter can be configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal, and the digital signal processing module can be configured to process the digital intermediate frequency signal to obtain a target parameter; the echo signal is a signal formed by the signal transmitted by the signal transmitting link being reflected and / or scattered by a target object; The receiving antenna connection ports of the two signal receiving links are respectively connected to the transmitting antenna connection ports of the signal transmitting link through the auxiliary circuit unit and the BIST module in turn, and the signal receiving link can be configured to calibrate the intermediate frequency part of the signal transmitting link.

20. A signal calibration link for a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on a generated signal according to a compensation coefficient, so as to achieve target detection and / or communication, wherein: The signal calibration link is configured to obtain current observation information of the main signal transmission path under the current compensation coefficient; And, when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient used in the signal transmission link compensation operation; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition.

21. The signal calibration link according to claim 20, characterized in that: The compensation coefficient includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.

22. The signal calibration link according to claim 20, characterized in that: The signal transmission main path and the signal calibration link are integrated into the same integrated circuit.

23. A signal compensation link, characterized in that: It comprises the signal transmission link as claimed in any one of claims 1 to 11, and a compensation unit, wherein the compensation unit can be configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion defects of the signal transmission link.

24. The signal compensation link according to claim 23, characterized in that The compensation unit may be configured to compensate the signal transmission link based on calibration data obtained by the signal calibration link according to any one of claims 15 to 22.

25. The signal compensation link of claim 24, wherein: The signal receiving link in the signal calibration link is integrated as an auxiliary calibration module in the vicinity of the transmitting link to be calibrated, so as to perform real-time calibration operations during the intervals when the transmitting link is working.

26. A method for compensating for unequal feeder lengths, characterized in that: In an antenna array of an electromagnetic wave sensor having at least two signal links, the method comprises: Using the shortest feeder of the at least two signal links as a reference link, and obtaining the delay difference of the remaining transmission links relative to the reference link; and Based on the time delay difference, compensation is performed on the antenna array in the digital domain for unequal feeder lengths.

27. An integrated circuit, characterized in that: It includes a radio frequency module, an analog signal processing module and a digital signal processing module connected in sequence; the radio frequency module is used to generate a radio frequency transmission signal and receive a radio frequency reception signal; the analog signal processing module is used to down-convert the radio frequency reception signal to obtain an intermediate frequency signal; the digital signal processing module is used to perform analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal; as well as The radio frequency module comprises a signal transmission link according to any one of claims 1 to 11, a signal transceiver link according to any one of claims 12 to 14, a signal calibration link according to any one of claims 15 to 22, and a signal compensation link according to any one of claims 23 to 25; and / or The digital signal processing module performs compensation in the digital domain based on the feeder unequal length compensation method described in claim 26.

28. The integrated circuit according to claim 27, characterized in that It also includes a data processing module, which is used to process the digital signal to achieve target detection and / or wireless communication.

29. The integrated circuit according to claim 27 or 28, characterized in that The integrated circuit is a millimeter wave chip.

30. The integrated circuit according to claim 27 or 28, characterized in that The radio frequency receiving signal is an echo signal formed by the radio frequency transmitting signal being emitted and / or scattered by a target, and the integrated circuit is a sensor chip.

31. An electromagnetic wave sensor, characterized in that: include: Carrier; An integrated circuit as claimed in any one of claims 27 to 30, arranged on a carrier; An antenna is arranged on the carrier, or the antenna and the integrated circuit are integrated into one device and arranged on the carrier; The integrated circuit is connected to the antenna and is used to transmit the radio frequency transmission signal and / or receive the radio frequency reception signal.

32. A device, characterized in that include: Equipment body; as well as The electromagnetic wave sensor as claimed in claim 31 is disposed on the device body; The electromagnetic wave sensor is used for target detection and / or communication to provide reference information for the operation of the device body.

33. A non-transitory computer-readable storage medium having computer-readable instructions stored thereon, which, when executed by a processor, cause the processor to perform the method of claim 26.

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