Current sensing apparatus, and magnetic modulation method

Abstract

A current sensing apparatus, and a magnetic modulation method. The current sensing apparatus comprises: a magnetic core (110), wherein a conductor which carries a current signal under test passes through the magnetic core (110); a secondary coil (120), which is wound on the magnetic core (110); a sampling resistor (130), which is connected in series to the secondary coil (120), and is used for providing a sampling voltage signal which corresponds to an excitation current; an excitation control circuit (140), which is connected to the sampling resistor (130), and is used for generating a floating flip threshold signal on the basis of the sampling voltage signal, and outputting an excitation polarity signal on the basis of a comparison result of the sampling voltage signal with the floating flip threshold signal; and an excitation driving circuit (150), which is connected to the excitation control circuit (140) and the secondary coil (120), and is used for flipping the polarity of an excitation driving signal on the basis of the excitation polarity signal output by the excitation control circuit (140), such that the magnetic core (110) oscillates between a positive saturation point and a negative saturation point. The magnetic modulation effect in which the precision is high, the power consumption is low and a frequency response is good is obtained by means of a relatively simple circuit.

Description

A current sensing device and magnetic modulation method

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS

[0002] This application claims priority to Chinese patent application No. 202311401632.4, filed on October 25, 2023, entitled “A Current Sensing Device and Magnetic Modulation Method,” the entire contents of which are incorporated herein for all purposes. Technical Field

[0003] The present invention generally relates to the field of current detection, and more particularly to a current sensing device and a magnetic modulation method. Background Art

[0004] Magnetic modulation technology is a high-precision magnetic field measurement technique, and current sensing is one of its most important applications. Traditional AC mutual induction technology uses the mutual inductance of the primary and secondary coils to detect current. However, when the measured current is DC or has a DC component, AC mutual induction technology cannot effectively detect the measured current. Hall effect sensors are another magnetic field detection technology that are also used for current detection. Hall effect sensors use the Hall effect principle to measure the magnetic field generated by the measured current. Although this technology can detect DC, its accuracy is lower than that of magnetic modulation technology and is mainly used in low-precision applications.

[0005] A typical magnetic modulation system primarily consists of a transformer, an excitation drive circuit, an excitation control circuit, and a detection circuit. The excitation control circuit uses a specific strategy to control the excitation drive circuit to drive the transformer, causing the transformer's magnetic core to oscillate within the positive and negative saturation regions. This allows the measured signal, including the DC component, to be superimposed on the excitation signal and visualized. This process is called modulation. The detection circuit uses a specific strategy to process the waveform containing both the measured signal and the excitation signal, removing the excitation signal to obtain the measured signal. This process is called demodulation. The essence of magnetic modulation technology is to exploit the saturation characteristics of magnetic materials through modulation-demodulation techniques to obtain information that would not otherwise be directly accessible.

[0006] The primary difference in magnetic modulation systems lies in how the excitation control circuitry causes the transformer's core to oscillate between positive and negative saturation regions. Current magnetic modulation excitation methods suffer from issues like poor accuracy and high power consumption with simple methods, and susceptibility to interference and difficulty in implementation with more complex methods.

[0007] Summary of the Invention

[0008] The Summary of the Invention introduces a series of simplified concepts that will be further described in the Detailed Description of the Invention. The Summary of the Invention of this application is not intended to limit the key features and essential technical features of the claimed technical solution, nor is it intended to determine the scope of protection of the claimed technical solution.

[0009] An embodiment of the present invention provides a current sensing device, comprising:

[0010] a magnetic core through which a conductor carrying a measured current signal passes;

[0011] a secondary coil wound on the magnetic core;

[0012] A sampling resistor is connected in series with the secondary coil and is used to provide a sampling voltage signal corresponding to the excitation current;

[0013] an excitation control circuit connected to the sampling resistor, configured to generate a floating inversion threshold signal according to the sampled voltage signal, and output an excitation polarity signal according to a comparison result between the sampled voltage signal and the floating inversion threshold signal;

[0014] An excitation drive circuit is connected to the excitation control circuit and the secondary coil, and is used to apply an excitation drive signal to the secondary coil, and flip the polarity of the excitation drive signal according to the excitation polarity signal output by the excitation control circuit, so that the magnetic core oscillates between a positive saturation point and a negative saturation point.

[0015] In one embodiment, generating a floating flip threshold signal according to the sampled voltage signal includes:

[0016] Obtaining a center offset value according to the sampled voltage signal;

[0017] A flip threshold is calculated according to a difference between a saturation depth setting value and the center offset value or a sum of the saturation depth setting value and the center offset value to generate the floating flip threshold signal.

[0018] In one embodiment, obtaining a center offset value according to the sampled voltage signal includes:

[0019] The sampled voltage signal is low-pass filtered, and the center offset value is obtained according to a result of the low-pass filtering.

[0020] In one embodiment, obtaining a center offset value according to the sampled voltage signal includes:

[0021] The center offset value of the current cycle is obtained according to the sampled voltage signal of the current cycle, or,

[0022] The center offset value of the current cycle is obtained according to the sampled voltage signal of the historical cycle.

[0023] In one embodiment, obtaining the center offset value of the current cycle according to the sampled voltage signal of the current cycle includes:

[0024] Obtaining the center offset value of the current cycle according to the fixed time value, average value or integrated value of the sampled voltage signal of the current cycle;

[0025] The obtaining of the center offset value of the current cycle according to the sampled voltage signal of the historical cycle includes:

[0026] The center offset value of the current cycle is obtained according to the fixed time value, average value or integrated value of the sampled voltage signal in the historical cycle.

[0027] In one embodiment, the excitation control circuit includes a filter, the filter is connected to the sampling resistor, and is used to filter the sampled voltage signal to obtain a filtered signal, and the filtered signal is an analog signal;

[0028] The excitation control circuit generates the floating flip threshold signal according to the filtered signal.

[0029] In one embodiment, the filter comprises a low-pass filter, and the filtered signal comprises a low-pass filtered signal; the excitation control circuit further comprises a threshold controller, a comparator, and a trigger, wherein:

[0030] The threshold controller is connected to the low-pass filter and is used to generate the floating flip threshold signal according to the low-pass filter signal, the excitation polarity signal and the saturation depth setting signal;

[0031] The comparator is connected to the threshold controller and is configured to output a flip request signal when the sampled voltage signal reaches the floating flip threshold signal according to the sampled voltage signal, the floating flip threshold signal and the excitation polarity signal;

[0032] The trigger is connected to the comparator and is configured to output the excitation polarity signal according to the flip request signal.

[0033] In one embodiment, the excitation control circuit includes an analog-to-digital converter and a digital processor, the analog-to-digital converter is connected to the sampling resistor, and the digital processor is connected to the analog-to-digital converter;

[0034] The analog-to-digital converter is used to convert the sampled voltage signal into discrete sampled data, and the digital processor is used to digitally process the sampled data and obtain the floating flip threshold signal in combination with a saturation depth setting signal.

[0035] In one embodiment, the digital processing frequency of the digital processor is higher than the analog-to-digital conversion frequency of the analog-to-digital converter.

[0036] In one embodiment, the digital processor includes an upsampling circuit, a floating threshold calculation circuit, and a flip controller;

[0037] The upsampling circuit is used to interpolate the sampled data to obtain interpolated data;

[0038] The floating threshold calculation circuit is connected to the upsampling circuit and is used to obtain the floating flip threshold signal according to the interpolation data and the saturation depth setting signal;

[0039] The flip controller is connected to the floating threshold calculation circuit and is used for outputting the excitation polarity signal according to the comparison result between the floating flip threshold signal and the interpolation data outside the flip masking time.

[0040] In one embodiment, the interpolation includes interpolating first sampling data obtained in a first calculation cycle and second sampling data obtained in a second calculation cycle to obtain interpolation data corresponding to at least one third calculation cycle, and the third calculation cycle is located between the first calculation cycle and the second calculation cycle.

[0041] In one embodiment, the interpolation includes performing interpolation based on a first-order derivative or a second-order derivative of the sampled data obtained in at least one calculation cycle.

[0042] In one embodiment, the analog-to-digital converter is further connected to the flip controller, and the flip controller is further configured to output a sampling synchronization signal to the analog-to-digital converter when the excitation polarity signal flips, so as to synchronize the analog-to-digital converter.

[0043] A second aspect of an embodiment of the present invention provides a magnetic modulation method, which is used in a current sensing device, wherein the current sensing device includes a magnetic core, a secondary coil wound around the magnetic core, and a sampling resistor connected in series with the secondary coil. The method includes:

[0044] Collecting a sampling voltage signal on the sampling resistor corresponding to the excitation current;

[0045] generating a floating flip threshold signal according to the sampled voltage signal;

[0046] outputting an excitation polarity signal according to a comparison result between the sampling voltage signal and the floating flip threshold signal;

[0047] An excitation drive signal is applied to the secondary coil, and the polarity of the excitation drive signal is reversed according to the excitation polarity signal, so that the magnetic core oscillates between a positive saturation point and a negative saturation point.

[0048] In one embodiment, generating a floating flip threshold signal according to the sampled voltage signal includes:

[0049] Filtering the sampled voltage signal to obtain a filtered signal, where the filtered signal is an analog signal;

[0050] The floating flip threshold signal is generated according to the filtered signal.

[0051] In one embodiment, generating a floating flip threshold signal according to the sampled voltage signal includes:

[0052] The sampled voltage signal is converted into discrete sampled data, and the sampled data is digitally processed to obtain the floating inversion threshold signal.

[0053] In one embodiment, the digitally processing the sampled data and combining the saturation depth setting signal to obtain the floating flip threshold signal includes:

[0054] interpolating the sampled data to obtain interpolated data;

[0055] obtaining the floating flip threshold signal according to the interpolation data and the saturation depth setting signal;

[0056] Outputting the excitation polarity signal according to the comparison result between the sampling voltage signal and the floating inversion threshold signal includes: outputting the excitation polarity signal according to the comparison result between the floating inversion threshold signal and the interpolation data outside the inversion masking time.

[0057] The current sensing device and magnetic modulation method of the embodiments of the present invention can achieve a magnetic modulation effect with high accuracy, low power consumption and good frequency response through a relatively simple circuit. BRIEF DESCRIPTION OF THE DRAWINGS

[0058] The above and other purposes, features, and advantages of the present application will become more apparent through a more detailed description of the embodiments of the present application in conjunction with the accompanying drawings. The accompanying drawings are intended to provide a further understanding of the embodiments of the present application and constitute a part of the specification. Together with the embodiments of the present application, they are used to explain the present application and do not constitute a limitation of the present application. In the drawings, the same reference numerals generally represent the same components or steps.

[0059] FIG1 is a schematic diagram of a current sensing device according to an embodiment of the present invention;

[0060] FIG2 is a schematic diagram of a current sensing device based on an analog circuit according to an embodiment of the present invention;

[0061] FIG3 is a schematic diagram of waveforms of various signals during operation of the current sensing device shown in FIG2 ;

[0062] FIG4 is a schematic diagram of a current sensing device based on a digital circuit according to an embodiment of the present invention;

[0063] FIG5 is a schematic diagram of waveforms of various signals during operation of the current sensing device shown in FIG4 ;

[0064] 6 is a schematic diagram of synchronization of an analog-to-digital converter and an excitation polarity signal according to an embodiment of the present invention;

[0065] FIG7 is a schematic diagram showing the configuration of a secondary coil current waveform according to an embodiment of the present invention;

[0066] FIG8 is a diagram showing the corresponding relationship between the secondary coil current waveform and the flipping time according to an embodiment of the present invention;

[0067] FIG9 is a schematic diagram showing the outer contour of the secondary coil current waveform floating along with the measured current according to an embodiment of the present invention;

[0068] FIG10 is a comparison diagram of the waveforms of the secondary coil current in fixed frequency flipping, fixed threshold flipping, and floating threshold flipping according to an embodiment of the present invention;

[0069] FIG11 is an integral nonlinear error curve of fixed frequency flip, fixed threshold flip and floating threshold flip according to an embodiment of the present invention;

[0070] FIG12 is a secondary coil current waveform of a large-amplitude AC current close to the excitation frequency using a floating threshold flip measurement according to one embodiment of the present invention;

[0071] FIG13 is a schematic flow chart of a magnetic modulation method according to an embodiment of the present invention. DETAILED DESCRIPTION

[0072] In the following description, numerous specific details are provided to provide a more thorough understanding of the present invention. However, it will be apparent to those skilled in the art that the present invention may be practiced without one or more of these details. In other instances, certain technical features well known in the art are not described to avoid confusion with the present invention.

[0073] It should be understood that the present invention can be implemented in various forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided to make the disclosure thorough and complete and to fully convey the scope of the invention to those skilled in the art. In the drawings, the dimensions and relative sizes of layers and regions may be exaggerated for clarity. Like reference numerals throughout represent like elements.

[0074] It should be understood that when an element or layer is referred to as being "on," "adjacent to," "connected to," or "coupled to" another element or layer, it can be directly on, adjacent to, connected to, or coupled to the other element or layer, or there can be intervening elements or layers. Conversely, when an element is referred to as being "directly on," "directly adjacent to," "directly connected to," or "directly coupled to" another element or layer, there are no intervening elements or layers. It should be understood that although the terms first, second, third, etc. may be used to describe various elements, components, regions, layers, and / or parts, these elements, components, regions, layers, and / or parts should not be limited by these terms. These terms are merely used to distinguish one element, component, region, layer, or part from another element, component, region, layer, or part. Thus, a first element, component, region, layer, or part discussed below may be represented as a second element, component, region, layer, or part without departing from the teachings of the present invention.

[0075] Spatially relative terms such as "under," "beneath," "below," "under," "above," "above," etc., may be used herein for convenience of description to describe the relationship of one element or feature shown in the figures to other elements or features. It should be understood that the spatially relative terms are intended to include different orientations of the device in use and operation in addition to the orientations shown in the figures. For example, if the device in the drawings is flipped, then the elements or features described as "under" or "beneath" or "beneath" the other elements will be oriented as "over" the other elements or features. Thus, the exemplary terms "under" and "under" may include both the upper and lower orientations. The device may be oriented otherwise (rotated 90 degrees or in other orientations) and the spatial descriptors used herein are interpreted accordingly.

[0076] The purpose of the terms used herein is only to describe specific embodiments and is not intended to limit the present invention. When used herein, the singular forms "a", "an", and "the" are intended to include the plural forms, unless the context clearly indicates otherwise. It should also be understood that the terms "comprising" and / or "comprising", when used in this specification, determine the presence of the features, integers, steps, operations, elements and / or parts, but do not exclude the presence or addition of one or more other features, integers, steps, operations, elements, parts and / or groups. When used herein, the term "and / or" includes any and all combinations of the relevant listed items.

[0077] In order to fully understand the present invention, a detailed structure will be provided in the following description to illustrate the technical solution proposed by the present invention. The preferred embodiments of the present invention are described in detail below. However, in addition to these detailed descriptions, the present invention may also have other implementations.

[0078] A relatively simple method of magnetic modulation excitation is to apply an excitation voltage at a fixed frequency to the secondary coil of the transformer to cause it to oscillate. Although this method is simple to implement, it cannot automatically adapt to changes in temperature or variations in the saturation point due to individual differences in the transformer, resulting in oversaturation in some cases and undersaturation in others. Oversaturation can cause transient excessive excitation currents, increasing system power consumption and even disrupting normal power supply. Undersaturation can lead to excessive residual magnetism in the core, affecting detection accuracy.

[0079] To enable the system to adapt to changes in the saturation point, closed-loop excitation technology was introduced. This technology monitors the secondary coil current through a sampling resistor and dynamically determines whether the core is saturated based on a specific strategy. If the core is saturated, the excitation voltage is reversed, achieving closed-loop control of the excitation oscillation. This closed-loop excitation technology can better overcome the effects of temperature changes and individual differences in transformers.

[0080] Fixed-threshold flipping is a common closed-loop excitation technique. This technique monitors the secondary coil current through a sampling resistor. When the excitation voltage is positive, if the secondary coil current exceeds a positive threshold, it indicates that the core has reached the set positive saturation point. At this point, the excitation voltage is flipped to a negative value, pulling the core out of the positive saturation zone and into the negative saturation zone. When the excitation voltage is negative, if the secondary coil current is less than a negative threshold, it indicates that the core has reached the set negative saturation point. At this point, the excitation voltage is flipped to a positive value, pulling the core out of the negative saturation zone and into the positive saturation zone.

[0081] While fixed-threshold flipping technology achieves closed-loop flipping, because the secondary coil current is the result of the superposition of the excitation signal and the measured signal, employing a fixed secondary coil current threshold results in complementary changes in the excitation depth when the measured signal changes. For example, when the measured signal is positive, the positive excitation depth decreases while the negative excitation depth increases. This asymmetry significantly reduces system linearity, with errors becoming more pronounced with larger measured signals. Furthermore, to ensure the system covers the measurement range, the excitation depth must be set very deep to prevent the excitation from flipping due to the measured current signal, which significantly increases system power consumption.

[0082] Differential flip is a more complex closed-loop excitation method than fixed threshold flip. Differential flip determines whether to flip based on the first-order derivative of the current value on the secondary coil: when the excitation signal is positive, if the first-order derivative is greater than a set positive threshold, the excitation signal is flipped to negative; when the excitation signal is negative, if the first-order derivative is less than a set negative threshold, the excitation signal is flipped to positive. Since the DC component of the measured waveform in the secondary coil is eliminated during the derivation process, when the measured current is DC, differential flip can overcome the asymmetric saturation point of fixed threshold flip and obtain better linearity. At the same time, differential flip technology does not require the saturation depth to cover the entire measurement range. Compared with fixed threshold flip, differential flip can use a shallower saturation depth, resulting in significantly lower system power consumption when the measured current is zero.

[0083] However, when the measured current is not DC and has a high frequency, it can easily cause interference with the differential reversal. For example, when the measured current instantly changes from 0 to a larger value, the secondary current surges. Even if the transformer has not yet reached saturation, it will immediately reverse because the first-order derivative of the secondary coil current exceeds the threshold. When the measured current is a high-frequency sine wave, as long as the amplitude is sufficient, it can cause the differential reversal circuit to fail and enter a continuous oscillation state. In addition, differential reversal technology requires more complex analog circuits. Due to the introduction of differentials, the system is more sensitive to voltage fluctuations and external conduction and radiation interference, which greatly increases the difficulty and complexity of circuit design.

[0084] The performance of a magnetic modulation system is also affected by the detection circuit. The detection circuit's function is to demodulate, extracting the measured current waveform from the secondary current waveform superimposed with the excitation signal waveform. The detection circuit typically filters the excitation signal waveform using a low-pass filter constructed from analog circuits. Only when the frequency difference between the excitation frequency and the measured current is sufficiently large can the low-pass filter effectively remove the excitation signal waveform, leaving the measured current waveform. A low-pass filter cannot completely eliminate the excitation signal frequency; it can only reduce its amplitude. This means that even when the measured current is zero or DC, the system output signal still contains AC ripple at the excitation signal frequency. There are several ways to reduce this ripple: first, increasing the filter order, which results in a more complex system, longer latency, and greater susceptibility to oscillation; second, increasing the excitation frequency, which increases system power consumption; and third, lowering the filter's cutoff frequency, which limits the system's detection frequency range. Consequently, the performance parameters of traditional low-pass filter-based detection circuits exhibit significant constraints, making it difficult to improve overall system performance.

[0085] In summary, current magnetic modulation excitation methods have the following problems: simple methods have poor precision and high power consumption, while complex methods are susceptible to interference and difficult to implement. To address the above problems, the embodiments of the present invention propose a current sensing device and magnetic modulation method. This device and method adopt a new magnetic modulation flip mechanism, which can achieve a magnetic modulation effect with higher precision, lower power consumption, and better frequency response than traditional methods through a relatively simple circuit, breaking through the limitations of fixed frequency flip, fixed threshold flip, and differential flip.

[0086] The current sensing device and magnetic modulation method according to the embodiments of the present invention are described in detail below with reference to the accompanying drawings. The features of the following embodiments and implementations may be combined with each other unless they conflict.

[0087] 1 , the current sensing device according to an embodiment of the present invention includes: a magnetic core 110 through which a conductor carrying a measured current signal passes; a secondary coil 120 wound around the magnetic core 110; a sampling resistor 130 connected in series with the secondary coil 120 for providing a sampling voltage signal corresponding to the excitation current; an excitation control circuit 140 connected to the sampling resistor 130 for generating a floating flip threshold signal based on the sampling voltage signal and outputting an excitation polarity signal based on a comparison result between the sampling voltage signal and the floating flip threshold signal; and an excitation drive circuit 150 connected to the excitation control circuit 140 and the secondary coil 120 for applying an excitation drive signal to the secondary coil 120 and flipping the polarity of the excitation drive signal based on the excitation polarity signal output by the excitation control circuit 140, thereby causing the magnetic core 110 to oscillate between a positive saturation point and a negative saturation point.

[0088] The method used by the excitation control circuit 140 in this embodiment of the present invention to determine when to flip the excitation polarity can be referred to as "floating threshold flipping." There are two key elements to floating threshold flipping: first, the decision to flip is based on the comparison between the sampled voltage signal and the floating flip threshold signal; second, the flip threshold signal fluctuates.

[0089] Specifically, when the excitation drive signal is positive, if the sampled voltage signal is greater than the threshold Th+, the excitation drive signal is flipped to negative. When the excitation drive signal is negative, if the sampled voltage signal is less than the threshold Th-, the excitation voltage is flipped to positive. The difference from fixed threshold flipping is that Th+ and Th- are fixed values ​​in fixed threshold flipping, while Th+ and Th- in floating threshold flipping are dynamically changed and closed-loop adjusted, allowing the saturation point to remain unchanged even when the measured current changes.

[0090] For example, a center offset value can be obtained based on the sampled voltage signal, and a saturation depth setting value of the saturation depth setting signal can be obtained. A floating flip threshold can be obtained based on the center offset value and the saturation depth setting value. There are various methods for determining the floating flip threshold based on the sampled voltage signal. The methods described below are merely examples and are not intended to limit the present invention.

[0091] In one embodiment, the floating flip threshold can be determined based on a low-pass filter. Specifically, a low-pass filter is used to filter the sampled voltage signal that can represent the secondary coil current fed back by the sampling resistor, and the filter output is used as the center offset value T of the floating flip threshold. offset . Preset a positive saturation depth setting value T sat , the positive and negative flip threshold is determined by the following formula: T th+ =T offset +T sat T th- =T offset –T sat

[0092] This method is relatively simple to implement. If the current sensor obtains the measurement result through a low-pass filter, the excitation part can directly use the output of the low-pass filter as the feedback input without adding additional circuits.

[0093] In another embodiment, the floating flip threshold of the current cycle can be determined based on the sampled voltage signal of the current cycle. The current cycle can be the current full cycle or the current half cycle. The current half cycle refers to the time period from the rising edge to the falling edge, or from the falling edge to the rising edge, of the sampled voltage signal. Specifically, the center offset value T of the flip threshold of the current half cycle can be determined based on the sampled voltage signal (or the secondary coil current signal) in the current half cycle. offset . Preset a positive saturation depth setting value T sat ,The positive and negative flip threshold is determined by the following formula: T th+ =T offset +T sat T th- =T offset –T sat

[0094] The center offset value T of the flip threshold of the current half cycle is determined according to the sampling voltage signal of the current half cycle. offset There are many ways to do this, the following are just examples.

[0095] Optionally, the center offset value T can be determined based on the sampled voltage signal at a fixed time after the start of the current half cycle. offsetFor example, the fixed moment is located close to the center of the linear region of the magnetic core material, where the magnetic permeability of the magnetic material is relatively large and the current change rate is small.

[0096] Optionally, the center offset value T can be determined based on the integral value of the sampled voltage signal of the current half cycle. offset By integrating, the interference of system voltage noise on the flip point can be effectively reduced.

[0097] Optionally, the center offset value T can be determined based on the window average value of the sampled voltage signal in the current half cycle. offset By averaging, the interference caused by system voltage noise on the flip point can be effectively reduced.

[0098] In another embodiment, the floating flip threshold can be determined based on a historical sampled voltage signal. The historical sampled voltage signal can be a sampled voltage signal of one or more previous full cycles, or a sampled voltage signal of one or more previous half cycles. Specifically, the center offset value T of the flip threshold of the current half cycle can be determined based on the historical sampled voltage signal. offset . Preset a positive saturation depth setting value T sat ,The positive and negative flip threshold is determined by the following formula: T th+ =T offset +T sat T th- =T offset –T sat

[0099] Determine the center offset value T based on the historical excitation current offset There are also multiple ways to do this, for example:

[0100] The center offset T is determined based on the fixed time value, integral value and / or average value of the sampled voltage signal in the full cycle consisting of the previous half cycle and the previous half cycle. offset ;

[0101] The center offset value T is determined based on the fixed time value, integral value and / or average value of the excitation current in the full cycle composed of the current half cycle and the previous half cycle. offset ;

[0102] The center offset value T is determined based on the fixed time value, integral value and / or average value of the sampled voltage signal of the previous half cycle, i.e., the half cycle with the excitation direction opposite to the current half cycle. offset ;

[0103] The center offset value T is determined based on the fixed time value, integral value and / or average value of the sampled voltage signal of the previous half cycle, i.e., the half cycle with the same excitation direction as the current half cycle. offset ;

[0104] The center offset value T is determined based on the fixed time value, integral value and / or average value of the sampled voltage signal of multiple completed half cycles and / or full cycles. offset .

[0105] In some embodiments, the excitation control circuit 140 can be implemented based on an analog circuit. In existing magnetic modulation systems, analog circuits are commonly used for excitation closed-loop control. Therefore, the use of analog circuits can reuse the circuit structure of existing magnetic modulation systems. For example, the excitation control circuit 140 based on an analog circuit can include a filter, which is connected to a sampling resistor and is used to filter the sampled voltage signal to obtain a filtered signal. The filtered signal is an analog signal, and the excitation control circuit 140 determines the floating flip threshold based on the analog signal. When determining the floating flip threshold based on a filter, it is particularly suitable to use an analog circuit implementation because traditional magnetic modulation systems usually include a filter as the system's output signal filter, without the need for additional circuits. It should be noted that the following mainly uses a low-pass filtering method as an example to describe the system architecture based on an analog circuit, but analog circuits can also be used when determining the floating flip threshold based on other methods. The advantage of using an analog circuit to determine the floating flip threshold is that the analog circuit is continuous in time and voltage, and there are no sampling rate and resolution issues.

[0106] Referring to FIG2 , a schematic diagram of a current sensing device based on an analog circuit is shown. In the example of FIG2 , the filter includes a low-pass filter, and the filtered signal includes a low-pass filtered signal. The excitation control circuit 140 also includes a threshold controller, a comparator, and a trigger. The threshold controller is connected to the low-pass filter and is configured to generate a floating flip threshold signal based on the low-pass filtered signal, an excitation polarity signal, and a saturation depth setting signal. The comparator is connected to the threshold controller and is configured to output a flip request signal based on the sampled voltage signal, the floating flip threshold signal, and the excitation polarity signal when the sampled voltage signal reaches the floating flip threshold signal. The trigger is connected to the comparator and is configured to output the excitation polarity signal based on the flip request signal.

[0107] Specifically, the sampling resistor 130 is connected in series with the secondary coil 120 so that the voltage drop across the sampling resistor 130 is proportional to the secondary coil current. The voltage drop across the sampling resistor is a sampling voltage signal, which can also represent the secondary coil current signal.

[0108] Sampling resistor 130 is connected to a low-pass filter. After the sampled voltage signal passes through the low-pass filter, the periodic excitation signal is substantially filtered out, leaving a remaining signal called a low-pass filtered signal. The amplitude of this signal is proportional to the measured current, and the low-pass filtered signal can be directly output as the measurement result of the measured current signal.

[0109] The low-pass filter is connected to the threshold controller. The threshold controller generates a floating flip threshold signal based on the excitation polarity signal, the low-pass filter signal, and the saturation depth setting signal. Its logic is as follows: when the excitation polarity signal is 1, the floating flip threshold signal = the low-pass filter signal + the saturation depth setting signal; when the excitation polarity signal is 0, the floating flip threshold signal = the low-pass filter signal – the saturation depth setting signal.

[0110] The threshold controller is connected to a comparator. The comparator is used to determine whether the current sampled voltage signal has reached the floating flip threshold signal based on the sampled voltage signal, the floating flip threshold signal, and the excitation polarity signal. When the sampled voltage signal reaches the floating flip threshold signal, the comparator outputs a flip request signal. The logic is as follows: when the excitation polarity signal is 1, if the sampled voltage signal is greater than the floating flip threshold signal, the flip request signal is 1; otherwise, it is 0. When the excitation polarity signal is 0, if the sampled voltage signal is less than the floating flip threshold signal, the flip request signal is 1; otherwise, it is 0.

[0111] The trigger is connected to the comparator and is used to flip the excitation polarity signal based on the flip request signal. Its logic is: when the flip request signal shows a rising edge, the excitation polarity signal is flipped. The trigger is also connected to the excitation drive circuit 150 and is used to output the excitation polarity signal to the excitation drive circuit 150, forming a closed-loop control.

[0112] The excitation drive circuit 150 amplifies the excitation polarity signal (a 1-bit logic signal) into an excitation drive signal with a specified voltage amplitude and strong current drive capability. Its waveform includes, but is not limited to, a square wave. Driven by the excitation drive signal, the current in the secondary coil 120 periodically increases and decreases, and the magnetic core 110 periodically oscillates within the positive saturation region and the negative saturation region. The logic is as follows: when the excitation polarity signal is 1, the excitation drive circuit 150 outputs a positive voltage, driving the secondary coil current to increase; when the excitation polarity signal is 0, the excitation drive circuit 150 outputs a negative voltage, driving the secondary coil current to decrease.

[0113] Referring to FIG3 , there is shown a schematic diagram of waveforms of main signals during the operation of the current sensing device of FIG2 .

[0114] In another embodiment, the excitation control circuit 140 can be implemented based on a digital circuit. Specifically, the sampled voltage signal on the sampling resistor is converted into a digital value by an analog-to-digital converter, and then the digital value is processed by a digital processor to determine the floating flip threshold. Referring to Figure 4, a schematic diagram of a current sensing device based on a digital circuit is shown. In the example of Figure 4, the excitation control circuit includes an analog-to-digital converter and a digital processor, the analog-to-digital converter is connected to the sampling resistor 130, and the digital processor is connected to the analog-to-digital converter; the analog-to-digital converter is used to convert the sampled voltage signal into discrete sampling data, and the sampling data is discrete in both value and time; the digital processor is used to digitally process the sampling data and combine it with the saturation depth setting signal to obtain the floating flip threshold signal.

[0115] Thanks to advances in semiconductor technology, the performance of analog-to-digital converters (ADCs) and digital processors such as FPGAs, DSPs, and MCUs has significantly improved, making it possible to implement floating threshold flipping using a fully digital approach. As the sampling frequency and data resolution of ADCs have increased, the frequency and accuracy limits of digital methods for determining flipping thresholds have also increased, thereby reducing the ADC's limitations on the frequency response range and measurement accuracy of the entire magnetic modulation system. Furthermore, current digital processors are capable of performing large amounts of calculations at high speed, enabling the realization of digital closed-loop systems with high sampling rates and high data resolution.

[0116] For example, an FPGA can be used as a digital processor. Compared with other devices, FPGA has the best real-time performance and can accurately determine the flip timing.

[0117] For example, an analog-to-digital converter based on the successive approximation principle can be used. The analog-to-digital converter based on the successive approximation principle can achieve a very high conversion rate, better flip time accuracy at a higher excitation frequency, and a higher frequency measurement range.

[0118] For example, an analog-to-digital converter based on the Σ-Δ principle may be used. The analog-to-digital converter based on the Σ-Δ principle can achieve high linearity and low noise, thereby achieving better measurement accuracy.

[0119] Because the output value of the analog-to-digital converter is discrete in time, the time resolution of the excitation control circuit 140 in determining the flipping moment is equal to the sampling period of the analog-to-digital converter. However, high-precision analog-to-digital converters typically have low conversion rates. This results in the excitation control circuit 140 being able to control the flipping of the excitation signal with only low time resolution. This results in large random fluctuations in the excitation period, leading to unstable and asymmetric excitation saturation points, fluctuations in the excitation current curve, and increased measurement ripple.

[0120] To address this issue, the digital processor's calculation frequency can be made higher than the analog-to-digital converter's conversion frequency. For example, the digital processor's calculation frequency can be an integer multiple of the analog-to-digital converter's conversion frequency. During periods when the digital processor does not receive sampled data from the analog-to-digital converter, it can upsample based on the received sampled data. This means that through interpolation, each digital processor calculation cycle has a measured or interpolated sampled data sequence, and subsequent floating flip threshold determinations are made based on this sampled data sequence. Interpolation technology can effectively increase the system's supported excitation frequency, broaden the acceptable frequency range of the measured signal, and reduce the technical requirements for the analog-to-digital converter.

[0121] Based on the above principles, the digital processor may include an upsampling circuit, a floating threshold calculation circuit, and a flip controller. The upsampling circuit is used to interpolate sampled data to obtain interpolated data; the floating threshold calculation circuit is connected to the upsampling circuit and is used to obtain a floating flip threshold signal based on the interpolated data and a saturation depth setting signal; and the flip controller is connected to the floating threshold calculation circuit and is used to output an excitation polarity signal based on a comparison result between the floating flip threshold signal and the interpolated data.

[0122] Specifically, the upsampling circuit upsamples the sampled data sequence, reducing the sample intervals and minimizing the impact of time discretization on rollover determination and reading calculation. The upsampling circuit interpolates the received sampled data to achieve upsampling. Therefore, the upsampling result (interpolated data) has a certain latency compared to the sampled data.

[0123] In one embodiment, the interpolation includes interpolating first sampled data obtained in a first calculation cycle and second sampled data obtained in a second calculation cycle to obtain interpolated data corresponding to at least one third calculation cycle, where the third calculation cycle is between the first calculation cycle and the second calculation cycle. In another embodiment, the interpolation includes interpolating based on first-order derivatives or second-order derivatives of the sampled data obtained in at least one calculation cycle.

[0124] For ease of understanding, assume that the sampling rate of the analog-to-digital converter is f and the calculation frequency of the digital processor is a*f, that is, every a calculation cycle of the digital processor, an output reading of the analog-to-digital converter will be obtained. The calculation cycle of the second-to-last reading is recorded as cycle 0, and the reading is recorded as R0; the calculation cycle of the most recent reading is recorded as cycle a, and the reading is recorded as R a The calculation cycle for the third to last reading is recorded as cycle-a, and the reading is recorded as R -a .

[0125] In the above example, three interpolation methods may be specifically adopted as follows.

[0126] The first interpolation method is to perform linear interpolation based on the most recent measured sampling data. Assume that the interpolation data obtained in each cycle between the two measured sampling data of the 0th cycle and the ath cycle is recorded as R n , then the R obtained by linear interpolation based on the two most recent measured sampling data is n It can be expressed as:

[0127] Among them, R n In fact, it is the interpolation result of a certain moment in the past, that is, this method is based on the interpolation of two obtained sampling data.

[0128] The second interpolation method is to predict the current interpolation result based on the first-order derivative of at least one recent measured sampling data. Assume that the current cycle is cycle n, that is, n cycles have passed since the second-to-last sampling data R0 was obtained; the interpolation data of the current cycle is recorded as R n , then the current interpolation result R is predicted based on the first-order derivative of the two most recent measured sampling data. n It can be expressed as:

[0129] This interpolation method can overcome the problem of delayed flipping time to a certain extent.

[0130] The third interpolation method is to predict the current interpolation data based on the second-order derivative of at least one recent measured sampling data. Assume that the current cycle is cycle n, that is, n cycles have passed since the penultimate sampling data R0 was obtained. The interpolation data of the current cycle is recorded as R n .

[0131] The current interpolated data R is predicted based on the second derivative of the three most recent measured readings. n It can be expressed as:

[0132] Compared to using the first-order derivative, the second-order derivative can better fit the saturation curve of the magnetic material.

[0133] Referring to FIG5 , FIG5 is a schematic diagram showing waveforms of main signals during operation of the current sensing device shown in FIG4 .

[0134] The sampled interpolated data is fed into a floating threshold calculation circuit. This circuit determines the flip threshold based on the saturation depth setting signal input by the user and the excitation polarity signal output by the flip controller, generating a floating flip threshold signal. For example, the logic is as follows: when the excitation polarity signal is 1, the flip threshold = the average value of the sampled data within the half-cycle + the saturation depth setting value; when the excitation polarity signal is 0, the flip threshold = the average value of the sampled data within the half-cycle - the saturation depth setting value. Because the average value of the sampled data within the half-cycle is calculated dynamically, the flip threshold is also calculated dynamically, meaning that the flip threshold changes over time within each half-cycle.

[0135] The flip controller is responsible for generating an excitation polarity signal based on the flip threshold signal, interpolated data, and flip masking time value. Specifically, the flip controller is used to output the excitation polarity signal based on the comparison result between the floating flip threshold signal and the interpolated data outside the flip masking time. Its logic is as follows: if the current excitation polarity is 1, the current half-cycle time is greater than the flip masking time value, and the interpolated data is greater than the flip threshold + saturation depth setting value, then the excitation polarity is output as 0; if the current excitation polarity is 0, the current half-cycle time is greater than the flip masking time value, and the interpolated data is less than or equal to the flip threshold - saturation depth setting value, then the excitation polarity is output as 1.

[0136] Exemplarily, the digital processor further includes a period value calculation circuit for integrating and counting interpolated data within a half-cycle. When the output polarity flips, i.e., at the end of a half-cycle, the period value calculation circuit terminates integration and divides the integration result by the count of interpolated data within the half-cycle to obtain a half-cycle average value. The period value calculation circuit buffers at least two half-cycle average values ​​and sums the two most recent half-cycle average values ​​to obtain a measurement output signal, which is provided to the user as a digital measurement result. The value of this digital value is proportional to the current value of the measured current signal.

[0137] If digital interpolation is used, the excitation polarity signal may flip at any calculation cycle between two sampling data outputs from the analog-to-digital converter, resulting in a continuously changing phase difference between the excitation cycle and the sampling cycle. Although this can be compensated for using the aforementioned interpolation method, the interpolation model cannot fully fit the excitation current waveform. Therefore, when the phase difference between the excitation cycle and the sampling cycle is constantly changing, the phase difference will introduce errors in the judgment of the flipping time, resulting in large fluctuations in the flipping time when there is no measured current.

[0138] To reduce fluctuations in the flipping moment, the ADC is also connected to a flipping controller. This controller is also used to output a sampling synchronization signal to the ADC whenever the excitation polarity signal flips, thereby synchronizing the ADC. As shown in Figure 6, the flipping moment is used as the time when the ADC outputs sampled data. This ensures that the excitation flip always occurs at a relatively fixed time after the most recently output sampled data, reducing fluctuations in the results caused by interpolation errors.

[0139] As shown in Figure 7, the secondary coil current curve of half a cycle consists of three parts: the first part is to pull the magnetic material out of the deep saturation region. In this part, the magnetic material is in a saturated state, with low magnetic permeability, low inductance, and a large current slope; the second part is the linear region. In this part, the magnetic material is in an unsaturated state, with high magnetic permeability and a small current slope; the third part is to push the magnetic material into the deep saturation region. In this part, the magnetic material is in a saturated state again, with low magnetic permeability, low inductance, and a large current slope.

[0140] The judgment of the flipping moment inevitably has a certain degree of randomness. As shown in Figure 8, when the flipping is too early, the saturation depth of the magnetic material is shallow, and the secondary coil current-time curve will shift to the left as a whole; when the flipping is too late, the saturation depth of the magnetic material is deep, and the secondary coil current-time curve will shift to the right as a whole. The center offset T that determines the flipping threshold is offset When the secondary coil current-time curve is shifted left and right, it will lead to the calculation error of the center offset.

[0141] Since the slope of the secondary coil current curve of the first part and the third part is large, when the flipping moment is too early or too late, the instantaneous current values ​​of these two parts have a large error relative to the instantaneous current value at the same moment of complete and accurate flipping, which has a greater impact on the center offset calculation. Therefore, optionally, when calculating the center offset of the flipping threshold, the unidirectional or bidirectional saturation period of the magnetic material (i.e., the aforementioned first part and / or third part) can be avoided. For example, the center offset of the flipping threshold can be calculated based on a sampled voltage signal that is less than or equal to 7 / 8 of the length of the half cycle in each half cycle. Avoiding the saturation zone before and after the flip can be achieved based on an analog method or a digital method.

[0142] In summary, the current sensing device according to the embodiment of the present invention has at least the following advantages:

[0143] (1) Low power consumption

[0144] In traditional fixed-frequency toggle methods, to ensure sufficient saturation depth even when the measured current changes, the excitation voltage is typically maintained for a period after the current reaches saturation. During this period, the excitation current is very high, resulting in significant power consumption even when the measured current is zero.

[0145] Compared to the fixed-frequency flip method, the fixed-threshold flip method reduces system power consumption to a certain extent. However, because the flip threshold is fixed, the system design must ensure not only core saturation but also sufficient dynamic range to balance the magnetic field generated by the measured current. Therefore, the current threshold for fixed-threshold flipping is typically set high.

[0146] As shown in Figure 9, the current sensor device of an embodiment of the present invention utilizes a floating threshold flipping mechanism. The threshold automatically adjusts in a closed-loop manner based on the measured current, allowing the secondary coil current profile to fluctuate along the measured current waveform. Consequently, the saturation depth of the floating threshold flipping mechanism can be set very shallow, resulting in minimal power consumption when the measured current is zero. Consequently, the power consumption of the floating threshold flipping mechanism is significantly lower than that of the two aforementioned methods.

[0147] Figure 10 shows the secondary coil current waveforms (same scale) for fixed-frequency flipping, fixed-threshold flipping, and floating-threshold flipping under identical test conditions (top: fixed-frequency flipping, middle: fixed-threshold flipping, bottom: floating-threshold flipping). Since the operating voltage is the same, the system power consumption ratio of the three methods is the ratio of the RMS values ​​of their respective secondary coil current waveforms. Calculations based on the raw data in Figure 10 show that the RMS power consumption of the fixed-threshold flipping method is 1.29 times that of the floating-threshold flipping method, and its peak value is 2.39 times that of the floating-threshold flipping method. The RMS power consumption of the fixed-frequency flipping method is 3.51 times that of the floating-threshold flipping method, and its peak value is 3.48 times that of the floating-threshold flipping method. This shows that floating-threshold flipping significantly reduces both the RMS and peak power consumption of the system.

[0148] (2) High precision

[0149] Floating threshold flipping ensures relatively stable positive and negative saturation points of the core at different measured current values, significantly improving the system's linear accuracy. Figure 11 shows the integral nonlinearity (INL) curves for the three flipping methods. The INL for floating threshold flipping is significantly lower than that for the other two methods. The maximum INL for fixed-frequency flipping is 14.14 times that of floating threshold flipping, while the maximum INL for fixed-threshold flipping is 2.12 times that of floating threshold flipping.

[0150] (3) Strong anti-interference ability

[0151] Floating threshold flipping is essentially a closed-loop flipping method. Compared to open-loop methods like fixed-frequency flipping, closed-loop flipping is more susceptible to external interference and false flipping. False flipping is particularly noticeable with differential flipping: Fluctuations in the measured current are coupled to the secondary coil current. As long as the rate of change of the measured current is large enough, even extremely small fluctuations can cause false flipping. While fixed threshold flipping can also be affected by fluctuations in the measured current, false flipping only occurs when the measured current undergoes a sudden change of sufficient magnitude. Because it also uses current as the threshold, floating threshold flipping has similar interference immunity to fixed threshold flipping, but is far superior to differential flipping.

[0152] Figure 12 shows the secondary coil current waveform when using floating threshold flipping to measure a large-amplitude AC current close to the excitation frequency. Even with a high rate of change in the measured current, the excitation can still operate normally, and the period varies little with the measured current waveform.

[0153] Another embodiment of the present invention provides a magnetic modulation method for a current sensing device, the method comprising a magnetic core, a secondary coil wound around the magnetic core, and a sampling resistor connected in series with the secondary coil. The current sensing device may be the current sensing device described above. Referring to FIG. 13 , the magnetic modulation method 1300 includes the following steps:

[0154] In step S1310, a sampling voltage signal corresponding to the excitation current on the sampling resistor is collected;

[0155] In step S1320, a floating flip threshold signal is generated according to the sampled voltage signal;

[0156] In step S1330, an excitation polarity signal is output according to a comparison result between the sampled voltage signal and the floating flip threshold signal;

[0157] In step S1340 , an excitation drive signal is applied to the secondary coil, and the polarity of the excitation drive signal is reversed according to the excitation polarity signal, so that the magnetic core oscillates between a positive saturation point and a negative saturation point.

[0158] In one embodiment, generating a floating flip threshold signal according to the sampled voltage signal includes: filtering the sampled voltage signal to obtain a filtered signal, where the filtered signal is an analog signal; and generating the floating flip threshold signal according to the filtered signal.

[0159] In one embodiment, generating a floating flip threshold signal according to the sampled voltage signal includes: converting the sampled voltage signal into discrete sampled data, digitally processing the sampled data, and combining the sampled data with a saturation depth setting signal to obtain the floating flip threshold signal.

[0160] In one embodiment, the sampling data is digitally processed to obtain the floating flip threshold signal, including: interpolating the sampling data to obtain interpolated data; obtaining the floating flip threshold signal based on the interpolated data and the saturation depth setting signal; and outputting the excitation polarity signal based on the comparison result of the sampling voltage signal and the floating flip threshold signal, including: outputting the excitation polarity signal based on the comparison result of the floating flip threshold signal and the interpolated data outside the flip masking time.

[0161] For more specific details about magnetic modulation method 1300, please refer to the above description of the current sensing device and are not repeated here. The magnetic modulation method 1300 of the embodiment of the present invention uses a floating threshold flip method for magnetic modulation, achieving a magnetic modulation effect with higher precision, lower power consumption, and better frequency response than traditional methods, overcoming the limitations of fixed frequency flip, fixed threshold flip, and differential flip.

[0162] In the description provided herein, numerous specific details are described. However, it is understood that embodiments of the present invention may be practiced without these specific details. In some instances, well-known methods, structures, and techniques are not shown in detail so as not to obscure the understanding of this description.

[0163] Similarly, it should be understood that in order to streamline the present invention and aid in understanding one or more of the various inventive aspects, in the description of exemplary embodiments of the present invention, the various features of the present invention are sometimes grouped together into a single embodiment, figure, or description thereof. However, this approach to the present invention should not be interpreted as reflecting the intention that the claimed invention requires more features than those explicitly recited in each claim. More precisely, as reflected in the corresponding claims, the inventive point is that the corresponding technical problem can be solved with fewer features than all the features of a single disclosed embodiment. Therefore, the claims following the detailed description are hereby expressly incorporated into the detailed description, with each claim itself serving as a separate embodiment of the present invention.

[0164] Those skilled in the art will understand that, except where mutually exclusive, all features disclosed in this specification (including the accompanying claims, abstract, and drawings) and all processes or units of any method or apparatus disclosed herein may be combined in any combination. Unless expressly stated otherwise, each feature disclosed in this specification (including the accompanying claims, abstract, and drawings) may be replaced by an alternative feature that provides the same, equivalent, or similar purpose.

[0165] Furthermore, those skilled in the art will appreciate that although some embodiments described herein include certain features included in other embodiments but not other features, combinations of features from different embodiments are intended to be within the scope of the present invention and to form different embodiments. For example, in the claims, any of the claimed embodiments may be used in any combination.

[0166] It should be noted that the above embodiments illustrate rather than limit the invention, and that those skilled in the art may devise alternative embodiments without departing from the scope of the appended claims. In the claims, any reference signs placed between brackets should not be construed as limiting the claims. The invention may be implemented by means of hardware comprising several different elements and by means of a suitably programmed computer. In a unit claim enumerating several means, several of these means may be embodied by the same item of hardware. The use of the words first, second, and third etc. does not indicate any order. These words may be interpreted as names.

[0167] The foregoing description is merely a specific embodiment of the present invention or an illustration of a specific embodiment. The scope of protection of the present invention is not limited thereto. Any modifications or substitutions that can be readily conceived by a person skilled in the art within the technical scope disclosed in the present invention are intended to be encompassed by the scope of protection of the present invention. The scope of protection of the present invention shall be subject to the scope of protection of the claims.

Claims

1. A current sensing device, characterized in that: The device comprises: a magnetic core through which a conductor carrying a measured current signal passes; A secondary coil, wound on the magnetic core; A sampling resistor is connected in series with the secondary coil to provide a sampling voltage signal corresponding to the excitation current; an excitation control circuit, connected to the sampling resistor, for generating a floating flip threshold signal according to the sampling voltage signal, and outputting an excitation polarity signal according to a comparison result between the sampling voltage signal and the floating flip threshold signal; An excitation drive circuit is connected to the excitation control circuit and the secondary coil, and is used to apply an excitation drive signal to the secondary coil, and flip the polarity of the excitation drive signal according to the excitation polarity signal output by the excitation control circuit, so that the magnetic core oscillates between a positive saturation point and a negative saturation point.

2. The device according to claim 1, characterized in that The step of generating a floating flip threshold signal according to the sampled voltage signal comprises: Obtaining a center offset value according to the sampled voltage signal; The flip threshold is calculated according to the difference between the saturation depth setting value and the center offset value or the sum of the saturation depth setting value and the center offset value to generate the floating flip threshold signal.

3. The device according to claim 2, characterized in that The obtaining of a center offset value according to the sampled voltage signal comprises: The sampled voltage signal is low-pass filtered, and the center offset value is obtained according to a result of the low-pass filtering.

4. The device according to claim 2, characterized in that The obtaining of a center offset value according to the sampled voltage signal comprises: The center offset value of the current cycle is obtained according to the sampled voltage signal of the current cycle, or, The center offset value of the current cycle is obtained according to the sampled voltage signal of the historical cycle.

5. The device according to claim 4, characterized in that The obtaining the center offset value of the current cycle according to the sampled voltage signal of the current cycle includes: Obtaining the center offset value of the current cycle according to the fixed time value, average value or integrated value of the sampled voltage signal of the current cycle; The step of obtaining the center offset value of the current cycle according to the sampled voltage signal of the historical cycle includes: The center offset value of the current cycle is obtained according to the fixed time value, average value or integrated value of the sampled voltage signal in the historical cycle.

6. The device according to claim 1, characterized in that The excitation control circuit includes a filter, which is connected to the sampling resistor and is used to filter the sampled voltage signal to obtain a filtered signal, where the filtered signal is an analog signal; The excitation control circuit generates the floating flip threshold signal according to the filtered signal.

7. The device according to claim 6, characterized in that The filter includes a low-pass filter, and the filter signal includes a low-pass filter signal; the excitation control circuit also includes a threshold controller, a comparator and a trigger, wherein: The threshold controller is connected to the low-pass filter and is used to generate the floating flip threshold signal according to the low-pass filter signal, the excitation polarity signal and the saturation depth setting signal; The comparator is connected to the threshold controller, and is used for outputting a flip request signal when the sampled voltage signal reaches the floating flip threshold signal according to the sampled voltage signal, the floating flip threshold signal and the excitation polarity signal; The trigger is connected to the comparator and is used to output the excitation polarity signal according to the flip request signal.

8. The device according to claim 1, characterized in that The excitation control circuit comprises an analog-to-digital converter and a digital processor, the analog-to-digital converter is connected to the sampling resistor, and the digital processor is connected to the analog-to-digital converter; The analog-to-digital converter is used to convert the sampled voltage signal into discrete sampled data, and the digital processor is used to digitally process the sampled data and combine it with a saturation depth setting signal to obtain the floating flip threshold signal.

9. The device according to claim 8, characterized in that The digital processing frequency of the digital processor is higher than the analog-to-digital conversion frequency of the analog-to-digital converter.

10. The device according to claim 9, characterized in that The digital processor includes an upsampling circuit, a floating threshold calculation circuit and a flip controller; The up-sampling circuit is used to interpolate the sampled data to obtain interpolated data; The floating threshold calculation circuit is connected to the upsampling circuit and is used to obtain the floating flip threshold signal according to the interpolation data and the saturation depth setting signal; The flip controller is connected to the floating threshold calculation circuit and is used for outputting the excitation polarity signal according to the comparison result between the floating flip threshold signal and the interpolation data outside the flip masking time.

11. The device according to claim 10, characterized in that The interpolation includes interpolating first sampling data obtained in a first calculation cycle and second sampling data obtained in a second calculation cycle to obtain interpolation data corresponding to at least one third calculation cycle, and the third calculation cycle is located between the first calculation cycle and the second calculation cycle.

12. The device according to claim 10, characterized in that The interpolation includes interpolating the first-order derivative or the second-order derivative of the sampling data obtained in at least one calculation cycle.

13. The device according to claim 10, characterized in that The analog-to-digital converter is also connected to the flip controller, and the flip controller is further used to output a sampling synchronization signal to the analog-to-digital converter when the excitation polarity signal is flipped, so as to synchronize the analog-to-digital converter.

14. A magnetic modulation method, characterized in that: The method is used for a current sensing device, which includes a magnetic core, a secondary coil wound on the magnetic core, and a sampling resistor connected in series with the secondary coil; the method includes: Collecting a sampling voltage signal on the sampling resistor corresponding to the excitation current; Generate a floating flip threshold signal according to the sampled voltage signal; Outputting an excitation polarity signal according to a comparison result between the sampling voltage signal and the floating flip threshold signal; An excitation driving signal is applied to the secondary coil, and the polarity of the excitation driving signal is reversed according to the excitation polarity signal, so that the magnetic core oscillates between a positive saturation point and a negative saturation point.

15. The method according to claim 14, characterized in that The step of generating a floating flip threshold signal according to the sampled voltage signal comprises: Obtaining a center offset value according to the sampled voltage signal; The flip threshold is calculated according to the difference between the saturation depth setting value and the center offset value or the sum of the saturation depth setting value and the center offset value to generate the floating flip threshold signal.

16. The method according to claim 15, characterized in that The obtaining of a center offset value according to the sampled voltage signal comprises: The sampled voltage signal is low-pass filtered, and the center offset value is obtained according to a result of the low-pass filtering.

17. The method according to claim 15, characterized in that The obtaining of a center offset value according to the sampled voltage signal comprises: The center offset value of the current cycle is obtained according to the sampled voltage signal of the current cycle, or, The center offset value of the current cycle is obtained according to the sampled voltage signal of the historical cycle.

18. The method according to claim 17, characterized in that The obtaining the center offset value of the current cycle according to the sampled voltage signal of the current cycle includes: Obtaining the center offset value of the current cycle according to the fixed time value, average value or integrated value of the sampled voltage signal of the current cycle; The step of obtaining the center offset value of the current cycle according to the sampled voltage signal of the historical cycle includes: The center offset value of the current cycle is obtained according to the fixed time value, average value or integrated value of the sampled voltage signal in the historical cycle.

19. The method of claim 14, wherein: The step of generating a floating flip threshold signal according to the sampled voltage signal comprises: Filtering the sampled voltage signal to obtain a filtered signal, wherein the filtered signal is an analog signal; The floating flip threshold signal is generated according to the filtered signal.

20. The method of claim 14, wherein: The step of generating a floating flip threshold signal according to the sampled voltage signal comprises: The sampled voltage signal is converted into discrete sampled data, the sampled data is digitally processed, and combined with a saturation depth setting signal to obtain the floating flip threshold signal.

21. The method of claim 20, wherein: The digital processing of the sampled data to obtain the floating flip threshold signal comprises: interpolating the sampled data to obtain interpolated data; Obtaining the floating flip threshold signal according to the interpolation data and the saturation depth setting signal; Outputting the excitation polarity signal according to the comparison result between the sampling voltage signal and the floating inversion threshold signal includes: outputting the excitation polarity signal according to the comparison result between the floating inversion threshold signal and the interpolation data outside the inversion masking time.

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