Varactor-based switch for high frequency signals

Abstract

An electronic switching circuit (510a) for switching electric signals, the electronic switching circuit, which is realised by a variable LC filter of third order, comprises a varactor (501a), a parallel inductor (502a) arranged in parallel with the varactor (501a), and a series inductor (503a) arranged in series with the parallel circuit of the varactor and the parallel inductor, wherein a first terminal of the varactor (501a) is connected to a signal ground (504a) with a control voltage of the varactor (506a) as Direct Current, DC, potential of the signal ground, and a second terminal of the varactor is connected to the series inductor. Alternatively the varactor may be connected in series with the series inductor which series circuit may be connected in parallel with the parallel inductor.

Description

[0001] VARACTOR-BASED SWITCH FOR HIGH FREQUENCY SIGNALS

[0002] TECHNICAL FIELD

[0003] Embodiments disclosed herein relate to a varactor-based switch for high- frequency signals.

[0004] BACKGROUND

[0005] There is a trend towards higher carrier frequencies in wireless communications. In 5G, mm-waves are used for cellular communication between base-stations and user devices. The higher frequencies provide more spectrum for increased capacity and also wider bandwidth channels for increased user data rates. In 6G, even higher carrier frequencies are foreseen, in the so-called sub-THz band above 100 GHz.

[0006] In high frequency circuits there is often a need for switch functionality. Using one or more switches a circuit may be reconfigured by switching in different alternative parts, like different amplifier stages, different phase shifters, or different impedances. A switch may also be used to select if the receiver or transmitter should be connected to an antenna, a so-called TX / RX or TRX switch.

[0007] Switches come in different forms and complexity, depending on how many inputs and outputs they have. One common situation is that one signal node should by the switch be connected to one of a number of alternative nodes, a so-called single-pole multiple throw switch. The number of "poles" is the number of electrically separate switches which are controlled together. For example, a "2-pole" switch has two separate, parallel sets of contacts that open and close in unison via the same control mechanism. The number of "throws" is the number of separate wiring path choices other than "open" that the switch can adopt for each pole. A single-throw switch has one pair of contacts that may either be closed or open. A double-throw switch has a contact that may be connected to either of two other contacts, a triple-throw has a contact which may be connected to one of three other contacts, etc.

[0008] The TRX switch is an example of such a switch, a single pole dual throw switch. But there are also cases where more throws are needed, for instance when selecting between more than two amplifiers or phase shifts in a high frequency circuit. Another example is in lens-based systems with switched beams, where a front-end is connected to one of many different antenna elements radiating through a lens. The lens increases antenna gain by focusing the energy in one dominant direction, where the direction depends on the antenna element used. To enable beam steering functionality, only one antenna element is enabled at the time. The connection of one of several antenna elements to the front-end may be performed by a single-pole multi-throw switch. Below we describe the TRX switch as an example of currently used high frequency switches, but the structures may also be extended to more than two throws, like for the switched-beam lens-based system example.

[0009] In time division duplex (TDD) systems, a power amplifier (PA) of a transmitter and a low noise amplifier (LNA) of a receiver are typically connected to a TX / RX (or TRX) switch that directs a signal either from the PA to the antenna in transmit (Tx) mode or from the antenna to the LNA in receive (Rx) mode. Since high voltage levels exist at the antenna node in Tx mode, the switch must isolate the LNA from the PA such that these high signal levels do not reach the LNA input.

[0010] The following describes a basic functionality of a Tx / Rx switch.

[0011] Figure 1a shows a TRX switch in Tx mode. The Tx side switch is ON, which means it has a low (ideally zero) series impedance and the Tx signal gets through to the antenna. However, the Rx side switch is OFF and presents a large series impedance (ideally infinite) compared to the receiver input impedance thus protecting the Rx input from the high voltage levels at the antenna node.

[0012] Figure 1b shows the Tx / Rx switch in Rx mode. The Tx side switch is OFF in order to stop the received antenna signal from going to a Tx port, which would increase the Rx insertion loss. The Rx side switch is ON, presenting a low series impedance to the signal from the antenna for a low-loss connection to an Rx port. A port may comprise two terminals, such as a first terminal and a ground terminal. A Tx port may have a Tx terminal and a Tx ground terminal. An Rx port may have an Rx terminal and a Rx ground terminal.

[0013] The resistors of Figure 1a and 1b represent the output impedance of the transmitter RTX, the input impedance of the receiver RRX, and the impedance of the antenna RAnt, respectively. When used in a transceiver a power amplifier will be connected to the Tx port, a low noise amplifier to the Rx port, and an antenna to the antenna port.

[0014] To work properly as part of a radio transceiver, a TRX switch must have low Tx insertion loss, low Rx insertion loss, high TX to Rx isolation, high linearity and, preferably, enough bandwidth to cover a whole band of operation. A high Tx insertion loss would result in loss of output power and efficiency of the transmitter while a high Rx insertion loss results in loss of sensitivity for the receiver. Due to the different requirements, many different switch architectures have been developed depending on the importance of different parameters, frequency and bandwidth of operation, available semiconductor technology, etc.

[0015] To improve isolation between ports, more switches may be included in the structure above. For instance, a shunt switch may be used at the Rx port, to shunt leakage through the Rx side switch to ground in Tx mode, improving Tx to Rx isolation. To reduce effects of parasitic capacitances in the switch transistors at high frequencies, in switches intended for certain frequency bands, inductors may be used to resonate them. To improve linearity, different strategies may be used. One such strategy is to use stacked switch devices to reduce the voltage over each device, another is to use quarterwavelength transmission-line based transformers, so that the Rx side switch can be closed rather than open in Tx mode, to prevent the large Tx voltage from appearing across it.

[0016] As mentioned above, there is a trend towards using higher frequencies in wireless and cellular communications, driven by increased demands for data-rate.

[0017] As the operating frequencies increase, the implementation of an efficient TRX switch becomes more difficult since the integrated transistors cannot behave as an effective open switch due to the off-state capacitance. To have low loss when in ON state, the device size must be large, but this results in a large off-state capacitance which causes the impedance to ground to become low, increasing the loss. A figure of merit that may be used to determine the switch performance of a transistor is the off-state capacitance multiplied by the on-state resistance, which should be as low as possible. A ratio of off-state impedance and on-state impedance at a certain operating frequency depends on this figure of merit, but it is also proportional to the inverse of the frequency, as the impedance of the capacitance drops with frequency. This will also be true in a tuned structure where an inductor is used to resonate with the off-state capacitance, assuming a frequency flat quality factor (Q). The ratio will then have the same behavior with operating frequency, where the circuit is tuned to resonate, but will be Q times higher.

[0018] Switch performance will therefore degrade with frequency, even with very advanced semiconductor technologies, and switches at sub-THz for coming 6G systems will have limited performance. If switches with sufficient performance cannot be implemented, it will limit the choices of transceiver architectures. For instance, instead of using TRX switches, separate Rx and Tx antennas may be used, increasing the size of the equipment.

[0019] There is therefore a large need for techniques for TRX switch structures that operate with good performance at high frequencies.

[0020] A prior art switch structure has been disclosed in WO 2018 / 145758 A1. The prior art switch structure uses an LC tank in series with a capacitor. Both capacitors, where one is part of the LC tank and the other is in series with the LC tank, are implemented as a combination of a fixed passive capacitor and a Complementary Metal-Oxide Semiconductor (CMOS) device acting as a series switch, i.e. a switched capacitor. However, the solution described in WO 2018 / 145758 A1 has a limited high-frequency performance.

[0021] SUMMARY

[0022] The circuit described in WO 2018 / 145758 A1 relies on the R0nC0ff product of the CMOS device acting as a switch. This limits the high-frequency performance of that solution.

[0023] In addition, in WO 2018 / 145758 A1 , since there are two variable / programmable capacitors present, at least one of them must be implemented such that none of its terminals are connected to a signal ground.

[0024] This will affect circuit performance since such a variable / programmable capacitor will have additional parasitic capacitances, including resistive losses, associated with it, from the transistor either used as switch or varactor. The parasitic is then present in a node with a high frequency signal, also reducing circuit tunability.

[0025] There is thus a need for a switch with better high-frequency performance.

[0026] An object of embodiments herein may be to obviate some of the problems related to high-frequency switches.

[0027] Embodiments disclosed herein use a varactor as a switching element instead of a switch in series with a passive capacitor. Its performance then depends on the ratio Con / Coff of the varactor, and the quality factor of the varactor and inductors. This enables operation with high performance at much higher frequencies, as will be explained in detail below.

[0028] According to a first aspect, the object is achieved by an electronic switching circuit. The switching circuit may be adapted for switching high frequency electric signals, such as millimetre waves or sub-THz frequencies. The electronic switching circuit comprises a varactor, a parallel inductor arranged in parallel with the varactor, and a series inductor arranged in series with the varactor.

[0029] A first terminal of the varactor is connected to a signal ground with a control voltage of the varactor as Direct Current, DC, potential of the signal ground. A second terminal of the varactor is connected to the series inductor.

[0030] According to a second aspect, the object is achieved by a wireless transceiver arrangement comprising the electronic switching circuit according to the first aspect above.

[0031] According to a third aspect, the object is achieved by a wireless transceiver node comprising the wireless transceiver arrangement according to the second aspect above.

[0032] A high-performance varactor, such as a varactor with high Q and high ratio between an on-capacitance (Con) and an off-capacitance (COff), combined with inductors in a resonating switch structure provides improved performance of switches at very high frequencies. The resonating switch structure has a resonance frequency given by the capacitance and the inductances.

[0033] The varactor connected with parallel and series inductors provide a low ohmic series resonance when the varactor presents its on-capacitance at the operating frequency, and a high ohmic parallel resonance when it presents the off-capacitance at the operating frequency.

[0034] Depending on whether the varactor is in on or off mode, a low or high impedance will then be presented, so that the structure with the varactor and the two inductors may be used as a switch. Often varactors have significant parasitic capacitances associated with one terminal, which is then to be connected to a control voltage, i.e. a signal ground. The switch then becomes a shunt switch to signal ground. Such a shunt switch may be used to realize a single pole multi throw switch, by connecting quarter wavelength transmission lines to a common port, such as an antenna port, and at each transmission line end, where the other ports are, place a shunt switch. In the case of a TRX switch, two transmission lines are needed, one for Tx and one for Rx, and at each Tx and Rx port there will be a shunt switch according to the above.

[0035] The two inductors may be arranged in two different ways in the switching circuit. In a first alternative a first inductor, i.e. the series inductor, is placed in series with the varactor, to provide a series resonance when the varactor is on, and then a second inductor, i.e. the parallel inductor, is placed in parallel with the series connected components, to provide a parallel resonance with the series connection of the varactor in its off-state and the first inductor. In the second alternative the second inductor is connected in parallel with the varactor, to provide a parallel resonance when the varactor is off, and then the first inductor is placed in series with the parallel connected components, to provide a series resonance with the parallel connection of the varactor in its on state and the second inductor. The two alternative circuits have the same property of providing a low impedance through series resonance when the varactor is on, and a high impedance through parallel resonance when the varactor is off.

[0036] Furthermore, at high frequencies the varactors will have a lower quality factor than inductors, and the overall quality factor will therefore be better using a single varactor and two inductors, compared to using two varactors and a single inductor. As will be seen below the quality factor of the switch structure helps in obtaining a high switch performance, and the switch structures of embodiments disclosed herein therefore have superior performance compared to the switch structure in the prior art, even if a switched capacitor in WO 2018 / 145758 A1 would be replaced by a varactor.

[0037] BRIEF DESCRIPTION OF THE DRAWINGS

[0038] In the figures, features that appear in some embodiments are indicated by dashed lines.

[0039] The various aspects of embodiments disclosed herein, including particular features and advantages thereof, will be readily understood from the following detailed description and the accompanying drawings, in which:

[0040] Figure 1a is a block diagram schematically illustrating a first operating mode of a transceiver (TRX) switch circuit according to prior art,

[0041] Figure 1b is a block diagram schematically illustrating a second operating mode of a transceiver (TRX) switch circuit according to prior art,

[0042] Figure 2a is a circuit diagram illustrating a switching circuit according to some embodiments herein,

[0043] Figure 2b is a circuit diagram symbol illustrating a switching circuit according to some other embodiments herein,

[0044] Figure 2c is a circuit diagram symbol illustrating a varactor according to some embodiments herein, Figure 2d is a circuit diagram illustrating a further varactor according to some other embodiments herein,

[0045] Figure 3 is a block diagram illustrating a switching circuit according to some embodiments herein,

[0046] Figure 4 is a graph of impedance characteristics of a filter circuit according to some embodiments herein,

[0047] Figure 5 is a circuit diagram illustrating a transceiver (TRX) arrangement according to some embodiments herein,

[0048] Figure 6 is an S-parameter graph illustrating circuit performance of a receiver switching circuit according to some embodiments herein,

[0049] Figure 7 is an S-parameter graph illustrating circuit performance of a transmitter switching circuit according to some embodiments herein,

[0050] Figure 8 is a graph of voltage levels vs. time at different nodes in a transceiver arrangement circuit according to some embodiments herein,

[0051] Figure 9 is a block diagram schematically illustrating a wireless transceiver node according to some embodiments herein,

[0052] Figure 10 is a block diagram schematically illustrating a wireless communications network in which embodiments herein may be implemented.

[0053] DETAILED DESCRIPTION

[0054] Embodiments herein relate to high-frequency switches for wireless communications.

[0055] Unlike existing solutions, embodiments disclosed herein rely on a varactor instead of a transistor switch. This makes it possible to avoid the limitations imposed by the reliance on the R0n*C0ff performance of the transistor switch. Instead of relying on low on- resistance and low off capacitance of the transistor switch, the embodiments disclosed herein work based on the capacitance ratio between the on-capacitance and the off- capacitance of the varactor. This enables very high frequency operation together with lower insertion loss of the switch. The varactor (variable reactor), sometimes also referred to as a varicap diode, varactor diode, variable capacitance diode, variable reactance diode or tuning diode, is a device whose reactance can be varied in a controlled manner with a bias voltage. It is a p-n junction with a special impurity profile, and its capacitance variation is very sensitive to reverse-biased voltage. An increase in reverse bias increases the width of a gap (d) which reduces the capacitance (C) of the PN junction. Therefore, the capacitance of the varactor is reduced with an increased applied reverse bias. The ratio of varactor capacitance to reverse-bias voltage change may be as high as 10 to 1 . The varactor capacitance ratio is the ratio of the diode capacitance at a minimum reverse voltage to the diode capacitance at a maximum reverse voltage.

[0056] Varactors may be implemented by MOS technology. An nMOS varactor may have the same structure as an nMOS transistor, with gate as the first terminal and drain, source, and bulk connected together to form the second terminal. MOS varactors operate in four main regions, based on the biasing point (voltage across the varactor terminals): accumulation, depletion, weak inversion, and strong inversion. Accumulation and strong inversion are two regions where most varactors are designed to operate in when in the on-state.

[0057] Varactors may be used as voltage-controlled capacitors. They are commonly used in voltage-controlled oscillators, parametric amplifiers, and frequency multipliers. Voltage- controlled oscillators have many applications such as frequency modulation for FM transmitters and phase-locked loops. Phase-locked loops are used for the frequency synthesizers that tune many radios, television sets, and cellular telephones.

[0058] A key advantage of embodiments disclosed herein comes from the structure operating in resonance both in the on-state and the off-state. A transistor switch may also operate in resonance, but only in the off-state when the impedance is capacitive. A parallel inductor may then be used to increase the off-state impedance by a factor equal to the quality factor (Q) of the resulting resonance circuit, limited by losses in both the inductor and the capacitor. In the on-state, however, the impedance is resistive, and it is not possible to reduce it by forming a resonance. In the varactor-based switch, however, the impedance is capacitive also in the on-state, and by forming a network providing series resonance in the on-state and parallel resonance in the off-state, also the on-state may be improved, having its impedance reduced by a factor equal to Q.

[0059] Figures 2a and 2b show two alternative varactor-based electronic switching circuits 200a, 200b for switching electric signals. In some embodiments herein the electric signals to be switched is in a frequency range of millimetre waves or sub-THz frequencies.

[0060] The switching circuits 200a, 200b may be configured to operate as shunt switches. The varactor-based electronic switching circuits 200a, 200b may be integrated circuits.

[0061] The switching circuits 200a, 200b each comprise a varactor 201a, 201b. The varactor 201a, 201b may be any one of: a Metal Oxide Semiconductor, MOS, device, a Micro-ElectroMechanical Systems (MEMS) device, or a pn junction diode, or a Schottky diode.

[0062] An advantage of the MOS device varactor is a high Q and a high Con / Coff ratio. A further advantage with the MOS device varactor is that it may be integrated with integrated electronics, such as an integrated transceiver circuit.

[0063] Figure 2c illustrates the varactor 201a, 201b as a circuit diagram symbol for a MOS device with two terminals: a first terminal 230 and a second terminal 240.

[0064] In some embodiments herein the terminals of the varactor 201a, 201b may be an anode and a cathode. Which terminal that is the anode and which terminal is the cathode depends on a diode device type. For example, for an nMOS type device the anode is the second terminal 240 and the cathode is the first terminal 230. For some MOS-based varactors the terminals of the varactor 201a, 201b may be a gate terminal and channel (or source and drain) terminal.

[0065] Figure 2d illustrates the varactor 201a, 201b as a circuit diagram symbol for a MOS device with three terminals: the first terminal 230 (e.g. source or drain), the second terminal 240 (e.g. gate) and a third terminal 250 (e.g. body or back-gate).

[0066] The varactor 201a, 201b may be controlled by e.g., changing the DC potential of the first terminal 230 of the varactor 201a, 201b.

[0067] Each switching circuit 200a, 200b further comprises a parallel inductor 202a, 202b arranged in parallel with the varactor 201a, 201b, and a series inductor 203a, 203b arranged in series with the varactor 201a, 201b.

[0068] The first terminal 230 of the varactor 201a, 201b is connected to a signal ground 204a, 204b. A control voltage of the varactor 201a, 201b is a Direct Current, DC, potential of the signal ground 203a, 203b. The second terminal 240 of the varactor 201a, 201b is connected to the series inductor 203a, 203b. The series inductor 203a, 203b is connected to a high-frequency signal. The electric signal, such as a high-frequency electric signal, may be applied to the second terminal 240 of the varactor. The electric signal is applied via the series inductor 203a, 203b.

[0069] This is a difference compared to the prior art disclosure of WO 2018 / 145758 A1 described above where there are two variable / programmable capacitors present, and at least one of them must be implemented such that none of its terminals are connected to a signal ground.

[0070] Embodiments herein offer better circuit performance compared to the performance of the switch of the prior art disclosure of WO 2018 / 145758 A1 since a varactor, such as a single varactor, is used with one terminal connected to signal ground 204a, 204b. This avoids having additional parasitic capacitances related to variable / programmable capacitors in a node with a high frequency signal.

[0071] The switching circuits 200a, 200b may each further comprise a first node N1a, N1b, a second node N2a, N2b and a third node N3a, N3b. The varactor 201a, 201b may be connected to the first node N1a, N1b and the second node N2a, N2b and the series inductor 203a, 203b may be connected to the second node N2a, N2b and the third node N3a, N3b.

[0072] The parallel inductor 202a, 202b may be connected to the first node N1a, N1b and the second node N2a, N2b or to the first node N1a, N1b and the third node N3a, N3b.

[0073] In both alternatives of the switching circuit 201 a, 201b the varactor 201 a, 201b is combined with the parallel inductor 202a, 202b with an inductance Lpa, LPb arranged in parallel with the varactor 201a, 201b and the series inductor 203a, 203b with an inductance Lsa, LSb arranged in series with the varactor 201a, 201b.

[0074] In both alternatives of the switching circuit 201a, 201b the parallel inductor 202a, 202b and the series inductor 203a, 203b are in series resonance with the varactor 201a, 201b at the operating frequency when the varactor 201a, 201b is set to its on-state capacitance. The on-state capacitance, or in short, the on-capacitance Conis a highest capacitance setting of the varactor obtainable by the control voltage. The control voltage may be both negative and positive at the highest capacitance setting. At the on- capacitance setting the switch is in a low-impedance state and is closed.

[0075] The parallel inductor 202a, 202b and the series inductor 203a, 203b are in parallel resonance with the varactor 201a, 201b when the varactor 201a, 201b is set to its off- state capacitance.

[0076] The off-state capacitance, or in short, the off-capacitance COff is a lowest capacitance setting of the varactor obtainable by the control voltage. The control voltage may be both negative and positive at the lowest capacitance setting. At the off- capacitance setting the switch is in a high-impedance state and is open.

[0077] In a communication context, the operating frequency may be the centre frequency of the communication frequency band, or some other frequency within that frequency band, e.g. close to the centre frequency. A first varactor-based electronic switching circuit 200a according to the first alternative will now be described with respect to Figure 2a.

[0078] In the first electronic switching circuit 200a the series inductor 203a is further arranged in series with the parallel inductor 202a.

[0079] In the first varactor-based electronic switching circuit 200a, when the varactor 201a is off, it has a low capacitance, COff. The impedance presented by the shunt switch, Za, is then as high as possible and the switch open. This is accomplished by resonating the varactor capacitance COff with the parallel inductance Lpafrom the parallel inductor 202a.

[0080] The inductance Lpaof the parallel inductor 202a is then given by: where COff is an off-capacitance of the varactor 201 a and co = 2TTf where f is a circuit operating frequency of the electronic switching circuit 200a, 200b.

[0081] The off-capacitance is a lowest capacitance value of the varactor obtainable by the control voltage.

[0082] In the first varactor-based electronic switching circuit 200a, when the varactor 201a is on, with a high capacitance Con, the impedance Zashould be as low as possible. The series inductance Lsashould then be in series resonance with the combination of Conand Lpa. Since the parallel inductor 202a is designed to be in resonance with COff, its reactance at the operating frequency corresponds to a negative capacitance COff. The parallel connection of Conand minus COff, yields a capacitance of C0n-C0ff, that should be in resonance with inductance Lsa, so that a notch in impedance Zais seen at the operating frequency. The series inductance Lsaof the series inductor 203a is then given by: where Conis an on-capacitance of the varactor 201a and COff is an off-capacitance of the varactor 201a.

[0083] A second varactor-based electronic switching circuit 200b according to the second alternative will now be described with respect to Figure 2b.

[0084] In the second electronic switching circuit 200b the parallel inductor 202b is further arranged in parallel with the series inductor 203b.

[0085] In the second varactor-based electronic switching circuit 200b, when the varactor 201 b is on, it has a high capacitance, Con. The impedance presented by the shunt switch, Zb, is then as low as possible. This is accomplished by resonating the varactor capacitance Conwith the series inductance LSb from the series inductor 203b. In the second electronic switching circuit 200b the series inductance LSb of the series inductor 203a is given by:

[0086] 1 Lsb =o>2■ C where Conis an on-capacitance of the varactor 201 b.

[0087] In the second varactor-based electronic switching circuit 200b, when the varactor 201 b is off, with a low capacitance COff, the impedance Zb should be as high as possible. The parallel inductance LPb should then be in parallel resonance with the combination of Coff and Lsb. Since the series inductor 203b (with inductance LSb) is designed to be in resonance with Con, its reactance at the operating frequency corresponds to a negative capacitance Con. The series connection of COff and minus Conyields an equivalent capacitance of where Conis the on-capacitance of the varactor 201b and where COff is the off- capacitance of the varactor 201 b.

[0088] This equivalent capacitance, Cpeq, should be in resonance with inductance LPb, so that a peak in impedance Zb is seen at the operating frequency. LPb may then be calculated as:

[0089] As indicated above, the varactor-based switch has advantages at very high frequencies, such as millimetre waves or sub-THz frequencies, where regular switches have problems achieving high performance in terms of insertion loss and isolation between ports. The following back-of-the-envelope calculations provide some guidance to what frequencies are suitable for the switches according to embodiments disclosed herein.

[0090] A prior art shunt switch with an on resistance Ronand an off capacitance COffSWis used as comparison. For the prior art switch it is assumed that a parallel inductor is used to increase the impedance of COffSWQ times at the frequency of operation, providing ROff. The quality factor Q is limited both by losses in the parasitic capacitance COffSWand in the inductor. Even if the Q would be very high, concerns about bandwidth and robustness to variations would limit the benefits. The ratio of off-resistance to on-resistance of the prior art shunt switch becomes:

[0091] For the varactor-based shunt switch according to embodiments herein (both the first varactor-based electronic switching circuit 200a and the second varactor-based electronic switching circuit 200b, the ratio becomes:

[0092] Let us assume a Q-value of 10 at 100 GHz in both cases, for a bandwidth suitable for a 6G system and some robustness. There are prior art varactors with Con / Coff ratio equal to 4, and Q of 7 at 200 GHz, so Q=10 at 100 GHz (including inductor losses) is achievable. Assume for the prior art switch an R0n*C0ff product of 100 fs, which is what may be achieved with 22 nm Fully Depleted Silicon On Insulator FDSOI technology. The ratio for the prior art switch then becomes 159, and for the varactor-based switch according to embodiments herein it becomes 300, clearly showing the benefit at this high frequency.

[0093] Simulations

[0094] Embodiments of the electronic switching circuit 200a, 200b will now be exemplified with a structure according to figure 2(a), here called a third order filter, designed for an operating frequency fo = 200 GHz. At these frequencies, typical transistor switches are unable to provide a high OFF / ON impedance ratio.

[0095] The varactor 201 a, 201b may be fabricated in a 55 nm Bipolar CMOS (BiCMOS) technology. When the varactor 201a, 201b operates at 200 GHz it may have a Q of 7 and a Con / Coff ratio of 4. These values have been used for the varactor model in the simulations described below to ensure realistic performance estimation of the TRX switch. Using the structure shown in Figure 2(a), the designed filter has values of

[0096] Cvar,OFF= 12 fF

[0097] Cvar,ON=48 fF

[0098] Lpa= 52.5 pH Lsa= 17.5 pH

[0099] A block diagram of a wireless transceiver arrangement 300 is illustrated in Figure 3. The transceiver arrangement 300 comprises the electronic switching circuit 200a, 200b according to Figure 2a or 2b.

[0100] In particular, the transceiver arrangement 300 comprises two electronic switching circuits 310a, 310b which each corresponds to the embodiments disclosed above of the varactor-based electronic switching circuits 200a, 200b of Figures 2a and 2b. Further, each electronic switching circuit 200a, 200b may be implemented as a third order filter.

[0101] A first receiver switching circuit 310a is arranged between an antenna 301 and an LNA 360 of a receiver and a second transmitter switching circuit 310b is arranged between the antenna 301 and a PA 370 of a transmitter. The transceiver arrangement 300 further comprises a quarter wavelength transmission line 313a, 313b between the antenna 301 and the respective switching circuit 310a, 310b.

[0102] The impedance characteristics of the third order filter are shown in Figure 4Error! Reference source not found.. As seen in Figure 4 the high impedance at 200 GHz is above 1 kQ when the varactor is OFF and the low impedance is around 5.5 Q at 200 GHz when the varactor is switched ON. The on / off ratio of the impedance is about 180.

[0103] Figure 5 illustrates a further wireless transceiver arrangement 500 in which embodiments disclosed herein may be implemented. Specifically, the wireless transceiver arrangement 500 may comprise a TRX switch based on the circuit structure of Figure 3.

[0104] The transceiver arrangement 500 comprises an antenna 501. The antenna 501 is connected to a transmitter terminal 570 illustrated on the right side and a receiver terminal 560 illustrated on the left side. The transceiver arrangement 500 further comprises a transmitter 90° transmission line 513a arranged between the transmitter terminal 570 and the antenna 501 and a receiver 90° transmission line 513b arranged between the receiver terminal 560 and the antenna 501. Each 90° transmission line 513a, 513b is modelled as two pi networks joined together.

[0105] The transceiver arrangement 500 further comprises a transmitter switch circuit 510a arranged between the transmitter terminal 570 and a transmitter side ground 504a. The transceiver arrangement 500 further comprises a receiver switch circuit 510b arranged between the receiver terminal 560 and a receiver side ground 504a. The switch circuits may also be referred to as filters. The transmitter switch circuit 510a comprises a transmitter varactor 501a. The receiver switch circuit 51 Ob comprises a receiver varactor 501 b.

[0106] The transmitter switch circuit 510a further comprises a transmitter parallel inductor 502a. The receiver switch circuit 510b further comprises a receiver parallel inductor 502b.

[0107] The transmitter switch circuit 510a further comprises a transmitter serial inductor 503a. The receiver switch circuit 510b further comprises a receiver serial inductor 503b.

[0108] The switch circuits 510a, 510b may each further comprise a decoupling capacitor 505a, 505b, and a driver circuit 506a, 506b.

[0109] It should be observed that the respective terminal of the two varactors 501a, 501b connected to the decoupling capacitor 505a, 505b may be connected to different DC potentials, but if both are signal ground, for signals they are considered as connected together.

[0110] In some embodiments herein the transceiver arrangement 300 comprises a common port, a first port and a second port. Each port may comprise a signal terminal and a ground terminal. Then the electronic switching circuit 200a, 200b of the transceiver arrangement 300 is configurable between two modes. In the first mode the electronic switching circuit 200a, 200b passes the electric signals between the first port and the common port.

[0111] In the second mode the electronic switching circuit 200a, 200b passes the electric signals between the second port and the common port.

[0112] In some embodiments herein the wireless transceiver arrangement 300, 500 comprises a single-pole multi-throw switch 320, 520. Then the common port of the electronic switching circuit 200a, 200b may be the pole. The first port and the second port may each be a throw.

[0113] The single-pole multi-throw switch 320, 520 may comprise, for each throw, the electronic switching circuit 310a, 310b and a quarter wavelength transmission line 313a, 313b. The quarter wavelength transmission line 313a, 313b, 513a, 513b is arranged between a pole port of the single-pole multi-throw switch 320, 520 and the electronic switching circuit 510a, 510b which is arranged between the quarter wavelength transmission line 313a, 313b 513a, 513b and a throw port of the single-pole multi-throw switch 320, 520. In some embodiments herein the wireless transceiver arrangement 300, 500 further comprises the antenna 501 , the transmitter port connected to a transmitter throw port of the single-pole multi-throw switch 320, 520, and a receiver port connected to a receiver throw port of the single-pole multi-throw switch 320, 520. Then the electronic switching circuit 510a, 510b comprises a transmitter switching circuit 510a and a transmitter quarter wavelength transmission line 513a both arranged between the transmitter port and the antenna 501 and a receiver switching circuit 510a and a receiver quarter wavelength transmission line 513b both arranged between the receiver port and the antenna 501.

[0114] Each quarter wavelength transmission line 513a, 513b of the wireless transceiver arrangement 300, 500 may comprise lumped inductors and capacitors.

[0115] In some embodiments herein the lumped inductors and capacitors of the quarter wavelength transmission line 513a, 513b form one or more pi networks, preferably two or three pi networks. With more inductors the quarter wavelength transmission line 513a, 513b behaves more like an ideal transmission line but also takes up more space. Therefore two pi networks is a good trade-off.

[0116] The single-pole multi-throw switch 320, 520 may be configurable between at least two operating modes, wherein for each operating mode the varactor of one of the electronic switching circuits 510a, 510b is configured to be in an off state and the respective varactor of the other of the electronic switching circuits 510a, 510b is configured to be in an on state.

[0117] In some embodiments herein the wireless transceiver arrangement 300, 500 is configurable between a transmitter operating mode and a receiver operating mode. In the transmitter operating mode the varactor 511a of the transmitter switching circuit 510a is configured to be in an off state and the varactor 511 b of the receiver switching circuit 510b is configured to be in an on state such that the single-pole multi-throw switch 320, 520 is configured to pass the electric signals from the transmitter port to the antenna 501. In the receiver operating mode the varactor 511b of the receiver switching circuit 510b is configured to be in an off state and the varactor 511a of the transmitter switching circuit 510a is configured to be in an on state, such that the single-pole multi-throw switch 320, 520 is configured to pass the electric signals from the antenna 501 to the receiver port. To switch between Tx and Rx operation mode, the varactor of one of the switch circuits is switched ON and the varactor of the other switch circuit is switched OFF. In other words, the varactor is switched ON in one filter and the other varactor is switched OFF in the other filter to switch between Tx and Rx operation mode. When the varactor is turned ON the impedance is low at the operating frequency, in this case 200 GHz, and the switch is closed and signals will be shunted to signal ground and not pass (will be significantly attenuated). When the varactor is turned OFF the impedance is high at the operating frequency, in this case 200 GHz, and the switch is open and the signals are not shunted to signal ground and may then pass with low insertion loss. For example, if the transmitter varactor 501a is ON and the receiver varactor 501b is OFF then signals to and from the receiver port 502a will pass with low insertion loss while signals to and from the transmitter port 502b will experience high insertion loss. The low resistance (when the varactor is ON) is a result of series resonance occurring at the operating frequency and the high resistance (when the varactor is OFF) is a result of the parallel resonance occurring at the operating frequency. The quarter wavelength line works as an impedance inverter causing low impedance at one end to look like high impedance at the other end and vice versa.

[0118] The transmission line inductors as well as the inductors in the filter structures may have a Q of 20. Inductors of similar inductance values have been reported in literature that have Q values of higher than 20 and a self-resonance frequency of above 500 GHz. Circuit performance of the receiver switch circuit 510b is shown in Figure 6. Circuit performance of the transmitter switch circuit 510a is shown in Figure 7. The performance in Tx and Rx modes is identical since the circuit of the transceiver arrangement 500 is symmetrical.

[0119] A Tx to Rx isolation of about 20 dB is achieved while the insertion loss is around 1.2 dB at 200 GHz. As can be seen in Figures 6 and 7, the insertion loss is low in a large bandwidth around 200 GHz. Thus, the antenna 301 is matched across a large bandwidth. The 1-dB bandwidth of the circuit is more than 40 GHz.

[0120] Figure 8 shows a harmonic balance simulation performed to verify voltage levels at different nodes in the circuit of the transceiver arrangement 500. As can be seen in Figure 8, when the transmitter is driving the transmitter port 502a with +15 dBm at 200 GHz, the maximum voltage level at the receiver port 502b is less than 0.2 V, which ensures that the LNA of the receiver will be protected while peak power is transmitted. The maximum voltage across the receiver varactor device is 0.6 V, which is easily tolerated by active varactor devices of most commercial technologies.

[0121] Figure 9 illustrates a wireless transceiver node 701 , such as a wireless communications device or a radio access node. The wireless transceiver node 701 comprises the wireless transceiver arrangement 300, 500.

[0122] The wireless transceiver node 701 may comprise a processor or one or more processors, such as the processor 704 of a processing circuitry in the wireless transceiver node 701 and depicted in Figure 7 together with computer program code for performing functions and actions of the embodiments herein, for example to apply voltages. The program code mentioned above may also be provided as a computer program product, for instance in the form of a data carrier carrying computer program code for performing the embodiments herein when being loaded into the wireless transceiver node 701. One such carrier may be in the form of a CD ROM disc. It is however feasible with other data carriers such as a memory stick. The computer program code may furthermore be provided as pure program code on a server or a cloud and downloaded to the wireless transceiver node 701.

[0123] The wireless transceiver node 701 may further comprise a memory 702 comprising one or more memory units. The memory comprises instructions executable by the processor in the wireless transceiver node 701.

[0124] The memory 702 is arranged to be used to store e.g. information, data, configurations, and applications to perform methods herein when being executed in the wireless transceiver node 701.

[0125] In some embodiments, a computer program 703 comprises instructions, which when executed by the at least one processor, cause the at least one processor of the wireless transceiver node 701 to perform the actions above.

[0126] In some embodiments, a carrier 705 comprises the computer program, wherein the carrier is one of an electronic signal, an optical signal, an electromagnetic signal, a magnetic signal, an electric signal, a radio signal, a microwave signal, or a computer- readable storage medium. The wireless transceiver node 701 may further comprise an input and output interface, I / O, 706 configured to communicate with other devices. The input and output interface 706 may comprise one or more transceivers, such as a wireless transceiver. The wireless transceiver may comprise the transceiver arrangement 500.

[0127] Those skilled in the art will also appreciate that the units described above may refer to a combination of analog and digital circuits, and / or one or more processors configured with software and / or firmware, e.g. stored in the wireless transceiver node 701 , that when executed by the respective one or more processors such as the processors described above cause the one or more processors to carry out actions described herein.

[0128] One or more of these processors, as well as the other digital hardware, may be included in a single Application-Specific Integrated Circuitry (ASIC), or several processors and various digital hardware may be distributed among several separate components, whether individually packaged or assembled into a system-on-a-chip (SoC).

[0129] Embodiments herein relate to wireless communications networks in general. Figure 10 is a schematic overview depicting a wireless communications network 170 wherein embodiments herein may be implemented.

[0130] The wireless communications network 170 may be a telecommunications network, such as a cellular network. For example, the wireless communications network 170 may comprise one or more Radio Access Networks (RAN) and one or more Core Networks (ON).

[0131] The wireless communications network 170 may use a number of different technologies, such as Wi-Fi, Long Term Evolution (LTE), LTE-Advanced, 5G, New Radio (NR), Wideband Code Division Multiple Access (WCDMA), Global System for Mobile communications / enhanced Data rate for GSM Evolution (GSM / EDGE), Worldwide Interoperability for Microwave Access (WiMax), or Ultra Mobile Broadband (UMB), just to mention a few possible implementations. Embodiments herein relate to recent technology trends that are of particular interest in a 5G context, however, embodiments are also applicable in further development of the existing wireless communication systems such as e.g. WCDMA and LTE and in future wireless communication systems, such as 6G systems.

[0132] Access nodes, such as a first radio access node 111, operate in the RAN of the wireless communications network 170. The first radio access node 111 provides radio coverage over a geographical area, a service area referred to as a cell 115, which may also be referred to as a beam or a beam group of a first radio access technology (RAT), such as 5G, LTE, Wi-Fi or similar. There may also be further cells for which radio coverage is provided by the first radio access node 111 , such as a second cell 116.

[0133] There may also be further radio access nodes, such as a second radio access node (not shown). The second radio access node may provide radio coverage over a third cell and / or a fourth cell (not shown).

[0134] The first radio access node 111 and the second radio access node may each be a NR-RAN node, transmission and reception point e.g. a base station, a radio access node such as a Wireless Local Area Network (WLAN) access point or an Access Point Station (AP STA), an access controller, a base station, e.g. a radio base station such as a NodeB, an evolved Node B (eNB, eNode B), a gNB, a base transceiver station, a radio remote unit, an Access Point Base Station, a base station router, a transmission arrangement of a radio base station, a stand-alone access point or any other network unit capable of communicating with a wireless device within the service area depending e.g. on the radio access technology and terminology used. The respective first and second radio access node may be referred to as a serving radio access node and communicates with a UE with Downlink (DL) transmissions to the UE and Uplink (UL) transmissions from the UE.

[0135] A number of wireless communications devices operate in the wireless communications network 170, such as a wireless communications device 121. The wireless communications device 121 may for example be a UE.

[0136] The wireless communications device 121 may be a mobile station, a non-access point (non-AP) STA, a STA, a user equipment and / or a wireless terminal, that communicates via one or more Access Networks (AN), e.g. RAN, e.g. via the first radio access node 111 to one or more core networks (CN) e.g. comprising a CN node 130, for example comprising an Access Management Function (AMF). It should be understood by the skilled in the art that “UE” is a non-limiting term which means any terminal, wireless communication terminal, user equipment, Machine Type Communication (MTC) device, Device to Device (D2D) terminal, or node e.g. smart phone, laptop, mobile phone, sensor, relay, mobile tablets or even a small base station communicating within a cell.

[0137] The wireless communications network 170 may further comprise an OAM node

[0138] 135. In some embodiments herein the transceiver arrangement 500 is comprised in the wireless communications device 121 or in the first radio access node 111 or in both. When using the word "comprise" or “comprising” it shall be interpreted as nonlimiting, i.e. meaning "consist at least of".

[0139] The embodiments herein are not limited to the above-described preferred embodiments. Various alternatives, modifications and equivalents may be used.

Claims

CLAIMS1 . An electronic switching circuit (200a, 200b) for switching electric signals, the electronic switching circuit (200a, 200b) comprising: a varactor (201a, 201b), a parallel inductor (202a, 202b) arranged in parallel with the varactor (201a, 201 b), and a series inductor (203a, 203b) arranged in series with the varactor (201a, 201 b), wherein a first terminal (230) of the varactor (201a, 201b) is connected to a signal ground (204a, 204b) with a control voltage of the varactor (201a, 201b) as Direct Current, DC, potential of the signal ground (204a, 204b), and a second terminal (240) of the varactor (201a, 201 b) is connected to the series inductor (203a, 203b).

2. The electronic switching circuit (200a) according to claim 1 , wherein the series inductor (203a) is further arranged in series with the parallel inductor (202a).

3. The electronic switching circuit (200a) according to claim 2, wherein an inductance Lpaof the parallel inductor (202a) is given by:where COff is an off-capacitance of the varactor (201a) and co = 2TTf where f is a circuit operating frequency of the electronic switching circuit (200a, 200b).

4. The electronic switching circuit (200a) according to any of the claims 2-3, wherein a series inductance Lsaof the series inductor (203a) is given by:where Conis an on-capacitance of the varactor (201a) and COff is the off-capacitance of the varactor (201a).

5. The electronic switching circuit (200b) according to claim 1 , wherein the parallel inductor (202b) is further arranged in parallel with the series inductor (203b).

6. The electronic switching circuit (200b) according to claim 5, wherein an inductance LSb of the series inductor (202b) is given by:where Conis an on-capacitance of the varactor (201 b) and co = 2TTf where f is a circuit operating frequency of the electronic switching circuit (200a, 200b).

7. The electronic switching circuit (200b) according to any of the claims 5-6, wherein an inductance LPb of the parallel inductor (203b) is given by:whereand where Conis the on-capacitance of the varactor (201 b) and where COff is an off- capacitance of the varactor (201b).

8. The electronic switching circuit (200a, 200b) according to any of the claims 1-7, wherein the parallel inductor (202a, 202b) and the series inductor (203a, 203b) are in series resonance with the varactor (201a, 201 b) at the operating frequency when the varactor (201a, 201 b) is set to its on-state capacitance, and in parallel resonance with the varactor (201a, 201b) when the varactor 201a, 201 b is set to its off-state capacitance.

9. The electronic switching circuit (200a, 200b) according to any of the claims 1-8, wherein the varactor (201a, 201b) is any one of: a Metal Oxide Semiconductor, MOS, device, a Micro-ElectroMechanical Systems, MEMS, device, a pn junction diode, or a Schottky diode.

10. The electronic switching circuit (200a, 200b) according to any of the claims 1-9, wherein the electric signals to be switched is in a frequency range of millimetre waves or sub-THz frequencies.11 . A wireless transceiver arrangement (300, 500) comprising the electronic switching circuit (200a, 200b) according to any of the claims 1-10.

12. The wireless transceiver arrangement (300, 500) according to claim 11, wherein the wireless transceiver arrangement (300, 500) comprises a single-pole multi-throw switch (320, 520) comprising, for each throw, the electronic switching circuit (310a, 310b, 510a, 510b) according to any of the claims 1-11 and a quarter wavelength transmission line (313a, 313b, 513a, 513b), wherein the quarter wavelength transmission line (313a, 313b, 513a, 513b) is arranged between a pole port of the single-pole multi-throw switch (320, 520) and the electronic switching circuit (510a, 510b), which electronic switching circuit (510a, 510b) is arranged between the quarter wavelength transmission line (313a, 313b 513a, 513b) and a throw port of the singlepole multi-throw switch (320, 520).

13. The wireless transceiver arrangement (300, 500) according to claim 12, further comprising an antenna (501), a transmitter port (570) connected to a transmitter throw of the single-pole multi-throw switch (320, 520), and a receiver port (560) connected to a receiver throw of the single-pole multi-throw switch (320, 520) and wherein the electronic switching circuit (510a, 510b)comprises a transmitter switching circuit (510a) and a transmitter quarter wavelength transmission line (513a) both arranged between the transmitter port (570) and the antenna (501) and a receiver switching circuit (510a)and a receiver quarter wavelength transmission line (513b) both arranged between the receiver port (560) and the antenna (501).

14. The wireless transceiver arrangement (300, 500) according to claim 12 or 13, wherein each quarter wavelength transmission line (513a, 513b) comprises lumped inductors and capacitors.

15. The wireless transceiver arrangement (300, 500) according to claim 14, wherein the lumped inductors and capacitors form one or more pi networks, preferably two or three pi networks.

16. The wireless transceiver arrangement (300, 500) according to any of the claims 12-15, wherein the single-pole multi-throw switch (320, 520) is configurable between at least two operating modes, wherein for each operating mode the varactor of one of the electronic switching circuits (510a, 510b) is configured to be in an off state and the respective varactor of the other of the electronic switching circuits (510a, 510b) is configured to be in an on state.

17. The wireless transceiver arrangement (300, 500) according to any of the claims 13-16, wherein the wireless transceiver arrangement (300, 500) is configurable between a transmitter operating mode and a receiver operating mode, wherein in the transmitter operating mode the varactor (511a) of the transmitter switching circuit (510a) is configured to be in an off state and the varactor (511 b) of the receiver switching circuit (510b) is configured to be in an on state such that the single-pole multi-throw switch (320, 520) is configured to pass the electric signals from the transmitter port (570) to the antenna (501), and wherein in the receiver operating mode the varactor (511b) of the receiver switching circuit (510b) is configured to be in an off state and the varactor (511a) of the transmitter switching circuit (510a) is configured to be in an on state, such that the single-pole multi-throw switch (320, 520) is configured to pass the electric signals from the antenna (501) to the receiver port (560).

18. A wireless transceiver node (121 , 111) comprising the wireless transceiver arrangement (300, 500) according to any of the claims 11-17.

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