Control device, motor device, drive device, electric power steering device, control method, and program

Abstract

A control device according to one aspect of the present invention controls a motor through vector control in which a current flowing through the motor is decomposed into a d-axis current, which is a current component that contributes to generation of a magnetic flux, and a q-axis current, which is a current component that contributes to generation of torque, and the d-axis current and the q-axis current are respectively controlled. The control device can execute non-interference control for compensating for at least a part of the influence of the q-axis current on the control of the d-axis current and the influence of the d-axis current on the control of the q-axis current. In the non-interference control, the control device calculates the compensation amount in consideration of the influence of the magnetic saturation generated at the motor.

Description

Control device, motor device, drive device, electric power steering device, control method, and program

[0001] The present application claims priority to U.S. Provisional Application No. 63 / 661,627, filed in the United States on June 19, 2024, the contents of which are incorporated herein by reference.

[0002] A control device for controlling a motor is known (for example, see Patent Document 1).

[0003] Patent No. 7063406

[0004] In the control device described above, it has been desired to improve the efficiency of the motor.

[0005] In view of the above circumstances, one object of the present invention is to provide a control device, a motor device, a drive device, an electric power steering device, a control method, and a program that can improve the efficiency of a motor.

[0006] One aspect of the control device of the present invention is a control device that decomposes a current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controls the motor by vector control that controls the d-axis current and the q-axis current, respectively. The control device is capable of performing decoupling control that compensates for at least a portion of the effect of the q-axis current on control of the d-axis current and the effect of the d-axis current on control of the q-axis current, and calculates a compensation amount in the decoupling control, taking into account the effect of magnetic saturation occurring in the motor.

[0007] One aspect of the control device of the present invention is a control device that decomposes a current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controls the motor by vector control that controls the d-axis current and the q-axis current, respectively. The control device estimates a temperature change in a back electromotive force constant of the motor based on a target value of a q-axis voltage calculated based on a target value of the q-axis current, a rotational angular velocity of the motor, and the q-axis current, estimates a temperature change in a torque constant of the motor based on the rotational angular velocity of the motor and the q-axis current, and estimates the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change of the back electromotive force constant and the estimated temperature change of the torque constant.

[0008] One aspect of a motor device of the present invention includes the above-described control device and the motor.

[0009] One aspect of a drive device of the present invention includes the above-described control device, the motor, and a gear mechanism connected to the motor.

[0010] One aspect of an electric power steering device of the present invention includes the above-described control device and a steering mechanism having the motor.

[0011] One aspect of the control method of the present invention is a control method that decomposes a current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controls the motor using vector control that controls the d-axis current and the q-axis current, respectively. The control method includes executing decoupling control that compensates for at least a portion of the effect of the q-axis current on control of the d-axis current and the effect of the d-axis current on control of the q-axis current, and calculating a compensation amount in the decoupling control, taking into account the effect of magnetic saturation occurring in the motor.

[0012] One aspect of the control method of the present invention is a control method that decomposes a current flowing through a motor into a d-axis current, which is a current component that contributes to generating magnetic flux, and a q-axis current, which is a current component that contributes to generating torque, and controls the motor by vector control that controls the d-axis current and the q-axis current, respectively. The control method includes the steps of estimating a temperature change of a back electromotive force constant of the motor based on a target value of a q-axis voltage calculated based on a target value of the q-axis current, a rotational angular velocity of the motor, and the q-axis current, estimating a temperature change of a torque constant of the motor based on the rotational angular velocity of the motor and the q-axis current, and estimating the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change of the back electromotive force constant and the estimated temperature change of the torque constant.

[0013] One aspect of the program of the present invention causes a computer to execute the above control method.

[0014] According to one aspect of the present invention, the efficiency of the motor can be improved.

[0015] Fig. 1 is a diagram schematically illustrating a drive device according to an embodiment. Fig. 2 is a cross-sectional view showing a rotor according to an embodiment, taken along line II-II in Fig. 1. Fig. 3 is a block diagram illustrating a control device and a motor according to an embodiment. Fig. 4 is a block diagram illustrating a motor according to an embodiment. Fig. 5 is a diagram schematically illustrating an electric power steering device according to an embodiment.

[0016] <Embodiment of Drive Device> A drive device 100 of this embodiment shown in FIG. 1 is mounted on a vehicle 1000. The vehicle 1000 on which the drive device 100 is mounted is a vehicle that uses a motor as a power source, such as a hybrid electric vehicle (HEV), a plug-in hybrid vehicle (PHV), or an electric vehicle (EV). The vehicle 1000 is a moving body that includes the drive device 100. The drive device 100 of this embodiment is used as a power source for the vehicle 1000 on which it is mounted. The drive device 100 rotates a drive shaft 73 of the vehicle 1000.

[0017] FIG. 1 illustrates an XYZ coordinate system as a three-dimensional Cartesian coordinate system. In the XYZ coordinate system, the Z-axis direction is the up-down direction. The side toward which the Z-axis arrow points (+Z side) is the upper side, and the opposite side of the Z-axis arrow (-Z side) is the lower side. The X-axis direction is a direction perpendicular to the Z-axis direction and corresponds to the front-to-rear direction of the vehicle 1000 on which the drive unit 1000 is mounted. In the following embodiments, the side toward which the X-axis arrow points (+X side) corresponds to the front side of the vehicle 1000, and the opposite side of the X-axis arrow (-X side) corresponds to the rear side of the vehicle 1000. The Y-axis direction is a direction perpendicular to both the X-axis and Z-axis directions and corresponds to the left-to-right direction of the vehicle 1000, i.e., the vehicle width direction. In this embodiment, the side toward which the Y-axis arrow points (+Y side) is the left side of the vehicle 1000, and the opposite side toward which the Y-axis arrow points (-Y side) is the right side of the vehicle 1000.

[0018] The positional relationship in the front-rear direction is not limited to the positional relationship in this embodiment, and the +X side may be the rear side of the vehicle 1000, and the -X side may be the front side of the vehicle 1000. In this case, the +Y side is the right side of the vehicle 1000, and the -Y side is the left side of the vehicle 1000. In addition, in this specification, the "parallel direction" includes a substantially parallel direction, and the "orthogonal direction" includes a substantially orthogonal direction.

[0019] The central axis J shown in FIG. 1 is a virtual axis extending in a direction intersecting the up-down direction. More specifically, the central axis J extends in the Y-axis direction, which is perpendicular to the up-down direction, i.e., in the left-right direction of the vehicle 1000. In the following description, unless otherwise specified, the direction parallel to the central axis J will be simply referred to as the "axial direction," the radial direction centered on the central axis J will be simply referred to as the "radial direction," and the circumferential direction centered on the central axis J will be simply referred to as the "circumferential direction." In the following description, the left side (+Y side) of the axial direction will be referred to as the "one axial side," and the right side (-Y side) of the axial direction will be referred to as the "other axial side." The up-down direction is, for example, the vertical direction, and the front-rear direction and left-right direction (axial direction) are, for example, horizontal directions perpendicular to the vertical direction.

[0020] As shown in FIG. 1 , the drive device 100 includes a motor 60, a gear mechanism 70, a housing 80, and a control device 10. The housing 80 accommodates the motor 60 and the gear mechanism 70 therein. The housing 80 has a motor housing 81 that accommodates the motor 60 therein, and a gear housing 82 that accommodates the gear mechanism 70 therein. In this embodiment, oil O is accommodated inside the motor housing 81 and the gear housing 82. In this embodiment, the drive device 100 includes a motor device 60a that includes the motor 60 and the control device 10. The motor device 60a has a motor housing 81 that accommodates the motor 60 therein.

[0021] The motor 60 includes a rotor 61 and a stator 62. The rotor 61 is rotatable about a central axis J. The rotor 61 includes a shaft 61a, a rotor core 61b, and a magnet 61c. The shaft 61a extends in the axial direction. The shaft 61a is generally cylindrical and centered on the central axis J. A portion of the shaft 61a, including an end portion on one axial side (+Y side), protrudes into the gear housing 82. The rotor core 61b is fixed to the outer peripheral surface of the shaft 61a. The magnet 61c is fixed to the rotor core 61b. As shown in FIG. 2 , in this embodiment, a plurality of magnets 61c are provided at intervals in the circumferential direction. The type of magnet 61c is not particularly limited. The magnet 61c may be, for example, a neodymium magnet or a ferrite magnet. The magnet 61c has, for example, a generally rectangular parallelepiped shape that is elongated in the axial direction. The magnet 61c extends, for example, from one axial end to the other axial end of the rotor core 61b.

[0022] The rotor core 61b has a plurality of magnet holding portions 61f arranged in a line in the circumferential direction. The plurality of magnet holding portions 61f are provided on the radially outer portion of the rotor core 61b. The plurality of magnet holding portions 61f are arranged at equal intervals around the circumference.

[0023] Each of the multiple magnet holding portions 61f has a pair of magnet holes 61g, 61h adjacent to each other in the circumferential direction. In this embodiment, the pair of magnet holes 61g, 61h penetrates the rotor core 61b in the axial direction. Note that the pair of magnet holes 61g, 61h may be holes having bottoms at their axial ends. When viewed in the axial direction, the pair of magnet holes 61g, 61h extend in directions that separate them in the circumferential direction from the radially inner side toward the radially outer side. When viewed in the axial direction, the pair of magnet holes 61g, 61h are arranged along a V-shape that widens in the circumferential direction toward the radially outer side.

[0024] The multiple magnets 61c include a pair of magnets 61d, 61e respectively disposed in a pair of magnet holes 61g, 61h. The pair of magnets 61d, 61e disposed in the pair of magnet holes 61g, 61h are arranged along a V-shape that widens circumferentially as it extends radially outward when viewed in the axial direction. Thus, in this embodiment, the multiple magnets 61c are disposed in the magnet holes 61g, 61h provided in the rotor core 61b. In other words, in this embodiment, the motor 60 is an interior permanent magnet motor (IPM motor). The motor 60 may also be an SPM motor (Surface Permanent Magnet Motor) in which magnets are fixed to the outer circumferential surface of the rotor core 61b.

[0025] A magnetic pole portion 61P is formed by one magnet holding portion 61f and multiple magnets 61c arranged in multiple magnet holes 61g, 61h provided in the single magnet holding portion 61f. The magnetic pole portions 61P are arranged at equal intervals around the circumference. In other words, the rotor 61 has multiple magnetic pole portions 61P arranged at intervals around the circumference. In this embodiment, eight magnetic pole portions 61P are provided. The multiple magnetic pole portions 61P include multiple magnetic pole portions 61N with north poles on the outer peripheral surface of the rotor core 61b and multiple magnetic pole portions 61S with south poles on the outer peripheral surface of the rotor core 61b. In this embodiment, four magnetic pole portions 61N and four magnetic pole portions 61S are provided. The four magnetic pole portions 61N and four magnetic pole portions 61S are arranged alternately in the circumferential direction. The magnetic pole portions 61P have the same configuration except that they have different magnetic poles on the outer circumferential surface of the rotor core 61b and are positioned differently in the circumferential direction.

[0026] In the magnetic pole portion 61P, the magnet hole 61g and the magnet hole 61h are arranged on either side of the magnetic pole center line AXd in the circumferential direction. The magnetic pole center line AXd is an imaginary line that passes through the circumferential center between the pair of magnet holes 61g, 61h and extends radially. The magnetic pole center line AXd is an imaginary line that passes through the circumferential center of the magnetic pole portion 61P. The circumferential center of the magnetic pole portion 61P is the circumferential center of the magnet holding portion 61f. A magnetic pole center line AXd is provided for each magnetic pole portion 61P. When viewed in the axial direction, the magnetic pole center line AXd passes through the d-axis of the rotor 61. The direction in which the magnetic pole center line AXd extends is the d-axis direction of the rotor 61. When viewed in the axial direction, the magnet hole 61g and the magnet hole 61h are arranged symmetrically with respect to the magnetic pole center line AXd as an axis of symmetry.

[0027] An inter-pole centerline AXq is shown in Figure 2. When viewed in the axial direction, the inter-pole centerline AXq is a virtual line that passes through the circumferential center between circumferentially adjacent magnet holding portions 61f and extends radially. When viewed in the axial direction, the inter-pole centerline AXq passes through the q-axis of the rotor 61. The direction in which the inter-pole centerline AXq extends is the q-axis direction of the rotor 61. The inter-pole centerline AXq is provided between each pair of magnet holding portions 61f. The extending directions of the magnetic pole centerline AXd and the inter-pole centerline AXq intersect with each other. The magnetic pole centerline AXd and the inter-pole centerline AXq are provided alternately in the circumferential direction.

[0028] As shown in Fig. 1, the stator 62 faces the rotor 61 with a gap therebetween. In this embodiment, the stator 62 is located radially outward of the rotor 61. The stator 62 has a stator core 62a and a plurality of coils 62b. The plurality of coils 62b are attached to the stator core 62a. Although not shown, the plurality of coils 62b are attached to the stator core 62a via, for example, insulators.

[0029] The gear mechanism 70 is connected to the motor 60. The gear mechanism 70 transmits rotation of the motor 60 to a drive shaft 73 of the vehicle 1000. The gear mechanism 70 includes a reduction gear 71 and a differential gear 72. The reduction gear 71 is connected to the rotor 61 of the motor 60. More specifically, the reduction gear 71 is connected to a portion of the shaft 61a of the rotor 61 that is located inside the gear housing 82. The reduction gear 71 may be connected to the rotor 61 via a gear shaft that is coupled to an end of the shaft 61a on one axial side (+Y side). In this case, the end of the shaft 61a on one axial side does not need to be located inside the gear housing 82. The differential gear 72 is connected to the reduction gear 71. A drive shaft 73 extending in the axial direction (Y-axis direction) is connected to the differential gear 72. A pair of drive shafts 73 is provided. A pair of tires 74A, 74B are respectively coupled to the pair of drive shafts 73.

[0030] When the rotation of the rotor 61 in the motor 60 is transmitted to the differential device 72 via the reduction gear device 71, a pair of drive shafts 73 connected to the differential device 72 rotates. In this way, the drive device 100 rotates the drive shafts 73 to which tires 74A, 74B of the vehicle 1000 are connected. The rotation of the pair of drive shafts 73 rotates the pair of tires 74A, 74B, causing the vehicle 1000 to travel.

[0031] The control device 10 controls the motor 60 by vector control. The vector control controls the current flowing through the motor 60 by dividing the current component that contributes to the generation of magnetic flux, i ｄ and the q-axis current i, which is a current component that contributes to the generation of torque T. ｑ and the d-axis current i ｄ and q-axis current i ｑ The control method of this embodiment is a control method performed by the control device 10, and is a control method for controlling the motor 60 by vector control. The control device 10 controls the drive device 100 by supplying power to the motor 60. The control device 10 is located, for example, on the upper side of the housing 80.

[0032] The control device 10 is a computer that executes the method for controlling the motor 60 of this embodiment. A program that causes the control device 10, which is a computer, to execute the method for controlling the motor 60 of this embodiment is installed in the control device 10. The control device 10 has a control unit 10a and a storage unit 10b. The control unit 10a is, for example, a processor such as a CPU (Central Processing Unit). At least some of the functions of each component of the control device 10 are realized, for example, by the control unit 10a executing a program, i.e., software, stored in the storage unit 10b. The control unit 10a may be, for example, a part configured by multiple processors.

[0033] At least some of the functions of the components of the control device 10 may be implemented by hardware including circuitry such as a large-scale integration (LSI), an application-specific integrated circuit (ASIC), a field-programmable gate array (FPGA), and a graphics processing unit (GPU), or may be implemented by a combination of software and hardware. The storage unit 10b, which stores a program that causes the control device 10, which is a computer, to execute the control method for the motor 60 of this embodiment, is implemented by a storage medium such as a random access memory (RAM), a read-only memory (ROM), a hard disk drive (HDD), or a flash memory. The storage unit 10b is not particularly limited as long as it can store a program that causes the computer to execute the control method of this embodiment, and may be a microcomputer or a disk medium such as a CD-ROM. The storage unit 10b may be provided separately from the control device 10. In this case, the control device 10 may communicate with the storage unit 10b via wired or wireless communication and execute the program stored in the storage unit 10b.

[0034] As shown in FIG. 3 , the control device 10 has an inverter circuit 16 that supplies power to the motor 60. Although not shown, the inverter circuit 16 has a plurality of switching elements. The switching elements are transistors such as IGBTs (Insulated Gate Bipolar Transistors) and MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors). A three-phase AC voltage is applied from the inverter circuit 16 to the plurality of coils 62 b of the stator 62. This causes a three-phase AC current to flow through the plurality of coils 62 b, causing the rotor 61 to rotate. The motor 60 has a mechanical angular velocity ω of the motor 60 relative to the voltage input from the inverter circuit 16. ｍ The mechanical angular velocity ω of the motor 60 can be expressed as a function block that outputs ｍis the mechanical angular velocity of the rotor 61. ｍ is the rotational angular velocity of the motor 60 in mechanical angle.

[0035] The motor 60 is shown, for example, in a functional block diagram such as that shown in Fig. 4. The motor 60 has an electric circuit unit 63. The electric circuit unit 63 is a circuit portion through which current flows in the motor 60. The electric circuit unit 63 includes a plurality of coils 62b. A three-phase AC voltage is input to the electric circuit unit 63 from the inverter circuit 16. In Fig. 4, the voltage applied to the electric circuit unit 63 and the current flowing through the electric circuit unit 63 are shown separated into a d-axis component and a q-axis component.

[0036] In this specification, the "d-axis component of a parameter" refers to the d-axis component when a parameter is divided into a d-axis component and a q-axis component. The "q-axis component of a parameter" refers to the q-axis component when a parameter is divided into a d-axis component and a q-axis component.

[0037] The d-axis voltage v is the d-axis component of the voltage applied to the motor 60 from the inverter circuit 16. ｄ is expressed by the following equation (1): Equation (1) is the voltage equation for the d axis.

[0038] In the above formula (1), s is the Laplace transformer, and R ｄ is the d-axis component of the resistance of the electrical circuit unit 63, i.e., the d-axis resistance, and L ｄ is the d-axis component of the inductance of the electric circuit unit 63, i.e., the d-axis inductance, and L ｑ is the q-axis component of the inductance of the electric circuit unit 63, i.e., the q-axis inductance, and ω ｅ is the electrical angular velocity of the motor 60, i.e., the electrical angular velocity of the rotor 61, and i ｄ is the d-axis current, and i ｑ is the q-axis current. Electrical angular velocity ω ｅ and mechanical angular velocity ω ｍ That is, ω ｅ =P ｒ ×ω ｍ The relationship between P ｒ is the number of pole pairs of the rotor 61. Electrical angular velocity ωｅ is the rotational angular velocity of the motor 60 in electrical angle.

[0039] (R ｄ +sL ｄ ) i ｄ wo v ｄａ Then, the following equation (2) is derived: ｄａ is the input voltage V input to the electrical circuit unit 63 ｍ is the d-axis component of

[0040]

[0041] d-axis component of motor voltage v ｄａ From d-axis current i ｄ Transfer function P ｄ (s) is the transfer function of the d-axis component of the electric circuit unit 63. Transfer function P ｄ (s) is expressed, for example, by the following equation (3).

[0042] In the above formula (3), s is the Laplace transformer, and L ｄ is the d-axis inductance of the electric circuit unit 63, and R ｄ is the d-axis resistance of the electrical circuit unit 63.

[0043] The q-axis voltage v is the q-axis component of the voltage applied to the motor 60 from the inverter circuit 16. ｑ is expressed by the following equation (4): Equation (4) is the voltage equation for the q axis.

[0044] In the above formula (4), s is the Laplace transformer, and R ｑ is the q-axis component of the resistance of the electrical circuit unit 63, i.e., the q-axis resistance, and L ｄ is the d-axis inductance of the electric circuit unit 63, and L ｑ is the q-axis inductance of the electrical circuit unit 63, and ω ｅ is the electrical angular velocity of the motor 60, and i ｄ is the d-axis current, and i ｑ is the q-axis current, and Ψ is the flux linkage of the magnet 61c. The flux linkage Ψ is the magnetic flux emitted from the magnet 61c and linking with the coil 62b. The value of the flux linkage Ψ is an effective value. The value of the flux linkage Ψ is determined by the back electromotive force constant Kｅ is equal to the value of ω ｅ Ψ is K ｅ ω ｅ In addition, in FIG. ｅ Ψ to K ｅ ω ｅ The back electromotive force constant K ｅ and electrical angular velocity ω ｅ And the back electromotive force E is E = K ｅ ω ｅ Satisfy the relationship.

[0045] (R ｑ +sL ｑ ) i ｑ wo v ｑａ Then, the following equation (5) is derived: ｑａ is the input voltage V input to the electrical circuit unit 63 ｍ is the q-axis component of

[0046]

[0047] Input voltage V ｍ The q-axis component of v ｑａ From q-axis current i ｑ Transfer function P ｑ (s) is the transfer function of the q-axis component of the electric circuit unit 63. ｑ (s) is expressed, for example, by the following equation (6).

[0048] In the above formula (6), s is the Laplace transformer, and L ｑ is the q-axis component of the inductance of the electrical circuit unit 63, and R ｑ is the q-axis resistance of the electrical circuit section 63.

[0049] The current flowing through the electric circuit unit 63 generates a torque T about the central axis J in the rotor 61, causing the rotor 61 to rotate. ｄ and q-axis current i ｑ The torque converter 64 is shown as a functional block that receives the d-axis current i as input and outputs the torque T. The torque T of the motor 60 is calculated by ｄ and q-axis current i ｑ Using the above, it is expressed by the following equation (7).

[0050] In the above formula (7), P ｒ is the number of pole pairs of the rotor 61, and L ｄ is the d-axis inductance of the electric circuit unit 63, and L ｑ is the q-axis inductance of the electrical circuit unit 63, and i ｄ is the d-axis current, and i ｑ is the q-axis current, and Ψ is the flux linkage of the magnet 61c.

[0051] Torque T is the disturbance torque T ｄ is applied to the rotor 61. In FIG. 4, the rotor 61 is subjected to a disturbance torque T ｄ The torque T to which the force is applied is input, and the mechanical angular velocity ω ｍ The disturbance torque T ｄ includes disturbances applied to the motor 60 from the gear mechanism 70, and torques transmitted to the motor 60 from the tires 74A, 74B of the running vehicle 1000. ｄ The frequency of the rotor 61 is, for example, about 10 to 100 Hz. The transfer function M(s) of the rotor 61 is expressed by, for example, the following equation (8).

[0052] In the above formula (8), s is the Laplace transformer, and J ｒ is the moment of inertia of the rotor 61, and B is the viscous friction coefficient of the rotor 61.

[0053] As shown in FIG. 3, the control device 10 receives a torque command value T ｒ is input to the host device 10. The host device is, for example, an ECU (Electronic Control Unit) mounted on the vehicle 1000. The ECU controls each part of the vehicle 1000. The control device 10 has a current conversion unit 11, PI control units 12d and 12q, a three-phase conversion unit 15, an inverter circuit 16, subtractors 40d, 40q, 41d, and 41q, and adders 42d, 42q, and 43q.

[0054] Torque command value T ｒ is the d-axis current command value i ｄｒ and the q-axis current command value i ｑｒ The d-axis current command value iｄｒ is the d-axis current i ｄ The q-axis current command value i ｑｒ is the q-axis current i ｑ The current converter 11 receives the mechanical angular velocity ω of the rotor 61. ｍ is input. Mechanical angular velocity ω ｍ is detected, for example, based on a rotation sensor (not shown). ｒ and mechanical angular velocity ω ｍ Based on this, the d-axis current command value i ｄｒ and the q-axis current command value i ｑｒ and is output.

[0055] The d-axis current command value i output from the current conversion unit 11 ｄｒ is input to the subtractor 40d. The subtractor 40d subtracts the d-axis current command value i ｄｒ From the actual current, the d-axis current i ｄ The q-axis current command value i output from the current converter 11 is subtracted from the q-axis current command value i ｑｒ is input to the subtractor 40q. The subtractor 40q subtracts the q-axis current command value i ｑｒ From the actual current, the q-axis current i ｑ The d-axis current i is subtracted from the d-axis current i and output to the PI control unit 12q. ｄ and q-axis current i ｑ The d-axis current i is calculated based on the detected current, which is detected by a current sensor (not shown) and flows through the electric circuit unit 63 of the motor 60. ｄ and q-axis current i ｑ Since the current also flows in the inverter circuit 16, the current flowing in the inverter circuit 16 is detected by a current sensor (not shown), and the d-axis current i ｄ and q-axis current i ｑ may be calculated.

[0056] The PI control unit 12d calculates the d-axis current i ｄ PI control is performed in which the d-axis voltage command value v ｄｒ The transfer function of the PI control unit 12d is, for example, K ｐｄ +K ｉｄ / s, where s is the Laplace transform and Kｐｄ is the proportional gain, and K ｉｄ is the integral gain. The d-axis voltage command value v output from the PI control unit 12d ｄｒ is input to the subtractor 41d. The subtractor 41d subtracts the d-axis voltage command value v ｄｒ The d-axis correction voltage value v output from the d-axis model following control unit 20d (described later) is ｄｆ is subtracted, and the value after subtraction is the d-axis voltage command value v ｄｒａ The adder 42d outputs the d-axis voltage command value v ｄｒａ , the d-axis voltage compensation amount v output from the decoupling control unit 50 (to be described later) ｄｃ The value after addition is used as the d-axis voltage command value v ｄｒｂ In this embodiment, the d-axis voltage command value v ｄｒａ is the d-axis current i ｄ The d-axis current command value i ｄｒ The d-axis voltage v calculated based on ｄ and the d-axis voltage compensation amount v ｄｃ is the target value to be added.

[0057] The PI control unit 12q controls the q-axis current i ｑ PI control is performed in which the q-axis voltage command value v ｑｒ The transfer function of the PI control unit 12q is, for example, K ｐｑ +K ｉｑ / s, where s is the Laplace transform and K ｐｑ is the proportional gain, and K ｉｑ is the integral gain. The q-axis voltage command value v output from the PI control unit 12q ｑｒ is input to the subtractor 41q. The subtractor 41q subtracts the q-axis voltage command value v ｑｒ The q-axis corrected voltage value v output from the q-axis model following control unit 20q (described later) is ｑｆ is subtracted, and the value after subtraction is the q-axis voltage command value v ｑｒａ The adder 42q outputs the q-axis voltage command value v ｑｒａ , the q-axis voltage compensation amount v output from the non-interference control unit 50 (to be described later) ｑｃIn this embodiment, the q-axis voltage command value v ｑｒａ is the q-axis current i ｑ The q-axis current command value i ｑｒ The q-axis voltage v is calculated based on ｑ and the q-axis voltage compensation amount v ｑｃ is the target value to be added.

[0058] The adder 43q adds K to the value input from the adder 42q. ｅ０ ×ω ｅ The counter electromotive force correction value expressed by the following formula is added to the electrical angular velocity ω, and the value after the addition is output to the three-phase conversion unit 15. ｅ is detected based on a rotation sensor (not shown). ｅ０ is the temperature t of the magnet 61c ｍ is the reference temperature t ｍ０ is the back electromotive force constant when ｅ０ The value of the temperature t ｍ is the reference temperature t ｍ０ is equal to the value of the interlinkage flux Ψ when the reference temperature t ｍ０ is not particularly limited. ｍ０ is, for example, a value between 10° C. and 30° C. The value input from the adder 43q to the three-phase conversion unit 15 is the q-axis voltage command value v ｑｒｂ The temperature t of the magnet 61c ｍ When the magnetic flux linkage Ψ of the magnet 61c changes, the magnetic flux linkage Ψ changes. ｅ Since it is equal to ｅ is the temperature t of the magnet 61c ｍ changes as the

[0059] The three-phase conversion unit 15 converts the d-axis voltage command value v ｄｒａ and the q-axis voltage command value v ｑｒａ into three-phase voltage command values ​​of U-phase, V-phase, and W-phase. The inverter circuit 16 applies voltages to the plurality of coils 62b of the stator 62 based on the three-phase voltage command values.

[0060] The control device 10 includes a model following control unit 20f. The model following control unit 20f is a controller that performs model following control by constraining the transfer function of the electric circuit unit 63, including the coil 62b of the motor 60, to a nominal model based on the electric circuit unit 63. In other words, the control method of this embodiment includes model following control. In this embodiment, the model following control unit 20f controls the d-axis current i ｄ and a d-axis model following control unit 20d that performs model following control based on the q-axis current i ｑ and a q-axis model following control unit 20q that performs model following control based on

[0061] In this embodiment, the d-axis model following control unit 20d calculates the transfer function P ｄ The d-axis model-following control unit 20d is a controller that performs model-following control to constrain the d-axis voltage command value v(s) to a nominal model based on the d-axis component of the electric circuit unit 63. In the following description, the nominal model based on the d-axis component of the electric circuit unit 63 may be referred to as the "d-axis nominal model." The d-axis model-following control unit 20d constrains the d-axis voltage command value v(s) to a nominal model based on the d-axis component of the electric circuit unit 63. ｄｒ The d-axis correction voltage value v ｄｆ The current I flowing through the electrical circuit section 63 ｍ The d-axis component of the current i ｄ and the d-axis nominal model. ｄｆ is the input voltage V input to the electrical circuit unit 63 ｍ is the correction voltage value that corrects the target value of the input voltage V ｍ The target value of is the d-axis voltage command value v ｄｒ and the q-axis voltage command value v ｑｒ voltage command value V ｒ The d-axis model following control unit 20d includes a d-axis inverse nominal model 21d, a filter 22d, and a subtractor 23d. In this embodiment, the filter 22d is a low-pass filter. The cutoff frequency of the filter 22d is, for example, 0.1 Hz or higher and 2 kHz or lower.

[0062] The d-axis inverse nominal model 21d is an inverse model of the d-axis nominal model. The transfer function P ｄｎ (s) and the transfer function P of the d-axis inverse nominal model 21d ｄｎ －１ (s) are expressed by the following equations (9) and (10), respectively.

[0063] In the above equations (9) and (10), s is the Laplace transformer, and L ｄｎ is the d-axis inductance L of the electrical circuit unit 63 ｄ is a parameter corresponding to R ｄｎ is the d-axis resistance R of the electrical circuit unit 63 ｄ This is the parameter equivalent to

[0064] In this embodiment, the transfer function P ｄｎ (s) is the transfer function P ｄ (s), the d-axis inductance L ｄ is the parameter L ｄｎ and the d-axis resistance R ｄ is the parameter R ｄｎ The transfer function P of the d-axis nominal model is expressed as ｄｎ (s) and the transfer function P of the d-axis inverse nominal model 21d ｄｎ －１ (s) is not limited to the examples shown in formulas (9) and (10), and is not particularly limited.

[0065] The d-axis inverse nominal model 21d includes a d-axis current i ｄ The d-axis inverse nominal model 21d is obtained by using the above equation (10) and the input d-axis current i ｄ Based on this, the voltage value v ｄ１ The voltage value v is output. ｄ１ is equal to the value of the voltage input to the d-axis nominal model when the output value of the d-axis nominal model is the same as the output value of the d-axis component of the electric circuit section 63.

[0066] The subtractor 23d subtracts the d-axis voltage command value v from the output of the d-axis inverse nominal model 21d. ｄｒａ Subtract the difference voltage value v ｄ２The d-axis voltage command value v ｄｒａ is the voltage command value from the output of the subtractor 41d to the input of the adder 42d. In other words, the subtractor 23d calculates the d-axis corrected voltage value v ｄｆ is fed back and before the correction by the non-interference control unit 50 is performed. ｄｒａ , the voltage value v ｄ１ Subtract the differential voltage value v ｄ２ The differential voltage value v ｄ２ includes a d-axis component of the disturbance voltage. The d-axis component of the disturbance voltage includes, for example, a d-axis component of noise caused by a dead time provided when the switching elements of the inverter circuit 16 are switched. The frequency of the d-axis component of the disturbance voltage is, for example, about several hundred Hz to several thousand Hz.

[0067] The differential voltage value v output from the subtractor 23d ｄ２ is input to the filter 22d and subjected to low-pass filtering. ｄ２ is the d-axis correction voltage value v ｄｆ and is input to the subtractor 41d. The cutoff frequency of the filter 22d is higher than the frequency of the d-axis component of the disturbance voltage. Therefore, the frequency of the d-axis component of the disturbance voltage passes through the filter 22d with almost no attenuation. As a result, the d-axis corrected voltage value v ｄｆ can be treated as an estimate of the d-axis component of the disturbance voltage. The cutoff frequency of the filter 22d is determined appropriately depending on the frequency band of the disturbance voltage to be estimated.

[0068] The d-axis model following control unit 20d calculates the modeling error Δ between the d-axis component of the electric circuit unit 63 and the d-axis nominal model. ｄ (s) complementary sensitivity function T ｃｄ The complementary sensitivity gain GT is the gain in the gain characteristic of (s) ｄ In the frequency band where is approximately 1, the transfer function P ｄ (s) is the transfer function P of the d-axis nominal model ｄｎ In other words, the model following control is configured to be constrained by the complementary sensitivity gain GTｄ In the frequency band where is approximately 1, the transfer function P ｄ (s) is the transfer function P of the d-axis nominal model ｄｎ (s) "Complementary sensitivity gain GT ｄ is approximately 1" means that the complementary sensitivity gain GT ｄ In addition to the case where the complementary sensitivity gain GT ｄ This also includes cases where the ratio is 0.8 or more and 1.2 or less.

[0069] Complementary sensitivity function T ｃｄ (s) is the complementary sensitivity function of the inner loop formed by the d-axis model following control unit 20d. ｃｄ Complementary sensitivity gain TG in (s) ｄ is the complementary sensitivity function T as a transfer function ｃｄ (s) and the complementary sensitivity function T ｃｄ The absolute value of (s) is the complementary sensitivity function T ｃｄ (s) is, for example, a gain of 0 [dB] in a frequency band below the cutoff frequency of the filter 22d, that is, a complementary sensitivity gain GT ｄ becomes 1. The transfer function P ｄ (s) is the transfer function P of the d-axis nominal model ｄｎ "Constrained by (s)" means that, for example, when looking at the input / output relationship, the transfer function P ｄ (s) is apparently the transfer function P of the d-axis nominal model ｄｎ This means that the electric circuit section 63 is controlled as shown in (s).

[0070] In this embodiment, the q-axis model following control unit 20q is a transfer function P ｑ The q-axis model-following control unit 20q is a controller that performs model-following control to constrain the q-axis voltage command value v (s) to a nominal model based on the q-axis component of the electric circuit unit 63. In the following description, the nominal model based on the q-axis component of the electric circuit unit 63 may be referred to as the "q-axis nominal model." The q-axis model-following control unit 20q constrains the q-axis voltage command value v ｑｒ The q-axis correction voltage value v ｑｆThe current I flowing through the electrical circuit section 63 ｍ The q-axis component of the ｑ and the q-axis nominal model. ｑｆ is the input voltage V input to the electrical circuit unit 63 ｍ The q-axis model following control unit 20q includes a q-axis inverse nominal model 21q, a filter 22q, and a subtractor 23q. In this embodiment, the filter 22q is a low-pass filter. The cutoff frequency of the filter 22q is, for example, 0.1 Hz or higher and 2 kHz or lower.

[0071] The q-axis inverse nominal model 21q is an inverse model of the q-axis nominal model. The transfer function P ｑｎ (s) and the transfer function P of the q-axis inverse nominal model 21q ｑｎ －１ (s) are expressed by the following equations (11) and (12), respectively.

[0072] In the above equations (11) and (12), s is the Laplace transformer, and L ｑｎ is the q-axis inductance L of the electrical circuit unit 63 ｑ is a parameter corresponding to R ｑｎ is the q-axis resistance R of the electrical circuit unit 63 ｑ This is the parameter equivalent to

[0073] In this embodiment, the transfer function P ｑｎ (s) is the transfer function P ｑ (s), the q-axis inductance L ｑ is the parameter L ｑｎ and the q-axis resistance R ｑ is the parameter R ｑｎ The transfer function P of the q-axis nominal model is ｑｎ (s) and the transfer function P of the q-axis inverse nominal model 21q ｑｎ －１ (s) is not limited to the examples shown in formulas (11) and (12), and is not particularly limited.

[0074] The q-axis inverse nominal model 21q includes the q-axis current i ｑ The q-axis inverse nominal model 21q is calculated by the above equation (12) and the input q-axis current i ｑ Based on this, the voltage value v ｑ１ The voltage value v is output. ｑ１ is equal to the value of the voltage input to the q-axis nominal model when the output value of the q-axis nominal model is the same as the output value of the q-axis component of the electric circuit section 63.

[0075] The subtractor 23q subtracts the q-axis voltage command value v from the output of the q-axis inverse nominal model 21q. ｑｒａ Subtract the difference voltage value v ｑ２ The q-axis voltage command value v ｑｒａ is the voltage command value from the output of the subtractor 41q to the input of the adder 42q. In other words, the subtractor 23q calculates the q-axis corrected voltage value v ｑｆ After the feedback, and the correction by the non-interference control unit 50 and the back electromotive force correction value K ｅ０ ω ｅ The q-axis voltage command value v before addition ｑｒａ , the voltage value v ｑ１ Subtract the differential voltage value v ｑ２ The differential voltage value v ｑ２ is the difference ΔK in the back electromotive force caused by the temperature change of the magnet 61c ｅ ω ｅ and the q-axis component of the disturbance voltage. ｅ is a value indicating the temperature change of the back electromotive force constant. ｅ is the temperature t of the magnet 61c ｍ is the reference temperature t ０ When the back electromotive force constant K ｅ０ Current back electromotive force constant K ｅ The q-axis component of the disturbance voltage includes, for example, the q-axis component of noise caused by a dead time provided when the switching elements of the inverter circuit 16 are switched. The frequency of the q-axis component of the disturbance voltage is, for example, about several hundred Hz to several thousand Hz.

[0076] The differential voltage value v output from the subtractor 23q ｑ２is input to the filter 22q and subjected to low-pass filtering. ｑ２ is the q-axis correction voltage value v ｑｆ and is input to the subtractor 41q. The cutoff frequency of the filter 22q is higher than the frequency of the q-axis component of the disturbance voltage. Therefore, the frequency of the q-axis component of the disturbance voltage passes through the filter 22q with almost no attenuation. As a result, the q-axis corrected voltage value v ｑｆ can be treated as an estimate of the q-axis component of the disturbance voltage. The cutoff frequency of the filter 22q is determined appropriately depending on the frequency band of the disturbance voltage to be estimated.

[0077] The q-axis model following control unit 20q calculates the modeling error Δ between the q-axis component of the electric circuit unit 63 and the q-axis nominal model. ｑ (s) complementary sensitivity function T ｃｑ The complementary sensitivity gain GT is the gain in the gain characteristic of (s) ｑ In the frequency band where is approximately 1, the transfer function P ｑ (s) is the transfer function P of the q-axis nominal model ｑｎ In other words, the model following control is configured to be constrained by the complementary sensitivity gain GT ｑ In the frequency band where is approximately 1, the transfer function P ｑ (s) is the transfer function P of the q-axis nominal model ｑｎ (s) "Complementary sensitivity gain GT ｑ is approximately 1" means that the complementary sensitivity gain GT ｑ In addition to the case where the complementary sensitivity gain GT ｑ This also includes cases where the ratio is 0.8 or more and 1.2 or less.

[0078] Complementary sensitivity function T ｃｑ (s) is the complementary sensitivity function of the inner loop formed by the q-axis model following control unit 20q. ｃｑ Complementary sensitivity gain TG in (s) ｑ is the complementary sensitivity function T as a transfer function ｃｑ (s) and the complementary sensitivity function T ｃｑThe absolute value of (s) is the complementary sensitivity function T ｃｑ (s) is the gain of 0 [dB] in the frequency band below the cutoff frequency of the filter 22q, that is, the complementary sensitivity gain GT ｑ becomes 1. The transfer function P ｑ (s) is the transfer function P of the q-axis nominal model ｑｎ "Constrained by (s)" means that, for example, when looking at the input / output relationship, the transfer function P ｑ (s) is apparently the transfer function P of the q-axis nominal model ｑｎ This means that the electric circuit section 63 is controlled as shown in (s).

[0079] As described above, the model following control unit 20f calculates the transfer function P ｄ (s) and the transfer function P of the q-axis component of the electric circuit unit 63 ｑ (s) are constrained to each nominal model. In this way, the model following control unit 20f calculates the complementary sensitivity function T ｃ The electric circuit unit 63 is configured so that the transfer function of the electric circuit unit 63 is constrained to the transfer function of the nominal model based on the electric circuit unit 63 in a frequency band where the gain in the gain characteristic of (s), i.e., the complementary sensitivity gain GT, is approximately 1. In other words, the model following control involves constraining the transfer function of the electric circuit unit 63 to the transfer function of the nominal model based on the electric circuit unit 63 in a frequency band where the complementary sensitivity gain GT of the complementary sensitivity function for the modeling error Δ(s) between the electric circuit unit 63 and the nominal model based on the electric circuit unit 63 is approximately 1. The phrase "the complementary sensitivity gain GT is approximately 1" not only includes the case where the complementary sensitivity gain GT is 1, but also includes, for example, the case where the complementary sensitivity gain GT is 0.8 or more and 1.2 or less.

[0080] The transfer function of the electric circuit unit 63 is, for example, the d-axis inductance L ｄis replaced with the inductance of the electrical circuit unit 63, and the d-axis resistance R ｄ is replaced with the resistance of the electric circuit unit 63. The transfer function of the electric circuit unit 63 is not limited to this formula, and is not particularly limited. The transfer function of the nominal model based on the electric circuit unit 63 is expressed by, for example, the parameter L in the above-mentioned formula (9): ｄｎ is replaced with a parameter corresponding to the inductance of the electrical circuit unit 63, and the parameter R ｄｎ is replaced with a parameter corresponding to the resistance of the electric circuit section 63. The transfer function of the nominal model based on the electric circuit section 63 is not limited to this equation and is not particularly limited.

[0081] By the model following control as described above, the model following control unit 20f controls the input voltage V ｍ The correction voltage value for correcting the target value of the d-axis correction voltage value v ｄｆ and the q-axis correction voltage value v ｑｆ The current I flowing through the electric circuit unit 63 ｍ and a nominal model based on the electric circuit unit 63. In other words, the model following control generates the d-axis correction voltage value v ｄｆ and the q-axis correction voltage value v ｑｆ The current I flowing through the electric circuit unit 63 ｍ and a nominal model based on the electric circuit section 63.

[0082] The control device 10 includes a decoupling control unit 50. The control device 10 can perform decoupling control by using the decoupling control unit 50. That is, the control method by the control device 10 includes performing decoupling control. The decoupling control is a method for controlling the d-axis current i ｄ The q-axis current i ｑ and the q-axis current i ｑ The d-axis current i ｄ The non-interference control unit 50 controls the compensation amount v ｃ The compensation amount v ｃ is the d-axis voltage compensation amount v ｄｃ and the q-axis voltage compensation amount v ｑｃ The d-axis voltage compensation amount vｄｃ is calculated by the adder 42d as the d-axis voltage command value v ｄｒａ The q-axis voltage compensation amount v ｑｃ is calculated by the adder 42q as the q-axis voltage command value v ｑｒａ That is, the q-axis voltage compensation amount v ｑｃ is the q-axis current i ｑ The target value (q-axis current command value i ｑｒ ) The q-axis voltage v ｑ The target value (q-axis voltage command value v ｑｒａ ) is added to

[0083] The non-interference control unit 50 calculates the compensation amount v in consideration of the influence of magnetic saturation occurring in the motor 60. ｃ That is, the control device 10 calculates the compensation amount v in consideration of the influence of magnetic saturation occurring in the motor 60 during non-interference control. ｃ In other words, the control method by the control device 10 calculates the compensation amount v in consideration of the influence of magnetic saturation occurring in the motor 60 during non-interference control. ｃ This calculation includes calculating the magnetic flux linkage Ψ. When the current flowing through the motor 60 becomes larger than a certain value determined for each motor 60, magnetic saturation occurs. When magnetic saturation occurs, the relationship between the interlinkage magnetic flux Ψ and the current becomes nonlinear, and the inductance of the motor 60 also changes according to the current. The magnetic saturation occurring in the motor 60 affects the voltage applied to the electric circuit unit 63 of the motor 60. More specifically, when magnetic saturation occurs in the motor 60, the input voltage V ｍ The d-axis component of v ｄａ is the q-axis current i due to magnetic saturation ｑ The input voltage V ｍ The q-axis component of v ｑａ is the d-axis current i due to magnetic saturation ｄ Therefore, when magnetic saturation occurs in the motor 60, the input voltage V ｍ The d-axis component of v ｄａ and the q-axis component v ｑａis a different value from the values ​​expressed by the above-mentioned equations (2) and (5). Therefore, in the motor 60 where magnetic saturation occurs, it is not possible to achieve sufficient decoupling by simply performing decoupling control using a general voltage equation, and there is a risk that the efficiency of the motor 60 will decrease. In contrast, according to this embodiment, the control device 10 determines the compensation amount v in decoupling control by taking into account the influence of magnetic saturation that occurs in the motor 60. ｃ Therefore, the precision of the non-interference control can be improved, and the precision of controlling the motor 60 can be improved. This improves the efficiency of the motor 60. Since magnetic saturation occurs more easily as the motor 60 becomes smaller, the effect of improving the efficiency of the motor 60 described above is particularly useful in small motors 60.

[0084] In this embodiment, the non-interference control unit 50 calculates the compensation amount v by taking into account the influence of cross-saturation caused by magnetic saturation in the motor 60. ｃ That is, in this embodiment, the control device 10 calculates the compensation amount v in consideration of the influence of cross saturation caused by magnetic saturation during non-interference control. ｃ In other words, the control method by the control device 10 calculates the compensation amount v in consideration of the influence of cross saturation caused by magnetic saturation in the non-interference control. ｃ The cross saturation is calculated by calculating the q-axis current i ｑ influences the q-axis interlinkage flux, and the d-axis current i ｄ Cross saturation is a phenomenon that is influenced by magnetic saturation. By performing decoupling control that takes into account the influence of cross saturation, the motor 60 can be controlled with greater precision. Therefore, the efficiency of the motor 60 can be further improved.

[0085] In this embodiment, the non-interference control unit 50 controls the rotation angle θ of the motor 60 when magnetic saturation occurs. ｅ Considering the effect of the compensation amount v ｃThat is, in this embodiment, the control device 10 calculates the rotation angle θ of the motor 60 when magnetic saturation occurs under non-interference control. ｅ Considering the effect of the compensation amount v ｃ In other words, the control method by the control device 10 calculates the rotation angle θ of the motor 60 when magnetic saturation occurs under non-interference control. ｅ Considering the effect of the compensation amount v ｃ When magnetic saturation occurs in the inductance of the electric circuit unit 63 of the motor 60, the rotation angle θ ｅ Therefore, the rotation angle θ of the motor 60 when magnetic saturation occurs is ｅ By performing non-interference control while taking into consideration the influence of the rotation angle θ of the motor 60, the motor 60 can be controlled with higher precision, and the efficiency of the motor 60 can be further improved. ｅ is the rotation angle of the rotor 61, which is an electrical angle.

[0086] Considering magnetic saturation, the q-axis component v ｑａ is expressed by the following equation (13). In the above formula (13), s is the Laplace transformer, and L ｄｑ is the q-axis current i with respect to the d-axis component of the interlinkage magnetic flux Ψ generated in the motor 60 ｑ is the static inductance as a coefficient representing the influence of ｑｄｉ is the d-axis current i for the q-axis component of the interlinkage magnetic flux Ψ ｄ is the dynamic inductance as a coefficient representing the influence of ｄａ is the d-axis current i ｄ is the amplitude of the a-th order component of ｄａ is the d-axis current i ｄ is the phase of the a-th order component of θ ｅ is the rotation angle of the motor 60. The static inductance is an inductance expressed as Ψ / I, where Ψ is the magnetic flux linkage and I is the current. The dynamic inductance is an inductance expressed as dΨ / dI, where Ψ is the magnetic flux linkage and I is the current.

[0087] The fourth, fifth, and sixth terms from the left on the right side of the above equation (13) represent the influence of magnetic saturation. The fourth and fifth terms from the left on the right side of the equation (13) represent the influence of cross saturation. The sixth term from the left on the right side of the equation (13) represents the rotation angle θ of the motor 60 when magnetic saturation occurs. ｅ This shows the influence of Σ in the sixth term from the left on the right side of equation (13). ａ is ω for multiple integers a ｅ L ｄａ sin(aθ ｅ +b ｄａ ) i ｄ The integers a include numbers indicating multiple orders that affect the voltage of the motor 60. The integers a are determined based on the number of poles and slots of the motor 60, etc.

[0088] q-axis voltage compensation amount v ｑｃ is expressed by the following equation (14). In the above formula (14), v ｃｑ１ is the first compensation amount, and v ｃｑ２ is the second compensation amount. That is, the q-axis voltage compensation amount v ｑｃ is the first compensation amount v ｃｑ１ and the second compensation amount v ｃｑ２ The sum of the first and second terms from the left on the right side of equation (14) is equal to the q-axis voltage compensation amount in decoupling control that does not take magnetic saturation into consideration. The q-axis voltage compensation amount v ｑｃ is the q-axis voltage compensation amount in the non-interference control that does not consider magnetic saturation, and the first compensation amount v ｃｑ１ and the second compensation amount v ｃｑ２ The compensation amount is the sum of

[0089] First compensation amount v ｃｑ１ is a compensation amount that compensates for at least a part of the influence of cross-saturation. ｃｑ１ is expressed by the following equation (15). In the above formula (15), ω ｅ is the electrical angular velocity of the motor 60, and i ｑ is the q-axis current, and i ｄis the d-axis current, s is the Laplace transformer, and L ｄｑ is the q-axis current i with respect to the d-axis component of the interlinkage magnetic flux Ψ generated in the motor 60 ｑ is the static inductance as a coefficient representing the influence of ｑｄｉ is the d-axis current i for the q-axis component of the interlinkage magnetic flux Ψ ｄ is the dynamic inductance as a coefficient that represents the influence of

[0090] As shown in equation (15), the first compensation amount v ｃｑ１ is the sum of the part of equation (13) that indicates the influence of cross saturation, that is, the fourth and fifth terms from the left on the right side of equation (13), with their positive and negative signs reversed. ｃｑ１ is the q-axis voltage command value v ｑｒａ Therefore, the motor 60 can be controlled more precisely and the efficiency of the motor 60 can be improved.

[0091] In this embodiment, the non-interference control unit 50 calculates the first compensation amount v by taking into consideration the likelihood of cross-saturation occurring, which varies depending on the rotation speed N of the motor 60. ｃｑ１ , i.e., the compensation amount v ｃ That is, the control device 10 calculates the first compensation amount v in consideration of the likelihood of cross-saturation occurring, which changes due to the rotation speed N of the motor 60, during non-interference control. ｃｑ１ In other words, the control method by the control device 10 calculates the first compensation amount v in consideration of the likelihood of cross-saturation occurring, which changes due to the rotation speed N of the motor 60, during non-interference control. ｃｑ１ This allows the influence of cross-saturation to be compensated for more accurately, improving the accuracy of controlling the motor 60. This also improves the efficiency of the motor 60. The rotation speed N of the motor 60 is the rotation speed of the rotor 61. In this embodiment, the static inductance L ｄｑ is expressed by the following equation (16).

[0092]

[0093] In the above formula (16), L ｄｑｘ is the q-axis current i with respect to the d-axis component of the interlinkage magnetic flux Ψ when the rotation speed N of the motor 60 is the first rotation speed N1. ｑ is the static inductance as a coefficient representing the influence of ｄｑｙ is the q-axis current i with respect to the d-axis component of the flux linkage Ψ when the rotation speed N of the motor 60 is a second rotation speed N2 that is greater than the first rotation speed N1. ｑ is the static inductance as a coefficient representing the influence of x, and x and y are variables that satisfy the relationship x+y=1.

[0094] The rotation speed N of the motor 60 is acquired based on a rotation sensor (not shown). The first rotation speed N1 is the rotation speed N of the motor 60 at which magnetic saturation is unlikely to occur in the motor 60. In this embodiment, the first rotation speed N1 is the rotation speed N at which magnetic saturation does not occur, or almost never occurs, in the motor 60. The first rotation speed N1 is, for example, the minimum value of the rotation speed used in the motor 60. The second rotation speed N2 is the rotation speed N of the motor 60 at which magnetic saturation is likely to occur in the motor 60. In this embodiment, the second rotation speed N2 is the rotation speed N at which magnetic saturation always occurs, or almost always occurs, in the motor 60. The second rotation speed N2 is, for example, the maximum value of the rotation speed used in the motor 60.

[0095] In this embodiment, the dynamic inductance L ｑｄｉ is expressed by the following equation (17).

[0096] In the above formula (17), L ｑｄｘｉ is the d-axis current i with respect to the q-axis component of the flux linkage Ψ when the rotation speed N of the motor 60 is the first rotation speed N1. ｄ is the dynamic inductance as a coefficient representing the influence of L ｑｄｙｉ is the d-axis current i with respect to the q-axis component of the interlinkage magnetic flux Ψ when the rotation speed N of the motor 60 is the second rotation speed N2. ｄ is the dynamic inductance as a coefficient that represents the influence of

[0097] Static inductance L ｄｑｘ , L ｄｑｙ and dynamic inductance L ｑｄｘｉ , L ｑｄｙｉis the d-axis current i ｄ and the value of the q-axis current i ｑ In this embodiment, the storage unit 10b stores the static inductance L ｄｑｘ , L ｄｑｙ and dynamic inductance L ｑｄｘｉ , L ｑｄｙｉ Each data map stores a data map for each of the d-axis current i ｄ and the value of the q-axis current i ｑ The control device 10 stores the values ​​of each inductance corresponding to the value of each data map and the d-axis current i ｄ and the value of the q-axis current i ｑ Based on the value of the static inductance L ｄｑｘ , L ｄｑｙ and dynamic inductance L ｑｄｘｉ , L ｑｄｙｉ The d-axis current i ｄ and the value of the q-axis current i ｑ For the value of static inductance L ｄｑｘ , L ｄｑｙ and dynamic inductance L ｑｄｘｉ , L ｑｄｙｉ As long as each of the above can be calculated, the data stored in the storage unit 10b may be in any format.

[0098] The variable x is 1 when the rotation speed N of the motor 60 becomes the first rotation speed N1, and is 0 when the rotation speed N of the motor 60 becomes the second rotation speed N2. The variable x decreases as the rotation speed N of the motor 60 increases from the first rotation speed N1 to the second rotation speed N2. The variable y is 0 when the rotation speed N of the motor 60 becomes the first rotation speed N1, and is 1 when the rotation speed N of the motor 60 becomes the first rotation speed N1. The variable y increases as the rotation speed N of the motor 60 increases from the first rotation speed N1 to the second rotation speed N2.

[0099] Cross-saturation is more likely to occur as the rotation speed N of the motor 60 increases. Therefore, by changing the variables x and y in accordance with the rotation speed N of the motor 60 as described above, the first compensation amount vｃｑ１ This allows the influence of cross-saturation to be compensated for more accurately, and the motor 60 to be controlled more accurately. Therefore, the efficiency of the motor 60 can be improved.

[0100] Second compensation amount v ｃｑ２ is the rotation angle θ of the motor 60 ｅ The second compensation amount v ｃｑ２ is expressed by the following equation (18).

[0101] In the above formula (18), ω ｅ is the electrical angular velocity of the motor 60, and θ ｅ is the rotation angle of the motor 60, and i ｑ is the q-axis current, and i ｄ is the d-axis current, a is an integer equal to or greater than 1, and L ｄａ is the amplitude of the a-th order component of the d-axis current, and b ｄａ is the phase of the a-th order component of the d-axis current.

[0102] As shown in equation (18), the second compensation amount v ｃｑ２ is the rotation angle θ of the motor 60 when magnetic saturation occurs in equation (13). ｅ Therefore, the second compensation amount v ｃｑ２ is the q-axis voltage command value v ｑｒａ When magnetic saturation occurs, the rotation angle θ of the motor 60 is ｅ Therefore, the motor 60 can be controlled with higher precision, and the efficiency of the motor 60 can be improved.

[0103] Considering magnetic saturation, the d-axis component v ｄａ is expressed by the following equation (19). In the above formula (19), L ｑｄ is the d-axis current i with respect to the q-axis component of the interlinkage magnetic flux Ψ generated in the motor 60 ｄ is the static inductance as a coefficient representing the influence of ｄｑｉ is the q-axis current i for the d-axis component of the interlinkage magnetic flux Ψ ｑis the dynamic inductance as a coefficient representing the influence of ｑａ is the q-axis current i ｑ is the amplitude of the a-th order component of ｑａ is the q-axis current i ｑ is the phase of the a-th order component of

[0104] The third, fourth, and fifth terms from the left on the right side of the above equation (19) represent the influence of magnetic saturation. The third and fourth terms from the left on the right side of the equation (19) represent the influence of cross saturation. The fifth term from the left on the right side of the equation (19) represents the rotation angle θ of the motor 60 when magnetic saturation occurs. ｅ This shows the influence of Σ in the fifth term from the left on the right side of equation (19). ａ is ω for multiple integers a ｅ L ｑａ sin(aθ ｅ +b ｑａ ) i ｑ The multiple integers a include numbers indicating multiple orders that affect the voltage of the motor 60. Note that the multiple integers a in equation (13) and the multiple integers a in equation (19) may include the same numbers or different numbers.

[0105] d-axis voltage compensation amount v ｄｃ is expressed by the following equation (20). The first term from the left on the right side of equation (20) is equal to the d-axis voltage compensation amount in decoupling control that does not take magnetic saturation into consideration. ｄｃ is the compensation amount obtained by adding the second, third, and fourth terms from the left on the right side of equation (20) to the d-axis voltage compensation amount in decoupling control that does not consider magnetic saturation. The second, third, and fourth terms from the left on the right side of equation (20) are the values ​​obtained by inverting the positive and negative signs of the third, fourth, and fifth terms from the left on the right side of equation (19), respectively. Therefore, the d-axis voltage compensation amount v ｄｃ is the d-axis voltage command value v ｄｒａAs a result, the rotation angle θ of the motor 60 when the influence of cross saturation and magnetic saturation occurs is reduced, as in the case of the q-axis component described above. ｅ Therefore, the motor 60 can be controlled with higher precision, and the efficiency of the motor 60 can be improved.

[0106] In this embodiment, the decoupling control unit 50 calculates the d-axis voltage compensation amount v ｄｃ is also calculated taking into consideration the likelihood of cross-saturation occurring, which varies depending on the rotation speed N of the motor 60. This allows for more accurate compensation for the effects of cross-saturation, thereby further improving the efficiency of the motor 60. In this embodiment, the static inductance L ｑｄ is expressed by the following equation (21), and the dynamic inductance L ｄｑｉ is expressed by the following equation (22).

[0107]

[0108] In the above formula (21), L ｑｄｘ is the d-axis current i with respect to the q-axis component of the flux linkage Ψ when the rotation speed N of the motor 60 is the first rotation speed N1. ｄ is the static inductance as a coefficient representing the influence of ｑｄｙ is the d-axis current i with respect to the q-axis component of the flux linkage Ψ when the rotation speed N of the motor 60 is a second rotation speed N2 that is greater than the first rotation speed N1. ｄ is a static inductance as a coefficient representing the influence of the q-axis component, and x and y are variables that satisfy the relationship x+y=1. The variables x and y are the same as the q-axis component described above.

[0109] In the above formula (22), L ｄｑｘｉ is the q-axis current i with respect to the d-axis component of the interlinkage magnetic flux Ψ when the rotation speed N of the motor 60 is the first rotation speed N1. ｑ is the dynamic inductance as a coefficient representing the influence of L ｄｑｙｉ is the q-axis current i with respect to the d-axis component of the interlinkage magnetic flux Ψ when the rotation speed N of the motor 60 is the second rotation speed N2. ｑ is the dynamic inductance as a coefficient that represents the influence of

[0110] Static inductance L ｑｄｘ , L ｑｄｙ and dynamic inductance L ｄｑｘｉ , L ｄｑｙｉ is the d-axis current i ｄ and the value of the q-axis current i ｑ In this embodiment, the storage unit 10b stores the static inductance L ｑｄｘ , L ｑｄｙ and dynamic inductance L ｄｑｘｉ , L ｄｑｙｉ Each data map stores a data map for each of the d-axis current i ｄ and the value of the q-axis current i ｑ The control device 10 stores the values ​​of each inductance corresponding to the value of each data map and the d-axis current i ｄ and the value of the q-axis current i ｑ Based on the value of the static inductance L ｑｄｘ , L ｑｄｙ and dynamic inductance L ｄｑｘｉ , L ｄｑｙｉ The d-axis current i ｄ and the value of the q-axis current i ｑ For the value of static inductance L ｑｄｘ , L ｑｄｙ and dynamic inductance L ｄｑｘｉ , L ｄｑｙｉ As long as each of the above can be calculated, the data stored in the storage unit 10b may be in any format.

[0111] As described above, in this embodiment, the control device 10 includes the model following control unit 20f. The model following control unit 20f controls the input voltage V ｍ The correction voltage value for correcting the target value of the d-axis correction voltage value v ｄｆ and the q-axis correction voltage value v ｑｆ based on the current Im flowing through the electric circuit unit 63 and a nominal model based on the electric circuit unit 63. The model-following control unit 20f generates a complementary sensitivity function T ｃIn the frequency band where the gain in the gain characteristic of (s), i.e., the complementary sensitivity gain GT, is approximately 1, the transfer function of the electric circuit unit 63 is constrained by the transfer function of the nominal model based on the electric circuit unit 63. As a result, if the nominal model is set as a model having desired characteristics, the electric circuit unit 63 can be treated as a plant model having desired characteristics. For example, in this embodiment, the d-axis corrected voltage value v ｄｆ and the q-axis correction voltage value v ｑｆ By calculating the current I, the disturbance voltage is compensated for, and the electric circuit unit 63 can be treated as a plant model in which no disturbance voltage occurs. ｍ to a desired waveform, thereby improving the efficiency of the motor 60. In particular, in this embodiment, as described above, the accuracy of decoupling control can be improved by taking into account the influence of magnetic saturation that occurs in the motor 60, thereby improving the accuracy of controlling the motor 60. As a result, even in a relatively high frequency band, the modeling error Δ(s) can be prevented from increasing, and the complementary sensitivity gain can be made approximately 1. Therefore, it is possible to stably constrain the electric circuit unit 63 to a desired nominal model over a wide frequency band, and it is possible to more suitably improve the controllability of the motor 60. Therefore, by combining the decoupling control of this embodiment with model-following control, the efficiency of the motor 60 can be further improved.

[0112] As shown in FIG. 3 , the control device 10 includes a temperature estimation unit 30. The temperature estimation unit 30 includes an inverse nominal model 31, a filter 32, a filter 35, a subtractor 33, a calculation unit 34, and the q-axis model following control unit 20q described above. The filters 32 and 35 are low-pass filters. The cutoff frequency of the filter 32 is, for example, 0.1 Hz or higher and 5 Hz or lower.

[0113] The filter 35 receives the q-axis corrected voltage value v output from the filter 22q. ｑｆis input. The cutoff frequency of the filter 35 is lower than the frequency of the q-axis component of the disturbance voltage and higher than the frequency of the temperature change of the magnet 61c. Therefore, the q-axis component of the disturbance voltage is attenuated by the filter 35, and the difference ΔK in the back electromotive force caused by the temperature change of the magnet 61c is ｅ ω ｅ passes through the filter 35 without or almost without attenuation. As a result, the value output from the filter 35 is the difference ΔK ｅ ω ｅ The value output from the filter 35, i.e., the difference ΔK ｅ ω ｅ is input to the calculation unit 34.

[0114] The inverse nominal model 31 is an inverse model of the nominal model based on the rotor 61. The transfer function M ｎ (s) and the transfer function M of the inverse nominal model 31 ｎ －１ (s) are expressed by the following equations (23) and (24), respectively.

[0115] In the above equations (23) and (24), s is the Laplace transformer, and J ｎｒ is a parameter corresponding to the moment of inertia of the rotor 61, and B ｎ is a parameter corresponding to the viscous friction coefficient of the rotor 61.

[0116] In this embodiment, the transfer function M of the nominal model based on the rotor 61 ｎ (s) is the moment of inertia J for the transfer function M(s). ｒ is the parameter J ｒｎ and the viscous friction coefficient B is replaced by parameter B ｎ The transfer function M of the nominal model based on the rotor 61 is expressed as ｎ (s) and the transfer function M of the inverse nominal model 31 ｎ －１ (s) is not limited to the examples shown in formulas (23) and (24), and is not particularly limited. ​

[0117] The inverse nominal model 31 includes the mechanical angular velocity ω ｍ The inverse nominal model 31 is constructed by using the above equation (24) and the input mechanical angular velocity ω ｍ Based on this, the torque value T １ The torque value T １ is equal to the torque value input to the nominal model based on the rotor 61 when the output value of the nominal model based on the rotor 61 is the same as the output value of the rotor 61. １ Torque command value T ｒ The differential torque value T ２ Generate.

[0118] Differential torque value T ２ is the torque difference ΔT caused by the temperature change of the magnet 61c ｔ and disturbance torque T ｄ where the torque difference ΔT ｔ is the torque constant K due to the temperature change of the magnet 61c. ｔ This occurs due to the change in the torque constant K ｔ and the torque T of the motor 60 and the current I flowing through the coil 62b of the motor 60. ｍ The relationship is T = K ｔ I ｍ Here, the torque constant K caused by the temperature change of the magnet 61c is ｔ The change in the torque difference ΔT occurs due to a change in the magnetic flux linkage Ψ caused by a change in the temperature of the magnet 61c. As shown in the above equations (1) and (4), in the d-axis voltage equation and the q-axis voltage equation, the magnetic flux linkage Ψ is included only in the q-axis voltage equation. ｔ is the torque constant K ｔ Temperature change ΔK ｔ and q-axis current i ｑ Using the above, ΔK ｔ i ｑ The temperature change ΔK ｔ is the temperature t of the magnet 61c ｍ is the reference temperature t ｍ０ When the torque constant K ｔ０ Current torque constant K ｔIn an IPM motor, the temperature change ΔK ｔ is the temperature change ΔΨ of the interlinkage magnetic flux Ψ and the number of pole pairs P of the motor 60 ｒ Therefore, ΔK ｔ =P ｒ It is expressed as ΔΨ. Note that the torque constant K ｔ and the back electromotive force constant K ｅ That is, K ｔ =P ｒ ×K ｅ Satisfy. P ｒ is the number of pole pairs of the rotor 61.

[0119] The differential torque value T output from the subtractor 33 ２ is input to the filter 32 and subjected to low-pass filtering. The cutoff frequency of the filter 32 is ｄ and higher than the frequency of the temperature change of the magnet 61c. ｄ is attenuated by the filter 32, and the torque difference ΔK ｔ i ｑ passes through the filter 32 unattenuated or almost unattenuated. As a result, the value output from the filter 32 is the difference ΔK ｔ i ｑ The difference ΔK is equal to or approximately equal to ｔ i ｑ is input to the calculation unit 34.

[0120] The calculation unit 34 calculates the difference ΔK ｅ ω ｅ and the difference ΔK ｔ i ｑ The calculation unit 34 calculates the difference ΔK ｅ ω ｅ Based on the back electromotive force constant K ｅ Temperature change ΔK ｅ The calculation unit 34 calculates the difference ΔK ｅ ω ｅ is the electrical angular velocity ω ｅ By dividing by the value of ｅ Calculate the temperature change ΔK ｅ is an estimated value. As mentioned above, the difference ΔK ｅ ω ｅis the q-axis voltage v ｑ The q-axis voltage command value v ｑｒａ and q-axis current i ｑ That is, the control device 10 calculates the q-axis current i ｑ The q-axis voltage command value v is calculated based on the target value of ｑｒａ , the electrical angular velocity ω, which is the rotational angular velocity of the motor 60 ｅ , and the q-axis current i ｑ Based on this, the back electromotive force constant K ｅ Temperature change ΔK ｅ In other words, the control method by the control device 10 estimates the q-axis voltage command value v ｑｒａ , the electrical angular velocity ω of the motor 60 ｅ , and the q-axis current i ｑ Based on this, the back electromotive force constant K ｅ Temperature change ΔK ｅ This includes estimating the

[0121] The calculation unit 34 calculates the difference ΔK ｔ i ｑ Based on the torque constant K ｔ Temperature change ΔK ｔ The calculation unit 34 calculates the difference ΔK ｔ i ｑ The q-axis current i ｑ By dividing by the value of ｔ Calculate the temperature change ΔK ｔ is an estimated value. As mentioned above, the difference ΔK ｔ i ｑ is the electrical angular velocity ω ｅ That is, the control device 10 calculates the electric angular velocity ω, which is the rotational angular velocity of the motor 60, in the temperature estimation unit 30. ｅ and q-axis current i ｑ Based on this, the torque constant K ｔ Temperature change ΔK ｔ In other words, the control method by the control device 10 estimates the electrical angular velocity ω of the motor 60. ｅ and q-axis current i ｑ Based on this, the torque constant K ｔ Temperature change ΔK ｔThis includes estimating the

[0122] The calculation unit 34 calculates the temperature change ΔK ｅ The calculated temperature change ΔK ｔ Based on this, the temperature t ｍ That is, the control device 10 calculates the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ The torque constant K is estimated as ｔ Temperature change ΔK ｔ Based on this, the temperature t of the magnet 61c provided in the rotor 61 of the motor 60 is calculated. ｍ In other words, the control method by the control device 10 is to estimate the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ The torque constant K is estimated as ｔ Temperature change ΔK ｔ Based on this, the temperature t ｍ This includes estimating the

[0123] In this embodiment, the calculation unit 34 calculates the back electromotive force constant K ｅ Temperature change ΔK ｅ Based on the temperature t of the magnet 61c ｍ The first estimate t ｍｅ That is, the control device 10 calculates the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ Based on the temperature t of the magnet 61c ｍ The first estimate t ｍｅ In other words, the control method by the control device 10 calculates the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ Based on the temperature t of the magnet 61c ｍ The first estimate t ｍｅ Here, the temperature t of the magnet 61c is calculated. ｍ The back electromotive force constant K changes depending on ｅ is expressed by the following equation (25).

[0124] In the above formula (25), t ｍ０ is the reference temperature, Kｅ０ is the temperature t of the magnet 61c ｍ is the reference temperature t ｍ０ is the back electromotive force constant when ｅ is a coefficient that indicates the temperature dependence of

[0125] Temperature change of the back electromotive force constant ΔK ｅ is ΔK ｅ =K ｅ -K ｅ０ Using this equation, the above equation (25) is transformed to obtain the estimated temperature t of the magnet 61c. ｍ is the first estimate t ｍｅ Then, the following equation (26) is obtained.

[0126]

[0127] Reference temperature t ｍ０ , back electromotive force constant K ｅ０ , and the coefficient α are stored in advance in the storage unit 10b, for example. The calculation unit 34 calculates the temperature change ΔK ｅ and into the above equation (26) to obtain the first estimated value t ｍｅ In this way, the control device 10 calculates the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ Based on this, the temperature t of the magnet 61c provided in the rotor 61 of the motor 60 ｍ In other words, the control method by the control device 10 is to estimate the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ Based on this, the temperature t of the magnet 61c ｍ This includes estimating the

[0128] In this embodiment, the calculation unit 34 calculates the torque constant K ｔ Temperature change ΔK ｔ The second estimated value t of the magnet 61c is calculated based on ｍｔ That is, the control device 10 calculates the estimated torque constant K ｔ Temperature change ΔK ｔ Based on the temperature t of the magnet 61c ｍ The second estimate t ｍｔIn other words, the control method by the control device 10 calculates the estimated torque constant K ｔ Temperature change ΔK ｔ Based on the temperature t of the magnet 61c ｍ The second estimate t ｍｔ Here, the temperature t of the magnet 61c is calculated. ｍ The torque constant K changes depending on ｔ is expressed by the following equation (27).

[0129] In the above formula (27), t ｍ０ is the reference temperature, K ｔ０ is the temperature t of the magnet 61c ｍ is the reference temperature t ｍ０ is the torque constant when ｔ The coefficient β is a coefficient that indicates the temperature dependency of the temperature. The coefficient β may be the same as the coefficient α, or may be a different value from the coefficient α.

[0130] Temperature change in torque constant ΔK ｔ is ΔK ｔ =K ｔ -K ｔ０ Using this equation, the above equation (27) is transformed to obtain the estimated temperature t of the magnet 61c. ｍ is the second estimate t ｍｔ Then, the following equation (28) is obtained.

[0131]

[0132] Reference temperature t ｍ０ , torque constant K ｔ０ , and the coefficient β are stored in advance in the storage unit 10b, for example. The calculation unit 34 calculates the temperature change ΔK ｔ and into the above equation (28) to obtain the second estimated value t ｍｔ In this way, the control device 10 calculates the estimated torque constant K ｔ Temperature change ΔK ｔ Based on this, the temperature t of the magnet 61c provided in the rotor 61 of the motor 60 ｍIn other words, the control method by the control device 10 is to estimate the estimated torque constant K ｔ Temperature change ΔK ｔ Based on this, the temperature t of the magnet 61c ｍ This includes estimating the

[0133] In this embodiment, the calculation unit 34 calculates the first estimated value t ｍｅ and the second estimate t ｍｔ and are different from each other, the first estimated value t ｍｅ and the second estimate t ｍｔ The higher value of the two is the temperature t ｍ The estimated value of t ｍｓ That is, in this embodiment, the control device 10 calculates the first estimated value t ｍｅ and the second estimate t ｍｔ and are different from each other, the first estimated value t ｍｅ and the second estimate t ｍｔ The higher value of the two is the temperature t ｍ The estimated value of t ｍｓ In other words, the control method by the control device 10 is to obtain the first estimated value t ｍｅ and the second estimate t ｍｔ and are different from each other, the first estimated value t ｍｅ and the second estimate t ｍｔ The higher value of the two is the temperature t ｍ The estimated value of t ｍｓ The first estimate t ｍｅ and the second estimate t ｍｔ and are the same value, the calculation unit 34 sets the same value as the estimated value t ｍｓ That is, the control device 10 calculates the first estimated value t ｍｅ and the second estimate t ｍｔ and are the same value, the first estimated value t ｍｅ and the second estimate t ｍｔ The same value as the estimated value t ｍｓ Obtain as.

[0134] The control device 10 includes a determination unit 14. The determination unit 14 determines the temperature t of the magnet 61c calculated by the temperature estimation unit 30. ｍ The estimated value of t ｍｓThe determination unit 14 receives the estimated value t ｍｓ The determining unit 14 determines whether or not it is necessary to limit the current supplied to the motor 60 based on the estimated value t ｍｓ If the estimated value t is greater than the predetermined threshold, the determining unit 14 determines that the current supplied to the motor 60 needs to be limited. ｍｓ is equal to or less than a predetermined threshold, it is determined that there is no need to limit the current supplied to the motor 60. The determination unit 14 outputs a signal indicating whether or not it is necessary to limit the current supplied to the motor 60. Based on this signal, the control device 10 limits the current supplied to the motor 60 or removes the limit.

[0135] Temperature t of magnet 61c ｍ If the temperature t of the magnet 61c rises too much, irreversible demagnetization occurs in the magnet 61c. ｍ It is necessary to limit the current supplied to the motor 60 before the temperature t of the magnet 61c becomes too high. ｍ If the estimation accuracy is low, it is necessary to lower the threshold value in the determination unit 14 to some extent and limit the current early. However, in this case, the current is more likely to be limited, and the output of the motor 60 decreases. This may require, for example, increasing the amount of magnet 61c used, which may increase the manufacturing cost of the motor 60.

[0136] In contrast to the above, according to this embodiment, the control device 10 controls the q-axis voltage v ｑ The target value (q-axis voltage command value v ｑｒａ ), and the electrical angular velocity ω, which is the rotational angular velocity of the motor 60 ｅ , and the q-axis current i ｑ Based on this, the back electromotive force constant K ｅ Temperature change ΔK ｅ The control device 10 estimates the electrical angular velocity ω, which is the rotational angular velocity of the motor 60. ｅ and q-axis current i ｑ Based on this, the torque constant K ｔ Temperature change ΔK ｔ The control device 10 estimates the estimated back electromotive force constant K ｅTemperature change ΔK ｅ The torque constant K is estimated as ｔ Temperature change ΔK ｔ Based on this, the temperature t of the magnet 61c provided in the rotor 61 of the motor 60 is calculated. ｍ Therefore, the back electromotive force constant K ｅ and torque constant K ｔ The temperature t of the magnet 61c is calculated by estimating only one of the temperature changes. ｍ Compared to estimating the temperature t of the magnet 61c, by using two estimated values ​​of the temperature change, ｍ Therefore, the threshold value in the determination unit 14 can be set close to the temperature at which irreversible demagnetization occurs in the magnet 61c, and the current can be prevented from being limited earlier than necessary. This prevents the output of the motor 60 from being reduced due to the current being limited, and improves the efficiency of the motor 60. Furthermore, it is possible to prevent an increase in the amount of magnet 61c used, and it is possible to prevent an increase in the manufacturing cost of the motor 60. Furthermore, it is possible to prevent the electrical angular velocity ω of the motor 60 from being increased. ｅ and q-axis current i ｑ Based on this, the torque constant K ｔ Temperature change ΔK ｔ Since the torque T is estimated, there is no need to provide a torque sensor that detects the torque T of the motor 60. Therefore, an increase in the number of parts of the motor device 60a can be suppressed, and an increase in the manufacturing cost of the motor device 60a can be suppressed.

[0137] In this embodiment, the control device 10 calculates the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ Based on the temperature t of the magnet 61c ｍ The first estimate t ｍｅ The control device 10 calculates the estimated torque constant K ｔ Temperature change ΔK ｔ Based on the temperature t of the magnet 61c ｍ The second estimate t ｍｔ The control device 10 calculates the first estimated value t ｍｅ and the second estimate t ｍｔ and are different from each other, the first estimated value t ｍｅ and the second estimate tｍｔ The higher value of the two is the temperature t ｍ The estimated value of t ｍｓ Therefore, the first estimated value t ｍｅ and the second estimate t ｍｔ The estimated temperature used in the determination unit 14 can be made higher than when the lower value of the two is acquired, and the actual temperature t ｍ This makes it easier to limit the current supplied to the motor 60 before the voltage Vcc becomes too high. Therefore, irreversible demagnetization of the magnet 61c can be suitably prevented.

[0138] In this embodiment, as described above, the control device 10 determines the compensation amount v in consideration of the influence of magnetic saturation occurring in the motor 60 during non-interference control. ｃ Therefore, the motor 60 can be controlled with high precision, and the q-axis current i ｑ This allows for accurate control of the q-axis voltage command value v ｑｒａ , the electrical angular velocity ω of the motor 60 ｅ , and the q-axis current i ｑ Based on this, the back electromotive force constant K ｅ Temperature change ΔK ｅ Therefore, the control device 10 can estimate the estimated back electromotive force constant K ｅ Temperature change ΔK ｅ Based on the temperature t of the magnet 61c ｍ By estimating the temperature t of the magnet 61c, ｍ This makes it possible to more accurately estimate the differential voltage value v by limiting the current, thereby preventing the output of the motor 60 from being reduced, and improving the efficiency of the motor 60. ｑ２ Since the disturbance voltage can be suitably removed from the temperature change ΔK ｅ Therefore, the temperature t of the magnet 61c can be estimated more accurately. ｍ can be estimated more accurately.

[0139] Furthermore, by taking into consideration the influence of magnetic saturation in the non-interference control, the control device 10 can calculate the electric angular velocity ω ｅand q-axis current i ｑ Based on this, the torque constant K ｔ Temperature change ΔK ｔ Therefore, the control device 10 can estimate the estimated torque constant K ｔ Temperature change ΔK ｔ Based on the temperature t of the magnet 61c ｍ By estimating the temperature t of the magnet 61c, ｍ This makes it possible to more accurately estimate the torque difference T ２ From the disturbance torque T ｄ can be suitably removed, so that the temperature change ΔK ｔ Therefore, the temperature t of the magnet 61c can be estimated more accurately. ｍ can be estimated more accurately.

[0140] The influence of magnetic saturation is more pronounced in the back electromotive force constant K than in the control of the motor 60. ｅ Temperature change ΔK ｅ and torque constant K ｔ Temperature change ΔK ｔ Therefore, the effect obtained by considering the influence of magnetic saturation in the non-interference control is that the temperature t ｍ As described above, the smaller the motor 60, the more likely it is that magnetic saturation will occur. Therefore, the effect obtained by considering the influence of magnetic saturation in the non-interference control is that the temperature t ｍ This is particularly useful when estimating

[0141] <Embodiment of Electric Power Steering Device> A control device similar to the control device 10 described in the embodiment of the drive device above can also be applied to an electric power steering device 2000 shown in FIG. 5. The electric power steering device 2000 of this embodiment shown in FIG. 5 is mounted on a vehicle. As shown in FIG. 5, the electric power steering device 2000 includes a steering mechanism 530 and a control device 210. The steering mechanism 530 includes a steering mechanism unit 520 and an assist mechanism unit 540. The electric power steering device 2000 controls the assist mechanism unit 540 using the control device 210 to generate an assist torque that assists the steering torque generated in the steering mechanism unit 520 when a driver of the vehicle steers a steering wheel 521. This assist torque reduces the driver's operational burden when operating the steering wheel 521. The driver of the vehicle is the helmsman who steers the steering wheel 521 of the vehicle.

[0142] The steering mechanism 520 has a handle 521, a steering shaft 522, universal joints 523A, 523B, an input shaft 524a, an output shaft 524b, a rack and pinion mechanism 525, a rack shaft 526, left and right ball joints 552A, 552B, tie rods 527A, 527B, knuckles 528A, 528B, and left and right tires 529A, 529B.

[0143] The steering shaft 522 extends from the steering wheel 521 operated by the driver. One end of the input shaft 524a is connected to the end of the steering shaft 522 opposite to the end connected to the steering wheel 521 via universal joints 523A and 523B. As a result, the steering wheel 521 is connected to the input shaft 524a via the universal joints 523A and 523B and the steering shaft 522. The output shaft 524b is connected to the input shaft 524a via a torsion bar 546, which will be described later. More specifically, one end of the output shaft 524b is connected to the other end of the input shaft 524a via the torsion bar 546. The other end of the output shaft 524b is connected to the rack shaft 526 via a rack-and-pinion mechanism 525.

[0144] The input shaft 524a and the output shaft 524b are arranged coaxially. The input shaft 524a and the output shaft 524b are rotatable about the same central axis. The input shaft 524a and the output shaft 524b are rotatable relative to each other within a range in which a torsion bar 546 (described later) can twist.

[0145] The auxiliary mechanism 540 includes a steering torque sensor 541, a steering angle sensor 542, a motor 260, a reduction gear mechanism 544, and a torsion bar 546. That is, the steering mechanism 530 includes the steering torque sensor 541, the steering angle sensor 542, the motor 260, the reduction gear mechanism 544, and the torsion bar 546. The torsion bar 546 connects the input shaft 524a and the output shaft 524b. The torsion bar 546 is disposed coaxially with the input shaft 524a and the output shaft 524b. In the following description, a virtual axis passing through the common central axis of the input shaft 524a, the output shaft 524b, and the torsion bar 546 is referred to as the rotation axis RJ. The torsion bar 546 can twist around the rotation axis RJ.

[0146] The steering torque sensor 541 detects the amount of twist of the torsion bar 546 about the rotation axis RJ, thereby detecting the steering torque in the steering mechanism 520. The steering torque is the torsion bar torque generated in the torsion bar 546, and is a torsional moment about the rotation axis RJ. The steering angle sensor 542 detects the rotation angle θ of the input shaft 524a about the rotation axis RJ. ａ The rotation angle θ of the input shaft 524a can be detected. ａ is the steering angle θ of the steering wheel 521 ｈ That is, the steering angle sensor 542 detects the rotation angle θ of the input shaft 524a. ａ By detecting the steering angle θ of the steering wheel 521, ｈ The rotation angle θ of the output shaft 524b can be detected based on the steering torque sensor 541 and the steering angle sensor 542. ｂ The rotation angle θ of the output shaft 524b can be detected. ｂ is the steering angle θ ｓ is.

[0147] The control device 210 controls the motor 260 in the same manner as the control device 10 in the embodiment of the drive device. The control device 210 and the motor 260 constitute a motor device 260a. The motor device 260a may be manufactured and sold independently of the parts of the electric power steering device 2000 other than the motor device 260a. Furthermore, the control device 210 may be manufactured and sold as a control device for controlling the electric power steering device 2000, independently of the parts of the motor device 260a other than the control device 210.

[0148] The present invention is not limited to the above-described embodiments, and other configurations and methods may be adopted within the scope of the technical concept of the present invention. The control device may perform any control as long as it calculates a compensation amount taking into account the influence of magnetic saturation occurring in the motor during decoupling control. The control device may calculate a compensation amount taking into account only one of the influence of cross-saturation caused by magnetic saturation and the influence of the motor rotation angle when magnetic saturation occurs during decoupling control. The control device may not take into account the influence of magnetic saturation occurring in the motor during decoupling control. The control device may estimate only one of the temperature change of the back electromotive force constant and the temperature change of the torque constant, and estimate the temperature of the magnet provided in the motor rotor based on only the estimated temperature change. The control device does not need to estimate the temperature of the magnet provided in the motor rotor. The motor controlled by the control device may be any type of motor. The application of the motor controlled by the control device is not particularly limited.

[0149] In this specification, "a value is estimated based on the rotational angular velocity of the motor" means that the electric angular velocity ω ｅ The value may be estimated based on the mechanical angular velocity ω of the motor. ｍ The certain value may be estimated based on the electrical angular velocity ω ｅ and mechanical angular velocity ω ｍ What is ω? ｅ =P ｒ ×ω ｍ In order to satisfy the relationship, for example, in the above-described embodiment, the electrical angular velocity ωｅ Instead of the mechanical angular velocity ω ｍ and the number of pole pairs P ｒ The above may be used to estimate various values.

[0150] The configurations and methods described in this specification can be combined as appropriate within the scope of not being mutually contradictory.

[0151] 10, 210...control device, 20f...model following control unit, 60, 260...motor, 60a, 260a...motor device, 61...rotor, 61c, 61d, 61e...magnet, 62b...coil, 63...electric circuit unit, 70...gear mechanism, 100...drive device, 530...steering mechanism, 2000...electric power steering device, a...integer, GT, GT ｄ , G.T. ｑ ... complementary sensitivity gain (gain), I ｍ ...Current, i ｄ ...d-axis current, i ｑ ...q-axis current, i ｑｒ ...q-axis current command value (target value), K ｅ , K ｅ０ ...back electromotive force constant, K ｔ , K ｔ０ ...torque constant, L ｄｑ , L ｄｑｘ , L ｑｄ , L ｑｄｘ ...static inductance, L ｄｑｉ , L ｄｑｘｉ , L ｑｄｉ , L ｑｄｘｉ ...dynamic inductance, N...number of revolutions, N1...first number of revolutions, N2...second number of revolutions, T ｃ (s), T ｃｄ (s), T ｃｑ (s) Complementary sensitivity function, T ｍ ...torque, t ｍ ...temperature, t ｍｅ ...first estimate, t ｍｓ ...estimated value, t ｍｔ ...second estimate, v ｃ ...Compensation amount, v ｃｑ１ ...first compensation amount, v ｃｑ２ ...Second compensation amount, v ｄａ ...d-axis component, v ｄｆ ...d-axis correction voltage value (correction voltage value), V ｍ …input voltage, vｑ ...q-axis voltage, v ｑａ ...q-axis component, v ｑｃ ...q-axis voltage compensation amount, v ｑｆ ...q-axis correction voltage value (correction voltage value), v ｑｒａ ...q-axis voltage command value (target value), V ｒ ...voltage command value (target value), x, y...variables, α, β...coefficients, Δ(s), Δ ｄ (s), Δ ｑ (s)...Modeling error, θ ｅ ...rotation angle, ω ｅ …Electrical angular velocity (rotational angular velocity)

Claims

1. A control device that decomposes the current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controls the motor using vector control that controls the d-axis current and the q-axis current, respectively, and is capable of performing non-interference control that compensates for at least part of the effect of the q-axis current on the control of the d-axis current and the effect of the d-axis current on the control of the q-axis current, and in the non-interference control, calculates the amount of compensation taking into account the effect of magnetic saturation that occurs in the motor.

2. The control device according to claim 1, wherein the compensation amount is calculated in the non-interference control by taking into account the influence of cross-saturation caused by the magnetic saturation.

3. The compensation amount includes a q-axis voltage compensation amount that is added to a target value of a q-axis voltage calculated based on the target value of the q-axis current, and the q-axis voltage compensation amount includes a first compensation amount, and the first compensation amount compensates for at least a part of the influence of the cross-saturation, and the first compensation amount is set to v ｃｑ１ When this is done, it is expressed by the following formula: In the above formula, ω ｅ is the electrical angular velocity of the motor, and i ｑ is the q-axis current, and i ｄ is the d-axis current, s is the Laplace transformer, and L ｄｑ is the static inductance as a coefficient representing the influence of the q-axis current on the d-axis component of the interlinkage magnetic flux generated in the motor, and L ｑｄｉ 3. The control device according to claim 2, wherein ∑j is a dynamic inductance as a coefficient representing an influence of the d-axis current on the q-axis component of the magnetic flux linkage.

4. In the non-interference control, the first compensation amount is calculated taking into consideration the likelihood of cross-saturation occurring, which varies depending on the rotation speed of the motor, and the static inductance L ｄｑ is expressed by the following formula: In the above formula, L ｄｑｘ is a static inductance as a coefficient representing the influence of the q-axis current on the d-axis component of the flux linkage when the rotation speed of the motor is a first rotation speed, and L ｄｑｙ is a static inductance as a coefficient representing the influence of the q-axis current on the d-axis component of the interlinkage magnetic flux when the rotation speed of the motor is a second rotation speed that is greater than the first rotation speed, and x and y are variables that satisfy the relationship x + y = 1; and the dynamic inductance L ｑｄｉ is expressed by the following formula: In the above formula, L ｑｄｘｉ is a dynamic inductance as a coefficient representing the influence of the d-axis current on the q-axis component of the flux linkage when the rotation speed of the motor is the first rotation speed, and L ｑｄｙｉ is a dynamic inductance as a coefficient representing the influence of the d-axis current on the q-axis component of the flux linkage when the rotation speed of the motor is the second rotation speed, and the static inductance L ｄｑｘ , L ｄｑｙ and the dynamic inductance L ｑｄｘｉ , L ｑｄｙｉ is a value that changes based on the value of the d-axis current and the value of the q-axis current; the variable x is 1 when the rotation speed of the motor becomes the first rotation speed, is 0 when the rotation speed of the motor becomes the second rotation speed, and decreases as the rotation speed of the motor increases from the first rotation speed to the second rotation speed; and the variable y is 0 when the rotation speed of the motor becomes the first rotation speed, is 1 when the rotation speed of the motor becomes the first rotation speed, and increases as the rotation speed of the motor increases from the first rotation speed to the second rotation speed.

5. The control device according to claim 2 or 3, wherein the compensation amount is calculated in the non-interference control, taking into consideration the likelihood of cross-saturation occurring, which varies depending on the rotation speed of the motor.

6. A control device as claimed in any one of claims 1 to 5, comprising a model following control section that constrains the transfer function of an electric circuit section including a coil of the motor to a nominal model based on the electric circuit section, wherein an input voltage is input to the electric circuit section, and the model following control section generates a correction voltage value that corrects a target value of the input voltage based on the current flowing through the electric circuit section and the nominal model, and is configured so that the transfer function of the electric circuit section is constrained to the transfer function of the nominal model in a frequency band where the gain in the gain characteristics of a complementary sensitivity function with respect to a modeling error between the electric circuit section and the nominal model is approximately 1.

7. A control device according to any one of claims 1 to 6, wherein the compensation amount is calculated in the non-interference control by taking into account the effect of the rotation angle of the motor when magnetic saturation occurs.

8. The compensation amount includes a q-axis voltage compensation amount that compensates for a target value of a q-axis voltage calculated based on the target value of the q-axis current, and the q-axis voltage compensation amount includes a second compensation amount, and the second compensation amount compensates for at least a part of the influence of the rotation angle of the motor, and the second compensation amount is set to v ｃｑ２ When this is done, it is expressed by the following formula: In the above formula, ω ｅ is the electrical angular velocity of the motor, and θ ｅ is the rotation angle of the motor, and i ｑ is the q-axis current, and i ｄ is the d-axis current, a is an integer equal to or greater than 1, and L ｄａ is the amplitude of the a-th order component of the d-axis current, and b ｄａ The control device according to claim 7 , wherein: is the phase of the a-th order component of the d-axis current.

9. A control device according to any one of claims 1 to 8, which estimates a temperature change in the back electromotive force constant of the motor based on a target value of the q-axis voltage calculated based on the target value of the q-axis current, the rotational angular velocity of the motor, and the q-axis current, and estimates the temperature of a magnet provided in the rotor of the motor based on the estimated temperature change in the back electromotive force constant.

10. A control device according to any one of claims 1 to 8, which estimates the temperature change of the torque constant of the motor based on the rotational angular velocity of the motor and the q-axis current, and estimates the temperature of a magnet provided in the rotor of the motor based on the estimated temperature change of the torque constant.

11. A control device as claimed in any one of claims 1 to 8, which estimates a temperature change in a back electromotive force constant in the motor based on a target value of a q-axis voltage calculated based on the target value of the q-axis current, the rotational angular velocity of the motor, and the q-axis current; estimates a temperature change in a torque constant in the motor based on the rotational angular velocity of the motor and the q-axis current; and estimates the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change in the back electromotive force constant and the estimated temperature change in the torque constant.

12. A control device that decomposes the current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controls the motor by vector control that controls the d-axis current and the q-axis current, respectively; estimates a temperature change in a back electromotive force constant in the motor based on a target value of a q-axis voltage calculated based on a target value of the q-axis current, the rotational angular velocity of the motor, and the q-axis current; estimates a temperature change in a torque constant in the motor based on the rotational angular velocity of the motor and the q-axis current; and estimates the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change in the back electromotive force constant and the estimated temperature change in the torque constant.

13. A control device as described in claim 11 or 12, which calculates a first estimated value of the temperature of the magnet based on the temperature change of the estimated back electromotive force constant, calculates a second estimated value of the temperature of the magnet based on the temperature change of the estimated torque constant, and if the first estimated value and the second estimated value differ from each other, obtains the higher of the first estimated value and the second estimated value as the estimated value of the temperature of the magnet.

14. A motor device comprising: a control device according to any one of claims 1 to 13; and the motor.

15. A drive device comprising: a control device according to any one of claims 1 to 13; the motor; and a gear mechanism connected to the motor.

16. An electric power steering device comprising: a control device according to any one of claims 1 to 13; and a steering mechanism having the motor.

17. A control method for decomposing the current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controlling the motor by vector control that controls the d-axis current and the q-axis current, the control method comprising: executing non-interference control that compensates for at least a portion of the effect of the q-axis current on the control of the d-axis current and the effect of the d-axis current on the control of the q-axis current; and calculating a compensation amount in the non-interference control, taking into account the effect of magnetic saturation that occurs in the motor.

18. The control method according to claim 17, wherein the non-interference control includes calculating the compensation amount in consideration of the influence of cross-saturation caused by the magnetic saturation.

19. The compensation amount includes a q-axis voltage compensation amount that is added to a target value of a q-axis voltage calculated based on the target value of the q-axis current, the q-axis voltage compensation amount includes a first compensation amount, the first compensation amount compensates for at least a part of the influence of the cross-saturation, and the first compensation amount is set to v ｃｑ１ When this is done, it is expressed by the following formula: In the above formula, ω ｅ is the electrical angular velocity of the motor, and i ｑ is the q-axis current, and i ｄ is the d-axis current, s is the Laplace transformer, and L ｄｑ is the static inductance as a coefficient representing the influence of the q-axis current on the d-axis component of the interlinkage magnetic flux generated in the motor, and L ｑｄｉ 19. The control method of claim 18, wherein: ##EQU1## is a dynamic inductance as a coefficient representing the effect of the d-axis current on the q-axis component of the flux linkage.

20. The non-interference control includes calculating the first compensation amount in consideration of the likelihood of cross-saturation occurring, which varies depending on the rotation speed of the motor, and the static inductance L ｄｑ is expressed by the following formula: In the above formula, L ｄｑｘ is a static inductance as a coefficient representing the influence of the q-axis current on the d-axis component of the flux linkage when the rotation speed of the motor is a first rotation speed, and L ｄｑｙ is a static inductance as a coefficient representing the influence of the q-axis current on the d-axis component of the interlinkage magnetic flux when the rotation speed of the motor is a second rotation speed that is greater than the first rotation speed, and x and y are variables that satisfy the relationship x + y = 1; and the dynamic inductance L ｑｄｉ is expressed by the following formula: In the above formula, L ｑｄｘｉ is a dynamic inductance as a coefficient representing the influence of the d-axis current on the q-axis component of the flux linkage when the rotation speed of the motor is the first rotation speed, and L ｑｄｙｉ is a dynamic inductance as a coefficient representing the influence of the d-axis current on the q-axis component of the flux linkage when the rotation speed of the motor is the second rotation speed, and the static inductance L ｄｑｘ , L ｄｑｙ and the dynamic inductance L ｑｄｘｉ , L ｑｄｙｉ is a value that changes based on the value of the d-axis current and the value of the q-axis current; the variable x is 1 when the rotation speed of the motor becomes the first rotation speed, is 0 when the rotation speed of the motor becomes the second rotation speed, and decreases as the rotation speed of the motor increases from the first rotation speed to the second rotation speed; and the variable y is 0 when the rotation speed of the motor becomes the first rotation speed, is 1 when the rotation speed of the motor becomes the first rotation speed, and increases as the rotation speed of the motor increases from the first rotation speed to the second rotation speed.

21. The control method according to claim 18 or 19, wherein the non-interference control includes calculating the compensation amount in consideration of the likelihood of cross-saturation occurring, which varies depending on the rotation speed of the motor.

22. A control method according to any one of claims 18 to 21, including model-following control that constrains a transfer function of an electric circuit section of the motor, including a coil, to a nominal model based on the electric circuit section, wherein an input voltage is input to the electric circuit section, and the model-following control includes: generating a correction voltage value that corrects a target value of the input voltage based on a current flowing through the electric circuit section and the nominal model; and constraining the transfer function of the electric circuit section to the transfer function of the nominal model in a frequency band where a gain in a gain characteristic of a complementary sensitivity function with respect to a modeling error between the electric circuit section and the nominal model is approximately 1.

23. A control method according to any one of claims 18 to 22, wherein the non-interference control includes calculating the compensation amount in consideration of the effect of the rotation angle of the motor when magnetic saturation occurs.

24. The compensation amount includes a q-axis voltage compensation amount that compensates for a target value of a q-axis voltage calculated based on the target value of the q-axis current, and the q-axis voltage compensation amount includes a second compensation amount, and the second compensation amount compensates for at least a part of the influence of the rotation angle of the motor, and the second compensation amount is set to v ｃｑ２ When this is done, it is expressed by the following formula: In the above formula, ω ｅ is the electrical angular velocity of the motor, and θ ｅ is the rotation angle of the motor, and i ｑ is the q-axis current, and i ｄ is the d-axis current, a is an integer equal to or greater than 1, and L ｄａ is the amplitude of the a-th order component of the d-axis current, and b ｄａ The control method according to claim 23, wherein: is the phase of the a-th order component of the d-axis current.

25. A control method according to any one of claims 18 to 24, comprising: estimating a temperature change in a back electromotive force constant in the motor based on a target value of a q-axis voltage calculated based on the target value of the q-axis current, the rotational angular velocity of the motor, and the q-axis current; and estimating the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change in the back electromotive force constant.

26. A control method according to any one of claims 18 to 24, comprising: estimating a temperature change in a torque constant in the motor based on the rotational angular velocity of the motor and the q-axis current; and estimating the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change in the torque constant.

27. A control method according to any one of claims 18 to 24, comprising: estimating a temperature change in a back electromotive force constant of the motor based on a target value of a q-axis voltage calculated based on the target value of the q-axis current, the rotational angular velocity of the motor, and the q-axis current; estimating a temperature change in a torque constant of the motor based on the rotational angular velocity of the motor and the q-axis current; and estimating the temperature of a magnet provided in a rotor of the motor based on the estimated temperature change in the back electromotive force constant and the estimated temperature change in the torque constant.

28. A control method for decomposing the current flowing through a motor into a d-axis current, which is a current component that contributes to the generation of magnetic flux, and a q-axis current, which is a current component that contributes to the generation of torque, and controlling the motor by vector control that controls the d-axis current and the q-axis current, respectively, the control method comprising: estimating a temperature change in the back electromotive force constant of the motor based on a target value of the q-axis voltage calculated based on the target value of the q-axis current, the rotational angular velocity of the motor, and the q-axis current; estimating a temperature change in the torque constant of the motor based on the rotational angular velocity of the motor and the q-axis current; and estimating the temperature of a magnet provided in the rotor of the motor based on the estimated temperature change in the back electromotive force constant and the estimated temperature change in the torque constant.

29. A control method as described in claim 27 or 28, comprising: calculating a first estimated value of the temperature of the magnet based on the estimated change in the back electromotive force constant with temperature; calculating a second estimated value of the temperature of the magnet based on the estimated change in the torque constant with temperature; and, if the first estimated value and the second estimated value differ from each other, obtaining the higher of the first estimated value and the second estimated value as the estimated value of the temperature of the magnet.

30. A program that causes a computer to execute the control method according to any one of claims 18 to 29.

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