Control method for resonant converter

Abstract

The present invention relates to a control method for a resonant converter, the method comprising: determining a first switch-off threshold range and a first switching frequency of the resonant converter in a present first operating mode; measuring an input voltage and an output voltage of the resonant converter, to determine a present voltage gain of the resonant converter; based on a load and the present voltage gain of the resonant converter, determining a second switching frequency of the resonant converter in a second operating mode to be switched to; based on the first switch-off threshold range, the first switching frequency, a capacitance of the resonant capacitor and the second switching frequency, calculating a second switch-off threshold range in the second operating mode; and acquiring voltage feedback across the resonant capacitor in real time, and when the voltage feedback exceeds the first switch-off threshold range, switching a switch-off threshold of the resonant converter to the second switch-off threshold range, and switching the resonant converter from the first operating mode to the second operating mode by enabling corresponding switches in the two phase bridge arms in the full-bridge conversion circuit.

Description

[0001]Description Title of the Invention Control method for resonant converter Technical Field The present invention relates to the technical field of resonant converters, and more specifically to a control method for a resonant converter. Background Art With the rapid development of power electronics technology, resonant DCDC converters are playing an increasingly important role in power electronics systems. Conventional wide-voltage-range DCDC converters mostly employ Buck or Fly- back topologies with duty cycle control. With increases in power density and cost requirements, the development of new power semiconductors and advances in core component materials and design technology, DCDC converters are developing towards high frequencies, and the advantages of the resonant DCDC conversion topology are becoming more and more obvious. A resonant DCDC converter achieves voltage and current control by utilizing the resonant characteristics of a resonant tank, and in this way, the efficiency and power density of the converter are increased. However, the operating principles of resonant converters limit their voltage gain range, which limits their application scope to a certain extent. An effective way to expand the voltage gain range is through topology restructuring. Topology restructuring allows the converter to switch between different operating modes, resulting in a wider voltage gain range. At present, topology restructuring control methods for resonant DCDC converters are mainly based on an optimal path control method of direct frequency or duty cycle control. These methods are capable of achieving topology restructuring, but have significant drawbacks: firstly, these methods require complex calculations, which increases the complexity of the control algorithms; secondly, these methods have high requirements regarding the precision of parameters of devices used in the topology, which may increase the cost and design difficulty of the system; and finally, direct frequency control has poor dynamic performance, which may have an impact on the stability and reliability of the converter. Summary of the Invention The present invention is intended to solve the problems in existing control methods for resonant DCDC converters, by providing a topology hot restructuring control method with high dynamic performance. The present invention is able to achieve more stable and reliable control in the topology restructuring process by using feedback control of resonant tank state variables, thereby improving the overall performance of the resonant DCDC converter. Specifically, the invention proposes a control method for a resonant converter, a primary side of the resonant converter comprising a full-bridge conversion circuit and a resonant circuit, the full-bridge conversion circuit being formed of two phase bridge arms combined in a parallel-connected or series-connected manner, and the resonant circuit consisting of a magnetizing inductance, a resonant capacitor and a resonant inductor, and the method being used to control switching of the resonant converter from a present first operating mode to a second operating mode, wherein the control method comprises the following steps:determining a first switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^] and a firstswitching frequency ^^_^^of the resonant converter in the present first operating mode; measuring an input voltage^^^and an output voltage ^^^^of the resonantconverter, to determine a present voltage gain ^^^^ of the resonant converter;based on a load ^^^^ and the present voltage gain ^^^^ of the resonantconverter, determining a second switching frequency ^^_^^of the resonant converter in the second operating mode to be switched to;based on the first switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^], the firstswitching frequency ^^_^^, a capacitance of the resonant capacitor ^^^and the second switching calculating a second switch-off threshold range of the resonant converter in the second operating mode;and acquiring voltage feedback VCr1 across the resonant capacitor in real time, and when the voltage feedback VCr1 exceeds the first switch-off threshold range[^^^_^^_^^_^ , ^^^_^^_^^_^], switching a switch-off threshold of the resonant converterto the second switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^], and switchingthe resonant converter from the first operating mode to the second operating mode by enabling corresponding switches in the two phase bridge arms in the full-bridge conversion circuit. According to an optional embodiment, the full-bridge conversion circuit is formed of the two phase bridge arms combined in a series-connected manner, and input voltages ^^^^^and ^^^^^of the two phase bridge arms are each half of an input voltage ^^^of the resonant converter. According to an optional embodiment, the method further comprises: whencalculating the second switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^],ensuring that an output power ^^^_^^and a voltage gain of the resonant converter in the first operating mode are the same as an output power ^^^_^^and a voltage gain of the resonant converter in the second operating mode. According to an optional embodiment, the first operating mode is a full-bridge mode, and the second operating mode is a half-bridge mode. According to an optional embodiment, the method further comprises: taking into account a DC offset ^^^_^^of the resonant capacitor when calculating the secondswitch-off threshold range ^^^_^^_^^_^].According to an optional embodiment, a DC offset ^^^_^^_^^of the resonant capacitor in the full-bridge mode is set to be 1 / 2 of the input voltage ^^^of the resonant converter, and a DC offset ^^^_^^_^^of the resonant capacitor in the half- bridge mode is set to be 1 / 4 of the input voltage ^^^of the resonant converter. According to an optional embodiment, in the full-bridge mode, the two phase bridge arms are caused to both respond to the first switch-off threshold range[^^^_^^_^^_^ , ^^^_^^_^^_^]; and in the half-bridge mode, a corresponding one of thetwo phase bridge arms is caused to respond to the second switch-off thresholdrange [^^^_^^_^^_^, ^^^_^^_^^_^] each time..According to an optional embodiment, a slope compensation value is added whencalculating the second switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^], theslope compensation value being determined on the basis of a given harmonic compensation gradient ^^^^^^and the second switching frequency ^^_^^. According to an optional embodiment, the second switching frequency ^^_^^is determined on the basis of a predefined lookup table or mathematical model. According to an optional embodiment, the voltage feedback VCr1 across the resonant capacitor is directly sampled by a voltage sensor, or calculated from a resonant capacitor current ICr1 sampled with the aid of a current sensor. The control method for a resonant converter of the present invention has a significant advantage over existing control methods, specifically high dynamic performance. Through the use of closed-loop feedback control of resonant tank state variables, the present invention enables more precise voltage and current control during topology restructuring, significantly reducing overregulation of output voltage and current. This control strategy not only improves the stability and reliability of the system, but also allows online hot restructuring of the system without downtime, thus improving the power supply continuity and efficiency of the system. In addition, the control method of the present invention simplifies the control algorithm, reduces the precision requirements for device parameters, and makes system design simpler and more convenient, with a lower cost. The control method for the resonant converter of the present invention has higher flexibility and adaptability than existing control methods, while also achieving a wide voltage gain range. The invention enables smooth switching between different operating modes through precise control of the switching state of each phase bridge arm, thus achieving a wider voltage gain range. In addition, the control method of the present invention also takes into account differences in DC offset voltage on the resonant capacitor, ensuring a smooth transition when switching between different modes through dynamic adjustment of switching frequency and duty cycle, and avoiding current surges and upward or downward surges in output voltage / current. This closed-loop control strategy based on state quantity feedback makes the control method of the present invention relatively insensitive to error in resonant tank device parameters, thus improving the robustness of the system and its ability to adapt to different load requirements. Through this control method, the present invention provides a solid technical foundation for realizing DCDC converters having high efficiency and high power density, and is particularly suitable for application fields with stringent requirements regarding a wide voltage gain range and high dynamic performance, such as electric vehicles, photovoltaic energy storage systems, etc. Brief Description of the Drawings Through the incorporation of the attached drawings hereof and the following specific embodiments which, together with the drawings, are used to illustrate certain principles of the present invention, other features and advantages of the method of the present invention will become clear or illustrated more specifically. Fig.1 shows a topology diagram of a conventional LLC resonant converter. Figs.2a and 2b show exemplary switch state diagrams of a series-connected full- bridge LLC resonant converter in half-bridge mode and full-bridge mode, respectively. Fig.3 shows a comparison of curves of voltage gain against switching frequency for the resonant converter in full-bridge mode and half-bridge mode. Figs.4a and 4b show signal waveforms of a series-connected full-bridge LLC resonant converter in half-bridge mode and full-bridge mode, respectively. Fig.5a shows control waveforms for the restructuring of a series-connected full- bridge LLC resonant converter from FB mode to HB mode, using the control method according to an exemplary embodiment of the present invention. Fig.5b shows control waveforms for the restructuring of a series-connected full- bridge LLC resonant converter from HB mode to FB mode, using the control method according to an exemplary embodiment of the present invention. Fig.6a shows control waveforms when slope compensation is superimposed in the process of controlling restructuring from FB mode to HB mode as shown in Fig.5a. Fig.6b shows control waveforms when slope compensation is superimposed in the process of controlling restructuring from HB mode to FB mode as shown in Fig.5b. Fig.7 shows control waveforms when slope compensation is superimposed and current integration feedback sampling is used in the process of controlling restructuring from FB mode to HB mode as shown in Fig.5a. Detailed Description of the Invention The control method for a resonant converter according to the present invention is described below through embodiments with reference to the drawings. Many specific details are expounded in the description below, to give those skilled in the art a more comprehensive understanding of the present invention. However, it will be obvious to those skilled in the art that the invention can be realized without some of these specific details. On the contrary, one may consider using any combination of the features and key elements below to implement the present invention, regardless of whether they relate to different embodiments. Therefore, the various aspects, features, embodiments and advantages below merely serve an illustrative purpose, and should not be regarded as key elements or definitions of the claims. In order to improve the dynamic performance of a resonant converter, a current- type control method with resonant capacitor voltage or current feedback can be used. However, none of the current methods are suitable for dynamic hot restructuring of more complex resonant converter topologies. In addition, existing control methods may cause overregulation of output voltage or current during topology restructuring, which may trigger a protection mechanism, causing systemshutdown, which impacts system continuity and reliability.Fig.1 shows a topology diagram of a conventional LLC resonant converter. This LLC resonant converter generally consists of two circuit parts, namely a primary side and a secondary side, i.e. a circuit part connected to a secondary winding of a transformer. The primary side is a circuit part directly connected to a primarywinding of the transformer, and is mainly responsible for energy input andpreliminary processing; the chief function of the secondary side is to receive energy transmitted from the transformer, and supply this energy to a load for use after rectification and filtering. As shown in Fig.1, the primary side of the resonant converter comprises a full-bridge conversion circuit and a resonant circuit, whereinthe full-bridge conversion circuit is formed of two phase bridge arms - a phase 1consisting of Q11 and Q12, and a phase 2 consisting of Q21 and Q22 - combinedin a parallel-connected or series-connected manner, and the resonant circuit consists of a magnetizing inductance Lm1, a resonant capacitor Cr1 and a resonant inductor Lr1. The resonant converter can operate in full-bridge mode (abbreviated herein as "FB" mode) or half-bridge mode (abbreviated herein as "HB" mode), depending on the on / off switching of switches in each phase bridge arm. Figs.2a and 2b show exemplary switch state diagrams of a series-connected full- bridge LLC resonant converter in half-bridge mode and full-bridge mode, respectively. As shown in Fig.2a, the input voltage ^^^of the resonant converter is equal to the sum of the input voltages ^^^^^, ^^^^^of the two phase bridge arms. In half-bridge mode, Q11 of phase 1 and Q22 of phase 2 are ON, while Q12 and Q21 are OFF, and the direction of current formed at the primary side of the resonant converter is as indicated by the arrows. As shown in Fig.2b, in full-bridge mode, Q11 and Q21 are ON, while Q12 and Q22 are OFF, and the direction of current formed at the primary side of the resonant converter is as indicated by the arrows. In these two operating modes, the voltage output gains capable of being achieved by the resonant converter are different. Fig.3 shows a comparison of curves of voltage gain against switching frequency for the resonant converter in FB mode and HB mode. As shown in Fig.3, at the same switching frequency, FB mode can achieve a wider gain range, while the gain range capable of being achieved in HB mode is narrower. In addition, the switching frequency required to achieve the same voltage gain is different in FB mode and HB mode. When performing voltage gain expansion, existing converter control methods often require complex calculations and highly precise device parameters, which not only increases system complexity and cost, but also limits system flexibility and adaptability. In addition, existing control methods often struggle to achieve efficient power conversion and stable voltage output when processing DC loads at different voltage levels. Figs.4a and 4b show signal waveforms of a series-connected full-bridge LLC resonant converter in half-bridge mode and full-bridge mode, respectively. Fig.4a shows operating voltages and operating currents of the resonant converter in FB mode; Fig.4b shows operating voltages and operating currents of the resonant converter in HB mode. G11, G12, G21 and G21 are drive voltage waveforms of switches Q11, Q12, Q13 and Q14 respectively, where high level means ON and low level means OFF. Is11 and Is12 are secondary-side rectification currents at the locations marked in Fig.1. ILr1 is the current flowing through the resonant inductor Lr1. Ilm1 is the magnetizing current of transformer T1. VCr1 is the voltage across the resonant capacitor Cr1. VTin1-VTin2 is the output differential-mode voltage of the two phase bridge arms, and is also the input voltage of the LLC resonant tank. G11 are G12 are switched complementarily. G21 are G22 are switched complementarily. In FB mode, phase 1 and phase 2 have the same frequency, and a phase difference of Ts_FB / 2. Suppose that the duty cycle of G11, G12, G21 and G21 is D. Then the voltage DC offset on Cr1 is VCr1_DC = D*Vin. Typically, D is 0.5, in which case, VCr1_DC is Vin / 2. In HB mode, phase 1 and phase 2 have the same frequency, and phase difference of Ts_HB / 2, but only one phase bridge arm is allowed to be ON each time. Suppose that the duty cycle of G11 and G12 is D1. The duty cycle of G21 and G22 is D2. The interval periods of phase 1 and phase 2 operate in a staggered manner. Then the voltage DC offset on Cr1 is VCr1_DC = [D1*VCin1 + (1-D2)*VCin2] / 2. Typically, D1 is 0.5, D2 is 0.5, and VCin1 = VCin2 = Vin / 2, in which case, VCr1_DC is Vin / 4. It can be seen that the DC offset voltages on the resonant capacitor are different in the two modes; consequently, if the duty cycles and switching frequencies of the switches are not controlled according to the present resonant capacitor voltage when switching between the two modes, a large current surge will be input into the resonant tank, resulting in an upward or downward surge in output voltage / current. The present invention aims to develop a novel control method for a resonant converter, to achieve high dynamic performance of a resonant DCDC converter over a wide voltage gain range. This control method should be able to reduce voltage and current overregulation during topology restructuring, and improve system stability and reliability. At the same time, the method should simplify the control algorithm, lower the requirements for device parameter precision, and improve system practicality and economy. In prior path optimization control strategies, the duration Ttrans of a conversion state is generally calculated using input voltage, output voltage and output current, etc., and control is performed directly. Direct control of Ttrans is still a type of open-loop control, and since it is not possible to perform closed-loop control and stabilization of the resonant tank state before and after conversion, resonant tank overcurrent is still likely to result during switching, with an upward or downward surge in output voltage / current. The present invention proposes that resonant capacitor voltage / current can be sampled to perform closed-loop feedback control of the switching process; this closed-loop feedback control method is presented below with reference to Figs.5a and 5b. Fig.5a shows control waveforms for the restructuring of a series-connected full- bridge LLC resonant converter from FB mode to HB mode, using the control method according to an exemplary embodiment of the present invention. The switches of respective phase bridge arms are controlled by comparing VCr1 feedback with respective control thresholds outputted by an output voltage / current feedback controller, as shown in Fig.5a. The switching time Ttrans is a result of closed-loop control, and not a control quantity. FB_TH_H is the switch-off threshold for Q11 of phase 1 and Q22 of phase 2 in FB mode. FB_TH_L is the switch-off threshold for Q12 of phase 1 and Q21 of phase 2 in FB mode. HB_TH_H is the switch-off threshold for Q11 of phase 1 and Q22 of phase 2 in HB mode. HB_TH_L is the switch-off threshold for Q12 of phase 1 and Q21 of phase 2 in HB mode. In HB mode, the switches of only one of the phase bridge arms are enabled each time. The switch bridge arm to be enabled next needs to be determined each time an HB_TH_H comparison is triggered. Taking into account input voltage / power balance, the bridge arm switches of phase 1 and phase 2 are normally triggered in turn at intervals, as shown in the figure. Depending on the voltage balancing or gain control requirements, each phase bridge arm may also use its own independent comparison threshold, thus giving rise to a phase difference between different bridge arms. This control method realizes switching control of the transistor switches of each phase bridge arm in FB mode and HB mode. A switching frequency Fs / switching period Ts are the result of closed-loop feedback control of resonant capacitor voltage, and not directly controlled by an output voltage / current loop controller. As shown in Figure 5a, when switching from FB mode to HB mode, a determination should be made before time t0 to perform mode switching. At time t0, when the event Vcr1 > FB_TH_H is triggered, the switch-off threshold for Q12 of phase 1 and Q21 of phase 2 (Q12 and Q21 being ON, Q11 and Q22 being OFF) will change from FB_TH_L to HB_TH_L, and one of the phase bridge armsis prohibited from responding to this switch-off threshold, thus preventingsynchronous switching of the two phase bridge arms. At time t1, when the event Vcr1 < HB_TH_L is triggered, the switch-off threshold for Q11 of phase 1 and Q22 of phase 2 will change from FB_TH_H to HB_TH_H. Fig.5b shows control waveforms for the restructuring of a series-connected full- bridge LLC resonant converter from HB mode to FB mode, using the control method according to an exemplary embodiment of the present invention. As shown in Fig.5b, when switching from HB mode to FB mode, a determination should be made before time t0 to perform mode switching. At time t0, when the event Vcr1 > HB_TH_H is triggered, the switch-off threshold for Q12 of phase 1 and Q21 of phase 2 will change from HB_TH_L to FB_TH_L, and all of the two phase bridge arms are enabled to respond to this switch-off threshold, so that the two phase bridge arms are switched synchronously. At time t1, when the event Vcr1 < FB_TH_L is triggered, the switch-off threshold for Q11 of phase 1 and Q22 of phase 2 will change from HB_TH_H to FB_TH_H. The abovementioned FB_TH_H, FB_TH_L, HB_TH_H and HB_TH_L are output by an output voltage / current closed loop controller before mode switching and upon completion of switching. Initial values after mode switching may be obtained from the currently required load current, input voltage, output voltage, etc., by looking up a table or by calculation according to an LLC mathematical model. The detailed process of calculating the switch-off thresholds for switching from FB mode to HB mode and for switching from HB mode to FB mode is described below. Neglecting losses and junction capacitance of power switching devices, the output power in HB mode is ^^^_^^, wherein the working output power of phase 1 is ^^^^_^^, and the working output power of phase 2 in HB mode is ^^^^_^^. The output power in FB mode is^^^_^^. Suppose that the output power needs to be kept the same before and after switching, and that the input voltages of the two phases are equal. This is shown in equations (1), (2), (3), (4) and (5).^^^is the resonant capacitor, of known capacitance.^^^_^^_^^_^is the actual voltage corresponding to the comparison threshold HB_TH_H,^^^_^^_^^_^is the actual voltage corresponding to the comparison threshold HB_TH_L, ^^^_^^_^^_^is the actual voltage corresponding to the comparison threshold FB_TH_H, and ^^^_^^_^^_^is the actual voltage corresponding to the comparison threshold FB_TH_L. ^^^^_^^ = ^^^^_^^ = ^^^_^^ Equation (4)^^^^= ^^^^^ = ^^^^^ Equation (5)Before switching from FB mode to HB mode,^^_^^is the currently known FB mode switching frequency, and ^^^_^^_^^_^and ^^^_^^_^^_^are the currently known output quantities of the output voltage / current closed-loop controller in FB mode. The switching frequency ^^_^^in HB mode after conversion may be obtained from the required load and voltage gain by looking up a calibrated lookup table, or obtained by calculation from a mathematical model of the LLC converter,^^_^^(^^^^, ^^^^). Using the known quantities mentioned above and equations (1)- (5), it is possible to obtain a threshold ^^^_^^_^^_^ − ^^^_^^_^^_^ needed in HBmode after switching, which is required in order to keep the power the same before and after switching, and to keep the gain the same before and after switching: Equation (6)As can be seen, since ^^_^^and ^^^_^^_^^_^, ^^^_^^_^^_^are present actual values, the main source of error is the precision of the LLC converter model (lookup table or mathematical calculation model) and some parasitic parameters of components. To achieve an equivalent duty cycle of 50% for the resonant tank in HB mode, the resonant capacitor DC offset ^^^_^^needs to be 1 / 4 of the input voltage. The HB mode switch-off thresholds can then be calculated from equations (7) and (8): Equation (8)Similarly, before switching from HB mode to FB mode, ^^_^^is the currently known HB mode switching frequency, and ^^^_^^_^^_^, ^^^_^^_^^_^are the currently known output quantities of the output voltage / current closed-loop controller in HB mode. The switching frequency ^^_^^in FB mode after conversion may be obtained from the required load and voltage gain by looking up a calibrated lookup table, or obtained by calculation from a mathematical model ofthe LLC converter, ^^_^^(^^^^, ^^^^). Using the known quantities mentionedabove and equations (1) - (5), it is possible to obtain a threshold ^^^_^^_^^_^ −^^^_^^_^^_^needed in FB mode after switching, which is required in order to keep the power the same before and after switching, and to keep the gain the same before and after switching: Equation (9) As can be seen, since ^^_^^and ^^^_^^_^^_^, ^^^_^^_^^_^are present actual values, the main source of error is the precision of the LLC converter model (lookup table or mathematical calculation model) and some parasitic parameters of components. To achieve an equivalent duty cycle of 50% for the resonant tank in FB mode, the resonant capacitor DC offset ^^^_^^needs to be 1 / 2 of the input voltage. The switch-off thresholds in FB mode can then be calculated from equations (10) and (11). Equation (11)The above method uses the output control quantities of the output voltage / currentloop controller directly as comparison thresholds, for direct comparison withvoltage feedback of the resonant capacitor Cr1. However, unwanted subharmonic oscillation may occur in resonant converters. In order to suppress this subharmonic oscillation, slope compensation can be superimposed on the output control quantities of the voltage / current loop controller on the basis of the above method. Fig.6a shows control waveforms when slope compensation is superimposed in the process of controlling restructuring from FB mode to HB mode as shown in Fig.5a. As shown in Fig.6, suppose that the gradient of the added harmonic compensation is ^^^^^^.In this case, the output control quantities of an output voltage / current outer loop controller are FB_TH_H_C, FB_TH_L_C, HB_TH_H_C, and HB_TH_L_C. When bridge arm switch-off is triggered, the actual comparison thresholds with slope compensation superimposed thereon are FB_TH_H, FB_TH_L, HB_TH_H, and HB_TH_L. In FB mode, when the event VCr1 > FB_TH_H is triggered, FB_TH_L begins to have a slope superimposed thereon from the initial value FB_TH_L_C. When the event VCr1 < FB_TH_L is triggered, FB_TH_H begins to have a slope superimposed thereon from the initial value FB_TH_H_C. As in the basic topology mode switching method already described, when switching from FB mode to HB mode, the switch-off comparison threshold for G12 of phase 1 and G21 of phase 2 needs to be updated to HB_TH_L at time t0, asshown in Equation (8). In view of the relative complexity of the calculation of Trans,no slope is superimposed on HB_TH_L for the first comparison after switching to HB mode. At time t1, the initial control quantities in HB mode HB_TH_H_C and HB_TH_L_C are updated; in view of the fact that slopes have been superimposed, the required slope compensation values should be superimposed on the basis of Equations (7) and (8), as shown in Equations (12) and (13), so that the power is the same before and after mode switching, and the voltage gain is the same before and after mode switching. When the mode switch is complete, the output voltage / current loop controller regulates HB_TH_H_C and HB_TH_L_C and continues to perform closed-loop control. Fig.6b shows control waveforms when slope compensation is superimposed in the process of controlling restructuring from HB mode to FB mode as shown in Fig.5b. Similarly, in HB mode, when the event VCr1 > HB_TH_H is triggered, HB_TH_L begins to have a slope superimposed thereon from the initial value HB_TH_L_C. When the event VCr1 < HB_TH_L is triggered, HB_TH_H begins to have a slope superimposed thereon from the initial value HB_TH_H_C. As in the basic topology mode switching method already described, when switching from HB mode to FB mode, the switch-off comparison threshold for G12 of phase 1 and G21 of phase 2 needs to be updated to FB_TH_L at time t0, asshown in Equation (11). In view of the relative complexity of the calculation ofTrans, no slope is superimposed on FB_TH_L for the first comparison after switching to FB mode. At time t1, the initial control quantities in FB modeFB_TH_H_C and FB_TH_L_C are updated; in view of the fact that slopes havebeen superimposed, the required slope compensation values that should be superimposed on the basis of Equations (10) and (11) are shown in Equations (14) and (15), so that the power is the same before and after mode switching, and the voltage gain is the same before and after mode switching. After the mode switch is complete, the output voltage / current loop controller regulates FB_TH_H_C, FB_TH_L_C and continues to perform closed-loop control. The voltage feedback VCr1 of the resonant capacitor Cr1 is used for comparison with the comparison thresholds in the control method described above. It should be pointed out that this voltage feedback VCr1 can be obtained by direct sampling by a voltage sensor, or by calculating from resonant capacitor current ICr1 (equal to ILr1) feedback sampled by a current transformer / sensor. The above control method can be used directly when the resonant capacitor voltage is sampled directly. For a sampling method using integration, the DC voltage offset on Cs is independent of the DC voltage offset on the resonant capacitor Cr1 in practical applications, and consequently, there will generally be adifference in the DC voltage offset between a feedback voltage VCr1_s obtainedby sampling and the actual resonant capacitor voltage VCr1; in this case, appropriate adjustments will need to be made to the topology operating mode switching control method described above. Fig.7 shows control waveforms when slope compensation is superimposed and current integration feedback sampling is used in the process of controlling restructuring from FB mode to HB mode as shown in Fig.5a. As shown in Fig.7, when operating steadily in FB mode, the Cr1 voltage correspondingly represented by the DC voltage offset on Cs is ^^^_^^_^^_^; the actual DC voltage offset of Cr1 is ^^^_^^_^^.When operating steadily in HB mode, the Cr1 voltage correspondingly represented by the DC voltage offset on Cs is ^^^_^^_^^_^; the actual DC voltage offset of Cr1 is^^^_^^_^^. ^^^_^^_^^_^and ^^^_^^_^^_^might use a bias source or bias load (or at least parasitic resistance present in the actual circuit) in actual use, and will thus slowly stabilize to a set DC offset voltage, such as 0 V or a reference voltage, after a non- set DC offset occurs. However, unless a corresponding change is sampled at Cs resulting from the actual Cr1 voltage DC offset changing suddenly due to a switch state change, such that , the DC offset voltages ^^^_^^_^^_^and ^^^_^^_^^_^on Cs will not change suddenly, and can be regarded as unchanging known constants for a relatively large number of switching periods. When the control objective is to achieve an equivalent resonant tank input voltage ^ ^ duty cycle of 50%, this corresponds to ^^^ ^^^^_^^_^^= ^ and ^^= ^ . When operating steadily in FB mode: ^ ^^^^_^^_^^_^_^_^ ^^_^^_^^_^_^_^ ^ = ^ + Equation (16)^^_^^_^^_^_^_^ ^^_^^_^^_^^ ^ ^^^^_^^_^^_^_^_^ ^^_^^_^^_^_^_^ ^ = ^ − Equation (17)^^_^^_^^_^_^_^ ^^_^^_^^_^^where^ − ^ is an output control quantity of an output^^_^^_^^_^_^_^ ^^_^^_^^_^_^_^voltage loop / current loop controller, which is superimposed on the known Cs DC voltage offset ^ , to obtain the required comparison thresholds^^_^^_^^_^^ , ^ . Also, after superimposing suitable slope^^_^^_^^_^_^_^ ^^_^^_^^_^_^_^compensation, a comparison is performed with the feedback voltage ^ , to^^^_^finally control the switches of phase 1 and the switches of phase 2 of the bridge arms. This is equivalent to comparing ^ , ^ with the^^_^^_^^_^_^ ^^_^^_^^_^_^ feedback voltage ^^^^after superimposing suitable slope compensation, when the Cr1 voltage is sampled directly. When operating steadily in HB mode: voltage loop / current loop controller, which is superimposed on a known Cs DC voltage offset ^^^_^^_^^_^, to obtain the required comparison thresholds and after superimposing suitable slope compensation, a comparison is performed with the feedback voltage ^^^^_^, to finally control the switches of phase 1 and the switches of phase 2 of the bridge arms. This is equivalent to comparing ^^^_^^_^^_^_^, ^^^_^^_^^_^_^with the feedback voltage ^^^^after superimposing suitable slope compensation, when the Cr1 voltage is sampled directly. When switching from FB mode to HB mode, it is necessary to control the drop in the Cr1 voltage DC offset: At time t1, the initial values of the switch-off threshold control quantities for G11 of phase 1 and G22 of phase 2 are updated to HB_TH_H_C_S and HB_TH_L_C_S: Equation (21) (20), (21) and (22) are known quantities in FB mode before switching. In the case where the determined equivalent resonant tank duty cycle objective is 50%,^^^^^^^^_^^^^ = ^^^⁄ 4 is known.When the mode switch is complete, the output voltage / current loop controlleroutputs the control quantity ^^^_^^_^^_^_^_^ − ^^^_^^_^^_^_^_^ in a closed loop. Also,the comparison thresholds ^^^_^^_^^_^_^_^and ^^^_^^_^^_^_^_^for the Cs voltage are calculated according to Equations (18) and (19). By the same principle, when switching from HB mode to FB mode, the increase ^^^^^^^^_^^^^in the Cr1 voltage DC offset needs to be controlled. At time t0, the switch-off threshold for G12 of phase 1 and G21 of phase 2 is updated to FB_TH_L_S:^ = ^ −^^^_^^_^^_^_^^^^^_^^_^^_^_^^^_^^_^^_^_^ ^^_^^_^^_^^= ^^^_^^_^^_^ + ^^^^^^^^_^^^^ − At time t1, the initial values of the switch-off threshold control quantities for G11 of phase 1 and G22 of phase 2 are updated to FB_TH_H_C_S and FB_TH_L_C_S: (23), (24) and (25) are known quantities in HB mode before switching. In the case where the determined equivalent resonant tank duty cycle objective is 50%,^^^^^^^^_^^^^ = ^^^⁄ 4 is known.When the mode switch is complete, the output voltage / current loop controlleroutputs the control quantity ^^^_^^_^^_^_^_^ − ^^^_^^_^^_^_^_^ in a closed loop. Also,the comparison thresholds ^^^_^^_^^_^_^_^and ^^^_^^_^^_^_^_^for the Cs voltage are calculated according to Equations (16) and (17). It should be pointed out that in order to avoid excessive redundancy of explanation, physical quantities of voltage and current on an actual resonant capacitor are used directly for explanation in the description of the control method above, and voltage and current feedback coefficients are not involved; in actual applications, detailed design should be carried out appropriately according to requirements. The control method described above is also applicable to parallel-connected full- bridge LLC topologies and other resonant converter topologies, the difference being that the resonant capacitor DC offset voltage will be different; this is not detailed further here. The control method for a resonant converter of the present invention has a significant advantage over existing control methods, specifically high dynamic performance. Through the use of closed-loop feedback control of resonant tank state variables, the present invention enables more precise voltage and current control during topology restructuring, significantly reducing overregulation of output voltage and current. This control strategy not only improves the stability and reliability of the system, but also allows online hot restructuring of the system without downtime, thus improving the power supply continuity and efficiency of the system. In addition, the control method of the present invention simplifies the control algorithm, reduces the precision requirements for device parameters, and makes system design simpler and more convenient, with a lower cost. The control method for the resonant converter of the present invention has higher flexibility and adaptability than existing control methods, while also achieving a wide voltage gain range. The invention enables smooth switching between different operating modes through precise control of the switching state of each phase bridge arm, thus achieving a wider voltage gain range. In addition, the control method of the present invention also takes into account differences in DC offset voltage on the resonant capacitor, ensuring a smooth transition when switching between different modes through dynamic adjustment of switching frequency and duty cycle, and avoiding current surges and upward or downward surges in output voltage / current. This closed-loop control strategy based on state quantity feedback makes the control method of the present invention relatively insensitive to error in resonant tank device parameters, thus improving the robustness of the system and its ability to adapt to different load requirements. Through this control method, the present invention provides a solid technical foundation for realizing DCDC converters having high efficiency and high power density, and is particularly suitable for application fields with stringent requirements regarding a wide voltage gain range and high dynamic performance, such as electric vehicles, photovoltaic energy storage systems, etc. Those skilled in the art will understand that the steps of the method according to the invention are not limited to execution in the sequence set out above. In addition, in the present invention, terms such as "include" and "comprise" mean that besides having steps that are directly and explicitly stated in the Description and claims, the technical solution of the present application does not rule out having other steps which are not directly or explicitly stated. Although the present invention has been disclosed above through preferred embodiments, the present invention is not limited to this. Various changes and modifications made by a person skilled in the art without departing from the spirit and scope of the present invention should be included in the scope of protection thereof. Thus, the scope of protection of the present invention shall be the scope defined by the claims.

Claims

Claims 1. A control method for a resonant converter, a primary side of the resonant converter comprising a full-bridge conversion circuit and a resonant circuit, the full- bridge conversion circuit being formed of two phase bridge arms combined in a parallel-connected or series-connected manner, and the resonant circuit consisting of a magnetizing inductance, a resonant capacitor and a resonant inductor, and the method being used to control switching of the resonant converter from a present first operating mode to a second operating mode, wherein the control method comprises the following steps:determining a first switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^] and a firstswitching frequency ^^_^^of the resonant converter in the present first operating mode; measuring an input voltage ^^^and an output voltage ^^^^of the resonantconverter, to determine a present voltage gain ^^^^ of the resonant converter;based on a load ^^^^ and the present voltage gain ^^^^ of the resonantconverter, determining a second switching frequency ^^_^^of the resonant converter in the second operating mode to be switched to;based on the first switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^], the firstswitching frequency ^^_^^, a capacitance ^^^of the resonant capacitor and the second switching frequency^^_^^, calculating a second switch-off threshold rangeof the resonant converter in the second operating mode;and acquiring voltage feedback VCr1 across the resonant capacitor in real time, wherein, when the voltage feedback VCr1 exceeds the first switch-off thresholdrange [^^^_^^_^^_^ , ^^^_^^_^^_^], a switch-off threshold of the resonant converter isswitched to the second switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^], andthe resonant converter is switched from the first operating mode to the secondoperating mode by enabling corresponding switches in the two phase bridge arms in the full-bridge conversion circuit.

2. The control method for a resonant converter as claimed in claim 1, wherein the full-bridge conversion circuit is formed of the two phase bridge arms combined in a series-connected manner, and input voltages ^^^^^and ^^^^^of the two phase bridge arms are each half of the input voltage ^^^of the resonant converter.

3. The control method for a resonant converter as claimed in claim 2, wherein the method further comprises: when calculating the second switch-off threshold range[^^^_^^_^^_^, ^^^_^^_^^_^], ensuring that an output power ^^^_^^ and a voltage gainof the resonant converter in the first operating mode are the same as an output power ^^^_^^and a voltage gain of the resonant converter in the second operating mode.

4. The control method for a resonant converter as claimed in any one of claims 1 to 3, wherein the first operating mode is a full-bridge mode, and the second operating mode is a half-bridge mode.

5. The control method for a resonant converter as claimed in claim 4, wherein the method further comprises: taking into account a DC offset ^^^_^^of the resonant capacitor when calculating the second switch-off threshold range[^^^_^^_^^_^, ^^^_^^_^^_^].

6. The control method for a resonant converter as claimed in claim 5, wherein a DC offset ^^^_^^_^^of the resonant capacitor in the full-bridge mode is set to be 1 / 2 of an input voltage ^^^of the resonant converter, and a DC offset ^^^_^^_^^of the resonant capacitor in the half-bridge mode is set to be 1 / 4 of the input voltage ^^^of the resonant converter.

7. The control method for a resonant converter as claimed in claim 4, wherein, in the full-bridge mode, the two phase bridge arms are caused to both respond to thefirst switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^]; and in the half-bridgemode, a corresponding one of the two phase bridge arms is caused to respond tothe second switch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^] each time.

8. The control method for a resonant converter as claimed in any one of claims 1 - 3, wherein a slope compensation value is added when calculating the secondswitch-off threshold range [^^^_^^_^^_^ , ^^^_^^_^^_^], the slope compensation valuebeing determined on the basis of a given harmonic compensation gradient ^^^^^^and the second switching frequency ^^_^^.

9. The control method for a resonant converter as claimed in any one of claims 1 - 3, wherein the second switching frequency ^^_^^is determined on the basis of a predefined lookup table or mathematical model.

10. The control method for a resonant converter as claimed in any one of claims 1- 3, wherein the voltage feedback VCr1 across the resonant capacitor is directlysampled by a voltage sensor, or calculated from a resonant capacitor current ICr1 sampled with the aid of a current sensor.

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