MIMO radar system with non-coherent TX integration and transmitters of differing strengths

Abstract

The invention relates to a method for a radar system and to a radar system for detecting surroundings, comprising: transmission means having a plurality of transmitting antennas operating in parallel for emitting transmission signals which contain one or more sequences of K individual signals; means for changing the phase position of the transmitted individual signals, via which a different phase progression for the transmitting antennas is realized via the K individual signals; receiving means having a plurality of receiving antennas for receiving transmission signals reflected at objects; and signal processing means for processing the received signal; wherein not all transmitting antennas emit the same average transmission level, a non-coherent integration is carried out via certain signal components (TX integration); and wherein this non-coherent TX integration is carried out multiple times under the assumption of different radial relative velocities of an object and, optionally, distances away at which an object is located, and in the non-coherent TX integration, signal components are weighted more heavily the higher the emitted transmission level of the associated transmitting antenna is, and the result of the non-coherent TX integration is used in order to identify the phase progression resulting from the object relative speed and, optionally, object distance, and / or to identify a variable derived therefrom.

Description

[0001] 202505405

[0002] 1

[0003] Description

[0004] Title of the invention

[0005] MIMO radar system with non-coherent TX integration and transmitters of varying power levels

[0006] The invention relates to a radar method and radar system for use in driver assistance systems in motor vehicles. The radar system has several transmitting and receiving antennas operating in parallel, which is referred to as MIMO (Multiple-Input Multiple-Output), and has different transmission levels on the transmitting antennas, which, according to the invention, is used for object identification in the case of non-coherent TX integration.

[0007] State of the art

[0008] Motor vehicles are increasingly equipped with driver assistance systems that use sensors to perceive the surroundings and derive automatic vehicle reactions from the detected traffic situation and / or instruct the driver, in particular by issuing warnings. A distinction is made between comfort and safety functions.

[0009] In current vehicle development, FSRA (Full Speed ​​Range Adaptive Cruise Control) plays an important role as a comfort feature. The vehicle regulates its own speed to the driver's desired speed, provided the traffic situation allows it; otherwise, the vehicle's speed is automatically adjusted to the traffic situation.

[0010] Safety features now come in a wide variety of forms. One group consists of functions for reducing braking or stopping distance in emergency situations, up to and including autonomous emergency braking. Another group comprises lane change functions: These warn the driver or intervene in the steering if the driver intends to make a dangerous lane change, i.e., if a vehicle in the adjacent lane is in the blind spot (designated as 202505405).

[0011] 2

[0012] BSD - “Blind Spot Detection” - or approaches quickly from behind (LCA - “Lane Change Assist”).

[0013] Nowadays, the driver is no longer just assisted, but the driver's task is increasingly performed autonomously by the vehicle, i.e., the driver is increasingly replaced; this is referred to as autonomous driving.

[0014] Radar sensors are used for systems of the type described above, often in combination with sensors from other technologies, such as camera sensors. Radar sensors have the advantage, among others, of operating reliably even in poor weather conditions and being able to directly measure not only the distance to objects but also their radial relative velocity via the Doppler effect. Currently, 77 GHz and 79 GHz are typically used as transmission frequencies.

[0015] The functions mentioned above require high detection quality, for which precise angle determination is essential. Therefore, MIMO radars are increasingly being used. These radars have multiple fully parallel transmitting and receiving antennas and utilize all possible combinations of transmitting and receiving antennas to achieve the best possible angle determination. For parallel operation of the transmitting antennas, their transmitted signals must be modulated differently so that the components they cause in the received signals can be separated. A commonly used approach employs a sequence of identical or similar transmitting signals, over which the phase is linearly changed, with the modulation rate—that is, the rate of change of the linear phase shift—varying across the transmitting antennas.To separate the signal components originating from the different transmitting antennas, the result of a discrete Fourier transform of the individual received signals is often used, since the different rates of linear phase change cause the components of the different transmit paths to lie at different frequency points. A challenge here is correctly assigning the power peaks occurring in this spectrum to the individual transmitting antennas (also known as TX demodulation). In WO 2018 / 137835 A1, it is proposed that, in addition to the phase modulation described, the transmitting antennas should be deliberately driven with different transmit levels.

[0016] 3

[0017] The system transmits signals, resulting in power peaks in the spectrum having different amplitudes; the assignment of power peaks to the transmitting antennas is based on the magnitude of the power peaks. However, this method has limited robustness – especially in situations with multiple objects whose power peaks overlap at least partially, and with small and / or distant objects, i.e., objects that result in a weak received signal.

[0018] Problem, solution and advantages of the invention

[0019] The object of the invention is to provide improved TX demodulation for automotive MIMO radar systems, which is also suitable for scenarios with multiple objects and objects with a low received signal.

[0020] This problem is fundamentally solved by a method for a radar system according to claim 1 and a corresponding radar system according to the dependent claim, wherein the radar system may also explicitly include the features described in the dependent claims. Advantageous embodiments of the invention are claimed in the dependent claims. The core idea is that integration is performed not only over the spectra of the received path signals, but also incoherent integration in the grids of the transmitted path signals expected in the spectra, wherein the weights used in the incoherent integration are based on the transmit level of the respective associated transmitting antenna.From the result, in which individual objects generate power peaks at several positions of an equidistant grid, the correct, i.e., actual frequency support points of objects are identified, in particular as the frequency support point where the maximum of the respective power peaks lies.

[0021] The advantages of the invention arise from the fact that high detection quality and sensitivity can be ensured, thereby potentially reducing hardware requirements. 202505405

[0022] 4

[0023] The radar system to which the inventive method for environmental detection relates comprises: 1) transmitting means with several parallel transmitting antennas for emitting signals, which contain one or more sequences of K individual signals, the general form of which is preferably the same or similar, wherein their frequency can change, in particular, linearly; 2) means for changing the phase of the transmitted individual signals, by which a phase response different for the transmitting antennas is achieved over the K individual signals, wherein this difference in the phase response for different transmitting antennas over the K individual signals is at least approximately linear, optionally excluding phase jumps due to the phase uniqueness range of 2π; 3) receiving means with one or more receiving antennas for receiving signals reflected from objects.wherein the signals received by an object are composed of components originating from different transmitting antennas, these components exhibiting a phase response that differs linearly across the K individual signals, as well as a similar phase progression, which is generated in particular by the relative motion of the object and, in the case of a frequency shift across the K individual signals, also depends on the object's distance, and 4) signal processing means for processing the received signals. The method is characterized in that 1) not all transmitting antennas radiate the same average transmit level, 2) a non-coherent integration is performed over signal components whose phase responses across the K individual signals lie within the expected grid of the signal components of an object originating from different transmitting antennas and exhibiting linearly different phase responses, which is hereinafter referred to as non-coherent TX integration.wherein this non-coherent TX integration is performed multiple times assuming different radial relative velocities and, if necessary, distances of an object, 3) in the non-coherent TX integration, signal components are weighted more heavily the higher the radiated transmit level of the respective assigned transmitting antenna is, and 4) the result of the multiple non-coherent TX integration is used to determine the value of the signal from the 202505405,

[0024] 5

[0025] to identify the resulting phase progression and / or a derived quantity based on object relative velocity and, if applicable, object distance.

[0026] Advantageously, the weighting of the signal components used in non-coherent TX integration correlates with the average transmit level of the respective assigned transmitting antenna, i.e., is proportional to the average transmit power or transmit amplitude of the respective assigned antenna, possibly minus a constant value, in order to advantageously achieve the most robust possible identification of the phase progression resulting from the object relative velocity and, if applicable, object distance, and / or a quantity derived therefrom, especially in the case that the signals contain contributions from multiple objects and / or significant noise.

[0027] Advantageously, prior to the non-coherent TX integration via the received signals to multiple receiving antennas or via signals derived from them, a coherent integration, in particular in the form of digital beamforming, hereinafter referred to as RX beamforming, can be carried out, and the result of the multiple non-coherent TX integration from different RX beamforming directions can be used to identify the phase progression resulting from the object relative velocity and, if applicable, object distance, and / or a quantity derived therefrom.

[0028] Furthermore, non-coherent integration can be performed over the received signals to multiple receiving antennas or over signals derived from them, which is expediently carried out before the non-coherent TX integration. The result of the entire non-coherent integration, performed multiple times under the assumption of different radial relative velocities and, if necessary, distances to an object, can then be used to identify the phase progression resulting from the object's relative velocity and, if applicable, its distance, and / or a quantity derived therefrom.

[0029] 6

[0030] Non-coherent TX integration can be implemented using power values. Preferably, it can be implemented using the magnitude values ​​of the signal components.

[0031] In an advantageous embodiment of the invention, the method can be characterized in that 1) the K individual signals preferably lie in an at least approximately equidistant grid, 2) a discrete Fourier transform of length L is performed on their K received signals or signals derived from them, optionally after padding with zeros, i.e., a so-called zero-padding, wherein the L frequency reference points of the discrete Fourier transform are hereinafter referred to as Doppler gates, 3) the phase response over the K individual signals includes a linear component, i.e., with a different slope, for each transmitting antenna, by which the signal components from the different transmitting antennas lie at different Doppler gates in the discrete Fourier transform.4) the non-coherent TX integration is performed via Doppler gates in the grid expected from the different slopes of the linear components, and 5) this non-coherent TX integration is performed for different positions of this grid, i.e., different, in particular all, Doppler gates as the first grid point and thus for different, in particular all possible, object relative velocities and, if applicable, object distances.

[0032] Furthermore, the linear components of the phase responses, when mapped to the phase uniqueness range of 0…2π, can have a common period P, which preferably represents an integer divisor of the number L of Doppler gates, whereby the Doppler gates corresponding to the transmitting antennas represent a subset of an equidistant grid with a distance L / P.

[0033] Advantageously, the summation necessary for non-coherent TX integration for preferably all P grid layers in an equidistant grid with integer spacing L / P can be realized by performing a cyclic correlation between a vector with the P values ​​of the discrete Fourier transform in this grid and an allocation vector of length P, wherein the values ​​of the allocation vector for all indices which 202505405

[0034] 7

[0035] for a transmitting antenna whose linear phase change within a period P corresponds to multiples of 2π, assumes non-zero values ​​and is otherwise 0, and wherein this cyclic correlation can preferably be carried out as a fast correlation, i.e. by multiplying two spectra in the frequency domain.

[0036] Advantageously, the non-zero values ​​of the occupancy vector correlate to the average transmission level of the respective assigned transmitting antenna, i.e., are in particular proportional to the average transmission power or transmission amplitude of the respective assigned transmitting antenna, possibly minus a constant value.

[0037] Furthermore, different transmission levels can be caused by hardware-related effects such as different lengths of the antenna leads via the transmitting antennas and / or can be generated specifically, in particular by a configurable transmission power generation, especially so that the period P can be chosen as small as possible, with the number of transmitting antennas representing the lower limit.

[0038] Advantageously, the results of the non-coherent TX integration of raster layers with spacings that are integer multiples of L / P are investigated and, in particular, compared to identify objects as well as their relative velocity and, if applicable, distance, preferably also using a detection threshold that depends in particular on the noise level, and, when using an RX beamforming, this is done in all its beam directions.

[0039] Advantageously, for object identification, the grid position in the grid with a distance L / P can be used where the incoherent TX integration has the highest amount and thus the highest performance.

[0040] Furthermore, for object identification, grid layers with a spacing of L / P can also be used where the performance of the non-coherent TX integration is not maximum, but is above a threshold which may depend on the distance of this grid layer to the grid layer of the maximum and in particular is at least a minimum distance above the value that is there 202505405

[0041] 8

[0042] is generated by an object that is located at the grid position of the maximum and generates the power level there.

[0043] If the transmitting antennas are located in different planes when viewed vertically, the transmission levels of the respective transmitting antennas can differ in at least some planes, preferably in such a way that the diversity of the transmission levels is similar in as many or all planes as possible, thereby reducing the influence of superposition effects caused by reflections on a road surface.

[0044] Furthermore, there can be several similar groups, in particular pairs of transmitting antennas, in each group where the different transmission levels of their antennas are at least approximately the same.

[0045] Furthermore, the individual transmitted signals can be linearly frequency-modulated, with their center frequency optionally changing successively and preferably linearly, or represent OFDM signals or be generated with pseudorandom fast phase modulation, in particular characterized in that a signal separation into gates corresponding to different distances, so-called distance gates, is realized from the received values ​​resulting for each individual transmitted signal by means of a transformation, and the procedures described in the claims above are applied in the different distance gates.

[0046] Brief description of the drawings

[0047] Fig. 1a shows an exemplary embodiment of a radar system; in Fig. 1b its arrangement of the 12 transmitting antennas and the 16 receiving antennas is shown in detail, and Fig. 1c shows the two-dimensional equidistant antenna array with 16x8 channels synthesized from eight of the transmitting antennas and all 16 receiving antennas.

[0048] Figure 2 shows the frequency modulation consisting of a sequence of frequency ramps. 202505405

[0049] 9

[0050] Fig. 3 shows, for a single object, the magnitude of the spectrum resulting after two-dimensional transformation to a transmit and a receive path in the object distance gate - in Fig. 3a without random phase modulation component, in Fig. 3b with random phase modulation component; in this case, inaccuracies occurring in real phase shifters are assumed, i.e., small phase and amplitude errors.

[0051] Fig. 4 shows the magnitude of the spectrum resulting after two-dimensional transformation for a receive path with all components of the 12 parallel transmit paths.

[0052] Fig. 5 shows the performance spectrum after incoherent integration over all reception paths.

[0053] Fig. 6 shows the power spectrum after non-coherent integration over all receive paths and all transmit paths, while still assuming only a single object.

[0054] Fig. 7 shows the power spectrum for the case of two objects after incoherent integration over all receive paths and all transmit paths.

[0055] Fig. 8 shows, for the now always considered case of two objects, the magnitude of the spectrum which results from the spectrum after two-dimensional transformation by beam shaping via the receive paths (RX beam shaping).

[0056] Figure 9 shows the power spectrum resulting after RX beamforming and incoherent integration over all transmit path components.

[0057] Fig. 10 shows the magnitude of the beamformer spectrum in the Doppler gate of the first object, which results from two-dimensional digital beamforming via antenna channels formed from all transmit and receive path signals according to Fig. 1c.

[0058] Fig. 11 shows the power of the maximum of the two-dimensional beamformer spectrum formed per Doppler port over all beam directions.

[0059] Example 202505405

[0060] 10

[0061] Consider the exemplary embodiment of a radar system according to Fig. 1a, whose antenna 1.1 has an antenna arrangement according to Fig. 1b. The radar system has M TX = 12 transmitting antennas TX0-11 for broadcasting signals and M RX= 16 receiving antennas RX0-RX15 for receiving transmitted signals reflected from objects. All antennas (transmitting and receiving) have the same beam pattern in elevation and azimuth. Four transmitting antennas TXO-3 and four transmitting antennas TX4-7 each form vertical arrays; the vertical spacing of the antennas (and thus their phase centers, i.e., radiation centers) is 5λ, where λ = c / 76.5GHz = 3.92mm is the mean wavelength of the transmitted signals in the used frequency band of 76-77GHz and c = 3*10 8m / s is the speed of light. These two arrays, each with four transmitting antennas, are located at the left (TXO-3) and right (TX4-7) edges of the antenna array and have a horizontal spacing of 24 λ. Additionally, there are two more transmitting antennas each on the left inner side (TX8-9) and right inner side (TX10-11), which are 3.5 λ and 2.5 λ away from the transmitting antennas at the respective edges of the antenna array, respectively, and have a vertical spacing of 5.5 λ between each other. The 16 receiving antennas are divided into two horizontal arrays (RXO-7 and RX8-15), each with eight equidistant antennas, located at the top and bottom of the antenna array, respectively, and have a vertical spacing of 20 λ; the horizontal spacing between the antennas is 3 λ.All combinations of the eight transmitting antennas on the right and left (TXO-3 and TX4-7) and the 16 receiving antennas at the bottom and top (RXO-7 and RX8-15) synthesize a two-dimensional equidistant array with 16 x 8 = 128 antenna channels in a horizontal grid of 3λ and a vertical grid of 5λ (see Fig. 1c), which is used to determine the azimuth and elevation angles of objects. Since these grids represent multiples of half the wavelength λ / 2, ambiguities arise in the angular range -90° to 90°. To resolve azimuth ambiguities, the antenna channels from the transmitting antennas TX1, TX8, TX10 and TX5 and all receiving antennas are used; to resolve elevation ambiguities, the antenna channels from the transmitting antennas TX8-TX11 and all receiving antennas are used.

[0062] As shown in Fig. 1a, the transmitted signals emitted by the transmitting antennas are derived from the high-frequency oscillator 1.2 in the 76-77 GHz range 202505405

[0063] 11

[0064] The oscillator's frequency can be modulated via a control voltage vcontrol. This control voltage is generated by the control elements 1.8, which include, for example, a phase-locked loop or a digital-to-analog converter. These elements are controlled such that the oscillator's frequency response corresponds to the desired frequency modulation. The phase of the transmitted signals can be individually set and varied for the MTX = 12 transmitting antennas using phase shifters 1.3. These phase shifters provide 64 at least approximately uniformly distributed phase values ​​over the phase uniqueness range O...2TT. These phase shifters modulate the transmitted signals from the different transmitting antennas differently, enabling parallel transmission on all transmitting antennas, i.e., MIMO operation. This is because, after demodulation in the received signals, the components originating from the different transmitting antennas can be separated.The signals received by the MRX = 16 receiving antennas are simultaneously down-converted into the low-frequency range in the real-valued mixers 1.4, along with the signal from the oscillator 1.2. The received signals then pass through the bandpass filters 1.5 with the transfer function shown, the amplifiers 1.6, and the analog-to-digital converters 1.7. Finally, they are further processed in the digital signal processing unit 1.9.

[0065] In order to measure the distance to objects, the frequency of the high-frequency oscillator, and thus the transmit signals, is changed linearly very quickly (from Tch = 51.2 ps to Bch = 600 MHz, where the center frequency f) – as shown in Fig. 2 – is fixed (Tch = 51.2 ps to Bch = 600 MHz). c= 76.5 GHz); this is referred to as a frequency ramp (often also called a "chirp"). The frequency ramps are repeated periodically in a fixed grid TD = 70 ps; in total there are K = 512 frequency ramps, all of which have the same frequency profile, i.e., the same frequency slope, the same frequency position (i.e., in particular, the same start and center frequencies), and the same duration. In recent years, this type of modulation has become increasingly widespread and established in radar systems for the environmental detection of vehicles. It allows for a long sensor range and speed resolution (due to long data acquisition time) as well as a high distance resolution (due to the use of a high modulation bandwidth). 202505405

[0066] 12

[0067] During each frequency ramp k=0,..., K-1, the received signals are sampled by each of the MRX = 16 A / D converters 1 = 2048 times at intervals of Ts = 25 ns (i.e., at 40 MHz), with sampling always starting at the same time relative to the start of the ramp (see Fig. 2). The resulting digital samples in the receive path IYIRX, with index i=0,..., 1-1, are denoted by s(i,k, IYIRX). Signal sampling is only meaningful in the time range where received signals from objects within the distance range of interest arrive. Therefore, after the ramp starts, at least the propagation delay corresponding to the maximum distance of interest must be waited for (for a maximum distance of interest of 200 m, this corresponds to 1.33 ps). It should be noted that here and in the following, "distance" always refers to the radial distance, and "relative velocity" to its radial component.

[0068] As is known from the prior art (see e.g. EP 2629113 B1) and can also be easily derived, the sampling signal s(i,k, IYITX, IYIRX) caused by the transmit signal of a transmitting antenna IYITX represents, in the case of a single point-like object at a distance d, a sinusoidal oscillation over the index i, which can be described to a very good approximation as follows:

[0069] s(i,k, IYITX, IYIRX) = A-sin[2iT-i / l-jo + <p v (k) + cppM, Tx(k, IYITX) + (päjx(mTx) + <pä, Rx(mRx)] (1)

[0070] with jo = d / (Meter)-Bch / 150MHz, (2) i.e. the frequency of the oscillation is proportional to the object distance d (jo is a normalized frequency). A radial relative movement of the object to the sensor causes a phase shift contribution cp that changes over the K = 512 frequency ramps. v (k) of the sinusoidal oscillation; for a motion with a constant radial velocity component v, the following results:

[0071] CP v(k) = 2TT-k / K-lo with Io = 2KT D vf c / c, (3) dh, a linear phase shift over the frequency ramps k, where the rate of change of the phase is proportional to the radial relative velocity v of the object. The phase contribution cppM, Tx(k, IYITX) in reference (1) above describes the phase modulation (also called TX modulation) realized with the phase shifters 1.3: 202505405

[0072] 13

[0073] (ppM, Tx(k,rriTx) = <ppM,iin(k, rriTx) + cppM,r(k), (4a) welche zwei Anteile aufweist: Zum einen den sich linear über die Frequenzrampen k ändernden Anteil

[0074] (ppM,iin(k,rriTx) = 2iT-kp(mTx) / P (4b) with different rate of change, i.e., modulation speed p(rriTx) / P, across the transmit paths with integer common period P and normalized modulation speeds P(I ITX) (normalized with respect to the slowest modulation speed 1 / P), which are considered here as integers; this component serves to later separate the components originating from the different transmit paths in the received signals (which will be explained in detail later). Secondly, a random or pseudorandom component (ppM,r(k)) across the frequency ramps k, which is identical for all transmit paths and which is generated by a random selection for each frequency ramp k from the 64 phase values ​​of the phase shifters 1.3; this component serves to suppress unwanted effects, which will be derived and explained later. The two trailing phase components (pä x(mTx) and (pä.Rx(mRx) represent the phase angles for the various transmit and receive paths, dependent on the azimuth angle OAZ and elevation angle OEI of the object. It should be noted that real-world phase values ​​lie in the range O...2TT, since, due to the cyclic nature of phases, all phase values ​​can be mapped into this range, which mathematically represents a modulo functionality.

[0075] Finally, with regard to reference (1), it should be mentioned that the amplitude A of the received signal is assumed to be independent of the transmit and receive paths, i.e., all transmit and receive paths are assumed to be of equal strength; this assumption has no influence on the following considerations.

[0076] In the digital signal processing unit 1.9, for the received signals s(i,k,rr)Rx) per frequency ramp k and receive path I IRX, a first discrete Fourier transform (DFT = Discrete Fourier Transform) over the time index i=0,..., 1-1 is performed after multiplication with a suitable window function wi(i), since this corresponds to optimal filtering for the signal shape according to reference (1); the DFT is expediently implemented with a fast Fourier transform (FFT = Fast Fourier Transform). (See also 202505405)

[0077] 14

[0078] sin(x) = (exp(jx) - exp(-jx)) / (2j), (5) where “exp” denotes the exponential function and j is the imaginary unit, the DFT, i.e. the spectrum Si(j,k,mTx,mRx) of the sampled signal s(i,k,mTx,rriRx) caused by a transmit path, is given by:

[0079] Si(j,k,mTx,rriRx) = A / (2j) •

[0080] [exp(j( Pv(k) + (ppM, Tx(k,rriTx) + (pä, Tx(mi ) + <pä, Rx(mRx))) • Wi(modj(j-jo)) - exp(-j(cpv(k) + (ppM, Tx(k,rriTx) + (pä x(mTx)y + <pä, Rx(mRx))) • Wi(modj(j+jo))], (6)

[0081] where j = 0,..., 1-1 is the sweep variable for the image domain, i.e., the frequency domain of the DFT, and represents the so-called distance gates (because the frequency of the received signal is proportional to the distance), Wi(j) is the spectrum of the window function wi(i) used, and "modj" represents the modulo function of the modulo J. The spectrum Wi(j) of the window function has a rather sharp power peak at j = 0, which extends over approximately three frequency values ​​j. According to equation (6), the spectrum Si(j,k,mTx,rriRx) caused by a single object therefore exhibits two power peaks at the frequencies jo and l-jo (assuming 0 < jo ^1, which is valid due to non-negative distances and the effect of the bandpass filters 1.5).If 0 < jo 1 / 2, then the power peak at l-jo, i.e. in the upper half of the spectrum, carries no additional information, which generally applies to the upper half of the spectrum because, due to the real-valued input signal, it is complexly inverted to the lower half; therefore, for further processing, only the lower half of the spectrum is considered, i.e., only the frequencies or distance gates j=0...1 / 2.

[0082] Before a second DFT over dimension k can be performed, the random phase component cppM.r(k) of the phase modulation (ppM, Tx(k,rriTx)) must be compensated according to equation (4a) (the remaining phase components represent a linear phase response over dimension k, which is a prerequisite for applying the DFT as optimal filtering). For the distance gate range of interest jo = 0.. J / 2 (i.e., objects located there), the first term in equation (6) is relevant (since only the lower half of the spectrum j = 0.. J / 2 is considered), thus 202505405

[0083] 15

[0084] that to compensate for the random phase modulation component with exp(-j- <ppM,r(k)) zu multiplizieren ist; dann ergibt sich:

[0085] Si,comp(j,k,mTx,rriRx) = A / (2j)- (7) [exp(j( Pv(k) + (ppM,iin(k,mTx) + (pä x(mTx) + <pä, Rx(mRx))) • Wi(modj(j-jo))

[0086] - exp(-j(cpv(k) + (ppM,iin(k,rriTx) + 2cppM,r(k) + (pä x(mTx) + <pä, Rx(mRx))) • Wi(modj(j+jo))

[0087] For each distance gate j and receive path I IRX, and after multiplication by a window function W2(k), a second DFT is performed, now over the frequency ramp index k (preferably again via an FFT); this yields the two-dimensional spectrum.

[0088] S2(j,l,mTx,mRx) = A / (2j)- [exp(j( Pä, Tx(mTx) + <pä, Rx(mRx))) • Wi2(modj(j-jo),modK(l-lo-p(m-rx) K / P))

[0089] - RpM(modK(l+lo+p(mTx) K / P)) • exp(-j( <pa, Tx(mTx) + <pä, Rx(mRx))) • Wi(modj(j+jo))], (8)

[0090] where l=0,..., K-1 is the running variable for the image domain, i.e., the frequency domain of the second DFT, and represents the so-called Doppler gates (because frequency is proportional to relative velocity, apart from the additional phase modulation component p(rriTx) K / P), Wi2(j,l) is the two-dimensional spectrum of the two-dimensional window function wi(i)-W2(k) used (has a power peak at j = 0 and I = 0), and RPM(I) represents the spectrum of the unit vector exp(-j2cppM,r(k)) exhibiting a random phase response (2cppM,r(k) randomly assumes values ​​from 32 different phase values ​​uniformly distributed over 2TT - the modulo property of the phase is already taken into account); thus, RPM(I) itself is also noise, which is Rayleigh-distributed. For an object in the distance gate range of interest jo=O... J / 2, the following results in the two-dimensional spectrum S2Ü,l,nriTx,mRx) according to the designation.(8) of the first term a power peak at distance gate j = jo and Doppler gate I = modK(lo+p(rriTx) K / P); the second term generates at j = -jo (and its immediate neighborhood) a power peak over all 202505405.

[0091] 16

[0092] Doppler gates k distributed noise, where this noise j lies outside the distance gate range of interest and consideration j=0... J / 2.

[0093] Now, an object above the distance gate range of interest is considered, i.e., in the range jo = (J / 2 + 1)... (1 - 1). Due to the relatively low attenuation of the transition range of the bandpass filters 1.5, objects can be received particularly well above jo = 1 / 2 – this is referred to as overreach. Then, in the two-dimensional spectrum S2Ü,l,nr)Tx,mRx), according to equation (8), the first term produces a power peak at j = jo above the considered distance gate range j = 0...J / 2, while the second term generates noise distributed across all Doppler gates k at the distance gate j = -jo in the considered distance gate range j = 0...J / 2. However, this noise will not lead to detection even if it is significantly above the system noise, since detections are only made for power peaks in dimension k that are significantly above the overall noise level there.Would it include the random phase modulation component? <ppM,r(k) nicht geben, so würde das zweidimensionale Spektrum S2Ü,l,nriTx,mRx) wie folgt lauten (ist aus Bez. (7) und (8) zu ersehen): S2(j,l,mTx,rriRx) = A / (2j)- [exp(j(cpä, Tx(mTx) + <pä, Rx(mRx))) • Wi2(modj(j-jo),modK(l-lo-p(mTx) K / P)) - exp(-j((pä, Tx(rriTx) + <pä, Rx(mRx))) • Wi2(modj(j+jo),modK(l+lo+p(mTx) K / P))]. (9).

[0094] This would mean that an overrange, i.e. an object in the range jo=(J / 2+1 )... (1-1 ) would now lead to a power peak at the distance gate j = -jo in the distance gate range of interest and consideration j=0... J / 2 due to the second term; thus, a detection would be erroneously formed which not only has an incorrect (too small) distance, but whose measured values ​​for relative velocity and angle are also incorrect (because of incorrect signs of the contributing quantities in the second term of reference (9)).

[0095] The random phase modulation component cppM.r(k) thus prevents false detections due to overreach; the noise generated instead has an average power of approximately 26dB below the power peak, which is 202505405

[0096] 17

[0097] without a random component (the 26 dB result from the DFT integration gain of 10 logio(K = 512) = 27 dB minus approximately 1 dB window loss). It should also be mentioned that the above considerations do not apply to a binary phase shifter, i.e., one with only the two states 0 and TT; because then the phase component 2cppM,r(k) in the second term of equation (7) is always 0 (phase 2TT corresponds to phase 0) and therefore has no effect, so that the second term of the two-dimensional spectrum S2Ü,l,nriTx,mRx) does not represent noise, but according to equation (9) also generates a power peak, which then leads to the undesired long-range effect. Therefore, at least three different phase values ​​are necessary to convert long-range effects into noise.

[0098] Problems caused by over-propagation arise from a convolution effect, i.e., the mapping of frequencies onto other frequencies. When frequencies retain certain power components at other frequencies, this is referred to as convolution – such effects are collectively known as spectral convolution effects. This convolution effect occurs with very close objects; for example, consider an object at distance gate jo = 0.5 (it should be noted that signals from such close objects can be received despite the high attenuation of the 1.5 bandpass filters, as the short distance results in a very strong signal at the receiving antennas). Without the random phase modulation component cppM,r(k), the two-dimensional spectrum would then be S2Ü,l,nr)Tx,mRx) according to the formula.(9) At distance gate j = 0, components of both terms are effective (the power peak with the shape of the two-dimensional window spectrum W12 has a certain width and typically extends over 3 distance gates); the first term provides the correct information, while the second term represents false information. These erroneously occurring components of the second term can either lead to a distortion of the measured values ​​of the object (if they overlap with the real components of the first term and then, for example, influence the distance interpolation) or to non-real detections, i.e., ghost detections (if they do not overlap with the real components of the first term and thus form independent power peaks). Due to the random phase modulation component cppM,r(k), the second term of the two-dimensional spectrum S2Ü,l,nriTx,mRx) 202505405.

[0099] 18

[0100] According to reference (8), noise is present, so that no ghost detections occur; and because the noise is far below the actual power peaks of the first term, the influence on the measured values ​​of the object (distance, relative velocity and angle) is also negligibly small.

[0101] Negative effects of spectral convolutions are therefore avoided by the random phase modulation component cppM,r(k).

[0102] A fundamental problem with phase modulation is that the means used for this purpose, especially phase shifters, are never ideal and therefore always exhibit certain errors. The 64 phase values ​​of the phase shifters considered here, which ideally are uniformly distributed over the phase range O...2TT = 0...360 0The phases should have a standard deviation of 5°; and additionally, the amplitude of the realized pointers should also have a standard deviation of 10%. The modulation period used should be P = 32, and for the considered transmission path IYITX, the modulation rate should be p(rriTx,o) = 8. Without a random modulation component, the four target phase values ​​0, 90°, 180°, and 270° are then to be repeated periodically according to equation (4b), which are generated with the phase shifter indices 0, 16, 32, and 48; these should have actual phase values ​​of -5°, 91°, 185°, and 263° and amplitude values ​​of 1.02, 1.1, 0.85, and 1.11, which deviate from the target values. For an object with a distance gate < 1 / 2 and a Doppler gate Io = 168, the resulting two-dimensional spectrum in the object distance gate and a transmit path I ITX. O and a receive path IYIRX. O, i.e. S2(jo,l, IYITX. O, IYIRX. O), is shown in Fig.Figure 3a shows the power output in dB; in addition to the regular power peak at I = I0+8K / P = 296, further power peaks occur at intervals of 8 K / P = 128 (harmonic frequencies with a period of P / 8 = 4). These are generated by the non-ideal phase shift values ​​and their periodic repetition, and can lead to false detections, i.e., ghost detections. If the harmonics coincide with the correct power peaks of other transmitting antennas, they can distort the angle calculation. It should also be noted that in Figure 3a, the noise, which lies significantly below the power peaks, originates from the system noise and is not represented in the formulas above. 202505405.

[0103] 19

[0104] When using the random superimposed phase component cppM,r(k) of the phase modulation cppM,Tx(k, IYITX, o) according to equation (4a), four phase values ​​are no longer repeated periodically, but rather quasi-random phase values ​​are used (the linear phase component is no longer visible); thus, the phase modulation errors (phase and amplitude) are also no longer periodic, but quasi-random, so that in the spectrum S2Üo,l,mTx,o,mRx.o) according to Fig. 3b, no power peaks of harmonics occur – their energy is distributed in noise. The superimposed random phase modulation component cppM,r(k) therefore avoids ghost detections due to the always occurring inaccuracies in the phase modulation means and thus also allows the use of rather poor phase shifters, which can lead to a reduction in costs.

[0105] In addition to the advantages already presented through the random phase modulation component cppM,r(k), this also causes emissions via the receiving antennas (due to their limited TX isolation), internal couplings between transmit and receive paths, and interference from other radar systems to be decorrelated and thus converted into noise - they therefore cannot significantly degrade the measurement quality of objects and cannot generate ghost detections.

[0106] So far, only the contribution from one transmission path IYITX has been considered. The total two-dimensional spectrum S2Ü,l,mRx) generated by all MTX = 12 transmission paths is the sum of the individual contributions represented in the formulas above (i.e., the sum over I ITX = 0,..., MTX-1). Because the random phase modulation component <ppM,r(k) konstant über alle MTX = 12 Sendepfade ist, braucht seine Kompensation und die zweite DFT nur einmal für alle Sendepfade gerechnet werden, also nicht für jeden separat (letzteres wäre nötig, wenn der zufällige Phasenmodulationsanteil zwischen den Sendepfaden unterschiedlich wäre, was in deutlich erhöhtem Rechenaufwand resultieren würde).The modulation used to differentiate the transmission paths, i.e., the different linear phase response (ppM, iin(k, I ITX)) between the transmission paths, does not require multiple second DFTs, as it does not need to be compensated before the DFT because, without compensation in the DFT, it simply leads to a corresponding shift of the power peaks. After this second DFT, performed jointly for all transmission paths (i.e., only once), MTX = 12 power peaks result at the transmission paths 202505405.

[0107] 20

[0108] (i.e., their linear phase response) corresponding positions; this allows the separation of the components caused by the different transmission paths, which is necessary for MIMO operation (i.e., using all combinations of transmit and receive antennas for angle formation when the transmit antennas are operating in parallel) – this will be explained in detail later. The advantages and effects of the random phase modulation component cppM,r(k) described above naturally remain in the overall spectrum of all MTX = 12 transmission paths (they apply to each individual transmission path and thus also to their sum).

[0109] So far, only a single point-like object has been considered. The above considerations remain valid even in the case of multiple and / or extended objects (since this simply involves a linear superposition of several individual signals). Up to this point, a real-valued mixer has been considered. In the case of a complex-valued mixer (also known as an IQ mixer), there is ideally only one power peak; however, in reality, the IQ generation is not entirely perfect, so there is also a smaller power peak at a negative frequency, which can lead to ghost detections or distortion of measured values. The random phase modulation component cppM,r(k) also converts this power peak into noise, thus preventing its negative effects.

[0110] Now, for the radar system shown in Fig. 1, it will be described how, in MIMO operation (i.e., using all combinations of transmitting and receiving antennas for angle formation with parallel operation of the transmitting antennas), the separation of the components caused by the different transmission paths can be realized. After the second DFT (i.e., after two-dimensional transformation), performed jointly for all transmission paths (i.e., only once), the spectrum S2G, I, I IRX) in the receive path IYIRX is obtained in the case of a single point-like object with a distance gate jo < / 2 and a Doppler gate Io as the sum of the partial spectra belonging to the individual transmission paths IYITX (§20,1, IYITX, IYIRX) according to reference (8):

[0111] S20,l, IYIRX) = A / (2j)- summTx[exp(j((p&, Tx(rriTx) + <pä, Rx(mRx))) • Wi2(modj(j-jo),modK(l-lo-p(mTx) K / P))] mit I ITX = 0,..., MTX-1, (10) 202505405

[0112] 21

[0113] where “surrimTx” means the sum over all IYITX = 0,..., MTX-1; the second term in reference (8), which generates noise distributed over all Doppler gates I at j = j-jo (and its immediate neighborhood), is omitted here because this j lies outside the distance gate range of interest and consideration j=0... J / 2.

[0114] This spectrum (S2G,l,rriRx) exhibits power peaks at the distance gate jo of the object MTX = 12 at the positions corresponding to the transmit paths (i.e., their linear phase response) and is shown in Fig. 4 as an example, expressed in dB and scaled for a receive path IYIRX. The scale is such that the power peaks would be exactly at OdB without noise. The object is located at the Doppler gate Io = 168, and the normalized modulation speeds P(I ITX) used for the MTX = 12 signal paths are...

[0115] p(m Tx = 0,...,11) = [0 4 6 8 10 14 17 20 22 25 27 29] (11) and the received signal exhibits system noise, which results in a noise floor in the spectrum that is clearly below the power peaks. For each of the MRX = 16 receive paths, the power peaks have the same position and the same shape. The phases of the complex values ​​at the line peaks vary across the associated transmit and receive paths - they depend on the azimuth and elevation angle of the object.

[0116] The positions resulting from section (10), i.e., double goals

[0117] lo(mTx) = modulo + p(mTx) K / P), IYITX = 0,..., MTX-1, (12) the power peaks depend on the generally unknown and to be determined Doppler gate Io of the object. In order to determine this Doppler gate Io of the object and the assignment of the power peaks to the transmit paths IYITX ZU, a non-coherent integration is first carried out according to the state of the art over the spectra S2(j,l,rriRx) resulting in the MRX = 16 receive paths according to Ref. (10):

[0118] PRX-NC(], I) = summRx[|S2(j,l,mRx) | 2 ] (13a)

[0119] = |A / 2| 2 • summRx[|summTx[exp(j((pä, Tx(rnTx)+(pä, Rx(rriRx))) •

[0120] Wi2(modj(j-jo),modK(l-lo-p(m T x) K / P))]| 2 ] 202505405

[0121] 22

[0122] m with IYIRX = 0,..., MTX-1, (13b) where, as is generally customary, the non-coherent integration is formed as the sum of the powers. The power spectrum PRX-NCÜO, I) resulting in the object distance gate jo is shown as an example in Fig. 5 in dB and scaled. As can also be seen by comparing Fig. 4 and Fig. 5, the non-coherent integration significantly reduces the variance within the noise floor; this allows for better detection of small and / or distant objects whose power peaks lie within or only slightly above the noise floor.To identify the object Doppler gate Io and correctly assign the power peaks to the transmit paths IYITX, a pattern comparison can be performed between the expected grid P(IYITX) K / P of the power peaks and the power peaks occurring in the measured power spectrum PRX-NCÜO, I); the pattern comparison is possible because the modulation speeds P(I ITX) are chosen such that they have a non-periodic population with respect to the modulation period P = 32: for each q = 0,..., P-1: {modp(q+p(0,..., MTX-1 ))}^ {modp(p(0,..., MTx-1))}, (14) where “{.}” denotes the set of the respective MTX values.

[0123] One approach according to the invention is to non-coherently integrate over the values ​​of PRX-NCGJ) in the expected grid I = IO(IYITX) of the power peaks for each possible Doppler gate Io = 0,..., K-1; with PRX-NCGJ) according to reference (13) and IO(IYITX) according to reference (12) the following results:

[0124] PRX-NC, TX-Nc(j, I) = SUmmTx[PRX-Nc(j,mOdK(l+p(m. Tx) K / P))] (15a)

[0125] = |A / 2| 2 • sum m Tx[summRx[|summTx[exp(j((pä, Tx(mTx)+(pä, Rx(rriRx))) •

[0126] Wi2(modj(j-jo),modK(l+p(m T x) K / P-lo-p(m T x) K / P))| 2 ]]

[0127] with mix = 0,..., MTX-1 and I ITX = 0,..., MTX-1; (15b) where the loop variable I is again used for the possible Doppler gates Io = 0,..., K-1, and since summation over the transmitting antennas is performed twice in reference (15b), two different loop variables rrn-x and IYITX must be used. The power spectrum resulting in the object distance gate jo is shown as an example: 202505405

[0128] 23

[0129] PRX-NC, TX-NCQO, I) are shown scaled (to maximum) in Fig. 6, with the object still located at the Doppler gate Io = 168. A total of 30 power peaks now occur, lying on an equidistant grid with grid length K / P = 16 (the fact that two grid points are unoccupied will be explained later) and exhibiting different level levels. The maximum level occurs at the Doppler gate Io = 168 of the object, since the power peaks of all MTX = 12 transmit paths are summed there (the power PRX-NC, TX-NCÜO, IO there is therefore 12 times higher than the power peaks in PRx-Ncüo o(mTx)); the other 29 power peaks of PRX-NC. TX-NCQO, I) are lower because, due to the choice of modulation speeds, P(I ITX) is only added over a part of the power peaks of PRX-NCÜO, I) according to reference (14) - a maximum of six power peaks, a minimum of 4.Thus, the Doppler gate lo of the object can now be identified as the position of the highest power peak; the object distance gate jo results from the position of the power peak in dimension j. The complex channel values ​​S2(jo,lo,mTx,rriRx) required for angle determination, i.e., the MRX MTX = 192 combinations of all transmit and receive paths, can be taken from the two-dimensional spectra S2G,l,rriRx) at position I = IO(ITITX) according to reference (12).

[0130] S2(jo,lo,mTx,rriRx) = S2(jo,lo(mTx),rriRx) = S2(jo,modK(lo+p(mTx) K / P),rriRx)

[0131] (16)

[0132] The described method fails if the power peaks in the power spectrum PRX-NC, TX-NCQ, I) resulting from two-dimensional non-coherent integration do not lie above the noise floor, or if there are multiple objects at the same distance that are in the same grid I = G g with

[0133] G g= {g+ (0,..., P-1) K / P} with g = 0,..., K / P-1 (17) generate line peaks. As an example, two point-like objects in the same distance gate jo are considered, with the Doppler gates lo,i = 168 and Io, 2 = 168+9*16 = 312, with the azimuth angles OAZ, I = 4.78° and OAZ,2 = 14.5° and the elevation angles OEI, I = 4.3° and OEI,2 = -4.3°, whereby the additional object 2 at Doppler gate Io, 2 = 312 is supposed to generate a received signal 6dB lower than the previously considered object 1 at lo,i = 168; the power spectrum PRX-NC. TX-NCGO,!) resulting after two-dimensional non-coherent integration is shown in 202505405

[0134] 24

[0135] Fig. 7 is shown scaled (with the same scaling as in Fig. 6). The power peaks of both objects lie in the same grid.

[0136] I = Gs = {8+(0,...,31) 16},

[0137] This means they overlap and are therefore inseparable. The highest power peak occurs at Doppler gate l0,i = 168 of object 1, which has a stronger received signal, making this object identifiable in the measurement. The other object, 2, remains unidentifiable even when the second-highest power peak is used – this occurs at Doppler gate I = 200, where no object is located.

[0138] To solve this problem, the following procedure according to the invention is used: In contrast to the above, the spectrum resulting after distance and Doppler transformation (S2G,l,rriRx) is now coherently integrated over the receive paths, for which digital beamforming is applied. Since the arrangement of the RX antennas according to Fig. 1b represents a two-dimensional array with MRX.AZ = 8 antennas in the horizontal direction and MRX.EI = 2 antennas in the vertical direction, two-dimensional digital beamforming is applied (for azimuth and elevation); for this purpose, the MRX = 16 receive paths I IRX = 0,..., 15 are arranged as a two-dimensional array in S2G,l,rriRx):

[0139] S2, Rx-BF(j,l,mRx, Az=0...7,mRx, Ei=0) = S2(j, I, m RX=0... 7) (18a) S2, Rx-BF(j,l,mRx, Az=0...7,mRx, Ei=1 ) = S2(j, I, m RX=8...15) (18b) For two-dimensional beamforming (hereinafter also referred to as RX-beamforming with the abbreviation RX-BF), a two-dimensional DFT preferably implemented via FFTs with 100% zeropadding (i.e., extension by MRX. AZ = 8 or MRX. EI = 2 zeros) is used, so that it has the dimension (NRX-BF, AZ=16)X(NRX-BF, EI=4), and the window functions W3, Rx-BF(rriRx, Az) are used for azimuth. (Chebyshev window with 30dB sidelobe suppression) and W4, Rx-BF(rriRx, Ei) for elevation (because there are only two elevation channels, W4, Rx-BF(rriRx, Ei) is constant):

[0140] SRX-BF(j, I, nRX-BF, Az, nRX-BF, El) =

[0141] DFTNRX-BF, El[W4, RX-BF(mRX, El) DFTNRX-BF, Az[W3, RX-BF(mRX, Az) S2, RX-BF(j, I, mRX, Az,m RX, El)]], 202505405

[0142] 25

[0143] (19a)

[0144] where riRx-BF, Az = 0,..., NRX-BF, AZ-1 is the sweep variable for the image domain, i.e., the frequency domain of the third discrete Fourier transform DFTNRX-BF, AZ, and represents the so-called RX-BF azimuths (because frequency is proportional to the electrical azimuth angle) and HRX-BF. EI = 0,..., NRX-BF, EI-1 is the sweep variable for the image domain of the fourth discrete Fourier transform DFTNRX-BF, EI, and represents the so-called RX-BF elevation stores; RX-BF azimuth and RX-BF elevation stores are also referred to as RX-BF azimuth and RX-BF elevation beams in the following.

[0145] With the four-dimensional spectrum Wi234, Rx-BFÜ,nRx-BF, Az, HRX-BF. EI) of the used four-dimensional window function wi(i) w2(k) w3, Rx-BF(mRx, Az) w4, Rx-BF(mRx, Ei), which is also formed with 100% zero padding in the third and fourth dimensions and which has its maximum at indices 0, and with S2(j,l,rriRx) according to Ref. (10) the four-dimensional spectrum SRx-BFÜ,nRx-BF, Az,nRx-BF, Ei) for a point-like object at the RX-BF azimuth beam HRX-BF. AZ. O and RX-BF elevation beam HRX-BF. EI. O TO SRX-BF(J, I, nRX-BF, Az, nRX-BF, El) =

[0146] A / (2j)-summTx[exp(j-(pä, Tx(mTx)) • Wi234, Rx-BF(modj(j-jo),modK(l-lo-p(mTx) K / P), mOdNRX-BF, Az(nRX-BF, Az-nRX-BF, Az,o),mOdNRX-BF, El(nRX-BF, EI-nRX-BF, EI,o))] with I ITX = 0,..., MTX-1; (19b) For the sake of completeness, it should also be mentioned that, firstly, the RX-BF azimuth beam HRX-BF. AZ. O and the RX-BF elevation beam HRX-BF. EI. O are related to the azimuth angle OAZ and the elevation angle OEI of the object as follows:

[0147] nRx-BF, Az,o = modNRx-BF, Az(sin(c(Az)-3-NRx-BF, Az) (20a) nRX-BF, EI,0 = mOdNRX-BF, El(sin(C(El)-20-NRX-BF, El), (20b) where the factors 3 and 20 come from the antenna spacings 3X (in the horizontal direction) and 20X (in the vertical direction), and secondly for the angle-related phase (pa. Rx(mRx) in reference (10) the following applies: 202505405

[0148] 26

[0149] (pä, Rx(rriRx=0...7) = 2iT-mRx-nRx-BF, Az,o / NRx-BF, Az (21a) (pä, Rx(rriRX=8... 15) = 2lT-((mRX-BF, Az, O-8)-nRX-BF, Az,o / NRX-BF, Az+nRX-BF, EI,o / NRX-BF, El) (21 b)

[0150] The spectrum SRx-BFÜ,nRx-BF, Az,nRx-BF, Ei) resulting after four-dimensional transformation of a single object exhibits, according to the above designation (19b), MTX = 12 power peaks, which lie at the distance gate jo, the RX-BF azimuth beam ORX-BF. AZ. O, the RX-BF elevation beam HRX-BF. EI. O and the 12 different Doppler gates modK(lo+p(rriTx) K / P). For the case of two objects considered above, this is at their distance gate jo and their identical RX-BF elevation beam HRX-BF. EI. The resulting spectrum SRx-BFÜo,l,nRx-BF, Az,nRx-BF, Ei,o) is shown in Fig. 8 with magnitudes in dB and scaled (to maximum). (Although the two elevation angles OEI,I = 4.3° and OEI,2 = -4.3° of the objects are different, they fall within the same RX-BF elevation beam HRX-BF.EI.O = 2 due to the large vertical distance and the associated high angular ambiguity, as per reference (20b).) Since the two objects have different azimuth angles OAZ,I = 4.78° and OAZ,2 = 14.Having 5°, they are located at the different RX-BF azimuth beams nRx-BF, Az,o,i = 4 and nRx-BF, Az,o,2 = 12 and are thus separated in the spectrum; at the two RX-BF azimuth beams, MTX = 12 power peaks occur, namely at the Doppler gates.

[0151] I = lo,i / 2(mTx) = modK(lo,i / 2+p(mTx) K / P)

[0152] with the object Doppler gates lo,i = 168 and Io, 2 = 312.

[0153] Analogous to equation (15), the RX-BF spectrum SRx-BFÜ,nRx-BF, Az,nRx-BF, Ei) is now integrated non-coherently over the values ​​in the expected grid I = IO(ITITX) = modK(lo+p(rriTx) K / P) of the power peaks for each possible Doppler gate Io = 0,..., K-1:

[0154] PRX-BF, TX-Nc(j,l,nRX-BF, Az,nRX-BF, El) =

[0155] summTx[|SRx-BF(j,modK(l+p(m. Tx) K / P),nRx-BF, Az,nRx-BF, Ei)| 2 ] with mix = 0,..., MTX-1; (22a)

[0156] Here, the loop variable I is again used for the possible Doppler gates Io = 0,..., K-1. For a single object, this results with reference (19b):

[0157] PRx-BF, Tx-Nc(j,l,nRx-BF, Az,nRx-BF, Ei) = |A / 2| 2 • summTx[|summTx[exp(j-(pä, Tx(rriTx)) • 202505405

[0158] 27

[0159] Wi234, Rx-BF(modj(j-jo),modK(l+p(mTx) K / P-lo-p(mTx) K / P),

[0160] mOdNRX-BF, Az(nRX-BF, Az-nRX-BF, Az,o),mOdNRX-BF, El(nRX-BF, EI-nRX-BF, EI,o))]| 2] with m. Tx = 0,..., MTX-1 and I ITX = 0,..., MTX-1. (22b) For the above example of the two objects, the power spectrum PRx-BF, Tx-Ncüo,l,nRX BF, Az,nRX BF, Ei,o) resulting at their distance gate jo and their RX-BF elevation beam HRX-BF. EI. O = 2 is shown scaled in Fig. 9; the 30 power peaks occurring in the same grid I = Gs are now separated, since they occur at different RX-BF azimuth beams nRx-BF, Az,o,i = 4 and nRx-BF, Az,o,2 = 12. This allows the Doppler gate lo,i = 168 and Io, 2 = 312 for both objects to be identified from the position of the highest power peak in the respective RX-BF azimuth beam; thus, both objects can be detected and - as will be shown later - their angles can be determined without mutual interference.By combining, according to the invention, a coherent integration over the receive paths and a subsequent non-coherent integration over the transmit paths, the problem that arises with completely non-coherent integration (i.e., over both receive and transmit paths) is solved.

[0161] The inventive approach also offers advantages with weak signals, since coherent integration allows small signals to be distinguished from noise significantly better than non-coherent integration: With coherent integration, the distance between the signal and the mean noise is increased (by a factor of N for N signals, i.e., by a factor of 16 in the case of MRX = 16 receive paths, which corresponds to 10 Iog10(16) = 12 dB); with non-coherent integration, the distance to the mean noise remains constant, only the variance within the noise floor is reduced (i.e., the noise assumes a flatter profile, as can be seen, for example, by comparing Fig. 5 and Fig. 6 (spectra with non-coherent integration) with Fig. 4 (spectrum before non-coherent integration), i.e.,The statistically induced peaks in the noise floor decrease, which does lead to smaller signals being better distinguished from noise, but this effect is several dB smaller than the gain in the signal-to-noise ratio resulting from coherent integration. 202505405.

[0162] 28

[0163] As mentioned above, object detection requires differentiating the power peaks they generate from noise or noise peaks; this necessitates determining the noise level. Since the noise level can vary not only across the distance gates but also across the RX-BF beams (e.g., due to interference and phase noise effects), it is determined individually for each distance gate and each of the NRX-BF, AZXNRX-BF, EI = 64 RX-BF beams. To do this, an ordered statistic (OS) is performed on the respective K = 512 values ​​of the RX-BF power spectrum resulting from non-coherent TX integration (PRx-BF, Tx-Ncü,nRx-BF, Az,nRx-BF, Ei), i.e., across Doppler dimension I, to find, for example, the 40% smallest value, i.e., the 205 smallest. The detection threshold for distinguishing between object power peaks and noise peaks is reduced by a factor of approximately 4.5, i.e., approximately 6.6dB above the noise level value determined in this way (then the probability that statistically caused noise peaks lie above the detection threshold and thus falsely lead to a detection is low enough).

[0164] For noise level determination, instead of the RX-BF power spectrum PRx-BF, Tx-Ncü,nRx-BF, Az,nRx-BF, Ei) after non-coherent integration over the transmit paths, the RX-BF power spectrum |SRx-BF(j,l,nRx-BF, Az,nRx-BF, Ei)| can also be used. 2before this non-coherent integration, which has two advantageous aspects: Firstly, the noise level can be determined in parallel with the non-coherent TX integration, provided the computer used has parallel computing units, which leads to a reduced latency.Secondly, in the power spectrum before non-coherent TX integration, only 12 power peaks per object are generated, while in the power spectrum after non-coherent TX integration, there are 30 power peaks per object; in particular, if power peaks from several objects occur in an RX-BF beam, the noise level is overestimated by the ordered statistic - this effect is greater the more power peaks occur in the spectrum and thus per object, so that when using the RX-BF power spectrum after non-coherent TX integration, there can be a greater overestimation of the noise level than when using the RX-BF power spectrum before non-coherent integration, which can lead to the non-detection of small objects (whose power peaks 202505405).

[0165] 29

[0166] (then below the noise threshold increased by overestimating the noise) - thus, using the power spectrum before non-coherent TX integration for noise level estimation leads to more robust detection in the case of multiple objects at similar distances with similar RX-BF beams.

[0167] If the noise level is estimated from the power spectrum before incoherent integration, it must be translated to the noise level after incoherent integration (since detection, i.e., checking for the noise threshold, takes place after incoherent integration); assuming an approximately constant and uncorrelated noise across the Doppler ports, the translation factor MTX = 12 (i.e., in the case without noise overestimation due to power peaks, the noise level after incoherent integration is 12 times higher than before).

[0168] In addition to the noise level, other criteria can be used to determine the detection threshold. In particular, it is useful to limit the dynamic range per RX-BF beam and per distance gate; that is, the detection threshold may be at most a certain value below the maximum value obtained in the respective RX-BF beam (via the K = 512 values) or in the respective distance gate (via the 51264 values). This prevents, for example, sidelobes from window functions from causing false detections.

[0169] As described above and shown in Fig. 9, a total of 30 power peaks per object are generated in the RX-BF power spectrum (PRx-BF, Tx-Ncü,nRx-BF, Az,nRx-BF, Ei) after non-coherent TX integration. The highest power peak is at the object's Doppler gate Io and is referred to below as the valid or correct power peak. The other 29 power peaks with lower levels are referred to as invalid power peaks or TX modulation artifacts. To detect objects, the valid power peak must be identified, while the invalid power peaks, i.e., the TX modulation artifacts, are to be ignored, as they do not indicate that an object is located at their Doppler gates. Thus, a simple approach is to use a grid G g According to reference (17) per RX-BF beam (and of course per distance gate, which will not be mentioned again below) only the highest 202505405

[0170] 30

[0171] The power peak is classified as valid, but this leads to the situation that in the case of two objects whose power peaks are in the same RX-BF beam and in the same grid G g In such cases, only the stronger object (i.e., the one with the stronger received signal) is detected. The approaches described below can be used to reduce this problem.

[0172] In the case of a single object, the largest TX modulation artifacts lie approximately 3 dB below the maximum, i.e., correct, power peak (provided this is significantly above the noise level); therefore, at least the second largest power peak can also be considered valid if its level is only slightly below the largest (whereby the permissible level difference can depend on the distance from the noise level) - however, it must be taken into account that if excessively large level differences are allowed (e.g., the 3 dB itself), even with power peaks far above the noise level, an incorrect identification of the Doppler gate of the smaller object can occur, since the objects in the complex RF-BF spectrum SRX-BF overlap and the power peaks there can reinforce or attenuate each other depending on the respective phase.

[0173] As can be seen in Fig. 9, in the respective grid G gThe power peaks do not occupy all 32 positions; rather, the two neighbors (left and right neighbors) to the correct, i.e., largest, power peak are unoccupied. Therefore, if power peaks also occur at these two positions, they must originate from a second object. They can be considered valid if the second largest power peak occurs there, or if their level is even a few dB below the second largest power peak. The fact that the two neighbors to the correct power peak in grid G g Having the TX modulation artifacts unoccupied has the following two advantages in particular:

[0174] - The detection of pedestrians in the presence of other strong reflections from the stationary environment (e.g., buildings, trees, traffic signs, parked vehicles, etc.) is important; due to the comparatively low speed of a pedestrian, their Doppler gate and the Doppler gate of infrastructure reflections can differ by a grid spacing K / P = 16, which is due to the comparatively low 202505405

[0175] 31

[0176] The reflectivity of pedestrians can lead to the following: G in the same grid gThe pedestrian's peak power is significantly lower than that of the infrastructure – but because the pedestrian's peak power is located at an unoccupied grid neighbor position of the significantly larger infrastructure peak power, the pedestrian can be detected. This prevents the peak power of two objects with different relative velocities from being detected in the same grid G ​​over many radar cycles. g Advantageously, the period TD of the K frequency ramps (see Fig. 2) is changed from radar cycle to radar cycle (i.e., from data acquisition to data acquisition) – so instead of TD = 70 ps, ​​for example TD = 72 ps is used for the next K frequency ramps. This changes the ratio to the grating spacing 16 of the gratings G. gThe corresponding relative velocity difference is reduced accordingly, i.e., by 2.8% in this numerical example, so that with two objects of different relative velocity, it is impossible for them to be in exactly the same grid G ​​in two successive radar cycles. gThe relative velocity difference between the two objects is greater (at least apart from rounding effects for the integer grids); the greater the relative velocity difference between the two objects, the further their two grids shift relative to each other due to a change in the ramp repetition time (TD). However, if the relative velocity difference is only about one grid spacing, then a moderate change in the ramp repetition time (TD) may not be sufficient to clearly separate the power peaks, because power peaks typically extend over about three Doppler gates (due to their window function used in the FFT); but since the two grid neighbors of the actual power peaks are each unoccupied, a lack of grid separation does not pose a significant problem.

[0177] So that in grid G gIf the two neighboring positions to the correct power peak are not occupied, the normalized modulation speeds P(I ITX) must be chosen so that they differ by at least 2 (also taking into account the periodic wrap-around caused by the modulo property):

[0178] for each (ITITX. I = 0,..., MTX -1) + (rriTx,2= 0,..., MTX -1): 202505405

[0179] 32

[0180] modp(p(rriTx,i) - p(rriTx,2), P) e {2,... P-2}; (23)

[0181] This condition is satisfied for the example considered here according to reference (11). Instead of a completely omitted occupation of the two adjacent positions, a significantly reduced occupation compared to the other TX modulation artifact positions could also be used; for example, exactly two modulation speeds P(I ITX) could be allowed to be directly adjacent, i.e., differing by only one.

[0182] For the most robust detection possible, the modulation speeds P(I ITX) are chosen such that, for a single point-like object, the difference between the largest and second-largest power peaks is as large as possible, possibly subject to further boundary conditions, e.g., the boundary condition considered here that the neighboring positions to the correct power peak are unoccupied; for this purpose, preferably all possible combinations of modulation speeds are searched, cleverly avoiding the repeated examination of the same set of modulation speeds (only with a different order), and excluding a priori combinations (here, those in which two p(mrx) have a distance of less than 2) (otherwise 12 would be 32 (to search combinations).

[0183] It should also be mentioned that in the case of non-coherent integration over the MTX = 12 transmission paths in total MTX 2= 144 power peaks are used, meaning the sum of the power peaks in the resulting power spectrum PRX-BF, TX-NC is 144 times the power of the 12 individual power peaks, assumed to be of equal magnitude, before non-coherent TX integration, i.e., in SRX-BF. Since the correct power peak of PRX-BF, TX-NC MTX = 12 power peaks are summed, the other MTX (MTX - I ) = 132 power peaks are distributed across the TX modulation artifact positions, so that a theoretical lower bound for the minimum number of summed power peaks in the largest TX modulation artifact can be determined - in the example under consideration, it would be five (132 / 29 increased to the nearest integer); however, as in the example considered here, there can be no combination, 202505405

[0184] 33

[0185] which realizes this theoretical minimum (here the best combinations have six power peaks in the two largest TX modulation artifacts).

[0186] For the non-coherent integration, power values ​​were used above, as is generally customary. Alternatively, and according to the invention, magnitudes (i.e., absolute values) can also be used; in particular, for very weak objects, this leads to a slightly reduced probability that the largest power peak is not the actual power peak, but rather a TX modulation artifact, resulting in a false identification. Whether calculating power or magnitudes requires less computational effort depends on the computing platform used.

[0187] The above describes how power peaks can be checked against a detection threshold (derived from the noise level and, if necessary, additional criteria, especially for limiting the dynamic range) and to ensure they are not TX modulation artifacts. This check is advantageously performed at each distance gate not only for power peaks but for every value, i.e., across all Doppler gates and RX-BF beams (in particular, it is not checked whether the value represents a local maximum in Doppler gate dimensions). The result is also needed in a later processing step and is stored as a bit field of length 64 for each Doppler gate (bits with the value one indicate that the corresponding RX-BF beam is valid in the respective Doppler gate, i.e., the value there is above the detection threshold and is not a TX modulation artifact).All Doppler gates in which at least one of the 64 RX-BF beams passes the detection threshold test and shows no TX modulation artifacts are then declared valid. If the local maximum were checked at this stage, objects that only become visible after full digital beamforming (i.e., using the transmit paths) could be missed; therefore, the local maximum test, i.e., the power peak test, is only performed after full digital beamforming.

[0188] The next section describes how the valid Doppler gates determined in this way are further processed. In the example above, with two objects located at the same distance gate but different RX-BF beams, the Doppler gates are 202505405.

[0189] 34

[0190] lo,i = 168 and Io, 2 = 312, and because power peaks extend across three Doppler gates there, the set L results. v the valid double goals to

[0191] Lv = {167, 168, 169, 311, 312, 313}.

[0192] First, the set Lv of valid Doppler gates is extended to include their direct neighbors, i.e., the Doppler gates before and after it, provided these are not already valid themselves – this is necessary to later check for a local maximum in the Doppler gate dimension; this is called the set L P the processed Doppler gates are referred to as:

[0193] Lp = {166, 167, 168, 169, 170, 310, 311, 312, 313, 314}.

[0194] As shown in Fig. 1c and explained above, the combinations of the eight transmitting antennas on the right and left (TXO-3 and TX4-7) and the 16 receiving antennas below and above (RXO-7 and RX8-15) synthesize a two-dimensional equidistant array with (MTXRX, AZ=16)X(MTXRX, EI=8) = 128 antenna channels in a horizontal grid of 3X and a vertical grid of 5X. The complex channel values ​​S2(jo,lp,mTx,rriRx) of these antenna channels are applied to the two-dimensional spectra S2G,l,rriRx) resulting after distance and Doppler transformation at the Doppler gate l to be processed in each case. P corresponding points I = Ip(rr)Tx) taken from reference (12):

[0195] S2(jo,lp,mTx,rriRx) = S2(jo,lp(mTx),rriRx) = S2(jo,modK(l P +p(mTx)) K / P,rriRx)

[0196] with FTIRX = 0,...,15 and mTx = 0,...,7 (24) These 128 values ​​are now subjected to two-dimensional digital beamforming (in contrast to the RX beamformer above, coherent integration is now performed not only over the signals of the receive paths, but also over the transmit paths); for this purpose, they are arranged according to the two-dimensional antenna array:

[0197] S2, TXRx-BF(jo, lp,rriTXRx, Az=0...7,mTXRx, Ei=0...3) = S2O0, lp, m TX=0...3,rriRx=0...7) T (25a)

[0198] S2, TXRx-BF(jo, lp,rriTXRx, Az=8...15,mTXRx, Ei=0...3) = S2(jo, lp, m TX=4...7,rriRx=0...7) T (25b) 202505405

[0199] 35

[0200] S2, TXRX-BF(JO, lp,rriTXRx, Az=0...7,rriTXRx, Ei=4...7) = S2(jo, Ip, m TX=0...3,rriRx=8... 15) T (25c)

[0201] S2, TXRX-BF(JO, Ip, rriTXRx, Az— 8... 15, rriTXRx, Ei—...7) = S2(jo,lp,mTx=4...7,mRx=8... 15) T . (25d)

[0202] For two-dimensional beamforming (hereinafter also referred to as TXRX-Beamforming, abbreviated TXRX-BF), a two-dimensional DFT with 100% zeropadding (i.e., extension by MTXRX.AZ = 16 or MTXRX.EI = 8 zeros), preferably implemented via FFTs, is used, giving it the dimension (NTXRX-BF, AZ=32) X (NTXRX-BF, AZ=16). The window functions W3, TXRx-BF(rriRx, Az) for azimuth (Chebyshev window with 30dB sidelobe suppression) and W4, TXRx-BF(m RX. EI) for elevation (also a Chebyshev window with 30dB sidelobe suppression) are used.

[0203] STXRX-BF(jo, Ip, nTXRX-BF. Az, HTXRX-BF. El) = DFTNTXRX-BF, El[W4, TXRX-BF(rriTXRX, El)' DFTNTXRX-BF, Az[W3, TXRX-BF(mTXRX, Az) ' S2, TXRX-BF(jo,lp,mTXRX, Az,mTXRX, El)]] (26a)

[0204] where nTXRx-BF, Az = 0,..., NTXRX-BF, AZ-1 is the sweep variable for the image domain, i.e., the frequency domain of the third discrete Fourier transform DFTNTXRX-BF, AZ, and represents the so-called TXRX-BF azimuths (because frequency is proportional to the electrical azimuth angle), and HTXRX-BF, EI = 0,..., NTXRX-BF, EI-1 is the sweep variable for the image domain of the fourth discrete Fourier transform DFTNTXRX-BF, EI, and represents the so-called TXRX-BF elevation stores; TXRX-BF azimuth and TXRX-BF elevation stores are also referred to as TXRX-BF azimuth and TXRX-BF elevation beams in the following.

[0205] The spectrum W34, TXRX-BF, HRX-BF, AZ, HRX-BF, EI) of the two-dimensional window function W3, TXRx-BF(mRx, Az)-W4, TXRx-BF(mRx, Ei) used for beamforming, which is also formed with 100% zero padding in the third and fourth dimensions and which has its maximum at the indices 0, yields the TXRX beamformer spectrum STXRx-BF(jo,lp,nTXRx-BF, Az,nTXRx-BF, Ei) for a point-like object at the distance gate jo, the Doppler gate Io, the TXRX-BF azimuth beam OTXRX-BF, AZ, O and TXRX-BF elevation beam HTXRX-BF, EI, O ZU 202505405

[0206] 36

[0207] STXRX-BF(jo, Ip, riTXRX-BF. Az, ClTXRX-BF. El) = A / (2j) • Wl2(0,mOdK(lp-lo)) ■ W34, TXRX-BF( mOdNTXRX-BF, Az(nTXRX-BF, Az-nRTXRX-BF, Az,o),mOdNTXRX-BF, El(nTXRX-BF, EI-riTXRX-BF, EI,o)); (26b)

[0208] For the sake of completeness, it should also be mentioned that the TXRX-BF azimuth beam HTXRX-BF. AZ. O and the TXRX-BF elevation beam HTXRX-BF. EI. O are related to the azimuth angle OAZ and the elevation angle OEI of the object as follows:

[0209] nTXRx-BF, Az,o = modNTXRx-BF, Az(sin(aAz)-3-NTXRx-BF, Az) (27a) nTXRX-BF, EI,0 = mOdNTXRX-BF, El(sin(C(El)-5-NTXRX-BF, El), (27b) where the factors 3 and 5 come from the antenna spacings 3X (in the horizontal direction) and 5X (in the vertical direction).

[0210] We now consider again the above example with two objects in the same distance gate jo and at the Doppler gates lo,i = 168 and Io, 2 = 312, whereby the signals corresponding to the transmit paths appear in the same grid Gs (see references (17) and (12)); The azimuth angles OAZ,I = 4.78° and OAZ,2 = 14.5° of the two objects lead, according to designation (27a), to the TXRX-BF azimuth beams nRTXRx-BF, Az,o,i = 8 and nRTXRx-BF, Az,o,2 = 24, and the elevation angles OEI,I = 4.3° and OEI,2 = -4.3°, according to designation (27b), to the TXRX-BF elevation beams nRTXRx-BF, Ei,o,i = 6 and nRTXRx-BF, Ei,o,2 = 10. In Fig. 10, the Doppler gate l PThe TXRX-BF spectrum (STXRx-BF0o,lo,i,nTXRx-BF, Az,nTXRx-BF, Ei) resulting from object 1 (= lo,i = 168) is shown in magnitude in dB and scaled; as expected, a power peak occurs in the TXRX-BF beam of object 1, i.e., at ORTXRX-BF. AZ. O. I = 8 and nRTXRx-BF, Ei,o,i = 6, but a power peak, particularly in the elevation direction, also occurs in the region of the TXRX-BF azimuth beam nRTXRx-BF, Az,o,2 = 24 of object 2, even though object 2 is located at a different Doppler gate (at Io, 2 = 312). This fragmented power peak arises because the Doppler gates IO, I(IT)TX) resulting from reference (12) of the transmit path signals of object 1 also partially correspond to transmit path signals of object 2 - at a total of four points, the Doppler gates IO,2(IT)TX) of object 2 correspond to the Doppler gates IO, I(IT)TX) of object 1; thus, some signals from object 2 are also included in the digital beamforming, however, they have an incorrect assignment to the 202505405

[0211] 37

[0212] Transmit paths and thus generally incorrect calibration (the previous considerations and formulas have not addressed the fact that the antenna channels generally also have different phase angles and slightly different amplitudes due to hardware-related effects such as different antenna feed line lengths; these differences can be determined, for example, during sensor production and then compensated for during signal processing using appropriate calibration factors). As a result, these signals from object 2 do not exhibit meaningful beamforming, particularly in the elevation direction (elevation is dominated by the transmit paths), while a blurred power peak is still visible in azimuth (azimuth is dominated by the receive paths).

[0213] To prevent these components of object 2 from later leading to detections in the TXRX-BF spectrum (STXRx-BFÜo,lo,i,nTXRx-BF, Az,nTXRx-BF, Ei) (which would have an incorrect Doppler gate and incorrect TXRX-BF beam), they must be identified as TX modulation artifacts. For this purpose, it is checked for each value in the TXRX-BF spectrum whether the RX-BF power spectrum PRx-BF, Tx-Ncüo,lo,i,nRx-BF, Az,nRx-BF, Ei) resulting from incoherent TX integration was valid at the corresponding position (the result of the check there for detection threshold and no TX modulation artifact was temporarily stored in bit fields); The necessary relationship between RX-BF-Beams and TXRX-BF-Beams is derived from references (20) and (27) to:

[0214] nRX-BF, Az = nTXRX-BF, Az NRX-BF, Az / NTXRX-BF, Az = nTXRX-BF, Az / 2 (28a) nRX-BF, EI = mOdNTXRX-BF, EI / NRX-BF, El(nTXRX-BF, Az) = ITIOd4(nTXRX-BF, Az) (28b) To identify power peaks, local maxima of the TXRX-BF spectrum STXRx-BFÜ,nTXRx-BF, Az,nTXRx-BF, Ei) resulting in four-dimensional space are determined, i.e., via TXRX-BF azimuth beams, via TXRX-BF elevation beams, via Doppler gates I (therefore the set L P (the processed Doppler gates are also extended to include the neighbors of the valid Doppler gates) and via the distance gates (for this, the corresponding values ​​of the TXRX-BF spectrum in the two adjacent distance gates must also be calculated). 202505405

[0215] 38

[0216] In addition, such identified power peaks must be checked against a detection threshold, which is based primarily on the noise level. It could be argued that this is implicitly included in the above check for validity in the RX-BF power spectrum PRX-BF, TX-NC resulting from non-coherent TX integration (since a detection threshold is also checked there), but the RX-BF beams used there correspond to several TXRX-BF beams of the TXRX-BF spectrum considered here (because TXRX beamforming uses many more channels, especially in elevation, and therefore has a better resolution). Therefore, especially with multiple TXRX-BF power peaks on the same RX-BF beam, it must be checked whether the smaller power peaks are also above the noise level. For a noise-derived detection threshold, the respective distance gate and corresponding RX-BF beam (according to the reference...) must be...(28)) that certain noises (before or after non-coherent TX integration) are translated to the TXRX beamformer during RX-BF beamforming; when using the RX-BF noise before non-coherent integration, this must be multiplied by the factor TNoise = [SUm(W3, TXRX-BF(mTXRX, Az). 2 ) SUm(W4, TXRX-BF(mTXRX, El) 2 )] /

[0217] [sum(w3, Rx-BF(mRx, Az) 2 ) sum(w4, Rx-BF(rriRx, Ei) 2)] (29) are multiplied to obtain the noise in the TXRX beamformer (assuming approximately constant and uncorrelated noise across all receive paths). The detection threshold is then set approximately 25 times higher than this noise (mean noise is 1.97 times higher than the OS40% noise calculated here, and the detection threshold is assumed to be 11 dB, i.e., 12.6 times higher than the mean noise). Additionally, the detection threshold could also include a dynamic range limitation per TXRX BF beam and / or across the entire distance gate. To prevent detections from sidelobes of the digital beamforming (due to the window functions used and imperfect sensors), so-called sidelobe thresholds are also employed. These thresholds prevent detections that are too far (e.g., more than 10 dB) below the maximum in the respective TXRX azimuth beam and the maximum in the respective TXRX elevation beam. 202505405

[0218] 39

[0219] The following criteria exist for identifying detections in the TXRX-BF spectrum: exceeding the detection threshold, validity in the non-coherently integrated RX beamformer, and a local maximum in four-dimensional space; these criteria can, of course, be checked in any order. As already mentioned, such detections in the TXRX-BF spectrum are still ambiguous with respect to azimuth and elevation angles, since the distances in the two-dimensional equidistant antenna array used, as shown in Fig. 1c, are greater than half the wavelength.To resolve the azimuth ambiguities, the antenna channels from the transmitting antennas TX1, TX8, TX10 and TX5 and all receiving antennas can be used; to resolve the elevation ambiguities, the antenna channels from the transmitting antennas TX8-TX11 and all receiving antennas can be used, since two of these transmitting antennas (TX8 and TX10 for azimuth and TX9 and TX11 for elevation) are located half a wavelength outside the respective equidistant grid. The complex channel values ​​S2Ü,l,nriTx,mRx) of the additionally required antenna channels are taken from the spectra S2G,l,rriRx) at the points I = IO(ITITX) corresponding to the Doppler gate Io to be processed, as per reference (24), in the same way as the channel values ​​required for TXRX beamforming.

[0220] S2G0, Io, IYITX, IYIRX) = S2(jo,lo(mTx),rriRx) = S2(jo,modK(lo+p(mTx) K / P),rriRx)

[0221] with mRx = 0,...,15 and mix = 8,...,11 (30) To resolve the azimuth ambiguities, digital beamforming is performed for the four transmit paths IYITX = 1,8,10,5 and all receive paths - either at all azimuth angles or, advantageously, only at the six hypotheses resulting from the TXRX beamforming (because of the 3X spacing, there are six hypotheses in -90°...+90°). For elevation, this can be performed for the four transmit paths I ITX = 8,9,10,11 and all receive paths, where the correct one is found among 10 hypotheses (whereby, due to the vertical extent of the individual antennas and the associated focusing, large elevation angles can be excluded a priori).

[0222] Now we consider again the example above with two objects in the same distance gate jo and at the Doppler gates lo,i = 168 and Io, 2 = 312, respectively, whereby the signals corresponding to the transmission paths appear in the same grid Gs. (See 202505405)

[0223] 40

[0224] As also explained with reference to Fig. 10, object 2 generates components in the TXRX-BF spectrum of the Doppler port 11, i1, which do not form a sharp power peak. Depending on the position of the two objects, these components from object 2 can extend into or beyond the power peak region of object 1, which can lead to errors, particularly when determining the angle of object 1. These angular errors (i.e., their magnitude and sign) depend on the phase relationship in which the components from both objects superimpose in the region of the TXRX-BF beam from object 1 (the superposition in the TXRX-BF spectrum takes place in the complex domain).To prevent the same or at least similar angular error from occurring in successive cycles (in which the angles of the objects do not change significantly) (then the angular averaging that takes place in so-called tracking, i.e. the tracking of detections over several radar cycles, would not be helpful), the phase modulation is suitable to be changed from radar cycle to radar cycle.

[0225] A first approach to this is that in the phase modulation according to reference (4) the start phase (po(rriTx,c) is different over the transmit paths I ITX = 0,..., MTX-1 and varies over the radar cycles c = 0,1,... (in the original reference (4b) all start phases were zero):

[0226] (ppM,iin(k,mTx) = (po(rriTx,c) + 2iT-kp(mTx) / P; (31) where the start phase, for example, assumes random values ​​from the entire phase range O...2TT. During TX demodulation, i.e., the extraction of the complex antenna channel values ​​from the TXRX-BF spectrum at the corresponding Doppler gates, the respective start phase must be compensated by multiplication with a corresponding complex unit vector. For demodulation of object 1, the signals resulting in the associated Doppler gates are thus multiplied by the complex unit vector that corresponds to the respective transmit path. For Doppler gates where there are also transmit path signals from object 2, these originate from a different transmit path than those of object 1, so that the start phases are incorrectly compensated for these transmit path signals from object 2. Thus, the phases of these 202505405

[0227] 41

[0228] The contributions originating from object 2 and incorporated into the TXRX beamforming of object 1 change over the radar cycles, so that their contributions in the spectrum change not only in phase but also in general form; thus it cannot happen that object 2 always has a maximum contribution in the area of ​​the TXRX-BF beam of object 1 over several cycles.

[0229] A second approach to changing the phase modulation across radar cycles can be achieved by varying the modulation velocities p(m_TX). This can generally involve using different sets of such modulation velocities (i.e., changing the set defined by the values ​​p(m_TX)), or simply changing the mapping of the transmit paths while using the same set. This alters the mapping of transmit paths from object 2 to the Doppler gates of object 1 (at least their mapping, and possibly also the Doppler gates themselves).

[0230] The same effect can be achieved by the third approach, which involves changing the period P of the linear modulation across the cycles.

[0231] When assigning the modulation speeds p(m) / P of the linear phase components to the transmission paths I ITX, the following effect should be considered: Due to the reflective properties of the road surface, the transmitted signal reaches an object both directly and via a reflected path on the road surface; depending on the phase difference of the two components, this can lead to amplification or attenuation up to total cancellation (at a 180° phase difference). The phase difference depends on the height of the transmitting antennas above the road surface (and of course also on the height of the object, which is not significant here); since in the antenna arrangement considered here according to Fig.1b Since the transmitting antennas are essentially arranged in four planes (apart from the additional offset of antennas TX9 and TX11), the effective transmit power at the object, and thus also the power of the corresponding received signals, will differ in the presence of road reflections. For example, if total cancellation occurs at transmitting antennas of the second and fourth planes (viewed from the bottom) (i.e., for antennas TX1, TX8, TX10 and TX5 in the second plane, and TX3 and TX7 in 202505405).

[0232] 42

[0233] fourth level) and the other six transmitting antennas are assigned to a TX modulation artifact (there are two TX modulation artifacts, each with six transmitting antennas assigned to it), then the level in this TX modulation artifact would be the same as in the correct Doppler gate, so that a unique object identification would no longer be possible.Therefore, the assignment of the modulation speeds p(m) / P of the linear phase components to the transmit paths m_TX should advantageously be such that, for the TX modulation artifacts, the distribution of the used transmit paths across the different vertical transmit antenna levels corresponds as closely as possible, in percentage terms, to the distribution in the actual Doppler gate (where, in the approach presented here, all transmit paths are included). For the TX modulation artifacts to which six transmit antennas are assigned, there should be two transmit antennas each from the second and third vertical levels, and one each from the first and fourth levels. In general, this cannot be achieved exactly for all TX modulation artifacts; by searching through all possible assignment combinations, the assignment that comes closest to this goal can be determined.

[0234] Another criterion for assigning the modulation speeds p(m) / P to the transmit paths I ITX can be the consideration of the transmit path strength (unlike the case of equally strong transmit paths considered here, real sensors exhibit slight differences due to hardware effects). Therefore, weaker transmit paths are advantageously used for the TX modulation artifacts to which the most transmit paths are assigned (six transmitting antennas in the example under consideration).

[0235] So far, only integer ratios K / P of the length K of the second FFT (across the frequency ramps) and period P of the linear component of the phase modulation have been considered (in particular the example K / P = 512 / 32 = 16); the power peaks of the different transmit paths in the second DFT and the power peaks in the spectra used for object identification (i.e., the power peaks at the correct Doppler gate and at the TX modulation artifact positions) then assume positions from an equidistant grid with grid spacing K / P. Now, we will consider what effects would result with non-integer K / P ratios.

[0236] 43

[0237] For example, with P = 24 and K = 512 unchanged, K / P = 21.33. To identify the correct Doppler gates, one would then have to search this grid. Since only values ​​for integer Doppler gates are available, either the nearest integer Doppler gates of the grid would have to be used (leading to varying distances of 21 and 22), or corresponding intermediate values ​​would have to be calculated by interpolation. However, neither the use of a non-equidistant grid nor interpolation is supported when using special hardware-implemented computational logics, which are frequently used in radar processing.Furthermore, non-integer K / P ratios can lead to adverse performance: If the nearest integer Doppler gate of the grid is used, the distance of the used values ​​to the actual position of the power peaks is not constant. For example, if the target is at the non-integer Doppler gate 10.33, the used values ​​are sometimes located exactly at the position of the respective power peak, but sometimes also a third of a Doppler gate before or after it. This means that the levels used do not correspond exactly to the ratio of the power peaks, which can lead to less accurate identification of the object Doppler gates. (With an integer K / P ratio, such as the K / P = 16 considered above, all used values ​​are located a third of a Doppler gate before the respective position of the power peaks, so that the level ratio used corresponds to that of the power peaks.)While interpolation can reduce this problem, errors can still occur because, for example, the shape of the power peaks assumed for interpolation differs from the actual shape (e.g., because the object has relative acceleration). If digital beamforming is also performed via the transmit paths, the required signals from the transmit paths of the second DFT must be taken at the corresponding positions, as explained and used above. However, since these positions lie on a grid with a non-integer grid spacing, the values ​​used (determined from the nearest Doppler gate or by interpolation) do not match exactly, exhibiting not only amplitude but also phase errors, which can lead to significant errors in the beamforming spectrum. 202505405.

[0238] 44

[0239] For the reasons stated above, it is not advisable to use non-integer K / P ratios. With the considered K = 512, however, this leads to the conclusion that only values ​​of P that are powers of two are possible. To reduce the number of TX modulation artifacts, a period P = 24 would also be desirable. For this purpose, zero-padding can be performed on the second DFT to a length L = 768, resulting in L = 768 Doppler gates and the relevant ratio L / P = 768 / 24 = 32, which is again an integer.

[0240] With a period P that is not a power of two, but phase shifters whose number of realizable values ​​in an equidistant grid is typically a power of two (64 phase values ​​in the example under consideration), many of the required phase values ​​2iT m / P, where m is an integer, cannot be realized exactly. One approach is to always use the next available value (instead of phase 2rr5 / 24, phase 2ir 13 / 64 is used); however, the resulting quantization error then repeats itself every period P (possibly even with a smaller period), which ultimately leads to harmonics, i.e., further power peaks in the spectrum (analogous to Fig. 3a for non-ideal phase shifters), which can lead in particular to false detections and errors in angle formation. To avoid this (analogous to Fig.3b), the quantization errors must be decorrelated, which can be achieved, for example, by a superimposed random phase component. The random component φ_PM,r(k) considered above can be used for this purpose; it is identical for all transmit paths and varies over the frequency ramps k = 0,..., K-1. However, unlike above, it is no longer defined by random selection from the 64 phase values ​​of the phase shifters (i.e., as a value from 2π·m / 64 with m = 0,...,63), but rather it takes on any arbitrary value from 0...2π. The phase shifter value used in each case is the one that is closest to the phase value calculated according to (4) (composed of the random component φ_PM,r(k) and the linear component φ_PM,lin(k, m_TX), possibly with a random start phase).

[0241] In the processing sequence described above, the RX beamforming is performed according to the two DFTs across the individual values ​​per frequency ramp and across the frequency ramps; the order can of course be arbitrary.

[0242] 45

[0243] can be changed - for example, the RX beam shaping could also take place right at the beginning.

[0244] The inventive method of RX beamforming for identifying valid Doppler gates (and thus for distinguishing TX modulation artifacts) exploits the fact that, with equidistant RX antennas, targets with different angles can be separated by beamforming. This applies not only to the two-dimensional RX array considered here, but also to one-dimensional RX arrays. At least reduced separation (i.e., with a reduced dynamic range) is still possible even if there are moderate deviations from an equidistant arrangement in the RX antenna array – either because many antennas have small offsets from an equidistant grid or only a few antennas have larger offsets.

[0245] In simpler radar sensors with fewer antennas, the RX antennas are often arranged in a highly non-equidistant manner (for example, they represent a so-called sparse array). If RX beamforming is then performed, large sidelobes appear in the beamforming spectrum in addition to the actual power peak. This means that if components from two or more objects occur in an RX-BF spectrum at angles to different RX-BF beams, they are no longer clearly distinguishable, which can lead to incorrect identification of their Doppler gates. Therefore, to identify the valid Doppler gates (as distinct from the TX modulation artifacts), it may be advantageous not to perform and utilize RX beamforming with subsequent non-coherent integration over the received signals, but rather, according to the invention, to use coherent integration over the receive and transmit paths, i.e., the two-dimensional beamforming according to reference (26), whereby this is then applied to all Doppler gates I = 0,..., to determine K-1 (and not just in a subset of L). P For this TXRX beam shaping STXRx-BFÜ,nTXRx-BF, Az,nTXRx-BF, Ei) the maximum power PTXRx-BF,max(j,l) is formed in each Doppler port I = 0,..., K-1 via the NTXRX-BF, AZ x NTXRX-BF, EI beams:

[0246] PTXRX-BF,max(j, I) = maXnTXRX-BF, Az & nTXRX-BF, EI (|STXRX-BF(j, I, HTXRX-BF. AZ, nTXRX-BF, El)| 2 ) (32) 202505405

[0247] 46

[0248] where “maxn&m” denotes the maximum over all combinations of the indices n and m; for the above example of the two objects, the resulting TXRX-BF power maximum PTXRx-BF,max(jo,l) in dB and scaled (to maximum) is shown in Fig. 11. The identification of the valid Doppler gates is now carried out in PTXRx-BF,max(jo,l) analogously to the procedure for the non-coherently integrated RX-BF, but here only in one vector and not in several vectors for different beams.

[0249] Even in the initially considered, non-coherently integrated power spectrum PRX-NC, TX-NCGOJ) across receive and transmit paths according to reference (15), the identification of valid Doppler gates only occurred in a single vector, i.e., without beamforming and thus without angular separation. In this case, the signal-to-noise ratio did not improve through integration; only the variance in the noise floor was reduced. In contrast, the coherent integration used to determine PTXRx-BF,max(jo,l) leads to a significantly improved signal-to-noise ratio (as can be seen from a comparison of Fig. 11 and Fig. 7); this allows for better identification of objects with a small received signal (distant objects and / or objects with low reflectivity).

[0250] Another advantage of the TXRX-BF power maximum PTXRx-BF,max(jo,l) is that the level of the TX modulation artifacts is generally not the full sum of the signal levels of the transmit paths located there, but is lower, because the transmit path signals do not add up in phase in any TXRX-BF beam; this is also evident by comparing Fig. 11 and Fig. 7, since in Fig. 11 for Pi Rx-BF.max the level difference of the TX modulation artifacts to the maximum is on average more than a factor of 2 lower than in Fig. 7 for PRX-NC. TX-NCGOJ) - a factor of 2 must be taken into account in the comparison, since for PTXRx-BF,max a signal sum is formed over the contributing transmit path signals, whereas for PRX-NC... TX-NCQO, I) a power sum, so that the reduced number of transmit path signals used in the TX modulation artifacts has a correspondingly different effect when powers are compared.This advantage also means that at the TXRX-BF power maximum PTXRx-BF,max(jo,l) the smaller object 2 at the Doppler gate Io, 2 = 312 is generally identifiable, since it develops the second largest power peak (see Fig. 11), whereas this is not the case at 202505405.

[0251] 47

[0252] The non-coherent integrated power spectrum PRX-NC. TX-NCGOJ) according to Fig. 7 is not the case (there the level of the TX modulation artifacts is higher at I = 136 and I = 200).

[0253] To prevent the recurring case over radar cycles in which, during TXRX beam shaping, many or all contributing transmit path signals overlap at least approximately coherently in a beam due to a TX modulation artifact, it is also advantageous to apply the variation of the start phases over the transmit paths (see Ref. (31)).

[0254] In general, even when using the TXRX-BF power maximum PTXRx-BF,max(jo,l), all previously presented, advantageous and inventive designs of the TX modulation, i.e., the different phase responses across the transmit paths, can be used.

[0255] It should be added that, of course, all previously presented, advantageous and inventive designs of TX modulation can be applied when using the non-coherently integrated power spectrum PRX-NC, TX-NC(]O, I).

[0256] Now, let us discuss an advantageous implementation of the non-coherent TX integration according to references (15) and (22). There, for each Doppler port lo = O,..., K-1, the values ​​of PRX-NCGJ) or | SRX-BFG, I, HRX-BF. AZ, nRx-BF, Ei)| are used. 2 The power peaks are integrated into the expected grid I = IO(IT)TX. For all P Doppler gates Io in a grid G gAccording to Equation (17), the values ​​to be summed also lie in this grid; the grid I = IO(IT)TX) of these values ​​is defined according to Equation (12) by the modulation velocities P(I ITX) and shifts successively with Io. Thus, the calculation of the P sums for this grid can be carried out as a cyclic correlation between the P values ​​in this grid and an occupancy vector b(n) with n=0,...,(P-1), where b(n) = 1 for all n corresponding to a P(I ITX), and otherwise b(n) = 0. The cyclic correlation can advantageously be carried out as a fast correlation, i.e., by an FFT of the P values ​​of the grid, a multiplication of this FFT with the pre-calculated complex conjugate of the FFT of the occupancy vector b(n), and a subsequent inverse FFT. If a normal FFT (i.e., not an inverse FFT) is used for the inverse transformation, then 202505405

[0257] 48

[0258] The conjugate complexes of the first FFT are multiplied by the pre-calculated FFT of the occupancy vector b(n), and a scaling factor P may need to be taken into account. Since the values ​​used for the summation are real-valued, the summation can be realized simultaneously for two different grids by fast correlation, by assigning the values ​​of one grid to the real part and those of the other grid to the imaginary part of a complex-valued vector (the FFT works with complex values). If a normal FFT (i.e., not an inverse FFT) is used for the inverse transformation, the conjugate complex of the result must then also be calculated.The computing units used for radar processing are generally optimized for FFTs, so this approach according to the invention can lead to a significant reduction in the required computing time for non-coherent TX integration; this is particularly important for the approach with prior RX beam shaping, since the non-coherent TX integration has to be performed in each RX-BF beam, i.e. 64 times, and therefore represents the most computationally intensive step of the entire calculation.

[0259] If the signals corresponding to the different transmission paths are not of equal strength, they can be used with different weights in non-coherent TX integration, for example, to optimize the signal-to-noise ratio. Such different weights can be implemented in the occupancy vector b(n) for cyclic correlation by ensuring that the occupied elements no longer all assume the value 1, but rather their respective weights.

[0260] Different transmission levels across the transmitting antennas can occur due to hardware-related effects such as different lengths of the antenna leads or tolerances in the electronic components generating the transmission power; these generally unintentional and unknown differences are determined during sensor production or even during sensor operation in order to ascertain the calibration factors required for signal processing.

[0261] Different transmission levels can also be specifically generated via the transmitting antennas (through appropriate configuration of the signal power generation) and 202505405

[0262] 49

[0263] The following will now be briefly explained: In the example considered so far, there are MTX = 12 transmitting antennas and a modulation period of P = 32 is used; to enable robust detection of the correct Doppler gate of the objects, the period P is significantly higher than the number of transmitting antennas. For even better environmental detection, future radar systems will have even more transmitting antennas, e.g., MTX = 32. Then a significantly larger modulation period P would also be necessary, which would increase the number of TX modulation artifacts and the probability that the power peaks of two objects in the same grid G g to come to rest, significantly increased; in addition, a large number of modulation artifacts also leads to an overestimation of the noise level, from which the detection threshold is derived, and thus to a reduced dynamic range.

[0264] In the approach considered so far, with at least approximately the same transmit level across the transmitting antennas, identifying the correct Doppler gate in the power peak grid requires unoccupied positions, thus creating an identifiable pattern. Alternatively, an identifiable pattern can also be generated by having the power peaks exhibit different levels (through different transmit levels on the respective transmitting antennas). For example, with MTX = 32 transmitting antennas, the modulation period P = 32 can still be used, with full transmit power and the modulation velocities p(rriTx) according to equation (11) for the first 12 transmitting antennas, and the remaining modulation velocities and half transmit power for the remaining 20 transmitting antennas.The non-coherent TX integration is then performed across all 32 positions in the grid, with the 20 positions corresponding to transmitting antennas at half power weighted at half the weight of the 12 positions corresponding to transmitting antennas at full power. The resulting occupancy vector b(n) then has the values ​​1 and 0.5. The line values ​​after non-coherent TX integration remain relatively unchanged (and thus also the identification of the correct Doppler gate) if a constant value is subtracted from the weights—for example, in the example under consideration, the value 0.5—resulting in an occupancy vector with only 12 non-zero values. 202505405.

[0265] 50

[0266] When using two levels for the transmit levels of the transmitting antennas and in the above case that the modulation period P corresponds to the number of transmitting antennas (which represents the smallest possible value for P), the most robust identification of the correct Doppler gate is achieved when both levels occur equally often (in this sense, the above example is therefore not quite optimal).

[0267] To increase the robustness of identifying the correct Doppler gate, more than two different transmit levels can be used, and the modulation period P can be designed to be larger than the number of transmitting antennas (for example, P = 48 with 32 transmitting antennas, resulting in 16 zeros in the occupancy vector).

[0268] To make the identification of the correct Doppler gate as robust as possible against noise contained in the received signals, the weighting used in the example above is optimal: the weights are chosen proportionally to the transmit level of the respective transmitting antenna.

[0269] The average transmit level across all frequency ramps is relevant for the magnitude of the power peaks. A reduction in the average transmit level can be achieved not only by an equal level reduction across all frequency ramps, but also by omitting transmission on some frequency ramps. However, this results in multiple power peaks per transmitting antenna in the Doppler spectrum (two power peaks occur if transmission only occurs on every other frequency ramp), which in principle necessitates a longer modulation period. This approach is primarily interesting when only binary phase shifters are available (i.e., only phases 0° and 180° are possible) in order to synthesize further linear phase profiles (phase profile 0, 90°, 180°, 270°, 0°,... with every other frequency ramp muted, requires only phases 0° and 180°).

[0270] In the antenna arrangement shown in Fig. 1b, the transmitting antennas are located in different planes when viewed vertically. Due to superposition effects caused by reflections off a road surface, the signal components originating from different planes can vary in level (because the object is seen from different directions due to the reflections, and the corresponding signal components superimpose) – thus, 202505405

[0271] 51

[0272] This is solely due to the different transmit levels resulting in different receive levels. Consequently, the level variation of the power peaks used for non-coherent TX integration does not correspond to the expected pattern, which can lead to errors in identifying the correct Doppler gate. To reduce or completely prevent the probability of such errors, the transmit levels of the respective transmitting antennas should differ in as many planes as possible, preferably such that the diversity of the transmit levels is similar in as many or all planes as possible.

[0273] Antenna arrays are often used in which several transmitting antennas are arranged in a grid at constant spacing. In such cases, particularly when dealing with multiple objects at the same distance and with the same relative speed, it can be advantageous for the distribution of the different transmission levels across these antennas to be irregular.

[0274] Some antenna arrays have several similar groups of transmitting antennas, especially antenna pairs (which are used differently in several signal processing steps, for example, firstly as a common, i.e., larger, antenna, and secondly for phase comparison of the signal components generated by the individual antennas). It can be advantageous to make the different transmit levels of the antennas in each group at least approximately the same.

[0275] A potential disadvantage is that deliberately different transmission levels of the transmitting antennas mean the sensor's full sensitivity potential is not utilized (because some transmitting antennas do not use their full transmission power). However, this is not critical, or only slightly so, for sensors with many transmitting antennas, as they already possess very high sensitivity, and a reduction in the overall transmission power may be necessary for regulatory reasons. Furthermore, reducing the transmission power also reduces the sensor's power consumption (less transmission power requires less electrical power to generate).

[0276] In the modulation considered so far according to Fig. 2, all frequency ramps have the same frequency position, i.e., starting frequency (and thus center frequency). For improved distance resolution, as described in DE 102020 210 079 B3 202505405

[0277] 52

[0278] As described, the starting frequency is changed linearly across the frequency ramps, with the spacing of the frequency ramps preferably also changing linearly. This has no effect on the approaches according to the invention and their advantages; they remain unchanged. The only difference is that the Doppler gate of an object is then not only determined by the relative velocity of the object, but also has a contribution from the object's distance, which increases linearly with this distance (due to a center frequency changing linearly across the frequency ramps, the phase changes linearly, since the number of wave trains that fit into the beam path to the object and back changes, and this change is more pronounced the longer the beam path is).

[0279] So far, a linear frequency ramp has been considered as the transmitted signal, which is repeated sequentially. However, other signal forms can be used instead of a frequency ramp, for example, a signal with pseudorandom binary phase modulation (i.e., the sign is changed very rapidly and pseudorandomly within the signal) or an OFDM signal (OFDM = Orthogonal Frequency Division Multiplexing). The sampled values ​​(after analog-to-digital conversion) cannot then be used directly as input values ​​for the first DFT described above; instead, the Fourier transform must first be calculated for each transmitted signal and receiving antenna and divided by the spectrum of the transmitted signal.

[0280] Finally, it should be noted that it is obvious to a person skilled in the art how the considerations and explanations according to the invention, illustrated by the above application examples, can be transferred to general dimensions and parameter designs, i.e., they can also be applied to other numerical values.

Claims

1. 202505405 2.53 3. Patent claims 1.

1. Method for a radar system for environmental sensing with 5. Transmitting devices with several parallel transmitting antennas for radiating transmitting signals, which contain one or more sequences of K individual signals whose general form is preferably the same or similar, wherein their frequency position can change, in particular, linearly; - means for changing the phase position of the transmitted individual signals, by which a phase response different for the transmitting antennas is realized over the K individual signals, wherein this difference in the phase response to different transmitting antennas over the K individual signals is at least approximately linear, optionally apart from phase jumps due to the phase uniqueness range of 2π, 6. Receiving means with one or more receiving antennas for receiving transmitted signals reflected from objects, wherein the signals received from an object are composed of components originating from the various transmitting antennas, wherein these components include a phase response that differs linearly across the K individual signals as well as a similar phase progression, which is generated in particular by the relative movement of the object and, in the case of a frequency shift across the K individual signals, also depends on the object distance, 7. and signal processing equipment for processing the received signals, 8. characterized by the fact that 9. - not all transmitting antennas radiate the same average transmission level, - a non-coherent integration is performed over signal components whose phase responses over the K individual signals lie in the expected grid of the signal components of an object originating from different transmitting antennas and exhibiting linearly different phase responses, which is referred to below as non-coherent TX integration, wherein this non-coherent TX integration is performed multiple times under assumption 202505405 10.54 11. different radial relative velocities and, if applicable, distances of an object are taken, 12. In non-coherent TX integration, signal components are weighted more heavily the higher the radiated transmit level of the respective assigned transmitting antenna is. 13.- and the result of multiple non-coherent TX integration is used to identify the phase progression resulting from the object's relative velocity and, if applicable, object's distance, and / or a quantity derived therefrom.

2. The method according to claim 1, wherein, in the case of non-coherent TX integration, the weighting of the signal components correlates with the mean transmit level of the respective assigned transmitting antenna, i.e., is in particular proportional to the mean transmit power or transmit amplitude of the respective assigned antenna, optionally less a constant value, in order to expediently achieve the most robust possible identification of the phase progression resulting from the object relative velocity and optionally object distance and / or a quantity derived therefrom, in particular in the case that contributions from several objects and / or significant noise are present in the signals.

3. Method according to claim 1 or 2, characterized in that, prior to the non-coherent TX integration via the received signals to multiple receiving antennas or via signals derived from them, a coherent integration, in particular in the form of digital beamforming, hereinafter referred to as RX beamforming, is carried out and the result of the multiple non-coherent TX integration of different RX beamforming directions is used to identify the phase progression resulting from the object relative velocity and, if applicable, object distance and / or a quantity derived therefrom. 202505405 16.55 4. Method according to claim 1 or 2, characterized in that a non-coherent integration is performed over the received signals to several receiving antennas or signals derived from them, which expediently takes place before the non-coherent TX integration, and the result of the entire non-coherent integration, performed multiple times under the assumption of different radial relative velocities and optionally distances of an object, is used to identify the phase progression resulting from the object's relative velocity and optionally object distance and / or a quantity derived therefrom.

5. Method according to one of the above claims, wherein the incoherent integration is performed over power values ​​or over magnitude values ​​of the signal components.

6. Method according to any one of the above claims, wherein 20. The K individual signals preferably lie in an equidistant grid, at least approximately.

21. A discrete Fourier transform of length L is performed on their K received signals or signals derived from them, if necessary after extension with zeros, i.e. a so-called zero-padding, wherein the L frequency reference points of the discrete Fourier transform are referred to below as Doppler gates, 22. The phase response across the K individual signals contains a different linear component for each transmitting antenna, i.e., with a different slope, which causes the signal components from the different transmitting antennas to lie at different Doppler gates in the discrete Fourier transform.

23. The non-coherent TX integration is performed via Doppler gates in the grid expected by the different slopes of the linear components, 24.- and this incoherent TX integration for different layers of this grid, i.e., different, especially all Doppler gates as the first 202505405 25.56 26. Raster point and thus for various, in particular all possible object relative velocities and, if necessary, object distances.

7. Method according to claim 6, wherein the linear components of the phase responses, when mapped to the phase uniqueness range of 0…2π, have a common period P, which preferably represents an integer divisor of the number L of Doppler gates, whereby the Doppler gates corresponding to the transmitting antennas represent a subset of an equidistant grating with a distance L / P.

8. Method according to claim 7, wherein the summation necessary for non-coherent TX integration for preferably all P grid layers in an equidistant grid with integer spacing L / P is realized by performing a cyclic correlation between a vector with the P values ​​of the discrete Fourier transform in this grid and an occupancy vector of length P, wherein the values ​​of the occupancy vector for all indices which correspond to the linear phase change of a transmitting antenna within one period P as multiples of 2π, assume non-zero values ​​and are otherwise 0, and wherein this cyclic correlation can preferably be performed as a fast correlation, i.e., by multiplying two spectra in the frequency domain.

9. Method according to claim 8, wherein the non-zero values ​​of the occupancy vector correlate to the mean transmission level of the respective associated transmitting antenna, i.e., in particular, are proportional to the mean transmission power or transmission amplitude of the respective associated transmitting antenna, optionally less a constant value.

10. Method according to one of the above claims, in which different transmission levels via the transmitting antennas are caused by hardware-related effects such as different lengths of the antenna leads and / or are specifically controlled, in particular by a configurable 202505405 31.57 32. Transmit power generation is generated, in particular so that the period P can be chosen as small as possible, with the number of transmitting antennas representing the lower limit.

11. Method according to one of claims 6-10, wherein the results of the non-coherent TX integration of raster layers with spacings which are integer multiples of L / P are examined and in particular compared in order to identify objects as well as their relative velocity and, if applicable, distance, wherein preferably a detection threshold which depends in particular on the noise level is also used and, when using an RX beamforming, this is done in all its beam directions.

12. Method according to claim 11, wherein for object identification the grid position is used that of those in the grid at a distance L / P in which the incoherent TX integration assumes the highest amount and thus the highest performance.

13. Method according to one of claims 6-12, in which, for object identification, grid layers in the grid with distance L / P are also used where the power of the non-coherent TX integration is not maximum, but is above a threshold which may depend on the distance of this grid layer to the grid layer of the maximum and in particular is at least a minimum distance above the value generated there by an object which is located at the grid layer of the maximum and generates the power level there.

14. Method according to one of the above claims, wherein the transmitting antennas are located in different planes when viewed vertically and the transmit levels of the respective transmitting antennas differ in at least some planes, preferably such that the diversity of the transmit levels is similar in as many or as many planes as possible, thereby reducing the influence of superposition effects caused by reflections on a road surface. 202505405 37.58 15. Method according to one of the above claims, wherein there are several similar groups, in particular pairs of transmitting antennas, wherein in each group the different transmit levels of their antennas are at least approximately equal.

16. A method according to one of the above claims, wherein the individual transmitted signals are linearly frequency-modulated, their center frequency optionally changing successively and preferably linearly, or representing OFDM signals or being generated with pseudorandom fast phase modulation, in particular characterized in that a signal separation into gates corresponding to different distances, so-called distance gates, is realized from the received values ​​resulting for each individual transmitted signal by means of a transformation, and the procedures described in the above claims are applied in the different distance gates.

17. Radar system for environmental sensing with 41. Transmitting devices with several parallel transmitting antennas for radiating transmitting signals, which contain one or more sequences of K individual signals whose general form is preferably the same or similar, wherein their frequency position can change, in particular, linearly; - means for changing the phase position of the transmitted individual signals, by which a phase response different for the transmitting antennas is realized over the K individual signals, wherein this difference in the phase response to different transmitting antennas over the K individual signals is at least approximately linear, optionally apart from phase jumps due to the phase uniqueness range of 2π, 42. Receiving devices with one or more receiving antennas for receiving transmitted signals reflected from objects, wherein the signals received from an object are composed of components originating from the various transmitting antennas, these components having a phase response that differs linearly across the individual signals 202505405 43.59 44. as well as a similar phase progression, which is generated in particular by the relative motion of the object and, in the case of a change in frequency position via the K individual signals, also depends on the object distance, 45.- and signal processing equipment for processing the received signals, 46. ​​characterized by the fact that 47. - not all transmitting antennas radiate the same average transmission level, - a non-coherent integration is carried out over signal components whose phase responses over the K individual signals lie in the expected grid of the signal components of an object originating from different transmitting antennas and exhibiting linearly different phase responses, which is referred to below as non-coherent TX integration, wherein this non-coherent TX integration is carried out several times under assumption of different radial relative velocities and, if necessary, distances of an object, 48. In non-coherent TX integration, signal components are weighted more heavily the higher the radiated transmit level of the respective assigned transmitting antenna is, 49.- and the result of multiple non-coherent TX integration is used to identify the phase progression resulting from the object's relative velocity and, if applicable, object's distance, and / or a quantity derived therefrom.

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