Electric power conversion method and electric power conversion device

Abstract

An electric power conversion device (20) comprises: inverter circuits (2A, 2B) for converting DC electric power from a DC power supply (4) into n-phase (n≥3) AC electric power, and driving an AC load (1; a smoothing capacitor (3) that is provided between the DC electric power supply (4) and the inverter circuits (2A, 2B); and a control unit (7) for driving the inverter circuits (2A, 2B). The control unit (7) is configured to calculate a modulation wave phase (θmu) on the basis of the operation state (θre) of the AC load (1) and a voltage command value that corresponds to the operation of the AC load (1), and perform control so as to change the carrier phase of a predetermined phase among the n carrier phases (Su, Sv, Sw) in a specific pattern on the basis of the modulation wave phase (θmu).

Description

Power Conversion Method and Power Conversion Device

[0001] The present invention relates to a power conversion method and a power conversion device.

[0002] Patent Document 1 describes a power conversion device applied to a motor control device that drives a multiphase motor using an inverter circuit. In a power conversion device such as that of Patent Document 1, a smoothing capacitor is provided between a DC power supply and an inverter circuit in order to reduce fluctuations in the input voltage of the inverter circuit during switching of the switching elements of the inverter circuit and to stably control the power supply to the load.

[0003] Japanese Unexamined Patent Application Publication No. 2009-232546

[0004] For example, a ripple current with a peak value of about 100 A flows through the smoothing capacitor, and losses occur due to the internal resistance of the smoothing capacitor. When the temperature of the smoothing capacitor rises due to this loss, the durability of the smoothing capacitor decreases, leading to a decrease in its lifespan.

[0005] Therefore, in the power conversion device of Patent Document 1, according to the operating point of the multiphase motor (power conversion), the phase difference between the carrier signals of each phase used for pulse width modulation (PWM) control is selectively and uniquely determined and changed, thereby reducing the ripple current flowing through the smoothing capacitor. However, depending on the form of power conversion and the operating point of power conversion, when the phase difference between the carrier signals of each phase is fixed, overlap occurs in the ripple current of each phase, resulting in a large peak value of the ripple current. Therefore, with the method of uniquely determining the phase difference between the carrier signals of each phase according to the operating point of power conversion as shown in Patent Document 1, it may not be possible to obtain a sufficient effect of reducing the peak value of the ripple current.

[0006] The present invention has been made in view of the above problems, and an object thereof is to provide a power conversion method and a power conversion device capable of reducing the peak value of the ripple current flowing through a smoothing capacitor.

[0007] A power conversion device according to one aspect of the present invention comprises an inverter circuit that converts DC power from a DC power source into n-phase (n≧3) AC power, a smoothing capacitor provided between the DC power source and the inverter circuit, and a control unit that drives the inverter circuit. The control unit is configured to calculate the modulated wave phase based on a voltage command value corresponding to the operation of the AC load and the operating state of the AC load, and to control the carrier phase of a predetermined phase among the n-phase carrier phases to change in a specific pattern based on the modulated wave phase.

[0008] According to the present invention, it is possible to provide a power conversion method and a power conversion device that can reduce the peak value of the ripple current flowing through a smoothing capacitor.

[0009] Figure 1 is a block diagram showing an overview of a motor drive device to which the power conversion device according to the embodiment is applied. Figure 2 is a block diagram showing an example of the configuration of the carrier phase selection unit shown in Figure 1. Figure 3 is a diagram showing an example of the logic for selecting the carrier phase in the embodiment. Figure 4A is a diagram showing the calculated waveforms of the ripple current of each phase flowing through the smoothing capacitor in the embodiment. Figure 4B is a diagram showing the calculated waveform of the total ripple current flowing through the smoothing capacitor in the embodiment. Figure 5A is a diagram showing the calculated waveforms of the ripple current of each phase flowing through the smoothing capacitor in the first comparative example. Figure 5B is a diagram showing the calculated waveform of the total ripple current flowing through the smoothing capacitor in the first comparative example. Figure 6A is a diagram showing the calculated waveforms of the ripple current of each phase flowing through the smoothing capacitor in the second comparative example. Figure 6B is a diagram showing the calculated waveform of the total ripple current flowing through the smoothing capacitor in the second comparative example.

[0010] The embodiments will be described with reference to the drawings. In the drawings, the same parts are denoted by the same reference numerals and their descriptions are omitted.

[0011] Referring to Figure 1, the configuration of the power converter 20 according to the embodiment will be described. In this embodiment, the power converter 20 will be described as an example of a case in which the power converter 20 is applied as a motor control device that drives a motor 1 as an open-winding motor as a multiphase AC load. The motor 1 is an n-phase electric motor, and in this embodiment, a three-phase electric motor with n=3 phases will be described as an example, but it is not limited to this, and the power converter 20 can be applied to electric motors with any number of phases, such as n=5, n=7, etc.

[0012] The power converter 20 includes a first inverter circuit 2A and a second inverter circuit 2B to which a DC power supply 4 is connected, a smoothing capacitor 3, a resolver 5, a control unit 7, and an ammeter side unit 17, all located between the DC power supply 4 and the first and second inverter circuits 2A and 2B. Since the motor 1 is an open-wound motor, two first inverter circuits 2A and 2B are used, but if the power converter 20 is applied to a multi-phase motor drive system other than an open-wound motor, one inverter circuit may be used. The DC power supply 4 is, for example, a rechargeable secondary battery such as a lithium-ion battery. The smoothing capacitor 3 smooths the voltage supplied from the DC power supply 4 to the first and second inverter circuits 2A and 2B and removes noise, etc.

[0013] The first inverter circuit 2A and the second inverter circuit 2B each convert the DC power from the DC power supply 4 into n-phase (three-phase in this embodiment) AC power to drive an n-phase motor 1 having n coils. The first end of the motor 1 is connected to the output terminal of the first inverter circuit 2A, and the second end of the motor 1 is connected to the output terminal of the second inverter circuit 2B. The control unit 7 controls the driving of the motor 1 via the first inverter circuit 2A and the second inverter circuit 2B.

[0014] Each of the first inverter circuits 2A and 2B has, for example, n arms (not shown) corresponding to n coils (not shown) of the motor 1. Each arm is connected in parallel to a DC power supply 4 and a smoothing capacitor 3. Each arm is configured by connecting two semiconductor switching elements, such as IGBTs and MOSFETs, in series, and the midpoint of the two semiconductor switching elements connected in series is connected to the corresponding phases (here, U-phase, V-phase, and W-phase) of the coils of the motor 1. The control electrode (gate) of each semiconductor switching element is connected to a control unit 7. The motor 1 is driven by controlling each semiconductor switching element with a gate signal from the control unit 7. Since the first inverter circuits 2A and 2B with such configurations can use known inverter circuits such as the inverter circuit described in Patent Document 1, for example, a more detailed explanation is omitted.

[0015] The resolver 5 measures the rotor position θre, which is the operating state of the motor 1, and feeds back the measured value of the rotor position θre to the control unit 7. The ammeter side unit 17 measures the phase currents flowing through each phase of the motor 1, namely the U-phase current Iu, the V-phase current Iv, and the W-phase current Iw, and feeds back the measured values ​​of the phase currents Iu, Iv, and Iw to the control unit 7.

[0016] The control unit 7 is a general-purpose computer equipped with a CPU (Central Processing Unit), memory, and an input / output unit. A computer program (control program) is installed in the control unit 7. By executing the computer program, the control unit 7 performs various functions.

[0017] The control unit 7 is a system that performs current control to realize the desired torque command value from the torque command unit 6 in the motor 1, and performs feedback control to supply the motor 1 with phase currents Iu, Iv, and Iw necessary to generate a predetermined torque. The control unit 7 comprises a current command unit 8, an n-phase / dq0 conversion unit 9, a current control unit 10, a dq0 / n-phase conversion unit 11, a carrier phase selection unit 12, a triangular wave generation unit 13, and a PWM generation unit 14.

[0018] The torque command value from the torque command unit 6 is input to the current command unit 8. The current command unit 8 converts the torque command value into current command values ​​on the dq0 axis, namely the d-axis current command value Id*, the q-axis current command value Iq*, and the 0-axis current command value I0*, and inputs them to the current control unit 10. The rotor position θre of the motor 1 measured by the resolver 5 and the phase currents Iu, Iv, and Iw measured by the ammeter side unit 17 are input to the n-phase / dq0 conversion unit 9. The n-phase / dq0 conversion unit 9 converts the measured phase currents Iu, Iv, Iw and rotor position θre into coordinates corresponding to the measured currents on the dq0 axis, namely the d-axis measured current Id, the q-axis measured current Iq, and the 0-axis measured current I0, and inputs them to the current control unit 10.

[0019] The current control unit 10 calculates the voltage command values ​​on the dq0 axis, namely the d-axis voltage command value Vd*, the q-axis voltage command value Vq*, and the 0-axis voltage command value V0*, from the difference between the current command values ​​Id*, Iq*, and I0* on the dq0 axis and the measured currents Id, Iq, and I0, using PI control or the like. The calculated voltage command values ​​Vd*, Vq*, and V0* on the dq0 axis are input to the dq0 / n-phase conversion unit 11 and the carrier phase selection unit 12.

[0020] The dq0 / n-phase conversion unit 11 performs a coordinate transformation of the voltage command values ​​Vd*, Vq*, and V0* on the dq0 axis into n-phase voltage command values ​​Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*, and inputs them to the PWM generation unit 14. The carrier phase selection unit 12 determines the optimal carrier phases Su, Sv, and Sw for each phase based on the d-axis voltage command value Vd*, the q-axis voltage command value Vq*, and the U-phase modulated wave phase θmu calculated from the rotor position θre, and inputs them to the triangular wave generation unit 13. The triangular wave generation unit 13 generates three-phase triangular wave carrier signals Cu, Cv, and Cw based on the carrier phases Su, Sv, and Sw, and inputs them to the PWM generation unit 14.

[0021] The PWM generation unit 14 performs a triangular wave comparison using the n-phase voltage command values ​​Vu*, Vv*, Vw* and the triangular wave carrier signals Cu, Cv, Cw to generate the first gate signals Gu1, Gv1, Gw1 and the second gate signals Gu2, Gv2, Gw2, which are PWM-controlled pulse waves. The first gate signals Gu1, Gv1, Gw1 control the switching operation of the first inverter circuit 2A, and the second gate signals Gu2, Gv2, Gw2 control the switching operation of the second inverter circuit 2B. The first gate signals Gu1, Gv1, Gw1 and the second gate signals Gu2, Gv2, Gw2 are pulse signals modulated from voltage sinusoidal wave commands with the same modulation rate but shifted phase. Phase currents Iu, Iv, Iw are supplied to each coil of the motor 1 from the first inverter circuit 2A and the second inverter circuit 2B.

[0022] Next, the operation of the carrier phase selection unit 12 will be explained in more detail using Figure 2. Figure 2 is a block diagram showing an example of the configuration of the carrier phase selection unit 12 shown in Figure 1. In the configuration shown in Figure 2, the carrier phase selection unit 12 comprises a voltage phase calculation unit 15 and a phase selection logic unit 16. The carrier phase selection unit 12 calculates the voltage phase θvol from the d-axis voltage command value Vd* and the q-axis voltage command value Vq* using the voltage phase calculation unit 15, and calculates the U-phase modulated wave phase θmu by adding the rotor position θre measured by the resolver 5 to the voltage phase θvol. Note that the carrier phase selection unit 12 does not only calculate the U-phase modulated wave phase θmu, but may also calculate the V-phase modulated wave phase θmv or the W-phase modulated wave phase θmw.

[0023] The U-phase modulated wave phase θmu is input to the phase selection logic unit 16. The phase selection logic unit 16 controls the carrier phases Su, Sv, and Sw to change in a specific pattern according to the U-phase modulated wave phase θmu. For example, the phase selection logic unit 16 controls two of the three carrier phases Su, Sv, and Sw to be in phase with the reference carrier, and one of the carrier phases to have a phase difference with respect to the reference carrier, according to the value of the U-phase modulated wave phase θmu. The one carrier phase that has a phase difference with respect to the reference carrier is controlled to switch in a specific pattern according to the value of the U-phase modulated wave phase θmu. For example, the one carrier phase that has a phase difference with respect to the reference carrier is controlled to switch sequentially according to the value of the U-phase modulated wave phase θmu.

[0024] By controlling the carrier phase of a predetermined phase among the n-phase carrier phases Su, Sv, and Sw to change in a specific pattern based on the U-phase modulated wave phase θmu, the peak value of the total ripple current Ir flowing through the smoothing capacitor 3 can be reduced. For example, by controlling one of the carrier phases having a phase difference with respect to the reference carrier to switch in a specific pattern according to the value of the U-phase modulated wave phase θmu, the peak value of the total ripple current Ir flowing through the smoothing capacitor 3 can be reduced.

[0025] Instead of controlling all carrier phases Su, Sv, and Sw to have an equal phase difference with respect to the reference carrier, for example, the carrier phases Su and Sv of the U and V phases are set to be in phase with the reference carrier, while the W phase carrier phase Sw is given a phase difference with respect to the reference carrier. This has the effect of reducing the peak value of the total ripple current Ir.

[0026] Furthermore, the system may be configured to operate when the modulation rate determined by current control exceeds a predetermined threshold. That is, when the modulation rate exceeds a predetermined threshold (for example, 50%), the system may be controlled to change the carrier phase of a predetermined phase among the carrier phases Su, Sv, and Sw in a specific pattern based on the U-phase modulated wave phase θmu. When the modulation rate is below the predetermined threshold (for example, 50%), the system may be controlled not to change the carrier phases Su, Sv, and Sw.

[0027] When the modulation rate exceeds a predetermined threshold (for example, 50%) and the pulse widths of the ripple currents Iur, Ivr, and Iwr flowing through the smoothing capacitor 3 become wider, the carrier phases Su, Sv, and Sw are changed. This maximizes the effect of reducing the peak value of the total ripple current Ir.

[0028] Figure 3 shows an example of the logic by which the phase selection logic unit 16 selects the carrier phases Su, Sv, and Sw of each phase according to the value of the U-phase modulated wave phase θmu. Here, two types of carriers are provided: a reference carrier and a carrier with a phase difference of 90° (π / 2 radians) from the reference carrier, and the logic is set up to decide which one to use for each phase. In other words, in a motor 1 in steady operation, the logic is such that the switching timing of only one of the three phases is shifted every 30° (π / 6 radians) of electrical angle.

[0029] In detail, when the U-phase modulated wave phase θmu is between π / 6 and 3π / 6 radians, the U-phase carrier phase Su and V-phase carrier phase Sv are used as reference carriers, and the W-phase carrier phase Sw is defined as a carrier with a phase difference of 90° (π / 2 radians) relative to the reference carrier. When the U-phase modulated wave phase θmu is between 3π / 6 and 5π / 6 radians, the V-phase carrier phase Sv and W-phase carrier phase Sw are used as reference carriers, and the U-phase carrier phase Su is defined as a carrier with a phase difference of 90° (π / 2 radians) relative to the reference carrier. When the U-phase modulated wave phase θmu is between 5π / 6 and 7π / 6 radians, the W-phase carrier phase Sw and U-phase carrier phase Su are used as reference carriers, and the V-phase carrier phase Sv is defined as a carrier with a phase difference of 90° (π / 2 radians) relative to the reference carrier. When the U-phase modulated wave phase θmu is between 7π / 6 and 9π / 6 radians, the U-phase carrier phase Su and V-phase carrier phase Sv are used as reference carriers, and the W-phase carrier phase Sw is defined as a carrier with a phase difference of 90° (π / 2 radians) relative to the reference carrier. When the U-phase modulated wave phase θmu is between 9π / 6 and 11π / 6 radians, the V-phase carrier phase Sv and W-phase carrier phase Sw are used as reference carriers, and the U-phase carrier phase Su is defined as a carrier with a phase difference of 90° (π / 2 radians) relative to the reference carrier. When the U-phase modulated wave phase θmu is between 11π / 6 and 12π / 6, and between 0 and π / 6 radians, the W-phase carrier phase Sw and U-phase carrier phase Su are used as reference carriers, and the V-phase carrier phase Sv is defined as a carrier with a phase difference of 90° (π / 2 radians) relative to the reference carrier.

[0030] In addition to the example in Figure 3, various other methods can be considered, such as dividing the interval of the U-phase modulated wave phase θmu into finer segments or defining the logic continuously in a function-like form. For example, in the case of the number of phases n = 5, the logic may be such that the switching timing of only one of the five phases is shifted, or the logic may be such that the switching timing of two of the five phases is shifted. For any number of phases n, the logic may be such that the switching timing of a number of phases less than n / 2 is shifted.

[0031] Next, using Figures 4A and 4B, we will explain the ripple currents Iur, Ivr, Iwr for each phase and the total ripple current Ir flowing through the smoothing capacitor 3 when the power conversion device 20 according to the embodiment is applied as a motor control device to drive the motor 1 as an open-winding motor. Figure 4A is a diagram showing the calculated waveforms of the ripple currents Iur, Ivr, Iwr for each phase flowing through the smoothing capacitor 3 in the power conversion device 20 according to the embodiment. Figure 4B is a diagram showing the calculated waveform of the total ripple current Ir = Iur + Ivr + Iwr flowing through the smoothing capacitor 3 in the power conversion device 20 according to the embodiment.

[0032] Furthermore, for comparison, Figures 5A and 5B show the first comparative example in which the switch timings of all three phases are synchronized, and Figures 6A and 6B show the second comparative example in which interleaving is performed to create a phase difference of 120° (2π / 3 radians) between the three phases. Figure 5A is a diagram showing the calculated waveforms of the ripple currents Iur, Ivr, and Iwr of each phase flowing through the smoothing capacitor 3 in the power converter 20 operated in the first comparative example. Figure 5B is a diagram showing the calculated waveform of the total ripple current Ir flowing through the smoothing capacitor 3 in the power converter 20 operated in the first comparative example. Figure 6A is a diagram showing the calculated waveforms of the ripple currents Iur, Ivr, and Iwr of each phase flowing through the smoothing capacitor 3 in the power converter 20 operated in the second comparative example. Figure 5B is a diagram showing the calculated waveform of the total ripple current Ir flowing through the smoothing capacitor 3 in the power converter 20 operated in the second comparative example.

[0033] In the embodiments shown in Figures 4A and 4B, the first comparative example shown in Figures 5A and 5B, and the second comparative example shown in Figures 6A and 6B, the operating point of the motor 1 on the horizontal axis (time axis) is the same. Furthermore, the calculation results are shown for a modulation rate around 50%, where the total ripple current Ir to the smoothing capacitor 3 is large.

[0034] As shown in Figure 5A, in the first comparative example where the switch timings of all three phases are synchronized, the peak value of the total ripple current Ir is approximately 200A, as shown in Figure 5B. Furthermore, in the second comparative example where interleaving is performed to create a phase difference of 120° (2π / 3 radians) between the three phases, as shown in Figure 6A, the peak value of the total ripple current Ir is approximately 175A, as shown in Figure 6B. Thus, it can be seen that in the second comparative example, the peak value of the total ripple current Ir is not reduced as much as in the first comparative example. This is because when the modulation rate is around 50%, the pulse width of the ripple currents Iur, Ivr, and Iwr of each phase becomes wider, causing overlap between the ripple currents Iur, Ivr, and Iwr of each phase.

[0035] On the other hand, when the embodiments shown in Figures 4A and 4B are applied, the peak value of the total ripple current Ir can be reduced to about 100A. This is because, in the embodiments, the waveforms of the ripple currents of the two phases with small current values ​​are allowed to overlap, and the waveform of the ripple current of only the phase with a large current value is controlled to continuously shift the pulse phase by 90° from the other two phases. Looking at the waveforms of the three phase ripple currents Iur, Ivr, and Iwr in Figure 4A, from 0ms to 0.1ms, the ripple current Ivr of the V phase is controlled to shift the pulse phase by 90° relative to the ripple currents Iwr and Iur of the other phases. Similarly, from 0.1ms to 0.3ms, the ripple current Iwr of the W phase is controlled to shift the pulse phase, and from 0.3ms onward, the ripple current Iur of the U phase is controlled to shift the pulse phase.

[0036] Thus, in this embodiment, the carrier phase of a predetermined phase among the carrier phases Su, Sv, and Sw is controlled to change in a specific pattern based on the U-phase modulated wave phase θmu. As a result, in this embodiment, a greater reduction effect on the peak value of ripple current can be obtained than, for example, the method of providing a phase difference as in the second reference example.

[0037] Furthermore, according to the embodiment, since the power conversion device 20 is applied to a motor control device that drives the motor 1 as an open-winding motor, the peak value of the total ripple current Ir of the smoothing capacitor 3 can be reduced when the motor 1 is driven.

[0038] In this embodiment, the power converter 20 was described as being used as a motor control device to drive a motor 1 as an open-winding motor, but it is not limited to this. For example, the same effects can be obtained when the power converter 20 is applied to multi-phase motor drive systems other than open-winding motors that perform DC / AC conversion, or to power conversion systems such as grid-connected systems.

[0039] As described above, embodiments of the present invention have been presented, but the statements and drawings that constitute part of this disclosure should not be understood as limiting the invention. Various alternative embodiments, examples, and operational techniques will become apparent to those skilled in the art from this disclosure.

[0040] 1. Motor 2A. First inverter circuit 2B. Second inverter circuit 3. Smoothing capacitor 4. DC power supply 5. Resolver 6. Torque command unit 7. Control unit 8. Current command unit 9. n-phase / dq0 conversion unit 10. Current control unit 11. dq0 / n-phase conversion unit 12. Carrier phase selection unit 13. Triangular wave generation unit 14. PWM generation unit 15. Voltage phase calculation unit 16. Phase selection logic unit 17. Ammeter side unit 20. Power conversion device

Claims

1. A power conversion method for a power conversion device for driving an n-phase AC load, comprising an inverter circuit that converts DC power from a DC power supply into n-phase (n≧3) AC power, and a smoothing capacitor provided between the DC power supply and the inverter circuit, the power conversion method comprising: generating an n-phase voltage command value corresponding to the operation of the AC load; driving the inverter circuit based on the voltage command value and a reference carrier; calculating the modulated wave phase based on the voltage command value and the operating state of the AC load; and controlling the carrier phase of a predetermined phase among the n-phase carrier phases to change in a specific pattern based on the modulated wave phase.

2. The power conversion method according to claim 1, wherein when the modulation rate exceeds a predetermined threshold, the carrier phase of the predetermined phase is controlled to change in a specific pattern based on the modulated wave phase, and when the modulation rate is below a predetermined threshold, the carrier phase of the predetermined phase is not changed.

3. The power conversion method according to claim 1 or 2, wherein the carrier phase of the predetermined phase is controlled to have a phase difference with respect to the reference carrier, the carrier phase of the other phases is controlled to be in phase with the reference carrier, and the carrier phase of the predetermined phase is controlled to switch between the carrier phases of the n phases in a specific pattern based on the modulated wave phase.

4. A power conversion method according to claim 1 or 2, further comprising: generating an n-phase carrier signal from an n-phase carrier phase controlled to change the carrier phase of a predetermined phase in a specific pattern; generating a pulse-width modulated n-phase gate signal for driving the inverter circuit from the voltage command value and the n-phase carrier signal; and driving the inverter circuit with the n-phase gate signal.

5. A power conversion device for driving an n-phase AC load, comprising: an inverter circuit that converts DC power from a DC power source into n-phase (n≧3) AC power to drive the AC load; a smoothing capacitor provided between the DC power source and the inverter circuit; and a control unit that drives the inverter circuit, wherein the control unit is configured to generate n-phase voltage command values ​​corresponding to the operation of the AC load, drive the inverter circuit based on the voltage command values ​​and a reference carrier, calculate the modulated wave phase based on the voltage command values ​​and the operating state of the AC load, and control the carrier phase of a predetermined phase among the n-phase carrier phases to change in a specific pattern based on the modulated wave phase.

6. The power conversion device according to claim 5, wherein the control unit is configured to generate an n-phase carrier signal from an n-phase carrier phase controlled to change the carrier phase of a predetermined phase in a specific pattern, generate a pulse-width modulated n-phase gate signal for driving the inverter circuit from the voltage command value and the n-phase carrier signal, and drive the inverter circuit with the n-phase gate signal.

7. The power conversion device according to claim 5 or 6, wherein the AC load is a motor.

8. The power conversion device according to claim 7, wherein the motor is an open-winding motor.

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