Discharging of a damping capacitor

Abstract

The present disclosure relates to a circuit (100) including: a supply terminal (102) configured to receive a supply voltage (104); a driver circuit (106) including an input terminal (108) coupled with the supply terminal (102) to receive the supply voltage (104) as input voltage for powering an operation of the driver circuit (106); a damping capacitor (114) coupled in parallel with the input terminal (108) and ground (116) such that the damping capacitor (114) is charged by the supply voltage (104); wherein the driver circuit (106) further comprises a discharge current generator circuit (130) configured to cause a flow of a discharge current through a discharge path (132) if an input voltage value of the input voltage at the input terminal (108) fulfills an overvoltage criterion, thereby causing a discharge of the damping capacitor (114), wherein the discharge current generator circuit (130) is configured such that the discharge current has a behavior over time that is inversely related to a behavior of the input voltage at the input terminal (108).

Description

DISCHARGING OF A DAMPING CAPACITORTechnical Field

[0001] The present disclosure relates generally to a circuit configured to implement a scheme for discharging a damping capacitor in case of an overvoltage condition, and to methods thereof (e.g., a method of discharging a damping capacitor in case of an overvoltage condition)Background

[0002] In general, protection of electronic circuits from events that may deteriorate, damage, or even destroy their components is of utmost importance for preventing the risk of failures of electronic devices. In particular, unexpected increases in supply voltage (so-called, “voltage surges”) may cause an electronic circuit to receive a voltage that greatly exceeds the nominal specified voltage range for the circuit, thus potentially damaging or breaking sensitive components. The occurrence of voltage surges may be a problem of particular relevance in the automotive context, e.g. for electric vehicles, in which the battery unit powers up several electronic components and circuits. Various surge protection components have been developed, which may include variable resistors, transient voltage suppressors, damping capacitors, and the like. In general, a surge protection component absorbs or diverts the excess electrical energy, thus protecting the electronic circuit from sudden peaks in the supply voltage. Improvements in the field of surge protection components for protecting electronic circuits may be of particular relevance for the further advancements of several technologies.Brief Description of the Drawings

[0003] In the drawings, like reference characters generally refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead generally being placed upon illustrating the principles of the invention. In the following description, various aspects of the invention are described with reference to the following drawings, in which:FIG.1 A and FIG. IB show a circuit configured to implement a scheme for discharging a damping capacitor, in a schematic representation according to various aspects;FIG.1C shows a graph associated with the discharge of the damping capacitor, according to various aspects;FIG.2A and FIG.2B show a realization of the discharge current generator circuit of the driver circuit, in a schematic representation according to various aspects;FIG.3 A shows a realization of the discharge current generator circuit of the driver circuit including a translinear loop, in a schematic representation according to various aspects;FIG.3B shows graphs associated with the operation of the discharge current generator circuit of FIG.3A, according to various aspects;FIG.4 shows a realization of the discharge current generator circuit of the driver circuit including a translinear loop and a digital multiplexer, in a schematic representation according to various aspects;FIG.5 A and FIG.5B show a realization of the discharge current generator circuit of the driver circuit including a translinear loop and an analog multiplexer, in a schematic representation according to various aspects;FIG.5C shows graphs associated with the operation of the discharge current generator circuit of FIG.5 A and FIG.5B, according to various aspects; andFIG.6 shows an exemplary realization of the circuit of FIG.1A, in a schematic representation according to various aspects.Description

[0004] The following detailed description refers to the accompanying drawings that show, by way of illustration, specific details and aspects in which the invention may be practiced. These aspects are described in sufficient detail to enable those skilled in the art to practice the invention. Other aspects may be utilized and structural, logical, and electrical changes may be made without departing from the scope of the invention. The various aspects are not necessarily mutually exclusive, as some aspects may be combined with one or more other aspects to form new aspects.

[0005] In general, electronic circuits may include surge protection components to absorb or divert excess electrical energy, which may be delivered to the electronic circuit in case of a surge of supply voltage. Surge protection components may be of various types, e.g., including high-voltage transistors, voltage-dependent resistors (VDR), transient voltage suppressors (TVS), damping capacitors, and the like. In this context, damping capacitors offer several benefits, e.g. in terms of a relatively low cost, a relatively low surface utilization, and a high tunability to the specific requirements of a certain circuit. Damping capacitors may be usually disposed at the input terminal of an electronic circuit to reduce noise and provide protection against voltage spikes.

[0006] Unexpected surges in the supply voltage may occur in many different scenarios and due to many different causes, so that surge protection plays a role in a variety of differentapplications. Of particular interest is the protection of electronic circuits and components in a vehicle (e.g., in an electric car), which is a type of environment in which sudden spikes in supply voltage may cause the malfunction of components that are critical for the driver’s safety. In a vehicle, a voltage surge may be the result of a load being turned off, and of the regulator being slow in adapting the generated power, so that there is an excess of power being delivered to the other (still on) electronic circuits and components. In this context, long cable connections from the battery are at the root of extremely large voltage disturbances superimposed to the supply line. A voltage surge in a vehicle may bring the supply voltage to rather high values, e.g. 120 V, well above the safe operating range of electronic circuits and components commonly employed to implement various functionalities of the vehicle. A voltage surge may thus cause an overvoltage of the electronic circuit.

[0007] In a conventional configuration, an electronic circuit may include a damping capacitor and other surge protection components coupled with the input terminal at which the electronic circuit receives a supply voltage. Furthermore, the electronic circuit may be configured to implement a surge protection function according to which the circuit is turned off in case of an overvoltage condition at the input terminal, e.g., triggered by the voltage increase, and / or in case of a corresponding increase of the operating temperature of the circuit. In case of a voltage surge, the excess energy is absorbed by the surge protection components and the circuit is turned off to prevent the risk of damages to the circuit.

[0008] Aspects of the present disclosure may be based on the realization that in case of a voltage surge the damping capacitor becomes charged at a high voltage, so that even after the voltage surge has finished and the voltage returned back to a safe operating range, a high voltage remains at the input terminal of the electronic circuit. This behavior may occur in case a discharge of the capacitor towards the supply side is prevented by the configuration of the supply source. This may occur, for example, if the supply source is asymmetric, i.e. if the supply source has the capability of supplying current but has limited capability for sinking current. Illustratively, the supply source may allow flow of current in the forward direction from the supply source towards the electronic circuit (and the capacitor), but may prevent the flow of current in the reverse direction from the electronic circuit (and the capacitor) towards the supply source.

[0009] This situation may occur in a variety of configurations. As a relevant example, the discharge of the capacitor towards the supply side may be prevented by a filter element (e.g., a diode) that provides reverse protection by blocking the negative portion of the supply voltage. The filter element may allow a flow of current in the forward direction (from supply source tocapacitor) and prevent the flow of current in the reverse direction (from capacitor to supply source). As another example, the supply source may be configured such that the supply source may supply a high amount of current but may absorb (in other words, sink) only a small amount of current. Illustratively, the supply source may supply a greater amount of current compared to the amount of current that the supply source may absorb. For example, this may be the case for a low-dropout regulator.

[0010] Illustratively, in the depicted scenario, the increase in the supply voltage may charge the damping capacitor, and the damping capacitor may remain charged for a certain period of time even after the supply voltage decreased back to its intended voltage range. The capacitor remains charged because its discharge towards the supply side is prevented by the configuration of the supply source (e.g., by the presence of a diode), and in a conventional system the discharge of the capacitor at the circuit side is slow because in case of an overvoltage the electronic circuit is turned off and draws almost no current.

[0011] Due to the slow capacitor discharge, the electronic circuit may thus still see a high voltage at the input terminal despite the fact that the supply voltage returned to a safe range. The circuit may thus refrain from resuming its operation (illustratively, from turning back on) due to the high voltage provided by the charged capacitor. The high voltage at the input terminal may thus cause the electronic circuit to remain turned off for an unnecessarily long period of time, illustratively the period of time it takes for the damping capacitor to discharge.

[0012] This undesired behavior of the electronic circuit may be particularly evident in case the surge of the supply voltage takes the form of a sequence of voltage pulses. Each voltage pulse may have a voltage value (e.g., at the peak) causing an overvoltage condition of the electronic circuit and thus causing the electronic circuit to turn off. Positive voltage peaks in bursts (surge pulses) load the damping capacitor (also referred to herein as filtering capacitor or smoothing capacitor) at a voltage level higher than the operational one, especially if there is a reverse diode in series, thus stressing or causing the chip protections to intervene and put in shutdown the analog sections and the external load. As mentioned, in certain circuit configurations the discharge of the damping capacitor may be slow such that the damping capacitor remains charged at a high voltage during the interval between consecutive voltage pulses. Thus, in this scenario the electronic circuit remains off for the whole duration of the sequence of voltage pulses, even though it would be possible to safely operate the circuit in the interval between voltage pulses, during which the supply voltage has a voltage value in a safe operating range.

[0013] Considering the scenario in which a diode is used, the diode in series is forward biased in nominal operation, and removes negative voltages at the supply pin, which would constitutea harmful condition for the device reliability. However, a critical drawback is immediately associated: as a large filter is present at the device terminals to improve Power Supply Rejection Ratio (PSRR), the combination of the capacitor and the diode acts as a peak detector and only the load current can discharge the capacitor to recover the normal operation. Trivial calculations report a critical condition: assuming a nominal supply voltage below 20 V, peaks can be as large as 100 V. In case the current through the load is left unaltered, the power dissipation is more than 5 times the value observed during normal operation. This stress lasts for the time it takes to discharge the filter capacitor. The driver current may be turned off to avoid the device stress. Unfortunately, the residual device current is usually so poor that makes excessive the time to discharge the capacitor. The possible remedy of reducing the capacitor size degrades PSRR performances and emphasizes possible resonant conditions.

[0014] Existing approaches to address the slow discharge of the capacitor still present some drawbacks. In particular, existing solutions miss the possibility to discharge the capacitor with a larger current when the supply voltage is not the worst case one.

[0015] As an example, a voltage analog-to-digital converter (ADC) and a current digital-to-analog converter (DAC) corresponding to different thresholds at the supply voltage may be used to select different values for the discharge current. This solution is straightforward but increases the device complexity. It is a matter to sense the battery voltage, compare it with different thresholds and, any time the associated comparator trips, a change in the load current is triggered. Once the threshold voltage is determined, the corresponding value of the load current to be set can be determined accordingly to the maximum value allowed by self-heating and reliability demands for that supply value. Unfortunately, also the drawbacks are relevant: besides a resistor string from supply to ground, it needs for as many comparators as the load current levels to be chosen. To avoid spurious transitions, a deglitcher should be implemented for each comparator too. This implies a relevant area occupation for the many filters to be adopted, one for each current step. In addition, the DAC current implementation is not straightforward, being non-linear the relationship between voltage and current to operate at constant dissipation.

[0016] Aspects of the present disclosure may be related to a strategy to provide a rapid and dynamic discharge of a damping capacitor, e.g., in case of a voltage surge, thus ensuring that an electronic circuit may resume its intended operation in a fast and efficient manner. In particular, the present disclosure may be based on the realization that it is beneficial to dynamically adjust the flow of discharge current according to the behavior of the supplyvoltage, thus enhancing the discharge efficiency compared to a “static” approach that makes use of a fixed discharge current.

[0017] In the configuration proposed herein, the discharge current may inversely follow the supply voltage. For example, the discharge current may be at a low current value when the supply voltage is at its peak during overvoltage, and then the discharge current may increase while the supply voltage starts decreasing from the peak value. The dynamic adaptation of the discharge current provides a faster discharge (in view of the higher discharge current), while ensuring that power dissipation (illustratively, the product of the discharge current and the voltage) remains substantially constant during the discharge. The proposed approach allows to discharge the damping capacitor quickly and without damaging the circuit due to prolonged overvoltage and higher power dissipation and avoiding over-temperature conditions.

[0018] According to various aspects, a circuit includes: a supply terminal configured to receive a supply voltage; a driver circuit comprising an input terminal coupled with the supply terminal to receive the supply voltage as input voltage for powering an operation of the driver circuit; a damping capacitor coupled in parallel with the input terminal and ground such that the damping capacitor is charged by the supply voltage; wherein the driver circuit further includes a discharge current generator circuit configured to cause a flow of a discharge current through a discharge path if an input voltage value of the input voltage at the input terminal fulfills an overvoltage criterion, thereby causing a discharge of the damping capacitor, wherein the discharge current generator circuit is configured such that the discharge current has a behavior over time that is inversely related to a behavior of the input voltage at the input terminal.

[0019] In a preferred configuration, the discharge current generator circuit may include a translinear loop. As generally known, the term “translinear loop” may describe a type of analog circuit designed to implement a nonlinear signal processing according to the translinear principle. A “translinear loop” may include an even number of translinear elements disposed within the circuit in such a way that the product of the currents through a first subset of translinear elements (e.g., a first half of the translinear element) is equal to the product of the currents through a second subset of translinear elements (e.g., a second half of the translinear elements). The translinear elements may include diode elements or transistor elements, such as bipolar junction transistors, BJT, or metal-oxide semiconductor field-effect transistors, MOSFETs.

[0020] Aspects of the present disclosure are based on the realization that a translinear loop provides a convenient analog realization of the discharge current generator circuit, to provide a discharge current that follows the supply voltage with an optimum behavior in terms of speedof discharge and reduced power consumption, as will be described in further detail below. It is however understood that a “translinear loop” is not the only option to implement the strategy proposed herein.

[0021] In general, the circuit may be configured such that, in operation, a discharge of the damping capacitor towards the supply terminal is prevented. For example, the supply terminal may be coupled with a supply source configured to supply a greater amount of current compared to an amount of current that the supply source is capable of absorbing. As another example, the circuit may further include a filter element coupled between the supply terminal and the input terminal and configured to block a negative portion of the supply voltage (and accordingly a flow of current in the reverse direction from the capacitor towards the supply terminal and supply source).

[0022] The proposed configuration may be implemented without introducing additional components at the input, such as high voltage transistors. In principle a high voltage NMOS transistor, where NMOS is an acronym for N-type metal-oxide-semiconductor, may be added in parallel to the damping capacitor to allow a discharge of the damping capacitor, e.g. between consecutive peaks in a sequence of (over)voltage pulses. Illustratively, TVS and VDR clamps used to reduce the maximum value of the voltage peak, or a diode used to clamp negative signal residues, are not capable of discharging the filtering capacity in a rapid manner, e.g. during time between one peak and another, so that an active discrete component such as a high voltage NMOS may be introduced.

[0023] However, such type of active discrete component is rather expensive and occupies a large silicon area on the chip, thus making it an unattractive solution, especially considering the constant trend towards cost reduction and miniaturization of electronic circuits. Furthermore, the use of clamp structures and the active clamp (like NMOS) inside the chip increases the power dissipation. In contrast thereto, the proposed scheme for a rapid discharge of the damping capacitor may be implemented at a lower cost and with a lower footprint compared to alternative solution that rely on expensive active discrete components.

[0024] The proposed approach may be of particular relevance for electronic circuits for use in a vehicle, which is an application scenario in which sequences of voltage pulses at a high voltage may occur more often than in other types of applications. The proposed strategy for a rapid discharge of a damping capacitor may thus be particularly suitable to ensure a robust and reliable operation of the electronic circuits of a vehicle. Thus, in the following particular reference may be made to the integration of a circuit configured as proposed herein in a vehicle, e.g. for coupling with a battery unit of the vehicle as source of supply voltage. It is howeverunderstood that the applications of a circuit configured as described herein are not limited to the automotive context, and the proposed circuit may be for use also in any suitable type of environments or device in which surge protection may be of relevance.

[0025] Furthermore, the proposed approach may be of particular relevance for implementation in an electronic for the driving of light emitting elements, e.g. light emitting diodes. The proposed strategy for a rapid discharge of a damping capacitor may ensure that the light emitting elements (e.g., in the headlight of a vehicle) are turned off for a short(er) period of time in case of an overvoltage, thus providing a more stable and reliable operation, which may provide an enhanced safety in the automotive context. Thus, in the following particular reference may be made to a configuration in which the load driven by the driver circuit includes one or more light emitting elements. It is however understood that the load (e.g., the external series element) of the proposed circuit is not limited to light emitting elements, and the load of the proposed circuit may include any suitable type of component.

[0026] FIG.1A and FIG.1B show an electronic circuit 100 configured to implement an adapted scheme for causing a discharge of a damping capacitor, in a schematic representation according to various aspects. As an exemplary use case scenario, the electronic circuit 100 may be for use in a vehicle. For example, a vehicle (e.g., a car, such as an electric car) may include one or more electronic circuits configured as the electronic circuit 100. In the following, the electronic circuit 100 may be referred to simply as circuit 100. It is understood that the representation of the circuit 100 in FIG.1 A and FIG. IB may be simplified for the purpose of illustration, and the circuit 100 may include additional components with respect to those shown.

[0027] The circuit 100 may include a supply terminal 102 configured to receive a supply voltage 104. The supply terminal 102 may be configured to be coupled with a supply source, e.g., a constant voltage source, that generates and delivers the supply voltage 104. For example, considering the integration of the circuit 100 in a vehicle, the supply terminal 102 may be coupled with a battery unit of the vehicle, and may receive the supply voltage 104 from the battery unit. A supply voltage from the battery unit may also be referred to herein as battery voltage, VBAT. In some aspects, the circuit 100 may further include a supply source coupled with the supply terminal 102, e.g. a battery unit coupled with the supply terminal 102.

[0028] The circuit 100 may further include a driver circuit 106 coupled with the supply terminal 102 to receive the supply voltage 104. The driver circuit 106 may in general be an electronic circuit configured to implement a certain functionality, for example the driver circuit 106 may be a circuit configured to drive one or more electronic elements coupled with the driver circuit 106 (e.g., a load coupled with a load terminal of the driver circuit 106), as discussed in furtherdetail below. It is however understood that the driver circuit 106 is not limited to a circuit that drives a component but may be in general any suitable circuit in which the scheme proposed herein may be implemented.

[0029] In some aspects, the driver circuit 106 may be an integrated circuit, illustratively the various components of the driver circuit 106 may be integrated on the same substrate, e.g. on a printed circuit board (PCB) substrate. The driver circuit 106 may also be referred to as integrated circuit device. The representation of the driver circuit 106 in FIG.1A may be simplified for the purpose of illustration, and the driver circuit 106 may include additional components with respect to those shown (see also FIG.6). As further exemplary components, the driver circuit 106 may include a digital processing circuit, a power management circuit, a control circuit, a temperature detection circuit, and the like.

[0030] The driver circuit 106 may include an input terminal 108 coupled with the supply terminal 102 to receive the supply voltage 104 as input voltage for powering an operation of the driver circuit 106. In various aspects, the input terminal 108 may be indirectly coupled with the supply terminal 102, e.g. over a protection circuit 110 that will be described in further detail below. In this configuration, the input voltage at the input terminal 108 may be the supply voltage 104 as output from the protection circuit 110. To avoid any misunderstanding possibly caused by the simplified and schematic representation in FIG.1A it is noted that the input terminal 108 is not directly connected to the reference (ground) terminal 117. Depending on the type of components of the protection circuit 110, one or more properties of the input voltage at the input terminal 108 (e.g., an amplitude, a sign, etc.) may be different from corresponding properties of the supply voltage 104 at the supply terminal 102. Illustratively, the input voltage at the input terminal 108 may in general have a behavior corresponding to the behavior of the supply voltage 104, but the input voltage may differ from the supply voltage 104 in view of the functionality of the components of the protection circuit 110. Aspects discussed in relation to a “supply voltage” may apply in a corresponding manner to an “input voltage” at the input terminal of the driver circuit, and vice versa.

[0031] In general, the protection circuit 110 may include one or more components to protect the driver circuit 106 from an undesired (and unexpected) behavior of the supply voltage 104. In an exemplary, simple configuration the protection circuit 110 may include a filter element 112 coupled between the supply terminal 102 and the input terminal 108, and a damping capacitor 114 coupled in parallel with the input terminal 108 and ground. It is however understood that, in principle, the protection circuit 110 may include additional components to further enhance the protection of the driver circuit 104. Further exemplary protectioncomponents, e.g. for surge protection, will be described in further detail below. It is also understood that, in principle, the protection circuit 110 may include only the damping capacitor 114 (without the filter element 112).

[0032] The filter element 112 may be configured to block a negative portion of the supply voltage 104. Illustratively, the filter element 112 may be configured to receive the supply voltage 104 from the supply terminal 102, e.g. a supply voltage 104 having positive and negative values, and to deliver an output voltage including only the positive portion of the supply voltage 104. The voltage at the output of the filter element 112 may thus include only positive values, e.g. only values equal to or greater than 0 V considering 0 V as reference voltage for the circuit 100. The filter element 112 may thus be connected in series with the supply terminal 102 and may be configured to provide reverse voltage protection by preventing negative (reversed) voltage to reach the input terminal 108 of the driver circuit 106. Illustratively, the filter element 112 may have a rectifying function on the supply voltage 104.

[0033] The filter element 112 may be realized in any suitable manner, e.g. as an individual component or as a combination of multiple components. As an exemplary configuration, the filter element 112 may be or include a diode, e.g. a Schottky diode. The diode may allow power (e.g., voltage) to reach the driver circuit 106 only when the diode is forward biased (by a positive supply voltage 104 or a positive portion of the supply voltage 104), and block the voltage in case of reverse bias. A diode may provide a simple and cost-effective reverse voltage protection. It is however understood that, in principle, the filter element 112 may be realized in a different manner, e.g. the filter element 112 may include a metal-oxide-semiconductor fieldeffect transistor (MOSFET), or an oRing controller in combination with a power MOSFET, as other examples.

[0034] The damping capacitor 114 may in general be configured to protect the driver circuit 106 from spikes in the supply voltage 104, e.g. spikes in the positive portion of the supply voltage 104 as output by the filter element 112. Illustratively, the damping capacitor 114 may be coupled in parallel with the input terminal 108 and a ground terminal 116, such that the supply voltage 104 at the supply terminal 102 may charge the damping capacitor 114, e.g. the positive portion of the supply voltage 104 as output by the filter element 112 may charge the damping capacitor 114. Illustratively, the voltage difference between the terminal at which the damping capacitor 114 sees the supply voltage 104 and the ground terminal 116 causes a current flow that charges the damping capacitor 114.

[0035] As generally known in the art, a damping capacitor (e.g., the damping capacitor 114) may absorb excess energy, thus preventing the excess energy to reach and damage othercomponents (e.g., the driver circuit 106). The damping capacitor may reduce the amplitude of a high-frequency signal, e.g. a high-frequency supply voltage.

[0036] The ground terminal 116 (and the ground terminal 117) may be a reference terminal at which a reference voltage for the circuit 100 is provided. The term “reference voltage” may be used herein to denote a base voltage for the circuit 100 (e.g., a base voltage for the driver circuit 106). In some aspects, the reference voltage may be also referred to as ground (GND) voltage, ground potential, virtual ground voltage, or zero volts (0 V). In an exemplary configuration the reference voltage may be 0 V, but the aspects described herein may apply in principle to any suitable value for the reference voltage.

[0037] The damping capacitor 114 may have any suitable configuration to implement the desired function, e.g. as a single component or as a combination of a plurality of components. For example, the damping capacitor 114 may be configured as a film capacitor, e.g. including a polymer as dielectric material, such as polyester, polypropylene or polycarbonate. As another example, the damping capacitor 114 may be configured as a ceramic capacitor including a ceramic material as dielectric. As a further example, the damping capacitor 114 may be configured as a tantalum capacitor including tantalum as dielectric material.

[0038] The properties of the damping capacitor 114, e.g. in terms of size, capacitance, and the like, may be adapted depending on fabrication constraints and depending on the intended operation of the circuit 100. For example, the capacitance of the damping capacitor 114 may be selected based on the frequency range of the circuit 100. Only as a numerical example, the damping capacitor 114 may have a capacitance in the range from 50 nF (nanofarad) to 500 pF (microfarad), for example in the range from 100 nF to 300 pF, for example a capacitance of about 220 nF.

[0039] In an exemplary configuration, the damping capacitor 114 may include a plurality of capacitors connected in series with one another. Illustratively, the damping capacitor 114 may be realized as a plurality of capacitors, so that in case one of the capacitors behaves as a short circuit, the other capacitors may still provide protection to the driver circuit 106. Such configuration may be of particular relevance as a safety feature in the automotive context, to ensure a safe and reliable operation of the circuit 100. The damping capacitor 114 may include any suitable number of series-connected capacitors, e.g., two in a simple and cost-effective configuration, or more than two, e.g., three, four, five, etc.

[0040] Considering a diode as filter element 112, the diode and the damping capacitor 114 may act as a peak detector. In case of an oscillating waveform of the supply voltage 104, the diode allows the positive portion (the positive cycles) of the supply voltage 104 to pass through andcharge the damping capacitor 114 to the peak voltage of the waveform. When the waveform falls below the peak value stored in the damping capacitor 114, the diode is reverse biased and blocks the current flow from the damping capacitor 114 back to the supply terminal 102 (and supply source). The damping capacitor 114 may thus be isolated and store (illustratively, memorize) the peak value of the waveform of the supply voltage 104, even when the waveform drops (e.g., back to zero).

[0041] As mentioned above, aspects of the present disclosure may address the fact that in case of a voltage surge of the supply voltage 104, the damping capacitor 114 may remain charged at the peak value of the waveform of the supply voltage 104, thus leading to unnecessarily prolonged “off-time” of the driver circuit 106.

[0042] As discussed, the damping capacitor 114 may remain charged in a scenario in which there is limited possibility for a current to flow away from the capacitor. In general, the circuit 100 may be configured such that, in operation, the discharge of the damping capacitor 114 towards the supply terminal 102 is limited or prevented. In the exemplary configuration in FIG.1 A, the filter element 112 (e.g., the diode) may limit the discharge of the damping capacitor 114 by preventing the current flow in the reverse direction towards the supply terminal 102. In another exemplary configuration, the circuit 100 may include or may be coupled with a supply source (at the supply terminal 102) that limits or prevents current flow towards the supply source. The supply source may be an asymmetric supply source configured to supply current at a first current value and to absorb current at a second current value, and the second current value may be (much) lower than the first current value, for example at least 5 times lower, at least 10 times lower, or at least 20 times lower. In this configuration, the damping capacitor 114 may have a discharge path only towards the circuit side, i.e. towards the driver circuit 106.

[0043] In case of a voltage surge, the supply voltage 104 may have a voltage value greater than an operational voltage of the driver circuit 106, illustratively the maximum voltage rated for the operation of the driver circuit 106. For example, in case of a sequence of pulses, the voltage value of the pulses may be greater than the operational voltage of the driver circuit 106. At least the peak voltage of the pulses may be greater than the operational voltage of the driver circuit 106, or the voltage of the pulses may be greater than the operational voltage also prior to reaching the peak. The operational voltage of the driver circuit 106 may be a voltage or voltage range in which the driver circuit 106 may operate without incurring in the risk of damaging or breaking the components of the driver circuit 106. In case of a voltage surge, the supply voltage 104 may thus have or assume voltage values outside of the safe operating range of the driver circuit 106.

[0044] In general, the driver circuit 106 may be configured to implement a protection mechanism to turn off (in other words, shut down) the driver circuit 106 in case of a surge of the supply voltage 104. As long as the driver circuit 106 receives an input voltage at the input terminal 108 greater than the rated operational voltage, the driver circuit 106 may refrain from turning on again. However, in a conventional configuration the damping capacitor 114 may have a relatively slow discharge, e.g., due to the fact that the driver circuit 106 is off and there is no path for the current to flow away rapidly from the damping capacitor 114. Thus, in a conventional configuration the damping capacitor 114 retains a voltage value greater than the operational voltage of the driver circuit 106 for a relatively long time, and such voltage is seen at the input terminal 108 thus preventing the driver circuit 106 from turning on again.

[0045] As mentioned, the driver circuit 106 may generally be configured to drive a load. Illustratively, the driver circuit 106 may be configured to provide a driving current to the load for driving the operation of the load. In this regard, the driver circuit 106 may include a load terminal 120 configured to be coupled with a load, and a reference current source 118 configured to generate a reference current for providing the driving current at the load terminal 120. Illustratively, the reference current source 118 may be configured to provide the reference current to be used for obtaining the driving current delivered to the load via the load terminal 120. As an exemplary implementation of the reference current source 118, the driver circuit 106 may include a bandgap voltage generator configured to generate a bandgap voltage, and a resistive element (e.g., a resistor), such that the reference current may be obtained by a drop of the bandgap voltage over the resistive element. As will be discussed in further detail below, the present disclosure is based on the realization that in addition to the reference current source 118 a further discharge current generator circuit 130 may be provided to operate in coordination with the reference current source 118 for providing a rapid discharge of the capacitor 114.

[0046] The load may be external to the driver circuit 106. In some aspects, as shown in FIG. IB, the circuit 100 may include a load 122 coupled to the load terminal 120. In general, the load 122 may be any suitable electronic component that the driver circuit 106 may drive (and control, as discussed in further detail below). In a preferred configuration the load 122 may include one or more light emitting elements, e.g., a plurality of light emitting elements, e.g., connected in series with one another. Aspects of the present disclosure may be based on the realization that the proposed approach may be of particular interest in the context of light emitting devices, e.g., for use in a vehicle, for example in the headlight of a vehicle or for the internal illumination of the vehicle. In this scenario, the load 122 may include any suitable number of light emitting elements, e.g., one, two, three, four, five, ten, or more than ten.

[0047] In principle, the light emitting elements may be of any suitable type. Considering the context of integrated circuits, the light emitting elements may be or include light emitting diodes (LEDs), e.g., the plurality of light emitting elements may include at least one light emitting diode. LEDs may be particularly suitable to allow a current flow even in case of an overvoltage condition. Illustratively, a LED may have a certain forward voltage, e.g., in the range from 1.8 V to 3.8 V depending on the color of the LED, but a LED may operate without suffering catastrophic damages even at higher voltages. As another example, the light emitting elements may be or include laser diodes, e.g., edge emitting laser diodes or vertical cavity surface emitting laser diodes.

[0048] The light emitting elements (e.g., the LEDs) may be configured to emit light having a predefined wavelength, for example in the visible range (e.g., from about 380 nm to about 700 nm), infrared and / or near-infrared range (e.g., in the range from about 700 nm to about 5000 nm), or ultraviolet range (e.g., from about 100 nm to about 400 nm). In some aspects, the light emitting elements may be configured to emit light in different wavelength ranges. For example, a first light emitting element may be configured to emit light in a first wavelength range (e.g., a first color, for example blue), a second light emitting element may be configured to emit light in a second wavelength range (e.g., a second color, for example red), and a third light emitting element may be configured to emit light in a third wavelength range (e.g., a third color, for example green), etc.

[0049] In some aspects, the load 122 may be or include a series element 124 (external to the driver circuit 106, referred to as external series element), e.g., one or more series elements. The series elements 124 may be coupled in series between the input terminal 108 and ground 117. The load 122 may thus include the series element(s) coupled in series with the reference current source 118. The reference current source 118 may determine the current drawn through the load 122, and the external series element(s) 124 may define the path through which the current flows to ground 117.

[0050] In some aspects, as shown in FIG.1A and FIG. IB, the driver circuit 106 may further include an internal series element 126. The internal series element 126 may be or include any suitable internal component of the driver circuit 106 capable of conducting current even in case of overvoltage. The internal series element 126 may be coupled with the input terminal 108 and with the current source 118 (and to ground 117). In some aspects, the internal series element 126 may be a dedicated component in the driver circuit 106, illustratively a component that does not carry out any other functionality other than allowing a current flow through it in case of an overvoltage. In other aspects, the internal series element 126 may be a component of thedriver circuit 106 that carries out a certain functionality of the driver circuit 106 and is additionally used to draw current through it in case of an overvoltage. This second configuration may provide a cost- and space-efficient approach by reusing components already present in the circuit 100. As an example, the internal series element 126 may be or include one or more transistors.

[0051] In some aspects, the driver circuit 106 may be configured to deliver the reference current as driving current. In a preferred configuration, the driver circuit 106 may be configured to deliver an amplified version of the reference current as driving current to the load terminal 120 (and to the load 122). This configuration may exploit the generation of a precise reference current (e.g., using a bandgap voltage generator) with the possibility of tailoring the amplification depending on the requirements of the load 122.

[0052] Various implementations may be provided in this regard. As a preferred realization, as shown in the inset 140 in FIG.1A, the driver circuit 106 may include a current mirror 142 configured to receive the reference current 144 from the reference current source 118 and to deliver a magnified version of the reference current 144 as driving current 146 at the load terminal 120. The current mirror 142 may be adapted to provide any suitable amplification factor, K, for the reference current 144.

[0053] As shown in FIG.1A, the current mirror 142 may include an input branch coupled with the reference current source 118, and an output branch coupled with the load terminal 120. The input branch of the current mirror 142 may thus be configured to receive the reference current 144, and the current mirror 142 may be configured to provide a replica of the reference current 144 at the output branch, amplified by a factor K, as output current 146 (and accordingly as driving current for driving the load 122). The output current 146 of the current mirror 142 may thus be a multiple of the reference current 144.

[0054] As an exemplary configuration, the current mirror 142 may include a first transistor element 143 at the input branch (e.g., a first NMOS transistor) and a second transistor element 145 at the output branch (e.g., a second NMOS transistor). The transistor elements 143, 145 may be dimensioned to obtain the desired amplification factor, K, for the current. Considering for example NMOS transistors, the drain of the first NMOS may be coupled with an input terminal 147 of the current mirror 142 at which the current mirror 142 receives the reference current 144. The source of the first NMOS may be coupled to the source of the second NMOS (and to the reference terminal 117). The gate of the first NMOS may be coupled to the gate of the second NMOS, and the drain of the second NMOS may be coupled to the load terminal 120.The first NMOS may be in a diode-connected configuration, and may have the gate coupled to the drain.

[0055] In general, the use of a current mirror 142 and the use of NMOS transistors to realize the current mirror 142 may represent a preferred configuration of the driver circuit 106. The current mirror 142 allows a flexible, yet efficient amplification of the reference current 144, whereas NMOS transistors may generally be realized having a smaller footprint compared to other transistor types (e.g., PMOS). Thus, in the following discussion regarding possible configurations of the discharge current generator circuit 130 (e.g., in FIG.2A to FIG.5C), without loss of generality the discharge current generator circuit is generally shown in combination with a current mirror 142, and the current mirror 142 is generally shown to include NMOS transistors.

[0056] It is however understood that the aspects discussed in relation to the possible configurations of the discharge current generator circuit 130 apply in a corresponding manner to a configuration in which the driver circuit 106 does not include the current mirror 142 and / or to a configuration in which the current mirror 142 is realized with different components (e.g., additional transistors, additional branches, different types of transistors such as PMOS transistors or BJT transistors, etc.).

[0057] Furthermore, considering the preferred configuration in which the discharge path 132 is provided by the load 122, in the following discussion and without loss of generality the discharge current generator circuit 130 is generally shown coupled with the load terminal 120, such that the discharge path 132 goes through the load terminal 120 (and accordingly through the load). It is however understood that the aspects discussed in relation to the possible configurations of the discharge current generator circuit 130 apply in a corresponding manner to a configuration in which a different discharge path 132 is provided (e.g., through the internal series elements 126, or through a different path).

[0058] As discussed above, aspects of the present disclosure may be based on the realization that a discharge of the damping capacitor 114 may be provided via a discharge current that has an inverse behavior with respect to the supply voltage (and accordingly, with respect to the input voltage at the input terminal 108), thus combining an increased speed for the discharge with a reduced power consumption. The quick discharge of the damping capacitor 114 ensures that the input voltage at the input terminal 108 returns to a voltage value in the operational range of the driver circuit 106 in a rapid manner, thus allowing the driver circuit 106 to resume its operation.

[0059] In this regard, the driver circuit 106 may include a discharge current generator circuit 130 configured to generate the discharge current for discharging the damping capacitor 114 in case of overvoltage. The discharge current generator circuit 130 may be understood to provide an overvoltage protection circuit, that triggers a discharge of the damping capacitor 114 if overvoltage occurs, thus allowing the driver circuit 106 to resume its normal operation in a time-efficient manner.

[0060] The discharge current generator circuit 130 may thus be configured to cause a flow of discharge current through a discharge path 132 if an input voltage value of the input voltage at the input terminal 108 fulfills an overvoltage criterion. For example, an overvoltage condition may include the input voltage value of the input voltage at the input terminal 108 being greater than the rated operational voltage of the driver circuit 106, e.g., the voltage value of the input voltage at the input terminal 108 being outside of the nominal operational voltage range of the driver circuit 106. The input voltage value may fulfill the overvoltage criterion if the input voltage value is greater than a predefined threshold voltage (illustratively, an overvoltage threshold).

[0061] The flow of discharge current caused by the discharge current generator circuit 130 may cause a discharge of the damping capacitor 114. Illustratively, current may flow away from the damping capacitor 114 through the discharge path 132, thereby discharging the damping capacitor 114. In some aspects, the discharge current generator circuit 130 may act as a sink to allow a discharge of the capacitor 114 through the discharge path 132. The discharge path 132 may thus be configured to enable sinking (illustratively, dissipating) the discharge current from the damping capacitor 114.

[0062] In this regard, there are various possible configurations for the discharge current generator circuit 130, which will be described in further detail in relation to FIG.2A to FIG.5C. In general, according to the proposed approach, the discharge current generator circuit 130 is configured such that the discharge current has a behavior over time that is inversely related to a behavior of the input voltage at the input terminal 108 (and accordingly inversely related to a behavior of the supply voltage 104 at the supply terminal 102). Illustratively, the discharge current generator circuit 130 may be configured such that a current value of the discharge current varies over time in an opposite manner with respect to the input voltage value of the input voltage at the input terminal 108.

[0063] For example, the discharge current generator circuit 130 may be configured such that for increasing voltage value of the input voltage, the current value of the discharge current may decrease, and such that for decreasing voltage value of the input voltage, the current value ofthe discharge current may increase. Illustratively, for a first voltage value of the input voltage the discharge current may have a first current value. For a second voltage value of the input voltage less than the first voltage value, the discharge current may have a second current value greater than the first current value. For a third voltage value of the input voltage less than the second voltage value, the discharge current may have a third current value greater than the second current value, etc.

[0064] For the discharge of the damping capacitor 114 any suitable discharge path may be used. Illustratively, the discharge path 132 may be any suitable electrically conductive path through which current may flow away from the damping capacitor 114 (e.g., towards ground 117). For example, the discharge path 132 may be disposed internally to the driver circuit 106 or externally to the driver circuit 106. In some aspects, the discharge path 132 may include one or more electrical components disposed along the discharge path 132, and the one or more electrical components may enable dissipation of the discharge current.

[0065] In a preferred configuration, the discharge current generator circuit 130 may be configured such that the load 122 provides the discharge path 132 for the discharge of the damping capacitor 114. Illustratively, the discharge current generator circuit 130 may be configured such that the discharge current flows through the load terminal 120 (and through the load 122, if coupled to the load terminal 120). In this configuration, the discharge current generator circuit 130 (e.g., an output terminal of the discharge current generator circuit 130) may be coupled with the load terminal 120, thereby providing the flow of discharge current at the load terminal 120. This approach may allow “re-using” elements generally present in the circuit 100 during operation (i.e., the load 122) to provide a discharge of the capacitor 114, thus enabling a resource-efficient configuration.

[0066] It is however understood that in principle the discharge current generator circuit 130 may be configured such that other elements provide the discharge path 132 for the discharge of the damping capacitor 114. As another example, the discharge current generator circuit 130 may be configured such that the internal series element(s) 126 provide the discharge path 132. As a further example, the discharge path 132 may be a dedicated electrically conductive path of the driver circuit 106, provided to allow the flow (and the sinking) of discharge current therethrough.

[0067] As mentioned above, the discharge current generator circuit 130 is configured to obtain an inverse relationship between the current value of the discharge current and the voltage value of the input voltage at the input terminal 108. In this regard, the relationship may have anysuitable profile, illustratively any suitable relation between the variation of the input voltage and the corresponding variation of the discharge current.

[0068] In a preferred configuration, the discharge current generator circuit 130 may be configured such that the discharge current is inversely related (e.g., inversely proportional) to the input voltage value according to a hyperbolic profile. Stated in a different fashion, the discharge current generator circuit 130 may be configured such that the discharge current varies over time in an inverse manner compared to the input voltage value following a hyperbolic relationship with the variation of the input voltage value. A hyperbolic relationship has been found to be the most suitable to combine the requirements of speed and power consumption.

[0069] This finding can be explained with reference to the graph 150 in FIG.1C, that shows a variation of the load current, ILOAD, with respect to the supply voltage, VDD. In this context, the voltage drop across the load 122 may be denoted as VLOAD. For example, considering the case in which the load 122 includes a plurality of LEDs (e.g., a string of LEDs), the load current, ILOAD, may be a LED current, ILED, and the voltage drop across the load 122 may be a drop across the LEDs, VLED.

[0070] In general, considering a load 122 coupled to the load terminal 120, the overall power dissipation PDISS) in the driver circuit 106 may be expressed as PDISS = (VDD-VLOAD)* ILOAD (e.g., as (VDD-VLED)*ILED), where VDD is the supply voltage (and accordingly, indicates the input voltage at the input terminal 108). At the same time, the device may be designed to have the maximum allowed dissipation when the supply (e.g., the battery) is at the overvoltage threshold, VDD th, and the driver forces the nominal current, ILOAD nom (e.g., ILED nom). In this scenario, there is a maximum of dissipated power, PDISS max = (VDD II-VLOAD)* ILOAD nom.

[0071] It follows, that the discharge of the damping capacitor 114 is optimally set once the above expressions are made equal to give ILOAD = ILOAD nom* (VDD th- VLOAD) / (VDD- VLOAD). This makes a hyperbolic relationship (drawn in FIG.1C as dotted line 152) for ILOAD VS. the value of the supply voltage (e.g., the battery voltage). The hyperbolic discharge 152 represents thus the optimal current for the capacitor discharge.

[0072] It is however understood that in principle the discharge current generator circuit 130 may be configured such that the discharge current has a different type of profile with respect to the input voltage (and supply voltage). As another example, the discharge current generator circuit 130 may be configured such that the discharge current is inversely related to the input voltage value according to a linear profile (e.g., as shown by the curve 154 in the graph 150). As a further example, the discharge current generator circuit 130 may be configured such that the discharge current is inversely related to the input voltage value according to a parabolicprofile. Stated in a different fashion, the discharge current generator circuit 130 may be configured such that the discharge current varies over time in an inverse manner compared to the input voltage value following a linear relationship or a parabolic relationship with the variation of the input voltage value. These other options may be less efficient than the hyperbolic profile in terms of power consumption, but may still enable a rapid discharge of the capacitor 114, and may be provided in some scenarios in which a slightly increased power consumption may be acceptable.

[0073] As mentioned above, the discharge current generator circuit 130 may cause the flow of discharge current if (e.g., as soon as) the input voltage value of the input voltage at the input terminal fulfills the overvoltage criterion. In this regard, the overvoltage criterion may include any suitable criterion to enable determining that an overvoltage has occurred or is occurring. In a preferred configuration, the overvoltage criterion may include the voltage value of the input voltage being greater than an overvoltage threshold (VDD th), such that the criterion is fulfilled if the input voltage is greater than the overvoltage threshold and is not fulfilled if the input voltage is less than the overvoltage threshold. The overvoltage threshold may thus be a voltage value not to be exceeded during an operation of the driver circuit. This approach allows designing the discharge current generator circuit 130 in such a way that the circuit 130 reacts “automatically” to the increased input voltage, as will be discussed in further detail in relation to FIG.2A to FIG.5C. Illustratively, the discharge current generator circuit 130 may be configured in such a way that the input voltage exceeding the overvoltage threshold activates the current generation by the discharge current generator circuit 130, without the need for additional components.

[0074] The overvoltage threshold and the threshold voltage value of the overvoltage threshold may be selected according to circuit considerations, e.g., based on the specific properties of the circuit 100 and driver circuit 106. In general, the threshold voltage value may have any suitable value that may represent an overvoltage condition of the circuit if the input voltage becomes greater than such threshold. In an exemplary configuration, the threshold voltage value may be equal to or greater than the (maximum) operational voltage of the driver circuit 106. Only as a numerical example, the threshold voltage value may be in the range from 10 V to 30 V, for example in the range from 15 V to 25 V, for example the threshold voltage value may be 20 V or 25 V.

[0075] According to various aspects, the discharge current generator circuit 130 may be configured such that the discharge current is inversely related (e.g., inversely proportional) to a difference between the input voltage value and the threshold voltage value of the overvoltagethreshold. Illustratively, the input voltage reaching the overvoltage threshold may trigger the operation of the discharge current generator circuit 130, and the discharge current itself may then vary depending on how much greater the input voltage is than the overvoltage threshold.

[0076] It is however understood that in principle also other overvoltage criteria may be used, e.g., in addition or in the alternative. As another example, the overvoltage criterion may include an input current value at the input terminal 108 being greater than a threshold current value. As a further example, the overvoltage criterion may include a temperature at or around the input terminal 108 being greater than a threshold temperature value, in which case the driver circuit 106 may trigger the operation of the discharge current generator circuit 130.

[0077] According to various aspects, the discharge current generator circuit 130 may be configured to interrupt the flow of discharge current if the input voltage stops fulfilling the overvoltage criterion, e.g., if the input voltage becomes less than the threshold voltage value. Illustratively, the discharge current generator circuit 130 may be configured such that as soon as the input voltage returns to the normal operating range, the generation of discharge current is stopped, thus allowing the driver circuit 106 to resume its normal operation.

[0078] As mentioned, the representation of the driver circuit 106 is simplified to focus on the aspects relevant for the present disclosure, and that the driver circuit 106 may include additional components with respect to those shown.

[0079] As an example, the driver circuit 106 may further include a temperature detection circuit. The temperature detection circuit may be configured to protect the driver circuit 106 in case the temperature of the driver circuit 106 increases to a level that could deteriorate or damage the components of the circuit. For example, the temperature detection circuit may be part of the internal series element 122, illustratively as circuit component through which current may flow to allow the discharge of the damping capacitor 114.

[0080] In general, the temperature detection circuit may be configured to sense the temperature of the driver circuit 106, e.g., the temperature at which the driver circuit 106 is operating, and may be configured to cause a shutdown of the driver circuit 106 if the sensed temperature fulfills an over-temperature criterion. In an exemplary configuration, the temperature detection circuit may be configured to cause the shutdown of the driver circuit 106 if the sensed temperature is in a predefined temperature range, e.g., if the sensed temperature is greater than a threshold temperature. For example, the temperature detection circuit may transmit a shutdown signal to one or more components of the driver circuit 106 to instruct an interruption of the respective operation.

[0081] The predefined temperature range and / or the threshold temperature may be adapted depending on circuit considerations, e.g., depending on the resistance to temperature of the circuit components. Only as a numerical example, the temperature range may be from 100°C to 200°C, for example from 120° to 180°C. As a further numerical example, the threshold temperature may be 100°C, 155°C, 160°C, 170°C, or any suitable temperature value.

[0082] As an exemplary implementation, the temperature detection circuit may include a temperature sensor configured to sense the temperature, and a processing circuit configured to receive the sensed temperature from the sensor and instruct the shutdown according to the sensed temperature. The temperature sensor may include, for example, a negative-temperature-coefficient thermistor, a positive-temperature-coefficient thermistor, or any suitable temperature sensitive component.

[0083] As a further example, the driver circuit 106 may further include a control circuit configured to control the driving of the load 120. The control circuit may be configured to control the operation of the load 120 including, e.g., a start of the operation, a duration of the operation, a stop of the operation, etc. For example, the control circuit 130 may be part of the internal series element 122, illustratively as circuit component through which current may flow to allow the discharge of the damping capacitor 114.

[0084] In an exemplary configuration, the control circuit may be configured to control the load 120 as a function of the voltage value of the input voltage at the input terminal 108. Illustratively, when the input voltage is in the safe operational range (and does not fulfill the overvoltage criterion), the control circuit may control the operation of the load 120 based on the received supply voltage 104 (as input voltage). As an example, considering the scenario in which the external series element 122 includes a plurality of light emitting elements, e.g., a plurality of LEDs, the control circuit may be configured to control a light emission by the light emitting elements. As an exemplary configuration, to turn on a light emitting element the control circuit may control a corresponding switch element, e.g., a transistor. The switch element associated with a light emitting element may selectively enable or disable a current flow through the light emitting element.

[0085] As mentioned above, the protection circuit 110 may include further protection elements to enhance the safety of the circuit 100. In some aspects, the protection circuit 110 may thus include one or more further protection components, e.g., coupled in parallel with the damping capacitor 114. The one or more further protection components may in general be configured to limit a peak voltage value of the supply voltage 104. Illustratively, the one or more furtherprotection components may be configured to act as a clamp for the supply voltage 104, to prevent transient voltages from passing through.

[0086] As an example, the protection circuit 110 may include a transient voltage suppression (TVS) diode in parallel with the damping capacitor 114, e.g., coupled between the supply terminal 102 and ground 116. As a further example, additionally or alternatively, the protection circuit 110 may include a voltage-dependent resistor in parallel with the damping capacitor 114, e.g., coupled between the supply terminal 102 and ground 116. In view of the proposed discharge method, the protection circuit 110 may in general be free of discrete components to facilitate the discharge of the damping capacitor 114, e.g., the protection circuit 110 may be free of a high voltage transistor (e.g., a high voltage NMOS). However, in some aspects, to further facilitate the capacitor discharge, the protection circuit 110 may include an analog discrete component, such as a high voltage transistor (e.g., a high voltage NMOS) in parallel with the damping capacitor 114, e.g., coupled between the supply terminal 102 and ground 116.

[0087] Turning now to the discharge current generator circuit 130, there may be various possible realizations to obtain the desired inverse relationship between the discharge current and the input voltage. In this regard, FIG.2A and FIG.2B show a first possible approach, in which the discharge current has a linear behavior with respect to the input voltage at the input terminal 108 of the driver circuit 106. FIG.2A and FIG.2B show thus a discharge current generator circuit 200, as a possible realization of the discharge current generator circuit 130. Further possible configurations for the discharge current generator circuit 130 will be discussed in relation to FIG.3 A to FIG.5C.

[0088] In general, to implement the “linear discharge”, the discharge current generator circuit 200 may be configured to receive an input current 202 that is directly proportional to the input voltage at the input terminal 108 of the driver circuit 106. Illustratively, the input current 202 may be the current corresponding to the input voltage, such that a current value of the input current 202 may be directly proportional to the input voltage value of the input voltage. For example, the input current 202 may be directly proportional to the supply voltage 104 at the supply terminal 102.

[0089] Providing an input current 202 that is directly proportional to the input voltage may be realized in various manners. For example, as shown in FIG.2A and FIG.2B, the discharge current generator circuit 200 may include a resistive element 204 (e.g., a resistor), and the input current 202 proportional to the input voltage may be obtained by a voltage drop of the input voltage across the resistive element 204. The use of a resistive element 204 may provide a simple implementation to obtain the input current 202, but it is understood that otherconfigurations may be provided. As another example, the discharge current generator circuit 200 may include an input current source configured to generate the input current 202 proportional to the input voltage.

[0090] The discharge current generator circuit 200 may be configured to start generating the discharge current if the input current 202 is greater than a threshold current 206 (Ith). The threshold current 206 may correspond to the threshold voltage of the overvoltage threshold. Illustratively, the threshold current 206 may be a current value for the input current 202 indicative of an overvoltage condition. Thus, if the input current 202 is greater than the threshold current 206, the overvoltage criterion is fulfilled, and the discharge current generator circuit 200 may cause the flow of discharge current.

[0091] In some aspects, the discharge current generator circuit 200 may further include a threshold current source 208 configured to generate the threshold current 206. As an exemplary realization, the threshold current source 208 may include a bandgap voltage generator and a resistive element (e.g., a resistor) to obtain the threshold current 206 as a voltage drop of the bandgap voltage across the resistive element.

[0092] According to the “linear approach”, the discharge current generator circuit 200 may be configured such that the discharge current is obtained by a subtraction of a difference between the input current 202 and the threshold current 206 from the reference current of the driver circuit 106. Illustratively, obtaining the discharge current may include subtracting, from the reference current 144, IREF, the difference between the input current 202 and the threshold current 206 (IIN - Ith). Considering the direct relationship between the input current 202 and the input voltage, for increasing values of the input voltage the discharge current may decrease as a greater current gets subtracted from the reference current 144.

[0093] As mentioned, in the preferred configuration the discharge path is provided by the load 122 of the driver circuit 106. Thus, in the preferred configuration the discharge current generator circuit 200 may be coupled with the load terminal 120 (e.g., over the current mirror 142), as shown in FIG.2 A and FIG.2B. It is however understood that the “linear approach” may be provided also for a different discharge path 132 (e.g., through the internal series elements, or through a different path).

[0094] As an exemplary realization of the linear approach, as shown in FIG.2A and FIG.2B, the discharge current generator circuit 200 may include a current mirror 210 having an input branch and an output branch.

[0095] The current mirror 210 may receive, at the input branch, the input current 202 proportional to the input voltage, and the threshold current 206. For example, the input branchof the current mirror 210 may include an input terminal 212 coupled with the threshold current source 208, and coupled with the resistive element 204 over which the drop of the input voltage is provided. Illustratively, the resistive element 204 may be coupled between the input terminal 108 of the driver circuit 106 and the input terminal 212 of the current mirror 210.

[0096] In some aspects, to improve the performance, the discharge current generator circuit 200 may include one or more switches to modify the value of the resistor when the input voltage at the input terminal of the driver circuit 106 is high. However, this approach increases the complexity of the arrangement, as each switch should be driven by a dedicated comparator.

[0097] The current mirror 210 may be configured such that no current is transferred from the input branch to the output branch as long as the input current 202 is (in other words, remains) below the threshold current 206. The current mirror 210 may be configured such that if the input current 202 is greater than the threshold current 206, the current mirror 210 transfers to the output branch an output current given by the difference between the input current 202 and the threshold current 206. The current mirror 210 may thus act as an “analog switch” that initiates the discharge of the damping capacitor in case of overvoltage.

[0098] The output branch of the current mirror 210 may thus be coupled with the discharge path 132 for the discharge current. The current mirror 210 may include an output terminal 214 at which the current mirror 210 provides the output current, and the output terminal 214 may be coupled with the discharge path 132. Illustratively, the current mirror 210 may transfer to the output terminal 214 the current given by the difference between the input current 202 and the threshold current 206, if the input current 202 is greater than the threshold current 206.

[0099] As shown in FIG.2A and FIG.2B, the output branch of the current mirror 210 may further be coupled with the reference current source 118, such that the discharge current may be obtained by a subtraction of the output current of the current mirror 210 (i.e., the current given by the difference between the input current 202 and the threshold current 206) from the reference current 144. For example, the output terminal 214 of the current mirror 210 may be further coupled with the reference current source 118, such that at the output the discharge current with the linear behavior is obtained.

[0100] As an exemplary configuration, the current mirror 210 may include a first transistor element 213 at the input branch (e.g., a first NMOS transistor) and a second transistor element 215 at the output branch (e.g., a second NMOS transistor). The “switching behavior” may be provided by the transistor elements 213, 215 that become conductive if the input current 202 is greater than the threshold current 206 and remain non-conductive if the input current 202 is less than the threshold current 206.

[0101] Considering for example NMOS transistors, the drain of the first NMOS may be coupled with the input terminal 212 of the current mirror 210 at which the current mirror 210 receives the input current 202 and the threshold current 206. The source of the first NMOS may be coupled to the source of the second NMOS (and to the reference terminal 117). The gate of the first NMOS may be coupled to the gate of the second NMOS, and the drain of the second NMOS may be coupled to the output terminal 214 (and to the input terminal 147 of the current mirror 142, as an example). The first NMOS may be in a diode-connected configuration, and may have the gate coupled to the drain.

[0102] Thus, using a linear profile for the discharge current, the current is linearly reduced starting from the overvoltage threshold until it reaches the value allowed when the supply is maximum. Starting from the reference current 144 IREF that, conveniently multiplied, generates the driving current 146 for the load, ILOAD, it is a matter to subtract a current that depends on the input voltage (e.g., the supply voltage) once the input voltage exceeds a given threshold. The implementation is simple and purely analog. In this regard, being the input voltage value generally much larger in the range of interest, the drop across the associated diode is negligible and the resistive element 204 may generate the input current 202 proportional to the input voltage (e.g., proportional to the supply voltage).

[0103] Once the constant threshold current 206, Ith, is subtracted, it follows that the load current, ILOAD (e.g., a LED current, ILED) is K times the reference current IREF until the input voltage becomes greater than R*Ith. For increasing values of the input voltage (e.g., at supply, VDD, increasing values), the load current, ILOAD, reduces of an amount K*(VDD-R*Ith) / R. In this expression it is already possible to recognize a potential weakness of the solution because the slope of the current is affected by the large spread (+ / - 25%) of the resistor.

[0104] As the current derivative for the optimal profile of the load current is progressively decreasing, if a resistor is adopted to derate it (as in the curve 154 in FIG.1C), the same slope for the two profiles in FIG.1C should be ensured when the input voltage is at the overvoltage threshold. The associated two main consequences may be easily understood, still considering FIG.1C. The current soon decreases largely below the allowed value, so that the discharge is far away from being optimal. The load current would decrease to zero at an input voltage largely lower than the maximum foreseen one.

[0105] With this in mind, as shown in FIG.2B the discharge current generator circuit 200 may further include a current limiter 220 configured to define a maximum current value for the current to be subtracted from the reference current 144 (to obtain the discharge current). For example, the current limiter 220 may be coupled with the output terminal 214 of the currentmirror 210, such that the current limiter 220 receives the output current of the current mirror 210 and imposes a maximum limit to the output current (then subtracted from the reference current to obtain the discharge current).

[0106] The current limiter 220 thus provides a clamping of the subtraction current to avoid that even at moderately large value of the supply the capacitor 114 is discharged only by the core current. The insertion of the current limiter 220 thus allows to keep a sufficiently large current to discharge the capacitor 114 when the input voltage value (e.g., the supply value) reaches the maximum allowed value. The current limiter 220 may thus be configured to set a predefined current limit for the current obtained by subtracting the threshold current from the input current.

[0107] As mentioned, the configuration of FIG.2A and FIG.2B does not provide the ideal profile for the discharge of the damping capacitor 114. Having in mind the profile to be obtained shown in FIG.1C, aspects of the present disclosure are related to a simple and more effective solution based on that theorical evaluation, i.e., to a configuration of the discharge current generator circuit capable of reproducing the hyperbolic current profile. In particular, aspects of the present disclosure are based on the realization that a translinear loop provides a current configuration that may achieve the hyperbolic behavior for the discharge current.

[0108] Thus, in a preferred configuration shown in FIG.3A a discharge current generator circuit 300 includes a translinear loop. Considering the configuration of FIG.3A, the discharge current has a hyperbolic behavior with respect to the input voltage at the input terminal 108 of the driver circuit 106. FIG.3A shows thus a discharge current generator circuit 300, as a possible realization of the discharge current generator circuit 130.

[0109] In general, to implement the “hyperbolic discharge” (or parabolic discharge), the discharge current generator circuit 300 may include a translinear loop. The translinear loop may include an even number of translinear elements, configured such that the product of the currents through a first subset (a first half) of translinear elements is equal to the product of the currents through a second subset (a second half) of translinear elements. Stated differently, the sum of the voltage drops across the first subset of translinear elements is equal to the sum of the voltage drops across the second subset of translinear elements. The first subset of translinear elements may thus contribute in an opposite manner with respect to the second subset of translinear elements to the equation of the circuit.

[0110] In particular, as shown in FIG.3A, the discharge current generator circuit 300 may include four translinear elements, i.e., a first translinear element 302, a second translinear element 304, a third translinear element 306, and a fourth translinear element 308. The firsttranslinear element 302 and the second translinear element 304 are configured such that a first current through the first translinear element 302 and a second current through the second translinear element 304 have a same first sign in the loop equation. The third translinear element 306 and the fourth translinear element 308 are configured such that a third current through the third translinear element 306 and a fourth current through the fourth translinear element 308 have a same second sign in the loop equation, opposite to the first sign.

[0111] Illustratively, the translinear loop may consists of four translinear elements, two of which 302, 304 contribute in a first manner to the equation of the circuit, and two others of which 306, 308 contribute in a second, opposite manner to the equation of the circuit. Stated differently, the sum of a first voltage drop across the first translinear element 302 and a second voltage drop across the second translinear element 304 may be equal to the sum of a third voltage drop across the third translinear element 306 and a fourth voltage drop across the fourth translinear element 308.

[0112] As generally known, a translinear element may be a type of electronic device having a transconductance that is linearly proportional to the current flowing through the translinear element. A translinear element may have an exponential current-voltage relationship, such that the current flowing through the translinear element has an exponential relationship with the voltage drop across the translinear element.

[0113] In a preferred configuration, the discharge current generator circuit 300 may include bipolar junction transistors as translinear elements, e.g., a first BJT 302, a second BJT 304, a third BJT 306, and a fourth BJT 308. For example, the first BJT 302 and the second BJT 304 may be in a diode-connected configuration, as discussed in further detail below. BJT may be the preferred type of translinear element as they have been found to enable a very precise operation, and accordingly a very precise “triggering” of the discharge. In the following discussion some terminology may be used that pertains to the BJT-context (e.g., base, emitter, etc.). It is however understood that in principle other types of translinear elements may be used. As another example, the discharge current generator circuit 300 may include MOSFETs as translinear elements. As a further example, the discharge current generator circuit 300 may include diodes as translinear elements.

[0114] Before describing the generation of the discharge current, a brief explanation of the operation of a translinear loop is provided. As mentioned, the loop is made of four devices and forces the sums of two element drops to be equal. Denoting the first translinear element 302 as QI, the second translinear element 304 as Q2, the third translinear element 306 as Q3, the fourth translinear element 308 as Q4, and considering BJTs, the sum of the base-emitter voltage(VBEI+VBE2) of QI and Q2 is equal to the sum of the base-emitter voltage (VBES+VBEQ of Q3 and Q4. This configuration allows setting a careful control of the biasing current. An exemplary implementation of this technique is the well-known push pull stage.

[0115] The same principle, assuming the use of B JT devices that offer a simpler mathematical demonstration, can be applied to generate a desired bias current dependence. Considering VBEI + VBE2 - VBE3 = VBE4, and considering that the currents across QI Q2 and Q3 are set, the bias for Q4 is automatically generated. The linear relationship between voltage drop VBE can be written in terms of current products and ratios, as I4 = Ii*l2 / l3, where li denotes the current through the i-th translinear element (also denoted herein as IQI, e.g., IQI, IQ2, IQ3, IQ4). Considering that VBE = kT / q*ln(I / Io), the relationship VBEI + VBE2 = VBE3 + VBE4 may be written as IQI*IQ2=IQ3*IQ4. Conveniently selecting the bias currents for the translinear elements 302, 304, 306, 308 (e.g., the BJT bias currents), it is possible to generate a current in respect to the desired relationship.

[0116] Being li, I2, and I3 independently set, they may be generated as proportional to a given reference voltage (e.g., by means of a resistive element, e.g., a resistor), so that I4 (illustratively, the output current of the translinear loop) may be inversely proportional to the reference that generates I3 and proportional to the references generating li and I2. In the particular case that li and I2 are from the same reference, I4 would result as proportional to the power of the associated reference. The reference may be a bandgap, just to give a fixed contribute, a PTAT current, where PTAT stands for proportional to absolute temperature (to generate either a PTAT or CTAT current, where CTAT stands for complementary to absolute temperature) of any other possible voltage. In this case the circuit allows obtaining the desired hyperbolic dependence for I4, and discharge the capacitor through the discharge path 132 (e.g., through the load 122).

[0117] Turning back to the equation above ILOAD = ILOAD nom*(VDD th-VLOAD) / (VDD-VLOAD), a BJT -based translinear loop may thus provide a suitable implementation to obtain the desired discharge current (e.g., a desired current through the load to discharge the damping capacitor).

[0118] Turning back to FIG.3A, the translinear loop may include an input branch 310, an output branch 330, and an intermediate branch 320 disposed between the input branch 310 and the output branch 330. The input branch 310 may be configured to receive the reference current, for example the input branch 310 may include an input terminal 312 coupled with a (second) reference current source 301 of the driver circuit 106. The (second) reference current source 301 may be a replica of the current source 118 and may generate a (second) reference current 303 (equal to the reference current 144).

[0119] The output branch 330 may be coupled with the discharge path 132 for the discharge current. For example, the output branch 330 may include an output terminal 332 coupled with the discharge path 132 (e.g., the output terminal 332 may be coupled with the load terminal 120). The intermediate branch 320 may be configured to receive a threshold current 322 and an input current 324 that is proportional to the input voltage at the input terminal 108 of the driver circuit 106.

[0120] The input branch 310 may include the first translinear element 302 and the second translinear element 304 in series with one another. The first translinear element 302 may be coupled with the input terminal 312, and the second translinear element 304 may be coupled with a reference terminal 350 for the circuit (e.g., the reference terminal 117 or another terminal at the same potential as the reference terminal 117). Considering the BJT implementation, the collector of the first BJT 302 may be coupled with the input terminal 312 to receive the reference current. The emitter of the first BJT 302 may be coupled with the collector of the second BJT 304. The emitter of the second BJT 304 may be coupled with the reference terminal 350. Considering the diode-connected configuration, the base and the collector of the first BJT 302 may be coupled with one another, and the base and the collector of the second BJT 304 may be coupled with one another.

[0121] The intermediate branch 320 may include the third translinear element 306 receiving the threshold current 322 and the input current 324. The third translinear element 306 may be configured such that the third current through the third translinear element 306 is obtained by subtracting the threshold current 322 from the input current 324. Considering the BJT implementation, the third BJT 306 may receive the threshold current 322 and the input current 324 at the emitter. The base of the third BJT 306 may be coupled with the base of the first BJT 302. The threshold current 322 may be representative of the overvoltage condition (as discussed for the threshold current 206 in FIG.2A), and the input current 324 may represent the input voltage, and accordingly the supply voltage (as discussed for the input current 202 in FIG.2A).

[0122] The output branch 330 may include the fourth translinear element 308, to provide the fourth current through the fourth translinear element 308, which is “automatically” set by the loop equation, as output current 334 at the discharge path 132. Considering the BJT implementation, the collector of the fourth BJT 308 may be coupled with the output terminal 332, and the emitter of the fourth BJT 308 may be coupled with the reference terminal 350. The base of the fourth BJT 308 may be coupled with the emitter of the third BJT 306.

[0123] By calculating the expression for the current across Q3, it holds that current generated by Q4 has the expression IQ4 = IQI*IQ2* / (IIN-ITH). The translinear loop may thus be configuredsuch that the output current 334 at the output branch 330 is proportional to a factor that includes a difference between the input current 324 and the threshold current 322. Considering the reference current at the input branch 310, the translinear loop may be configured such that the output current 334 at the output branch 330 is given by the reference current multiplied by the factor that includes the difference between the input current 324 and the threshold current 322, as discussed in further detail below. This relationship allows obtaining the hyperbolic profile for the discharge.

[0124] Turning to the generation of the bias currents for the translinear elements 302, 304, 306, a preferred configuration may include using resistive elements (e.g., resistors) over which a voltage drop is provided. Such configuration allows tuning the resistance values of the resistive elements to tailor the output current 334.

[0125] According to various aspects, the discharge current generator circuit 300 may include a first resistive element coupled with the intermediate branch 320, e.g., the first resistive element may be coupled with the emitter of the third BJT 306. The first resistive element (e.g., a first resistor) may have a first resistance value R1. In this configuration, the discharge current generator circuit 300 may be configured such that the input current 324 is given by a drop of the input voltage across the first resistive element (e.g., a drop of the supply voltage across the first resistive element VDD / R1). The voltage drop across the first resistive element may illustratively define an input current source 326 providing the input current 324.

[0126] In some aspects, the discharge current generator circuit 300 may further include a threshold current source 328 configured to generate the threshold current 322. As an exemplary realization, the threshold current source 328 may include a bandgap voltage generator and a resistive element (e.g., a resistor) to obtain the threshold current 322 as a voltage drop of the bandgap voltage across the resistive element. In an exemplary configuration, the threshold current is given by a drop of a threshold voltage across a resistive element equal to the first resistive element (e.g., a resistive element having the same resistance value Ri, thus providing a threshold current as Vth / Ri, where Vth is the threshold voltage of the overvoltage condition).

[0127] In this scenario, turning back to the expression for the current across Q3, it follows that the current generated by Q4 has the expression IQ4 = lQi*lQ2*Ri / (VDD-Vth). This expression shows the inverse proportionality to the difference between the supply voltage and a threshold.

[0128] Turning to the first current through the first translinear element 302 and the second current through the second translinear element 304, a further useful relationship may be provided. In particular, the first current through the first translinear element 302 may be the reference current IREF, and the second current through the second translinear element 304 maybe defined by a voltage drop across a second resistive element (e.g., a second resistor), e.g., having a resistance value R2. For example, the second current may be defined by a bandgap voltage drop across the second resistive element. In this scenario, the (bandgap) voltage generator and the second resistive element may define a further current source 314 for delivering the (second) current to the second translinear element 304. Q2 may thus receive a current that depends on a precise voltage (e.g., a bandgap voltage, VBG, which is precise and temperature stable, being a silicon physic property) divided by the value of a resistive element, R2.

[0129] In this scenario, the translinear loop may thus be configured such that the factor for multiplying the reference current further includes a ratio of a first resistance value of the first resistive element to a second resistance value of the second resistive element. Considering this configuration, the current generated by Q4 may be expressed as IQ4 = IREF* Ri / R2*[VBG / (VDD-Vth)]. Thus, considering that the voltage VDD th-VroAD may be expressed as a multiple (R1 / R2) of the bandgap voltage, the translinear loop allows obtaining the ideal expression for the discharge current (e.g., ILOAD through the load, e.g., a LED discharge current ILED).

[0130] A closer look to this expression shows that IQ4 is a scaled replica of the reference current 303, IREF, sharing the same temperature and process dependence, being the scaling factor inversely proportional to the supply voltage. In addition, the ratio between R1 and R2 is independent on the production process spread, giving another relevant advantage over the solution proposed in FIG.2A. Illustratively, although resistance values may vary due to process variations, they do so in the same manner on the same chip, so that exploiting a ratio between resistance values cancels out possible process-related variations, and enhances the accuracy of the proposed approach.

[0131] As discussed in relation to FIG.1 A, the reference current 144, 303 IREF may be a scaled down version of the nominal current for driving the load (e.g., the nominal LED current), which is then scaled up to obtain the driving current (e.g., via a current mirror 142). It follows that if the nominal current for driving the load is trimmed, also the current IQ4 through the fourth translinear element 308 will be trimmed. Furthermore, if the load current benefits of a temperature compensation drift or a derating factor for reliability strength at high temperature, the same feature is automatically transferred to the current IQ4 through the fourth translinear element 308.

[0132] From the perspective of a practical design, considering that the current IQ4 through the fourth translinear element 308 may be used to bias the load 122 (e.g., a LED array), thecontribute Vth / Ri in FIG.3A may be considered as the mean to shift along the X-axis the dependence of the generated current vs. the input voltage or supply voltage (VDD), while the term R1 / R2 determines a scaling factor to properly set the IQ4 current at a given voltage (e.g., a given battery voltage).

[0133] In this regard, FIG.3B shows a first graph 360a illustrating various settings of the properties of the circuit of FIG.3A. The load current ILOAD is represented as a function of the input voltage (e.g., a battery voltage). Starting from the first curve 362, the current Vth / Ri is firstly increased to shift the origin of the hyperbolic profile. The second curve 364 is obtained. Then, the corner point at 20 V is recovered by means of an adjustment in the value of R2 (third curve 366). The comparison between the original curve 362 and the new one 366 shows that the same corner threshold can be associated to different deratings of the load current at supply above the corner.

[0134] The second graph 360b provides a comparison of the two solutions in terms of power dissipation. The original solution (represented by the first curve 372) has a derating factor that keeps the power consumption constant, while the larger derating associated to the new solution (second curve 374), produces a progressive reduction of the power dissipation at increased voltages (e.g., increased battery voltages).

[0135] Considering the preferred scenario in which the discharge path 132 is provided by the load 122, the discharge current generator circuit 130 of the present disclosure may be further configured to selectively deliver to the load terminal 120 either the current for driving the load or the discharge current for discharging the damping capacitor 114. Illustratively, the discharge current generator circuit 130 may be configured to cause the flow of discharge current through the load (e.g., inside the LEDs) only in case the input voltage fulfills the overvoltage criterion (e.g., only when the supply exceeds the overvoltage threshold), while ensuring that the load current remains at the nominal value during normal operation. In this regard, a digital approach (FIG.4) or an analog approach (FIG.5 and FIG.6A) may be provided.

[0136] Considering the digital approach, the driver circuit 106 may further include a multiplexer (e.g., as part of the discharge current generator circuit 130) configured to selectively allow either a flow of the reference current towards the load terminal 120, or a flow of the output current of the discharge current generator circuit 130 towards the load terminal 120. The multiplexer may illustratively act as a switch to deliver to the load 122 the reference current (e.g., its amplified version) during normal operation, or the discharge current during overvoltage.

[0137] FIG.4 shows a discharge current generator circuit 400 including a digital multiplexer 402. In the configuration in FIG.4, the multiplexer is shown with reference to the configuration of the discharge current generator circuit 300 with a translinear loop, as this may represent the most relevant scenario. It is however understood that in principle the multiplexer may also be introduced in the discharge current generator circuit 200 of FIG.2A and FIG.2B.

[0138] As shown in FIG.4, the discharge current generator circuit 400 may be generally configured as the discharge current generator circuit 300, and may further include a multiplexer 402 configured to selectively allow either a flow of the reference current 144 towards the load terminal 120 (e.g., towards the current mirror 142) for driving the load 122, or a flow of the output current 334 from the output branch 330 of the translinear loop towards the load terminal 120 (e.g., towards the current mirror 142) for causing the discharge of the damping capacitor 114 through the load 122.

[0139] Illustratively, the multiplexer 402 may be configured to selectively connect with the load terminal 120 (e.g., with the input terminal 147 of the current mirror 142) either an electrically conductive path to the reference current source 118, or an electrically conductive path to the output branch 330 of the translinear loop (e.g., an electrically conductive path to the output terminal 332 coupled with the fourth translinear element 308).

[0140] In this digital implementation, the multiplexer 402 receives the reference current 144 from the reference current source 118 (e.g., the nominal current for the load ILOAD nom) and the output current from the discharge current generator circuit 400 (e.g., the output current from the translinear loop in the configuration of FIG.4), and the multiplexer 402 allows selecting which of the two currents crosses the load 122 (e.g., which of the two currents crosses the LED string). As discussed above, the proposed configuration allows making the two currents nominally equal to one another at the overvoltage threshold, just playing on the Ri / R2 factor. In this way, the transition from one current to the other does not include any jump, at least nominally, and this prevents any switching noise.

[0141] This solution shows also a remarkable advantage. The selection between the reference current 144 IREF and the output current 234 IQ4 allows to have, in nominal condition, the value for the load current ILOAD (e.g., ILED) that is not influenced by the presence of the fast discharge circuitry, bringing evident benefits to the overall accuracy in the load drive.

[0142] In the configuration with the multiplexer, the discharge current generator circuit 130, 400 may further include a control element configured to send a control signal to the multiplexer 402 to cause the selection of either the reference current 144 or the output current 434 for delivery to the load terminal 120. Illustratively, the control element may be configured to senda first control signal to the multiplexer 402 to cause selection of the output current 434 for delivery to the load terminal 120 if the input voltage value fulfills the overvoltage criterion, and may be configured to send a second control signal to the multiplexer 402 to cause selection of the reference current 144 for delivery to the load terminal 120 if the input voltage value does not fulfill the overvoltage criterion.

[0143] As an exemplary realization, the control element may include a comparator configured to compare the input voltage (or supply voltage) with the overvoltage threshold. The comparator may deliver a first output signal as first control signal to the multiplexer 402 if the input voltage is equal to or greater than the threshold voltage associated with the overvoltage, and may deliver a second output signal as second control signal to the multiplexer 402 if the input voltage is less than the threshold voltage associated with the overvoltage. For example, the first output signal may be a logic 1 (e.g., a high voltage), and the second output signal may be a logic 0 (e.g., a low voltage), although the definition may be arbitrary.

[0144] Although efficient, the digital multiplexing may be the source of an unwanted phenomenon. In case the current provided as output by the discharge current generator circuit (e.g., the output current 334, IQ4) is appreciably larger than the reference current 144 IREF at the trip point Vth, the current delivered to the load is increased as soon as the discharge current is selected. For example, considering LEDs, this may lead to a sudden increase in light emission. An analog implementation, shown in FIG.5A and FIG.5B, provides a solution to this problem, although with a less intuitive design compared to the digital approach.

[0145] FIG.5A and FIG.5B show a discharge current generator circuit 500 including a current mirror 502 acting as analog multiplexer. In the configuration in FIG.5A and FIG.5B, the current mirror 502 is shown with reference to the configuration of the discharge current generator circuit 300 with a translinear loop, as this may represent the most relevant scenario. It is however understood that in principle the analog multiplexer 502 may also be introduced in the discharge current generator circuit 200 of FIG.2 A and FIG.2B.

[0146] As shown in FIG.5 A and FIG.5B, the discharge current generator circuit 400 may be generally configured as the discharge current generator circuit 300, and may further include a current mirror 502 acting as analog multiplexer configured to selectively allow either a flow of the reference current 144 towards the load terminal 120 (e.g., towards the current mirror 142) for driving the load 122, or a flow of the output current 334 from the output branch 330 of the translinear loop towards the load terminal 120 (e.g., towards the current mirror 142) for causing the discharge of the damping capacitor 114 through the load 122. It is to be noted that the current mirror 502 may be configured to have a unitary gain, but it is trivial to scale up if desired.

[0147] In this configuration, the driver circuit 106 may further include (e.g., as part of the discharge current generator circuit 500) a current mirror 502 coupled between the output of the discharge current generator circuit 130, 300, 500 and the load terminal 120. For example, the current mirror 502 may be coupled between the output branch 330 of the translinear loop and the load terminal 120 (e.g., the input terminal 147 of the current mirror 142).

[0148] The current mirror 502 may include input branch configured to receive a replica of the reference current 512, and the output current of the discharge current generator circuit 500 (e.g., the output current 334 of the translinear loop). For example, the input branch may be coupled with a (third) reference current source 510 configured to generate the (third) reference current 512 as a replica of the reference current 144 provided by the reference current source 118. The (third) reference current source 510 and reference current source 118 may be configured such that the replica of the reference current 512 and the reference current 144 are equal to one another. The replica of the reference current 512 may also be referred to herein simply as reference current 512 (or third reference current 512).

[0149] The current mirror 502 may include an output branch coupled with the load terminal 120 (e.g., an output branch coupled with the input terminal 147 of the current mirror 142), and may be configured to transfer a difference between the reference current 512 and the output current at the output branch, if the output current of the discharge current generator circuit 500 is less than the reference current 512.

[0150] Illustratively, the current mirror 502 may be configured to be conductive if the output current of the discharge current generator circuit 500 (e.g., the output current 334) is less than the reference current 512, thereby delivering at towards the load terminal 120 a current given by subtracting the output current of the discharge current generator circuit 500 from the reference current 512 (e.g., IREF-IQ4). The current mirror 502 may be configured to be not conductive if the output current of the discharge current generator circuit 500 is greater than the reference current 512 (as this may indicate a non-overvoltage condition).

[0151] As mentioned in relation to FIG.1 A, the reference current source 118 may be coupled with the load terminal 120 (or with the input terminal 147 of the current mirror 142), such that when the current mirror 502 is not conductive, the reference current 144 from the (first) reference current source 118 is delivered towards the load terminal 120. On the other hand, when the current mirror 502 is or becomes conductive, the current at the output branch of the current mirror 502 is subtracted from the reference current 144 from the reference current source 118, thus providing towards the load terminal 120 a discharge current corresponding to the output current of the discharge current generator circuit, illustratively a current given bysubtracting from the reference current 144 the difference between the replica of the reference current 512 and the output current 334, IREF-(IREF-IQ4) = IQ4.

[0152] As an exemplary realization, the current mirror 502 may include a first transistor element 504 (e.g., a first NMOS, Mn) at the input branch, and a second transistor element 506 (e.g., a second NMOS, Mg) at the output branch. Considering for example NMOS transistors, the drain of the first NMOS may be coupled with the output terminal 332 of the translinear loop and with a source of reference current. The source of the first NMOS may be coupled to the source of the second NMOS (and to the reference terminal 350). The gate of the first NMOS may be coupled to the gate of the second NMOS, and the drain of the second NMOS may be coupled to the load terminal 120 (e.g., over the input terminal 147 of the current mirror 142). The first NMOS may be in a diode-connected configuration, and may have the gate coupled to the drain.

[0153] The current mirror 502 may thus provide an analog implementation to select IQ4 instead of IREF. If IQ4 is larger than IREF, the first transistor element 504 Mn is off so that the load current ILOAD is defined by the reference current IREF. AS soon as IQ4 gets lower than IREF, Mn is turned ON to drive a current IREF-IQ4 at the output branch of the current mirror 502. A replica subtracts this current from IREF SO that the load is biased with a current defined by IQ4 (e.g., a current equal to IQ4).

[0154] As discussed above, the current IQ4 may be very large when the supply voltage is small, then IQ4 rapidly decreases for increasing supply voltages. So, at first, the ratio between the resistance value of the first resistive element to the resistance value of the second resistive element (R1 / R2) may be selected to make IQ4 equal to IREF at the overvoltage threshold. It follows that the current generator Mg is totally OFF when the supply voltage is below the overvoltage threshold. In this case, the load current ILOAD may be provided as a multiple (by a factor K) of the reference current IREF. Thanks to the rectification from the current mirror 502, this value cannot be exceeded in any way and in any condition. Therefore, the desired protection against excessive currents is implemented and ensured.

[0155] In addition, as in the case of the digital multiplexer 402, the Mg in the OFF status ensures that the accuracy of the load current depends only on the reference current source 118 in nominal conditions. Thus, the nominal accuracy is the same, regardless of the presence of the fast discharge circuit.

[0156] When the supply voltage rises above the overvoltage threshold, IQ4 drops lower than IREF and the current mirror 502 starts conducting a current equal to IREF-IQ4. Subtracting from IREF, this makes the load current equal to K*IQ4, as desired, to discharge the overshoot. In theanalog implementation, the transition from IREF to IQ4 is not determined by any switch and it is automatic and continuous. This prevents any switching noise, and may thus be implemented without the need for smoothing filters.

[0157] The analog implementation of FIG.5 A may thus provide various benefits. For example, it ensures that the load current is defined by the reference current IREF when the supply voltage is below the overvoltage threshold. It ensures that the load current can never exceed the reference current IREF (e.g., so that no increase of light emission can be observed). Finally, the modulation of the load current at the crossover point between IREF and IQ4 is continuous and smooth.

[0158] An evolution of the discharge current generator circuit 500 is shown in the configuration 500b in FIG.5B.

[0159] In general, a driver circuit 106 (e.g., a LED driver) may be designed to generate a set of possible driving currents, not a single specific one. Thus, in various aspects, the profile of the discharge current (e.g., the profile of the load current, e.g., the LED current) to discharge the overshoot may be corrected accordingly.

[0160] It may appear straightforward to scale of the same amount the three fixed current generators, i.e., the three reference current sources 118, 301, 510 in FIG.5A (Ia, L, lb). The load current will be scaled down in the same way. However, this solution is not optimal, because in case of a small nominal load current, the discharge during an overvoltage event will take proportionally longer. At the same time, no appreciable advantages would be obtained regarding reliability issues, because the package thermal resistance is the same for all the settings. This suggests not to scale the profile of the current in overvoltage conditions and keep it independent from the current settings of the load.

[0161] With this in mind, the following observations regarding the configuration of FIG.5A may be of interest. First, the load current cannot exceed the one supplied by the first reference current source 118 (Ia). Second, at low supply voltage the current mirror 502 (e.g., the second transistor element 506) is totally OFF, and the first reference current source 118 sets the load current for the nominal case. Third, above the overvoltage threshold, the load current is set by IQ4 considering that the reference current 512 (lb) from the third reference current source 510 is equal to the reference current 144 (Ia) from the first reference current source 118.

[0162] In this way, it is sufficient that Iaand lb track the setting for the load current while the second reference current 303 (L) from the second reference current source 301 is left independent from the setting for the load current, and corresponds to the maximum load current value. In this scenario, IQ4 is the same for all the configurations and corresponds to themaximum allowed value by reliability. This ensures the optimal discharge of the supply overshoot in all current settings. The reliability of the load cannot be affected by a value of IQ4 that is not scaled down. Indeed, the load current cannot exceed the la value in any case.

[0163] The configuration 500b of FIG.5B may thus be a preferred implementation of the proposed approach, in which the driver circuit 106 includes the first reference current source 118 configured to generate the (first) reference current 144b, the third reference current source 510 configured to generate the (third) reference current 512b replicating the (first) reference current 144b. Illustratively, the (first) reference current 144b and the (third) reference current 512b may have the same value, e.g., a fraction of the driving current for driving the load 122 (e.g., a fraction by a factor K, provided by the current mirror 142), ILOAD / K.

[0164] The driver circuit 106 may further include the second reference current source 301 configured to generate the (second) reference current 303b, and the second reference current source 301 may be configured such that the (second) reference current 303 has a greater current value compared to the (first) reference current 144b and the (third) reference current 512b. In particular, the (second) reference current 303 may be related to the maximum current value allowed by the load. For example, the (second) reference current 303b may be a fraction of such maximum current value, (e.g., a fraction by a factor K, provided by the current mirror 142), ILOAD max / K (for example, lLED_max / K). Illustratively, the second reference current source 301 may be configured to deliver a current defined by the maximum current value allowed by the programming of the load 122, e.g., the maximum regulation allowed by the package.

[0165] This setting for the (second) reference current 303b reflects in the output current of the discharge current generator circuit 500 (e.g., the output current 334 of the translinear loop), which is also increased compared to the scenario of FIG.5A with respect to the value of the reference current.

[0166] The resulting waveforms are shown in the graph 550 in FIG.5C, which illustrates the current profile for different load current settings. The graph 550 shows the load current profile (left sub-window) and power dissipation curves (right sub-window) for different settings of the nominal load current. As it can be appreciated, the derating profile is the same while the nominal values are different. A remarkable consequence is that, to make both conditions hold, the supply voltage where the current starts being reduced moves at a higher value when the load current is set at a smaller value. The current varies according to the setting in the nominal supply range while it becomes identical at large supply voltages, to keep the same power dissipation and make the discharge performance optimal. Remarkably, the load current profile is monotonic, as expected by the device implementation, to comply with the load reliability.

[0167] FIG.6 shows a circuit 600 in a schematic representation, according to various aspects. The circuit 600 may be an exemplary realization of the circuit 100. In general, the circuit 600 may include a supply terminal 602 to receive a supply voltage 604 (e.g., a battery voltage VBAT). The circuit 600 may further include a driver circuit 606 (an exemplary realization of the driver circuit 106) with an input terminal 608 to receive the supply voltage 604 as input voltage. The circuit 600 may further include a protection circuit 610 disposed between the supply terminal 602 and the input terminal 608 to protect the driver circuit 606 from an undesired behavior of the supply voltage 604.

[0168] In the exemplary configuration in FIG.6, the protection circuit 610 may include a diode 612 (as filter element) to block a negative portion of the supply voltage 604, a damping capacitor 614, CFILT coupled in parallel between the input terminal 608 and ground 616. The protection circuit 610 may further include a further diode 632 in parallel with the damping capacitor 614, as additional protection element to block residuals of the negative portion of the supply voltage 604.

[0169] As another exemplary configuration, the protection circuit 610 may include the diode 612 and the damping capacitor 614, and a further protection element, e.g. a further diode, a TVS, a VDR, or the like. The protection circuit 610 may further include a high voltage transistor, e.g., a high voltage MOS (such as a NMOS), and a differentiator configured to determine a variation of the supply voltage 604 over time, and generate a control signal for the transistor based on the determined variation. The high voltage transistor may provide a further discharge path for the damping capacitor 614 in case of overvoltage.

[0170] Turning now to the driver circuit 606, the driver circuit 606 may include, in addition to the input terminal 608, various further terminal for coupling with external circuit and components. For example, the driver circuit 606 may include one or more data terminals 640 for data transmission and reception. The data terminals 640 may include, for example, a chip select terminal, CS, to receive a chip select signal; a clock terminal, SCLK, to receive a clock signal, a multiple input single output, MISO, to receive data, and a multiple output single input, MOSI, to output data. As another example, the driver circuit 606 may include one or more test terminals 642 at which test signals may be provided to carry out a testing of the driver circuit 606, e.g., a testing of one or more components of the driver circuit 606. The driver circuit 606 may be coupled with ground 617.

[0171] The driver circuit 606 may further include a digital processing circuit 644 (a digital core) coupled with the data terminals 640 and configured to carry out digital processing, e.g. for selecting current levels for the operation of an external series element 624, to implementderating functions, to bypass part of the drivers, etc. For example, the digital processing circuit 644 may be coupled with a memory 658 configured to store instructions for the digital processing circuit 644. For example, the memory 658 may store data and parameters for an operation of the driving circuit 606. As an exemplary implementation, the memory 658 may be a one-time programmable (OTP) memory.

[0172] The driver circuit 606 may further include a reversal protection circuit 646 configured to protect the driver circuit 606 from reverse input voltage. Illustratively, the reversal protection circuit 646 may be configured to block a negative portion of the input voltage at the input terminal 608. Considering the scenario in which the circuit 600 is coupled with a battery as supply source, the reversal protection circuit 646 may be referred to as battery reversal. The driver circuit 606 may further include an overvoltage detection circuit 626 configured to monitor the input voltage (e.g., as output by the reversal protection circuit 646). The driver circuit 606 may further include a discharge current generator circuit 670, e.g., configured as any of the discharge current generator circuits 130, 200, 300, 400, 500 described above.

[0173] The driver circuit 606 may further include a power management circuit 650 coupled with the reversal protection circuit 646 and the overvoltage detection circuit 626. The power management circuit 650 may be configured to control an operation of the reversal protection circuit 646 and the overvoltage detection circuit 626, e.g. to adjust one or more operating parameters of such circuits. The power management circuit 650 may be further coupled with a further capacitor 652, CLDO, disposed externally to the driver circuit 606. The further capacitor 652 may be referred to herein as low-dropout capacitor, CLDO. The internal power management circuit 650 may be an internal low-dropout regulator to generate a fixed voltage independently of the input voltage, thus providing a controlled voltage value for powering the components that require a small supply voltage, or in general a smaller supply voltage than other components of the driver circuit 606.

[0174] Regarding temperature protection, the driver circuit 606 may include a temperature sensor 654 and a thermal shutdown circuit 656. The temperature sensor 654 may be configured to sense a temperature of the driver circuit 606, and the thermal shutdown circuit 656 may be a hardwired circuit configured to force a shutdown of the driver circuit 606 if the sensed temperature indicates an over-temperature condition, e.g., if the sensed temperature is greater than a threshold temperature. Additionally or alternatively, the driver circuit 606 may include an over-temperature protection circuit configured to instruct a shut down if the internal temperature of the driver circuit 606 exceeds the maximum safe operating temperature. With respect to the thermal shutdown circuit 656, the over-temperature protection circuit may operateat the digital / software level, e.g., the over-temperature protection circuit may send an over-temperature signal to the digital processing circuit 644 based on the temperature sensed by the temperature sensor 654. The over-temperature signal may instruct the digital processing circuit 644 to initiate a shutdown of the driver circuit 606.

[0175] The driver circuit 606 may further include a control circuit 630 configured to control the external series element 624. In the exemplary configuration in FIG.6, the external series element 624 coupled with the driver device 606 may include a plurality of light emitting diodes 660, e.g. four LEDs as an example. In this configuration the control circuit 630 may be referred to as LED control circuit, and may be configured to control a light emission of the LEDs. For example, the control circuit 630 may be configured to control a corresponding transistor 662 for each LED 660 to enable or disable the current flow through the LEDs 660. For example, the transistors 662 may be an example of internal series element of the driver circuit 606.

[0176] As discussed above, in a conventional configuration, during positive surge pulses the voltage present after the reverse protection (that blocks negative peaks), e.g. after the diode 612, and the damping capacity 614 tends to remain high, stressing or making the driver 606 not work due to the intervention of the overvoltage 648 or thermal protections 654, 656. Illustratively, the overvoltage comparator above the maximum operating VBAT turns off the current of series of LEDs (II) to avoid thermal damage to the device. In this condition, the discharge current of filter capacitor 614 is the constant absorption current (few mA) of internal blocks as power management 650, monitors 648 and digital block 644. On the contrary, in the proposed approach, the filter capacitance 614 may be discharged using the current flowing through the discharge path using the discharge current from the discharge current generator circuit 670.

[0177] The terms “processor”, “processing circuit”, or “control circuit” as used herein may be understood as any kind of technological entity that allows handling of data. The data may be handled according to one or more specific functions that the processor / processing circuit / control circuit may execute. Further, a processor / processing circuit / control circuit as used herein may be understood as any kind of circuit, e.g., any kind of analog or digital circuit. A processor / processing circuit / control circuit may thus be or include an analog circuit, digital circuit, mixed-signal circuit, logic circuit (e.g., a hard-wired logic circuit or a programmable logic circuit), microprocessor, Central Processing Unit (CPU), Graphics Processing Unit (GPU), Digital Signal Processor (DSP), Field Programmable Gate Array (FPGA), integrated circuit, Application Specific Integrated Circuit (ASIC), etc., or any combination thereof. It is understood that any two (or more) of the processors / processing circuits / control circuits detailed herein may be realized as a single entity with equivalent functionality or the like, and converselythat any single processor / processing circuit / control circuit detailed herein may be realized as two (or more) separate entities with equivalent functionality or the like.

[0178] The term “connected” may be used herein with respect to terminals, integrated circuit elements, devices, and the like, to mean electrically connected, which may include a direct connection or an indirect connection, wherein an indirect connection may only include additional structures in the current path that do not influence the substantial functioning of the described circuit or device. The term “electrically conductively connected” that is used herein to describe an electrical connection between one or more terminals, devices, regions, contacts, etc., may be understood as an electrically conductive connection with, for example, ohmic behavior, e.g. provided by a metal or degenerate semiconductor in absence of p-n junctions in the current path. The term “electrically conductively connected” may be also referred to as “galvanically connected”. The term “coupled” may be used herein in the same manner as the term “connected”.

[0179] The term “terminal” may be used herein to describe a location (e.g., a point) or structure of a device or of an element of the device at which a signal (e.g., an analog signal, for example a current or a voltage) may be provided and / or to which another device or element may be connected. Illustratively, a terminal may be a location or a structure that is electrically conductively connected with the device or the element. A terminal may also be referred to herein as port, pin, contact, or contact point.

[0180] The word “exemplary” is used herein to mean “serving as an example, instance, or illustration”. Any embodiment or design described herein as “exemplary” is not necessarily to be construed as preferred or advantageous over other embodiments or designs.

[0181] The phrase “at least one” and “one or more” may be understood to include a numerical quantity greater than or equal to one (e.g., one, two, three, four, [...], etc.). The phrase “at least one of’ with regard to a group of elements may be used herein to mean at least one element from the group consisting of the elements. For example, the phrase “at least one of’ with regard to a group of elements may be used herein to mean a selection of: one of the listed elements, a plurality of one of the listed elements, a plurality of individual listed elements, or a plurality of a multiple of individual listed elements.

[0182] All acronyms defined in the above description additionally hold in all claims included herein.

[0183] While the invention has been particularly shown and described with reference to specific aspects, it should be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention asdefined by the appended claims. The scope of the invention is thus indicated by the appended claims and all changes, which come within the meaning and range of equivalency of the claims, are therefore intended to be embraced.List of reference signs 206 Threshold current208 Threshold current source100 Circuit 210 Current mirror100b Circuit configuration 212 Input terminal100c Circuit configuration 213 First transistor element102 Supply terminal 214 Output terminal104 Supply voltage 215 Second transistor element106 Driver circuit 220 Current limiter108 Input terminal 300 Discharge current generator circuit 110 Protection circuit 301 Reference current source112 Filter element 302 First translinear element114 Damping capacitor 303 Reference current116 Ground terminal 303b Reference current117 Ground terminal 304 Second translinear element 118 Reference current source 306 Third translinear element120 Load terminal 308 Fourth translinear element 122 Load 310 Input branch124 External series element 312 Input terminal126 Internal series element 314 Further current source130 Discharge current generator circuit 320 Intermediate branch132 Discharge path 322 Threshold current140 Inset 324 Input current142 Current Mirror 326 Input current source143 First transistor 328 Threshold current source144 Reference current 330 Output branch144b Reference current 332 Output terminal145 Second Transistor 334 Output current146 Driving current 350 Reference terminal150 Graph 360a Graph152 Curve 360b Graph154 Curve 362 First curve200 Discharge current generator circuit 364 Second curve202 Input current 366 Third curve204 Resistive element 372 First curveSecond curve 656 Thermal shutdown circuit Discharge current generator circuit 658 MemoryMultiplexer 660 Light emitting diode Discharge current generator circuit 662 Transistorb Discharge current generator circuit 670 Discharge current generator circuit Current mirrorFirst transistor elementSecond transistor elementReference current sourceReference currentb Reference currentGraph(s)CircuitSupply terminalSupply voltageDriver circuitInput terminalProtection circuitDiodeDamping capacitorGround terminalGround terminalCurrent sourceExternal series elementOvervoltage detection circuitControl circuitDiodeData terminalsTest terminalsDigital processing circuitReversal protection circuitPower management circuitCapacitorTemperature sensor

Claims

1. Claims1. A circuit (100) compri sing :3.a supply terminal (102) configured to receive a supply voltage (104); a driver circuit (106) comprising an input terminal (108) coupled with the supply terminal (102) to receive the supply voltage (104) as input voltage for powering an operation of the driver circuit (106);4.a damping capacitor (114) coupled in parallel with the input terminal (108) and ground (116) such that the damping capacitor (114) is charged by the supply voltage (104);5.wherein the driver circuit (106) further comprises a discharge current generator circuit (130) configured to cause a flow of a discharge current through a discharge path (132) if an input voltage value of the input voltage at the input terminal (108) fulfills an overvoltage criterion representative of an overvoltage condition of the driver circuit (106), thereby causing a discharge of the damping capacitor (114),6.wherein the discharge current generator circuit (130) is configured such that the discharge current has a behavior over time that is inversely related to a behavior of the input voltage at the input terminal (108).

2. The circuit (100) according to claim 1,8.wherein the discharge current generator circuit (130) is configured such that the discharge current is inversely proportional to the input voltage value according to a linear profile, a parabolic profile, or a hyperbolic profile.

3. The circuit (100) according to claim 1 or 2,10.wherein the overvoltage criterion comprises the input voltage value being greater than an overvoltage threshold, and wherein the discharge current generator circuit (130, 200) is configured such that the discharge current is inversely related to a difference between the input voltage value and the overvoltage threshold.

4. The circuit (100) according to any one of claims 1 to 3,12.wherein the driver circuit (106) further comprises:13.a load terminal (120) configured to be coupled with a load (122), and a reference current source (118) configured to generate a reference current (144) for providing a driving current (146) for driving the load (122), wherein the discharge current generator circuit (130) is configured such that the load (122) provides the discharge path (132) for the discharge of the damping capacitor (114).

5. The circuit (100) according to claim 4,15.wherein the discharge current generator circuit (130, 200) is configured to: receive an input current (202) proportional to the input voltage at the input terminal (108) of the driver circuit (106); and16.start generating the discharge current if the input current (202) is greater than a threshold current (206) representative of the overvoltage condition of the driver circuit (106),17.wherein the discharge current generator circuit (130, 200) is configured such that the discharge current is obtained by a subtraction of a difference between the input current (202) and the threshold current (206) from the reference current (144).

6. The circuit (100) according to claim 5,wherein the discharge current generator circuit (130, 200) further comprises a current limiter (220) configured to set a maximum current value for the current to be subtracted from the reference current (144).

7. The circuit (100) according to any one of claims 1 to 4,20.wherein the discharge current generator circuit (130, 300) comprises a translinear loop, wherein the translinear loop comprises:21.an input branch (310) configured to receive the reference current (303); an output branch (330) coupled with the discharge path (132) for the discharge current; and22.an intermediate branch (320) disposed between the input branch (310) and the output branch (330) and configured to receive a threshold current (322) representative of the overvoltage condition of the driver circuit (106) and an input current (324) proportional to the input voltage at the input terminal (108) of the driver circuit (106),23.wherein the translinear loop is configured such that an output current (334) at the output branch (330) is given by the reference current (303) multiplied by a factor that comprises a difference between the input current (324) and the threshold current (322).24.The circuit (100) according to claim 7,25.wherein the input branch (310) comprises a first translinear element (302) and a second translinear element (304),26.wherein the intermediate branch (320) comprises a third translinear element (306), and27.wherein the output branch (330) comprises a fourth translinear element (308),28.wherein the first translinear element (302), the second translinear element (304), the third translinear element (306), the fourth translinear element (308) are configured such that a product of a first current through the first translinear element (302) and a second current through the second translinear element (304) is equal to a product of a third current through the third translinear element (306) and a fourth current through the fourth translinear element (308).29.The circuit (100) according to claim 8,30.wherein the first translinear element (302), the second translinear element (304), the third translinear element (306), the fourth translinear element (308) are bipolar junction transistors.31.The circuit (100) according to any one of claims 7 to 9,32.wherein the discharge current generator circuit (130, 300) further comprises a first resistive element coupled with the intermediate branch (320); and wherein the input current (324) proportional to the input voltage at the input terminal (108) of the driver circuit (106) is given by a drop of the input voltage across the first resistive element.33.The circuit (100) according to any one of claims 8 to 10,34.wherein the first current through the first translinear element (302) corresponds to the reference current (303),35.wherein the second current through the second translinear element (304) is defined by a voltage drop across a second resistive element, and36.wherein the third current through the third translinear element (306) is given by the difference between the input current (324) and the threshold current (322), and37.wherein the translinear loop is configured such that the factor for multiplying the reference current (303) further comprises a ratio of a first resistance value of the first resistive element to a second resistance value of the second resistive element.

12. The circuit (100) according to any one of claims 8 to 11,38.wherein the discharge current generator circuit (130, 400) further comprises a digital multiplexer (402) configured to selectively allow either a flow of the reference current (144) to the load terminal (120) for driving the load (122), or a flow of the output current (334) from the output branch (330) of the translinear loop to the load terminal (120) for causing the discharge of the damping capacitor (114) through the load (122).

13. The circuit (100) according to claim 12,40.wherein the discharge current generator circuit (130, 400) further comprises a control element configured to:41.send a first control signal to the digital multiplexer (402) to cause a selection of the output current (334) at the output branch (330) of the translinear loop for delivery to the load terminal (120) if the input voltage value fulfills the overvoltage criterion; and42.send a second control signal to the digital multiplexer (402) to cause a selection of the reference current (144) for delivery to the load terminal (120) if the input voltage value does not fulfill the overvoltage criterion.

14. The circuit (100) according to any one of claims 8 to 11,44.wherein the discharge current generator circuit (130, 500) further comprises a current mirror (502) coupled between the output branch (330) of the translinear loop and the load terminal (120),45.wherein the current mirror (502) comprises an input branch configured to receive a replica of the reference current (512) and the output current (334) of the translinear loop, and an output branch coupled with the load terminal (120), and wherein the current mirror (502) is configured to transfer to the output branch a difference between the replica of the reference current (512) and the output current (334) of the translinear loop, if the output current (334) of the translinear loop is less than the replica of the reference current (512).

15. The circuit (100) according to claim 14,47.wherein the reference current (303) at the input branch of the translinear loop is greater than the reference current (144) provided by the reference current source (118) and greater than the replica of the reference current (512) at the input branch of the current mirror (502).

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