Microstrip reflective array antenna and design method thereof

Abstract

The application discloses a microstrip reflective array antenna and a design method thereof, and belongs to the field of antenna technology.The microstrip reflective array antenna comprises a feed source and a planar reflective array antenna which is oppositely arranged with the feed source through a support.The planar reflective array antenna is a single-layer patch structure which is coupled to a back microstrip line through a gap for phase shifting, and is used for realizing a standing wave bandwidth of 15%, satisfying a dual-polarization working characteristic, and stably working at an angle of 45 degrees and having an aperture efficiency of not less than 30%.The microstrip reflective array antenna and the design method thereof are used, and through design based on simulation analysis, the obtained sample has excellent aperture efficiency.

Description

Technical Field

[0001] This invention relates to the field of antenna technology, and in particular to a microstrip reflective array antenna and its design method. Background Technology

[0002] Microstrip reflector antennas consist of a feed antenna and a metasurface. They integrate the advantages of reflector antennas and array antennas and are widely used in satellite communication systems and radar systems. Their operating frequency band has been extended from the microwave band to the terahertz band. At the same time, due to the powerful beam control characteristics of reflector antennas, antenna engineers have developed reflector antennas with different functions, such as dual-frequency high-gain antennas, dual-polarized reflector antennas, circularly polarized reflector antennas, dual-linearly polarized tunable reflector antennas, X-band ultra-wideband antennas, beam scanning antennas, and multi-beam high-gain antennas.

[0003] In the design of microstrip reflective array antennas, there is a lack of research on the broadband, slant angle stability and dual-polarization operation of metasurface elements, as well as the impact of feed on the gain and aperture efficiency of planar reflective array antennas. Summary of the Invention

[0004] To address the aforementioned problems, this invention provides a microstrip reflective array antenna and its design method. The design, based on simulation analysis, yields a sample with excellent aperture efficiency.

[0005] To achieve the above objectives, the present invention provides a microstrip reflector antenna, including a feed and a planar reflector antenna arranged opposite to the feed via a support. The planar reflector antenna is a unit structure in which a single-layer patch is coupled to a back microstrip line through a slot for phase shifting, achieving a 15% standing wave bandwidth, meeting the operating characteristics of dual polarization, being able to operate stably at a 45° tilt angle, and having an aperture efficiency of not less than 30%.

[0006] Preferably, the unit structure includes, from top to bottom, a first dielectric layer, a first radiating patch layer, a second dielectric layer, a first slot coupling layer, a third dielectric layer, a stripline feeder layer, a fourth dielectric layer, a ground plane layer, a fifth dielectric layer, and a microstrip phase-shifting layer, which are bonded together by a PP layer.

[0007] The first dielectric layer is used to enable the feed to better match the unit structure under oblique angle incidence, expand the bandwidth, and protect the first radiating patch layer.

[0008] The first radiating patch layer is used to receive electromagnetic waves emitted by the feed source and to radiate electromagnetic waves outward after phase shifting.

[0009] The second dielectric layer is used to extend the bandwidth of the gap coupling;

[0010] The third dielectric layer is used for coupling bilinear polarization with the stripline;

[0011] The fourth dielectric layer is used to isolate the stripline feed layer and the ground plane layer;

[0012] The fifth dielectric layer is used to achieve phase shifting by loading microstrip lines of different lengths.

[0013] Preferably, the first dielectric layer is made of Tacnoic RF-35A2 with a thickness of h1, a dielectric constant of 3.5, and a loss tangent of 0.0018;

[0014] A single-layer square patch with a side length of l is attached to the first radiating patch layer with a side length of p;

[0015] The second dielectric layer is made of F4B-M220 with a thickness of h2, a dielectric constant of 2.2, and a loss tangent of 0.0007.

[0016] Two orthogonally arranged first I-shaped slots are formed on the first slot coupling layer for coupling the dual-polarization and the stripline feed layer. The length of the middle section of the first I-shaped slot is g_l, and the length and width of the rectangles at both ends of the first I-shaped slot are b_l and b_w, respectively.

[0017] The third dielectric layer uses Tacnoic RF-35A2 with a thickness of h3;

[0018] Two striplines are orthogonally arranged on the stripline feeder layer, with a length of c_l and a width of c_w.

[0019] The fourth dielectric layer uses F4B-M220 with a thickness of h4;

[0020] The thickness of the fifth dielectric layer is h5;

[0021] Two orthogonally arranged microstrip delay lines are disposed on the microstrip phase shifting layer. The length of the microstrip delay lines is m_l and the width is m_w.

[0022] The PP layer uses Rogers 4450 prepreg with a dielectric constant close to that of the first radiative patch layer and the microstrip phase-shifting layer, and its dielectric constant is 3.52.

[0023] Preferably, a first metallized via is provided on the first slot coupling layer, the third dielectric layer, the stripline feed layer, the fourth dielectric layer, the ground layer, the fifth dielectric layer, and the microstrip phase shifting layer to reduce the coupling between array elements;

[0024] Multiple second metallized vias are arranged in a matrix around the positions of the first metallized vias on the stripline feed layer and around the positions of the first metallized vias on the microstrip phase-shifting layer, for conducting and suppressing the generation of cavity modes.

[0025] Preferably, the transition structure consisting of a fourth dielectric layer, a ground layer, and a fifth dielectric layer is used to separate the stripline feeder layer and the microstrip phase-shifting layer;

[0026] Within the 9–18 GHz range, the modulus of the transmission coefficient of the transition structure is not less than -0.5 dB, while the modulus of the reflection coefficient of the transition structure is less than -15 dB. Within the frequency range of 12.5–14.5 GHz in the operating frequency band, the modulus of the transmission coefficient is greater than -0.3 dB, and the modulus of the reflection coefficient is less than -15 dB.

[0027] Preferably, the feed source is a microstrip quad array antenna with an operating frequency band of 12.5-14.5GHz. The microstrip quad array antenna includes a second radiating patch layer, a sixth dielectric layer, a seventh dielectric layer, a second slot coupling layer, an eighth dielectric layer and a feed layer arranged sequentially from top to bottom.

[0028] Multiple side-length p-type radiating patches are attached to a second radiating patch layer with side length a in a rectangular array. l A single-layer square patch;

[0029] The second slot coupling layer has an array of second I-shaped slots to increase the coupling between the second radiating patch layer and the feed layer, thereby increasing the bandwidth.

[0030] The length of the middle section of the second I-shaped gap is g. l Width is g w The length of the rectangles at both ends of the second I-shaped gap is b. l Width is b w ;

[0031] The feed layer uses a T-shaped power divider, which includes an input section and an output section connected to the second slot coupling layer and the SMA connector, respectively, as well as a λ / 4 impedance transformation section connected between the input and output sections. The microstrip impedance of the input and output sections is 50Ω, and the microstrip line width is 1.7mm. The microstrip line width of the λ / 4 impedance transformation section is 0.9mm.

[0032] The sixth dielectric layer has a thickness of h. p The F4B-M220;

[0033] The eighth dielectric layer has a thickness of h. m The Tacnoic RF-35A2;

[0034] The seventh dielectric layer has a thickness of h. air The air layer or PMI foam layer.

[0035] Preferably, the feed source is a horn antenna, which includes a rectangular waveguide and a circular gradient connected in sequence. A 50Ω SMA connector is connected to the rectangular waveguide, and a metal disk with an open circuit termination is provided at the top of the probe of the 50Ω SMA connector.

[0036] The overall length of the horn antenna is 137mm;

[0037] The rectangular waveguide has a length and a width of 22.1 mm and 13.2 mm, respectively.

[0038] The maximum diameter of the circular gradient is 35mm;

[0039] The 50Ω SMA connector has inner and outer radii of 1.27mm and 4.1mm, respectively, and is filled with polytetrafluoroethylene between itself and the metal disc.

[0040] A design method for a microstrip reflector array antenna includes the following steps:

[0041] S1. Design the unit structure to achieve a 15% standing wave bandwidth, meet the dual-polarization operating characteristics, and be able to work stably at a 45° tilt angle and have an aperture efficiency of not less than 30%.

[0042] S2. Design the feed source, selecting a microstrip quad array antenna or a horn antenna as the feed source, with a size of 120×120mm. 2 With a focal diameter ratio of 0.8, its 3dB beamwidth is between 40 and 50° within the operating frequency band;

[0043] S3. Estimating antenna aperture efficiency: Based on the unit structure designed in step S1 and the feed pattern designed in step S2, make a preliminary estimate of the overall planar reflective array antenna aperture efficiency.

[0044] S4. Design the array configuration;

[0045] S5. First, simulate the array in CST MICROWAVE STUDIO simulation software, analyze the results and optimize, then process the experimental sample and test it in a microwave anechoic chamber to verify the feasibility of the design.

[0046] Preferably, step S4 specifically includes the following steps:

[0047] S41. Using 3-bit phase quantization, the phase from 0 to 360° is discretized into 8 phase intervals. These intervals are then connected using 8 microstrip delay lines of different lengths to form a reflection unit. Each unit represents a phase interval.

[0048] The phase range corresponding to the 0° unit is 315-360°; the phase range corresponding to the 45° unit is 270-315°; the phase range corresponding to the 90° unit is 215-270°; the phase range corresponding to the 135° unit is 180-215°; the phase range corresponding to the 180° unit is 135-180°; the phase range corresponding to the 215° unit is 90-135°; the phase range corresponding to the 270° unit is 45-90°; and the phase range corresponding to the 315° unit is 0-45°.

[0049] S42. Design the array configuration based on the compensation phase required for the beam deflection angle of the array antenna:

[0050] Design an array with 12×12 elements, i.e., the array face size is 120×120mm. 2 The focal diameter ratio is selected as 0.8, and the feed source is selected as a spatial side feed. Taking the lower left corner of the array as the origin, the coordinates of the feed source phase center are selected as (20, 60, 84). Then, according to the phase compensation formula, the phase distribution of the beam exit surface at 0° is calculated, and the form of the planar reflection array is designed according to the phase distribution.

[0051] Preferably, step S5 specifically includes the following steps:

[0052] S51. Spatial side-feed is performed using a microstrip quad array antenna and a horn antenna, respectively. The feed source is obliquely incident at different angles with the antenna array surface and then simulated.

[0053] S52. Use CST's time-domain solver to solve the problem and obtain the simulation results;

[0054] S53. The horn antenna is processed using 3D printing technology and integrated with the planar reflector array and bracket for installation and fixation; and the microstrip quad array antenna is processed using PCB technology and integrated with the planar reflector array and bracket for installation and fixation.

[0055] S54. Experimentally test the radiation patterns of different feed plane reflector arrays at frequencies of 12.5 GHz and 14.5 GHz, and obtain the test results;

[0056] S55. Compare the test results with the simulation results to verify the feasibility.

[0057] The present invention has the following beneficial effects:

[0058] 1. A unit structure with broadband, slant angle stability and dual polarization operation was proposed. The effects of different types of feed sources on the gain and aperture efficiency of the planar reflector array antenna were analyzed and compared. Finally, the functionality and rationality of the design were verified by experiments. The fabricated sample has excellent aperture efficiency.

[0059] 2. Utilizing the concept of phase quantization, the phase from 0° to 360° was quantized into eight phase states, and eight corresponding reflective elements were designed. Using the principle of phase compensation, the eight reflective elements were arranged into a planar reflective array with different beam deflection capabilities. Horn feed sources and microstrip four-element array feed sources were fabricated using 3D printing technology and PCB manufacturing processes. The differences in the feeding effects of different feed sources on the planar reflective array were compared. Finally, the working capability of the planar reflective array under beam pointing at 0°, 15°, 30°, and 45° was designed using four-element array feeding.

[0060] The technical solution of the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. Attached Figure Description

[0061] Figure 1 This is a flowchart of a microstrip reflector array antenna design method according to the present invention;

[0062] Figure 2 This is a schematic diagram of the structure of a planar reflective array antenna according to the present invention;

[0063] Figure 3 This is a schematic diagram of the transition structure of a planar reflective array antenna for a microstrip reflective array antenna according to the present invention.

[0064] Figure 4 The simulation results of the transmission and reflection coefficients of the transition structure between the stripline and the microstrip phase shifter in an embodiment of the present invention are shown in the figure.

[0065] Figure 5 This is a distribution diagram of the reflection coefficient of the feed port under different modes of an embodiment of the present invention;

[0066] Figure 6 This is a distribution diagram of the magnitude of the coupling coefficient at the feed port under different modes according to an embodiment of the present invention;

[0067] Figure 7 This is a distribution diagram of the power supply port transmission coefficient under different modes according to an embodiment of the present invention;

[0068] Figure 8 This is a distribution diagram of the polarization coupling coefficient magnitude under different modes in embodiments of the present invention;

[0069] Figure 9 This is a schematic diagram of a microstrip quad array antenna structure for a planar reflective array antenna of the present invention.

[0070] Figure 10 The reflection coefficient and radiation pattern at different frequencies of the microstrip four-element array antenna according to an embodiment of the present invention are simulated.

[0071] Figure 11This is a schematic diagram of the structure of a horn antenna of a microstrip reflector array antenna according to the present invention;

[0072] Figure 12 The reflection coefficient and radiation pattern at different frequency points of the horn antenna simulation for an embodiment of the present invention;

[0073] Figure 13 The figure shows the test results of the microstrip quad array according to an embodiment of the present invention;

[0074] Figure 14 This is a diagram showing the test results of a horn antenna according to an embodiment of the present invention;

[0075] Figure 15 This is a diagram showing the states of eight reflective units according to an embodiment of the present invention;

[0076] Figure 16 The diagram shows the S-parameters and phase difference results for different unit states in an embodiment of the present invention.

[0077] Figure 17 This is a phase distribution diagram of the 0° exit array surface according to an embodiment of the present invention;

[0078] Figure 18 This is a diagram showing the array element arrangement of an embodiment of the present invention;

[0079] Figure 19 This is a comparison of the gains of different feed-source planar reflector arrays according to embodiments of the present invention, and a graph showing the relationship between gain and frequency.

[0080] Figure 20 The diagram shows the phase distribution and array arrangement of the 15°, 30° and 45° exit arrays according to an embodiment of the present invention;

[0081] Figure 21 These are deflection patterns of a planar reflection array at different frequencies and angles according to an embodiment of the present invention.

[0082] Figure 22 This is a schematic diagram of sample processing according to an embodiment of the present invention;

[0083] Figure 23 This is a schematic diagram of the four-element array feed support structure according to an embodiment of the present invention;

[0084] Figure 24 This is a schematic diagram of the speaker feed bracket structure according to an embodiment of the present invention;

[0085] Figure 25 This is a comparison diagram of experimental and simulation radiation patterns of different feed sources at different frequencies according to embodiments of the present invention. Detailed Implementation

[0086] The present invention will be further described below with reference to the accompanying drawings. It should be noted that this embodiment is based on the present technical solution and provides detailed implementation methods and specific operation processes, but the protection scope of the present invention is not limited to this embodiment.

[0087] For linearly polarized broadband planar reflector arrays, common forms include square single-ring and double-ring structures, square patches, and multi-layer patch structures. This embodiment proposes a unit structure in which a single-layer patch is coupled to a back-side microstrip line via a slot for phase shifting. This unit structure is simple, and the modulation phase and the receiving patch are located on different layers, reducing the possibility of coupling. Most importantly, the reflection phase can be easily achieved within 360° by designing the required microstrip delay line length.

[0088] A microstrip reflector antenna includes a feed and a planar reflector antenna arranged opposite to the feed via a support. The planar reflector antenna is a unit structure in which a single-layer patch is coupled to a back microstrip line through a slot to achieve a 15% standing wave bandwidth, meet the operating characteristics of dual polarization, can operate stably at a 45° tilt angle, and has an aperture efficiency of not less than 30% for phase shifting.

[0089] Specifically, the unit structure includes, from top to bottom, a first dielectric layer bonded together with a PP layer, a first radiating patch layer, a second dielectric layer, a first slot coupling layer, a third dielectric layer, a stripline feed layer, a fourth dielectric layer, a ground plane layer, a fifth dielectric layer, and a microstrip phase-shifting layer. The first dielectric layer is used to better match the feed source with the unit structure under oblique angle incidence, expand the bandwidth, and protect the first radiating patch layer. The first radiating patch layer is used to receive electromagnetic waves emitted by the feed source and to radiate electromagnetic waves outward after phase shifting. The second dielectric layer is used to expand the bandwidth of the slot coupling. The third dielectric layer is used for bilinear polarization coupling with the stripline. The fourth dielectric layer is used to isolate the stripline feed layer and the ground plane layer. The fifth dielectric layer is used to achieve phase shifting by loading microstrip lines of different lengths.

[0090] Preferably, the first dielectric layer is made of Tacnoic RF-35A2 with a thickness of h1, a dielectric constant of 3.5, and a loss tangent of 0.0018; a single-layer square patch with a side length of l is attached to the first radiating patch layer with a side length of p; the second dielectric layer is made of F4B-M220 with a thickness of h2, a dielectric constant of 2.2, and a loss tangent of 0.0007; two orthogonally arranged first I-shaped slots are formed on the first slot coupling layer for coupling the dual-polarization to the stripline feed layer, the middle section of the first I-shaped slot has a length of g_l, and the length and width of the rectangles at both ends of the first I-shaped slot are b_l and b_w respectively; the third dielectric layer is made of Tacnoic with a thickness of h3. RF-35A2; Two orthogonally arranged striplines are set on the stripline feed layer, with a length of c_l and a width of c_w; The fourth dielectric layer is F4B-M220 with a thickness of h4; The fifth dielectric layer has a thickness of h5; Two orthogonally arranged microstrip delay lines are set on the microstrip phase-shifting layer, with a length of m_l and a width of m_w; The PP layer is Rogers 4450 prepreg with a dielectric constant close to that of the first radiating patch layer and the microstrip phase-shifting layer, and its dielectric constant is 3.52.

[0091] In this embodiment, a slot-radiatively coupled metasurface unit operating in the Ku band is designed based on multilayer PCB technology. This unit will be used for broadband planar reflective array antennas and planar folded reflective array antennas.

[0092] Table 1 shows the structural dimensions of the Ku-band array unit (unit: mm).

[0093]

[0094]

[0095] Preferably, a first metallized via is provided on the first slot coupling layer, the third dielectric layer, the stripline feed layer, the fourth dielectric layer, the ground layer, the fifth dielectric layer, and the microstrip phase shifting layer to reduce the coupling between array elements;

[0096] Multiple second metallized vias arranged in a matrix are formed around the positions of the first metallized vias on both the stripline feed layer and the microstrip phase-shifting layer. These vias are used for conduction and to suppress cavity mode generation. Since the four isolating second metallized vias penetrate the entire layer, the PCB board only needs to be laminated once, reducing manufacturing difficulty. However, the additional via length is equivalent to a parallel inductive load, directly affecting the performance of the transition structure. This impact can be mitigated through simulation optimization. Furthermore, to make the simulation more closely resemble manufacturing testing, pads were added at the corresponding positions of the vias for simulation.

[0097] Preferably, the transition structure consisting of a fourth dielectric layer, a ground layer, and a fifth dielectric layer is used to separate the stripline feeder layer and the microstrip phase-shifting layer;

[0098] Within the 9–18 GHz range, the modulus of the transmission coefficient of the transition structure is not less than -0.5 dB, while the modulus of the reflection coefficient of the transition structure is less than -15 dB. Within the frequency range of 12.5–14.5 GHz in the operating frequency band, the modulus of the transmission coefficient is greater than -0.3 dB, and the modulus of the reflection coefficient is less than -15 dB.

[0099] Transition structures can be used to ensure the effective transmission of electromagnetic waves between striplines and microstrip phase shifters. Therefore, performance simulation of the transition structure is necessary before designing the unit structure.

[0100] Preferably, since the resonance of a microstrip antenna is similar to that of a high-Q resonant circuit, the problem is that the bandwidth is narrow, generally only reaching about 5-10% of the relative bandwidth. For the operating frequency band of 12.5-14.5GHz, it is difficult to achieve the bandwidth of ordinary microstrip antennas. Therefore, the feed source in this embodiment is a microstrip four-element array antenna with an operating frequency band of 12.5-14.5GHz. The microstrip four-element array antenna includes, from top to bottom, a second radiating patch layer, a sixth dielectric layer, a seventh dielectric layer, a second slot coupling layer, an eighth dielectric layer, and a feed layer. Multiple antennas with side length p are attached to the second radiating patch layer with side length a in a rectangular array. l The first layer consists of a single-layer square patch; the second slot coupling layer has arrayed second I-shaped slots to increase the coupling between the second radiating patch layer and the feed layer, thereby increasing the bandwidth; the middle section of the second I-shaped slot has a length of g. l Width is g w The length of the rectangles at both ends of the second I-shaped gap is b. l Width is b w The feed layer employs a T-shaped power divider, which includes an input section and an output section connected to the second slot coupling layer and the SMA connector, respectively, as well as a λ / 4 impedance transformation section connecting the input and output sections. The microstrip impedance of both the input and output sections is 50Ω, and their microstrip line width is 1.7mm. The microstrip line width of the λ / 4 impedance transformation section is 0.9mm. The sixth dielectric layer uses a thickness of h. p F4B-M220; the eighth dielectric layer uses a thickness of h m The Tacnoic RF-35A2; the seventh dielectric layer has a thickness of h. air The air layer or PMI foam layer. In this embodiment, seven through holes of the same size are opened on each layer, with a diameter of 4mm. The two layers of dielectric plates and the PMI foam layer are fixed with nylon screws.

[0101] Table 2 shows the structural parameters of the microstrip quad array antenna. Unit: mm

[0102] a <![CDATA[p l ]]> <![CDATA[g l ]]> 30 6.5 3.2 <![CDATA[g w ]]> <![CDATA[b l ]]> <![CDATA[b w ]]> 0.9 1.8 0.7 <![CDATA[h p ]]> <![CDATA[h m ]]> <![CDATA[h air ]]> 0.5 0.8 2

[0103] Simulations were performed on the designed microstrip quaternary array. To make the simulation results closer to the actual sample test results, a 50Ω SMA connector was also designed at the microstrip feed port during the simulation. Figure 10 (a) shows the reflection coefficient and standing wave ratio of the four-element array. Within the operating frequency band, the reflection coefficient at its port is less than -15dB, and the standing wave ratio is within 1.3, indicating that surface electromagnetic waves can be effectively radiated within the operating frequency band. Other simulation results to consider include the shape of its radiation pattern and the 3dB beamwidth within the operating frequency band. Five operating frequencies—12.5GHz, 13GHz, 13.5GHz, 14GHz, and 14.5GHz—were selected for analysis. Figure 10 As shown in (b, c, d, e, f), the 3dB beamwidth of the E-plane and H-plane at these 5 frequency points is basically between 45° and 55°. The radiation patterns of the E-plane and H-plane have good symmetry and the radiation directions are consistent. Moreover, the sidelobe levels are all less than -20dB.

[0104] Preferably, the feed source is a horn antenna, which includes a rectangular waveguide and a circular gradient connected in sequence. A 50Ω SMA connector is connected to the rectangular waveguide, and a metal disk with an open-circuit termination is provided at the probe top of the 50Ω SMA connector. The overall length of the horn antenna is 137mm. The length and width of the rectangular waveguide are 22.1mm and 13.2mm, respectively. The maximum aperture of the circular gradient is 35mm. The inner and outer radii of the 50Ω SMA connector are 1.27mm and 4.1mm, respectively, and polytetrafluoroethylene is filled between it and the metal disk.

[0105] The simulation results of the horn antenna are as follows: Figure 12 As shown, its reflection coefficient is less than -15dB in the operating frequency band, and its gain is 12.1dBi at the lowest frequency of 12.5GHz. The 3dB beamwidth in both the E-plane and H-plane is 47°. At the highest frequency of 14.5GHz, its gain is 13.5dBi, and the 3dB beamwidths in the E-plane and H-plane are 38° and 40°, respectively. The radiation patterns in the E-plane and H-plane at different frequencies exhibit perfect symmetry. Since a reasonable beamwidth can enable the planar reflector array to achieve high aperture efficiency, the horn antenna in this embodiment takes into account the size of the 3dB beamwidth in its design. Therefore, the gain of the horn at different frequencies is not required to be too high; it is sufficient to meet the requirements of good radiation characteristics and directivity within the 3dB beamwidth. Therefore, this horn antenna meets the operating requirements of the feed of the planar reflector array in the range of 12.5 to 14.5GHz.

[0106] Comparing the two types of feed sources, the simulation results show that the horn antenna has a stable standing wave and perfect E-plane and H-plane radiation pattern symmetry. In addition, the horn antenna does not have a strong back lobe and the energy radiation direction is concentrated. The advantage of the microstrip quad array is that it has a lower profile, which can significantly reduce the overall height of the planar reflective array antenna.

[0107] This embodiment further presents the experimental test results and comparisons of two different feed sources:

[0108] Meanwhile, a microstrip quad array and a horn antenna were fabricated.

[0109] The test results for the microstrip quad array are as follows: Figure 13 As shown, the experimentally measured reflection coefficient S11 differs significantly from the simulation. Although all three resonant points are retained, the resonant frequencies shift towards higher frequencies and the resonant intervals increase. This phenomenon is attributed to the use of PMI foam with a dielectric constant of 1.15 instead of the air layer and the larger nylon rivets, both of which have some impact. However, within the operating frequency band, S11 remains below -10dB. Regarding its radiation pattern, its 3dB beamwidth is between 40-50°, meeting the requirements of a planar reflection array.

[0110] The horn antenna was fabricated using 3D printing technology. The printed part forms the main body of the horn antenna, and the printed metal material is aluminum alloy. The SMA connector for the feed port was purchased directly, and the printed main body was then welded to the SMA connector. A sample of the fabricated antenna is shown below. Figure 14 As shown, its reflection coefficient and E-plane radiation pattern were also tested. It can be seen that the experimental and simulated reflection coefficient curves are basically consistent, but there is a slight difference at 13.5 GHz. This is because the 3D-printed metal surface has some unevenness, causing inconsistencies in the resonant points. However, in the operating frequency band of the planar reflector array, the reflection coefficient of this horn is less than -15 dB, exhibiting good radiation characteristics. The antenna radiation patterns at 12.5 GHz and 14.5 GHz were measured in a microwave anechoic chamber. Within the operating frequency band, its 3 dB beamwidth is between 40 and 50°, meeting the requirements of a planar reflector array feed.

[0111] A design method for a microstrip reflector array antenna includes the following steps:

[0112] S1. Design the unit structure to achieve a 15% standing wave bandwidth, meet the dual-polarization operating characteristics, and be able to work stably at a 45° tilt angle and have an aperture efficiency of not less than 30%.

[0113] S2. Design the feed source, selecting a microstrip quad array antenna or a horn antenna as the feed source, with a size of 120×120mm. 2With a focal diameter ratio of 0.8, its 3dB beamwidth is between 40 and 50° within the operating frequency band;

[0114] S3. Estimating antenna aperture efficiency: Based on the unit structure designed in step S1 and the feed pattern designed in step S2, make a preliminary estimate of the overall planar reflective array antenna aperture efficiency.

[0115] In this embodiment, the process of calculating the aperture efficiency needs to consider multiple factors of the antenna, including the antenna radiation mode, antenna aperture, beamwidth and gain, etc.: (1) First, we need to calculate the theoretical gain G of the antenna. t :G t =4πA e / λ, where A e λ is the effective aperture area of ​​the antenna, and λ is the wavelength of the signal; (2) Next, obtain the actual gain G of the antenna. The methods to obtain this gain include: actual measurement or finding it in the antenna specifications provided by the antenna manufacturer; (3) Calculate the aperture efficiency:

[0116] In this embodiment, for the feed source, if its radiation pattern is narrow, the gain received by the array elements at the aperture edge will be much smaller than that received by the array elements at the aperture center, resulting in a decrease in aperture efficiency. Aperture efficiency can be improved by increasing the appropriate focal-to-aperture ratio. Similarly, for the array element radiation pattern, if its pattern is too narrow, the gain received by the edge array elements will also be less than that received by the center array elements. Therefore, the orientation of the array elements... Figure 1 It is generally designed in a wider form.

[0117] S4. Design the array configuration;

[0118] Preferably, step S4 specifically includes the following steps:

[0119] S41. Larger arrays result in higher gain and stronger directivity, while smaller arrays exhibit significant cross-polarization and larger sidelobes. Therefore, an appropriate array size should be designed according to requirements. In this embodiment, 3-bit phase quantization is used to discretize the phase from 0 to 360° into eight phase intervals, which are then connected using eight microstrip delay lines of different lengths to form a reflection unit. Each unit represents a phase interval.

[0120] like Figure 15As shown, the phase range corresponding to the 0° unit is 315-360°; the phase range corresponding to the 45° unit is 270-315°; the phase range corresponding to the 90° unit is 215-270°; the phase range corresponding to the 135° unit is 180-215°; the phase range corresponding to the 180° unit is 135-180°; the phase range corresponding to the 215° unit is 90-135°; the phase range corresponding to the 270° unit is 45-90°; and the phase range corresponding to the 315° unit is 0-45°.

[0121] Table 3 shows the dimensions of eight types of reflective units (unit: mm).

[0122] Cell state <![CDATA[l1]]> <![CDATA[l2]]> <![CDATA[l3]]> <![CDATA[l4]]> <![CDATA[l5]]> <![CDATA[l6]]> <![CDATA[l7]]> 0° 1.7 1.1 4.9 2 2.3 6.4 3.8 45° 1.5 1.5 4.4 2 2.1 6.3 3.6 90° 6.5 4.4 3.2 \ \ \ \ 135° 5.8 4.4 2.5 \ \ \ \ 180° 1.7 2.6 6.8 \ \ \ \ 225° 3 1 1.3 4.9 1.5 \ \ 270° 3 4.8 \ \ \ \ \ 315° 1.7 6 6.4 3 \ \ \

[0123] All element simulations are based on the periodic boundary conditions of the CST simulation software. Since the planar reflective array is composed of eight types of reflective elements, the coupling of other elements must be considered during element design. According to Floquet theory, periodic boundary conditions can simulate an infinitely large array environment. Therefore, the coupling of other elements can be considered in the element simulation. By designing the excitation ports and boundary conditions, the S-parameters of each element and the phase difference between elements can be obtained through simulation. It is important to note that the time harmonic factor in the CST simulation software is e. -iωt Therefore, the 0° element is actually loaded with a longer microstrip delay line. The Floquet mode was selected as the excitation, and simulations were performed on all eight elements.

[0124] Simulation results are as follows Figure 16 As shown, the cross-polarization conversion rates of the eight states are all within -0.7dB in the operating frequency band, and the common polarization conversion rates are all less than -10dB, demonstrating good common polarization isolation performance. The phase difference is based on the reflection phase of the 315° reflector unit, and designed with the highest bandwidth frequency of 14.5GHz. It can be seen that at 14.5GHz, the phase differences of the eight reflector units are 0°, 45°, 90°, 135°, 180°, 225°, 270°, and 315°, respectively, meeting the array design requirements. It can be seen that the 0° and 45° reflector units have more resonant points and higher insertion losses than other reflector units. This is because the microstrip phase shift of the 0° and 45° reflector units is more complex than other reflector units. Due to the space constraints on the back side of the unit, coupling between microstrip lines and some reflection at microstrip corners are inevitable during routing. Multiple reflections form a Fabry-Perot resonance-like effect, thus resulting in more resonant points. This insertion loss has been minimized by optimizing the microstrip line routing.

[0125] S42. Design the array configuration based on the compensation phase required for the beam deflection angle of the array antenna:

[0126] like Figure 17 As shown, an array of 12×12 elements is designed, meaning the array surface size is 120×120mm. 2 With a focal diameter ratio of 0.8 and a spatial side-feed feed selected, and the lower left corner of the array as the origin, the coordinates of the feed phase center are chosen as (20, 60, 84). Then, according to the phase compensation formula (the reflection phase of a planar reflective array element should compensate for the spatial phase delay φ from the feed phase center to the element),... pd =-k0R i , where R i Let k be the distance from the feed phase center to the i-th element, and k0 be the wave number in vacuum. Calculate the phase distribution of the beam exiting the array at 0°. Then, design the planar reflector array based on the phase distribution. (A key step in designing a planar reflector array antenna is selecting a suitable phase tuning method (loading a phase delay line) so that the planar reflector elements of the planar reflector array can achieve the required phase tuning range. Once the phase tuning method is selected, the element characteristics can be determined, and finally, the antenna's radiation characteristics can be achieved by arranging the element characteristics.)

[0127] S5. First, simulate the array in CST MICROWAVE STUDIO simulation software, analyze the results and optimize, then process the experimental sample and test it in a microwave anechoic chamber to verify the feasibility of the design.

[0128] In this embodiment, simulation is performed using CST MICROWAVE STUDIO software. Periodic boundary conditions are applied to the element structure to simulate an infinitely large periodic structure. For dual-polarized elements, the reflection coefficient of the port under different incident modes is the primary focus, as the port reflection coefficient reflects the quality of the matching between the element and the feed source. For example... Figure 5 As shown, under normal incidence in both TE and TM modes, the port reflection coefficient is less than -15dB in the 12–15GHz range. With increasing incidence angle, the reflection coefficient of the TE and TM mode ports gradually increases. When the incidence angle increases to 45°, the port reflection coefficient increases significantly, but remains less than -10dB in the 12–15GHz range, meeting the requirements for units operating in the 12.5–14.5GHz range.

[0129] Secondly, the coupling coefficient between feed ports of different modes is also an important indicator. A low coupling coefficient ensures that the cell will not receive electromagnetic energy with reverse polarization under single-polarization incident radiation. For example... Figure 6 As shown, under TE mode with normal incidence, the coupling coefficient of the port is less than -25dB in the 12–15GHz range. For TM mode, as the incident angle increases, the coupling coefficient of the TM mode port gradually increases to -22dB, which meets the requirements for units operating in the 12.5–14.5GHz range.

[0130] The most important aspect of cell design is the transmission coefficient from the feed to the cell, as its value reflects the strength of the feed energy received by the cell. In TE and TM feeding modes, the feed energy is received via a patch, coupled through a slot to the stripline, and finally transmitted to the microstrip phase shifter layer behind it via a via. A matching port is attached to the end of the microstrip line, such as... Figure 7 As shown, the received energy is better than -1dB in the 12–15GHz range, and the transmission coefficient remains stable even when the incident angle increases to 45°. For the array's operating frequency band of 12.5–14.5GHz, the transmission coefficient is less than -0.7dB, and this unit meets the array design requirements.

[0131] Finally, the coupling coefficients between the microstrip line receiving polarizations under different incident modes are as follows: Figure 8 As shown, simulation results indicate that under normal incidence, the polarization coupling coefficient is relatively large at high frequencies, but less than -25dB within the unit's operating frequency band. With increasing incidence angle, the polarization coupling coefficient at the receiver port gradually decreases under both incident modes, demonstrating good polarization isolation performance.

[0132] Preferably, step S5 specifically includes the following steps:

[0133] S51. Spatial side-feed is performed using a microstrip quad array antenna and a horn antenna, respectively. The feed source is obliquely incident at different angles with the antenna array surface and then simulated.

[0134] After the array arrangement was completed, spatial side feeding was performed using a horn antenna and a microstrip four-element array, with the feed source incident at a 25° angle to the antenna array surface. During simulation, the far-field radiation patterns at 12.5 GHz, 13.5 GHz, and 14.5 GHz were observed, and their far-field radiation pattern comparisons are as follows: Figure 19As shown, the planar reflector array exhibits strong beam concentration at 0°. At 12.5 GHz, the gains of the horn and quaternary array feeds are 16.7 and 18.1 dBi, respectively; at 13.5 GHz, the gains are 19.6 and 20.6 dBi; and at 14.5 GHz, the gains are 22.2 and 22.0 dBi. The planar reflector array using a horn feed has a significantly lower gain at low frequencies than the one using a quaternary array feed, while the gains of both feeds are roughly the same at high frequencies. However, the sidelobes of the horn-feed planar reflector array are superior to those of the quaternary array feed. Using the formula for antenna aperture efficiency, the aperture efficiencies at these three different frequencies are calculated. The aperture efficiencies using a horn feed at 12.5 GHz, 13.5 GHz, and 14.5 GHz are 14.88%, 24.89%, and 39.26%, respectively. Using a quaternary array as a feed, the aperture efficiencies at 12.5GHz, 13.5GHz, and 14.5GHz were 20.55%, 31.33%, and 37.49%, respectively. The low aperture efficiency at low frequencies is due to the fact that the phase delay of the reflector unit was designed based on the highest frequency of 14.5GHz, without achieving dispersion correction across the entire frequency band. This results in the low-frequency gain being less than the theoretical gain, thus leading to the low aperture efficiency at low frequencies.

[0135] like Figure 20 As shown, in order to verify the operation of the planar reflective array at oblique angles, planar reflective arrays with angles of 15°, 30° and 45° were designed. Since the normal-emission four-element array is better as a feed source than the horn feed source, the oblique-emission planar reflective arrays were simulated using four-element arrays as feed sources.

[0136] S52. Use CST's time-domain solver to solve the problem and obtain the simulation results;

[0137] like Figure 21 As shown, the array is located in the XOY plane, and its radiation pattern is observed entirely in the plane where Phi = 0, i.e., the E-plane radiation pattern. The results show that the beams are emitted in the designed direction. As the beam deflection angle increases, the gain of the planar reflector array decreases. At the lowest operating frequency of 12.5 GHz, the maximum gain of the obliquely emitted beam is 17.2 dBi, and the minimum gain is 15.5 dBi. At the highest operating frequency of 14.5 GHz, the maximum gain of the obliquely emitted beam is 22.6 dBi, and the minimum gain is 21.3 dBi.

[0138] S53. The horn antenna is processed using 3D printing technology and integrated with the planar reflector array and bracket for installation and fixation; and the microstrip quad array antenna is processed using PCB technology and integrated with the planar reflector array and bracket for installation and fixation.

[0139] For the physical testing of the Ku-band planar reflector array, a sample with normal emission was selected for fabrication. Before sending it for fabrication, the mounting connectors and space feed support for the planar reflector array antenna must be designed. As the array element design shows, the entire array consists of 5 PCB layers. During fabrication, prepreg is used to press and fix each layer together. Rogers 4450 prepreg with a dielectric constant of 3.52 and a pressing thickness of 0.1mm is selected. Since the dielectric substrate of the phase-shifting layer is 0.127mm thick, air bubbles may occur when pressing it with the 0.1mm thick prepreg. Therefore, a 0.2mm thick prepreg is used to press it with the phase-shifting layer during sample fabrication to prevent air bubbles. It is also important to note that the prepreg must be included in the simulation design to achieve consistency between simulation and experiment. A schematic diagram of the designed sample is shown below. Figure 22 As shown, the dielectric substrate and ground plane are extended outward by 2.7mm in the top, bottom, and right directions of the planar reflector array, and by 8.7mm in the left direction. Five through holes with a diameter of 4mm and a spacing of 15mm are drilled at the center of the left extension for fixing the planar reflector array to the support. The planar reflector array antenna support is fabricated using 3D printing technology. The material used is white resin, which has a low dielectric constant and sufficient strength to support the weight of the planar reflector array and feed source. It has minimal impact on the beamout of the planar reflector array, avoiding strong reflection and absorption.

[0140] To verify the differences in the effects of the two feed sources on the planar reflector array, a feed support for both a microstrip four-element array and a horn was designed. The fabricated planar reflector array and the feed source were then fixed and assembled together using nylon rivets. The structural form is as follows: Figure 23 and Figure 24 As shown, radiation pattern tests were also conducted simultaneously in the microwave anechoic chamber of Shanghai University. The overall size of the anechoic chamber is 10×6×4mm. 3 The environment was suitable for pattern testing. A standard gain horn antenna was used to receive the beam gain of the planar reflector array, and the pattern of the planar reflector array with different feed sources was tested at frequencies of 12.5 GHz and 14.5 GHz.

[0141] S54. Experimentally test the radiation patterns of different feed plane reflector arrays at frequencies of 12.5 GHz and 14.5 GHz, and obtain the test results;

[0142] S55. Compare the test results with the simulation results to verify the feasibility.

[0143] The test results are as follows Figure 25As shown, the test results are compared with the simulation results. At 12.5 GHz and 14.5 GHz, the beam deflection direction is entirely biased towards the designed direction. The maximum gain of the main lobe is 21.2 dBi, and its sidelobe level is 0 dBi. It can also be seen that the experimentally measured main lobe gain is slightly lower than the simulation result. This is due to certain experimental errors and some absorption by the support structure, resulting in a lower actual gain. The aperture efficiency is calculated to be 16.3% and 38.5% at 12.5 GHz and 14.5 GHz, respectively.

[0144] Therefore, the present invention adopts the above-mentioned microstrip reflective array antenna and its design method, and through design based on simulation analysis, the obtained sample has excellent aperture efficiency.

[0145] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and not to limit them. Although the present invention has been described in detail with reference to preferred embodiments, those skilled in the art should understand that modifications or equivalent substitutions can still be made to the technical solutions of the present invention, and these modifications or equivalent substitutions cannot cause the modified technical solutions to deviate from the spirit and scope of the technical solutions of the present invention.

Claims

1. A microstrip reflector array antenna, comprising a feed and a planar reflector array antenna mounted opposite to the feed via a support, characterized in that: The planar reflector antenna is a unit structure in which a single-layer patch is coupled to the back microstrip line through a slot to achieve a 15% standing wave bandwidth, meet the dual polarization operating characteristics, can operate stably at a 45° tilt angle, and has an aperture efficiency of not less than 30% for phase shifting. The unit structure includes, from top to bottom, a first dielectric layer, a first radiating patch layer, a second dielectric layer, a first slot coupling layer, a third dielectric layer, a stripline feeder layer, a fourth dielectric layer, a ground plane layer, a fifth dielectric layer, and a microstrip phase-shifting layer, all bonded together by a PP layer. The first dielectric layer is used to enable the feed to better match the unit structure under oblique angle incidence, expand the bandwidth, and protect the first radiating patch layer. The first radiating patch layer is used to receive electromagnetic waves emitted by the feed source and to radiate electromagnetic waves outward after phase shifting. The second dielectric layer is used to extend the bandwidth of the gap coupling; The third dielectric layer is used for coupling bilinear polarization with the stripline; The fourth dielectric layer is used to isolate the stripline feed layer and the ground plane layer; The fifth dielectric layer is used to achieve phase shifting by loading microstrip lines of different lengths; A single-layer square patch is attached to the first radiating patch layer; Two orthogonally arranged first I-shaped slots are provided on the first slot coupling layer for coupling the bilinear polarization with the stripline feed layer. Two striplines are orthogonally arranged on the stripline feeder layer; Two orthogonally arranged microstrip delay lines are set on the microstrip phase-shifting layer; First metallized vias are provided on the first slot coupling layer, the third dielectric layer, the stripline feed layer, the fourth dielectric layer, the ground layer, the fifth dielectric layer, and the microstrip phase shifting layer to reduce the coupling between array elements. Multiple second metallized vias are arranged in a matrix around the positions of the first metallized vias on the stripline feed layer and around the positions of the first metallized vias on the microstrip phase-shifting layer, for conducting and suppressing the generation of cavity modes. The transition structure, consisting of a fourth dielectric layer, a ground layer, and a fifth dielectric layer, is used to separate the stripline feeder layer and the microstrip phase-shifting layer. For the switching structure operating in the 9–18 GHz range, the modulus of its transmission coefficient is not less than [value missing]. Meanwhile, the reflection coefficient modulus of the transition structure is less than 100%. Within the operating frequency range of 12.5–14.5 GHz, the magnitude of the transmission coefficient is greater than... The magnitudes of the reflection coefficients are all less than .

2. The microstrip reflective array antenna according to claim 1, characterized in that: The first dielectric layer has a thickness of The Tacnoic RF-35A2 has a dielectric constant of 3.5 and a loss tangent of 0.0018. The second dielectric layer has a thickness of The F4B-M220 has a dielectric constant of 2.2 and a loss tangent of 0.0007. The length of the middle section of the first I-shaped gap is The lengths of the rectangles at both ends of the first I-shaped gap are Width is ; The third dielectric layer has a thickness of The Tacnoic RF-35A2; The length of the strip is Width is ; The fourth dielectric layer has a thickness of The F4B-M220; The thickness of the fifth dielectric layer is ; The length of the microstrip delay line is Width is ; The PP layer uses Rogers 4450 prepreg with a dielectric constant close to that of the first radiative patch layer and the microstrip phase-shifting layer, and its dielectric constant is 3.

52.

3. A microstrip reflective array antenna according to claim 1, characterized in that: The feed source is a microstrip quad array antenna with an operating frequency band of 12.5-14.5 GHz. The microstrip quad array antenna includes, from top to bottom, a second radiating patch layer, a sixth dielectric layer, a seventh dielectric layer, a second slot coupling layer, an eighth dielectric layer, and a feed layer. The side length is The second radiating patch layer has multiple side lengths arranged in a rectangular array. A single-layer square patch; The second slot coupling layer has an array of second I-shaped slots to increase the coupling between the second radiating patch layer and the feed layer, thereby increasing the bandwidth. The length of the middle section of the second I-shaped gap is Width is The lengths of the rectangles at both ends of the second I-shaped gap are Width is ; The feed layer employs a T-shaped power divider, which includes an input section and an output section connected to the second slot coupling layer and the SMA connector, respectively, as well as a connection between the input and output sections. Impedance transformation section; the microstrip impedance of both the input and output sections is 50Ω, and the microstrip linewidth is... ; The microstrip line width of the impedance transformation section is ; The sixth dielectric layer has a thickness of The F4B-M220; The eighth dielectric layer has a thickness of The Tacnoic RF-35A2; The seventh dielectric layer has a thickness of The air layer or PMI foam layer.

4. A microstrip reflective array antenna according to claim 1, characterized in that: The feed source is a horn antenna, which includes a rectangular waveguide and a circular gradient connected in sequence. A 50Ω SMA connector is connected to the rectangular waveguide, and a metal disk with an open circuit is set at the top of the probe of the 50Ω SMA connector. The overall length of the horn antenna is 137 mm; The rectangular waveguide has a length and a width of 22.1 mm and 13.2 mm, respectively; The maximum diameter of the circular gradient is 35 mm; The 50Ω SMA connector has inner and outer radii of 1.27 mm and 4.1 mm, respectively, and is filled with polytetrafluoroethylene between itself and the metal disk.

5. A design method for a microstrip reflector array antenna as described in any one of claims 1-4, characterized in that: Includes the following steps: S1. Design the unit structure to achieve a 15% standing wave bandwidth, meet the dual-polarization operating characteristics, and be able to work stably at a 45° tilt angle and have an aperture efficiency of not less than 30%. S2. Design the feed source, selecting a microstrip four-element array antenna or a horn antenna as the feed source, and its size is... With a focal diameter ratio of 0.8, its 3 dB beamwidth is between 40 and 50° within the operating frequency band; S3. Estimating antenna aperture efficiency: Based on the unit structure designed in step S1 and the feed pattern designed in step S2, make a preliminary estimate of the overall planar reflective array antenna aperture efficiency. S4. Design the array configuration; S5. First, simulate the array in CST MICROWAVE STUDIO simulation software, analyze the results and optimize, then process the experimental sample and test it in a microwave anechoic chamber to verify the feasibility of the design.

6. The design method of a microstrip reflector array antenna according to claim 5, characterized in that: Step S4 Specifically, the following steps are included: S41. Using 3-bit phase quantization, the phase from 0 to 360° is discretized into 8 phase intervals. These intervals are then connected using 8 microstrip delay lines of different lengths to form a reflection unit. Each unit represents a phase interval. The phase range corresponding to the 0° unit is 315-360°; the phase range corresponding to the 45° unit is 270-315°; the phase range corresponding to the 90° unit is 215-270°; the phase range corresponding to the 135° unit is 180-215°; the phase range corresponding to the 180° unit is 135-180°; the phase range corresponding to the 215° unit is 90-135°; the phase range corresponding to the 270° unit is 45-90°; and the phase range corresponding to the 315° unit is 0-45°. S42. Design the array configuration based on the compensation phase required for the beam deflection angle of the array antenna: design An array of array elements, i.e., an array face of size . The focal diameter ratio is selected as 0.8, and the feed source is selected as a spatial side feed. Taking the lower left corner of the array as the origin, the coordinates of the feed source phase center are selected as (20, 60, 84). Then, according to the phase compensation formula, the phase distribution of the beam exit surface at 0° is calculated, and the planar reflection array is designed according to the phase distribution.

7. The design method of a microstrip reflector array antenna according to claim 5, characterized in that: Step S5 specifically includes the following steps: S51. Spatial side-feed is performed using a microstrip quad array antenna and a horn antenna, respectively. The feed source is obliquely incident at different angles with the antenna array surface and then simulated. S52. Use CST's time-domain solver to solve the problem and obtain the simulation results; S53. The horn antenna is processed using 3D printing technology and integrated with the planar reflector array and bracket for installation and fixation; and the microstrip quad array antenna is processed using PCB technology and integrated with the planar reflector array and bracket for installation and fixation. S54. Experimentally test the radiation patterns of different feed plane reflector arrays at frequencies of 12.5 GHz and 14.5 GHz, and obtain the test results; S55. Compare the test results with the simulation results to verify the feasibility.

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