Design method of high-order matching network applied to ultra-wideband amplifier

Abstract

This invention provides a design method for a high-order matching network applied to an ultra-wideband amplifier. It involves deriving equivalent modules for the single-port preamplifier input and output stages; determining the internal structure of the high-order matching network and the transformer coupling coefficient; analyzing and obtaining the mismatch coefficient of the high-order matching network; and transforming the calculation of the load resonator quality factor into solving the constraint conditions based on the constraints between the transformer coupling coefficient, the high-order matching network mismatch coefficient, and the load resonator quality factor. The source resonator quality factor is then obtained based on the mismatch relationship between the load resonator quality factor and the high-order matching network. Finally, the parameter values ​​of the internal components of the high-order matching network are determined, resulting in the final designed high-order matching network. Compared to existing technologies, this invention achieves impedance matching between the source and load over an ultra-wideband range and enables high-gain flatness amplification within the ultra-wideband range.

Description

Technical Field

[0001] This invention belongs to the field of radio frequency integrated circuit design, specifically relating to a design method for a high-order matching network applied to an ultra-wideband amplifier. Background Technology

[0002] In recent years, with the explosive growth of wireless communication services, traditional communication frequency bands have not only shown severe bandwidth congestion, but their data transmission rates also fail to meet the requirements of iteratively updated mobile communication technologies. Shannon's theorem, with its concise formula, demonstrates the positive correlation between data transmission rate and bandwidth, pointing the way for the advancement of next-generation wireless communication technologies. Furthermore, ultra-wideband RF amplifiers can transmit and receive various communication signals with extremely strong bandwidth coverage, finding applications in fields such as active phased array radar, 5G communication, and smart antennas.

[0003] Unlike traditional narrowband amplification, the design of ultra-wideband RF amplifiers needs to consider the gain flatness at different frequencies within the operating frequency domain. The worse the in-band gain flatness, the greater the signal gain fluctuation of the ultra-wideband RF amplifier at different frequencies, which will distort the waveform of the RF signal, leading to signal distortion and seriously affecting the signal quality and accuracy. In the system, if the gain varies at different frequencies, the system will have difficulty processing strong and weak signals from different frequencies simultaneously, which will limit the dynamic range of the system and reduce its performance.

[0004] In the design of ultra-wideband high-gain flatness amplifiers, the design of the matching network is of paramount importance. The bandwidth and in-band gain flatness of the matching network often determine the performance of the module. While conventional high-order matching networks can provide frequency-variable dual poles to extend the bandwidth, they do not fully utilize the ultra-wideband performance of high-order matching networks and cannot provide in-band high-gain flatness conditions, or can only provide rough in-band high-gain flatness conditions. They also cannot accurately quantify in-band gain flatness, which limits their practical application.

[0005] Therefore, there is an urgent need to develop high-order matching networks that can fully utilize ultra-wideband performance and provide accurate in-band high gain flatness conditions, and it is also urgent to master the corresponding high-order matching network design methods. Summary of the Invention

[0006] To address the aforementioned problems in the prior art, this invention provides a design method for a high-order matching network applied to ultra-wideband amplifiers. The technical problem to be solved by this invention is achieved through the following technical solution:

[0007] This invention provides a design method for a high-order matching network applied to an ultra-wideband amplifier, comprising:

[0008] S100 obtains a single-port pre-stage input equivalent module by conjugating the complex impedance connected in series with the pre-stage input module, and obtains a single-port post-stage output equivalent module by paralleling the inductor-adjusted post-stage impedance.

[0009] S200, design the internal structure of the high-order matching network based on the single-port pre-amplifier equivalent module and the single-port post-amplifier equivalent module, and determine the coupling coefficient of the transformer contained in the high-order matching network based on the operating frequency of the ultra-wideband amplifier and the applied process.

[0010] S300, by jointly analyzing the expressions of the low-frequency poles, high-frequency poles, source-end resonant cavity resonant frequency and load resonant cavity resonant frequency of the high-order matching network in the frequency domain, the mismatch coefficient of the high-order matching network is obtained.

[0011] S400, based on the constraints between the coupling coefficient of the transformer, the mismatch coefficient of the high-order matching network and the quality factor of the load resonant cavity, the calculation of the quality factor of the load resonant cavity is transformed into solving the constraints, and the quality factor of the source resonant cavity is obtained based on the mismatch relationship between the quality factor of the load resonant cavity and the high-order matching network.

[0012] S500, determine the parameter values ​​of the internal components of the high-order matching network based on the quality factor of the source resonant cavity and the quality factor of the load resonant cavity, and obtain the final designed high-order matching network.

[0013] This invention provides an ultra-wideband amplifier that includes a high-order matching network, which is obtained by a design method for high-order matching networks applied to ultra-wideband amplifiers.

[0014] This invention provides a design method for a high-order matching network applied to an ultra-wideband amplifier. It involves deriving equivalent modules for the single-port preamplifier input and output stages; determining the internal structure of the high-order matching network and the coupling coefficient of the transformer; analyzing and obtaining the mismatch coefficient of the high-order matching network; and transforming the calculation of the load resonator quality factor into solving the constraint conditions based on the constraints between the transformer coupling coefficient, the mismatch coefficient of the high-order matching network, and the load resonator quality factor. The source resonator quality factor is then obtained based on the mismatch relationship between the load resonator quality factor and the high-order matching network. Finally, the parameter values ​​of the internal components of the high-order matching network are determined, resulting in the final designed high-order matching network. Compared to existing technologies, this invention achieves impedance matching between the source and load over an ultra-wideband range and enables high-gain flatness amplification within the ultra-wideband range.

[0015] The present invention will be further described in detail below with reference to the accompanying drawings and embodiments. Attached Figure Description

[0016] Figure 1 This is a flowchart illustrating the design method of a high-order matching network for an ultra-wideband amplifier provided by the present invention.

[0017] Figure 2 This is a schematic diagram of the structure of a high-order matching network designed by the design method of the high-order matching network for ultra-wideband amplifiers provided by the present invention.

[0018] Figure 3 The MCMCR high-order matching network Z provided in this embodiment of the invention does not consider gain flatness. 21 A schematic diagram of the simulation;

[0019] Figure 4 This is a schematic diagram illustrating the simulation of the impact of different coupling coefficient (k) values ​​on bandwidth provided in an embodiment of the present invention;

[0020] Figure 5 This is a schematic diagram illustrating the simulation of the impact of different mismatch coefficient (α) values ​​on bandwidth provided in an embodiment of the present invention;

[0021] Figure 6 This is a schematic diagram of a simulation of 6GHz-18GHz bandwidth and in-band gain flatness provided in an embodiment of the present invention. Detailed Implementation

[0022] The present invention will be further described in detail below with reference to specific embodiments, but the implementation of the present invention is not limited thereto.

[0023] In the design of ultra-wideband amplifiers, impedance matching networks are crucial for achieving bandwidth extension. Higher-order matching networks, capable of providing two or more frequency-domain poles, can broaden the operating frequency domain and are an important structural element for ultra-wideband amplification. For a fourth-order matching network that provides two poles, the distribution of pole positions in the frequency domain represents the degree of bandwidth extension, and the positive transfer impedance Z at the poles... 21 The difference in amplitude represents the potential relative magnitude of flatness within the ultra-wideband band. Therefore, obtaining the accurate frequency and forward transfer network expression at the double poles of the fourth-order matched network is a prerequisite for realizing ultra-wideband high-gain flatness amplification.

[0024] Combination Figure 1 and Figure 2 This invention provides a design method for a high-order matching network applied to an ultra-wideband amplifier, comprising:

[0025] S100 obtains a single-port pre-stage input equivalent module by conjugating the complex impedance connected in series with the pre-stage input module, and obtains a single-port post-stage output equivalent module by paralleling the inductor-adjusted post-stage impedance.

[0026] refer to Figure 2 The single-port preamplifier equivalent module includes a first resistor R. S and the third capacitor C S The single-port output equivalent module includes a second resistor R. L and the fourth capacitor C L ;

[0027] Among them, the first resistor R S The positive terminal and the third capacitor C S The positive terminal of the first resistor Rs is connected to the positive terminal of the input of the two-port fourth-order matching network, and the negative terminal of the first resistor Rs is connected to the third capacitor C. S The negative terminal is connected to the second resistor R, which is then electrically connected to the negative terminal of the input of the two-port fourth-order matching network. L The positive terminal and the fourth capacitor C L The positive terminal is connected to the positive terminal of the second resistor R, and then electrically connected to the positive terminal of the output of the two-port fourth-order matching network. L The negative terminal and the fourth capacitor C L Connect the negative terminal to the output of the two-port fourth-order matched network, and then electrically connect it to the negative terminal of the output of the two-port fourth-order matched network.

[0028] This invention extracts the parallel complex impedance R of the single-port preamplifier input module. S C S Parallel complex impedance R of the single-port output module L C L To maximize transmission power, R S and C S Conjugate matching has been achieved through conjugate processing. R L and C L It is composed of the first inductor L g The adjusted parallel complex impedance ensures that the mismatch between the input and output parallel complex impedances is within an acceptable range. R L and C L It is the parallel equivalent impedance after adjusting the reactance through the first inductor Lg. The input end completes conjugate matching, and the output end completes load line matching, both in order to increase the power transmission capability.

[0029] To maximize power transfer capability, conjugate matching is performed at the input of the high-order matching network to transmit the signal from the preceding stage without loss; load matching is performed at the output of the high-order matching network to maximize output power. Therefore, the parallel complex impedance R of the single-port preamplifier input module... S C S This is the impedance form required for actual simulation after conjugation and series-parallel conversion; assuming the series input impedance after conjugation is Z. in =R in -j·X in Formulas (1) and (2) can be used to convert the impedance into the corresponding parallel form:

[0030]

[0031]

[0032] Considering the parasitic capacitance C of the transistor in the subsequent active module gs The influence of series output impedance Z out =R out -j·X out The reactance component is usually large. If directly converted into parallel impedance, the real component of the parallel impedance will also increase, thus affecting the quality factor of the load resonant cavity and the bandwidth of the overall high-order matching network. The first inductor Lg is connected in series with the subsequent active module to form a series LC resonant circuit. By adjusting the inductance value of Lg to control the resonant frequency of the series LC, the above-mentioned negative effects can be avoided. The final adjusted series output impedance is Z. out =R out -j·X out +j·ω·L g Similarly, there are transformation formulas (3) and (4): where f is the frequency.

[0033]

[0034]

[0035] S200: The internal structure of the high-order matching network is designed based on the single-port pre-amplifier input equivalent module and the single-port post-amplifier output equivalent module. The coupling coefficient of the transformer contained in the high-order matching network is determined based on the operating frequency of the ultra-wideband amplifier and the applied process.

[0036] refer to Figure 2 The high-order matching network is designed based on the MCMCR (Mismatch-Consistent Magnetically Coupled Resonator) architecture. The high-order matching network includes: a two-port fourth-order matching network, a single-port pre-amplifier input equivalent module, and a single-port power stage output equivalent module. The single-port pre-amplifier input equivalent module is electrically connected to the two-port fourth-order matching network, forming a source resonant cavity. The single-port power stage output equivalent module is also electrically connected to the two-port fourth-order matching network, forming a load resonant cavity. These components are coupled together through a compact on-chip transformer formed by the on-chip inductors, enabling signal transmission. The two-port fourth-order matching network includes: a first capacitor C1, and a transformer primary coil L... P Transformer secondary coil L S The second capacitor C2; the positive terminal of the first capacitor C1 is connected to the primary coil L of the transformer. PThe positive terminal of the capacitor is connected to the positive terminal of the transformer and serves as the positive terminal of the input to the higher-order matching network. The negative terminal of the first capacitor C1 is connected to the secondary coil L of the transformer. S The negative terminal of the first capacitor is connected to the second capacitor C2, which serves as the negative input of the two-port fourth-order matching network; the positive terminal of the second capacitor C2 is connected to the transformer secondary coil L. S The positive terminal of the first capacitor is connected to the second capacitor C2, which serves as the positive terminal of the output of the two-port fourth-order matching network. The negative terminal of the second capacitor C2 is connected to the secondary coil L of the transformer. S The negative end is connected and serves as the negative end of the output of the two-port fourth-order matching network.

[0037] refer to Figure 2 , Figure 2 The inductor Lg is implemented using an on-chip inductor or a bonding wire, and the first capacitor C1 and the second capacitor C2 are implemented using on-chip capacitors, on-chip variable capacitors, or parasitic capacitance between inductors.

[0038] The coupling coefficient k of the transformer is directly determined by the operating frequency of the required ultra-wideband amplifier and the application process. Under the premise that other parameters remain unchanged, the larger the value of k, the wider the bandwidth of the MCMCR and the greater the passive gain. Therefore, under the condition that the first transformer does not experience significant self-resonance in the operating frequency domain, the value of k should be as close as possible to the maximum value of the applied process.

[0039] Figure 4 The simulation diagram shows the impact of different coupling coefficients k on the bandwidth of the higher-order matching network. It can be seen that, with other parameters remaining constant, the larger the value of k, the wider the bandwidth of the higher-order matching network. To allow for margins in other parameters, the value of k is usually chosen to be the maximum value of the process used in the desired frequency band. It is worth noting that the self-resonant frequency of the transformer must be carefully maintained, especially the fluctuation of the primary and secondary coil inductance values ​​across the entire ultra-wideband operating frequency domain. Generally, the inductance fluctuation needs to be controlled within 20%, which requires a good understanding of the process used.

[0040] S300: The mismatch coefficient of the high-order matching network is obtained by jointly analyzing the expressions of the low-frequency poles, high-frequency poles, source-end resonant cavity resonant frequency and load resonant cavity resonant frequency in the frequency domain.

[0041] The low-frequency poles and high-frequency poles of the MCMCR can be represented as (5) and (6), respectively:

[0042]

[0043]

[0044] in, The resonant frequency of the source-end resonant cavity is . f is the resonant frequency of the load resonant cavity. cHere, k represents the center frequency of the ultra-wideband amplifier's operating bandwidth. k is the transformer coupling coefficient. With k determined, the mismatch coefficient α can be determined using the high-frequency or low-frequency pole formula; different values ​​of α will strongly affect the bandwidth. Comparative analysis shows that the α value obtained using the high-frequency pole formula is often close to 1, while the α value obtained using the low-frequency pole formula is far from 1. Figure 5 The diagram shows the simulation results of the impact of different values ​​of α on the bandwidth of a higher-order matched network. Only the range greater than 1 is shown here, because α>1 corresponds to the same bandwidth and gain obtained in the inverse mapping for values ​​less than 1, only the self-inductance of the corresponding transformer coil is different. It can be seen that the further α is from 1, the larger the bandwidth of the higher-order matched network, and the lost passive gain can be compensated by active modules. Therefore, the low-frequency pole formula is usually used to solve for the mismatch coefficient α of the MCMCR.

[0045] Joint f L The expressions for f1, f2 and other known parameters f c f lower , k, can be used to obtain the expression The expression contains two real solutions, which are reciprocals of each other; the only difference is the L value of MCMCR. P L S The values ​​of the coil inductance and the capacitance of the parallel capacitors C1 and C2 can be chosen based on the actual difficulty of the transformer design.

[0046] S400 transforms the calculation of the load resonant cavity quality factor into solving the constraint conditions based on the constraints between the coupling coefficient of the transformer, the mismatch coefficient of the high-order matching network and the quality factor of the load resonant cavity, and obtains the source-end resonant cavity quality factor based on the mismatch relationship between the load resonant cavity quality factor and the high-order matching network.

[0047] like Figure 3 As shown, this is because only when the precondition f1 / f2 = Q is satisfied... S / Q L =α magnetically coupled resonator, fourth-order matched network at the double-pole frequency Z 21 Only when the amplitudes are strictly equal will they be a necessary condition for improving in-band flatness. Otherwise, in the frequency domain, Z... 21 It will exhibit two peaks of unequal amplitude, and the band itself has an insurmountable flatness problem, making it unworthy of analysis. Among them, Q... L It is the quality factor of the load resonator, Q. S It is the quality factor of the source-end resonant cavity.

[0048] This step provides a constraint to achieve high gain flatness over the ultra-wideband frequency range between the high and low frequency poles. This is because the Z-axis of the two pole frequencies of the MCMCR... 21They are the same, sharing the same expression (7):

[0049]

[0050] At the trough frequency, there is (8):

[0051]

[0052] Equal expressions (7) and (8) represent the Z values ​​of peak and trough points within the band. 21 Equal, mathematically speaking, the passive gain is completely consistent at all frequencies between the high and low frequency poles, and has a strict flatness. The equation obtained by solving it is the in-band gain-free ripple constraint condition of MCMCR (9):

[0053]

[0054] Since the values ​​of k and α are already determined, the load resonant cavity quality factor Q can be solved by applying the constraints. L Analytical solution Prerequisites for applying MCMCR Q S / Q L The quality factor Q of the source-end resonant cavity can be determined by α. S .

[0055] S500 determines the parameter values ​​of the internal components of the high-order matching network based on the quality factors of the source resonant cavity and the load resonant cavity, thus obtaining the final designed high-order matching network.

[0056] The parameter values ​​of the internal components of the high-order matching network include: the parameter value of the first capacitor C1, and the parameter value of the transformer primary coil L. P The parameter values ​​are determined based on the quality factor definitions of the source-end resonant cavity and the load resonant cavity. This allows us to determine the parameter values ​​of each component in the high-order matching network, including C1 and L. P L S And C2, transformer secondary coil L S The parameter values ​​of capacitor C1 and capacitor C2 are expressed as follows:

[0057]

[0058]

[0059]

[0060]

[0061] Similarly, Figure 3 As shown, there is a trough frequency f between the two pole frequencies of the MCMCR. VThe difference in Z21 amplitude between the trough frequency and the double-pole frequency is called gain ripple, which is a parameter for measuring the in-band flatness of the proposed fourth-order matched network. To improve the in-band flatness of the MCMCR, the trough frequency f needs to be increased. V Z at the location 21 Amplitude, until it is related to f H f L Equalization at the point of view can reduce or even eliminate gain ripple. Based on this, a design method is found to enable high-order matching networks based on the MCMCR structure to possess ultra-wideband and high gain flatness.

[0062] This invention provides an ultra-wideband amplifier, including a high-order matching network designed using a design method for an ultra-wideband amplifier.

[0063] The other structures of the ultra-wideband amplifier of the present invention are the same as those of the prior art, but the matching network adopts the high-order matching network designed in the present invention. In this way, the ultra-wideband amplifier can achieve impedance matching between the source and the load in the ultra-wideband range and achieve high gain flatness amplification in the ultra-wideband range.

[0064] The following simulation experiments illustrate the performance of the high-order matching network designed by the design method of the high-gain flatness high-order matching network provided by this invention.

[0065] Simulation conditions:

[0066] Within the frequency range of 6GHz-18GHz, the SMIC 40nm CMOS process model is used, where R... S =29.5Ω, R L =9.8Ω, C S =-6.5Ff, C L =826.7GHz, f c =12GHz, negative capacitance is only a process quantity in the analysis, the actual model is C S The capacitor connected in parallel with C1 is a positive capacitor, which is simulated using the Cadence simulation tool in this example.

[0067] Simulation content:

[0068] Under the above conditions, the high-order matching network with ultra-wideband high-gain flatness proposed in this application and its design method are simulated and verified. For the bandwidth simulation of the high-order matching network, please refer to [link to relevant documentation]. Figure 6 As shown in the figure, this application achieves excellent ultra-wideband high-gain flatness amplification. Specifically, the 1dB bandwidth extends to 3.7GHz-22.8GHz, and within the required 6GHz-18GHz range, the in-band gain ripple is less than 0.15dB, achieving high-flatness amplification. Finally, L is obtained. P =229.8 pH, LS =633.2pH, C1=356.2fF, C2=226.3fF.

[0069] In existing fourth-order matching network schemes, there is no precise expression to quantify bandwidth and in-band flatness, or some imprecise expressions can only be used under certain special conditions, limiting their practical application. This invention derives the equivalent module of the single-port pre-amplifier input and the equivalent module of the single-port post-amplifier output; then determines the internal structure of the high-order matching network and the coupling coefficient of the transformer; analyzes and obtains the mismatch coefficient of the high-order matching network; based on the constraints between the transformer coupling coefficient, the mismatch coefficient of the high-order matching network, and the load resonant cavity quality factor, the calculation of the load resonant cavity quality factor is transformed into solving the constraints, and the source-end resonant cavity quality factor is obtained based on the mismatch relationship between the load resonant cavity quality factor and the high-order matching network; finally, the parameter values ​​of the internal components of the high-order matching network are determined, resulting in the final designed high-order matching network. Compared to existing technologies, this invention can achieve impedance matching between the source and load over an ultra-wideband range and achieve high-gain flatness amplification within the ultra-wideband range.

[0070] Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of technical features indicated. Thus, a feature defined as "first" or "second" may explicitly or implicitly include one or more of that feature. In the description of this invention, "a plurality of" means two or more, unless otherwise explicitly specified.

[0071] Although this application has been described herein in conjunction with various embodiments, those skilled in the art will understand and implement other variations of the disclosed embodiments by reviewing the accompanying drawings, the disclosure, and the appended claims in carrying out the claimed application. In the claims, the word "comprising" does not exclude other components or steps, and "a" or "an" does not exclude a plurality.

[0072] The above description, in conjunction with specific preferred embodiments, provides a further detailed explanation of the present invention. It should not be construed that the specific implementation of the present invention is limited to these descriptions. For those skilled in the art, various simple deductions or substitutions can be made without departing from the concept of the present invention, and all such modifications and substitutions should be considered within the scope of protection of the present invention.

Claims

1. A design method for a high-order matching network applied to an ultra-wideband amplifier, characterized in that, include: S100 obtains a single-port pre-stage input equivalent module by conjugating the complex impedance connected in series with the pre-stage input module, and obtains a single-port post-stage output equivalent module by paralleling the inductor-adjusted post-stage impedance. S200, design the internal structure of the high-order matching network based on the single-port pre-amplifier equivalent module and the single-port post-amplifier equivalent module, and determine the coupling coefficient of the transformer contained in the high-order matching network based on the operating frequency of the ultra-wideband amplifier and the applied process. S300, by jointly analyzing the expressions of the low-frequency poles, high-frequency poles, source-end resonant cavity resonant frequency and load resonant cavity resonant frequency of the high-order matching network in the frequency domain, the mismatch coefficient of the high-order matching network is obtained. S400, based on the constraints between the coupling coefficient of the transformer, the mismatch coefficient of the high-order matching network and the quality factor of the load resonant cavity, the calculation of the quality factor of the load resonant cavity is transformed into solving the constraints, and the quality factor of the source resonant cavity is obtained based on the mismatch relationship between the quality factor of the load resonant cavity and the high-order matching network. S500, determine the parameter values ​​of the internal components of the high-order matching network based on the quality factor of the source resonant cavity and the quality factor of the load resonant cavity, and obtain the final designed high-order matching network.

2. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 1, characterized in that, The high-order matching network described in S200 is designed based on a mismatched consistent magnetic coupling resonant architecture. The high-order matching network includes: a dual-port fourth-order matching network, a single-port pre-stage input equivalent module, and a single-port post-stage output equivalent module. The single-port pre-stage input equivalent module is electrically connected to the dual-port fourth-order matching network to form a source resonant cavity; the single-port post-stage output equivalent module is electrically connected to the dual-port fourth-order matching network to form a load resonant cavity.

3. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 2, characterized in that, The dual-port fourth-order matching network includes: a first capacitor C1, and a transformer primary coil L. P Transformer secondary coil L S Second capacitor C2; Wherein, the positive terminal of the first capacitor C1 is connected to the primary coil L of the transformer. P The positive terminal of the first capacitor C1 is connected to the positive terminal of the transformer secondary coil L, and serves as the positive terminal of the higher-order matching network input. S The negative terminal of the first capacitor is connected to the second capacitor C2, which serves as the negative input of the two-port fourth-order matching network; the positive terminal of the second capacitor C2 is connected to the transformer secondary coil L. S The positive terminal of the first capacitor is connected to the second capacitor C2, which serves as the positive terminal of the output of the two-port fourth-order matching network. The negative terminal of the second capacitor C2 is connected to the secondary coil L of the transformer. S The negative end is connected and serves as the negative end of the output of the two-port fourth-order matching network.

4. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 2, characterized in that, The single-port pre-amplifier equivalent module includes a first resistor R. S and the third capacitor C S The single-port output equivalent module includes a second resistor R. L and the fourth capacitor C L ; Among them, the first resistor R S The positive terminal and the third capacitor C S The positive terminal of the first resistor Rs is connected to the positive terminal of the input of the two-port fourth-order matching network, and the negative terminal of the first resistor Rs is connected to the third capacitor C. S The negative terminal is connected to the negative terminal of the second resistor R, and then electrically connected to the negative terminal of the input terminal of the two-port fourth-order matching network. L The positive terminal and the fourth capacitor C L The positive terminal is connected to the positive terminal of the second resistor R, and then electrically connected to the positive terminal of the output of the two-port fourth-order matching network. L The negative terminal and the fourth capacitor C L The negative terminal is connected to the negative terminal of the output of the two-port fourth-order matched network, and then electrically connected to the negative terminal of the output of the two-port fourth-order matched network.

5. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 2, characterized in that, Low-frequency poles of high-order matching networks in S300 Represented as: ； Among them, the resonant frequency of the source-end resonant cavity The resonant frequency of the load resonant cavity , The center frequency of the ultra-wideband amplifier. The coupling coefficient of the transformer. This is the mismatch coefficient.

6. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 5, characterized in that, The mismatch coefficient expression in S300 is: ； in, This is the lower limit of the operating frequency band for ultra-wideband amplifiers.

7. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 5, characterized in that, The constraints in S400 are expressed as follows: ； The mismatch relationship in a high-order matching network is represented as follows: ； in, It is the quality factor of the load resonant cavity. It is the quality factor of the source-end resonant cavity.

8. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 6, characterized in that, The parameter values ​​of the internal components of the high-order matching network in S500 include: the parameter value of the first capacitor C1, the parameter value of the transformer primary coil L... P The parameter values ​​of the transformer secondary coil L S The parameter values ​​of capacitor C1 and capacitor C2 are expressed as follows: ； ； ； 。 9. The design method of a high-order matching network for an ultra-wideband amplifier according to claim 3, characterized in that, The first capacitor C1 and the second capacitor C2 are implemented using on-chip capacitors, on-chip variable capacitors, or parasitic capacitance between inductors.

10. An ultra-wideband amplifier, characterized in that, It includes a high-order matching network, which is obtained by the design method of a high-order matching network for an ultra-wideband amplifier as described in any one of claims 1 to 9.

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