Power conversion circuit device

Abstract

The power conversion circuit device is provided with an inverter circuit between input terminals (T01, T02) and output terminals (T11, T12). The first connection point (N11) is connected to the first output terminal (T11) via a first inductor (L21), and the second connection point (N12) is connected to the second output terminal (T12) via a second inductor (L22). A first impedance element (A1) is connected between the first connection point (N11) and the second input terminal (T02), a second impedance element (A2) is connected between the second connection point (N12) and the second input terminal (T02), a third impedance element (B1) is connected between the first output terminal (T11) and the second input terminal (T02), and a fourth impedance element (B2) is connected between the second output terminal (T12) and the second input terminal (T02).

Description

Technical Field

[0001] The present disclosure relates to a power conversion circuit device including, for example, an inverter circuit. Background Art

[0002] Industrial equipment such as processing machines and assembly machines are usually equipped with motor systems. The power section of the motor system consists of the following five blocks.

[0003] (1) Connect the power cable to the AC power supply in the factory.

[0004] (2) A rectifier circuit that converts AC power into DC power.

[0005] (3) An inverter circuit that generates a PWM (Pulse Width Modulation) signal from DC power.

[0006] (4) Motor.

[0007] (5) Motor cables connecting the inverter circuit to the motor.

[0008] Among them, in the inverter circuit of (3) above, PWM control is achieved by switching actions of IGBT (Insulated Gate Bipolar Transistor) or MOSFET (Metal Oxide Semiconductor Field-Effect Transistor). As with power conversion circuits such as DCDC converters, electromagnetic noise is generated during switching actions, so it is necessary to use noise filters and the like as countermeasures.

[0009] Electromagnetic noise is classified into conducted noise and radiated noise. Radiated noise is easily radiated into space from a long conductor such as a cable. In factories, there is a lack of space in the equipment where the motor is assembled, so sometimes the rack that houses the inverter circuit is set apart from the equipment. In this case, the motor cable becomes longer, so the dominant source of radiated noise becomes the motor cable.

[0010] In addition, it is well known that common mode noise components are more likely to cause radiation noise than differential mode noise components. Therefore, in order to suppress the radiation noise of the motor system in the factory environment, it is important to reduce the "common mode" noise components flowing in the "motor cable".

[0011] Patent Document 1 discloses a circuit configuration that uses active elements such as transistors to reduce common mode noise in an inverter circuit.

[0012] Prior art literature

[0013] Patent Literature

[0014] Patent Document 1: Japanese Patent No. 6803478 Summary of the invention

[0015] Problem that the invention aims to solve

[0016] The advantage of using active components to eliminate noise is that the noise reduction performance is high. On the other hand, due to the frequency response performance limit of active components and the nonlinear frequency characteristics of magnetic components used for noise detection, the performance is prone to deterioration in high-frequency bands where radiation noise becomes a problem. In addition, the cost of components with good frequency characteristics is high. Therefore, it is required to use "passive components" to reduce the "high-frequency" "common mode" noise components flowing in the "motor cable".

[0017] An object of the present disclosure is to provide a power conversion circuit device including, for example, an inverter circuit, which can reduce common mode noise flowing in a motor cable without using expensive active components or large anti-noise components, thereby suppressing radiation noise.

[0018] Solutions for solving problems

[0019] In a power conversion circuit device according to one embodiment of the present disclosure,

[0020] An inverter circuit including a bridge circuit is provided between the first input terminal and the second input terminal and the first output terminal and the second output terminal, wherein the bridge circuit is formed by connecting a first switching element connected in series with a second switching element and a third switching element and a fourth switching element connected in series with each other in parallel, wherein:

[0021] Two ends of the series circuit of the first switch element and the second switch element, and two ends of the series circuit of the third switch element and the fourth switch element are defined as a first connection point and a second connection point, respectively, and the first connection point is connected to the first input terminal via the first inductor, and the second connection point is connected to the second input terminal via the second inductor,

[0022] A third connection point between the first switch element and the second switch element is connected to the first output terminal via a third inductor.

[0023] A fourth connection point between the third switch element and the fourth switch element is connected to the second output terminal via a fourth inductor.

[0024] The power conversion circuit device comprises:

[0025] a first impedance element connected between the third connection point and the first input terminal or the second input terminal;

[0026] a second impedance element connected between the fourth connection point and the first input terminal or the second input terminal;

[0027] A third impedance element connected between the first output terminal and the first input terminal or the second input terminal; and

[0028] A fourth impedance element is connected between the second output terminal and the first input terminal or the second input terminal.

[0029] Effects of the Invention

[0030] Therefore, according to a power conversion circuit device involved in one embodiment of the present disclosure, it is possible to reduce common mode noise flowing in a motor cable of a power supply circuit related to an inverter circuit, for example, thereby suppressing radiation noise. As a result, compared with conventional inverter circuits, it is possible to reduce anti-noise components, thereby achieving miniaturization, lightness, and cost reduction of the device. BRIEF DESCRIPTION OF THE DRAWINGS

[0031] Figure 1 1 is a circuit diagram showing a configuration of a power conversion circuit device related to a single-phase inverter circuit according to Comparative Example 1.

[0032] Figure 2 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the first embodiment.

[0033] Figure 3 yes Figure 2 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0034] Figure 4A It is shown Figure 2 1 is a circuit diagram of a configuration example 1 of an impedance element A1 of a power conversion circuit device.

[0035] Figure 4B It is shown Figure 2 1 is a circuit diagram of a second structural example of an impedance element A1 of a power conversion circuit device.

[0036] Figure 4C It is shown Figure 2 1 is a circuit diagram of a third structural example of the impedance element A1 of the power conversion circuit device.

[0037] Figure 4D It is shown Figure 2 1 is a circuit diagram of a fourth structural example of an impedance element A1 of a power conversion circuit device.

[0038] Figure 4E It is shown Figure 4D An equivalent circuit diagram of a structural example of a ferrite bead BD1.

[0039] Figure 5 yes Figure 2 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0040] Figure 6 It is shown Figure 1 and Figure 2 Spectrum diagram comparing the common-mode current suppression effects in power conversion circuit devices.

[0041] Figure 7 This is a circuit diagram showing a configuration example of a power conversion circuit device according to a modification of the first embodiment.

[0042] Figure 8 yes Figure 7 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0043] Fig. 9 yes Figure 7 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0044] Fig.10 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the second embodiment.

[0045] Fig.11 yes Fig.10 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0046] Fig.12 yes Fig.10 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0047] Fig.13 It is shown Figure 1 and Fig.10 Spectrum diagram comparing the common-mode current suppression effects of power conversion circuit devices.

[0048] Fig.14 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the third embodiment.

[0049] Fig.15 This is a circuit diagram showing a configuration example of a power conversion circuit device according to a fourth embodiment.

[0050] Fig.16 : is a circuit diagram showing the structure of a power conversion circuit device related to a three-phase inverter circuit of Comparative Example 2.

[0051] Fig.17 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the fifth embodiment.

[0052] Fig.18yes Fig.17 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0053] Fig.19 yes Fig.17 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0054] Fig. 20 It shows that Fig.16 and Fig.17 Spectrum diagram comparing the common-mode current suppression effects of power conversion circuit devices.

[0055] Fig.21 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the sixth embodiment.

[0056] Fig. 22 yes Fig.21 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0057] Fig.23 yes Fig.21 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0058] Fig.24 yes Fig.21 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0059] Fig.25 yes Fig.21 Equivalent circuit diagram of the power conversion circuit device in the radiation noise band domain.

[0060] Fig.26 It is shown Fig.16 and Fig.21 Spectrum diagram comparing the common-mode current suppression effects of power conversion circuit devices.

[0061] Fig. 27 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the seventh embodiment. DETAILED DESCRIPTION

[0062] Hereinafter, embodiments and modifications of the present disclosure will be described with reference to the accompanying drawings. In addition, the same or similar components are denoted by the same reference numerals.

[0063] (Opinion of the inventor)

[0064] Figure 1 1 is a circuit diagram showing a configuration of a power conversion circuit device related to a single-phase inverter circuit according to output comparison example 1. Figure 1In the embodiment, a smoothing circuit and a switching inverter circuit such as a PWM method are inserted between the input terminals T01 and T02 and the output terminals T11 and T12.

[0065] A DC voltage of DC power is applied between the input terminal T01 and the input terminal T02. The input terminal T01 is connected to the node N01 via the inductors L01 and L11, and the input terminal T02 is connected to the node N02 via the inductors L02 and L12. The connection point between the inductor L01 and the inductor L11 is connected to the connection point between the inductors L02 and L12 via the capacitor C1, and the node N01 is connected to the node N02 via the capacitor C2. The input terminal T01 is grounded via the capacitor C3, and the input terminal T02 is grounded via the capacitor C4.

[0066] Here, capacitors C1 and C2 are X capacitors, for example, capacitor C1 is a smoothing capacitor, and capacitor C2 is a buffer capacitor. Capacitors C3 and C4 are Y capacitors for noise suppression. Inductors L01, L02, L11, L12, L21, and L22 are wiring inductors (Japanese: wiring inductors) or choke coils. Switching elements S1 to S4 are, for example, N-channel MOS transistors, and switching elements S1 and S2, S3 and S4 are connected in series to form a full-bridge circuit. Here, node N01 is connected to node N02 via the drain and source of switching element S1, and the drain and source of switching element S2. In addition, node N01 is connected to node N02 via the drain and source of switching element S3, and the drain and source of switching element S4. A node N11 as a connection point between the source of the switching element S1 and the drain of the switching element S2 is connected to the output terminal T11 via the inductor L21 , and a node N12 as a connection point between the source of the switching element S3 and the drain of the switching element S4 is connected to the output terminal T12 via the inductor L22 .

[0067] The control circuit 1 generates known command signals SS1 to SS4 for performing PWM switching operations (two phases) and outputs them to the gates of the switching elements S1 to S4, respectively, thereby switching the switching elements S1 to S4 between on and off to generate the desired PWM voltage at the output terminals T11 and T12.

[0068] At this time, when a common mode current is generated that flows in the same phase in the inductors L21 and L22 and is transmitted from the output terminals T11 and T12 to, for example, a motor cable connected to a motor of the load, the radiation noise increases. In order to suppress the common mode current, it is necessary to take measures such as providing a ferrite core in the motor cable, which not only generates additional costs but also increases the size and weight of the equipment.

[0069] The inventors of the present invention have proposed power conversion circuit devices according to the following embodiments and modifications in view of these findings to solve the problems.

[0070] (Implementation Method 1)

[0071] Figure 2 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the first embodiment. Figure 2 Power conversion circuit device and Figure 1 Compared with the power conversion circuit device of , it has the following differences.

[0072] (1) Impedance elements A1, A2, B1, and B2 are further provided.

[0073] The differences are described below.

[0074] exist Figure 2 In FIG. 1 , impedance element A1 is connected between node N11 and input terminal T02, and impedance element A2 is connected between node N12 and input terminal T02. Impedance element B1 is connected between node N02 and output terminal T11, and impedance element B2 is connected between node N02 and output terminal T12.

[0075] In the frequency band of 30 MHz to 300 MHz (radiation noise band) where radiation noise becomes a problem, the impedance Z (=1 / (2πfC)) of capacitors C1 to C4 is sufficiently small compared to other elements and can be approximated as a short circuit (Z≈0).

[0076] Figure 3 This is the approximate state Figure 2 Equivalent circuit diagram of the radiation noise band of the power conversion circuit device. Figure 2 In the case where the capacitor C2 is approximately short-circuited, the potentials of the nodes N01 and N02 are equal, so the drain-source voltage of the switching element S1 is consistent with the source-drain voltage of the switching element S2. Figure 3 In the figure, it is represented by voltage source VS12. Figure 3 In the figure, the focus is on the noise transmission from the pair of switching elements S1 and S2 (hereinafter referred to as the "switching element pair") of the first phase, so the switching element pair S3 and S4 of the second phase is considered to be short-circuited. This is because according to the superposition theorem, the voltage source other than the voltage source of interest can be considered to be short-circuited. In addition, the reference symbol "X / / Y" in the figure represents a parallel connection circuit of element X and element Y. In addition, in Figure 3 In the embodiment, since the capacitor C3 and the capacitor C4 are approximately short-circuited, the input terminal T01 and the input terminal T02 are regarded as the same node in the "radiation noise band", and the same shall apply hereinafter.

[0077] according to Figure 3It is clear that the structure of the bridge circuit is formed from the voltage source VS12 to the output terminal T11 of the first phase by adding impedance elements A1 and B1. Therefore, although the detailed principle will be described later, the impedance balancing method can be used to reduce the noise transmission from the voltage source VS12 to the output terminal T11. Similarly, the noise transmission from the switching element pair S3 and S4 of the second phase to the output terminal T12 can also be reduced. As a result, the common mode current transmitted in the wiring extending from the output terminals T11 and T12 can be suppressed, the radiation noise can be reduced, and the cost reduction and miniaturization and lightness can be achieved by reducing the counter components.

[0078] Figure 4A to Figure 4D It is shown Figure 2 The circuit diagrams of the structure examples 1 to 4 of the impedance element A1 of the power conversion circuit device are shown in FIG. As the implementation methods of the impedance elements A1, A2, B1, and B2, the series circuits of inductors and capacitors are discussed respectively. For example, Figure 4A The impedance element A1 is expressed in this way, and the impedance elements A2, B1, and B2 can be expressed in the same way.

[0079] That is, in Figure 4A In FIG. 1 , the impedance element A1 is represented by a series circuit of a capacitor CA1 and an inductor LA1, for example. Similarly,

[0080] (1) The impedance element A2 is represented by, for example, a series circuit of a capacitor CA2 and an inductor LA2.

[0081] (2) The impedance element B1 is represented by, for example, a series circuit of a capacitor CB1 and an inductor LB1.

[0082] (3) The impedance element B2 is represented by, for example, a series circuit of a capacitor CB2 and an inductor LB2.

[0083] exist Figure 4A In the embodiment, the terminals of the impedance element A1 are insulated from direct current by the presence of the capacitor CA1, so that the same PWM voltage generation as the inverter circuit involved in the comparative example 1 can be achieved. In addition, in the radiation noise band, the impedance of the capacitor CA1 is sufficiently small compared to other elements, so it can be approximated as a short circuit. Therefore, in the radiation noise band, the impedance of the inductor LA1 is dominant. At this time, Figure 3 The equivalent circuit can be specifically expressed as Figure 5 Express it that way.

[0084] In this specification, the inductance of each inductor L is represented by the same reference numeral L, and the capacitance value of each capacitor C is represented by the same reference numeral C. In addition, the inductance L01 is different from the inductance L1, and the inductance L02 is different from the inductance L2.

[0085] Figure 5 yes Figure 2 Equivalent circuit diagram of the radiation noise band of the power conversion circuit device. Figure 5 In the embodiment, the only component of the bridge circuit is the inductor, so a significant noise reduction effect can be obtained in a wideband region by using the impedance balance method. In addition, the design becomes easier. In addition, in order to make the impedance of the inductor LA1 dominant in the radiation noise band as described above, the series resonance frequency of the capacitor CA1 and the inductor LA1 needs to be designed to be below the radiation noise band.

[0086] It is also possible to add a resistance element connected in series or in parallel with the inductor to the impedance elements A1, A2, B1, and B2.

[0087] (1) Figure 4B As shown, the impedance element A1 may be formed by a series circuit of a resistor R1, an inductor LA1, and a capacitor CA1, and the impedance elements A2, B1, and B2 are similarly formed.

[0088] (2) Figure 4C As shown, the impedance element A1 may be formed by a series circuit of a capacitor CA1 and a resistor R1 and an inductor LA1 connected in parallel, and the impedance elements A2, B1, and B2 are similarly formed.

[0089] This stabilizes the impedance of the impedance elements A1, A2, B1, and B2 at their self-resonant frequencies, and prevents the generation of noise peaks. In other words, the noise reduction effect can be obtained in a wideband region.

[0090] In addition, as an impedance element including an inductor, a ferrite bead BD1 may also be used. Figure 4D As shown in FIG. 1 , the impedance element A1 can also be formed by a series circuit of a ferrite bead BD1 and a capacitor CA1, and the same applies to the impedance elements A2, B1, and B2. In general, the equivalent circuit of the ferrite bead BD1 is a circuit obtained by connecting another resistor R2 in series to the parallel circuit of the inductor LA1 and the resistor R1 ( Figure 4E ) etc. As shown in this way, when the ferrite bead BD1 is used, since a resistance component is included therein, the noise reduction effect can be stably obtained without using a resistance element.

[0091] Next, conditions for maximizing the noise reduction effect are shown.

[0092] exist Figure 5 In the equivalent circuit of , the upper left component of the bridge circuit for output terminal T11 is represented by LA2 / / (L01 / / L02+L11 / / L12). The inductance of the composite inductor is set to inductor L1. The lower left, upper right, and lower right components of the bridge circuit are inductors LA1, LB1, and L21, respectively. Figure 5When the circuit is on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+LA1)×VS12. The input terminals T01 and T02 are at ground potential in the radiation noise band, so the potential at the nodes N01, N02, and N12 is L1 / (L1+LA1)×VS12.

[0093] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the upper right and lower right inductors of the bridge circuit. That is, the voltage applied to the upper right inductor of the bridge circuit is LB1 / (LB1+L21)×VS12. Here, when L1 / (L1+LA1)×VS12=LB1 / (LB1+L21)×VS12, the potential of the output terminal T11 is 0V. Since the ground potential is 0V, it means that no common mode voltage is generated. Therefore, when this condition is met, a significant noise reduction effect can be expected.

[0094] In addition, the above-mentioned design conditional expression can be expressed in the form of an inductance ratio as L1:LA1=LB1:L21. This is the impedance balancing method. L01 / / L02+L11 / / L12 is the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the node N01 and the node N02 are short-circuited.

[0095] Therefore, the optimal conditions are as follows: The inductance of the impedance elements A1 and A2 or the inductance of the magnetic beads is set to L2, the inductance of the parallel inductor of the inductor L2 and the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the nodes N01 and N02 are short-circuited is set to inductance L1, the inductance of the impedance elements B1 and B2 or the inductance of the magnetic beads is set to L3, and the inductance of the inductors L21 and L22 is set to L4. In this case, L1:L2=L3:L4.

[0096] In any embodiment, a common mode choke coil in which inductors L01 and L02 are coupled may be used, or a common mode choke coil in which inductors L11 and L12 are coupled may be used. In this case, the design may be based on the above-mentioned "effective inductance between input terminal T01 and node N01 when input terminal T01 and input terminal T02 are short-circuited and nodes N01 and N02 are short-circuited". For example, when using a common mode choke coil in which L01 and L02 (= L01) are coupled with a coupling degree k0 and a common mode choke coil in which L11 and L12 (= L11) are coupled with a coupling degree k1, the effective inductance between input terminal T01 and the node in the bridge circuit is L01×(1+k0) / 2+L11×(1+k1) / 2.

[0097] Although not as good as the above optimal conditions, noise reduction effects can be expected under the following conditions. Figure 5 The voltage of the upper right component of the bridge circuit is, for example, in the range of 0.5 to 1.5 times the voltage applied to the upper left component of the bridge circuit.

[0098] (1) At 0.5 times, L1 / (L1+LA1)×VS12=0.5×LB1 / (LB1+L21)×VS12.

[0099] (2) When it is 1.5 times, L1 / (L1+LA1)×VS12=1.5×LB1 / (LB1+L21)×VS12.

[0100] Therefore, in the formula L1 / (L1+LA1)×VS12=a×LB1 / (LB1+L21)×VS12, that is, in the formula L1 / (L1+LA1)=a×LB1 / (LB1+L21), when the coefficient a is 0.5≤a≤1.5 (when the value of each component is set in such a way that the formula holds true), a noise reduction effect can be expected.

[0101] The effect of the power conversion circuit device according to Embodiment 1 is shown using circuit simulation. The circuit parameters are as follows: The parameters in parentheses are parasitic components considered in the series connection.

[0102] C1=1mF(10nH,10mΩ),

[0103] C2=100nF(1nH,10mΩ),

[0104] C3=C4=4.7nF (10nH, 0.2Ω),

[0105] L01=L02=L11=L12=0.5μH,

[0106] L21=L22=0.5μH.

[0107] In addition, impedance elements A1 and A2 were set to a series circuit of 0.5μH, 200pF and 30Ω, and impedance elements B1 and B2 were set to a series circuit of 0.25μH, 200pF and 30Ω. A DC voltage of 282V was input between input terminal T01 and input terminal T02, and the switching frequency (carrier frequency) of switching elements S1 to S4 was set to 12kHz. The common-mode current was compared when a 5m long motor cable (T-type equivalent circuit) and a motor as a load (winding inductance of 1μH, parasitic capacitance between winding and housing of 0.5nF) were connected to the output terminals T11 and T12 of the inverter circuit.

[0108] Figure 6 It is shown Figure 1 and Figure 2 Spectra of the comparison of common mode current in the power conversion circuit device. Figure 6 It is clear that a suppression effect of 10dB to 20dB can be confirmed at the noise peak that is prone to cause problems in the radiation noise band. In particular, at about 50MHz, the suppression effect is DP11 (= P01-P11), and at about 70MHz, the suppression effect is DP12 (= P02-P12). Here, P01 and P02 are the peak values ​​of the common mode current of Comparative Example 1, and P11 and P12 are the peak values ​​of the common mode current of Implementation Example 1.

[0109] The same effect is achieved when the impedance elements A1 and A2 are connected to the input terminal T01 instead of the input terminal T02 . The same effect is achieved when the impedance elements B1 and B2 are connected to the node N01 instead of the node N02 .

[0110] (Variation of Implementation Example 1)

[0111] Figure 7 This is a circuit diagram showing a configuration example of a power conversion circuit device according to a modification of the first embodiment. Figure 7 Power conversion circuit device and Figure 2 Compared with the power conversion circuit device of , it has the following differences.

[0112] (1) Impedance elements A3, A4, B3, and B4 are further provided.

[0113] The differences are described below.

[0114] exist Figure 7In FIG. 1 , impedance element A3 is connected between node N11 and input terminal T01, impedance element A4 is connected between node N12 and input terminal T01, impedance element B3 is connected between node N01 and output terminal T11, impedance element B4 is connected between node N01 and output terminal T12.

[0115] In the above structure Figure 7 In the variation of Figure 2 The noise reduction effect can also be obtained for the same reason as in Embodiment 1. However, the design conditional expression is different, so it will be described below.

[0116] The impedance elements A1 to A4 and B1 to B4 are connected to both the positive and negative electrodes, but in the equivalent circuit, the input terminal T01 and the input terminal T02 are considered to have the same potential, and the node N01 and the node N02 are considered to have the same potential. Therefore, the equivalent circuit has the same circuit structure as that of the first embodiment. Figure 8 express.

[0117] Figure 8 yes Figure 7 Equivalent circuit diagram of the radiation noise band of the power conversion circuit device. Figure 8 Clearly, with Figure 3 Similarly, the structure of the bridge circuit is formed from the voltage source VS12 to the output terminal T11 of the first phase. Therefore, the noise transmission from the voltage source VS12 to the output terminal T11 can be reduced by using the impedance balancing method. Similarly, the noise transmission from the switching element pair S3 and S4 of the second phase to the output terminal T12 can also be reduced. As a result, the common mode current transmitted in the wiring extending from the output terminals T11 and T12 can be suppressed, the radiation noise can be reduced, and the cost reduction and compactness and lightness can be achieved by reducing the counter components.

[0118] As a method of realizing the impedance elements A1 to A4 and B1 to B4, the series circuits of inductors and capacitors are discussed respectively. For example, if the impedance element A1 is shown in a circuit diagram, for example, Figure 4A The impedance elements A2 to A4 and B1 to B4 are also represented in the same way.

[0119] (1) The impedance element A3 is represented by, for example, a series circuit of a capacitor CA3 and an inductor LA3.

[0120] (2) The impedance element A4 is represented by, for example, a series circuit of a capacitor CA4 and an inductor LA4.

[0121] (3) The impedance element B3 is represented by, for example, a series circuit of a capacitor CB3 and an inductor LB3.

[0122] (4) The impedance element B4 is represented by, for example, a series circuit of a capacitor CB4 and an inductor LB4.

[0123] In the impedance element A1, the terminals of the impedance element A1 are insulated from direct current by the presence of the capacitor CA1, so that the same PWM voltage generation as that of the inverter circuit involved in the comparative example 1 can be achieved. In addition, in the radiation noise band, the impedance of the capacitor CA1 is sufficiently small compared to other elements, so it can be approximated as a short circuit. Therefore, in the radiation noise band, the impedance of the inductor LA1 is dominant. At this time, Figure 8 The equivalent circuit can be specifically expressed as Fig. 9 Express it that way.

[0124] Fig. 9 yes Figure 7 Equivalent circuit diagram of the radiation noise band of the power conversion circuit device. Fig. 9 It is clear that the only component of the bridge circuit is the inductor, so it is possible to achieve a significant noise reduction effect in a wideband range by using the impedance balance method. In addition, the design is also made easier.

[0125] Furthermore, in order to make the impedance of the inductor LA1 dominant in the radiation noise band as described above, the series resonance frequency of the capacitor CA1 and the inductor LA1 needs to be designed to be equal to or lower than the radiation noise band.

[0126] Here, you can also Figure 4B to Figure 4D As shown in the figure, a resistor element is further added to the impedance elements A1 to A4, B1 to B4 in series or in parallel with the inductor. In this way, the impedance of the impedance element at the self-resonant frequency can be stabilized, and the peak of the noise can be avoided. That is, the noise reduction effect can be obtained in a wide band. In addition, as the impedance elements A1 to A4, B1 to B4 including the inductor, the ferrite bead BD1 can also be used. In general, the equivalent circuit of the ferrite bead BD1 is, for example, a circuit obtained by connecting another resistor R2 in series to the parallel circuit of the inductor LA1 and the resistor R1 ( Figure 4E ) etc. As shown in this way, when the ferrite bead BD1 is used, since a resistance component is included therein, the noise reduction effect can be stably obtained without using a resistance element.

[0127] Next, conditions for maximizing the noise reduction effect are shown.

[0128] exist Fig. 9In the equivalent circuit of , the upper left component of the bridge circuit for the output terminal T11 is represented by LA2 / / LA4 / / (L01 / / L02+L11 / / L12). The inductance of the composite inductor is set to inductance L1. The inductance of the composite inductor of the lower left component LA1 / / LA3 of the bridge circuit is set to inductance L2. The inductance of the composite inductor of the upper right component LB1 / / LB3 of the bridge circuit is set to inductance L3. The lower right component of the bridge circuit is inductor L21.

[0129] When to pay attention Fig. 9 When the circuit is on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+L2)×VS12. The input terminals T01 and T02 are at the ground potential in the radiation noise band, so the potential at the nodes N01, N02, and N12 becomes L1 / (L1+L2)×VS12.

[0130] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the inductors on the upper right and lower right sides of the bridge circuit. That is, the voltage applied to the inductor on the upper right side of the bridge circuit is L3 / (L3+L21)×VS12. Here, when L1 / (L1+L2)×VS12=L3 / (L3+L21)×VS12, the potential of the output terminal T11 is 0V. Since the ground potential is 0V, it means that no common mode voltage is generated. Therefore, when this condition is met, it is expected to achieve a significant noise reduction effect.

[0131] In addition, the above-mentioned design conditional expression can be expressed in the form of an inductance ratio as L1:L2=L3:L21. This is the impedance balance method. L01 / / L02+L11 / / L12 is the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the nodes N01 and N02 are short-circuited. Therefore, if the above-mentioned optimal conditions are explained, it is as follows. The inductance of impedance elements A1 to A4 or the inductance of magnetic beads is set to "2×L2", the parallel inductance of the above-mentioned inductor L2 and the effective inductance between input terminal T01 and node N01 when the input terminal T01 and input terminal T02 are short-circuited and the node N01 and node N02 are short-circuited is set to L1, the inductance of impedance elements B1 to B4 or the inductance of magnetic beads is set to "2×L3", and the inductance of inductors L21 and L22 is set to L4. At this time, L1:L2=L3:L4.

[0132] Although not as good as the above optimal conditions, noise reduction effects can be expected under the following conditions. Fig. 9The voltage of the upper right component of the bridge circuit is, for example, in the range of 0.5 to 1.5 times the voltage applied to the upper left component of the bridge circuit.

[0133] (1) At 0.5 times, L1 / (L1+L2)×VS12=0.5×L3 / (L3+L21)×VS12.

[0134] (2) When it is 1.5 times, L1 / (L1+L2)×VS12=1.5×L3 / (L3+L21)×VS12.

[0135] Therefore, in the formula of L1 / (L1+L2)×VS12=a×L3 / (L3+L21)×VS12, that is, in the formula of L1 / (L1+L2)=a×L3 / (L3+L21), when the coefficient a is 0.5≤a≤1.5 (this is when the value of each component is set in such a way that the formula holds true), a noise reduction effect can be expected.

[0136] exist Figure 2 In the first embodiment of the present invention, no impedance element is connected between the element terminals of the switching element S1 (between the drain and source in the case of a MOS transistor and between the collector and emitter in the case of an IGBT), but impedance elements A1 and B1 are connected between the terminals of the switching element S2. Figure 7 In the modified example, impedance elements A3 and B3 are connected between the element terminals of the switch element S1 (between the drain and source in the case of a MOS transistor and between the collector and emitter in the case of an IGBT), and impedance elements A1 and B1 are connected between the element terminals of the switch element S2. As a result, the elements of the switch elements S1 and S3 connected to the high voltage side and the switch elements S2 and S4 connected to the low voltage side are balanced, so that the switching operation time is the same degree of time. As a result, it is possible to achieve uniformity of switching operation loss and simplification of control design such as dead time adjustment.

[0137] exist Figure 7 In the power conversion circuit device, it is preferable to set the impedances of the impedance elements A1 to A4 and B1 to B4 to be substantially the same.

[0138] (Implementation Method 2)

[0139] Fig.10 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the second embodiment. Fig.10 Power conversion circuit device and Figure 2 Compared with the power conversion circuit device of , it has the following differences.

[0140] (1) Impedance elements D1 and D2 are further provided.

[0141] The differences are described below.

[0142] exist Fig.10 In the embodiment, the impedance element D1 is connected between the node N11 and the output terminal T12, and the impedance element D2 is connected between the node N12 and the output terminal T11.

[0143] In the above structure Fig.10 In the second embodiment, a more significant noise reduction effect can be obtained compared with the first embodiment. The principle will be described below.

[0144] Fig.11 yes Fig.10 The equivalent circuit diagram of the power conversion circuit device in the radiation noise band of FIG. The following principle is used in the same way as in the first embodiment: in the radiation noise band, the impedance of capacitors C1 to C4 is sufficiently small compared to other elements, so they can be approximated as a short circuit. However, since the focus is on the noise transmission from the switching element pair S1 and S2 of the first phase, the voltage of the switching element pair S1 and S2 is represented by the voltage source VS12, and the switching element pair S3 and S4 of the second phase is regarded as a short circuit.

[0145] according to Fig.11 It is clear that the voltage source VS12 to the output terminal T11 of the first phase is formed into a bridge circuit structure by adding impedance elements A1 and B1. Therefore, the noise transmission from the voltage source VS12 to the output terminal T11 can be reduced by using the impedance balancing method. As a result, the common mode current transmitted in the wiring extending from the output terminal T11 can be suppressed, the radiation noise can be reduced, and the cost can be reduced and the size and weight can be reduced by reducing the counter components.

[0146] Furthermore, the voltage source VS12 to the output terminal T12 of the second phase is also configured as a bridge circuit by adding impedance elements A1 and D1. Therefore, the impedance balancing method can also be used to reduce the noise transmission from the voltage source VS12 to the output terminal T12. Similarly, the noise transmission from the switch element pair S3 and S4 of the second phase to the output terminals T11 and T12 can also be reduced. As a result, the common mode current transmitted in the wiring extending from the output terminals T11 and T12 can be further suppressed, the radiation noise can be further reduced, and the cost reduction and compactness and lightness can be achieved by reducing the counter components.

[0147] As a method of realizing the impedance elements A1, A2, B1, B2, D1, and D2, a series circuit of an inductor and a capacitor is discussed. For example, if the impedance element A1 is shown in a circuit diagram, it is shown as shown in FIG4. The same is true for the impedance elements A2, B1, B2, D1, and D2. Here,

[0148] (1) The impedance element D1 is represented by, for example, a series circuit of a capacitor CD1 and an inductor LD1.

[0149] (2) The impedance element D2 is represented by, for example, a series circuit of a capacitor CD2 and an inductor LD2.

[0150] In the impedance element A1, the terminals of the impedance element are insulated from direct current by the presence of the capacitor CA1, so that the same PWM voltage generation as the inverter circuit involved in the comparative example 1 can be achieved. In addition, in the radiation noise band, the impedance of the capacitor CA1 is smaller than that of other elements, so it can be approximated as a short circuit. Therefore, in the radiation noise band, the impedance of the inductor LA1 is dominant. At this time, Fig.11 The equivalent circuit can be specifically expressed as Fig.12 Express it that way.

[0151] Fig.12 yes Fig.10 Equivalent circuit diagram of the radiation noise band of the power conversion circuit device. Fig.12 It is clear that the only component of the bridge circuit is the inductor, so it is possible to achieve a significant noise reduction effect in a wideband range by using the impedance balance method. In addition, the design is also made easier.

[0152] Furthermore, in order to make the impedance of the inductor LA1 dominant in the radiation noise band as described above, the series resonance frequency of the capacitor CA1 and the inductor LA1 needs to be designed to be equal to or lower than the radiation noise band.

[0153] Here, you can also Figure 4B to Figure 4D As shown in the figure, a resistance element connected in series or in parallel with the inductor is further added to the impedance elements A1, A2, B1, B2, D1, and D2. As a result, the impedance of the impedance element at the self-resonant frequency can be stabilized, and the peak of the noise can be avoided. That is, the noise reduction effect can be obtained in a wide band. In addition, as the impedance elements A1, A2, B1, B2, D1, and D2 including the inductor, the ferrite bead BD1 can also be used. In general, the equivalent circuit of the ferrite bead BD1 is, for example, a circuit obtained by connecting another resistor R2 in series to the parallel circuit of the inductor LA1 and the resistor R1 ( Figure 4E ) etc. As shown in this way, when the ferrite bead BD1 is used, since a resistance component is included therein, the noise reduction effect can be stably obtained without using a resistance element.

[0154] Next, conditions for maximizing the noise reduction effect are shown.

[0155] exist Fig.12In the equivalent circuit of , the bridge for output terminal T11 is first discussed. The upper left component of the bridge circuit for input terminal T11 is represented by LA2 / / (L01 / / L02+L11 / / L12). The inductance of the composite inductor is set to L1. The lower left component of the bridge circuit is LA1. The upper right component LB1 / / LD2 of the bridge circuit is set to L3a. The lower right component of the bridge circuit is L21. When focusing on the circuit on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+LA1)×VS12. The input terminals T01 and T02 are at ground potential in the radiation noise band, so the potential at nodes N01, N02, and N12 is L1 / (L1+LA1)×VS12.

[0156] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the upper right and lower right inductors of the bridge circuit. That is, the voltage applied to the upper right inductor of the bridge circuit is L3a / (L3a+L21)×VS12. When L1 / (L1+LA1)×VS12=L3a / (L3a+L21)×VS12, the potential of the output terminal T11 is 0V. The ground potential is 0V, which means that no common mode voltage is generated. Therefore, when this condition is met, it is expected to achieve a significant noise reduction effect.

[0157] exist Fig.12 Next, the bridge circuit for the output terminal T12 is discussed in the equivalent circuit.

[0158] The upper left and lower left components of the bridge circuit for input terminal T12 are the same as those of the bridge circuit for input terminal T11. The upper right component L22 / / LB2 of the bridge circuit is set to L3b. The lower right component of the bridge circuit is LD1. When focusing on the circuit on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+LA1)×VS12. T01 and T02 are at ground potential in the radiation noise band, so the potential at nodes N01 and N02 is L1 / (L1+LA1)×VS12.

[0159] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the upper right and lower right inductors of the bridge circuit. That is, the voltage applied to the upper right inductor is L3b / (L3b+LD1)×VS12. When L1 / (L1+LA1)×VS12=L3b / (L3b+LD1)×VS12, the potential of the output terminal T12 is 0V. The ground potential is 0V, which means that no common mode voltage is generated. Therefore, when this condition is met, it is expected to achieve a significant noise reduction effect.

[0160] Here, when the inductances L21 and L22 of the output wiring are equal to the inductances of the impedance elements D1 and D2 or the inductances LD1 and LD2 of the magnetic beads, the above two conditions are consistent. That is, due to the voltage source VS12, no common mode voltage is generated at the output terminal T11 and the output terminal T12. Similarly, due to the voltage source VS34, no common mode voltage is generated at the output terminal T11 and the output terminal T12. Embodiment 1 can suppress the transmission of noise from the switching elements S1 to S4 to the output terminal T11 or T12 of the current phase. In contrast, embodiment 2 can not only suppress the transmission of noise from the switching elements S1 to S4 to the output terminal T11 or T12 of the current phase, but also suppress the transmission of noise to the output terminal T12 or T11 of other phases.

[0161] In addition, the above-mentioned design conditional expression can be expressed in the form of an inductance ratio as L1:LA1=L3a:L21 or L1:LA1=L3b:LD1. This is an impedance balancing method. L01 / / L02+L11 / / L12 is the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the node N01 and the node N02 are short-circuited.

[0162] Therefore, the optimal conditions are as follows: Let the inductance of the impedance elements A1 and A2 or the inductance of the magnetic beads be L2, let the parallel inductance of the inductor L2 and the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the nodes N01 and N02 are short-circuited be L1, let the parallel inductance of the inductance of the inductor of the impedance element B1 or B2 and the inductance of the inductor of the impedance element D1 or D2 be L3, let the inductance of the inductor L21 and L22 be L4, in this case, L1:L2=L3:L4.

[0163] Although not as good as the above optimal conditions, noise reduction effects can be expected under the following conditions. Fig.12The voltage of the upper right component of the bridge circuit is in the range of 0.5 to 1.5 times the voltage applied to the upper left component of the bridge circuit. When it is 0.5 times, L1 / (L1+LA1)×VS12=0.5×L3a / (L3a+L21)×VS12. When it is 1.5 times, L1 / (L1+LA1)×VS12=1.5×L3a / (L3a+L21)×VS12. Therefore, in the formula of L1 / (L1+LA1)×VS12=a×L3a / (L3a+L21)×VS12, that is, in the formula of L1 / (L1+LA1)=a×L3a / (L3a+L21), when the coefficient a is 0.5≤a≤1.5 (when the values ​​of each component are set in such a way that the formula is established), it is expected to obtain a noise reduction effect.

[0164] Furthermore, the effect of the second embodiment is shown by circuit simulation. The circuit parameters are as follows: The parameters in parentheses are parasitic components considered in the series connection.

[0165] C1=1mF(10nH,10mΩ),

[0166] C2=100nF(1nH,10mΩ),

[0167] C3=C4=4.7nF (10nH, 0.2Ω),

[0168] L01=L02=L11=L12=0.5μH,

[0169] L21=L22=0.5μH.

[0170] In addition, the impedance elements A1, A2, B1, B2, D1, and D2 were all set to a series circuit of 0.5μH, 200pF, and 30Ω. A DC voltage of 282V was input between the input terminal T01 and the input terminal T02, and the switching frequency (carrier frequency) of the switching elements S1 to S4 was set to 12kHz. The common-mode current was compared when a 5m long motor cable (T-type equivalent circuit) and a motor (winding inductance 1μH, parasitic capacitance between winding and housing 0.5nF) were connected to the output terminals T11 and T12 of the inverter circuit.

[0171] Fig.13 It is shown Figure 1 and Fig.10 This is a spectrum diagram comparing the common-mode current suppression effects of power conversion circuit devices. Fig.13It is clear that a suppression effect of about 20 dB can be stably obtained in the radiation noise band. In particular, at about 50 MHz, the suppression effect is DP21 (= P01-P21), and at about 70 MHz, the suppression effect is DP22 (= P02-P22). Here, P01 and P02 are the peak values ​​of the common mode current of Comparative Example 1, and P21 and P22 are the peak values ​​of the common mode current of Implementation Example 2.

[0172] The above-mentioned embodiments or modified examples may be used in combination as appropriate. For example, the impedance elements A3, A4, B3, and B4 shown in the modified example of embodiment 1 may be added to embodiment 2. In this case, the same noise suppression effect can be obtained.

[0173] As mentioned above, in Fig.10 In the power conversion circuit device, it is preferable to set the inductance of the inductors L21 and L22 and the inductance included in the impedance elements D1 and D2 to be substantially equal to each other.

[0174] (Implementation method 3)

[0175] Fig.14 is a circuit diagram showing a configuration example of a power conversion circuit device according to Embodiment 3. Fig.14 In the embodiment, the power conversion circuit device includes a noise filter circuit 3, an inverter circuit 2, and a noise filter circuit 5.

[0176] exist Fig.14 In the embodiment, input terminals T01 and T02 of the inverter circuit 2 are connected to a DC power source 4 via a noise filter circuit 3. Output terminals T11 and T12 of the inverter circuit 2 are connected to an AC power source 6 (or an AC load) via a noise filter circuit 5.

[0177] The power conversion circuit device according to the third embodiment configured as described above is a single-phase DC-AC conversion circuit that performs power conversion between DC power and AC power in one direction or two directions. Therefore, by applying any of the above-mentioned embodiments as the inverter circuit 2, the AC side common mode current of the single-phase DC-AC conversion circuit can be suppressed. As a result, the radiation noise of the single-phase DC-AC conversion circuit can be reduced, and the cost can be reduced and the size and weight can be reduced by reducing the counter components.

[0178] (Implementation 4)

[0179] Fig.15 This is a circuit diagram showing a configuration example of a power conversion circuit according to a fourth embodiment. Fig.15 The power conversion circuit is configured to include an inverter circuit 2 , an isolation transformer 7 , and a rectifier circuit 8 .

[0180] exist Fig.15In the embodiment, output terminals T11 and T12 of the inverter circuit 2 are connected to the rectifier circuit 8 via the isolation transformer 7. Here, the rectifier circuit 8 has output terminals T21 and T22.

[0181] The power conversion circuit involved in the embodiment 4 configured as above is a DC-DC conversion circuit that converts a DC voltage unidirectionally or bidirectionally. Therefore, by applying any of the above embodiments to the inverter circuit 2, the common mode current of the DC-DC conversion circuit can be suppressed. As a result, the radiation noise of the DC-DC conversion circuit can be reduced, and the cost can be reduced and the size and weight can be reduced by reducing the counter components.

[0182] (Comparative Example 2)

[0183] Fig.16 : is a circuit diagram showing the structure of a power conversion circuit device related to a three-phase inverter circuit of Comparative Example 2. Fig.16 Power conversion circuit device and Figure 1 Compared with the power conversion circuit device of , it has the following differences.

[0184] (1) A control circuit 1A for a three-phase inverter circuit is provided instead of the control circuit 1 for a two-phase inverter circuit.

[0185] (2) The device further includes switching elements S5 and S6, an inductor L23, and an output terminal T13.

[0186] The differences are described below.

[0187] exist Fig.16 Node N01 is connected to node N02 via the drain and source of switching element S5 and the drain and source of switching element S6. Node N13, which is a connection point between the source of switching element S5 and the drain of switching element S6, is connected to output terminal T13 via inductor L23.

[0188] Here, a DC voltage of DC power is input between input terminal T01 and input terminal T02. Capacitors C1 and C2 are X capacitors, for example, C1 is a smoothing capacitor, and C2 is a buffer capacitor. Capacitors C3 and C4 are Y capacitors for noise suppression. Inductors L01, L02, L11, L12, L21 to L23 are routing inductors or choke coils.

[0189] The switch elements S1 to S6 are, for example, N-channel MOS transistors, and the switch elements S1 to S6 form a full-bridge circuit. The control circuit 1A generates known command signals SS1 to SS6 for performing PWM switching (three phases) and outputs them to the gates of the switch elements S1 to S6, respectively, thereby switching the switch elements S1 to S6 between on and off to generate desired PWM voltages at the output terminals T11, T12, and T13.

[0190] At this time, when common mode currents are generated in the inductors L21 to L23 and are transmitted from the output terminals T11 to T13 to, for example, the motor cables, the radiation noise increases. In order to suppress the common mode currents, measures such as installing ferrite cores in the motor cables are required, which not only generates additional costs but also makes the equipment larger and heavier.

[0191] The inventors of the present invention have proposed power conversion circuit devices according to the following embodiments and modifications in view of these findings to solve the problems.

[0192] (Implementation 5)

[0193] Fig.17 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the fifth embodiment. Fig.17 Power conversion circuit device and Fig.16 Compared with the power conversion circuit device of , it has the following differences.

[0194] (1) Impedance elements A1 to A3 and B1 to B3 are further provided.

[0195] The differences are described below.

[0196] exist Fig.17 In the embodiment, the node N11 is connected to the input terminal T02 via the impedance element A1, the node N12 is connected to the input terminal T02 via the impedance element A2, and the node N13 is connected to the input terminal T02 via the impedance element A3. In addition, the output terminal T11 is connected to the node N02 via the impedance element B1, the output terminal T12 is connected to the node N02 via the impedance element B2, and the output terminal T13 is connected to the node N02 via the impedance element B3.

[0197] According to the fifth embodiment configured as described above, in the frequency band of 30 MHz to 300 MHz (radiation noise band) where radiation noise becomes a problem, the impedance of the capacitors C1 to C4 is small and thus they can be approximated as a short circuit.

[0198] Fig.18 yes Fig.17Equivalent circuit diagram in the radiation noise band of the power conversion circuit device. Here, the noise transmission from the switching element pair S1 and S2 of the first phase is focused on, so the voltage of the switching element pair S1 and S2 is represented by the voltage source VS12, and the switching element pair S3 and S4 of the second phase and the switching element pair S5 and S6 of the third phase are considered to be short-circuited. This is because according to the superposition theorem, voltage sources other than the voltage source of focus can be considered to be short-circuited.

[0199] according to Fig.18 It is clear that the structure of the bridge circuit is formed by adding impedance elements A1 and B1 from the voltage source VS12 to the output terminal T11 of the first phase. Therefore, although the detailed principle will be described later, the impedance balancing method can be used to reduce the noise transmission from the voltage source VS12 to the output terminal T11. As a result, the common mode current transmitted in the wiring extending from the output terminal T11 can be suppressed, the radiation noise can be reduced, and the cost reduction and small size and light weight can be achieved by reducing the counter components.

[0200] As a method of realizing impedance elements A1 to A3 and B1 to B3, the series circuits of inductors and capacitors are discussed respectively. For example, if the impedance element A1 is shown in a circuit diagram, then Figure 4A The impedance elements A2, A3, B1 to B3 are also represented in the same way. Since the capacitor CA1 is present between the terminals of the impedance element, direct current is insulated, so the generation of PWM voltage similar to that of the inverter circuit involved in Comparative Example 2 can be achieved. In addition, in the radiation noise band, the impedance of the capacitor CA1 is sufficiently small compared to other elements, so it can be approximated as a short circuit. Therefore, in the radiation noise band, the impedance of the inductor LA1 is dominant. At this time, Fig.18 The equivalent circuit can be specifically expressed as Fig.19 Express it that way.

[0201] Fig.19 yes Fig.17 Equivalent circuit diagram of the radiation noise band of the power conversion circuit device. Fig.19 It is clear that the only component of the bridge circuit is the inductor, so it is possible to achieve a significant noise reduction effect in a wideband range by using the impedance balance method. In addition, the design is also made easier.

[0202] Furthermore, in order to make the impedance of the inductor LA1 dominant in the radiation noise band as described above, the series resonance frequency of the capacitor CA1 and the inductor LA1 needs to be designed to be equal to or lower than the radiation noise band.

[0203] Here, you can also Figure 4B or Figure 4CAs shown, a resistor element is further added to the impedance elements A1 to A3, B1 to B3 in series or in parallel with the inductor. This stabilizes the impedance of the impedance element at the self-resonant frequency and avoids the generation of noise peaks. In other words, the noise reduction effect can be obtained in a wideband domain. In addition, as the impedance elements A1 to A3, B1 to B3 including the inductor, the ferrite bead BD1 can also be used. In general, the equivalent circuit of the ferrite bead BD1 is, for example, a circuit obtained by connecting another resistor R2 in series to the parallel circuit of the inductor LA1 and the resistor R1 ( Figure 4E ) etc. As shown in this way, when the ferrite bead BD1 is used, since a resistance component is included therein, the noise reduction effect can be stably obtained without using a resistance element.

[0204] Next, the conditions for maximizing the noise reduction effect are shown. Fig.19 In the equivalent circuit, LA2 / / LA3 / / (L01 / / L02+L11 / / L12) is used to represent the upper left component of the bridge circuit for the output terminal T11. The composite inductance is set to L1. The lower left, upper right and lower right components of the bridge circuit are inductors LA1, LB1, and L21, respectively. When focusing on the circuit on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+LA1)×VS12. The input terminals T01 and T02 are at ground potential in the radiation noise band, so the potential at the nodes N01 and N02 is L1 / (L1+LA1)×VS12.

[0205] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the upper right and lower right inductors of the bridge circuit. That is, the voltage applied to the upper right inductor of the bridge circuit is LB1 / (LB1+L21)×VS12. Here, when L1 / (L1+LA1)×VS12=LB1 / (LB1+L21)×VS12, the potential of the output terminal T11 is 0V. The ground potential is 0V, which means that no common mode voltage is generated. Therefore, when this condition is met, a significant noise reduction effect can be expected.

[0206] In addition, the above-mentioned design conditional expression can be expressed in the form of an inductance ratio as L1:LA1=LB1:L21. This is the impedance balance method. L01 / / L02+L11 / / L12 is the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the node N01 and the node N02 are short-circuited. Therefore, if the above-mentioned optimal conditions are explained, it is as follows. Let the inductance of impedance elements A1 to A3 or the inductance of magnetic beads be L2, let the parallel inductance of the above L2 / 2 and the effective inductance between input terminal T01 and node N01 when the input terminal T01 and input terminal T02 are short-circuited and the node N01 and node N02 are short-circuited be L1, let the inductance of impedance elements B1 to B3 or the inductance of magnetic beads be L3, let the inductance of inductors L21 to L23 be L4, at this time, L1:L2=L3:L4.

[0207] Although not as good as the above optimal conditions, noise reduction effects can be expected under the following conditions. Fig.19 The voltage of the upper right component of the bridge circuit is, for example, in the range of 0.5 to 1.5 times the voltage applied to the upper left component of the bridge circuit. At 0.5 times, L1 / (L1+LA1)×VS12=0.5×LB1 / (LB1+L21)×VS12. At 1.5 times, L1 / (L1+LA1)×VS12=1.5×LB1 / (LB1+L21)×VS12. Therefore, in the formula of L1 / (L1+LA1)×VS12=a×LB1 / (LB1+L21)×VS12, that is, in the formula of L1 / (L1+LA1)=a×LB1 / (LB1+L21), when the coefficient a is 0.5≤a≤1.5 (when the values ​​of each component are set in such a way that the formula is established), a noise reduction effect can be expected.

[0208] The effect of Embodiment 5 is shown by circuit simulation. The circuit parameters are as follows: The parasitic components considered in the series connection are in brackets.

[0209] C1=1mF(10nH,10mΩ),

[0210] C2=100nF(1nH,10mΩ),

[0211] C3=C4=4.7nF (10nH, 0.2Ω),

[0212] L01=L02=L11=L12=0.5μH,

[0213] L21=L22=0.5μH.

[0214] In addition, impedance elements A1 and A2 were set to a series circuit of 0.5μH, 200pF and 30Ω, and B1 and B2 were set to a series circuit of 0.167μH, 200pF and 30Ω. A DC voltage of 282V was input between input terminal T01 and input terminal T02, and the switching frequency (carrier frequency) of switching elements S1 to S6 was set to 12kHz. The common-mode current was compared when a 5m long motor cable (T-type equivalent circuit) and a motor as a load (winding inductance of 1μH, parasitic capacitance between winding and housing of 0.5nF) were connected to the output terminals T11 to T13 of the inverter circuit.

[0215] Fig. 20 It is shown Fig.16 and Fig.17 This is a spectrum diagram comparing the common-mode current suppression effects of power conversion circuit devices. Fig. 20 It is clear that a suppression effect of 10dB to 20dB can be confirmed at the noise peak that is prone to cause problems in the radiation noise band. In particular, at about 50MHz, the suppression effect is DP51 (= P03-P51), and at about 70MHz, the suppression effect is DP52 (= P03-P52). Here, P03 and P04 are the peak values ​​of the common mode current of Comparative Example 2, and P51 and P52 are the peak values ​​of the common mode current of Implementation Example 5.

[0216] (Implementation 6)

[0217] Fig.21 This is a circuit diagram showing a configuration example of a power conversion circuit device according to the sixth embodiment. Fig.21 Power conversion circuit device and Fig.17 Compared with the power conversion circuit device of , it has the following differences.

[0218] (1) Impedance elements D1 to D6 are further provided.

[0219] The differences are described below.

[0220] exist Fig.21 In the embodiment, the node N11 is connected to the output terminal T13 via the impedance element D1, the node N11 is connected to the output terminal T12 via the impedance element D2, and the node N12 is connected to the output terminal T13 via the impedance element D3. In addition, the node N12 is connected to the output terminal T11 via the impedance element D4, the node N13 is connected to the output terminal T11 via the impedance element D5, and the node N13 is connected to the output terminal T12 via the impedance element D6.

[0221] According to the sixth embodiment configured as described above, a greater noise reduction effect can be obtained compared with the fifth embodiment. The principle thereof will be described below.

[0222] In the radiation noise band, the impedance of capacitors C1 to C4 is sufficiently small compared to other components, so they can be approximated as short circuits. Fig.21 Since the focus is on the noise transmission from the first-phase switch element pair S1 and S2, the voltage of the switch element pair S1 and S2 is represented by the voltage source VS12, and the second-phase switch element pair S3 and S4 and the third-phase switch element pair S5 and S6 are regarded as short-circuited. At this time, the overall equivalent circuit becomes complicated, so Fig. 22 The equivalent circuit (bridge circuit) for the output terminal T11 is shown in FIG. Fig.23 . ...

[0223] Fig. 22 and Fig.23 They are for output terminals T11 and T12 respectively. Fig.21 The equivalent circuit diagram in the radiation noise band of the power conversion circuit device of FIG. In addition, the equivalent circuit for the output terminal T13 is the same as the equivalent circuit for the output terminal T12. That is, it is classified into an equivalent circuit for the output terminal of the current phase and an equivalent circuit for the output terminal of the other phase.

[0224] according to Fig. 22 It is clear that the voltage source VS12 to the output terminal T11 of the first phase is formed into a bridge circuit structure by adding impedance elements A1 and B1. Therefore, the noise transmission from the voltage source VS12 to the output terminal T11 can be reduced by using the impedance balancing method. As a result, the common mode current transmitted in the wiring extending from the output terminal T11 can be suppressed, the radiation noise can be reduced, and the cost reduction and compactness and lightness brought about by the reduction of countermeasure components can be achieved.

[0225] And, according to Fig.23 It is clear that the voltage source VS12 to the output terminal T12 of the second phase is also configured as a bridge circuit by adding impedance elements B2 and D2. Therefore, the impedance balancing method can also be used to reduce the noise transmission from the voltage source VS12 to the output terminal T12. As a result, the common mode current transmitted before the output terminal can be further suppressed, the radiation noise can be further reduced, and the cost reduction and small size and light weight can be achieved by reducing the counter components.

[0226] As a method of realizing the impedance elements A1 to A3, B1 to B3, and D1 to D6, the series circuits of inductors and capacitors are discussed respectively. For example, if the impedance element A1 is shown in a circuit diagram, for example, Figure 4A The impedance elements A2, A3, B1 to B3, and D1 to D6 are also represented in the same manner.

[0227] (1) The impedance element D3 is represented by, for example, a series circuit of a capacitor CD3 and an inductor LD3.

[0228] (2) The impedance element D4 is represented by, for example, a series circuit of a capacitor CD4 and an inductor LD4.

[0229] In the impedance element A1, the terminals of the impedance element are insulated from direct current by the presence of the capacitor CA1, so that the same PWM voltage generation as the inverter circuit involved in the comparative example 2 can be achieved. In addition, in the radiation noise band, the impedance of the capacitor CA1 is small, so it can be approximated as a short circuit. Therefore, in the radiation noise band, the impedance of the inductor LA1 is dominant. At this time, Fig. 22 and Fig.23 The equivalent circuits of Fig.24 and Fig.25 Express it that way.

[0230] Fig.24 and Fig.25 The equivalent circuit diagrams of the power conversion circuit device in the radiation noise band for the output terminals T11 and T12 are shown in FIG. Fig.24 and Fig.25 It is clear that the only component of the bridge circuit is the inductor, so it is possible to achieve a significant noise reduction effect in a wideband range by using the impedance balance method. In addition, the design is also made easier.

[0231] Furthermore, in order to make the impedance of the inductor LA1 dominant in the radiation noise band as described above, the series resonance frequency of the capacitor CA1 and the inductor LA1 needs to be designed to be equal to or lower than the radiation noise band.

[0232] Here, you can also Figure 4B or Figure 4C As shown in the figure, a resistor element connected in series or in parallel with the inductor is further added to the impedance elements A1 to A3, B1 to B3, D1 to D6. This can stabilize the impedance at the self-resonant frequency of the impedance element and avoid the peak of noise. In other words, the noise reduction effect can be obtained in a wide band. In addition, as the impedance elements A1 to A3, B1 to B3, D1 to D6 including the inductor, Figure 4D Generally speaking, the equivalent circuit of the ferrite bead BD1 is a circuit obtained by connecting another resistor R2 in series to the parallel circuit of the inductor LA1 and the resistor R1 ( Figure 4E ) etc. As shown in this way, when the ferrite bead BD1 is used, since a resistance component is included therein, the noise reduction effect can be stably obtained without using a resistance element.

[0233] Next, conditions for maximizing the noise reduction effect are shown.

[0234] First, discuss Fig.24 The equivalent circuit of the bridge for the output terminal T11. The upper left component of the bridge circuit for T11 is represented by LA2 / / LA3 / / (L01 / / L02+L11 / / L12). The synthetic inductance is set to L1. The lower left component of the bridge circuit is LA1. The upper right component of the bridge circuit LB1 / / LD4 / / LD5 is set to L3a. The lower right component of the bridge circuit is L21. When focusing on the circuit on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+LA1)×VS12. The input terminals T01 and T02 are at ground potential in the radiation noise band, so the potential at the nodes N01 and N02 is L1 / (L1+LA1)×VS12.

[0235] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the upper right and lower right inductors of the bridge circuit. That is, the voltage applied to the upper right inductor of the bridge circuit is L3a / (L3a+L21)×VS12. When L1 / (L1+LA1)×VS12=L3a / (L3a+L21)×VS12, the potential of the output terminal T11 is 0V. The ground potential is 0V, which means that no common mode voltage is generated. Therefore, when this condition is met, it is expected to achieve a significant noise reduction effect.

[0236] Next, discuss Fig.25 The equivalent circuit of the bridge for output terminal T12. The upper left and lower left components of the bridge circuit for T12 are the same as those of the bridge circuit for T11. The upper right component L22 / / LB2 / / LD6 of the bridge circuit is set to L3b. The lower right component of the bridge circuit is LD2. When focusing on the circuit on the left side of the bridge circuit, the voltage source VS12 is divided by the upper left and lower left inductors of the bridge circuit. That is, the voltage applied to the upper left inductor of the bridge circuit is L1 / (L1+LA1)×VS12. T01 and T02 are at ground potential in the radiation noise band, so the potential at N01 and N02 is L1 / (L1+LA1)×VS12.

[0237] Next, when focusing on the circuit on the right side of the bridge circuit, the voltage source VS12 is divided by the upper right and lower right inductors of the bridge circuit. That is, the voltage applied to the upper right inductor of the bridge circuit is L3b / (L3b+LD2)×VS12. When L1 / (L1+LA1)×VS12=L3b / (L3b+LD2)×VS12, the potential of the output terminal T12 is 0V. The ground potential is 0V, which means that no common mode voltage is generated. Therefore, when this condition is met, it is expected to achieve a significant noise reduction effect.

[0238] Here, when the inductance L21 to L23 of the output wiring is equal to the inductance of the impedance elements D1 to D6 or the inductance LD1 to LD6 of the magnetic beads, the above two conditions are consistent. That is, due to the voltage source VS12, no common mode voltage is generated at the output terminals T11 and T12 (and T13). Similarly, due to the voltage source VS34, no common mode voltage is generated at the output terminals T11 and T12, and no common mode voltage is generated at the output terminal T13. The same is true for the voltage source VS56. Implementation 5 can suppress the transmission of noise from the switching element to the output terminal of the current phase. In contrast, implementation 6 can not only suppress the transmission of noise from the switching element to the current phase, but also suppress the transmission of noise to the output terminals of other phases.

[0239] In addition, the above-mentioned design conditional expression can be expressed in the form of an inductance ratio as L1:LA1=L3a:L21 or L1:LA1=L3b:LD2. This is an impedance balancing method. L01 / / L02+L11 / / L12 is the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the node N01 and the node N02 are short-circuited.

[0240] Therefore, the above-mentioned optimal conditions are described as follows: Let the inductance of the impedance elements A1 to A3 or the inductance of the magnetic beads be L2, let L1 be the parallel inductance of L2 / 2 and the effective inductance between the input terminal T01 and the node N01 when the input terminal T01 and the input terminal T02 are short-circuited and the node N01 and the node N02 are short-circuited, let L3 be the parallel inductance of half the inductance of the inductors of the impedance elements D1 to D6 and the inductance of the inductors of the impedance elements B1 to B3, let L4 be the inductance of the inductors L21 to L23, in this case, L1:L2=L3:L4.

[0241] Here, the voltage applied to the upper right component of the bridge circuit is discussed in the range of, for example, 0.5 to 1.5 times the voltage applied to the upper left component of the bridge circuit. When it is 0.5 times, L1 / (L1+LA1)×VS12=0.5×L3a / (L3a+L21)×VS12. When it is 1.5 times, L1 / (L1+LA1)×VS12=1.5×L3a / (L3a+L21)×VS12. Therefore, in the formula of L1 / (L1+LA1)×VS12=a×L3a / (L3a+L21)×VS12, that is, in the formula of L1 / (L1+LA1)=a×L3a / (L3a+L21), when the coefficient a is 0.5≤a≤1.5 (when the values ​​of each component are set so that the formula is established), it is expected that a noise reduction effect can be obtained.

[0242] Circuit simulation is used to illustrate the effect of Embodiment 6. The circuit parameters are as follows: The parameters in parentheses are parasitic components considered in the series connection.

[0243] C1=1mF(10nH,10mΩ),

[0244] C2=100nF(1nH,10mΩ),

[0245] C3=C4=4.7nF (10nH, 0.2Ω),

[0246] L01=L02=L11=L12=0.5μH,

[0247] L21=L22=0.5μH.

[0248] In addition, the impedance elements A1 to A3, B1 to B3, and D1 to D6 are all set to a series circuit of 0.5μH, 200pF, and 30Ω. A DC voltage of 282V is input between the input terminal T01 and the input terminal T02, and the switching frequency (carrier frequency) of the switching elements S1 to S6 is set to 12kHz. The common-mode current is compared when a 5m long motor cable (T-type equivalent circuit) and a motor as a load (winding inductance is 1μH, and the parasitic capacitance between the winding and the housing is 0.5nF) are connected to the output terminals T11 to T13 of the inverter circuit.

[0249] Fig.26 It is shown Fig.16 and Fig.21 This is a spectrum diagram comparing the common-mode current suppression effects of power conversion circuit devices. Fig.26 It is clear that a suppression effect of about 20 dB can be stably obtained in the radiation noise band. In particular, at about 50 MHz, the suppression effect is DP61 (= P03-P61), and at about 70 MHz, the suppression effect is DP62 (= P03-P62). Here, P03 and P04 are the peak values ​​of the common mode current of Comparative Example 2, and P61 and P62 are the peak values ​​of the common mode current of Implementation Example 6.

[0250] (Implementation 7)

[0251] Fig. 27 is a circuit diagram showing a configuration example of a power conversion circuit device according to Embodiment 7. Fig. 27 In the embodiment, the power conversion circuit device is configured to include a noise filter circuit 3 and an inverter circuit 2.

[0252] exist Fig. 27In the embodiment, input terminals T01 and T02 of the inverter circuit 2 are connected to a DC power supply 4 via a noise filter circuit 3. Output terminals T11 to T13 of the inverter circuit 2 are connected to a three-phase motor 9 as a load example.

[0253] The power conversion circuit device involved in Embodiment 7 is a three-phase motor inverter device that uses DC power to drive a three-phase motor. Therefore, by applying any of the above-mentioned embodiments as the inverter circuit 2, the motor-side common-mode current of the three-phase motor inverter device can be suppressed. As a result, the radiation noise of the three-phase motor inverter device can be reduced, and the cost can be reduced and the size and weight can be reduced by reducing the counter components.

[0254] In embodiment 7, the DC power supply 4 may also be composed of an AC power supply and a rectifier circuit. In this case, the connection order of the rectifier circuit and the noise filtering circuit 3 may also be exchanged. That is, the DC power supply 4 may be connected to the input terminal of the noise filtering circuit 3, the output terminal of the noise filtering circuit 3 may be connected to the input terminal of the rectifier circuit, and the output circuit of the rectifier circuit may be connected to T01 and T02. In this case, it is a three-phase motor inverter device that uses AC power to drive a three-phase motor. Therefore, by applying any of the above-mentioned embodiments as the inverter circuit 2, the common mode current on the motor side of the three-phase motor inverter device using AC power can be suppressed. As a result, the radiation noise of the three-phase motor inverter device using AC power can be reduced, and low cost and compactness and lightness can be achieved by reducing counteracting components.

[0255] (Other Modifications)

[0256] In the above embodiment, the ferrite beads BD1 are used in the modified example, but the present disclosure is not limited thereto, and other types of beads such as chip beads may be used.

[0257] Industrial Applicability

[0258] The power conversion circuit device according to the present disclosure is useful for realizing a power conversion circuit device used in vehicle-mounted equipment, industrial equipment, etc. in a low-noise, small-sized, and low-cost manner.

[0259] Description of Reference Numerals

[0260] 1. 1A: control circuit; 2: inverter circuit; 3. 5: noise filtering circuit; 4: DC power supply; 6: AC power supply; 7: insulation transformer; 8: rectifier circuit; 9: motor; A1~A4, B1~B4, D1~D6: impedance elements; BD1: ferrite bead; C1~C4, CA1: capacitors; L01~L23, LA1~LA4, LB1~LB4, LD1~LD6: inductors; N01~N13: nodes; R1~R2: resistors; S1~S6: switching elements; T01~T02: input terminals; T11~T13, T21~T22: output terminals; VS12: voltage source.

Claims

1. A power conversion circuit device, comprising an inverter circuit including a bridge circuit between first and second input terminals and first and second output terminals, wherein the bridge circuit is formed by connecting a first switching element and a second switching element connected in series with each other and a third switching element and a fourth switching element connected in series with each other in parallel, wherein: Two ends of the series circuit of the first switch element and the second switch element, and two ends of the series circuit of the third switch element and the fourth switch element are defined as a first connection point and a second connection point, respectively, and the first connection point is connected to the first input terminal via a first inductor, and the second connection point is connected to the second input terminal via a second inductor, A third connection point between the first switch element and the second switch element is connected to the first output terminal via a third inductor. A fourth connection point between the third switch element and the fourth switch element is connected to the second output terminal via a fourth inductor. The power conversion circuit device comprises: a first impedance element connected between the third connection point and the first input terminal or the second input terminal; a second impedance element connected between the fourth connection point and the first input terminal or the second input terminal; a third impedance element connected between the first output terminal and the first input terminal or the second input terminal; as well as A fourth impedance element is connected between the second output terminal and the first input terminal or the second input terminal.

2. The power conversion circuit device according to claim 1, wherein: The first impedance element to the fourth impedance element are any of the following: (1) Series circuit of inductor and capacitor; (2) a circuit further comprising a resistor connected in series or in parallel with the inductor in the series circuit; and (3) Series circuit of magnetic beads and capacitors.

3. The power conversion circuit device according to claim 1 or 2, wherein: The inductance included in the first impedance element and the second impedance element is set to L2. The parallel inductance of the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The inductance included in the third impedance element and the fourth impedance element is set to L3. The third inductor and the fourth inductor are set to L4, At this time, the inductors L1 to L4 are set so that L1:L2=L3:L4.

4. The power conversion circuit device according to claim 1 or 2, wherein: The inductance included in the first impedance element is set to L2, The parallel inductance of the inductance L2 and the effective inductance between the first input terminal and the third connection point when the first input terminal and the second input terminal are short-circuited and the third connection point and the fourth connection point are short-circuited is set to L1. The inductance included in the third impedance element is set to L3, The third inductor and the fourth inductor are set to L4, The inductances L1 to L4 are set so that the coefficient a in the equation (1) L1 / (L1+L2)=a×L3 / (L3+L4) satisfies 0.5≤a≤1.

5.

5. The power conversion circuit device according to claim 1, further comprising: a fifth impedance element connected between the first input terminal and the third connection point; a sixth impedance element connected between the first input terminal and the fourth connection point; a seventh impedance element connected between the first connection point and the first output terminal; and An eighth impedance element is connected between the first connection point and the second output terminal.

6. The power conversion circuit device according to claim 5, wherein: The impedances of the first impedance element, the second impedance element, the fifth impedance element, and the sixth impedance element are substantially the same, and the impedances of the third impedance element, the fourth impedance element, the seventh impedance element, and the eighth impedance element are substantially the same.

7. The power conversion circuit device according to claim 5 or 6, wherein: The inductance included in the first impedance element, the second impedance element, the fifth impedance element, and the sixth impedance element is set to 2×L2, The parallel inductance of the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The inductance included in the third impedance element, the fourth impedance element, the seventh impedance element, and the eighth impedance element is set to 2×L3, The third inductor and the fourth inductor are set to L4, At this time, the inductors L1 to L4 are set so that L1:L2=L3:L4.

8. The power conversion circuit device according to claim 5 or 6, wherein: The inductance included in the first impedance element, the second impedance element, the fifth impedance element, and the sixth impedance element is set to 2×L2, The parallel inductance of the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The inductance included in the third impedance element, the fourth impedance element, the seventh impedance element, and the eighth impedance element is set to 2×L3, The third inductor and the fourth inductor are set to L4, At this time, the inductances L1 to L4 are set so that the coefficient a in the equation L1 / (L1+L2)=a×L3 / (L3+L4) satisfies 0.5≤a≤1.

5.

9. The power conversion circuit device according to claim 1, further comprising: a ninth impedance element connected between the third connection point and the second output terminal; and A tenth impedance element is connected between the fourth connection point and the first output terminal.

10. The power conversion circuit device according to claim 9, wherein: The ninth impedance element and the tenth impedance element are respectively any of the following: (1) Series circuit of inductor and capacitor; (2) a circuit further comprising a resistor connected in series or in parallel with the inductor in the series circuit; and (3) Series circuit of magnetic beads and capacitors.

11. The power conversion circuit device according to claim 9 or 10, wherein: The third inductor and the fourth inductor are substantially the same as the inductors included in the ninth impedance element and the tenth impedance element.

12. The power conversion circuit device according to claim 9 or 10, wherein: The inductance included in the first impedance element and the second impedance element is set to L2. The parallel inductance of the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The parallel inductance of the inductance included in the third impedance element and the fourth impedance element and the inductance included in the ninth impedance element and the tenth impedance element is defined as L3. The third inductor and the fourth inductor are set to L4, At this time, the inductors L1 to L4 are set so that L1:L2=L3:L4.

13. The power conversion circuit device according to claim 9 or 10, wherein: The inductance included in the first impedance element and the second impedance element is set to L2. The parallel inductance of the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The parallel inductance of the inductance included in the third impedance element and the fourth impedance element and the inductance included in the ninth impedance element and the tenth impedance element is defined as L3. The third inductor and the fourth inductor are set to L4, At this time, the inductances L1 to L4 are set so that the coefficient a in the equation L1 / (L1+L2)=a×L3 / (L3+L4) satisfies 0.5≤a≤1.

5.

14. The power conversion circuit device according to any one of claims 1, 5 and 9, further comprising: A first noise filtering circuit connected to the first input terminal and the second input terminal of the power conversion circuit device; and The second noise filtering circuit is connected to the first output terminal and the second output terminal of the power conversion circuit device.

15. The power conversion circuit device according to any one of claims 1, 5 and 9, further comprising: an isolation transformer having an input terminal connected to the first output terminal and the second output terminal of the power conversion circuit device; and A rectifier circuit is connected to the output terminal of the isolation transformer.

16. A power conversion circuit device, comprising an inverter circuit including a bridge circuit between first and second input terminals and first and second output terminals, the bridge circuit being formed by connecting in parallel a first switching element and a second switching element connected in series, a third switching element and a fourth switching element connected in series, and a fifth switching element and a sixth switching element connected in series, wherein: Two ends of the series circuit of the first switch element and the second switch element, two ends of the series circuit of the third switch element and the fourth switch element, and two ends of the fifth switch element and the sixth switch element are respectively defined as a first connection point and a second connection point, and the first connection point is connected to the first input terminal via a first inductor, and the second connection point is connected to the second input terminal via a second inductor, A third connection point between the first switch element and the second switch element is connected to the first output terminal via a third inductor. A fourth connection point between the third switch element and the fourth switch element is connected to the second output terminal via a fourth inductor. A fifth connection point between the fifth switch element and the sixth switch element is connected to the third output terminal via a fifth inductor. The power conversion circuit device comprises: a first impedance element connected between the third connection point and the first input terminal or the second input terminal; a second impedance element connected between the fourth connection point and the first input terminal or the second input terminal; a third impedance element connected between the fifth connection point and the first input terminal or the second input terminal; a fourth impedance element connected between the first output terminal and the first input terminal or the second input terminal; a fifth impedance element connected between the second output terminal and the first input terminal or the second input terminal; as well as A sixth impedance element is connected between the third output terminal and the first input terminal or the second input terminal.

17. The power conversion circuit device according to claim 16, wherein: The first impedance element to the sixth impedance element are any of the following: (1) Series circuit of inductor and capacitor; (2) a circuit further comprising a resistor connected in series or in parallel with the inductor in the series circuit; and (3) Series circuit of magnetic beads and capacitors.

18. The power conversion circuit device according to claim 16 or 17, wherein: The inductance included in the first to third impedance elements is set to L2, The parallel inductance of the inductance of half the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The inductance included in the fourth to sixth impedance elements is set to L3, The third to fifth inductors are set as L4, At this time, the inductors L1 to L4 are set so that L1:L2=L3:L4.

19. The power conversion circuit device according to claim 16 or 17, wherein: The inductance included in the first to third impedance elements is set to L2, The parallel inductance of the inductance of half the inductance L2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The inductance included in the first to third impedance elements is set to L2, The inductance included in the fourth to sixth impedance elements is set to L3, The third to fifth inductors are set as L4, At this time, the inductances L1 to L4 are set so that the coefficient a in the equation L1 / (L1+L2)=a×L3 / (L3+L4) satisfies 0.5≤a≤1.

5.

20. The power conversion circuit device according to claim 16, wherein: The power conversion circuit device further comprises: a seventh impedance element connected between the third connection point and the third output terminal; an eighth impedance element connected between the third connection point and the second output terminal; a ninth impedance element connected between the fourth connection point and the third output terminal; a tenth impedance element connected between the fourth connection point and the first output terminal; an eleventh impedance element connected between the fifth connection point and the first output terminal; and A twelfth impedance element is connected between the fifth connection point and the second output terminal.

21. The power conversion circuit device according to claim 20, wherein: The seventh to twelfth impedance elements are any of the following: (1) Series circuit of inductor and capacitor; (2) a circuit further comprising a resistor connected in series or in parallel with the inductor in the series circuit; and (3) Series circuit of magnetic beads and capacitors.

22. The power conversion circuit device according to claim 20 or 21, wherein: The inductances included in the third to fifth inductors and the seventh to twelfth impedance elements are substantially the same as each other.

23. The power conversion circuit device according to claim 20 or 21, wherein: The inductance included in the first to third impedance elements is set to L2, The parallel inductance of L2 / 2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The parallel inductance of the inductance included in the fourth to sixth impedance elements and the inductance half of the inductance included in the seventh to twelfth impedance elements is defined as L3. The third to fifth inductors are set as L4, At this time, the inductors L1 to L4 are set so that L1:L2=L3:L4.

24. The power conversion circuit device according to claim 20 or 21, wherein: The inductance included in the first to third impedance elements is set to L2, The parallel inductance of L2 / 2 and the effective inductance between the first input terminal and the first connection point when the first input terminal and the second input terminal are short-circuited and the first connection point and the second connection point are short-circuited is set to L1. The parallel inductance of the inductance included in the fourth to sixth impedance elements and the inductance half of the inductance included in the seventh to twelfth impedance elements is defined as L3. The third to fifth inductors are set as L4, At this time, the inductances L1 to L4 are set so that the coefficient a in the equation L1 / (L1+L2)=a×L3 / (L3+L4) satisfies 0.5≤a≤1.

5.

25. A power conversion circuit device comprising: The power conversion circuit device according to claim 16 or 20; a noise filtering circuit connected to an input terminal of the power conversion circuit device; and A three-phase motor is connected to the output terminal of the power conversion circuit device.

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