Two-channel wireless energy supply system and communication-free power control method thereof

Abstract

The invention discloses a dual-channel wireless energy supply system and a communication-free power control method thereof, and relates to the technical field of wireless power transmission, the system comprises a primary side transmitting end and a secondary side receiving end, the primary side transmitting end and the secondary side receiving end jointly construct a fundamental wave main power channel CH1 and a third harmonic channel CH2, energy and signal coupling transmission is realized between the primary side transmitting end and the secondary side receiving end through a dual-channel magnetic coupling structure. According to the method, the phase of third harmonic current is adjusted through an active rectification circuit of a secondary side receiving end, the load state is transmitted to a primary side transmitting end in a phase deviation mode, and after the primary side transmitting end detects phase information, the duty ratio of a full-bridge inverter is adjusted through a closed loop, and self-adaptive control over the output power of the system is achieved. According to the invention, non-communication transmission of a load state is completed by using a harmonic channel, closed-loop adjustment is carried out and the power range of the ZVS is expanded in combination with phase information, and stable and efficient wireless power transmission control is realized.

Description

Technical Field

[0001] The present invention relates to the technical field of wireless power transfer, and particularly to a dual-channel wireless power supply system and a communication-free power control method therefor. Background Art

[0002] Wireless Power Transfer (WPT) technology has broad application prospects in the fields of electric vehicle charging, industrial automation, consumer electronics, etc. Especially in the field of electric vehicles, wireless charging can eliminate mechanical contact and improve charging convenience and system durability, and is becoming an important development direction. However, to achieve reliable power control, most existing wireless power transfer systems still rely on a communication link between the primary side (transmitter) and the secondary side (receiver), such as wireless communication methods like Bluetooth, Wi-Fi, radio frequency identification, etc., for transmitting load information or feedback power commands. Such communication methods are vulnerable to interference in a complex electromagnetic environment, resulting in control failure; at the same time, their communication modules increase the hardware cost and control complexity of the system. In application scenarios that require high reliability and low system complexity, such as rail transit, industrial mobile devices, etc., it poses limitations to the existing communication-dependent WPT control schemes.

[0003] On the other hand, existing WPT systems usually use a fundamental wave resonance channel for main power transmission, but there are significant challenges in achieving zero voltage switching (ZVS) of a full-bridge inverter in the medium and low power range. If ZVS cannot be achieved, it will significantly increase the switching loss of the converter and reduce the overall efficiency and lifespan of the system. In addition, in application scenarios such as electric vehicles, the system also needs to face non-ideal conditions such as dynamic changes in the battery load impedance and coil position offset, further exacerbating the adaptability problem of the control system.

[0004] To solve the above problems, existing research has proposed power regulation ideas without communication, but most methods still rely on inferring the load state based on voltage or current amplitude, with insufficient robustness and regulation accuracy. At the same time, there are still technical bottlenecks in switch optimization and low-power operation stability.

[0005] Therefore, there is an urgent need for a control method that does not require communication, has high dynamic adaptability and low-complexity implementation, and can achieve power regulation and switch optimization through a novel feedback mechanism without changing the main topology of the system. Summary of the Invention

[0006] To solve the problems of the existing wireless power transmission system, such as high dependence on the communication module, complex power regulation method, poor system stability, and difficulty in achieving zero-voltage switching of the inverter in the low-power range, the present invention proposes a dual-channel wireless power supply system and its communication-free power control method, which can achieve stable and reliable power regulation without communication equipment, while expanding the working range of zero-voltage switching of the inverter, improving system efficiency and load adaptability, so as to solve the above problems.

[0007] This application discloses a dual-channel wireless power supply system, including a primary-side transmitter and a secondary-side receiver; The primary-side transmitter includes a full-bridge inverter composed of a first bridge arm H1 and a second bridge arm H2. A DC voltage is connected in parallel on the input side of the full-bridge inverter and a first capacitor . The midpoint of the first bridge arm H1 is connected to the midpoint of the second bridge arm H2 by a primary fundamental main power channel, and the midpoint of the first bridge arm H1 is connected to the reference ground g by a primary third-harmonic channel; The secondary-side receiver includes a diode rectifier half-bridge H3 and an active rectifier half-bridge H4 connected in parallel. A series circuit of a second capacitor and a third capacitor is connected in parallel on the output side of the secondary-side receiver. The midpoint of the diode rectifier half-bridge H3 is connected to the midpoint of the series circuit of the second capacitor , the third capacitor by a secondary fundamental main power channel, and the midpoint of the active rectifier half-bridge H4 is connected to the midpoint of the series circuit of the second capacitor , the third capacitor , the third capacitor by a secondary third-harmonic channel; Energy and signal coupling transmission are realized between the primary-side transmitter and the secondary-side receiver through a dual-channel magnetic coupling structure. Preferably, the primary fundamental main power channel and the secondary fundamental main power channel constitute a fundamental main power channel CH1, and the primary third-harmonic channel and the secondary third-harmonic channel constitute a third-harmonic channel CH2;

[0008] A fourth capacitor and a first primary-side coil are connected between the midpoint of the first bridge arm H1 and the midpoint of the second bridge arm H2 . A fifth capacitor and a second primary-side coil are connected between the midpoint of the first bridge arm H1 and the reference ground; , the midpoint of the diode rectifier half-bridge H3 and the second capacitor , the third capacitor The midpoint of the series circuit There is a sixth capacitor connected between and the first secondary coil , the midpoint of the active rectifier half-bridge H4 and the second capacitor , the third capacitor The midpoint of the series circuit There is a seventh capacitor connected between and the second secondary coil .

[0009] Preferably, the dual-channel magnetic coupling structure includes a first primary coil , a second primary coil , a first secondary coil and a second secondary coil ; The first primary coil and the first secondary coil are square coils, and the second primary coil and the second secondary coil are figure-eight coils. The first primary coil and the second primary coil are stacked, and the first secondary coil and the second secondary coil are stacked.

[0010] Preferably, the primary third-harmonic channel is sequentially connected with a first zero-crossing sampling module ZCD1 and a primary controller FPGA-A; The secondary fundamental power channel is sequentially connected with a second zero-crossing sampling module ZCD2 and a secondary controller FPGA-B; The secondary third-harmonic channel is sequentially connected with a third zero-crossing sampling module ZCD3 and a secondary controller FPGA-B.

[0011] Preferably, the output end of the secondary receiving end is connected with a voltage sensor VS and a current sensor CS, and both the voltage sensor VS and the current sensor CS are connected with the secondary controller FPGA-B.

[0012] This application also discloses a communication-free power control method for a dual-channel wireless power supply system, which is applicable to the above-mentioned dual-channel wireless power supply system, and includes the following steps: S1. At the secondary receiving end, according to the actual output power and the set target power The error power signal between Generate a phase-shifting angle control command, and use the phase-shifting angle control command to make the secondary three-harmonic current in the secondary three-harmonic channel Phase Generate a phase shift ; S2. The secondary receiving end transmits the phase shift obtained in S1 to the primary transmitting end through the three-harmonic channel CH2 This transmission depends on the frequency response characteristics of the three-harmonic channel and does not require a communication device; S3. In the primary transmitting end, detect and calculate the primary three-harmonic current and the port voltage of the primary three-harmonic channel The phase difference between , and compare it with the reference phase difference to obtain the phase error signal of the primary three-harmonic current ; ; S4. According to the phase error signal of the primary three-harmonic current , adjust the duty cycle of the full-bridge inverter to achieve closed-loop regulation of the power of the fundamental wave main power channel CH1; S5. By superimposing the three-harmonic current and the fundamental wave current, expand the zero-voltage switching working range of the full-bridge inverter at the primary transmitting end. Preferably, S1 includes the following steps:

[0013] Preferably, S1 includes the following steps: S11. Use the voltage sensor VS and the current sensor CS to collect the rectified output voltage and the rectified output current in real time; S12. Calculate the actual output power , and compare it with the set target power to obtain the error power signal ; S13. Input the error power signal into the proportional-integral control module in the secondary controller FPGA-B to generate a phase-shifting angle control command and ; S14. According to the phase-shifting angle control command and control the conduction timing of the switching devices in the active rectifier half-bridge H4, so that the secondary three-harmonic current in the secondary three-harmonic channel Phase Generate a phase shift .

[0014] Preferably, S3 includes the following steps: S31. Collect the primary side third harmonic current in the primary side third harmonic channel of the primary side transmitter and the port voltage of the primary side third harmonic channel ; S32. Adopt a method combining zero-crossing detection and low-pass filtering to convert the zero-crossing points of the primary side third harmonic current into a digital pulse sequence and obtain the phase . The phase of the port voltage of the primary side third harmonic channel is replaced by the phase of the switching signal output by the corresponding controller. According to the pulse sequence of the primary side third harmonic current and the phase of the port voltage of the primary side third harmonic channel calculate the phase difference between the voltage and the current ; S33. Compare the phase difference with the set reference phase difference to calculate and obtain the phase error signal of the primary side transmitter . .

[0015] Preferably, the S4 includes the following steps: Input the phase error signal into the proportional-integral control module in the primary side controller FPGA-A to generate the change value of the phase shift duty cycle , and adjust the duty cycle of the full-bridge inverter in the primary side transmitter to achieve real-time power regulation of the fundamental wave main power channel CH1.

[0016] Preferably, the S5 includes the following steps: The primary side third harmonic current is injected via the first bridge arm H1 and superimposed with the primary side fundamental wave current to enable the first bridge arm H1 to have the ZVS conduction current condition in the low power range and expand the zero voltage switching working range of the full-bridge inverter.

[0017] Advantages of the present invention: (1) Without a communication module, the structure is simpler. Using the harmonic current phase shift as feedback information, without communication devices such as Bluetooth and Wi-Fi, reduces the system complexity and cost, and improves the anti-interference ability; (2) Strong power adaptive regulation ability. The secondary side receiver adjusts the harmonic phase through error drive, and the primary side (transmitter) detects and controls the power output in real time, forming a complete closed loop, which is suitable for application scenarios with frequent load changes; (3) Assist in achieving zero - voltage switching. By introducing the third - harmonic current, the zero - voltage - switching (ZVS) operating range of the primary - side emitter inverter in the low - power region is extended, effectively reducing the switching loss and improving the system efficiency; (4) Strong structural versatility. This method is applicable to typical dual - channel wireless power transfer systems, compatible with the primary - side emitter H - type full - bridge and secondary - side receiver hybrid - rectifier structures, facilitating integration and popularization; (5) Wide range of applicable scenarios. It is suitable for high - power wireless power supply systems such as electric - vehicle wireless charging, industrial robots, and non - contact power supply for rail transit, with good adaptability and robustness. Description of the Drawings

[0018] Figure 1 Schematic diagram of the topology structure of the dual - channel wireless power transfer system to which the present invention is applicable for this control method; Figure 2 Schematic diagram of the structure of the supporting dual - channel magnetic coupler of the present invention; Figure 3 Control flow chart of the wireless power transfer power control method based on third - harmonic phase feedback of the present invention; Figure 4 Implementation scheme of zero - crossing detection and digital low - pass filtering of the present invention; Figure 5 Operating waveform diagram of the system of the present invention achieving the zero - voltage - switching (ZVS) state within the full - power range; Figure 6 Experimental waveform for validating the effectiveness of the wireless power transfer power control method with third - harmonic phase feedback of the present invention. Detailed Embodiment

[0019] To make the purpose, technical solutions and advantages of the present application clearer, the following examples are given with reference to the accompanying drawings to further elaborate on the present application in detail.

[0020] An embodiment of the present application discloses a dual - channel wireless power supply system, including a primary - side emitter and a secondary - side receiver. The primary - side emitter and the secondary - side receiver jointly construct a fundamental - wave main - power channel CH1 and a third - harmonic channel CH2. The fundamental - wave main - power channel CH1 is used for main energy transfer, and the third - harmonic channel CH2 is used for feedback control and undertakes part of the power - transfer function. The specific structure of the dual - channel wireless power supply system in this embodiment is as Figure 1 shown.

[0021] The primary - side emitter includes a full - bridge inverter composed of a first bridge arm H1 and a second bridge arm H2 to drive two resonant branches of the fundamental - wave main - power channel CH1 and the third - harmonic channel CH2. The first bridge arm H1 includes a first switching device connected in series and a second switching device , the midpoint of the first bridge arm H1 is located between the first switching device and the second switching device The second bridge arm H2 includes a third switching device and a fourth switching device connected in series and the fourth switching device , the midpoint of the second bridge arm H2 is located between the third switching device and the fourth switching device The midpoint of the first bridge arm H1 and the midpoint of the second bridge arm H2 are connected with a fourth capacitor and a first primary coil , the fourth capacitor and the first primary coil are in series. The midpoint of the first bridge arm H1 and the reference ground are connected with a fifth capacitor and a second primary coil , there is a fifth capacitor and the second primary coil are in series. The circuit between the midpoint of the first bridge arm H1 and the midpoint of the second bridge arm H2 is the primary fundamental main power channel, and the circuit between the midpoint of the first bridge arm H1 and the reference ground g is the primary third harmonic channel. A DC voltage and a first capacitor are connected in parallel on the input side of the full-bridge inverter, and the DC voltage and the first capacitor are in parallel. The primary DC side input current charges the first capacitor and supplies power to the first bridge arm H1 and the second bridge arm H2.

[0022] The secondary receiving end includes a diode rectifier half-bridge H3 and an active rectifier half-bridge H4 connected in parallel. Among them, the diode rectifier half-bridge H3 includes a first diode and a second diode connected in series and the second diode , the midpoint of the diode rectifier half-bridge H3 is located between the first diode and the second diode The active rectifier half-bridge H4 includes a first active switching device and a second active switching device connected in series and the second active switching device , the midpoint of the active rectifier half-bridge H4 is located between the first active switching device and the second active switching device A second capacitor and the third capacitor of the series circuit, the midpoint of the diode rectifier half-bridge H3 and the second capacitor , the third capacitor The midpoint of the series circuit is connected with a sixth capacitor and the first secondary coil , the sixth capacitor and the first secondary coil are in series, the midpoint of the active rectifier half-bridge H4 and the second capacitor , the third capacitor The midpoint of the series circuit is connected with a seventh capacitor and the second secondary coil , the seventh capacitor and the second secondary coil are in series. The midpoint of the diode rectifier half-bridge H3 and the second capacitor , the third capacitor The midpoint of the series circuit The circuit between them is the secondary fundamental wave main power channel, and the midpoint of the active rectifier half-bridge H4 and the second capacitor , the third capacitor The midpoint of the series circuit The circuit between them is the secondary third harmonic channel. The secondary DC side input current is for the load to supply energy.

[0023] The primary fundamental wave main power channel and the secondary fundamental wave main power channel form the fundamental wave main power channel CH1, and the primary third harmonic channel and the secondary third harmonic channel form the third harmonic channel CH2. Among them, the fundamental wave main power channel CH1 is driven by a full-bridge inverter and transmits the fundamental wave power through the fundamental wave resonant network ( , , , ). The third harmonic channel CH2 is powered by the half-bridge inverter H1 and transmits the third harmonic feedback signal through the third harmonic resonant network ( , , , ). The secondary receiving end uses the diode half-bridge rectifier H3 to rectify the fundamental wave power and rectifies the fundamental wave frequency electrical energy into a DC voltage for the load . By actively rectifying the half-bridge H4 to regulate the phase of the third harmonic channel CH2, the non-communication closed-loop power regulation is realized. The DC voltage At the input side of the primary transmitting end, a support capacitor (the first capacitor to provide a stable DC voltage, and the second capacitor is used on the output side of the secondary receiving end and the third capacitor in series, which reduces the output voltage ripple and provides a mid-potential point .

[0024] The energy and signal coupling transmission between the primary transmitting end and the secondary receiving end is realized through a dual-channel magnetic coupling structure. The specific structure is as Figure 2 shown. The dual-channel magnetic coupling structure includes a first primary coil , a second primary coil , a first secondary coil and a second secondary coil . The primary transmitting end and the secondary receiving end are respectively composed of two independent winding systems. Among them, the first primary coil and the first secondary coil are square coils, which form the fundamental wave main power channel CH1. The second primary coil and the second secondary coil are figure-eight coils, which form the third harmonic channel CH2 for phase feedback and partial power transmission. Stack the fundamental wave channel coil of the primary transmitting end, the first primary coil , and the third harmonic channel coil, the second primary coil . Stack the fundamental wave channel coil of the secondary receiving end, the first secondary coil , and the third harmonic channel coil, the second secondary coil . The stacking scheme reduces the overall floor area. At the same time, due to the figure-eight coil characteristics of the third harmonic channel, the mutual inductance influence between different channels is eliminated, avoiding mutual interference between different channels and affecting the transmission power and signal transmission.

[0025] At the same time, the mutual inductance between the primary and secondary sides of the same channel is not affected. As Figure 1 and Figure 2 shown, where is the main mutual inductance of the fundamental wave main power channel CH1 to ensure the power transmission efficiency; is the mutual inductance of the third harmonic channel CH2 for signal feedback and partial harmonic power transmission. To make CH1 work at the fundamental wave frequency and CH2 work at the third harmonic frequency, the system resonance parameters need to be configured as follows:

[0026] Among them, is the angular frequency of the fundamental wave main power current, with the unit of rad / s. Similarly, the angular frequency of the third harmonic current is .

[0027] To provide a hardware basis for subsequent control methods, a signal acquisition device is configured in the dual-channel wireless power supply system in this embodiment, specifically as follows Figure 1 shown. At the primary side transmitter, the primary side third harmonic channel is sequentially connected to a first zero-crossing sampling module ZCD1 and a primary side controller FPGA-A. At the secondary side receiver, the secondary side third harmonic channel is sequentially connected to a third zero-crossing sampling module ZCD3 and a secondary side controller FPGA-B. The load is set at the output end of the secondary side receiver. The output end of the secondary side receiver is also connected to a voltage sensor VS and a current sensor CS, and both the voltage sensor VS and the current sensor CS are connected to the secondary side controller FPGA-B.

[0028] In another embodiment of the present application, a communication-free power control method for a dual-channel wireless power supply system is also disclosed. This method is applicable to the above dual-channel wireless power supply system, and the specific process is as follows Figure 3 shown. When the dual-channel wireless power supply system starts, the full-bridge inverter (H1 and H2) at the primary side transmitter operates at the fundamental frequency , driving the fundamental resonance network ( , , , ) to achieve wireless power transmission. The diode rectifier half-bridge H3 at the secondary side receiver converts the received fundamental power into a DC voltage , for powering the load. At the same time, the third harmonic channel CH2 is driven by the active rectifier half-bridge H4 to establish a feedback signal path. Specifically, the communication-free power control method for the dual-channel wireless power supply system disclosed in this embodiment includes two stages: stage 1 and stage 2.

[0029] Stage 1 As shown in (a) of Figure 3 , the primary side controller FPGA-A has a corresponding reference clock of Clock A. The duty cycle is initially set to (the recommended value is 0.3), generating a control signal to drive the switching devices in the full-bridge inverter at the primary side transmitter, generating the primary side fundamental main power channel port voltage and the primary side third harmonic channel port voltage . At the primary side transmitter, the primary side controller FPGA-A is responsible for outputting the switching waveforms of the first switching device and the second switching device . The zero-crossing detection and digital low-pass filtering (ZCD-DLPF) are used to perform zero-crossing detection on the primary side third harmonic current , and the period of the primary side third harmonic current is obtained.

[0030] The current zero-crossing detection and digital low-pass filtering scheme is as follows Figure 4 As shown, the current entering the current sampling is , through a current transformer TA with a ratio of 1:40, the sampling current is reduced to 1 / 40, and the sampled output current is obtained. The sampled output current is converted into a voltage signal through the subsequent resistor-capacitor circuit. The voltage signal is combined with the comparator TLV3502 to obtain the zero-crossing pulse of the current , and is input to the I / O of the FPGA. After the FPGA detects the level change of the corresponding I / O, it passes through a low-pass digital filter DLPF to reduce the interference of the zero-crossing pulse. The zero-crossing phase and period collected are output as the zero-crossing phase and period after passing through the low-pass digital filter DLPF. In this embodiment, the FPGA model used is EP3C25E144I7N.

[0031] Continuously sample the period of the above-mentioned primary side third harmonic current 2048 times, and judge whether its change is less than 5% of the third harmonic period (1 / 3 of the fundamental wave period ). If the condition is satisfied for 2048 consecutive samples, it is determined that the initialization of the primary side transmitter is completed. Otherwise, recalculate the number of samples until the condition that the change value of the current period for 2048 consecutive samples is less than 1 / 3 of the fundamental wave period is satisfied. The change value of the primary side third harmonic current period is expressed as:

[0032] where, and are two adjacent periods detected by the primary side third harmonic current after passing through the low-pass digital filter DLPF, so as to calculate the change value of the two adjacent periods.

[0033] As shown in (b) of Figure 3 , the secondary side controller FPGA-B has a corresponding reference clock of Clock B. The first diode , the second diode in the driving diode rectifier half-bridge H3, and the first active switch device , the second active switch device , put it in the diode rectification mode (DRM), at this time the secondary side third harmonic channel only serves as a power channel. Use zero-crossing detection and digital low-pass filtering (ZCD-DLPF) to sample the secondary side fundamental current and the secondary side third harmonic current of the zero-crossing period and . Similarly, when the above two periods and are respectively kept within a 5% error range of the fundamental period and the third harmonic period within 2048 samplings, it is determined that the initialization of the secondary side receiving end is completed. The period change values of each current are:

[0034] Among them, and are two adjacent periods detected after the secondary side fundamental current passes through the low-pass digital filter DLPF, in order to calculate the change value of the two adjacent secondary side fundamental current periods. and are two adjacent periods detected after the secondary side third harmonic current passes through the low-pass digital filter DLPF, in order to calculate the change value of the two adjacent secondary side third harmonic current periods.

[0035] After the initialization is completed, the phase difference between the secondary side fundamental current and the secondary side third harmonic current is stored in the memory of the FPGA controller, stored as . Among them the calculation formula is as follows:

[0036] Among them, is the real-time phase of the secondary side third harmonic current , is the real-time phase of the secondary side fundamental current .

[0037] Stage 2 Stage 2 is the real-time power regulation stage, as shown in (c), (d), (e) in Figure 3 , including the following steps: S1. At the secondary side receiving end, according to the actual output power and the set target power the error power signal between Generate a phase-shifting angle control command, and use the phase-shifting angle control command to make the secondary three-harmonic current in the secondary three-harmonic channel of the real-time phase produce a phase shift .

[0038] S11. Use the voltage sensor VS and the current sensor CS to collect the rectified output voltage and the rectified output current .

[0039] S12. Use the rectified output voltage and the rectified output current to calculate the actual output power , and compare the actual output power with the set target power to obtain an error power signal .

[0040] S13. Input the error power signal into the proportional-integral (PI) control module in the secondary controller FPGA-B to generate a phase-shifting angle control command and .

[0041] S14. According to the phase-shifting angle control command and control the conduction timing of the switching devices in the active rectifier half-bridge H4, so that the real-time phase of the secondary three-harmonic current in the secondary three-harmonic channel produces a phase shift .

[0042] Specifically, according to the phase-shifting angle control command and calculate the phase-shifting angle of the secondary harmonic port voltage relative to the secondary three-harmonic current . And based on the phase of the secondary fundamental current detected at the secondary receiving end as a reference, combined with the stored phase difference , calculate the real-time phase of the secondary three-harmonic current , and the calculation formula is as follows:

[0043] Then, combined with the obtained by the proportional-integral control module, adjust in real time and , and then adjust the first active switch device in real time and the second active switch device phase, so as to generate the real-time phase of the secondary side third harmonic current Generate a phase shift Generate a phase shift . The phase shift is used to feedback the power change of the secondary side receiving end. At the same time, the third harmonic channel CH2 has the ability to transmit electric energy and is used to transmit part of the power to the load .

[0044] S2. The secondary side receiving end transmits the phase shift obtained by S1 through the third harmonic channel CH2 to the primary side transmitting end. The transmission depends on the frequency response characteristics of the third harmonic channel and does not require a communication device.

[0045] The third harmonic channel CH2 transmits independently phase shift information. Due to the series resonance characteristic, this phase shift information is directly transmitted to the primary side transmitting end along with the current waveform, so no communication device is required.

[0046] S3. In the primary side transmitting end, detect and calculate the primary side third harmonic current and the voltage at the primary side third harmonic channel port phase difference between , and compare with the reference phase difference to obtain the phase error signal of the primary side third harmonic current .

[0047]

[0047] S31. Collect the primary side third harmonic current in the primary side third harmonic channel of the primary side transmitting end and the voltage at the primary side third harmonic channel port .

[0048] S32. The primary side transmitting end detects the zero crossing point of the primary side third harmonic current through the first zero crossing sampling module ZCD1. By combining zero crossing detection and low-pass filtering, the zero crossing point of the primary side third harmonic current is converted into a digital pulse sequence and the phase is obtained. The phase of the voltage at the primary side third harmonic channel port is replaced by the phase of the switching signal ( and ) output by the corresponding controller. Calculate the phase difference between the voltage at the primary side third harmonic channel port and the primary side third harmonic current , and the calculation formula is as follows:

[0049] S33. Compare the phase difference with the reference phase difference (dynamically varying according to the ZVS boundary or power requirement), and obtain the primary side phase error signal after comparison . The purpose of the reference phase difference is to select an appropriate harmonic phase according to the current duty cycle to improve the ZVS condition, ensuring that the system can achieve ZVS at different power levels. In this embodiment, the calculation formula of the reference phase difference is as follows:

[0050] S4. According to the phase error signal , adjust the duty cycle of the full-bridge inverter to achieve closed-loop regulation of the power of the fundamental wave main power channel CH1.

[0051] Input the primary side phase error signal into the proportional-integral control module in the primary side controller FPGA-A. This proportional-integral control module outputs an adjustment instruction for the duty cycle of the full-bridge inverter, which is used to control the third switching device and the fourth switching device of the second bridge arm H2 to generate a new duty cycle of the full-bridge inverter, achieving closed-loop regulation of the fundamental wave main power channel CH1.

[0052] S5. Expand the zero-voltage switching operating range of the primary side transmitting full-bridge inverter by superimposing the third harmonic current and the fundamental wave current.

[0053] The third harmonic channel CH2 is connected to the midpoints of the bridge arms (point and ) of the primary side transmitting full-bridge inverter through an independent resonant network ( and point ), forming a composite current path that is superimposed on the primary side fundamental wave current of the fundamental wave main power channel CH1, thereby providing sufficient discharge current before the full-bridge inverter switches and achieving zero crossing of the device port voltage. The primary side third harmonic current is injected via the first bridge arm H1 and superimposed on the primary side fundamental wave current , enabling the first bridge arm H1 to have the ZVS conduction current condition in the low power range, expanding the zero-voltage switching operating range of the full-bridge inverter and improving the system efficiency. The resonant frequency of the third harmonic channel CH2 is three times that of the fundamental wave main power channel CH1, ensuring that the two channels are decoupled from each other, achieving independent operation, and avoiding the third harmonic current ( and ) Interfere with the fundamental power transmission.

[0054] As Figure 5 shown, in this embodiment, the zero-voltage switching (ZVS) characteristics of the full-bridge inverter are optimized by the third-harmonic current, enabling the switching devices to achieve zero-voltage switching within a wider power range, reducing the switching losses, and improving the system efficiency. In Fig. (a), the ZVS situation of the traditional WPT system is shown. The switching devices at the primary transmitter side Figure 5 only have fundamental current , and fail to achieve ZVS when the duty cycle is relatively small, resulting in additional switching losses. In Fig. (b), the proposed scheme in this application is shown. Under the action of the primary third-harmonic current Figure 5 , the total current of the first bridge arm H1 changes, significantly expanding the ZVS operating range of the first bridge arm, thereby improving the system energy efficiency. The system automatically adjusts the phase of the primary third-harmonic current according to Equation (7), such that at different duty cycles , the superposition current formed by and satisfies the ZVS condition of the first bridge arm H1, achieving wide-range soft switching.

[0055] Through the above control method, the system not only realizes the dynamic power closed-loop regulation without communication but also significantly expands the ZVS operating range of the primary transmitter-side inverter, improving the system operation efficiency and reliability, and is applicable to complex application scenarios.

[0056] In a specific embodiment, the effectiveness of this application is verified through experiments. The test conditions of the experimental platform are as follows: the fundamental frequency is configured as 100 kHz, the third-harmonic frequency is 300 kHz, the input voltage is 400 V, and the rated power of this experiment is 1.7 kW. As Figure 6 shown in Fig. (a), after the system starts, it completes the initialization in the short stage 1 and then enters the dynamic power control stage (stage 2). In stage 2, when the system changes the given voltage , using the non-communication power control method of the dual-channel wireless power supply system proposed in the embodiment of this application, the system output voltage can adjust the voltage to output 252 V - 180 V - 252 V, realizing the non-communication closed-loop control of power. As Figure 6 shown in Fig. (b), after the system starts and completes the initialization in stage 1, it enters the dynamic power control stage (stage 2). In stage 2, when the system load When dynamic changes occur, the non-communication power control method of the dual-channel wireless power supply system proposed in the application embodiment is used, and the system output voltage can ensure a stable voltage output of 252V, realizing non-communication closed-loop power control. When the load power exceeds 50%, the system efficiency can reach over 93.5%, and the peak efficiency reaches 94.39%. Under light load conditions, the system still maintains the ZVS conduction of the inverter, verifying the effectiveness of the harmonic-assisted ZVS mechanism.

[0057] The non-communication power control method of the dual-channel wireless power supply system proposed in this application is applicable to wireless power transmission systems without communication devices, and can achieve adaptive power regulation under non-ideal working conditions such as load changes and coupling offsets, and maintain the stability and high efficiency of the system output. In addition, the dual-channel wireless power supply system proposed in this application is applicable to high-power application scenarios including but not limited to wireless charging of electric vehicles, wireless power supply of mobile robots, and non-contact power supply of rail transit. The dual-channel magnetic coupling structure adopted can reduce the dependence on the precise alignment of the primary and secondary receiving coils under the condition of simultaneous transmission of power and control information, and improve the practicability, efficiency and deployment flexibility of the wireless power supply system.

[0058] The basic principles, main features and advantages of the present invention have been shown and described above. Those skilled in the art should understand that the present invention is not limited by the above embodiments, and what is described in the above embodiments and the specification is only to illustrate the principle of the present invention. Without departing from the spirit and scope of the present invention, the present invention will have various changes and improvements, and these changes and improvements all fall within the scope of the present invention claimed. The scope of protection claimed by the present invention is defined by the appended claims and their equivalents.

Claims

1. A dual-channel wireless energy supply system, characterized in that: Including the primary side transmitting end and the secondary side receiving end; The primary transmitting end includes a full-bridge inverter composed of a first bridge arm H1 and a second bridge arm H2, and a DC voltage is connected in parallel at the input side of the full-bridge inverter. and the first capacitor , the midpoint of the first bridge arm H1 and the midpoint of the second bridge arm H2 The primary fundamental wave main power channel is connected between them, and the midpoint of the first bridge arm H1 A primary third harmonic channel is connected to the reference ground g; The secondary side receiving end includes a diode rectifier half-bridge H3 and an active rectifier half-bridge H4 connected in parallel, and a second capacitor is connected in parallel on the output side of the secondary side receiving end. and the third capacitor The series circuit of the diode rectifier half bridge H3 midpoint With the second capacitor , the third capacitor Midpoint of a series circuit The secondary fundamental wave main power channel is connected between the active rectifier half bridge H4 midpoint With the second capacitor , the third capacitor Midpoint of a series circuit A secondary third harmonic channel is connected between them; The primary transmitting end and the secondary receiving end realize the coupling transmission of energy and signal through a dual-channel magnetic coupling structure.

2. The dual-channel wireless energy supply system according to claim 1, characterized in that: The primary side fundamental wave main power channel and the secondary side fundamental wave main power channel constitute the fundamental wave main power channel CH1, and the primary side third harmonic channel and the secondary side third harmonic channel constitute the third harmonic channel CH2; The midpoint of the first bridge arm H1 and the midpoint of the second bridge arm H2 A fourth capacitor is connected between and the first primary coil , the midpoint of the first bridge arm H1 A fifth capacitor is connected between the reference ground and the second primary coil , diode rectifier half bridge H3 midpoint With the second capacitor , the third capacitor Midpoint of a series circuit A sixth capacitor is connected between and the first secondary coil , active rectifier half bridge H4 midpoint With the second capacitor , the third capacitor Midpoint of a series circuit A seventh capacitor is connected between and the second secondary coil .

3. The dual-channel wireless energy supply system according to claim 2, characterized in that: The dual-channel magnetic coupling structure includes a first primary coil , Second primary coil , the first secondary coil and the second secondary coil ; The first primary coil and the first secondary coil It is a square coil, the second primary coil and the second secondary coil It is an 8-shaped coil, the first primary coil and the second primary coil Stacked, first secondary coil and the second secondary coil Stack.

4. The dual-channel wireless energy supply system according to claim 3 is characterized in that: The primary third harmonic channel is sequentially connected to a first zero-crossing sampling module ZCD1 and a primary controller FPGA-A; The secondary side fundamental wave main power channel is connected to the second zero crossing sampling module ZCD2 and the secondary side controller FPGA-B in sequence; The secondary side third harmonic channel is sequentially connected to a third zero-crossing sampling module ZCD3 and a secondary side controller FPGA-B.

5. The dual-channel wireless energy supply system according to claim 4, characterized in that: The output end of the secondary side receiving end is connected to a voltage sensor VS and a current sensor CS, and both the voltage sensor VS and the current sensor CS are connected to the secondary side controller FPGA-B.

6. A non-communication power control method for a dual-channel wireless energy supply system, characterized in that: The dual-channel wireless energy supply system according to any one of claims 1 to 5 comprises the following steps: S1, at the secondary receiving end, according to the actual output power With setting target power The error power signal between Generate a phase shift angle control instruction, and use the phase shift angle control instruction to make the secondary third harmonic current of the secondary third harmonic channel Phase Phase shift ; S2, the secondary side receiving end uses the third harmonic channel CH2 to offset the phase obtained by S1 Transmitting to the primary transmitting end, the transmission relying on the frequency response characteristics of the third harmonic channel, without the need for a communication device; S3. In the primary transmitting end, detect and calculate the primary third harmonic current And the primary third harmonic channel port voltage The phase difference between , and the phase difference with the reference Compare and obtain the primary third harmonic current The phase error signal ; S4, according to the primary third harmonic current The phase error signal , adjust the duty cycle of the full-bridge inverter , realize closed-loop regulation of the fundamental wave main power channel CH1 power; S5. By superimposing the third harmonic current and the fundamental current, the zero voltage switching operating range of the primary transmitting end full-bridge inverter is extended.

7. The non-communication power control method of the dual-channel wireless energy supply system according to claim 6, characterized in that: The S1 comprises the following steps: S11, use the voltage sensor VS and current sensor CS to collect the rectifier output voltage in real time and rectified output current ; S12. Calculate the actual output power and set the target power Compare and get the error power signal ; S13, the error power signal Input the proportional-integral control module in the secondary controller FPGA-B to generate the phase shift angle control instruction and ; S14, according to the phase shift angle control instruction and Control the conduction timing of the switch devices in the active rectifier half-bridge H4, so that the secondary third harmonic current in the secondary third harmonic channel Phase Phase shift .

8. The non-communication power control method of the dual-channel wireless energy supply system according to claim 7, characterized in that: The S3 comprises the following steps: S31, collect the primary third harmonic current in the primary third harmonic channel of the primary transmitting end And the primary third harmonic channel port voltage ; S32, using the method of combining zero-crossing detection with low-pass filtering, the primary third harmonic current The zero crossing point is converted into a digital pulse sequence and the phase , the primary third harmonic channel port voltage Phase The phase of the switching signal output by the corresponding controller is replaced by the phase of the primary third harmonic current. The pulse sequence and the primary third harmonic channel port voltage Phase Calculate the phase difference between voltage and current ; S33, the phase difference Phase difference from the set reference By comparison, the phase error signal of the primary transmitting end is calculated. .

9. The non-communication power control method of the dual-channel wireless energy supply system according to claim 8, characterized in that: The S4 comprises the following steps: The phase error signal Input the proportional-integral control module in the primary controller FPGA-A to generate the change value of the phase shift duty cycle , adjust the duty cycle of the full-bridge inverter in the primary transmitter , to achieve real-time power regulation of the fundamental main power channel CH1.

10. The non-communication power control method of the dual-channel wireless energy supply system according to claim 9, characterized in that: The S5 comprises the following steps: The primary third harmonic current Injected through the first bridge arm H1, and the primary fundamental current The superposition is performed so that the first bridge arm H1 has the ZVS conduction current condition in the low power range, thereby extending the zero voltage switching operating range of the full-bridge inverter.

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