Method and system for continuous glucose monitoring

Abstract

A method for optional external calibration of a calibration-free glucose sensor uses a measured value of working electrode current (Isig) and EIS data to calculate a final sensor glucose (SG) value. The counter electrode voltage (Vcntr) may also be used as an input. The raw Isig and Vcntr values may be pre-processed, and low-pass filtering, averaging, and / or feature generation may be applied. SG values may be generated using one or more models for predicting SG calculations. Complex redundancy may be employed to take advantage of the operational advantages of different characteristics of two or more different or different sensors, including, for example, characteristics related to hydration, stability, and durability of such sensors. The use of such redundant glucose sensors, devices, and sensor systems may be achieved using fusion algorithms, EIS, and advanced application specific integrated circuits (ASICs) in a manner that can bridged the gap between quick start-up, sensor life, and calibration-free algorithm accuracy.

Description

[0001] This application is a divisional application of the application patent application with the application number 201880080621.1, the title "Method and system for continuous glucose monitoring" and the filing date of 15 October 2018. TECHNICAL FIELD

[0002] Embodiments of the present invention generally relate to sensor technology, including sensors and sensor devices for sensing a variety of physiological parameters, such as glucose concentration. More specifically, embodiments of the present invention relate to optional calibration in a calibration-free system, device, and method, as well as the use of complex redundancy in glucose sensors, devices, and sensor systems including closed-loop insulin infusion systems, as well as fusion algorithms, electrochemical impedance spectroscopy (EIS), and application specific integrated circuits (ASICs) for implementing such redundant glucose sensors, devices, and sensor systems. BACKGROUND

[0003] Subjects (e.g., patients) and medical personnel desire to monitor readings of physiological conditions in the subjects. Illustratively, subjects desire to continuously monitor blood glucose levels in the subjects. Currently, a patient can measure his / her blood glucose (BG) using a BG measurement device (i.e., a blood glucose meter), such as a test strip meter, a continuous glucose measurement system (or continuous glucose monitor), or a hospital hemacue. The BG measurement device measures the patient's BG level using various methods, such as using a sample of the patient's blood, a sensor in contact with a bodily fluid, an optical sensor, an enzyme sensor, or a fluorescence / fluorescence quenching sensor. When the BG measurement device has produced a BG measurement, the measurement is displayed on the BG measurement device.

[0004] Infusion pump devices and systems are relatively well known in the medical arts for use in delivering or dispensing prescribed medication, such as insulin, to a patient. In one form, such devices include a relatively compact pump housing adapted to receive a syringe or reservoir carrying the prescribed medication for administration to the patient through an infusion tube and associated catheter or infusion set. A programmable controller can operate the infusion pump continuously or at periodic intervals to achieve a tightly controlled and accurate delivery of medication over an extended period of time. Such infusion pumps are used to administer insulin and other medications, with exemplary pump configurations shown and described in U.S. Patent Nos. 4,562,751; 4,678,408; 4,685,903; 5,080,653; and 5,097,122, which are incorporated herein by reference.

[0005] Everyone has a basal insulin requirement, which in a diabetic individual can generally be maintained by administering a basal amount of insulin to the patient using an infusion pump continuously or on a regular basis. However, when additional glucose (i.e., above the basal level) is present in the diabetic individual, for example when the individual consumes a meal, the amount and timing of insulin to be administered must be determined so as to adequately account for the additional glucose while avoiding over-infusion of insulin. Typically, a bolus of insulin is administered to compensate for the meal (i.e., a meal bolus). Diabetics generally determine the amount of insulin they cover for an anticipated meal based on the carbohydrate content of the meal.

[0006] In recent years, a variety of electrochemical glucose sensors have been developed for obtaining an indication of the blood glucose level of a diabetic patient. Such readings are useful in monitoring and / or adjusting a treatment regimen that typically includes periodic administration of insulin to the patient. Typically, periodic readings can be obtained over an extended period of time using a small and flexible electrochemical sensor. In one form, a flexible subcutaneous sensor is constructed in accordance with a thin film mask technology. Typical thin film sensors are described in commonly assigned U.S. Patent Nos. 5,390,671; 5,391,250; 5,482,473; and 5,586,553, which are incorporated herein by reference.

[0007] These electrochemical sensors have been applied in telemetered characteristic monitoring systems. As described, for example, in commonly assigned U.S. Patent No. 6,809,653 ("the '653 patent"), which is incorporated herein by reference in its entirety, a telemetered system includes a remotely located data receiver, a sensor for producing a signal indicative of a characteristic of a user, and a transmitter device for processing the signal received from the sensor and for wirelessly transmitting the processed signal to the remotely located data receiver. The data receiver can be a characteristic monitor, a data receiver that provides data to another device, an RF programmer, a medication delivery device such as an infusion pump, or the like.

[0008] Current continuous glucose measurement systems include subcutaneous (or short-term) sensors and implantable (or long-term) sensors. With each of the short-term sensors and the long-term sensors, the patient must wait a certain amount of time for the continuous glucose sensor to stabilize and provide accurate readings. In many continuous glucose sensors, the subject must wait three hours for the continuous glucose sensor to stabilize before any glucose measurement is utilized. This is inconvenient for the patient and, in some cases, can cause the patient to not utilize the continuous glucose measurement system.

[0009] In addition, when a glucose sensor is first inserted into a patient's skin or subcutaneous layer, the glucose sensor is not operating in a steady state. The sensor electrical readings indicative of the patient's glucose level vary over a wide range of readings. In the past, sensor stabilization often took several hours. Techniques for sensor stabilization are described in detail, for example, in the '653 patent, where the initialization process for sensor stabilization can be reduced to about an hour. A high voltage (e.g., 1.0 to 1.2 volts) can be applied for 1 to 2 minutes to stabilize the sensor, and then a low voltage (e.g., 0.5 to 0.6 volts) can be applied for the remainder of the initialization process (e.g., about 58 minutes).

[0010] There is also a need to sufficiently "wet" or hydrate the electrodes of a sensor before utilizing the electrodes of the sensor. If the electrodes of a sensor are not sufficiently hydrated, the result can be inaccurate readings of the patient's physiological condition. Users of current blood glucose sensors can be instructed not to power up the sensor immediately. Such blood glucose sensors can not operate in an optimal or efficient manner if utilized too early.

[0011] Many existing technologies for continuous glucose monitoring (CGM) are primarily supplementary, meaning that the readings provided by a CGM device (including, for example, an implantable or subcutaneous sensor) cannot be used to make a clinical decision without a reference value. The reference value, in turn, must be obtained from a finger stick using, for example, a BG meter. The reason for the need for a reference value is that the amount of information available from a sensor / sensing assembly is limited. Specifically, the sensing assembly can only provide raw sensor values (i.e., sensor current or Isig) and voltage for processing. Thus, during analysis, if the raw sensor signal appears abnormal (e.g., if the signal is decreasing), the only way to distinguish between a sensor malfunction and a physiological change (i.e., a change in glucose level) in the user / patient's body can be by taking a reference glucose value from a finger stick. It is well known that reference finger sticks are also used to calibrate the sensor.

[0012] The art has sought ways to eliminate or at least minimize the number of finger sticks needed to calibrate and assess sensor health. However, given the number and level of complexity of multiple sensor failure modes, a satisfactory solution has not been found. Only diagnostic methods based on direct assessment of Isig or comparison of two Isigs have been developed. In either case, because Isig tracks glucose levels in the body, Isig is clearly not analyte independent. Thus, Isig itself is not a reliable source of information for sensor diagnostics nor a reliable predictor of sensor continued performance.

[0013] Another limitation existing in the art to date is the lack of sensor electronics that not only can run the sensor, but also perform real-time sensor and electrode diagnostics, and can diagnose redundant electrodes, redundant sensors, complementary sensors, as well as redundant and complementary sensors, while managing the power supply of the sensor. It is certain that the concept of electrode redundancy has existed for some time. However, in the past, using electrode redundancy (and / or complementary and redundant electrodes) not only made it nearly impossible to obtain more than one reading at a time, but also made it nearly impossible to assess the relative health of the redundant electrodes, the overall reliability of the sensor, and the frequency of calibration reference values, if any, that were needed.

[0014] Additionally, even when redundant sensing electrodes have been used, the number has typically been limited to two. Again, this is attributed to the lack of advanced electronics that can run, assess, and manage multiple, independently working electrodes (e.g., up to 5 or more) in real-time. However, another reason is the limited view that redundant electrodes are used in order to obtain "independent" sensor signals, and that two redundant electrodes are sufficient for this purpose. As noted, while this is one function of utilizing redundant electrodes, it is not the only function. SUMMARY

[0015] According to embodiments of the present application, a method for optional external calibration of a calibration-free glucose sensor for measuring glucose levels in a user, wherein the glucose sensor comprises physical sensor electronics, a microcontroller, and a working electrode, the method comprising: periodically measuring, by the physical sensor electronics, an electrode current (Isig) signal of the working electrode; performing, by the microcontroller, an electrochemical impedance spectroscopy (EIS) procedure to generate EIS-related data of the working electrode; calculating, by the microcontroller, a respective sensor glucose (SG) value for each of a plurality of calibration-free SG prediction models based on the Isig signal and the EIS-related data and the plurality of calibration-free SG prediction models; calculating, by the microcontroller, a SG variance estimate value for each respective SG value; determining, by the microcontroller, whether an external blood glucose (BG) value is available, and when available, incorporating the BG value into the calculation of the SG value; fusing, by the microcontroller, the respective SG values from the plurality of SG prediction models to obtain a single fused SG value; applying, by the microcontroller, an unscented Kalman filter to any of the SG values; and calculating, by the microcontroller, a calibrated SG value to be displayed to the user.

[0016] According to other embodiments of the present application, a glucose monitoring system includes a glucose sensor device for determining glucose concentration levels in a user's body over a total sensor device wear time, wherein the total sensor device wear time includes a first time window, a subsequent second time window, and a transition period between the first time window and the second time window, and the glucose sensor device comprises a first glucose sensor and a second glucose sensor, wherein the first glucose sensor and the second glucose sensor have different characteristics in at least one of hydration, stability, and durability. The glucose sensor device further includes sensor electronics, wherein the sensor electronics include at least one physical microprocessor configured to: (a) periodically receive a respective first output signal from the first glucose sensor indicative of glucose concentration levels in the user's body; (b) calculate glucose concentration levels in the user's body during the first time window based entirely on the first output signal; (c) periodically receive a respective second output signal from the second glucose sensor indicative of glucose concentration levels in the user's body; (d) calculate glucose concentration levels during the transition period based on both the first output signal and the second output signal; and (e) calculate glucose concentration levels in the user's body during the second time window based entirely on the second output signal. BRIEF DESCRIPTION OF DRAWINGS

[0017] Embodiments of the present application will be described in detail with reference to the drawings, wherein like reference numerals identify corresponding parts.

[0018] FIG. 1 is a perspective view of a subcutaneous sensor insertion device and a block diagram of sensor electronics according to embodiments of the present application.

[0019] FIG. 2A shows a substrate with two sides, the first side containing an electrode configuration and the second side containing electronic circuitry.

[0020] FIG. 2B shows a general block diagram of electronic circuitry for sensing output of a sensor.

[0021] FIG. 3 shows a block diagram of sensor electronics and a sensor including multiple electrodes according to embodiments of the present application.

[0022] FIG. 4 shows an alternative embodiment of the present application including a sensor and sensor electronics according to embodiments of the present application.

[0023] FIG. 5 shows an electronic block diagram of sensor electrodes and voltages applied to the sensor electrodes according to embodiments of the present application.

[0024] FIG. 6A A method of applying a pulse during a stabilization time frame to reduce the stabilization time frame is shown according to an embodiment of the application.

[0025] FIG. 6B A method of stabilizing a sensor is shown according to an embodiment of the application.

[0026] FIG. 6C The use of feedback in stabilizing a sensor is shown according to an embodiment of the application.

[0027] FIG. 7 The effect of stabilizing a sensor is shown according to an embodiment of the application.

[0028] FIG. 8A A block diagram of a sensor electronics and a sensor containing a voltage generating device is shown according to an embodiment of the application.

[0029] FIG. 8B A voltage generating device for implementing this embodiment of the application is shown.

[0030] FIG. 8C A voltage generating device to generate two voltage values is shown according to an embodiment of the application.

[0031] FIG. 8D A voltage generating device with three voltage generating systems is shown according to an embodiment of the application.

[0032] FIG. 9A A sensor electronics containing a microcontroller for generating a voltage pulse is shown according to an embodiment of the application.

[0033] FIG. 9B A sensor electronics containing an analysis module is shown according to an embodiment of the application.

[0034] FIG. 10 A block diagram of a sensor system containing a hydration electronics is shown according to an embodiment of the application.

[0035] FIG. 11 An embodiment of the application containing a mechanical switch for assisting in determining a hydration time is shown.

[0036] FIG. 12 A method of detecting hydration is shown according to an embodiment of the application.

[0037] FIG. 13A A method of hydrating a sensor is shown according to an embodiment of the application.

[0038] FIG. 13B A further method for verifying hydration of a sensor is shown according to an embodiment of the application.

[0039] FIG. 14A 、 FIG. 14B and FIG. 14C shows a method of combining hydration of a sensor with stabilizing the sensor according to embodiments of the application.

[0040] FIG. 15A shows EIS-based analysis of system response to application of a periodic AC signal according to embodiments of the application.

[0041] FIG. 15B shows a known circuit model for electrochemical impedance spectroscopy.

[0042] FIG. 16A shows an example of a Nyquist plot according to embodiments of the application in which an AC voltage plus a DC voltage (DC bias) is applied to the working electrode for a selected frequency spectrum from 0.1 Hz to 1000 Mhz.

[0043] FIG. 16B shows another example of a Nyquist plot having a linear fit for relatively lower frequencies and an intercept at relatively higher frequencies approaching the real impedance value.

[0044] FIG. 16C and FIG. 16D shows infinite and finite glucose sensor responses to a sinusoidal working potential, respectively.

[0045] FIG. 16E shows a Bode plot of magnitude according to embodiments of the application.

[0046] FIG. 16F shows a Bode plot of phase according to embodiments of the application.

[0047] FIG. 17 shows a sensor impedance Nyquist plot changing as a sensor ages according to embodiments of the application.

[0048] FIG. 18 shows a method of applying EIS techniques in stabilizing a sensor and detecting sensor life according to embodiments of the application.

[0049] FIG. 19 shows a schedule for performing an EIS procedure according to embodiments of the application.

[0050] FIG. 20 shows a method of using an EIS procedure in conjunction with remedial measures to detect and repair a sensor according to embodiments of the application.

[0051] FIG. 21A and FIG. 21BAn example of a sensor remedy according to an embodiment of the present invention is shown.

[0052] FIG. 22 The Nyquist plot of a normally operating sensor is shown, where the Nyquist slope gradually increases and the intercept gradually decreases as the sensor is worn for a period of time.

[0053] FIG. 23A The original current signals (Isig) from two redundant working electrodes according to an embodiment of the present invention are shown, as well as the corresponding real impedance of the electrodes at 1 kHz.

[0054] FIG. 23B It shows FIG. 23A Nyquist plot of the first working electrode (WE1).

[0055] FIG. 23C It shows FIG. 23A Nyquist plot of the second working electrode (WE2).

[0056] FIG. 24 An example is shown of a signal drop of two redundant working electrodes according to an embodiment of the present invention and the corresponding real impedance of the electrodes at 1 kHz.

[0057] FIG. 25A The fundamental glucose independence of the real impedance, virtual impedance, and phase of a normally functioning glucose sensor according to an embodiment of the present invention at relatively high frequencies is shown.

[0058] FIG. 25B An illustrative example of the variation level of glucose correlation of real impedance at relatively low frequencies is shown according to an embodiment of the present invention.

[0059] FIG. 25C An illustrative example of the variation level of glucose correlation of phase at relatively low frequencies is shown according to an embodiment of the present invention.

[0060] FIG. 26 The trends of 1kHz real impedance, 1kHz virtual impedance, and relatively high-frequency phase are shown in an embodiment of the present invention when the glucose sensor loses sensitivity due to hypoxia at the sensor insertion site.

[0061] FIG. 27 The isig and phase of hypoxia simulated in vitro at different glucose concentrations according to an embodiment of the present invention are shown.

[0062] FIG. 28A to FIG. 28C An example of sensitivity loss due to hypoxia and EIS-based parameters of the electrodes in the case of redundant working electrodes WE1 and WE2, according to an embodiment of the present invention, is shown.

[0063] FIG. 28D shows an example of EIS-induced spikes in raw Isig for FIG. 28A to FIG. 28C

[0064] FIG. 29 shows an example of sensitivity loss due to occlusion according to an embodiment of the application.

[0065] FIG. 30A to FIG. 30C shows an example of sensitivity loss due to bio-fouling and redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes according to an embodiment of the application.

[0066] FIG. 30D shows an example of EIS-induced spikes in raw Isig for FIG. 30A to FIG. 30C

[0067] FIG. 31 shows a diagnostic procedure for sensor fault detection according to an embodiment of the application.

[0068] FIG. 32A and FIG. 32B shows another diagnostic procedure for sensor fault detection according to an embodiment of the application.

[0069] FIG. 33A shows a top level flowchart involving current (Isig) based fusion algorithm according to an embodiment of the application.

[0070] FIG. 33B shows a top level flowchart involving sensor glucose (SG) based fusion algorithm according to an embodiment of the application.

[0071] FIG. 34 shows details of sensor glucose (SG) based fusion algorithm for FIG. 33B according to an embodiment of the application.

[0072] FIG. 35 shows details of current (Isig) based fusion algorithm for FIG. 33A according to an embodiment of the application.

[0073] FIG. 36 is a graphical representation of calibration of a sensor in steady state according to an embodiment of the application.

[0074] FIG. 37 is a graphical representation of calibration of a sensor in transition according to an embodiment of the application.

[0075] FIG. 38A ​​This is a diagram illustrating the EIS-based dynamic slope (and slope adjustment) according to an embodiment of the present invention for sensor calibration.

[0076] FIG. 38B A flowchart of EIS auxiliary sensor calibration involving low-start detection is shown according to an embodiment of the present invention.

[0077] FIG. 39 The sensor current (Isig) and 1kHz impedance values ​​are shown in an external simulation of an interfering object very close to the sensor, according to an embodiment of the present invention.

[0078] FIG. 40A and FIG. 40B They respectively showed the targets for FIG. 39 The simulated phase and impedance Bode plots are shown.

[0079] FIG. 40C Showing the target FIG. 39 The simulated Nyquist plot is shown.

[0080] FIG. 41 Another in vitro simulation of the interfering object is shown according to an embodiment of the present invention.

[0081] FIG. 42A and FIG. 42B An ASIC block diagram according to an embodiment of the present invention is shown.

[0082] FIG. 43 A potentiostat configuration with redundant working electrodes for a sensor according to an embodiment of the present invention is shown.

[0083] FIG. 44 It shows having FIG. 43 The equivalent AC electrode circuit of the sensor configured in the potentiostat shown.

[0084] FIG. 45 Some key blocks of the EIS circuitry in the analog front-end IC of a glucose sensor according to an embodiment of the present invention are shown.

[0085] FIG. 46A to FIG. 46F It shows FIG. 45 The signal of the EIS circuit system shown is a simulation of the current multiplied by the 0-degree phase.

[0086] FIG. 47A to FIG. 47F It shows FIG. 45 The signal of the EIS circuit system shown is a simulation of the current multiplied by the 0-degree phase and the 90-degree phase.

[0087] FIG. 48 A circuit model according to an embodiment of the present invention is shown.

[0088] FIG. 49A to FIG. 49C A graphical representation of a circuit model according to an alternative embodiment of the application is shown.

[0089] FIG. 50A A Nyquist plot of a superimposed equivalent circuit simulation according to an embodiment of the application.

[0090] FIG. 50B An enlarged view of the high frequency portion of FIG. 50A

[0091] A Nyquist plot showing an increase of Cdl in the direction of arrow A according to an embodiment of the application. FIG. 51

[0092] A Nyquist plot showing an increase of a in the direction of arrow A according to an embodiment of the application. FIG. 52

[0093] A Nyquist plot showing an increase of Rp in the direction of arrow A according to an embodiment of the application. FIG. 53

[0094] A Nyquist plot showing an increase of the Warburg admittance in the direction of arrow A according to an embodiment of the application. FIG. 54

[0095] A Nyquist plot showing an increase of l in the direction of arrow A according to an embodiment of the application. FIG. 55

[0096] A Nyquist plot showing the influence of the membrane capacitance according to an embodiment of the application. FIG. 56

[0097] A Nyquist plot showing an increase of the membrane resistance in the direction of arrow A according to an embodiment of the application. FIG. 57

[0098] A Nyquist plot showing an increase of Rsol in the direction of arrow A according to an embodiment of the application. FIG. 58

[0099] A Nyquist plot showing the variation of EIS parameters related to the circuit elements during start-up and calibration according to an embodiment of the application. FIG. 59A to FIG. 59C

[0100] A Nyquist plot showing the variation of a different set of EIS parameters related to the circuit elements during start-up and calibration according to an embodiment of the application. FIG. 60A to FIG. 60C

[0101] A Nyquist plot showing the variation of another different set of EIS parameters related to the circuit elements during start-up and calibration according to an embodiment of the application. FIG. 61A to FIG. 61C ​

[0102] FIG. 62 EIS response for multiple electrodes is shown in accordance with embodiments of the application.

[0103] FIG. 63 Nyquist plot showing the effect of Isig calibration by increasing glucose in accordance with embodiments of the application.

[0104] FIG. 64 Nyquist plot showing the effect of oxygen (Vcntr) response in accordance with embodiments of the application.

[0105] FIG. 65 Shift in Nyquist plot due to temperature change in accordance with embodiments of the application.

[0106] FIG. 66 Relationship between Isig and blood glucose in accordance with embodiments of the application.

[0107] FIG. 67A to FIG. 67B Sensor drift in accordance with embodiments of the application.

[0108] FIG. 68 Increase in membrane resistance during loss of sensitivity in accordance with embodiments of the application.

[0109] FIG. 69 Drop in Weber admittance during loss of sensitivity in accordance with embodiments of the application.

[0110] FIG. 70 Calibration curve in accordance with embodiments of the application.

[0111] FIG. 71 High frequency semicircle becoming visible on Nyquist plot in accordance with embodiments of the application.

[0112] FIG. 72A And FIG. 72B Vcntr drop (rail) and Cdl decrease in accordance with embodiments of the application.

[0113] FIG. 73 Changing slope of calibration curve in accordance with embodiments of the application

[0114] FIG. 74 Changing length of Nyquist plot in accordance with embodiments of the application.

[0115] FIG. 75 Nyquist plot showing FIG. 74 Zoomed in view of low and high frequency regions of the Nyquist plot of

[0116] FIG. 76A And FIG. 76BThe combined effect of the increase in membrane resistance, decrease in Cdl and drop in Vcntr according to embodiments of the application is shown.

[0117] FIG. 77 The relative Cdl values of two working electrodes according to embodiments of the application are shown.

[0118] FIG. 78 The relative Rp values of two working electrodes according to embodiments of the application are shown.

[0119] FIG. 79 The combined effect of the varying EIS parameters on the calibration curve according to embodiments of the application is shown.

[0120] FIG. 80 It is shown that the length of the Nyquist plot in the low frequency region is longer than in the region where sensitivity loss exists according to embodiments of the application.

[0121] FIG. 81 is a flowchart of sensor self-calibration based on detection of sensitivity change according to embodiments of the application.

[0122] FIG. 82 The horizontal shift in the Nyquist plot due to sensitivity loss according to embodiments of the application is shown.

[0123] FIG. 83 The method of developing a heuristic EIS metric based on the Nyquist plot according to embodiments of the application is shown.

[0124] FIG. 84 The relationship between Rmand the calibration factor according to embodiments of the application is shown.

[0125] FIG. 85 The relationship between Rmand normalized Isig according to embodiments of the application is shown.

[0126] FIG. 86 The Isig curves of various glucose levels over time according to embodiments of the application are shown.

[0127] FIG. 87 The Cdl curves of various glucose levels over time according to embodiments of the application are shown.

[0128] FIG. 88 The second inflection point of the plot of Rmvs time according to embodiments of the application is shown. FIG. 86 The second inflection point of the plot of Rmvs time according to embodiments of the application is shown.

[0129] FIG. 89 The second inflection point of the plot of Rmvs time according to embodiments of the application is shown. FIG. 88 The second inflection point of the plot of Rmvs time according to embodiments of the application is shown.

[0130] FIG. 90 One graph showing the relationship between the calibration factor (CF) and Rmem+Rsol according to embodiments of the application.

[0131] FIG. 91A is a graph showing in-vivo results of MARD for all valid BGs within approximately the first 8 hours of sensor life according to embodiments of the application.

[0132] FIG. 91B is a graph showing the median ARD number for all valid BGs within approximately the first 8 hours of sensor life according to embodiments of the application.

[0133] FIG. 92A to FIG. 92C shows calibration factor adjustment according to embodiments of the application.

[0134] FIG. 93A to FIG. 93C shows calibration factor adjustment according to embodiments of the application.

[0135] FIG. 94A to FIG. 94C shows calibration factor adjustment according to embodiments of the application.

[0136] FIG. 95 shows an illustrative example of initial decay of Cdl according to embodiments of the application.

[0137] FIG. 96 shows removal of non-faradaic current effects on Isig according to embodiments of the application.

[0138] FIG. 97A shows calibration factor before removal of non-faradaic current of two working electrodes according to embodiments of the application.

[0139] FIG. 97B shows calibration factor after removal of non-faradaic current of two working electrodes according to embodiments of the application.

[0140] FIG. 98A and FIG. 98B shows the effect of removal of non-faradaic current on MARD according to embodiments of the application.

[0141] FIG. 99 is a graph of double layer capacitance over time according to embodiments of the application.

[0142] FIG. 100 shows shift of Rmem+Rsol and appearance of high frequency semicircle during loss of sensitivity according to embodiments of the application.

[0143] FIG. 101A shows a flowchart of using combinatorial logic to detect loss of sensitivity according to embodiments of the application.

[0144] FIG. 101B A flowchart showing use of combinatorial logic to detect loss of sensitivity according to another embodiment of the application.

[0145] FIG. 102 An illustrative method for using Nyquist slope as a marker to distinguish between new and old sensors according to an embodiment of the application is shown.

[0146] FIG. 103A to FIG. 103C An illustrative example of Nyquist plots with different lengths for different sensor configurations according to an embodiment of the application is shown.

[0147] FIG. 104 Nyquist plot length over time for a sensor of FIG. 103A to FIG. 103C is shown.

[0148] FIG. 105 A flowchart for blanking sensor data or terminating a sensor according to an embodiment of the application is shown.

[0149] FIG. 106 A flowchart for sensor termination according to an embodiment of the application is shown.

[0150] FIG. 107 A flowchart for signal sag detection according to an embodiment of the application is shown.

[0151] FIG. 108A Isig and Vcntr over time are shown, and FIG. 108B glucose over time according to an embodiment of the application is shown.

[0152] FIG. 109A calibration ratio over time is shown, and FIG. 109B glucose over time according to an embodiment of the application is shown.

[0153] FIG. 110A and FIG. 110B calibration factor trend over time according to an embodiment of the application is shown.

[0154] FIG. 111 A flowchart for first day calibration (FDC) according to an embodiment of the application is shown.

[0155] FIG. 112 A flowchart for EIS-based calibration according to an embodiment of the application is shown.

[0156] FIG. 113 A flowchart of an existing calibration method is shown.

[0157] FIG. 114A calibration flowchart is shown according to embodiments of the application.

[0158] FIG. 115 A calibration flowchart is shown according to other embodiments of the application.

[0159] FIG. 116 A calibration flowchart is shown according to yet other embodiments of the application.

[0160] FIG. 117 A calibration flowchart is shown according to other embodiments of the application.

[0161] FIG. 118 A table showing comparative MARD values calculated based on embodiments of the application is shown.

[0162] FIG. 119 A flowchart for calculating raw fusion weights according to embodiments of the application is shown.

[0163] FIG. 120 A sensor glucose (SG) fusion logic diagram according to embodiments of the application is shown.

[0164] FIG. 121 A flowchart of a calibration-free back-calculation algorithm according to embodiments of the application is shown.

[0165] FIG. 122 A decision tree model according to embodiments of the application is shown.

[0166] FIG. 123 A decision tree model for blanking data according to embodiments of the application is shown.

[0167] FIG. 124 A table showing examples of parameters for blanking algorithms according to embodiments of the application is shown.

[0168] FIG. 125 Fusion, filtering, and blanking results according to embodiments of the application are shown.

[0169] FIG. 126 A flowchart of optional calibration logic according to embodiments of the application is shown.

[0170] FIG. 127 A comparison table between two different glucose sensor designs is shown.

[0171] FIG. 128 An example of complex redundancy according to embodiments of the application is shown.

[0172] FIG. 129 A block diagram including calibrated and uncalibrated models according to embodiments of the application is shown.

[0173] FIG. 130 A diagram showing fusion logic according to embodiments of the application.

[0174] FIG. 131 A diagram showing fusion logic with one calibrated model and one uncalibrated model according to embodiments of the application.

[0175] FIG. 132 A diagram showing fusion logic with two uncalibrated models according to embodiments of the application.

[0176] FIG. 133 A diagram showing fusion logic with two calibrated models according to embodiments of the application.

[0177] FIG. 134 A diagram showing fusion logic with multiple calibrated models and / or multiple uncalibrated models according to embodiments of the application. DETAILED DESCRIPTION

[0178] In the following description, reference is made to the accompanying drawings which form a part hereof, and in which are shown by way of illustration several embodiments of the present application. It is to be understood that other embodiments can be utilized and structural and operational changes can be made without departing from the scope of the present application.

[0179] The present application is described below with reference to flowchart illustrations of methods, systems, apparatuses, devices, and programming and computer program products. It should be understood that each block of the flowchart illustrations, and combinations of blocks in the flowchart illustrations, can be implemented by programming instructions contained in computer-readable memory (e.g., any menu screens described in the figures). These program instructions might be provided to a computer or other programmable data processing device (e.g., a controller, microcontroller, or processor in a sensor electronics device) to produce a machine, such that the instructions, which execute on the computer or other programmable data processing device, create means for implementing the functions specified in the flowchart block or blocks. These program instructions might also be stored in computer-readable memory that can direct a computer or other programmable data processing device to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instructions which implement the function specified in the flowchart block or blocks. The program instructions might also be loaded onto a computer or other programmable data processing device to cause a series of operational steps to be performed on the computer or other programmable device to produce a computer-implemented process such that the instructions which execute on the computer or other programmable device provide steps for implementing the functions specified in the flowchart block or blocks, and / or the menus presented herein. Program instructions can also be stored in and / or implemented by electronic circuitry, including integrated circuits (ICs) and application-specific integrated circuits (ASICs) used in conjunction with sensor devices, apparatuses, and systems.

[0180] FIG. 1 is a perspective view of a subcutaneous sensor insertion device and a block diagram of a sensor electronics device according to embodiments of the present application. As shown in FIG. 1 the subcutaneous or transcutaneous portion of the sensor set 10 includes a hollow slotted insertion needle 14 and a cannula 16. The needle 14 is used to facilitate quick and easy subcutaneous placement of the cannula 16 at a subcutaneous insertion site. A sensing portion 18 of the sensor 12 is inside the cannula 16, which is used to expose one or more sensor electrodes 20 to the bodily fluids of the user through a window 22 formed in the cannula 16. In embodiments of the present application, the one or more sensor electrodes 20 can include a counter electrode, a reference electrode, and one or more working electrodes. After insertion, the insertion needle 14 is withdrawn, leaving the cannula 16, and the sensing portion 18 and sensor electrodes 20 in place at the selected insertion site.

[0181] In particular embodiments, the subcutaneous sensor set 10 facilitates the precise placement of a flexible thin film electrochemical sensor 12 of the type used to monitor a particular blood parameter indicative of a user's condition. The sensor 12 monitors glucose levels in the body and can be used in conjunction with an external or implantable type of automated or semi-automated medication infusion pump, as described in U.S. Patent Nos. 4,562,751; 4,678,408; 4,685,903; or 4,573,994, to control insulin delivery to a diabetic patient.

[0182] Particular embodiments of the flexible electrochemical sensor 12 are constructed in accordance with thin film masking techniques to include an elongated thin film conductor embedded or coated between selected insulating materials, such as polyimide film or sheet, and a membrane. When the sensing portion 18 (or active portion) of the sensor 12 is placed subcutaneously at an insertion site, the sensor electrode 20 at the tip of the sensing portion 18 is exposed through one of the insulating layers for direct contact with the patient's blood or other bodily fluids. The sensing portion 18 is joined to a connecting portion 24 that terminates in a conductive contact pad or the like that is also exposed through one of the insulating layers. In alternative embodiments, other types of implantable sensors can be used, such as chemical-based sensors, optical-based sensors, and the like.

[0183] As is known in the art, the connecting portion 24 and contact pad are generally adapted for direct wired electrical connection to suitable monitor or sensor electronics 100 for monitoring the user's condition in response to signals derived from the sensor electrode 20. Further description of this general type of flexible thin film sensor can be found, for example, in U.S. Patent No. 5,391,250, which is incorporated herein by reference. The connecting portion 24 can be conveniently electrically connected to the monitor or sensor electronics 100, or connected through a connector block 28 (or the like) as shown and described, for example, in U.S. Patent No. 5,482,473, which is also incorporated herein by reference. Thus, in accordance with embodiments of the present application, the subcutaneous sensor set 10 can be configured or formed to work with wired or wireless property monitoring systems.

[0184] The sensor electrode 20 can be used in a variety of sensing applications and can be configured in a variety of ways. For example, the sensor electrode 20 can be used in some types of physiological parameter sensing applications in which a biomolecule is used as a catalyst. For example, the sensor electrode 20 can be used in a glucose and oxygen sensor having glucose oxidase (GOx) that catalyzes a reaction with the sensor electrode 20. The reaction produces glucose acid (C6H 12 O7) and hydrogen peroxide (H2O2) in proportion to the amount of glucose present.

[0185] The sensor electrode 20 and the biomolecules or some other catalyst can be placed in the human body in a vascular or non-vascular environment. For example, the sensor electrode 20 and the biomolecules can be placed in a vein and subjected to blood flow, or can be placed in a subcutaneous or peritoneal region of the human body.

[0186] The monitor 100 can also be referred to as a sensor electronics 100. The monitor 100 can include a power source 110, a sensor interface 122, processing electronics 124, and data formatting electronics 128. The monitor 100 can be coupled to the sensor set 10 with a cable 102 through a connector of the connector block 28 of the connection portion 24. In alternative embodiments, the cable can be omitted. In this embodiment of the present application, the monitor 100 can include a connector adapted to directly connect to the connector portion 104 of the sensor set 10. The sensor set 10 can be modified to have the connector portion 104 located in a different location, such as on top of the sensor set, to facilitate placement of the monitor 100 over the sensor set.

[0187] In embodiments of the present application, the sensor interface 122, the processing electronics 124, and the data formatting electronics 128 are formed as separate semiconductor chips, however, alternative embodiments can combine the various semiconductor chips into a single or multiple custom semiconductor chips. The sensor interface 122 is connected with the cable 102, which is connected with the sensor set 10.

[0188] The power source 110 can be a battery. The battery can include three silver oxide 357 batteries connected in series. In alternative embodiments, different battery chemistries can be used, such as lithium-based chemistries, alkaline batteries, nickel metal hydride, etc., and different numbers of batteries can be used. The monitor 100 provides power to the sensor set through the power source 110 via the cable 102 and the cable connector 104. In embodiments of the present application, the power is a voltage provided to the sensor set 10. In embodiments of the present application, the power is a current provided to the sensor set 10. In embodiments of the present application, the power is a voltage provided to the sensor set 10 at a particular voltage.

[0189] FIG. 2A and 2B An implantable sensor and electronics for driving the implantable sensor according to embodiments of the present application are shown. FIG. 2A A substrate 220 with two sides is shown, a first side 222 containing an electrode configuration and a second side 224 containing electronic circuitry. As in FIG. 2AAs seen in the figure, the first side 222 of the substrate includes two pairs of electrodes - working electrode pairs 240, 242, 244, 246 on opposite sides of a reference electrode 248. The second side 224 of the substrate includes electronic circuitry. As shown, the electronic circuitry can be enclosed in a hermetically sealed housing 226, providing a protective shell for the electronic circuitry. This allows the sensor substrate 220 to be inserted into a vascular environment or other environment that can subject the electronic circuitry to fluids. By sealing the electronic circuitry in a hermetically sealed housing 226, the electronic circuitry can operate without the risk of being shorted out by surrounding fluids. FIG. 2A Also shown in the figure are pads 228 to which input and output lines of the electronic circuitry can be connected. The electronic circuitry itself can be fabricated in a variety of ways. According to embodiments of the present application, the electronic circuitry can be fabricated as an integrated circuit using techniques common in the industry.

[0190] FIG. 2B A general block diagram of an electronic circuit for sensing the output of a sensor according to embodiments of the present application is shown. At least one pair of sensor electrodes 310 can interface with a data converter 312, the output of which can interface with a counter 314. The counter 314 can be controlled by control logic 316. The output of the counter 314 can be connected to a line interface 318. The line interface 318 can be connected to input and output lines 320, and can also be connected to the control logic 316. The input and output lines 320 can also be connected to a power rectifier 322.

[0191] The sensor electrodes 310 can be used in a variety of sensing applications, and can be configured in a variety of ways. For example, the sensor electrodes 310 can be used in some types of physiological parameter sensing applications in which a biomolecule is used as a catalyst. For example, the sensor electrodes 310 can be used in a glucose and oxygen sensor having glucose oxidase (GOx) that catalyzes a reaction with the sensor electrodes 310. The sensor electrodes 310, along with the biomolecule or some other catalyst, can be placed in a human body in a vascular or non-vascular environment. For example, the sensor electrodes 310 and biomolecule can be placed in a vein and subjected to blood flow.

[0192] FIG. 3A block diagram of a sensor electronics device and a sensor comprising a plurality of electrodes according to embodiments of the application is shown. The sensor set or system 350 comprises a sensor 355 and a sensor electronics device 360. The sensor 355 comprises a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronics device 360 comprises a power source 380, a regulator 385, a signal processor 390, a measurement processor 395, and a display / transmission module 397. The power source 380 provides power (in the form of voltage, current, or voltage comprising current) to the regulator 385. The regulator 385 transmits a regulated voltage to the sensor 355. In embodiments of the application, the regulator 385 transmits a voltage to the counter electrode 365 of the sensor 355.

[0193] The sensor 355 generates a sensor signal indicative of the concentration of the physiological property being measured. For example, the sensor signal can be indicative of a blood glucose reading. In embodiments of the application utilizing a subcutaneous sensor, the sensor signal can be representative of the level of hydrogen peroxide in the subject. In embodiments of the application utilizing a blood or cranial sensor, the amount of oxygen is measured by the sensor and represented by the sensor signal. In embodiments of the application utilizing an implantable or long-term sensor, the sensor signal can be representative of the level of oxygen in the subject. The sensor signal can be measured at the working electrode 375. In embodiments of the application, the sensor signal can be a current measured at the working electrode. In embodiments of the application, the sensor signal can be a voltage measured at the working electrode.

[0194] After the sensor signal (e.g., the measured current or voltage) is measured at the sensor 355 (e.g., the working electrode), the signal processor 390 receives the sensor signal. The signal processor 390 processes the sensor signal and generates a processed sensor signal. The measurement processor 395 receives the processed sensor signal and calibrates the processed sensor signal with a reference value. In embodiments of the application, the reference value is stored in a reference memory and provided to the measurement processor 395. The measurement processor 395 generates a sensor measurement. The sensor measurement can be stored in a measurement memory (not shown). The sensor measurement can be sent to a display / transmission device for display on a display in the housing along with the sensor electronics or transmitted to an external device.

[0195] The sensor electronics 360 can be a monitor that includes a display to display physiological characteristic readings. The sensor electronics 360 can also be installed in a desktop computer, a pager, a television that includes communication capabilities, a notebook computer, a server, a network computer, a personal digital assistant (PDA), a portable telephone that includes computer functionality, an infusion pump that includes a display, a glucose sensor that includes a display, and / or a combination infusion pump / glucose sensor. The sensor electronics 360 can be housed in a cellular telephone, a smart phone, a network appliance, a home network appliance, and / or other device connected to a home network.

[0196] FIG. 4 An alternative embodiment including a sensor and sensor electronics is shown. The sensor set or sensor system 400 includes the sensor electronics 360 and the sensor 355. The sensor includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensor electronics 360 includes a microcontroller 410 and a digital-to-analog converter (DAC) 420. The sensor electronics 360 can also include a current-to-frequency converter (I / F converter) 430.

[0197] The microcontroller 410 includes software program code or programmable logic that, when executed, causes the microcontroller 410 to transmit a signal to the DAC 420, where the signal represents a voltage level or voltage value to be applied to the sensor 355. The DAC 420 receives the signal and generates a voltage value at the level indicated by the microcontroller 410. In embodiments of the present application, the microcontroller 410 can change the representation of the voltage level in the signal frequently or infrequently. Illustratively, the signal from the microcontroller 410 can instruct the DAC 420 to apply a first voltage value for one second and a second voltage value for two seconds.

[0198] The sensor 355 can receive a voltage level or voltage value. In embodiments of the application, the counter electrode 365 can receive the output of an operational amplifier having as inputs a reference voltage from the DAC 420 and the voltage value. The application of the voltage level causes the sensor 355 to generate a sensor signal indicative of the concentration of the physiological characteristic being measured. In embodiments of the application, the microcontroller 410 can measure the sensor signal (e.g., current value) from the working electrode. Illustratively, the sensor signal measurement circuit 431 can measure the sensor signal. In embodiments of the application, the sensor signal measurement circuit 431 can include a resistor, and a current can flow through the resistor to measure the value of the sensor signal. In embodiments of the application, the sensor signal can be a current level signal, and the sensor signal measurement circuit 431 can be a current-to-frequency (I / F) converter 430. The current-to-frequency converter 430 can measure the sensor signal from the current reading, convert the sensor signal to a frequency-based sensor signal, and transmit the frequency-based sensor signal to the microcontroller 410. In embodiments of the application, the microcontroller 410 can be able to more easily receive the frequency-based sensor signal than a non-frequency-based sensor signal. The microcontroller 410 receives the frequency-based or non-frequency-based sensor signal and determines a value of the physiological characteristic of the subject, e.g., a blood glucose level. The microcontroller 410 can include program code that, when executed or run, is able to receive the sensor signal and convert the sensor signal to a physiological characteristic value. In one embodiment, the microcontroller 410 can convert the sensor signal to a blood glucose level. In some embodiments, the microcontroller 410 can utilize measurement values stored in internal memory in order to determine the blood glucose level of the subject. In some embodiments, the microcontroller 410 can utilize measurement values stored in memory external to the microcontroller 410 to assist in determining the blood glucose level of the subject.

[0199] After the microcontroller 410 determines the physiological characteristic value, the microcontroller 410 can store the measurement of the physiological characteristic value for a number of time periods. For example, a blood glucose value can be sent from the sensor to the microcontroller 410 every second or every five seconds, and the microcontroller can save the sensor measurement for five or ten minutes of the BG reading. The microcontroller 410 can communicate the measurement of the physiological characteristic value to a display on the sensor electronics 360. For example, the sensor electronics 360 can be a monitor that includes a display that provides a blood glucose reading of the subject. In one embodiment, the microcontroller 410 can communicate the measurement of the physiological characteristic value to an output interface of the microcontroller 410. The output interface of the microcontroller 410 can communicate the measurement of the physiological characteristic value, e.g., a blood glucose value, to an external device, such as an infusion pump, a combination infusion pump / blood glucose meter, a computer, a personal digital assistant, a pager, a network appliance, a server, a cellular telephone, or any computing device.

[0200] FIG. 5 An electronic diagram of a sensor electrode and the voltage applied to the sensor electrode is shown in accordance with one embodiment. In FIG. 5 In the illustrated embodiment, an operational amplifier 530 or other servo control device can be connected to the sensor electrode 510 through a circuit / electrode interface 538. The operational amplifier 530, using feedback through the sensor electrode, attempts to maintain a specified voltage between the reference electrode 532 and the working electrode 534 by adjusting the voltage at the counter electrode 536 (the DAC can desire that the voltage applied be the specified voltage). Current can then flow from the counter electrode 536 to the working electrode 534. This current can be measured to determine an electrochemical reaction between the sensor electrode 510 and biological molecules of a sensor that have been placed near the sensor electrode 510 and serve as a catalyst. FIG. 5 The circuitry disclosed in the background can be used in long-term or implantable sensors, or can be used in short-term or subcutaneous sensors.

[0201] In embodiments of long-term sensors, where glucose oxidase (GOx) enzymes are used as a catalyst in the sensor, current can flow from the counter electrode 536 to the working electrode 534 only when oxygen is present near the enzyme and sensor electrode 510. Illustratively, if the voltage set at the reference electrode 532 is maintained at about 0.5 volts, the amount of current flowing from the counter electrode 536 to the working electrode 534 has a fairly linear relationship with the amount of oxygen present in the region surrounding the enzyme and electrode. Thus, by maintaining the reference electrode 532 at about 0.5 volts and utilizing this region of the current-voltage curve, the accuracy of determining the amount of oxygen in the blood can be improved. Different embodiments can utilize different sensors with biological molecules other than glucose oxidase, and thus can set voltages at the reference electrode other than 0.5 volts.

[0202] As discussed above, during the initial implantation or insertion of the sensor 510, the sensor 510 can provide inaccurate readings due to the subject's adjustment to the sensor and the electrochemical byproducts produced by the catalyst used in the sensor. Many sensors require a stabilization period in order for the sensor 510 to provide accurate readings of the subject's physiological parameter. During the stabilization period, the sensor 510 does not provide accurate blood glucose measurements. Users and manufacturers of sensors can desire to improve the stabilization time frame of the sensor so that the sensor can be utilized quickly after being inserted into the subject or the subject's subcutaneous layer.

[0203] In previous sensor electrode systems, the stabilization period or time frame can range from one to three hours. To reduce the stabilization period or time frame and improve the timeliness of the accuracy of the sensor, instead of applying one pulse to the sensor (or electrode of the sensor), another voltage is applied. FIG. 6A One method of applying pulses during the stabilization time frame to reduce the stabilization time frame is shown. In this embodiment, the voltage applying device applies 600 a first voltage to the electrode for a first time or time period. In one embodiment, the first voltage can be a DC constant voltage. This causes an anodic current to be generated. In an alternative embodiment, a digital to analog converter or another voltage source can supply a voltage to the electrode for the first time period. An anodic current means that electrons are driven to the electrode to which the voltage is applied. In certain embodiments, the applying device can apply a current instead of a voltage. In embodiments where a voltage is applied to the sensor, after the first voltage is applied to the electrode, the voltage regulator can wait (i.e., not apply a voltage) for a second time, time frame, or time period 605. In other words, the voltage applying device waits until the second time period has passed. Not applying a voltage causes a cathodic current, which causes the electrode to which the voltage is not applied to gain electrons. This is repeated 610 applying the first voltage to the electrode for the first time period and then not applying a voltage for the second time period for a number of iterations. This can be referred to as an anodic and cathodic cycle. In one embodiment, the total number of iterations of the stabilization method is three, i.e., the voltage is applied for the first time period three times and not applied for the second time period after each application. In embodiments, the first voltage can be 1.07 volts. In further embodiments, the first voltage can be 0.535 volts, or can be approximately 0.7 volts.

[0204] Repeating the application of the voltage and not applying the voltage causes the sensor (and thus the electrode) to undergo an anodic-cathodic cycle. The anodic-cathodic cycle causes a reduction in electrochemical byproducts that are generated due to the reaction of the patient’s body to the insertion of the sensor or the implantation of the sensor. The electrochemical byproducts cause the generation of a background current, which results in inaccurate measurements of the physiological parameter of the subject. Under certain operating conditions, the electrochemical byproducts can be removed. Under other operating conditions, the electrochemical byproducts can be reduced or significantly reduced. A successful stabilization method will cause the anodic-cathodic cycle to reach equilibrium, the electrochemical byproducts to be significantly reduced, and the background current to be minimized.

[0205] In one embodiment, the first voltage applied to the electrode of the sensor can be a positive voltage. In alternative embodiments, the first voltage applied can be a negative voltage. Further, the first voltage can be applied to the working electrode. In some embodiments, the first voltage can be applied to the counter electrode or the reference electrode.

[0206] In some embodiments, the duration of the voltage pulse and the duration of no voltage application can be equal, for example, three minutes each. In other embodiments, the duration of the voltage application or the voltage pulse can be different values, for example, the first time and the second time can be different. In one embodiment, the first time period can be five minutes and the wait time period can be two minutes. In a variation, the first time period can be two minutes and the wait time period (or second time frame) can be five minutes. In other words, the duration of the first voltage application can be two minutes and no voltage can be applied for five minutes. This time frame is illustrative and should not be limiting. For example, the first time frame can be two minutes, three minutes, five minutes, or ten minutes and the second time frame can be five minutes, ten minutes, twenty minutes, etc. The time frames (e.g., the first time and the second time) can depend on the unique characteristics of the different electrodes, the sensor, and / or the physiological characteristics of the patient.

[0207] In connection with the foregoing, more or less than three pulses can be utilized to stabilize the glucose sensor. In other words, the number of iterations can be greater than 3 or less than 3. For example, four voltage pulses (e.g., high voltage followed by no voltage application) can be applied to one of the electrodes or six voltage pulses can be applied to one of the electrodes.

[0208] Illustratively, three consecutive 1.07 volt pulses (followed by the respective wait times) can be sufficient for a subcutaneously implanted sensor. In one embodiment, three consecutive 0.7 volt voltage pulses can be utilized. For a sensor implanted into the blood or cranial fluid, such as a long-term sensor or a permanent sensor, the three consecutive pulses can have a higher or lower voltage value: a negative voltage or a positive voltage. Additionally, more than three pulses (e.g., five, eight, twelve) can be utilized to create an anode-cathode cycle between the anode current and the cathode current in any of the subcutaneous, blood, or cranial fluid sensors.

[0209] FIG. 6B A method of stabilizing a sensor according to an embodiment of the present application is shown. In FIG. 6B In the illustrated embodiment, a voltage application device can apply 630 a first voltage to the sensor for a first time to start an anodic cycle at the electrodes of the sensor. The voltage application device can be a DC power source, a digital-to-analog converter, or a voltage regulator. After the first time period elapses, a second voltage is applied 635 to the sensor for a second time to start a cathodic cycle at the electrodes of the sensor. Illustratively, as shown in FIG. 6, the first voltage is a positive voltage and the second voltage is a negative voltage. In other embodiments, the first voltage can be a negative voltage and the second voltage can be a positive voltage. In other words, the first voltage and the second voltage can be of opposite polarity. FIG. 6AIn the method shown in FIG. 6, a different voltage (different than the first voltage) is applied to the sensor during the second time frame, rather than no voltage being applied. In embodiments of the present application, the application of the first voltage for the first time and the second voltage for the second time is repeated for several iterations. In certain embodiments, the application of the first voltage for the first time and the second voltage for the second time can each be applied for a stabilization time frame, such as 10 minutes, 15 minutes, or 20 minutes, rather than several iterations. This stabilization time frame is the entire time frame of the stabilization sequence, such as until the sensor (and electrodes) is stabilized. The benefit of this stabilization method is that the sensor is broken in faster, has less background current (in other words, some background current is suppressed), and has a better glucose response.

[0210] In one embodiment, the first voltage can be 0.535 volts applied for five minutes, the second voltage can be 1.070 volts applied for two minutes, the first voltage of 0.535 volts can be applied for five minutes, the second voltage of 1.070 volts can be applied for two minutes, the first voltage of 0.535 volts can be applied for five minutes, and the second voltage of 1.070 volts can be applied for two minutes. In other words, in this embodiment, there are three iterations of the voltage pulse scheme. The pulse method can be varied, as the second time frame, such as the time frame of applying the second voltage, can be extended from two minutes to five minutes, ten minutes, fifteen minutes, or twenty minutes. Additionally, in this embodiment of the present application, after the three iterations are applied, a nominal operating voltage of 0.535 volts can be applied.

[0211] 1.070 and 0.535 volts are illustrative values. Other voltage values can be selected based on a variety of factors. These factors can include the type of enzyme utilized in the sensor, the membrane utilized in the sensor, the operating period of the sensor, the length of the pulse, and / or the magnitude of the pulse. In certain operating conditions, the first voltage can be in the range of 1.00 to 1.09 volts, and the second voltage can be in the range of 0.510 to 0.565 volts. In other operating embodiments, the range including the first voltage and the second voltage can have a higher range, such as 0.3 volts, 0.6 volts, 0.9 volts, depending on the voltage sensitivity of the electrodes in the sensor. In other operating conditions, the voltage can be in the range of 0.8 volts to 1.34 volts, and the other voltage can be in the range of 0.335 to 0.735. In other operating conditions, the range of the higher voltage can be less than the range of the lower voltage. Illustratively, the higher voltage can be in the range of 0.9 to 1.09 volts, and the lower voltage can be in the range of 0.235 to 0.835 volts.

[0212] In embodiments, the first voltage and the second voltage can be positive voltages, or in other embodiments, can be negative voltages. In another embodiment, the first voltage can be positive and the second voltage can be negative, or, the first voltage can be negative and the second voltage can be positive. For each iteration, the first voltage can be a different voltage level. In addition, the first voltage can be a DC constant voltage. Further, the first voltage can be a ramp voltage, a sinusoidal voltage, a step voltage, or other commonly used voltage waveforms. In embodiments, the second voltage can be a DC constant voltage, a ramp voltage, a sinusoidal voltage, a step voltage, or other commonly used voltage waveforms. In alternative embodiments, the first voltage or the second voltage can be an AC signal on a DC waveform. Generally, the first voltage can be one type of voltage, such as a ramp voltage, and the second voltage can be a second type of voltage, such as a sinusoidal voltage, and the first voltage (or the second voltage) can have a different waveform shape for each iteration. For example, if there are three loops in the stabilization method, then in the first loop, the first voltage can be a ramp voltage, in the second loop, the first voltage can be a constant voltage, and in the third loop, the first voltage can be a sinusoidal voltage.

[0213] In embodiments, the duration of the first time frame and the duration of the second time frame can have the same value, or, the duration of the first time frame and the second time frame can have different values. For example, the duration of the first time frame can be two minutes and the duration of the second time frame can be five minutes, and the number of iterations can be three. As discussed above, the stabilization method can include a number of iterations. In various embodiments, the duration of each of the first time frames can change and the duration of each of the second time frames can change during different iterations of the stabilization method. Illustratively, during the first iteration of the anodic-cathodic cycle, the first time frame can be 2 minutes and the second time frame can be 5 minutes. During the second iteration, the first time frame can be 1 minute and the second time frame can be 3 minutes. During the third iteration, the first time frame can be 3 minutes and the second time frame can be 10 minutes.

[0214] In one embodiment, a first voltage of 0.535 volts is applied to the electrodes in the sensor for two minutes to start an anodic cycle, then a second voltage of 1.07 volts is applied to the electrodes for five minutes to start a cathodic cycle. Then the first voltage of 0.535 volts is applied again for two minutes to start an anodic cycle and a second voltage of 1.07 volts is applied to the sensor for five minutes. In the third iteration, 0.535 volts is applied for two minutes to start an anodic cycle, then 1.07 volts is applied for five minutes. Next, during the actual working time frame of the sensor, for example, when the sensor is providing readings of a physiological characteristic of a subject, the voltage applied to the sensor is 0.535.

[0215] In FIG. 6A and FIG. 6B embodiments, shorter duration voltage pulses can be utilized. The shorter duration voltage pulses can be used to apply the first voltage, the second voltage, or both. In one embodiment, the magnitude of the shorter duration voltage pulses for the first voltage is -1.07 volts, and the magnitude of the shorter duration voltage pulses for the second voltage is approximately half of the high magnitude, e.g., -.535 volts. Alternatively, the magnitude of the shorter duration pulses for the first voltage can be 0.535 volts, and the magnitude of the shorter duration pulses for the second voltage is 1.07 volts.

[0216] In embodiments utilizing short duration pulses, the voltage can be applied discontinuously throughout the first time period. Alternatively, the voltage application device can transmit several short duration pulses during the first time period. In other words, several small width or short duration voltage pulses can be applied to the electrodes of the sensor over the first time period. Each small width or short duration pulse can have a width of several milliseconds. Illustratively, this pulse width can be 30 milliseconds, 50 milliseconds, 70 milliseconds, or 200 milliseconds. These values are intended to be illustrative and not limiting. In one embodiment, for example, in the embodiment shown in FIG. 2, these short duration pulses are applied to the sensor (electrodes) over the first time period, and then no voltage is applied over the second time period. FIG. 6A

[0217] Each short duration pulse can have the same duration over the first time period. For example, each short duration voltage pulse can have a temporal width of 50 milliseconds, and each pulse delay between pulses can be 950 milliseconds. In this example, if the measured time of the first time frame is two minutes, then 120 short duration voltage pulses can be applied to the sensor. Alternatively, each of the short duration voltage pulses can have a different duration. In various embodiments, each of the short duration voltage pulses can have the same amplitude value, or can have different amplitude values. By utilizing short duration voltage pulses instead of continuously applying a voltage to the sensor, the same anodic and cathodic cycles can be performed, and the total energy or charge of the sensor (e.g., electrodes) is reduced over time. Less power is utilized using short duration voltage pulses as compared to applying a continuous voltage to the electrodes, as less energy is applied to the sensor (and electrodes).

[0218] FIG. 6C ​The use of feedback in stabilizing a sensor is shown in accordance with one embodiment. A sensor system can include a feedback mechanism for determining if additional pulses are needed to stabilize the sensor. In one embodiment, a sensor signal generated by an electrode (e.g., a working electrode) can be analyzed to determine if the sensor signal is stable. A first voltage is applied 630 to the electrode for a first time frame to begin an anodic cycle. A second voltage is applied 635 to the electrode for a second time frame to begin a cathodic cycle. In embodiments of the present invention, an analysis module can analyze the sensor signal (e.g., the current emitted by the sensor signal, the resistance at a particular point in the sensor, the impedance at a particular node in the sensor) and determine if a threshold measurement has been reached 637 (e.g., determine if the sensor is providing accurate readings by comparing to a threshold measurement). If the sensor readings are determined to be accurate, this indicates that the electrode (and thus the sensor) has stabilized 642, then no additional first and / or second voltages can be applied. If stability is not achieved, then additional anodic / cathodic cycles can be started by applying 630 the first voltage to the electrode for a first time period and then applying 635 the second voltage to the electrode for a second time period.

[0219] In some embodiments, the analysis module can be employed after three anodic / cathodic cycles of applying the first voltage and the second voltage to the electrodes of the sensor. However, the analysis module can be employed after one application of the first voltage and the second voltage, as shown. FIG. 6C

[0220] The analysis module can be utilized to measure the voltage emitted after a current has been introduced across one electrode or across a pair of electrodes. The analysis module can monitor the voltage level at the electrode or at the receiving level. In one embodiment, if the voltage level is above a certain threshold, this can mean that the sensor has stabilized. In one embodiment, if the voltage level drops below a threshold level, this can indicate that the sensor has stabilized and is ready to provide readings. In one embodiment, a current can be introduced to an electrode or across a pair of electrodes. The analysis module can monitor the current level emitted from the electrode. In this embodiment, the analysis module is able to monitor the current if the current differs from the sensor signal current by an order of magnitude. If the current is above or below a current threshold, this can indicate that the sensor has stabilized.

[0221] ​In embodiments of the application, the analysis module can measure the impedance between two electrodes of the sensor. The analysis module can compare the impedance to a threshold or target impedance value, and if the measured impedance is below the target or threshold impedance, then the sensor (and thus the sensor signal) can be stabilized. In one embodiment, the analysis module can measure the electrical resistance between two electrodes of the sensor. In this embodiment of the application, if the analysis module compares the electrical resistance to a threshold or target resistance value and the measured resistance value is less than the threshold or target resistance value, then the analysis module can determine that the sensor is stabilized and that the sensor signal can be utilized.

[0222] FIG. 7 The effect of stabilizing a sensor according to embodiments of the application is shown. Line 705 represents the blood glucose sensor readings of a glucose sensor utilizing a previous single pulse stabilization method. Line 710 represents the blood glucose readings of a glucose sensor to which three voltage pulses are applied (e.g., 3 voltage pulses of 2 minutes in duration, with no voltage applied for 5 minutes after each voltage pulse). The x-axis 715 represents the amount of time. Points 720, 725, 730, and 735 represent the measured glucose readings, which are taken with a finger prick and then input into a blood glucose meter. As shown by the graph, the previous single pulse stabilization method takes approximately 1 hour and 30 minutes to stabilize to a desired glucose reading, such as 100 units. In contrast, the three pulse stabilization method takes only approximately 15 minutes to stabilize the glucose sensor and significantly improves the stabilization time frame.

[0223] FIG. 8A A block diagram of a sensor electronics device and a sensor containing a voltage generating device is shown. The voltage generating or applying device 810 contains electronics, logic, or circuitry that generates voltage pulses. The sensor electronics device 360 can also contain an input device 820 for receiving reference values and other useful data. In one embodiment, the sensor electronics device can contain a measurement memory 830 for storing sensor measurements. In this embodiment, a power source 380 can power the sensor electronics device. The power source 380 can power a regulator 385, which supplies a regulated voltage to the voltage generating or applying device 810. The connection terminal 811 represents that in the embodiment shown, the connection terminal couples or connects the sensor 355 to the sensor electronics device 360.

[0224] In FIG. 8AIn the illustrated embodiment, voltage generating or applying device 810 supplies a voltage, such as a first voltage or a second voltage, to an input terminal of operational amplifier 840. Voltage generating or applying device 810 can also supply voltage to the working electrode 375 of sensor 355. Another input terminal of operational amplifier 840 is connected to the reference electrode 370 of the sensor. Applying voltage from voltage generating or applying device 810 to operational amplifier 840 will drive the voltage measured at counter electrode 365 to be close to or equal to the voltage applied at working electrode 375. In this embodiment, voltage generating or applying device 810 can be used to apply a desired voltage between counter electrode and working electrode. This can be done by applying a fixed voltage directly to counter electrode.

[0225] In such FIG. 6A and FIG. 6B In one embodiment shown, voltage generating device 810 generates a first voltage to be applied to the sensor during a first time frame. Voltage generating device 810 transmits this first voltage to operational amplifier 840, which drives the voltage at the counter electrode 365 of sensor 355 to the first voltage. In some embodiments, voltage generating device 810 may also transmit the first voltage directly to the counter electrode 365 of sensor 355. FIG. 6A In the illustrated embodiment, the voltage generating device 810 then does not transmit the first voltage to the sensor 355 during the second time frame. In other words, the voltage generating device 810 is turned off or disconnected. The voltage generating device 810 can be programmed to continue cycling between applying and not applying the first voltage over several iterations or stable time frames, for example, over twenty minutes. FIG. 8B A voltage generating apparatus for implementing this embodiment of the invention is shown. A voltage regulator 385 delivers a regulated voltage to the voltage generating apparatus 810. A control circuit 860 controls the closing and opening of a switch 850. If switch 850 is closed, then a voltage is applied. If switch 850 is open, then no voltage is applied. A timer 865 provides a signal to the control circuit 860 to instruct the control circuit 860 to turn switch 850 on and off. The control circuit 860 includes logic that can instruct the circuit to close and open switch 850 a number of times (to match necessary iterations). In one embodiment, the timer 865 may also transmit a stabilization signal to identify that a stabilization sequence has been completed, i.e., a stabilization time frame has elapsed.

[0226] In one embodiment, the voltage generating device generates a first voltage in a first time frame and a second voltage in a second time frame. FIG. 8CA voltage generating device for generating two voltage values to implement this embodiment is shown. In this embodiment, a two position switch 870 is utilized. Illustratively, if the control circuit 860 is instructed by the timer 865 to turn on or close the first switch position 871, the voltage generating device 810 generates a first voltage for a first time frame. After the first voltage has been applied for the first time frame, the timer sends a signal to the control circuit 860 that the first time frame has elapsed and the control circuit 860 directs the switch 870 to move to the second position 872. When the switch 870 is in the second position 872, the regulated voltage is directed to a voltage step down or buck converter 880 to reduce the regulated voltage to a smaller value. The smaller value is then delivered to the operational amplifier 840 for a second time frame. After the timer 865 has sent a signal to the control circuit 860 that the second time period has elapsed, the control circuit 860 moves the switch 870 back to the first position. This continues until the desired number of iterations have been completed or the sensor stabilization time frame has elapsed. In embodiments of the present invention, after the sensor stabilization time frame has elapsed, the sensor transmits the sensor signal 350 to the signal processor 390.

[0227] FIG. 8D A voltage application device 810 for performing a more complex application of voltage to the sensor is shown. The voltage application device 810 can include a control device 860, a switch 890, a sinusoidal voltage generating device 891, a ramp voltage generating device 892, and a constant voltage generating device 893. In other embodiments, the voltage application can generate an AC wave on top of a DC signal or other various voltage pulse waveforms. In this embodiment, the control device 860 is instructed by the timer 865 to turn on or close the first switch position 891. The sinusoidal voltage generating device 891 generates a sinusoidal voltage for a first time frame. After the sinusoidal voltage has been applied for the first time frame, the timer sends a signal to the control circuit 860 that the first time frame has elapsed and the control circuit 860 directs the switch 890 to move to the second position 892. When the switch 890 is in the second position 892, the regulated voltage is directed to a voltage step down or buck converter 880 to reduce the regulated voltage to a smaller value. The smaller value is then delivered to the operational amplifier 840 for a second time frame. After the timer 865 has sent a signal to the control circuit 860 that the second time period has elapsed, the control circuit 860 moves the switch 890 back to the first position. This continues until the desired number of iterations have been completed or the sensor stabilization time frame has elapsed. In embodiments of the present invention, after the sensor stabilization time frame has elapsed, the sensor transmits the sensor signal 350 to the signal processor 390. FIG. 8DIn the illustrated embodiment, the control device 860 can cause the switch to move to one of three voltage generation systems 891 (sine curve), 892 (ramp), 893 (constant DC). This causes each of the voltage generation systems to generate the identified voltage waveform. Under certain operating conditions, for example, where a sine pulse is to be applied for three pulses, the control device 860 can cause the switch 890 to connect the voltage from the voltage regulator 385 to the sine voltage generator 891 in order for the voltage application device 810 to generate a sine voltage. Under other operating conditions, for example, when a ramp voltage is to be applied as the first voltage for a first pulse of three pulses to the sensor, a sine voltage is to be applied as the first voltage for a second pulse of three pulses to the sensor, and a constant DC voltage is to be applied as the first voltage for a third pulse of three pulses to the sensor, the control device 860 can cause the switch 890 to move between, during a first time frame in the anode / cathode cycle, connecting the voltage from the voltage generation or application device 810 to the ramp voltage generation system 892, then to the sine voltage generation system 891, and then to the constant DC voltage generation system 893. In this embodiment, the control device 860 can also direct or control the switch to connect certain voltage generation subsystems to the voltage from the regulator 385 during a second time frame, for example, during the application of the second voltage.

[0228] FIG. 9A A sensor electronics device is shown that includes a microcontroller for generating voltage pulses. An advanced sensor electronics device can include a microcontroller 410 (see FIG. 4 ), a digital-to-analog converter (DAC) 420, an operational amplifier 840, and a sensor signal measurement circuit 431. In one embodiment, the sensor signal measurement circuit can be a current-to-frequency (I / F) converter 430. In FIG. 9A In the illustrated embodiment, software or programmable logic in the microcontroller 410 provides instructions for transmitting a signal to the DAC 420, which in turn instructs the DAC 420 to output a particular voltage to the operational amplifier 840. As FIG. 9Athe working electrode 375. As discussed above, applying a particular voltage to the operational amplifier 840 and the working electrode 375 can drive the voltage measured at the counter electrode to a particular voltage magnitude. In other words, the microcontroller 410 outputs a signal indicating the voltage or voltage waveform to be applied to the sensor 355 (e.g., the operational amplifier 840 coupled to the sensor 355). In alternative embodiments, a fixed voltage can be set by directly applying a voltage from the DAC 420 between the reference electrode and the working electrode 375. Similar results can also be obtained by applying a voltage to each of the electrodes, the voltage difference being equal to the fixed voltage applied between the reference electrode and the working electrode. Additionally, a fixed voltage can be set by applying a voltage between the reference electrode and the counter electrode. Under certain operating conditions, the microcontroller 410 can generate a pulse of a particular magnitude that the DAC 420 interprets as indicating that a voltage of the particular magnitude is to be applied to the sensor. After a first time frame, the microcontroller 410 (via programming or programmable logic) outputs a second signal that indicates that the DAC 420 is to output no voltage (for a sensor electronic device 360 operating according to the method described in FIG. 6A FIG. 6B After the second time frame has elapsed, the microcontroller 410 then repeats the cycle of sending a signal indicating a first voltage to be applied (for a first time frame) and then sending the signal to indicate no voltage is to be applied or a second voltage is to be applied (for a second time frame).

[0229] Under other operating conditions, the microcontroller 410 can generate a signal to the DAC 420 indicating that the DAC outputs a ramping voltage. Under other operating conditions, the microcontroller 410 can generate a signal to the DAC 420 indicating that the DAC 420 outputs a voltage of an analog sinusoidal voltage. These signals can be incorporated into any of the pulsing methods discussed above in the previous paragraphs or earlier in the application. In one embodiment, the microcontroller 410 can generate a series of instructions and / or pulses that the DAC 420 receives and interprets as meaning that a particular sequence of pulses is to be applied. For example, the microcontroller 410 can transmit a series of instructions (via signals and / or pulses) indicating that the DAC 420 is to generate a constant voltage for a first iteration of a first time frame, a ramping voltage for a first iteration of a second time frame, a sinusoidal voltage for a second iteration of the first time frame, and a square wave with two values for a second iteration of the second time frame.

[0230] ​The microcontroller 410 can include programmable logic or programming for continuing this cycle for a number of iterations within a stabilization timeframe or for a number of iterations. Illustratively, the microcontroller 410 can include counting logic for identifying when the first timeframe or the second timeframe has elapsed. Additionally, the microcontroller 410 can include counting logic for identifying when the stabilization timeframe has elapsed. After any of the aforementioned timeframes have elapsed, the counting logic can instruct the microcontroller to send a new signal or to stop transmitting signals to the DAC 420.

[0231] The use of the microcontroller 410 allows for the application of a number of voltage magnitudes in a number of sequences over a number of durations. In an embodiment of the present application, the microcontroller 410 can include control logic or programming for instructing the digital-to-analog converter 420 to transmit a voltage pulse having a magnitude of approximately 1.0 volts for a first period of 1 minute and then transmit a voltage pulse having a magnitude of approximately 0.5 volts for a second period of 4 minutes, and repeat this cycle for four iterations. In one embodiment, the microcontroller 420 can be programmed to transmit signals to cause the DAC 420 to apply the same magnitude of voltage pulse for each first voltage in each iteration. The microcontroller 410 can be programmed to transmit signals to cause the DAC to apply a different magnitude of voltage pulse for each first voltage in each iteration. In this embodiment, the microcontroller 410 can also be programmed to transmit signals to cause the DAC 420 to apply a different magnitude of voltage pulse for each second voltage in each iteration. Illustratively, the microcontroller 410 can be programmed to transmit signals to cause the DAC 420 to apply a first voltage pulse of approximately 1.0 volts in the first iteration, a second voltage pulse of approximately 0.5 volts in the first iteration, a first voltage of 0.7 volts and a second voltage of 0.4 volts in the second iteration, and a first voltage of 1.2 volts and a second voltage of 0.8 volts in the third iteration.

[0232] The microcontroller 410 can also be programmed to instruct the DAC 420 to provide a number of short duration voltage pulses within the first time frame. In this embodiment of the present application, a number of shorter duration pulses can be applied to the sensor instead of one voltage for the entire first time frame (e.g., two minutes). In this embodiment, the microcontroller 410 can also be programmed to instruct the DAC 420 to provide a number of short duration voltage pulses to the sensor within the second time frame. Illustratively, the microcontroller 410 can send a signal to cause the DAC to apply a number of short duration voltage pulses, where the short duration is 50 milliseconds or 100 milliseconds. Between these short duration pulses, the DAC can not apply a voltage, or the DAC can apply a minimum voltage. The microcontroller can cause the DAC 420 to apply short duration voltage pulses within the first time frame (e.g., two minutes). The microcontroller 410 can then send a signal to cause the DAC to not apply any voltage or to apply short duration voltage pulses to the sensor within the second time frame at a second magnitude of voltage, e.g., the second voltage can be 0.75 volts and the second time frame can be 5 minutes. In one embodiment, the microcontroller 410 can send a signal to the DAC 420 to cause the DAC 420 to apply a different magnitude voltage for each short duration pulse in the first time frame and / or the second time frame. In an embodiment, the microcontroller 410 can send a signal to the DAC 420 to cause the DAC 420 to apply a pattern of voltage magnitudes to the short duration voltage pulses within the first time frame or the second time frame. For example, the microcontroller can transmit a signal or a pulse instructing the DAC 420 to apply thirty 20 millisecond pulses to the sensor during the first time frame. Each of the thirty 20 millisecond pulses can have the same magnitude or can have different magnitudes. In this embodiment, the microcontroller 410 can instruct the DAC 420 to apply short duration pulses during the second time frame or can instruct the DAC 420 to apply another voltage waveform during the second time frame.

[0233] Although the disclosure in FIGS. 6-8 discloses the application of a voltage, a current can also be applied to the sensor to start the stabilization process. Illustratively, in FIG. 6BIn the illustrated embodiment, a first current can be applied during a first time frame to initiate an anodic or cathodic response, and a second current can be applied during a second time frame to initiate an opposite anodic or cathodic response. The application of the first and second currents can continue for several iterations or can continue for a stabilization time frame. In one embodiment, a first current can be applied during a first time frame, and a first voltage can be applied during a second time frame. In other words, one of the anodic or cathodic cycles can be triggered by applying a current to the sensor, and one of the anodic or cathodic cycles can be triggered by applying a voltage to the sensor. As described above, the applied current can be a constant current, a ramped current, a stepped pulsed current, or a sinusoidal current. Under certain operating conditions, the current can be applied as a series of short duration pulses during the first time frame.

[0234] FIG. 9B A sensor and sensor electronics utilizing an analysis module to perform feedback over a stabilization period according to an embodiment of the present application is shown. FIG. 9B An analysis module 950 is introduced to the sensor electronics 360. The analysis module 950 utilizes feedback from the sensor to determine whether the sensor is stable. In one embodiment, the microcontroller 410 can contain instructions or commands to control the DAC 420 to cause the DAC 420 to apply a voltage or current to a portion of the sensor 355. FIG. 9B A voltage or current that can be applied between the reference electrode 370 and the working electrode 375 is shown. However, the voltage or current can be applied between the electrodes or directly to one of the electrodes, and the present application should not be limited to the FIG. 9B embodiment shown. The application of the voltage or current is shown by the dashed line 955. The analysis module 950 can measure the voltage, current, resistance, or impedance in the sensor 355. FIG. 9BMeasurements are shown being taken at the working electrode 375, but this should not limit the application as other embodiments can measure voltage, current, resistance, or impedance between the electrodes of the sensor or directly at the reference electrode 370 or counter electrode 365. The analysis module 950 can receive the measured voltage, current, resistance, or impedance and can compare the measurement to a stored value (e.g., a threshold value). The dotted line 956 represents the analysis module 950 reading or taking a measurement of the voltage, current, resistance, or impedance. Under certain operating conditions, if the measured voltage, current, resistance, or impedance is above the threshold value, then the sensor is stabilized and the sensor signal provides an accurate reading of the patient's physiological condition. Under other operating conditions, if the measured voltage, current, resistance, or impedance is below the threshold value, then the sensor is stabilized. Under other operating conditions, the analysis module 950 can verify that the measured voltage, current, resistance, or impedance is stable over a certain time frame, e.g., one or two minutes. This can indicate that the sensor 355 is stabilized and that the sensor signal is transmitting an accurate measurement of the subject's physiological parameter, e.g., blood glucose level. After the analysis module 950 has determined that the sensor is stabilized and that the sensor signal is providing accurate measurements, the analysis module 950 can transmit a signal (e.g., a sensor stabilized signal) to the microcontroller 410 indicating that the sensor is stabilized and the microcontroller 410 can begin using or receiving the sensor signal from the sensor 355. This is represented by the dotted line 957.

[0235] FIG. 10 A block diagram of a sensor system including a hydration sensor is shown. The sensor system includes a connector 1010, a sensor 1012, and a monitor or sensor electronics 1025. The sensor 1012 includes electrodes 1020 and a connection portion 1024. In one embodiment, the sensor 1012 can be connected to the sensor electronics 1025 through the connector 1010 and a cable. In other embodiments, the sensor 1012 can be directly connected to the sensor electronics 1025. In some embodiments, the sensor 1012 can be incorporated into the same physical device as the sensor electronics 1025. The monitor or sensor electronics 1025 can include a power source 1030, a regulator 1035, a signal processor 1040, a measurement processor 1045, and a processor 1050. The monitor or sensor electronics 1025 can also include a hydration detection circuit 1060. The hydration detection circuit 1060 interfaces with the sensor 1012 to determine whether the electrodes 1020 of the sensor 1012 are sufficiently hydrated. If the electrodes 1020 are not sufficiently hydrated, the electrodes 1020 do not provide accurate glucose readings, so it is important to know when the electrodes 1020 are sufficiently hydrated. Once the electrodes 1020 are sufficiently hydrated, accurate glucose readings can be obtained.

[0236] InFIG. 10 In the illustrated embodiment, the hydration detection circuit 1060 can include a delay or timer module 1065 and a connection detection module 1070. In embodiments utilizing short-term or subcutaneous sensors, after the sensor 1012 is inserted into subcutaneous tissue, the sensor electronics or monitor 1025 is connected to the sensor 1012. The connection detection module 1070 recognizes that the sensor electronics 1025 has been connected to the sensor 1012 and sends a signal to the timer module 1065. This is shown in FIG. 10 by arrow 1084, which represents the detector 1083 detecting the connection and sending a signal to the connection detection module 1070 indicating that the sensor 1012 has been connected to the sensor electronics 1025. In embodiments utilizing implantable or long-term sensors, the connection detection module 1070 recognizes that the implantable sensor has been inserted into the body. The timer module 1065 receives the connection signal and waits for a set or established hydration time. Illustratively, the hydration time can be two minutes, five minutes, ten minutes, or 20 minutes. These examples are illustrative and not limiting. The time frame need not be a set number of minutes and can include any number of seconds. In one embodiment, after the timer module 1065 has waited for the set hydration time, the timer module 1065 can notify the processor 1050 that the sensor 1012 is hydrated by sending a hydration signal, as shown by line 1086. FIG. 10

[0237] In this embodiment, the processor 1050 can receive the hydration signal and only begin utilizing the sensor signal (e.g., sensor measurements) after the hydration signal has been received. In another embodiment, the hydration detection circuit 1060 can be coupled between the sensor (sensor electrodes 1020) and the signal processor 1040. In this embodiment, the hydration detection circuit 1060 can prevent the sensor signal from being sent to the signal processor 1040 until the timer module 1065 has notified the hydration detection circuit 1060 that the set hydration time has elapsed. This is shown by the dashed lines labeled 1080 and 1081. Illustratively, the timer module 1065 can transmit the connection signal to a switch (or transistor) to turn the switch on and advance the sensor signal to the signal processor 1040. In alternative embodiments, the timer module 1065 can transmit the connection signal to turn on (or close) a switch 1088 in the hydration detection circuit 1060 to allow the voltage from the regulator 1035 to be applied to the sensor 1012 after the hydration time has elapsed. In other words, in this embodiment, the voltage from the regulator 1035 is not applied to the sensor 1012 until after the hydration time has elapsed.

[0238] FIG. 11 ​Embodiments are shown that include a mechanical switch to assist in determining hydration time. In one embodiment, a single housing can contain the sensor assembly 1120 and the sensor electronics 1125. In another embodiment, the sensor assembly 1120 can be in one housing and the sensor electronics 1125 can be in a separate housing, but the sensor assembly 1120 and the sensor electronics 1125 can be connected together. In this embodiment, the connection detection mechanism 1160 can be a mechanical switch. The mechanical switch can detect that the sensor 1120 is physically connected to the sensor electronics 1125. When the mechanical switch 1160 detects that the sensor 1120 is connected to the sensor electronics 1125, the timer circuit 1135 can also be activated. In other words, the mechanical switch can close and a signal can be transmitted to the timer circuit 1135. Once the hydration time has elapsed, the timer circuit 1135 transmits a signal to the switch 1140 to allow the regulator 1035 to apply a voltage to the sensor 1120. In other words, the voltage is not applied until the hydration time has elapsed. In one embodiment, once the hydration time has elapsed, a current can be applied to the sensor instead of a voltage. In an alternative embodiment, power can first be applied to the sensor 1120 when the mechanical switch 1160 recognizes that the sensor 1120 has been physically connected to the sensor electronics 1125. Sending power to the sensor 1120 causes a sensor signal to be output from the working electrode in the sensor 1120. The sensor signal can be measured and sent to the processor 1175. The processor 1175 can contain a counter input. After a set hydration time has elapsed from the time the sensor signal is input into the processor 1175, the processor 1175 can begin processing the sensor signal as an accurate measurement of glucose in the body of the subject under certain operating conditions. In other words, the processor 1170 has been receiving a sensor signal from the potentiostat circuit 1170 for a certain amount of time, but does not process the signal until receiving an instruction from the processor's counter input that recognizes that the hydration time has elapsed. In an embodiment, the potentiostat circuit 1170 can contain a current-to-frequency converter 1180. In this embodiment, the current-to-frequency converter 1180 can receive the sensor signal as a current value and can convert the current value to a frequency value, which is easier for the processor 1175 to process.

[0239] The mechanical switch 1160 can also inform the processor 1175 when the sensor 1120 has been disconnected from the sensor electronics 1125. This is shown by the dashed line from the mechanical switch 1160 to the processor 1175. In one embodiment, the mechanical switch 1160 can be a normally closed switch. In this embodiment, the mechanical switch 1160 can be closed when the sensor 1120 is not connected to the sensor electronics 1125. When the sensor 1120 is connected to the sensor electronics 1125, the mechanical switch 1160 can open. In another embodiment, the mechanical switch 1160 can be a normally open switch. In this embodiment, the mechanical switch 1160 can be open when the sensor 1120 is not connected to the sensor electronics 1125. When the sensor 1120 is connected to the sensor electronics 1125, the mechanical switch 1160 can close. FIG. 11The dashed line 1176 indicates this. This may cause the processor 1170 to de-energize or reduce the power of several components, chips, and / or circuits of the sensor electronics 1125. If the sensor 1120 is not connected, the battery or power source may be depleted while the components or circuits of the sensor electronics 1125 are powered on. Therefore, if the mechanical switch 1160 detects that the sensor 1120 has been disconnected from the sensor electronics 1125, the mechanical switch can indicate this to the processor 1175, and the processor 1175 can de-energize or reduce the power of one or more of the electronic circuits, chips, or components of the sensor electronics 1125.

[0240] FIG. 12 An electrical method for detecting hydration according to an embodiment of the present invention is illustrated. In one embodiment, an electrical detection mechanism for a connection to a detection sensor can be utilized. In this embodiment, hydration detection electronics 1250 may include an AC source 1255 and a detection circuit 1260. Hydration detection electronics 1250 may be located within sensor electronics 1225. Sensor 1220 may include a counter electrode 1221, a reference electrode 1222, and a working electrode 1223. FIG. 12 As shown, the AC source 1255 is connected to the voltage setting device 1275, the reference electrode 1222, and the detection circuit 1260. In this embodiment, an AC signal from the AC source is applied to the reference electrode connection, such as... FIG. 12 As shown by dashed line 1291 in the diagram. An AC signal can be impedance-connected to sensor 1220, and if sensor 1220 is connected to sensor electronics 1225, the connected signal will be significantly attenuated. Therefore, a low-level AC signal exists at the input of detection circuit 1260. This can also be referred to as a highly attenuated signal or a signal with a high attenuation level. Under certain operating conditions, the voltage level of the AC signal can be Vapplied*(Ccoupling) / (Ccoupling+Csensor). If detection circuit 1260 detects a high-level AC signal (low-attenuation signal) at its input terminal, no interrupt is sent to microcontroller 410 because sensor 1220 is not sufficiently hydrated or activated. For example, the input of detection circuit 1260 could be a comparator. If sensor 1220 is sufficiently hydrated (or wetted), an effective capacitance is formed between the counter electrode and the reference electrode (e.g., ...). FIG. 12 The capacitor C in r-c ), and an effective capacitance is formed between the reference electrode and the working electrode (e.g., FIG. 12 The capacitor C in w-rIn other words, the effective capacitance is related to the capacitance formed between the two nodes and does not represent the placement of an actual capacitor in the circuit between the two electrodes. In one embodiment, the AC signal from the AC source 1255 is capacitively coupled to the sensor 1120 through the capacitance C r-c and C w-r is sufficiently attenuated, and the detection circuit 1260 detects the presence of a low level or highly attenuated AC signal from the AC source 1255 at the input terminal of the detection circuit 1260. This embodiment is important because it takes advantage of the existing connections between the sensor 1120 and the sensor electronics 1125 to reduce the number of connections to the sensor. In other words, FIG. 11 The mechanical switch disclosed in the '932 patent requires a switch and associated connections between the sensor 1120 and the sensor electronics 1125. It is advantageous to remove the mechanical switch because the size of the sensor 1120 is continually shrinking, and removing components helps to achieve this size reduction. In alternative embodiments, the AC signal can be applied to a different electrode (e.g., the counter electrode or the working electrode), and the present invention can operate in a similar manner.

[0241] As described above, after the detection circuit 1260 has detected the presence of a low level AC signal at the input terminal of the detection circuit 1260, the detection circuit 1260 can later detect the presence of a high level AC signal with low attenuation at the input terminal. This indicates that the sensor 1220 has been disconnected from the sensor electronics 1225 or that the sensor is not operating properly. If the sensor has been disconnected from the sensor electronics 1225, then the AC source can be small or low attenuation coupled to the input of the detection circuit 1260. As described above, the detection circuit 1260 can generate an interrupt to the microcontroller. This interrupt can be received by the microcontroller, and the microcontroller can reduce or remove power to one or several components or circuits in the sensor electronics 1225. This can be referred to as a second interrupt. Again, this helps to reduce the power consumption of the sensor electronics 1225, particularly when the sensor 1220 is not connected to the sensor electronics 1225.

[0242] In alternative embodiments, the AC signal can be applied to the reference electrode 1222, as indicated by reference numeral 1291, and the impedance measurement device 1277 can measure the impedance of a region in the sensor 1220. Illustratively, the region can be the region between the reference electrode and the working electrode, as indicated by reference numeral 1292. In this embodiment, the sensor 1220 is not connected to the sensor electronics 1225, and the sensor electronics 1225 can be powered down or have reduced power consumption. This is advantageous because it reduces the power consumption of the sensor electronics 1225 when the sensor 1220 is not connected to the sensor electronics 1225. FIG. 12under certain operating conditions, the impedance measurement device 1277 can transmit a signal to the detection circuit 1260 if the measured impedance has decreased below an impedance threshold or other set criteria. This indicates that the sensor is sufficiently hydrated. Under other operating conditions, the impedance measurement device 1277 can transmit a signal to the detection circuit 1260 once the impedance is above an impedance threshold. The detection circuit 1260 then transmits an interrupt to the microcontroller 410. In another embodiment, the impedance measurement device 1277 can transmit the interrupt or signal directly to the microcontroller.

[0243] In an alternative embodiment, the AC source 1255 can be replaced by a DC source. If a DC source is utilized, a resistance measurement element can be utilized in place of the impedance measurement element 1277. In an embodiment utilizing a resistance measurement element, once the resistance drops below a resistance threshold or set criteria, the resistance measurement element can transmit a signal to the detection circuit 1260 (represented by dashed line 1293) or directly to the microcontroller indicating that the sensor is sufficiently hydrated and power can be applied to the sensor.

[0244] In FIG. 12 In the embodiment shown, an interrupt is generated to the microcontroller 410 if the detection circuit 1260 detects a low level or highly attenuated AC signal from the AC source. This interrupt indicates that the sensor is sufficiently hydrated. In this embodiment, in response to the interrupt, the microcontroller 410 generates a signal that is communicated to the digital to analog converter 420 to instruct or cause the digital to analog converter 420 to apply a voltage or current to the sensor 1220. This is described above in FIG. 6A 、 FIG. 6B or FIG. 6CAny of the different sequences of pulses or pulses of short duration described in the associated text describing the pulse application can be applied to the sensor 1220. Illustratively, the voltage from the DAC 420 can be applied to an operational amplifier 1275, the output of which is applied to the counter electrode 1221 of the sensor 1220. This causes the sensor, e.g., the working electrode 1223 of the sensor, to produce a sensor signal. Because the sensor is sufficiently hydrated, as identified by the interrupt, the sensor signal produced at the working electrode 1223 is accurately measuring glucose. The sensor signal is measured by the sensor signal measurement device 431 and the sensor signal measurement device 431 transmits the sensor signal to the microcontroller 410 where the parameter of the subject's physiological condition is measured. The production of the interrupt indicates that the sensor is sufficiently hydrated and that the sensor 1220 is now supplying accurate glucose measurement values. In this embodiment, the hydration period can depend on the type and / or manufacturer of the sensor, and on the sensor's reaction to being inserted or implanted into the subject. Illustratively, one sensor 1220 can have a five minute hydration time and one sensor 1220 can have a one minute, two minute, three minute, six minute, or 20 minute hydration time. Again, any amount of time can be an acceptable hydration time amount for a sensor, but less amounts of time are preferred.

[0245] If the sensor 1220 has been connected but is not sufficiently hydrated or wetted, the effective capacitance C r-c and C w-r may not attenuate the AC signal from the AC source 1255. The electrodes in the sensor 1120 are dry before insertion and because the electrodes are dry, there is no good electrical path (or conductive path) between the two electrodes. Thus, the detection circuit 1260 can still detect a high level AC signal or a low attenuated AC signal and can not produce an interrupt. Once the sensor is inserted, the electrodes are immersed in the conductive body fluid. This creates a leaky path with a lower DC resistance. Also, a boundary layer capacitor is formed at the metal / fluid interface. In other words, a substantial capacitance is formed between the metal / fluid interface and this large capacitance looks like two capacitors in series between the electrodes of the sensor. This can be referred to as the effective capacitance. In effect, the conductivity of the electrolyte above the electrodes is being measured. In some embodiments of the invention, a glucose limiting membrane (GLM) also shows an impedance that hinders electrical efficiency. An unhydrated GLM produces a high impedance while a highly hydrated GLM produces a low impedance. Accurate sensor measurements require a low impedance.

[0246] FIG. 13AA method of hydrating a sensor according to embodiments of the application is shown. In one embodiment, the sensor can be physically connected 1310 to the sensor electronics. After connection, in one embodiment, a timer or counter can be started to count 1320 the hydration time. After the hydration time has elapsed, a signal can be transmitted 1330 to a subsystem in the sensor electronics to start applying a voltage to the sensor. As discussed above, in one embodiment, a microcontroller can receive the signal and instruct a DAC to apply a voltage to the sensor, or in another embodiment of the application, a switch can receive the signal which allows a regulator to apply a voltage to the sensor. The hydration time can be five minutes, two minutes, ten minutes, and can vary depending on the subject as well as the type of sensor.

[0247] In an alternative embodiment, after the sensor is connected to the sensor electronics, an AC signal (e.g., a low voltage AC signal) can be applied 1340 to the sensor, such as the reference electrode of the sensor. The AC signal can be applied because the connection of the sensor to the sensor electronics allows the AC signal to be applied to the sensor. After the AC signal is applied, an effective capacitance is formed 1350 in the sensor between the electrode to which the voltage is applied and the other two electrodes. The detection circuit determines 1360 what level of AC signal is present at the input of the detection circuit. If a low level AC signal (or a highly attenuated AC signal) is present at the input of the detection circuit, then because the effective capacitance forms a good electrical conduit between the electrodes and the AC signal is attenuated, the detection circuit generates 1370 an interrupt and the interrupt is sent to the microcontroller.

[0248] The microcontroller receives the interrupt generated by the detection circuit and transmits 1380 a signal to the digital to analog converter instructing or causing the digital to analog converter to apply a voltage to an electrode of the sensor, such as to the counter electrode. The application of the voltage to the electrode of the sensor causes the sensor to generate or produce a sensor signal 1390. The sensor signal measurement device 431 measures the produced sensor signal and transmits the sensor signal to the microcontroller. The microcontroller receives 1395 the sensor signal from the sensor signal measurement device coupled to the working electrode and processes the sensor signal to extract a measurement of a physiological characteristic of the subject or patient.

[0249] FIG. 13B An additional method for verifying hydration of a sensor according to embodiments of the application is shown. In FIG. 13BIn the illustrated embodiment, the sensor is physically connected 1310 to the sensor electronics. An AC signal is applied 1341 to an electrode in the sensor, e.g., the reference electrode. Alternatively, in another embodiment, a DC signal is applied 1341 to an electrode in the sensor. If an AC signal is applied, then the impedance measuring element measures 1351 the impedance at a point within the sensor. Alternatively, if a DC signal is applied, then the resistance measuring element measures 1351 the resistance at a point within the sensor. If the resistance or impedance, respectively, is below a resistance threshold or impedance threshold (or other set criteria), then the impedance (or resistance) measuring element transmits 1361 (or allows transmission of) a signal to the detection circuit, and the detection circuit will identify that the sensor has hydrated and transmit an interrupt to the microcontroller. Reference numerals 1380, 1390, and 1395 are the same in FIG. 13A and FIG. 13B as they represent the same action.

[0250] The microcontroller receives the interrupt and transmits 1380 a signal to the digital to analog converter to apply a voltage to the sensor. In an alternative embodiment, the digital to analog converter can apply a current to the sensor, as discussed above. The sensor, e.g., the working electrode, generates 1390 a sensor signal, which represents a physiological parameter of the patient. The microcontroller receives 1395 the sensor signal from a sensor signal measuring device, which measures the sensor signal on an electrode in the sensor, e.g., the working electrode. The microcontroller processes the sensor signal to extract a measurement of a physiological characteristic of the subject or patient, e.g., the patient’s blood glucose level.

[0251] FIG. 14A and FIG. 14B A method of combining hydration of a sensor with stabilization of the sensor according to an embodiment of the application is shown. In FIG. 14AIn the illustrated embodiment of the application, the sensor connection 1405 is connected to the sensor electronics. An AC signal is applied 1410 to the electrodes of the sensor. The detection circuit determines 1420 what level of AC signal is present at the input of the detection circuit. If the detection circuit determines that a low level of AC signal is present at the input (indicating a high level of attenuation of the AC signal), then an interrupt is sent 1430 to the microcontroller. Once the interrupt is sent to the microcontroller, the microcontroller knows to start or initiate 1440 a stabilization sequence, i.e., to apply a number of voltage pulses to the electrodes of the sensor, as described above. For example, the microcontroller can cause a digital-to-analog converter to apply three voltage pulses to the sensor (of magnitude +0.535 volts), with each of the three voltage pulses followed by a period of three voltage pulses of magnitude 1.07 volts to be applied. This can be referred to as a stabilization sequence of transmission voltages. The microcontroller can cause this by executing a software program in read-only memory (ROM) or random access memory. After the stabilization sequence is completed, the sensor can generate 1450 a sensor signal, which is measured and transmitted to the microcontroller.

[0252] The detection circuit can determine 1432 that a high level of AC signal continues to be present at the input of the detection circuit (e.g., the input of the comparator), even after a hydration time threshold has passed. For example, the hydration time threshold can be 10 minutes. After 10 minutes has passed, the detection circuit can still detect that a high level of AC signal is present. At this point in time, the detection circuit can transmit 1434 a hydration assist signal to the microcontroller. If the microcontroller receives the hydration assist signal, then the microcontroller can transmit 1436 a signal to cause the DAC to apply a voltage pulse or series of voltage pulses to assist the sensor in hydrating. In one embodiment, the microcontroller can transmit a signal to cause the DAC to apply a portion of the stabilization sequence or other voltage pulses to assist the sensor in hydrating. In this embodiment, the application of the voltage pulses can cause a low level of AC signal (or highly attenuated signal) to be detected 1438 at the detection circuit. At this point, the detection circuit can transmit an interrupt, as disclosed in step 1430, and the microcontroller can start the stabilization sequence.

[0253] FIG. 14BA second embodiment showing a combination of hydration and stabilization methods, where feedback is utilized in the stabilization process. A sensor is connected 1405 to sensor electronics. An AC signal (or DC signal) is applied 1411 to the sensor. In embodiments, the AC signal (or DC signal) is applied to an electrode of the sensor, such as a reference electrode. An impedance measurement device (or resistance measurement device) measures 1416 the impedance (or resistance) within a designated area of the sensor, such as between the reference electrode and a working electrode. The measured impedance (or resistance) can be compared 1421 to an impedance or resistance value to see if the impedance (or resistance) in the sensor is low enough, which indicates that the sensor is hydrated. If the impedance (or resistance) is below the impedance (or resistance) value or other set criteria (which can be a threshold), then an interrupt is transmitted 1431 to the microcontroller. Upon receiving the interrupt, the microcontroller transmits 1440 a signal to the DAC instructing the DAC to apply a stabilization sequence of voltages (or currents) to the sensor. After the stabilization sequence is applied to the sensor, a sensor signal is generated in the sensor (e.g., at the working electrode), measured by the sensor signal measurement device, transmitted by the sensor signal measurement device, and received 1450 by the microcontroller. Because the sensor is hydrated, and a stabilization sequence of voltages has been applied to the sensor, the sensor signal is accurately measuring the physiological parameter (i.e., blood glucose).

[0254] FIG. 14CA third embodiment is shown in which the stabilization method and hydration method are combined. In this embodiment, a sensor is connected 1500 to sensor electronics. After the sensor is physically connected to the sensor electronics, an AC signal (or DC signal) is applied 1510 to an electrode of the sensor (e.g., the reference electrode). At or about the same time, the microcontroller transmits a signal to cause the DAC to apply 1520 a stabilization voltage sequence to the sensor. In alternative embodiments, a stabilization current sequence can be applied to the sensor instead of a stabilization voltage sequence. The detection circuit determines 1530 what level of AC signal (or DC signal) is present at the input terminal of the detection circuit. If a low level AC signal (or DC signal) representing a highly attenuated AC signal (or DC signal) is present at the input terminal of the detection circuit, then an interrupt is transmitted 1540 to the microcontroller. Because the microcontroller has already started the stabilization sequence, the microcontroller receives the interrupt and sets 1550 a first indicator that the sensor is sufficiently hydrated. After the stabilization sequence is complete, the microcontroller sets 1555 a second indicator that the stabilization sequence is complete. The application of the stabilization sequence voltage causes the sensor, e.g., the working electrode, to produce 1560 a sensor signal that is measured by the sensor signal measurement circuit and sent to the microcontroller. If the second indicator that the stabilization sequence is complete is set and the first indicator that the hydration is complete is set, then the microcontroller can utilize 1570 the sensor signal. If one or both of the indicators are not set, then the microcontroller can not be able to utilize the sensor signal because the sensor signal can not represent an accurate measurement of the subject's physiological measurement.

[0255] In general, the above-described hydration and stabilization processes can be used as part of a larger continuous glucose monitoring (CGM) method. The current state of the art of continuous glucose monitoring is largely ancillary, meaning that a reference value is required to make a clinical decision using the readings provided by a CGM device (including, e.g., an implantable or subcutaneous sensor). The reference value, in turn, must typically be obtained from a finger stick using, e.g., a BG meter. The reason for the need for a reference value is that the amount of information available from the sensor / sensing assembly is limited. Specifically, the sensing assembly can only provide raw sensor values (i.e., sensor current or Isig) and a counter voltage (e.g., see FIG. 5 ) as the voltage between the counter electrode and the reference electrode for processing. Thus, during analysis, if the raw sensor signal appears abnormal (e.g., if the signal is decreasing), the only way to distinguish between a sensor malfunction and a physiological change (i.e., a change in glucose level) in the user / patient can be to take a reference glucose value via a finger stick. It is well known that reference finger sticks are also used to calibrate the sensor.

[0256] Embodiments of the invention described herein are directed to advances and improvements in continuous glucose monitoring, resulting in more autonomous systems and related devices and methods, in which the need for reference finger pricking can be minimized or eliminated, and from which clinical decisions can be made with great reliability based on information derived from sensor signals alone. From a sensor design perspective, autonomy can be achieved according to embodiments of the invention through electrode redundancy, sensor redundancy (including complex redundancy between, for example, two or more sensors), sensor diagnostics, and Isig and / or sensor glucose (SG) fusion.

[0257] In the discussion herein, and for the purposes of the invention, "redundancy" refers to the presence / use of two or more electrodes, whether contained on / within a single probe (or "flex"), or contained on / within two or more flexes, and "complex redundancy" refers to the presence / use of two (or more) sensors, in which at least two of the sensors are not identical. Thus, "redundant" electrodes can be contained on / within a single flex, two or more identical flexes, or two or more non-identical flexes. As will be explored further below, redundancy can be achieved, for example, by using multiple working electrodes to produce multiple signals indicative of a patient's blood glucose (BG) level. The multiple signals, in turn, can be used to produce a fused glucose value, as well as to assess the relative health of the (working) electrodes, the overall reliability of the sensor, and the frequency with which a calibration reference value is needed, if at all.

[0258] For example, it is known that acquiring signals from multiple electrochemical sensors can provide improved performance in the form of simple redundancy, either through multiple electrodes on the same probe (or flex), or by utilizing spatial separation and two separate probes. For example, Medtronic sells a hospital glucose sensor that contains two probes, with two working electrodes on each probe, resulting in four independent glucose signals.

[0259] In contrast to simple redundancy, orthogonal redundancy can be defined as two devices that employ two different technologies to achieve the same goal, in which the failure modes of the two devices are completely unique and disjoint. Thus, orthogonal redundancy can be produced by combining technologies such as optical sensing and electrochemical sensing. Clearly, the advantage of orthogonal redundancy is that two types of sensors, for example optical and electrochemical ("echem") sensors, are subject to different types of interference, failure modes, and body responses. On the other hand, the use of two completely different technologies introduces an additional layer of design and computational complexity to the measurement and analysis of glucose levels in the patient's body.

[0260] On the other hand, pseudo-orthogonal redundancy can be implemented by exploiting the same technology, but with variations, in order to produce complementary glucose measurements while minimizing additional design and / or computational complexity. For example, two or more electrochemical sensors can be employed, where one (or more) sensor can be a traditional peroxide-based sensor, and one (or more) sensor can measure glucose by calculating the oxygen difference between two working electrodes (typically on the same sensor).

[0261] In yet another particular type of redundancy, as will be explored in greater detail below, in embodiments of the present application, a sensor system employing complex redundancy can include two (or more) sensors, where (at least) two sensors differ from one another in design (and can also employ different chemistries and / or sizes). Here, one (or more) of the sensors can be designed to have, for example, significantly better hydration and / or stabilization characteristics, but can not last 2 or 3 days. On the other hand, the other sensor can have long-lasting durability, but slower initial hydration and / or stabilization. In this case, an algorithm can be designed in which the first sensor is used to generate glucose data during the early wear period, after which the first sensor can be used to calibrate the second sensor, and then the second sensor can be switched to for generating glucose data during the remaining life of the glucose sensor system.

[0262] Sensor diagnostics include the use of additional (diagnostic) information, which can provide real-time insight into the health of the sensor. In this regard, it has been discovered that electrochemical impedance spectroscopy (EIS) provides such additional information in the form of sensor impedance and impedance-related parameters at different frequencies. Moreover, advantageously, it has been further discovered that for certain frequency ranges, the impedance and / or impedance-related data is substantially glucose-independent. This glucose-independence not only enables a variety of EIS-based markers or indicators to produce robust, highly reliable sensor glucose values (through a fusion approach, described in greater detail below), but also enables the assessment of individual electrode and overall sensor condition, health, longevity, and efficiency substantially independent of glucose-related Isig.

[0263] For example, analysis of the glucose-independent impedance data uses values such as 1 kHz real impedance, 1 kHz imaginary impedance, and Nyquist slope to provide information about the efficiency of the sensor in terms of its hydration and readiness for data acquisition (to be described in more detail below). In addition, the glucose-independent impedance data provides information about potential occlusions that can be present on the surface of the sensor membrane that can temporarily prevent glucose from entering the sensor and thus cause a signal drop (using values such as 1 kHz real impedance). Further, the glucose-independent impedance data uses values such as phase angle and / or imaginary impedance at 1 kHz and higher frequencies to provide information about loss of sensor sensitivity during long wear - possibly due to local hypoxia at the insertion site.

[0264] In the context of (electrode) redundancy and EIS, a fusion algorithm can be used to take the diagnostic information provided by EIS for each redundant electrode and independently assess the reliability of each electrode. A weight as a measure of reliability can then be added to each independent signal, and a single fused signal can be calculated that can be used to produce the sensor glucose value seen by the patient / subject. As can be seen from the foregoing, the combined use of redundancy, sensor diagnostics using EIS, and an EIS-based fusion algorithm results in a more reliable overall CGM system. In addition, the EIS diagnostics can automatically check the health of each electrode without reference to glucose values (finger prick), thereby reducing the number of reference values needed.

[0265] EIS or AC impedance methods study the response of a system to the application of a periodic, small amplitude AC signal. This is shown schematically in FIG. 15A where E is the applied potential, I is the current, and impedance (Z) is defined as ΔE / ΔI. However, while impedance itself can be mathematically simply defined as ΔE / ΔI, to date there has not been commercial success in applying EIS techniques to continuous glucose monitoring. This is partly because glucose sensors are very complex systems, and to date no mathematical model has been developed that can fully account for the complexity of the EIS output of a glucose sensor.

[0266] FIG. 15B One simplified circuit model that has been used to describe electrochemical impedance spectroscopy is shown in d is the double layer capacitance, R p is the polarization resistance, Z w is the Warburg impedance, and R s is the solution resistance. The latter four components - the double layer capacitance (C d ), the Warburg impedance (Z w), and solution resistance (R p ) and solution resistance (R s ) can each play an important role in sensor performance and can be measured by applying low or high frequency alternating working potentials, respectively. For example, the Warburg impedance is closely related to the diffusion impedance of an electrochemical system, primarily the low frequency impedance, and thus exists in all diffusion-limited electrochemical sensors. Thus, by associating one or more of these components with one or more components and / or layers of a glucose sensor, EIS techniques can be used as a sensor diagnostic tool.

[0267] It is well known that impedance can be defined in terms of its magnitude and phase, where the magnitude (|Z|) is the ratio of the voltage difference amplitude to the current amplitude, and the phase (θ) is the phase shift of the current ahead of the voltage. When a circuit is driven with direct current (DC) only, impedance is the same as resistance, i.e., resistance is a special case of impedance with zero phase angle. However, as a complex number, impedance can also be represented by its real and imaginary parts. In this regard, the real and imaginary impedances can be derived from the impedance magnitude and phase using the following equations:

[0268] Real impedance (ω) = magnitude (ω) x cos(phase (ω) / 180 x π)

[0269] Imaginary impedance (ω) = magnitude (ω) x sin(phase (ω) / 180 x π)

[0270] where ω represents the input frequency at which the magnitude (in ohms) and phase (in degrees) are measured. The relationship between the impedance of one aspect and the current and voltage of another aspect, containing how impedance can be calculated from measured values of current and voltage, will be explored more fully below in connection with sensor electronics containing an application specific integrated circuit (ASIC) that has been developed for use in embodiments of the application described herein.

[0271] Continuing FIG. 15B with the circuit model shown, the total system impedance can be simplified to:

[0272]

[0273] where Z w (ω) is the Warburg impedance, ω is the angular velocity, j is the imaginary unit (instead of the traditional "i" to avoid confusion with current), and C d , R p , and R s are the double layer capacitance, polarization resistance, and solution resistance, respectively (as previously defined). The Warburg impedance can be calculated as

[0274]

[0275]

[0276] where D is the diffusion rate, L is the sensor film thickness, C is the peroxide concentration, and m: 1 / 2 corresponds to the 45° Nyquist slope.

[0277] A Nyquist plot is a graphical representation in which the real part of impedance (Real Z) is plotted against the imaginary part of impedance (Img Z) over the entire frequency spectrum. FIG. 16A A general example of a Nyquist plot is shown, where the X value is the real part of impedance, and the Y value is the imaginary part of impedance. The phase angle is the angle between the impedance point (X, Y) - which defines a vector of magnitude |Z| - and the X axis.

[0278] FIG. 16A The Nyquist plot of the is generated by applying an AC voltage plus a DC voltage (DC bias) between the working electrode and the counter electrode at a selected frequency (i.e., sweeping frequency) from 0.1 Hz to 1000 MHz. Starting from the right, the frequency starts at 0.1 Hz and increases. For each frequency, the real and imaginary impedances can be calculated and plotted. As shown, a typical Nyquist plot of an electrochemical system can look like a semicircle connected to a straight line at the knee, where the semicircle and the straight line represent the plotted impedance. In certain embodiments, the impedance at the knee is of particular interest because it is the most easily identified in the Nyquist plot and can define the intercept. Typically, the knee is near the X axis, and the X value of the knee approximates the sum of the polarization resistance and the solution resistance (R p + R s ).

[0279] Referring to FIG. 16B , the Nyquist plot can generally be described in terms of a low frequency region 1610 and a high frequency region 1620, where the labels “high frequency” and “low frequency” are used in a relative sense and are not meant to be limiting. Thus, for example, the low frequency region 1610 can exemplarily contain data points obtained for a frequency range between about 0.1 Hz and about 100 Hz (or higher), and the high frequency region 1620 can exemplarily contain data points obtained for a frequency range between about 1 kHz (or lower) and about 8 kHz (or higher). In the low frequency region 1610, the Nyquist slope represents the gradient of a linear fit 1630 of the low frequency data points in the Nyquist plot. As shown, in the high frequency region 1620, the value of the imaginary impedance is minimal and can become negligible. Thus, the intercept 1600 is essentially the value of the real impedance at higher frequencies (e.g., in this case, approximately in the 1 kHz to 8 kHz range). In FIG. 16B , the intercept 1600 is at about 25 kilo-ohms.

[0280] FIG. 16C andFIG. 16D illustrates how a glucose sensor responds to a sinusoidal (i.e., alternating) working potential. In these figures, GLM is the glucose limiting membrane of the sensor, AP is the adhesion promoter, HSA is human serum albumin, GOX is glucose oxidase (layer), E dc is the DC potential, E ac is the AC potential, and C' 过氧化物 is the peroxide concentration during AC application. As FIG. 16C shown, if the sensor diffusion length is smaller than the membrane (GOX) length as a function of AC potential frequency, molecular diffusivity, and membrane thickness, then the system gives a linear response with respect to constant phase angle (i.e., infinite). Conversely, if the diffusion length is equal to the membrane (GOX) length, then the system response will become finite, resulting in a semi-circular Nyquist plot, as FIG. 16D shown. The latter is typically applicable to low frequency EIS where non-faradaic processes are negligible.

[0281] In performing EIS analysis, an AC voltage at various frequencies and a DC bias can be applied, for example, between the working electrode and the reference electrode. In this regard, EIS is an improvement over previous methods that can limit the application to simple DC current or single frequency AC voltage. Although EIS can generally be performed at frequencies ranging from μHz to MHz, in embodiments of the present invention described herein, a narrower frequency range, for example, between about 0.1 Hz to about 8 kHz, can be sufficient. Thus, in some embodiments, an AC potential can be applied in a frequency range between about 0.1 Hz and about 8 kHz, with a programmable amplitude of up to at least 100 mV, and preferably about 50 mV.

[0282] Within the above frequency range, the capacitive properties of the sensor are examined using relatively higher frequencies, i.e., typically between about 1 kHz and about 8 kHz. Depending on the thickness and permeability of the membrane, the typical range of impedance at relatively higher frequencies can be, for example, between about 500 ohms and 25 kilo-ohms, and the typical range of phase can be, for example, between 0 degrees and -40 degrees. On the other hand, the resistive properties of the sensor are examined using relatively lower frequencies, i.e., typically between about 0.1 Hz and about 100 Hz. Here, depending on the electrode design and degree of metallization, the typical operating range for output real impedance can be, for example, between about 50 kilo-ohms and 300 kilo-ohms, and the typical range for phase can be between about -50 degrees and about -90 degrees. The above illustrative ranges are shown, for example, in the Bode plots of FIG. 16E and FIG. 16F .

[0283] As previously stated, the phrases “high frequency” and “low frequency” are used relative to each other, not in an absolute sense, and they, along with the typical impedance and phase ranges mentioned above, are intended to be illustrative, not restrictive. However, the fundamental principle remains unchanged: the capacitive and resistive characteristics of a sensor can be examined by analyzing impedance data across the entire spectrum, where, generally, lower frequencies provide information about more resistive components (e.g., electrodes), while higher frequencies provide information about capacitive components (e.g., membranes). However, the actual frequency range in each case depends on the overall design, including, for example, the type of electrode, the surface area of ​​the electrode, the membrane thickness, the membrane permeability, etc. For a general correspondence between high-frequency circuit components and sensor membranes, and between low-frequency circuit components and Faraday processes involving, for example, electrodes, see also [link to relevant documentation]. FIG. 15B .

[0284] EIS can be used in sensor systems where the sensor contains a single working electrode, as well as in sensor systems where the sensor contains multiple (redundant) working electrodes. In one embodiment, EIS provides valuable information about the sensor's lifetime (or aging). Specifically, it shows the magnitude of impedance and phase angle changes at different frequencies. FIG. 17 As seen above, the sensor impedance, especially the sum of Rp and Rs, reflects the sensor's lifespan and operating conditions. Therefore, as from... FIG. 17 As seen in the different figures, new sensors typically have higher impedance than used sensors. In this way, by considering the sum of Rp and Rs and the value of X, a threshold can be used to determine when the sensor's lifespan exceeds its specified operating life. It should be noted that, although for... FIG. 17 As shown in Figure 21 and the illustrative examples discussed below, the value of the real impedance at the inflection point (i.e., Rp+Rs) is used to determine the aging, condition, stability, and hydration of the sensor. However, alternative embodiments may use other EIS-based parameters besides or in place of the real impedance, such as virtual impedance, phase angle, Nyquist slope, etc.

[0285] FIG. 18An example of a Nyquist plot over the life of a sensor is shown. The arrowed points are the respective inflection points in the spectrum for each scan. For example, prior to initialization (at time t = 0), Rs+Rp is above 8.5 kilo-ohms, and after initialization (at time t = 0.5 hr), the value of Rs+Rp drops below 8 kilo-ohms. Over the next six days, Rs+Rp continues to decrease, and thus, at the end of the specified sensor lifetime, Rs+Rp drops below 6.5 kilo-ohms. Based on such examples, a threshold can be set to specify when the value of Rs+Rp indicates the end of the specified working life of the sensor. Thus, the EIS technique allows for a workaround to allow reuse of the sensor after a specified operating time. In other words, if a patient attempts to reuse the sensor after it has reached its specified operating time by disconnecting and then reconnecting the sensor, the EIS will measure an abnormally low impedance, thus allowing the system to reject the sensor and prompt the patient to replace a new sensor.

[0286] Additionally, the EIS can enable detection of sensor failure by detecting when the impedance of the sensor drops below a low impedance threshold level, thus indicating that the sensor can be excessively worn and unable to operate properly. The system can then terminate the sensor prior to the specified working life. As will be explored in greater detail below, the sensor impedance can also be used to detect other sensor failure (modes). For example, when the sensor enters a low current state (i.e., sensor failure) for various reasons, the sensor impedance can also increase beyond a certain high impedance threshold. The system can also terminate the sensor prior to the specified sensor working life if the impedance becomes abnormally high during sensor operation, for example, due to protein or polypeptide fouling, macrophage attachment, or any other factor.

[0287] FIG. 18 An example of how the EIS technique can be applied during sensor stabilization, as well as in detecting the life of a sensor, is shown in accordance with certain embodiments. FIG. 19 The logic of FIG. 18 begins at 1800 after the aforementioned hydration procedure and sensor initialization procedure have been completed. In other words, the sensor has been deemed sufficiently hydrated, and the first initialization procedure has been applied to initialize the sensor. The initialization procedure can preferably take the form of a voltage pulse, as previously described in the detailed description. However, in alternative embodiments, a different waveform can be used for the initialization procedure. For example, a sinusoidal wave can be used instead of a pulse to accelerate wetting or conditioning of the sensor. Additionally, it can be necessary to have a portion of the waveform greater than the normal operating voltage of the sensor, i.e., 0.535 volts.

[0288] At block 1810, an EIS procedure is applied and the impedance is compared to both a first high threshold and a first low threshold. Examples of the first high threshold and the first low threshold are 7 kilo-ohms and 8.5 kilo-ohms, respectively, but these can be set higher or lower as desired. If the impedance (e.g., Rp+Rs) is higher than the first high threshold, then the sensor undergoes an additional initialization procedure (e.g., application of one or more additional pulses) at block 1820. Ideally, the total number of initialization procedures applied to initialize the sensor will be optimized to limit the impact on the sensor battery life and the total amount of time required to stabilize the sensor. Thus, by applying EIS, a lesser initialization can be performed first, and the number of initialization can be added incrementally to provide only the correct number of initialization to prepare the sensor for use. Similarly, in alternative embodiments, EIS can be applied to the hydration procedure to minimize the number of initialization required to assist the hydration process, as shown in FIGS. 13-14.

[0289] On the other hand, if the impedance (e.g., Rp+Rs) is lower than the first low threshold, then the sensor will be determined to be faulty and will terminate immediately at block 1860. A message will be brought to the user to replace the sensor and begin the hydration process again. If the impedance is within the high and low thresholds, then the sensor will begin normal operation at block 1830. The logic then proceeds to block 1840, where an additional EIS is performed to check the life of the sensor. The first time the logic reaches block 1840, the microcontroller will perform an EIS to measure the life of the sensor, thereby closing a loophole where a user could insert and remove the same sensor. In future iterations of the EIS procedure, when the logic returns to block 1840, the microprocessor will perform an EIS at a fixed interval during the specified life of the sensor. In one preferred embodiment, the fixed interval is set to every 2 hours, however, longer or shorter time periods can be readily used.

[0290] At block 1850, the impedance is compared to a second set of high and low thresholds. Examples of such second high and low thresholds can be 5.5 kilo-ohms and 8.5 kilo-ohms, respectively, but the values can be set higher or lower as desired. So long as the impedance value remains within the second high and low thresholds, the logic proceeds to block 1830, where the sensor is operating normally until a specified sensor lifetime, e.g., 5 days, is reached. Of course, as described with respect to block 1840, the EIS will be performed at regularly scheduled intervals throughout the specified sensor lifetime. However, if after performing the EIS it is determined at block 1850 that the impedance has dropped below the second lower threshold or risen above the second higher threshold, then the sensor is terminated at block 1860. In other alternative embodiments, a second check can be implemented on the faulty sensor reading. For example, if the EIS indicates that the impedance is outside the range of the second high and low thresholds, then the logic can perform a second EIS to confirm that the second set of thresholds are indeed not met before determining that the sensor is finished at block 1860 (and confirming that the first EIS was performed correctly).

[0291] FIG. 19 The above description relied upon and detailed possible schedules for performing diagnostic EIS procedures. Each diagnostic EIS procedure is optional, and any diagnostic EIS procedure can be scheduled or any combination of one or more diagnostic EIS procedures can be scheduled as desired. FIG. 18 The schedule of FIG. 19 begins at the insertion of the sensor at point 1900. After the sensor is inserted, the sensor undergoes a hydration period 1910. This hydration period is important because a sensor that is not sufficiently hydrated can give inaccurate readings to the user, as previously described. During this hydration period 1910, a first optional diagnostic EIS procedure at point 1920 is scheduled to ensure that the sensor is sufficiently hydrated. The first diagnostic EIS procedure 1920 measures the sensor impedance value to determine whether the sensor has been sufficiently hydrated. If the first diagnostic EIS procedure 1920 determines that the impedance is within a set high and low threshold, which indicates sufficient hydration, then the sensor controller will allow the sensor to be started at point 1930. Conversely, if the first diagnostic EIS procedure 1920 determines that the impedance is outside the set high and low threshold, which indicates insufficient hydration, then the sensor hydration period 1910 can be extended. After the extended hydration, once a certain capacitance is reached between the electrodes of the sensor, which means that the sensor is sufficiently hydrated, the start at point 1930 can be made.

[0292] A second optional diagnostic EIS procedure 1940 is scheduled after sensor activation at point 1930 but before sensor initialization begins at point 1950. This second diagnostic EIS procedure 1940 scheduled at this time can detect if the sensor is being reused before initialization begins at 1950. The test to determine if the sensor is being reused is described in detail in the description of FIG. 18 However, unlike the previous description regarding FIG. 19 the aging test is shown in FIG. 19 as being performed before initialization. It is important to recognize that FIG. 20 the timeline of the EIS procedure described in

[0293] A third optional diagnostic EIS procedure 1960 is scheduled after initialization begins at point 1950. The third diagnostic EIS procedure 1960 tests the impedance value of the sensor to determine if the sensor is fully initialized. The third diagnostic EIS procedure 1960 should be performed in the minimum amount of time necessary to fully initialize any sensor. When performed at this time, sensor life is maximized by limiting the time a fully initialized sensor is not in use and over-initialization is avoided by confirming full initialization of the sensor before too much initialization is performed. Preventing over-initialization is important because over-initialization leads to current suppression which leads to inaccurate readings. However, under-initialization is also a problem, so if the third diagnostic EIS procedure 1960 indicates that the sensor is under-initialized, then an optional initialization at point 1970 can be performed to fully initialize the sensor. Under-initialization is disadvantageous because it produces excessive current that is not related to the actual glucose concentration. Because of the dangers of under- and over-initialization, the third diagnostic EIS procedure plays an important role in ensuring that the sensor is functioning properly when in use.

[0294] Additionally, optional periodic diagnostic EIS procedures 1980 can be scheduled at a time after the sensor is fully initialized. The EIS procedures 1980 can be scheduled at any set interval. As will be discussed in greater detail below, the EIS procedures 1980 can also be triggered by other sensor signals, such as an abnormal current or an abnormal counter electrode voltage. Additionally, fewer or more EIS procedures 1980 can be scheduled as desired. In the preferred embodiment, the EIS procedures used during the hydration process, the sensor life check, the initialization process, or the periodic diagnostic test are the same procedures. In alternative embodiments, the EIS procedures can be shortened or lengthened (i.e., check fewer or more ranges of frequencies) for the various EIS procedures depending on the need to focus on a particular impedance range. The periodic diagnostic EIS procedures 1980 monitor the impedance values to ensure that the sensor continues to operate at an optimal level.

[0295] If the sensor current is decreasing due to a contaminant species, sensor age, or a combination of the two, the sensor can not be operating at an optimal level. Sensors that have aged beyond a certain length of time are no longer useful, but sensors that are hindered by a contaminant species can be repaired. Contaminant species can reduce the surface area of the electrode or the diffusion path of the analyte and reaction byproducts, resulting in a decrease in sensor current. These contaminant species are charged and gradually accumulate on the electrode or membrane surface at a certain voltage. Previously, contaminant species would destroy the effectiveness of the sensor. Now, if the periodic diagnostic EIS procedures 1980 detect impedance values that indicate the presence of a contaminant species, remedial action can be taken. With respect to FIG. 21A The description of when remedial action is taken. Thus, the periodic diagnostic EIS procedures 1980 become very useful because they can trigger sensor remedial action, which can restore the sensor current to a normal level and extend the life of the sensor. Two possible embodiments of sensor remedial action are described below in the description of FIG. 21B and FIG. 20 The two possible embodiments of sensor remedial action.

[0296] Additionally, any scheduled diagnostic EIS procedure 1980 can be suspended or rescheduled when certain events are determined to be imminent. Such events can include any situation that requires the patient to check the sensor reading, including, for example, when the patient uses a test strip meter to measure his or her BG level to calibrate the sensor, when the patient is warned of a calibration error and needs to use a test strip meter to measure his or her BG level again, or when a hyperglycemic or hypoglycemic warning has been issued but not acknowledged.

[0297] FIG. 19 A method of combining diagnostic EIS procedures with sensor remedial action is shown. Block 2000 the diagnostic procedure can be as described in FIG. 20Any periodic diagnostic EIS procedure 1980 is described in detail. The logic of this method begins when a diagnostic EIS procedure is performed at block 2000 to detect the impedance value of the sensor. As noted, in certain embodiments, the EIS procedure applies a combination of a DC bias voltage and an AC voltage of variable frequency, where the impedance detected by performing the EIS procedure is mapped on a Nyquist plot, and the inflection point in the Nyquist plot approximates the sum of the polarization resistance and the solution resistance (i.e., the real impedance value). After the diagnostic EIS procedure at block 2000 detects the impedance value of the sensor, the logic moves to block 2010.

[0298] At block 2010, the impedance value is compared to the set high and low thresholds to determine if the impedance value is normal. If the impedance is within the set limits of the high and low thresholds at block 2010, then normal sensor operation is resumed at block 2020, and FIG. 21A The logic will end until the time for another diagnostic EIS procedure is scheduled. Conversely, if the impedance is determined to be abnormal at block 2010 (i.e., outside the set limits of the high and low thresholds), then remedial action at block 2030 is triggered. Examples of high and low thresholds that can be acceptable during the life of the sensor are 5.5 kilo-ohms and 8.5 kilo-ohms, respectively, but these can be set higher or lower as desired.

[0299] The remedial action of block 2030 is performed to remove any contaminant species that can have caused the abnormal impedance value. In the preferred embodiment, the remedial action is performed by applying a reverse current or reverse voltage between the working electrode and the reference electrode. Details of the remedial action will be described in more detail with respect to FIG. 21. After the remedial action is performed at block 2030, the impedance value is again tested at block 2040 by a diagnostic EIS procedure. Then, when the impedance value from the diagnostic EIS procedure of block 2040 is compared to the set high or low threshold, it is determined at block 2050 that the remedial action was successful. As in block 2010, if the impedance is within the set threshold, this is considered normal, and if the impedance is outside the set threshold, this is considered abnormal.

[0300] If it is determined at block 2050 that the impedance value of the sensor has returned to normal, then normal sensor operation at block 2020 will proceed. If the impedance remains abnormal, indicating that sensor aging is the cause of the impedance anomaly or that the remedial measures failed to successfully remove the contaminant species, then the sensor is terminated at block 2060. In an alternative embodiment, instead of immediately terminating the sensor, the sensor can generate a sensor message that first requests the user to wait and then perform further remedial measures after a set period of time has elapsed. This alternative step can be combined with separate logic to determine whether, after performing the initial remedial measures, the impedance value is increasingly approaching within the bounds of the high and low thresholds. For example, if no change in the sensor impedance value is found, then the sensor can decide to terminate. However, if the sensor impedance value is increasingly approaching the preset bounds, but still outside the bounds after the initial remedial measures, then additional remedial measures can be performed. In yet another alternative embodiment, the sensor can generate a message requesting the user to calibrate the sensor by taking a finger prick meter measurement to further confirm whether the sensor is indeed malfunctioning. All of the above embodiments are intended to prevent the user from using a malfunctioning sensor that produces inaccurate readings.

[0301] FIG. 21B One embodiment of the foregoing sensor remedial measures is shown. In this embodiment, the blockage created by the contaminant species is removed by reversing the voltage applied across the sensor between the working electrode and the reference electrode. The reversed DC voltage lifts the charged contaminant species from the electrode or membrane surface, thereby clearing the diffusion path. Through the cleared path, the current of the sensor returns to normal levels, and the sensor can provide accurate readings. Thus, the remedial measures save the user time and money associated with replacing a sensor that would have otherwise worked.

[0302] FIG. 16B An alternative embodiment of the previously mentioned sensor remedial measures is shown. In this embodiment, the reversed DC voltage applied between the working electrode and the reference electrode is coupled with an AC voltage. By adding an AC voltage, certain tightly adsorbed species or species on the surface layer can be removed because the AC voltage can extend the force of the AC voltage further from the electrode and penetrate all layers of the sensor. The AC voltage can take any number of different waveforms. Some examples of waveforms that can be used include a square wave, a triangle wave, a sine wave, or a pulse. As with the previous embodiment, once the contaminant species is cleared, the sensor can return to normal operation, and both sensor life and accuracy are improved.

[0303] While the above examples primarily show the use of real impedance data in sensor diagnostics, embodiments of the application described herein are also directed to the use of other EIS-based and substantially analyte-independent parameters (in addition to real impedance) in sensor diagnostic procedures. For example, as previously described, analysis of (substantially) glucose-independent impedance data, such as the values of 1 kHz real impedance and 1 kHz imaginary impedance and the Nyquist slope, provides information about the efficiency of the sensor in terms of its hydration and readiness for data acquisition. In addition, (substantially) glucose-independent impedance data, such as the value of 1 kHz real impedance, provides information about potential occlusions that can be present on the surface of the sensor membrane that can temporarily prevent glucose from entering the sensor and thus cause a signal dip.

[0304] In addition, (substantially) glucose-independent impedance data, such as the values of high frequency phase angle and / or imaginary impedance at 1 kHz and higher frequencies, provides information about loss of sensor sensitivity during long wear, which can be due to local hypoxia at the insertion site. In this regard, the underlying mechanism of sensitivity loss due to hypoxia can be described as follows: when local hypoxia occurs, the sensor output (i.e., Isig and SG) will depend on oxygen rather than glucose, and thus, the sensor will lose sensitivity to glucose. Other markers including 0.1 Hz real impedance, counter electrode voltage (Vcntr), and EIS-induced spikes in Isig can also be used to detect sensitivity loss due to hypoxia. In addition, in sensor systems with redundant electrodes, the relative differences in 1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biofouling.

[0305] According to embodiments of the invention described herein, EIS-based sensor diagnostics requires consideration and analysis of EIS data relating to one or more of at least three main factors, namely potential sensor failure modes: (1) signal initiation; (2) signal descent; and (3) sensitivity loss. Importantly, it has been found that most impedance-related parameters used in such diagnostic analyses and procedures can be studied at or within a frequency range, wherein these parameters are substantially independent of the analyte, a finding that allows sensor diagnostic procedures to be performed independently of analyte levels in the patient. Thus, while EIS-based sensor diagnostics can be triggered by, for example, large fluctuations in the isig related to the analyte, the impedance-related parameters used in such sensor diagnostic procedures are themselves substantially independent of analyte levels. As will be discussed in more detail below, it has also been found that in most cases where it can be seen that glucose has an effect on the magnitude (or other characteristics) of EIS-based parameters, this effect is usually small enough, for example, there is at least an order of magnitude difference between the EIS-based measurement and the effect of glucose on said measurement, so that glucose can be screened out of the measurement, for example, by software in the IC.

[0306] By definition, "start-up" refers to the integrity of the sensor signal during the first few hours after insertion (e.g., t = 0 to 6 hours). For example, in many current devices, the signal is considered unreliable during the first 2 hours after insertion, and therefore, the patient / user cannot see the sensor glucose value. In cases where the sensor takes a longer time to hydrate, the sensor signal is low for several hours after insertion. Using EIS, additional impedance information can be obtained immediately after sensor insertion (by running the EIS procedure). In this regard, the principle of low-start-up detection using 1kHz real impedance can be explained using the total impedance equation. At relatively high frequencies (in this case, 1kHz and above), the virtual impedance is very small (as confirmed by in vivo data), therefore the total impedance decreases to:

[0307]

[0308] As the sensor gradually completes wetting, the double-layer capacitor (C) d The total impedance will decrease as a result of the increase in C. This is because, as shown in the equation above, the total impedance is related to C. d Inversely proportional. For example... FIG. 15B The intercept of the real impedance axis, 1600, is shown in the form shown. Importantly, the 1kHz virtual impedance can also be used for the same purpose because it also contains a capacitive component and is inversely proportional to it.

[0309] Another marker for low-startup detection is the Nyquist slope, which relies only on the relatively low frequency impedance, which in turn corresponds to the Warburg impedance component of the total impedance (see, e.g., FIG. 22 ). FIG. 23A A Nyquist plot of a sensor in normal operation is shown, with arrow A indicating the time progression from t=0, i.e., sensor wear time. Thus, EIS is performed at a relatively low frequency immediately after sensor insertion (time t=0), which yields real and imaginary impedance data that are plotted with a first linear fit 2200 having a first (Nyquist) slope. A second (lower) sweep is run at a time interval after t=0, which yields a second linear fit 2210 having a second (Nyquist) slope that is greater than the first Nyquist slope, and so on. As the sensor becomes more hydrated, the Nyquist slope increases, and the intercept decreases, as reflected by lines 2200, 2210, etc., which move with the intercept becoming steeper and closer to the Y-axis. In conjunction with low-startup detection, clinical data indicates that the Nyquist slope typically increases significantly after sensor insertion and initialization, and then stabilizes to some level. One explanation for this is that as the sensor gradually gets wet, the species diffusion rates, as well as the concentrations, undergo significant changes, which are reflected in the Warburg impedance.

[0310] In FIG. 23A , the Isig 2230 for the first working electrode WE1 starts lower (at about 10 nA) than expected, and takes some time to catch up to the Isig 2240 for the second working electrode WE2. Thus, in this particular example, WE1 is represented as having a low start-up. The EIS data reflects this low start-up in two ways. First, as shown in FIG. 23C , the real impedance at 1 kHz (2235) for WE1 is much higher than the 1 kHz real impedance 2245 for WE2. Second, the Nyquist slope for WE1 ( FIG. 23B ) starts lower, with a larger intercept 2237, and takes more time to stabilize, when compared to the Nyquist slope for WE2 ( FIG. 24 ). As will be discussed later, these two features, 1 kHz real impedance and Nyquist slope, can be used as diagnostic inputs in a fusion algorithm to decide which of the two electrodes can carry a higher weight in computing the fused signal. In addition, one or both of these markers can be used in a diagnostic procedure to determine whether the sensor is acceptable overall, or whether it should be terminated and replaced.

[0311] By definition, a signal (or Isig) dip refers to an instance of low sensor signal that is mostly temporary in nature, e.g., on the order of a few hours. Such low signals can be due to, e.g., some form of biological occlusion on the sensor surface or due to pressure applied at the insertion site (e.g., when sleeping on one's side). During this time period, the sensor data is considered unreliable; however, the signal does eventually recover. This type of signal dip, as opposed to signal dips caused by changes in blood glucose in the patient, is reflected in the 1 kHz real impedance data as shown in FIG. 24 .

[0312] In particular, in FIG. 24 , the Isig 2250 for the first working electrode WE1 and the Isig 2260 for the second working electrode WE2 both start at about 25 nA at the far left (i.e., 6 PM). Over time, both Isigs fluctuate, which reflects glucose fluctuations in the vicinity of the sensor. For the first 12 or so hours (i.e., until about 6 AM), both Isigs are fairly stable, as are their respective 1 kHz real impedances 2255, 2265. However, between about 12 and 18 hours, i.e., between 6 AM and noon, the Isig 2260 for WE2 starts to dip and continues to trend downward over the next few hours, until about 9 PM. During this time period, the Isig 2250 for WE1 also exhibits some dip, but compared to the Isig 2260 for WE2, the Isig 2250 is much more stable and the dip magnitude is much smaller. The characteristics of the Isigs for WE1 and WE2 are also reflected in their respective 1 kHz real impedance data. Thus, as shown in FIG. 25A , while the 1 kHz real impedance for WE1 (2255) remains fairly stable during the aforementioned time period, the 1 kHz real impedance for WE2 (2265) increases significantly.

[0313] By definition, a loss of sensitivity refers to a situation where the sensor signal (Isig) goes low and does not respond for a long period of time, and typically does not recover. A loss of sensitivity can occur for various reasons. For example, electrode poisoning greatly reduces the effective surface area of the working electrode, severely limiting the current amplitude. A loss of sensitivity can also occur at the insertion site due to low or no oxygen. In addition, a loss of sensitivity can occur due to some form of extreme surface occlusion (i.e., a more permanent form of signal dip caused by biology or other factors) that limits the passage of glucose and oxygen through the sensor membrane, thereby reducing the number / frequency of chemical reactions that generate current in the electrode and ultimately the sensor signal (Isig). It should be noted that the various reasons for the above loss of sensitivity apply to both short-term (7-10 day wear) and long-term (6 month wear) sensors.

[0314] In EIS data, sensitivity loss is typically preceded by an increase in the absolute value of the phase (|phase|) and imaginary impedance (|imaginary impedance|) at a relatively higher frequency range, e.g., 128 Hz and above and 1 kHz and above, respectively. FIG. 25A An example of a normally functioning glucose sensor is shown, where the sensor current 2500 responds to glucose, i.e., Isig 2500 tracks glucose fluctuations, but all of the relevant impedance outputs, e.g., 1 kHz real impedance 2510, imaginary 1 kHz impedance 2530, and phase at frequencies equal to or above about 128 Hz (2520) remain stable, because they are essentially glucose independent.

[0315] In particular, FIG. 25B The top plot in FIG. 25A shows that after the first few hours, the 1 kHz real impedance 2510 remains fairly stable at about 5 kilo-Ohms (and the 1 kHz imaginary impedance 2530 remains fairly stable at about -400 Ohms). In other words, at 1 kHz, the real impedance data 2510 and imaginary impedance data 2530 are essentially glucose independent, and thus they can be used as a characteristic or independent indicator of the health, condition, and ultimately the reliability of the particular sensor under analysis. However, as mentioned previously, different impedance-related parameters can exhibit glucose independence at different frequency ranges, and the ranges can depend on the overall sensor design in each case, e.g., electrode type, electrode surface area, membrane thickness, permeability of the membrane, etc.

[0316] Thus, in the example FIG. 25B In FIG. 25B, for a 90% short tubeless electrode design, the top plot again shows that the sensor current 2501 responds to glucose, and after the first few hours, the 1 kHz real impedance 2511 remains fairly stable at about 7.5 kilo-Ohms. FIG. 25A The bottom plot in FIG. 25A shows the real impedance data at frequencies between 0.1 Hz (2518) and 1 kHz (2511). As can be seen, the real impedance data at 0.1 Hz (2518) is very glucose dependent. However, as shown by reference numerals 2516, 2514, and 2512, as the frequency increases from 0.1 Hz to 1 kHz, i.e., for impedance data measured at frequencies close to 1 kHz, the real impedance becomes increasingly glucose independent.

[0317] Returning to FIG. 25C, the middle plot illustrates that the phase 2520 at relatively higher frequencies is essentially glucose independent. However, it should be noted that "relatively higher frequencies" in relation to this parameter (phase) of the sensor under analysis means frequencies of 128 Hz and above. In this regard, the plot shows that the phase for all frequencies between 128 Hz and 8 kHz is stable throughout the time period shown. On the other hand, as can be seen from the bottom plot for FIG. 25C , while the phase 2522 at 128 Hz (and above) is stable, the phase 2524 fluctuates at frequencies increasingly smaller than 128 Hz, i.e., the phase becomes increasingly glucose dependent and varies to different degrees. It should be noted that the electrode design for the example of FIG. 25B is the same as the electrode design for FIG. 26 , and the top plot for the former is the same as the top plot for the latter.

[0318] FIG. 26 shows an example of loss of sensitivity due to hypoxia at the insertion site. In this case, the insertion site becomes hypoxic (indicated by the dark vertical line in FIG. 26 ) after day 4, which causes the sensor current 2600 to go low and unresponsive. The 1 kHz real impedance 2610 remains stable, which indicates that there is no physical occlusion on the sensor. However, as indicated by the corresponding downward arrows, the changes in the phase 2622 at relatively higher frequencies and the 1 kHz imaginary impedance 2632 are consistent with the loss of sensitivity, which indicates that this loss is due to hypoxia at the insertion site. Specifically, FIG. 27 shows that the phase at higher frequencies (2620) and 1 kHz imaginary impedance (2630) becomes more negative (indicated by the dark vertical line) before the sensor loses sensitivity, and continues the downward trend as the sensor loses sensitivity continuously. Thus, as mentioned above, this loss of sensitivity is preceded or predicted by increasing the absolute values of the phase (|phase|) and imaginary impedance (|imaginary impedance|) at relatively higher frequency ranges (e.g., 128 Hz and above and 1 kHz and above, respectively).

[0319] The above features can be verified by in vitro testing, an example of which is shown in FIG. 27 . FIG. 28A through FIG. 28DResults of in vitro testing of a sensor are shown in which anoxia at different glucose concentrations was simulated. In the top plot, Isig fluctuates with glucose concentration as the glucose concentration is increased from 100 mg / dl (2710) to 200 mg / dl (2720), 300 mg / dl (2730), and 400 mg / dl (2740) and then decreased back to 200 mg / dl (2750). In the bottom plot, the phase at the relatively higher frequency is generally stable, indicating that the phase is independent of glucose. However, at very low oxygen concentrations, such as at 0.1% O2, the phase at the relatively high frequency fluctuates, as indicated by the circled areas and arrows 2760, 2770. Note that the magnitude and / or direction (i.e., positive or negative) of the fluctuations depends on various factors. For example, the higher the ratio of glucose concentration to oxygen concentration, the greater the magnitude of the phase shift. In addition, the particular sensor design, as well as the sensor age (i.e., measured by time post-implantation), affects this fluctuation. Thus, for example, the older the sensor, the more susceptible it is to interference.

[0320] FIG. 28A Another example of anoxia-induced loss of sensitivity for redundant working electrodes WE1 and WE2 is shown. As shown in FIG. 28B even though the sensor current 2800 fluctuates and eventually becomes unresponsive, the 1 kHz real impedance 2810 is stable. Also, as previously noted, the change in 1 kHz imaginary impedance 2820 is consistent with the loss of sensitivity of the sensor. However, in addition, FIG. 28C Real and imaginary impedance data at 0.105 Hz are shown (2830 and 2840, respectively). The imaginary impedance data can be more commonly referred to as "0.1 Hz data," which indicates that while the imaginary impedance at 0.1 Hz appears to be fairly stable, the 0.1 Hz real impedance 2830 increases significantly as the sensor loses sensitivity. Moreover, as shown in FIG. 28A through FIG. 28B the V cntr 2850 drops to 1.2 volts.

[0321] In summary, the figures show the following findings: anoxia-induced loss of sensitivity is accompanied by lower 1 kHz imaginary impedance (i.e., the latter becomes more negative), higher 0.105 Hz real impedance (i.e., the latter becomes more positive), and V cntr drop. Moreover, the anoxia process and V cntr drop are generally accompanied by an increase in the capacitive component in the electrochemical circuit. Note that in some diagnostic procedures, described later, the 0.105 Hz real impedance can not be used because the real impedance data at this relatively low frequency can appear to be analyte-related.

[0322] Finally, in conjunction FIG. 28DFor example, it should be noted that impedance measurements at frequencies of 1 kHz or higher typically cause spikes in the Isig due to EIS. This is in FIG. 29 The diagram shows the original Isig of WE2 plotted at the control time. The sharp increase in Isig at the onset of the spike is a non-Radaic process due to the double-layer capacitance charge. Therefore, the sensitivity loss caused by hypoxia may also be accompanied by a spike caused by a higher EIS, as well as a lower virtual impedance at 1 kHz, a higher real impedance at 0.105 Hz, and V... cntr The decline, as discussed above.

[0323] FIG. 24 This illustrates another example of sensitivity loss. This situation can be considered in conjunction with the above. FIG. 25A This is an extreme version of the described Isig drop. Here, the sensor current 2910 is observed to be low from the start of insertion time, indicating a problem with the insertion procedure, leading to electrode blockage. FIG. 30A through FIG. 30D Compared to the same parameter values ​​of a normally operating sensor, the 1kHz real impedance 2920 is significantly higher, while the relatively higher frequency phase 2930 and 1kHz virtual impedance 2940 have shifted to significantly more negative values. The shift in the relatively higher frequency phase 2930 and 1kHz virtual impedance 2940 indicates that the sensitivity loss may be due to oxygen deficiency, which could be caused by blockage on the sensor surface.

[0324] FIG. 30D Data from another redundant sensor is shown, where the relative differences between the 1kHz real impedance and 1kHz virtual impedance, and the 0.1Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biofouling. In this example, WE1 exhibits a greater sensitivity loss than WE2, as evidenced by WE2's higher 1kHz real impedance 3010, lower 1kHz virtual impedance 3020, and much higher real impedance at 0.105Hz (3030). However, in this example, V cntr 3050 will not decrease. Furthermore, if... FIG. 31 As shown, the height of the spikes in the original Isig data does not change significantly over time. This indicates that V... cntr The decrease and the increase in peak height are related. In addition, the fact that the peak height in the original Isig data does not change much over time indicates that the capacitive component of the circuit does not change significantly over time. Therefore, the sensitivity loss due to biofouling is related to the resistive component of the circuit (i.e., diffusion).

[0325] Various impedance-related parameters described above can be used, alone or in combination, as inputs to the following: (1) EIS-based sensor diagnostic procedures; and / or (2) fusion algorithms. Regarding the former, FIG. 31 It is shown how EIS-based data, i.e., impedance-related parameters or characteristics, are used in diagnostic procedures to determine in real-time whether a sensor is functioning properly, or whether it should be replaced.

[0326] The diagnostic procedures shown in the flowchart of FIG. 31 The diagnostic procedures shown in the flowchart of

[0327] The time frequency of EIS implementation and data collection can be determined by various factors. For example, each implementation of EIS consumes an amount of power, which is typically provided by the sensor's battery, i.e., the battery that runs the sensor electronics, including the ASIC described later. Thus, the battery capacity, as well as the remaining sensor life, can contribute to determining the number of times EIS is run, as well as the frequency breadth sampled at each such run. In addition, certain conditions can require monitoring of EIS parameters at certain frequencies (e.g., real impedance at 1 kHz) based on a first schedule (e.g., once every few seconds or minutes), while other parameters and / or the same parameters at other frequencies can be monitored based on a second schedule (e.g., less frequently). In these cases, the diagnostic procedures can be customized for a particular sensor and requirements, such that battery power can be conserved, and unnecessary and / or redundant EIS data acquisition can be avoided.

[0328] It is noted that in certain embodiments, for example FIG. 31The diagnostic procedure shown requires a series of individual "tests" that are implemented in order to perform real-time monitoring of the sensor. The multiple tests, or markers, are implemented because each time EIS is run, i.e., each time the EIS procedure is performed, data can be collected about multiple impedance-based parameters or characteristics that can be used to detect sensor status or quality, including, for example, whether the sensor has failed or is failing. In performing sensor diagnostics, sometimes there can be diagnostic tests that can indicate a failure, while other diagnostics can indicate no failure. Thus, the availability of multiple impedance-related parameters and the implementation of a multiple test procedure is advantageous because some of the multiple tests can act as a validity check for certain other tests. Thus, real-time monitoring using a multiple marker procedure can include a degree of built-in redundancy.

[0329] In view of the above, FIG. 32A The logic of the diagnostic procedure shown begins at 3100 after the sensor has been inserted / implanted and an EIS run has been performed, in order to provide EIS data as input. At 3100, using the EIS data as input, it is first determined whether the sensor is still in place. Thus, if it is found that the |Z| slope is constant over the frequency band (or range) being tested, and / or the phase angle is about -90°, then it is determined that the sensor is no longer in place, and a warning is sent to the patient / user, for example, indicating that sensor pull-out has occurred. The particular parameters (and their respective values) described herein for detecting sensor pull-out are based on the finding that once the sensor leaves the body, and the membrane is no longer hydrated, the impedance spectral response behaves like a capacitor.

[0330] If it is determined that the sensor is still in place, then the logic moves to step 3110 to determine whether the sensor is properly initialized. As shown, the "Init. Check" is performed by determining the following conditions: (i) whether |(Z n - Z1) / Z1| > 30%, where Z1is the real impedance measured the first time, and Z n is the impedance measured at the next interval, as discussed above; and (2) whether the phase angle change at 0.1 Hz is greater than 10°. If the answer to either of the questions is "yes," then the test is satisfied, i.e., test 1 is passed. Otherwise, test 1 is marked as failed.

[0331] At step 3120, test 2 asks whether the frequency difference (f2- fl) between two consecutive EIS runs at a phase angle of -45° is greater than 10 Hz. Again, an answer of "no" is marked as failed; otherwise, test 2 is fully satisfied.

[0332] Test 3 at step 3130 is a hydration test. Here, it is asked whether the current impedance Z nIs the initialized post-impedance Z less than 1 kHz pi If so, then this test is satisfied; otherwise, test 3 is flagged as failed. Test 4 in step 3140 is also a hydration test, but this time at a lower frequency. Thus, this test asks whether Z n Is less than 300 kilo-ohms at 0.1 Hz. Again, a "no" answer indicates that the sensor failed test 4.

[0333] At step 3150, test 5 asks whether the low frequency Nyquist slope increases overall from 0.1 Hz to 1 Hz. As discussed previously, for a normally operating sensor, the Nyquist slope at relatively low frequencies should increase over time. Thus, if the answer to the inquiry is "yes," then this test is satisfied; otherwise, the test will be flagged as failed.

[0334] Step 3160 is the last test of this embodiment of the diagnostic procedure. Here, the inquiry is whether the real impedance decreases overall. Here, as discussed previously, in a normally operating sensor, the real impedance is expected to decrease over time. Thus, an answer of "yes" here means that the sensor is operating properly; otherwise, the sensor fails test 6.

[0335] Once all 6 tests have been implemented, a decision is made at 3170 as to whether the sensor is operating properly or whether a fault has occurred. In this embodiment, if the sensor passes at least 3 of the 6 tests, then the sensor is determined to be operating properly (3172). In other words, for the sensor to be determined as faulty (3174), it must fail at least 4 of the 6 tests. In alternative embodiments, different rules can be used to assess proper operation versus sensor failure. Additionally, in some embodiments, each test can be weighted in determining overall sensor operation (proper versus failed), such that the assigned weight reflects, for example, the importance of the test, or the importance of the particular parameter that the test inquires about. For example, one test can be weighted twice as much as another test, but only half as much as a third test, etc.

[0336] In other alternative embodiments, a different number of tests and / or a different set of EIS-based parameters can be used for each test. FIG. 32B and FIG. 32A An example of a diagnostic procedure for real-time monitoring is shown that includes 7 tests. Reference is made to FIG. 31, the logic starts at 3200 after the sensor has been inserted / implanted and the EIS procedure has been performed, in order to provide EIS data as input. At 3200, using the EIS data as input, it is first determined whether the sensor is still in place. Thus, if it is found that the |Z| slope is constant over the frequency band (or range) under test, and / or the phase angle is about -90°, then it is determined that the sensor is no longer in place, and a warning is sent to the patient / user, for example, indicating that sensor pull-out has occurred. On the other hand, if it is determined that the sensor is in place, then the logic moves to start diagnostic checks (3202).

[0337] At 3205, Test 1 is similar to Test 1 of the diagnostic procedure discussed above in connection with FIG. 31 , but this Test 1 provides for a subsequent measurement Z n at 2 hours after the first measurement. Thus, in this example, Z n = Z 2hr . More specifically, Test 1 compares the real impedance at 2 hours after sensor implantation and initialization to the value before initialization. Similarly, a second part of Test 1 asks whether the phase at 2 hours after initialization differs from the phase between the initialization and pre- initialization stages by more than 10° at 0.1 Hz. As before, if the answer to either query is positive, then the sensor is determined to be normally hydrated and initialized, and Test 1 is satisfied; otherwise, the sensor fails this test. It should be noted that even though this test queries about impedance and phase changes at 2 hours after initialization, the time interval between any two consecutive EIS runs can be shorter or longer, depending on a variety of factors, including, for example, sensor design, level of electrode redundancy, extent to which the diagnostic procedure includes redundant tests, battery power, etc.

[0338] Moving to 3210, the logic next performs a sensitivity loss check by asking whether the impedance magnitude percent change at 1 kHz after a 2-hour interval (n+2) and the impedance magnitude percent change in Isig are greater than 30%. If the answer to both queries is "yes", then the sensor is determined to be losing sensitivity, and, therefore, Test 2 is determined to fail. It should be noted that although Test 2 is shown herein based on a preferred percent difference of 30%, in other embodiments, the percent difference in impedance magnitude at 1 kHz and in Isig can range from 10% to 50% for purposes of conducting this test.

[0339] Test 3 (at 3220) is similar to FIG. 33ATest 5 of the illustrated algorithm. Here, as before, the question is whether the low frequency Nyquist slope increases monotonically from 0.1 Hz to 1 Hz. If so, then this test passes; otherwise, the test fails. As shown in 3220, this test can also set a threshold or acceptable range for the percent change in the low frequency Nyquist slope, beyond which the sensor can be deemed to be malfunctioning, or at least trigger additional diagnostic tests. In embodiments of the application, such a threshold / acceptable range for the percent change in the low frequency Nyquist slope can be in the range of about 2% to about 20%. In some preferred embodiments, the threshold can be about 5%.

[0340] The logic next moves to 3230, which is another low frequency test, this time involving the phase and impedance magnitude. More specifically, the phase test asks whether the phase at 0.1 Hz is increasing monotonically over time. If so, then the test fails. As with the other tests that monitor trends in parameters, the low frequency phase test of test 4 can also set a threshold or acceptable range for the percent change in the low frequency phase, beyond which the sensor can be deemed to be malfunctioning, or at least cause a problem. In some preferred embodiments, such a threshold / acceptable range for the percent change in the low frequency phase can be in the range of about 5% to about 30%. In some preferred embodiments, the threshold can be about 10%.

[0341] As noted, test 4 also includes a low frequency impedance magnitude test, in which the question is whether the impedance magnitude at 0.1 Hz is increasing monotonically over time. If so, then the test fails. Note that if either the phase test or the impedance magnitude test fails, then test 4 is deemed to have failed. The low frequency impedance magnitude test of test 4 is also suitable for setting a threshold or acceptable range for the percent change in the low frequency impedance magnitude, beyond which the sensor can be deemed to be malfunctioning, or at least cause a problem. In some preferred embodiments, such a threshold / acceptable range for the percent change in the low frequency impedance magnitude can be in the range of about 5% to about 30%. In some preferred embodiments, the threshold can be about 10%, where the range of impedance magnitudes for a typical sensor is generally between about 100 kilo-ohms and about 200 kilo-ohms.

[0342] Test 5 (at 3240) is another sensitivity loss check that can be considered a complement to test 2. Here, if both the percent change in Isig and the percent change in magnitude of impedance at 1 kHz are greater than 30%, then the sensor is determined to be recovering from a loss of sensitivity. In other words, even if test 2 does not detect a loss of sensitivity for some reason, the sensor is determined to have previously experienced some loss of sensitivity. As with test 2, while test 5 is shown based on a preferred percent difference of 30%, in other embodiments, the percent difference in Isig and the magnitude of impedance at 1 kHz can range from 10% to 50% for purposes of conducting this test.

[0343] Moving to 3250, test 6 provides specific failure criteria for the sensor functionality test, which have been determined based on observed data and a particular sensor design. Specifically, in one embodiment, if at least two of the following three criteria are met, then the sensor can be determined to have failed and thus is unlikely to respond to glucose: (1) Isig is less than 10 nA; (2) imaginary impedance at 1 kHz is less than -1500 ohms; and (3) phase at 1 kHz is less than -15°. Thus, if any two of (1) to (3) are not met, then test 6 is determined to have passed. It should be noted that in other embodiments, the Isig prong of this test can fail if Isig is less than about 5 nA to about 20 nA. Similarly, the second prong can fail if the imaginary impedance at 1 kHz is less than about -1000 ohms to about -2000 ohms. Finally, the phase prong can fail if the phase at 1 kHz is less than about -10° to about -20°.

[0344] Finally, step 3260 provides another sensitivity check, in which a parameter is evaluated at a low frequency. Thus, test 7 asks whether the magnitude of the difference between, on one hand, the ratio of imaginary impedance to Isig at 0.1 Hz (n+2) and, on the other hand, the permeable value of the ratio, is greater than 30% of the magnitude of the previous value of the ratio. If so, the test fails; otherwise, the test passes. Here, while test 7 is shown based on a preferred percent difference of 30%, in other embodiments, the percent difference can range from 10% to 50% for purposes of conducting this test.

[0345] Once all 7 tests have been implemented, a decision is made at 3270 as to whether the sensor is functioning properly or whether a warning should be issued indicating that a sensor failure (or possible failure) has occurred. As shown, in this embodiment, if the sensor passes at least 4 of the 7 tests, then the sensor is determined to be functioning properly (3272). In other words, in order to be determined to have failed or at least to have caused a problem (3274), the sensor must fail at least 4 of the 7 tests. If the sensor is determined to be "bad" (3274), a warning of the effect can be sent, for example, to the patient / user. As previously mentioned, in alternative embodiments, different rules can be used to assess the relationship between proper operation and sensor failure / problem. In addition, in some embodiments, each test can be weighted in determining overall sensor operation (proper vs. failed), such that the assigned weight reflects, for example, the importance of the test, or the importance of the particular parameter interrogated by the test.

[0346] As previously mentioned, in embodiments of the application described herein, various impedance-related parameters described above can be used, individually or in combination, as inputs to one or more fusion algorithms to produce more reliable sensor glucose values. In particular, as is well known, unlike single sensor (i.e., single working electrode) systems, multiple sensing electrodes provide glucose readings that are more reliable because the multiple signals obtained from two or more working electrodes can be fused to provide a single sensor glucose value. This signal fusion utilizes the quantitative inputs provided by EIS to calculate the most reliable output sensor glucose value from the redundant working electrodes. It should be noted that although the subsequent discussion can describe various fusion algorithms with respect to a first working electrode (WE1) and a second working electrode (WE2) as redundant electrodes, this is merely illustrative and not limiting, as the algorithms described herein and their underlying principles are applicable to and can be used in redundant sensor systems having more than 2 working electrodes. In addition, the redundant electrodes can be contained in (identical) sensors on / in a single flexible portion or multiple flexible portions, or the redundant electrodes can be contained in different sensors on / in a single flexible portion or multiple flexible portions (e.g., in a complex redundant sensor system having two or more sensors, where at least two of the sensors have different designs from one another).

[0347] FIG. 33B and FIG. 33A Top-level flowcharts are shown for two alternative methods, each of which includes a fusion algorithm. In particular, FIG. 33B is a flowchart involving a current (Isig) based fusion algorithm, and FIG. 33A is a flowchart for sensor glucose (SG) fusion. As can be seen from the figures, the main difference between the two methods is the calibration time. Thus, FIG. 33AIt is shown that for Isig fusion, calibration 3590 is performed after fusion 3540 is complete. That is, the redundant Isigs from WE1 to WEn are fused into a single Isig 3589, which is then calibrated to produce a single sensor glucose value 3598. On the other hand, for SG fusion, calibration 3435 is performed for each individual Isig from WE1 to WEn to produce calibrated SG values (e.g., 3436, 3438) for each working electrode. Thus, the SG fusion algorithm enables independent calibration of each of the multiple Isigs, which can be preferred in some embodiments of the application described herein. After calibration, the multiple calibrated SG values are fused into a single SG value 3498.

[0348] Importantly, it should be noted that, FIG. 33B and FIG. 34 Each of the flowcharts shown contains a spike filtering process (3520, 3420). As described in the discussion above regarding sensitivity loss, impedance measurements at 1 kHz or higher frequencies often induce EIS-induced spikes in the Isig. Thus, once an EIS procedure has been performed on each of the electrodes WE1 to WEn, it is preferred that for both SG fusion and Isig fusion, the Isigs 3410, 3412, etc. and 3510, 3512, etc. are first filtered to obtain the corresponding filtered Isigs 3422, 3424, etc. and 3522, 3524, etc. The filtered Isigs are then used for Isig fusion, or first calibrated and then used for SG fusion, as described below. It will become apparent in the subsequent discussion that both fusion algorithms require weights to be calculated and assigned based on various factors.

[0349] FIG. 33A through FIG. 35 Details of the fusion algorithm 3440 for SG fusion are shown. Essentially, four factors need to be checked before determining the fusion weights. First, integrity check 3450 involves determining whether each of the following parameters is within a specified range for normal sensor operation (e.g., predetermined lower and upper threshold values): (i) Isig; (ii) 1 kHz real and imaginary impedances; (iii) 0.105 Hz real and imaginary impedances; and (iv) Nyquist slope. As shown, integrity check 3450 contains boundary check 3452 and noise check 3456, where for each check, the above-mentioned parameters are used as input parameters. It should be noted that for brevity, only the 1 kHz real and imaginary impedances are presented in the figures as input parameters for the integrity check. However, it should be understood that the 0.105 Hz real and imaginary impedances and the Nyquist slope can also be used as input parameters for the integrity check. FIG. 33A The real and / or imaginary impedances at one or more frequencies, referred to simply as “Imp” to refer to impedance, are presented in the figures above for brevity. In addition, the magnitude and phase of the impedance (also shown as input in the FIG. 34 and 33B figures above can also be used to calculate both the real and imaginary impedances.

[0350] The output from each of the boundary check 3452 and the noise check 3458 is a respective reliability indicator (RI) for each of the redundant working electrodes. Thus, the output from the boundary check contains, for example, RI_bound_Wei (3543) and RI_bound_We2 (3454). Similarly, for the noise check, the output contains, for example, RI_noise_Wei (3457) and RI_noise_We2 (3458). The boundary and noise reliability indicators for each working electrode are calculated based on consistency with the above-described ranges for normal sensor operation. Thus, if any of the parameters are outside of the specified range for a particular electrode, the reliability indicator for that particular electrode is decreased.

[0351] It should be noted that the thresholds or ranges for the above-described parameters can depend on various factors, including the particular sensor and / or electrode design. However, in one preferred embodiment, typical ranges for some of the above-described parameters can be, for example, as follows: boundary threshold for 1 kHz real impedance = [0.3e+4 2e+4]; boundary threshold for 1 kHz imaginary impedance = [-2e+3, 0]; boundary threshold for 0.105 Hz real impedance = [2e+4 7e+4]; boundary threshold for 0.105 Hz imaginary impedance = [-2e+5 -0.25e+5]; and boundary threshold for Nyquist slope = [2 5]. The noise can be calculated, for example, using a second-order central difference method, where the noise is considered to be outside of the noise range if it is higher than a certain percentage (e.g., 30%) of the median of each variable buffer.

[0352] Second, sensor current (Isig) and 1 kHz real impedance can be used to detect sensor dropout. Thus, as shown in FIG. 34, Isig and “Imp” are used as inputs to dropout detection 3460. Here, the first step is to determine whether there is any divergence between Isig, and whether any such divergence is reflected in the 1 kHz real impedance data. This can be accomplished by using a mapping 3465 between an Isig similarity indicator (RI_sim_isig12) 3463 and a 1 kHz real impedance similarity indicator (RI_sim_imp12) 3464. This mapping is important because it helps to avoid false positives in cases where dropout is not true. When Isig divergence is true, the algorithm will select the sensor with the higher Isig. FIG. 35

[0353] According to one embodiment, divergence / convergence of two signals (e.g., two Isig, or two 1 kHz real impedance data points) can be calculated as follows:

[0354] diff_val = abs(val - (val + val2) / 2);

[0355] ​diff_va2 = abs(va2 - (va1 + va2) / 2);

[0356] RI_sim = 1 - (diff_va1 + diff_va2) / (mean(abs(va1 + va2)) / 4)

[0357] where va1 and va2 are two variables, and RI_sim (similarity index) is an index that measures the convergence or divergence of the signal. In this embodiment, RI_sim must be limited between 0 and 1. Thus, if RI_sim as calculated above is less than 0, it is set to 0, and if RI_sim is greater than 1, it is set to 1.

[0358] Mapping 3465 is performed by using ordinary linear regression (OLR). However, when OLR does not work well, robust median slope linear regression (RMSLR) can be used. For example, for the Isig similarity index and the 1 kHz real impedance index, two mapping procedures are needed: (i) mapping the Isig similarity index to the 1 kHz real impedance similarity index; and (ii) mapping the 1 kHz real impedance similarity index to the Isig similarity index. These two mapping procedures result in two residuals: res12 and res21. Each of the dip reliability indices 3467, 3468 can then be calculated as follows:

[0359] RI_dip = 1 - (res12 + res21) / (RI_sim_isig + RI_sim_1K_real_impedance).

[0360] A third factor is sensitivity loss 3470, which can be detected using, for example, the 1 kHz imaginary impedance trend over the past 8 hours. If the trend of one sensor becomes negative, the algorithm will rely on the other sensor. If both sensors lose sensitivity, a simple average will be taken. The trend is calculated by smoothing the 1 kHz imaginary impedance using a strong low-pass filter (which tends to have noise), and by using, for example, the correlation coefficient or linear regression over the past 8 hours, to determine if the correlation coefficient is negative or the slope is negative. Each of the sensitivity loss reliability indices 3473, 3474 is then assigned a binary value of 1 or 0.

[0361] The total reliability index (RI) for each of we1, we2,..., wen is calculated as follows:

[0362] RI_we1 = RI_dip_we1 x RI_sensitivity_loss_we1 x RI_bound_we1 x RI_noise_we1

[0363] RI_we2 = RI_dip_we2 x RI_sensitivity_loss_we2 x RI_bound_we2 x RI_noise_we2

[0364] RI_we3 = RI_dip_we3 x RI_sensitivity_loss_we3 x RI_bound_we3 x RI_noise_we3

[0365] RI_we4 = RI_dip_we4 x RI_sensitivity_loss_we4 x RI_bound_we4 x RI_noise_we4 . . .

[0369] RI_we n = RI_dip_we n x RI_sensitivity_loss_we n x RI_bound_we n x RI_noise_we n

[0370] After calculating the respective reliability indices of the individual working electrodes, the weight of each electrode can be calculated as follows:

[0371] weight_we1 = RI_we1 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 +... + RI_we n )

[0372] weight_we2 = RI_we2 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 +... + RI_we n )

[0373] weight_we3 = RI_we3 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 +... + RI_we n )

[0374] weight_we4 = RI_we4 / (RI_we1 + RI_we2 + RI_we3 + RI_we4 +... + RI_we n ) . . .

[0378] weight_we n = RI_wen / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0379] Based on the above, the fused SG 3498 is then calculated as follows:

[0380] SG=weight_we1×SG_we1+weight_we2×SG_we2+weight_we3×SG_we3+

[0381] weight_we4×SG_we4+...+weight_we n ×SG_we n

[0382] The final factor relates to artifacts in the final sensor readings, which may be caused by instantaneous weight changes during sensor fusion. This can be avoided by applying a low-pass filter 3480 to smooth the RI of each electrode, or by applying a low-pass filter to the final SG. When using the former approach, the filtered reliability metrics, such as RI_We1* and RI_We2* (3482, 3484), are used to calculate the weights of each electrode and thus to calculate the fused SG 3498.

[0383] FIG. 34 Details of the fusion algorithm 3540 used for Isig fusion are shown. It can be seen that this algorithm is essentially similar to... FIG. 33B The algorithm for SG fusion is shown, but with two differences. First, as previously stated, for Isig fusion, the calibration process is the final step, in which the Isig 3589 of the individual fusions is calibrated to produce a single sensor glucose value 3598. See also FIG. 33A through FIG. 35 Secondly, although SG fusion uses the SG values ​​of multiple electrodes to calculate the final SG value 3498, the fused Isig value 3589 is calculated using the filtered Isig values ​​(3522, 3524, etc.) of multiple electrodes.

[0384] In a closed loop study involving a non-diabetic population, the aforementioned fusion algorithm has been found to achieve a significant improvement in mean absolute relative difference (MARD) on day 1 (when the issue of low start is most severe and therefore can have a significant impact on the accuracy and therefore reliability of the sensor) and overall (i.e., over the 7-day lifetime of the sensor). The study evaluated data from an 88% distributed layout design utilizing high current density (nominal) plating using three different methods: (1) calculating one sensor glucose value (SG) by fusion using the Ferrari algorithm 1.0 by Medtronic Minimed (which is the SG fusion algorithm as discussed above); (2) calculating one SG by using 1 kHz EIS data to identify better ISIG values (by the Isig fusion algorithm discussed above); and (3) calculating one SG by using higher ISIG values (i.e., without using EIS). The details of the study data are as follows:

[0385] (1) SG based on Ferrari 1.0 Alg for 88% distributed layout utilizing high current density (nominal) plating

[0386]

[0387]

[0388]

[0389]

[0390] (2) SG based on better ISIG, using 1 kHz EIS for 88% distributed layout utilizing high current density (nominal) plating

[0391]

[0392]

[0393]

[0394]

[0395]

[0396]

[0397] (3) SG based on higher ISIG for 88% distributed layout utilizing high current density (nominal) plating

[0398]

[0399] ​

[0400]

[0401]

[0402]

[0403] From the above data, it has been found that for the first method, the MARD (%) on day 1 was 19.52% and the overall MARD was 12.28%. For the second method, the MARD on day 1 was 15.96% and the overall MARD was 11.83%. Finally, for the third method, the MARD on day 1 was 17.44% and the overall was 12.26%. Thus, for this design with redundant electrodes, it appears that calculating the SG based on better ISIGs using 1 kHz EIS (i.e., the second method) provides the greatest advantage. In particular, the lower day 1 MARD can be due to, for example, better low start-up detection using EIS. In addition, in this study, the overall MARD percentage was more than 1% lower than the overall average MARD of 13.5% for WE1 and WE2. It should be noted that in the above methods, data transitions can be handled, for example, by a filtering method that minimizes the severity of transitions, such as by using a low pass filter as discussed above in connection with FIG. 33B. FIG. 35

[0404] It is worth emphasizing that sensor diagnostics that include, for example, assessment of low start-up, sensitivity loss, and signal drop-out events depend on various factors, including sensor design, number of electrodes (i.e., redundancy), electrode distribution / configuration, etc. Thus, the actual frequency or frequency range based on EIS parameters that can be substantially glucose independent, and thus independent marker or predictor of one or more of the above failure modes, can also depend on the particular sensor design. For example, although it has been found that sensitivity loss can be predicted using imaginary impedance at a relatively high frequency - where the imaginary impedance is substantially glucose independent - as described above, the particular frequency range that is glucose level dependent and thus used as a marker for sensitivity loss can shift (higher or lower) depending on the actual sensor design.

[0405] More specifically, as sensor designs increasingly trend towards using redundant working electrodes, the size of the redundant working electrodes must increasingly decrease in order to maintain the overall size of the sensor. The size of the electrodes, in turn, affects the frequency at which certain diagnostics can be queried. In this regard, it is important to note that the diagnostics described herein and illustrated in FIG. 33A through FIG. 36 ​The fusion algorithms shown are considered illustrative, rather than limiting, as each algorithm can be modified as needed based on the type of sensors in the analysis to use EIS-based parameters at frequencies that exhibit minimal glucose dependence.

[0406] In addition, experimental data indicates that human tissue structure can also affect glucose dependence at different frequencies. For example, for children, it has been found that the real impedance at 0.105 Hz is a substantially glucose independent indicator for low start-up detection. This can be attributed to changes in tissue structure for children, such as Weber impedance changes, which are primarily related to the real component. See also the subsequent discussion on interferent detection.

[0407] Embodiments of the present invention described herein are also directed to the use of EIS in optimizing sensor calibration. As background, in current methods, the BG versus Isig slope used to calibrate subsequent Isig values is calculated as follows:

[0408]

[0409] where a is an exponential function of the time constant, β is a function of the blood glucose variance, and the offset is a constant. For sensors in stable conditions, this method provides reasonably accurate results. As shown, for example, in FIG. 37 BG and Isig follow a reasonably linear relationship, and the offset can be considered a constant.

[0410] However, there are instances where the above linear relationship does not hold, such as during periods when the sensor is experiencing a transition. As shown, for example, in FIG. 37 It is clear that pairs 1 and 2 are significantly different from pairs 3 and 4 in terms of the Isig versus BG relationship. For these types of conditions, using a constant offset tends to produce inaccurate results.

[0411] To address this issue, one embodiment is directed to the use of a dynamic offset based on EIS, where EIS measurements are used to define a sensor state vector as follows:

[0412] V = {real_imp_1K, img_imp_1K, Nyquist_slope, Nyquist_R_square}

[0413] where all of the elements in the vector are substantially independent of BG. Note that Nyquist_R_square is the R square of the linear regression used to calculate the Nyquist slope, i.e., the square of the correlation coefficient between the real and imaginary impedance at relatively low frequencies, and a low R square indicates abnormal sensor performance. For each Isig-BG pair, a state vector is assigned. If a significant difference in the state vector is detected, for example,FIG. 37 |V2-V3| of the example shown, then 3 and 4 are assigned different offset values when compared to 1 and 2. Thus, by using this dynamic offset method, it is possible to maintain the linear relationship between Isig and BG.

[0414] In a second embodiment, an EIS-based segmentation method can be used for calibration. Using the example and vector V, it can be determined that the sensor state during 1 and 2 is significantly different from the sensor state during 3 and 4. Thus, the calibration buffer can be segmented into two segments, as follows: FIG. 38A

[0415] Isig_buffer1 = [Isig1, Isig2]; BG_buffer1 = [BG1, BG2]

[0416] Isig_buffer2 = [Isig3, Isig4]; BG_buffer2 = [BG3, BG4]

[0417] Thus, when the sensor is operating during 1 and 2, Isig_buffer1 and BG_buffer1 will be used for calibration. However, when the sensor is working during 3 and 4, i.e. during the transition period, Isig_buffer2 and BG_buffer2 will be used for calibration.

[0418] In yet another embodiment, an EIS-based dynamic slope method can be used for calibration purposes, in which EIS is used to adjust the slope. FIG. 38A An example is shown of how this method can be used to improve sensor accuracy. In this figure, data points 1 to 4 are discrete blood glucose values. From FIG. 38A It can be seen that there is a sensor drop 3810 between data points 1 and 3, which can be detected using the sensor state vector V described above. During the drop, the slope can be adjusted upwards to reduce underreading, as indicated by reference numeral 3820 in FIG. 38B

[0419] In another embodiment, EIS diagnostics can be used to determine the timing of sensor calibration, which is useful for e.g. low-start events, loss of sensitivity events and other similar situations. As is known, most current methods require periodic calibration based on a preset schedule, e.g. 4 times per day. However, with the use of EIS diagnostics, calibration becomes event-driven, and thus calibration is only performed when necessary and most effective. Here again, the state vector V can be used to determine when the sensor state has changed, and to request calibration if indeed it has changed.

[0420] ​​More specifically, in the illustrative example, FIG. 33A through FIG. 35 A flowchart showing EIS-assisted sensor calibration involving low-startup detection is shown. By using the Nyquist slope, 1 kHz real impedance, and boundary checks 3850 (e.g., in conjunction with the fusion algorithm of FIG. 38B the aforementioned boundary checks and associated thresholds based on EIS-based parameters), a reliability index 3853 can be developed to proceed to startup such that when the 1 kHz real impedance 3851 and Nyquist slope 3852 are below their corresponding upper bounds, RI startup = 1, and the sensor is ready for calibration. In other words, the reliability index 3853 is "high" (3854), and the logic can proceed to calibration at 3860.

[0421] On the other hand, when the 1 kHz real impedance and Nyquist slope are above their corresponding upper bounds (or thresholds), RI startup = 0 (i.e., is "low"), and the sensor is not yet ready for calibration (3856), i.e., there can be a low-startup issue. Here, the trends of the 1 kHz real impedance and Nyquist slope can be used to predict when both parameters are within range (3870). If it is estimated that this will only take a very short time (e.g., less than an hour), then the algorithm will wait until the sensor is ready, i.e., until the aforementioned EIS-based parameters are within bounds (3874), at which time the algorithm proceeds to calibration. However, if the wait time is relatively long (3876), then the sensor can be calibrated now, and then the slope or offset can be adjusted gradually (3880) according to the trends of the 1 kHz real impedance and Nyquist slope. It should be noted that by performing the adjustment, severe over-reading or under-reading due to low-startup can be avoided. As previously mentioned, the EIS-based parameters and related information used in the on-the-fly calibration algorithm are essentially glucose-independent.

[0422] It should be noted that although the fusion algorithm of FIG. 39The above description of the EIS shows a single working electrode and the calculation of a reliability index for the activation of the working electrode, but this is only by way of illustration and not limitation. Thus, in a redundant sensor comprising two or more working electrodes, a boundary check can be performed for each of the plurality of (redundant) working electrodes and a start-up reliability index can be calculated. Then, based on the respective reliability indices, at least one working electrode can be identified that can proceed to obtain a glucose measurement. In other words, in a sensor with a single working electrode, if the working electrode exhibits a low start-up, then the actual use of the sensor (for measuring glucose) can have to be delayed until the low start-up period is over. This period can typically be on the order of one hour or more, which is clearly disadvantageous. In contrast, in a redundant sensor, the use of the methods described herein allows for an adaptive or "smart" start-up, in which the electrode that can proceed to data collection can be identified in a considerably shorter order, e.g., within a few minutes. This, in turn, will reduce the MARD, since a low start-up typically increases the MARD by about 1 / 2%.

[0423] In yet another embodiment, EIS can assist in adjusting the calibration buffer. For existing calibration algorithms, the buffer size is always 4, i.e., 4 Isig-BG pairs, and the weights are based on a and β, as previously described, a is an exponential function of the time constant, and β is an exponential function of the blood glucose variance. Here, EIS can help determine when to flush the buffer, how to adjust the buffer weights, and determine the appropriate buffer size.

[0424] In some embodiments, EIS can also be used for interferent detection. In particular, it can be desirable to provide a medication infusion set comprising a combination sensor and medication infusion catheter, wherein the sensor is placed within the infusion catheter. In such a system, the physical location of the infusion catheter relative to the sensor can cause some problems, primarily due to the potential impact (i.e., interference) on the sensor signal that can be caused by the medication being infused and / or its inactive components.

[0425] For example, diluents used with insulin contain meta-cresol as a preservative. In in vitro studies, it has been found that meta-cresol has an adverse effect on glucose sensors if infused in close proximity to the sensor (and thus the meta-cresol). Thus, a system that combines a sensor and an infusion catheter in a single needle must be able to detect and adjust for the impact of meta-cresol on the sensor signal. Since meta-cresol impacts the sensor signal, a way to detect this interferent independently of the sensor signal itself would be preferred.

[0426] Experiments have shown that the effect of meta-cresol on the sensor signal is temporary, and thus reversible. However, when insulin infusion is performed too close to the sensor, meta-cresol tends to "poison" the electrodes, making them no longer able to detect glucose until the insulin (and meta-cresol) has been absorbed into the patient's tissue. In this regard, it has been found that there is typically a period of about 40 minutes between the start of insulin infusion and the sensor regaining the ability to detect glucose. Advantageously, however, it has also been found that, over the same time period, the 1 kHz impedance magnitude increases substantially, completely independently of the glucose concentration.

[0427] In particular, FIG. 40A Isig and impedance data are shown for an in vitro experiment in which the sensor was placed in a glucose solution of 100 mg / dL, and the 1 kHz impedance was measured every 10 minutes, as shown by the circled data points 3920. Meta-cresol was then added to bring the solution to 0.35% meta-cresol (3930). As can be seen, after the addition of meta-cresol, the Isig 3940 initially increased significantly, and then began to drift downward. Then, by adding another 100 mg / dL of glucose, the concentration of glucose in the solution was doubled. However, this had no effect on the Isig 3940, as the electrodes were unable to detect glucose.

[0428] In another aspect, meta-cresol has a significant effect on both the impedance magnitude and phase. FIG. 40B A Bode plot of the phase is shown, and FIG. 40C A Bode plot of the impedance magnitude is shown, before and after the addition of meta-cresol. As can be seen, after the addition of meta-cresol, the impedance magnitude 4010 increased by at least an order of magnitude from its initialized value 4020 across the spectrum. At the same time, the phase 4030 changed completely compared to its initialized value 4040. In FIG. 39 on the Nyquist plot. Here, the pre- initialized curve 4050 and the post- initialized curve 4060 appear as expected for a normally operating sensor. However, after the addition of meta-cresol, the curve 4070 becomes completely different.

[0429] The above experiments identify an important practical deficiency in continuing to rely on the Isig after meta-cresol has been added. Referring back to FIG. 41The patient / user monitoring the sensor signal can be misled into thinking that his glucose level just reached a peak and that he should administer a bolus. The user then administers a bolus, at which point the Isig has already started to drift back down. In other words, everything appears normal to the patient / user. However, in reality, what has actually happened is that the patient has just administered an unnecessary dose of insulin, depending on the patient's glucose level prior to administering the bolus, which can put the patient at risk of experiencing a hypoglycemic event. This situation makes it even more desirable to have a method of detecting interferents that is as glucose independent as possible.

[0430] FIG. 41 Another experiment is shown in which the sensor is initialized to a 100 mg / dL glucose solution, after which the glucose is raised to 400 mg / dL over the course of an hour, and then returned to 100 mg / dL. M-cresol is then added to raise the concentration to 0.35%, and the sensor is left in this solution for 20 minutes. Finally, the sensor is placed in a 100 mg / dL glucose solution to allow the Isig to recover after exposure to m-cresol. As can be seen, after initialization, the 1 kHz impedance magnitude 4110 is about 2 kilo-ohms. When m-cresol is added, the Isig 4120 spikes, as does the impedance magnitude 4110. Furthermore, when the sensor is returned to a 100 mg / dL glucose solution, the impedance magnitude 4110 also recovers to near normal levels.

[0431] As can be seen from the above experiment, EIS can be used to detect the presence of an interfering agent - in this case, m-cresol. Specifically, since the interferent affects the sensor in a way that increases the impedance magnitude across the spectrum, the impedance magnitude can be used to detect the interference. Once the interference is detected, the sensor operating voltage can be changed to a point where the interferent is not measured, or data reporting can be suspended, and the sensor will indicate to the patient / user that the sensor is unable to report data due to the administration of the drug (until the measured impedance recovers to pre-infusion levels). It should be noted that since the effect of the interferent is due to the preservative contained in the insulin, the impedance magnitude will exhibit the same behavior as described above regardless of whether the insulin is infused rapidly or slowly.

[0432] Importantly, as described above, the impedance magnitude, and of course the magnitude at 1 kHz, is essentially glucose independent. Reference is made to FIG. 42AAs can be seen, as the glucose concentration increased from 100 mg / dL to 400 mg / dL - a four-fold increase - the 1 kHz impedance magnitude increased from about 2000 ohms to about 2200 ohms, or about 10%. In other words, the effect of glucose on the impedance magnitude measurement appears to be about an order of magnitude smaller than the measured impedance. This "signal-to-noise" level is typically small enough to allow the noise (i.e., the glucose effect) to be filtered out, such that the resulting impedance magnitude is essentially independent of glucose. In addition, it should be emphasized that the impedance magnitude exhibits even greater glucose independence in actual human tissue, as compared to the buffered solution used in the above-described in vitro experiments.

[0433] Embodiments of the application described herein are also directed to an analog front end integrated circuit (AFE IC), which is a custom application specific integrated circuit (ASIC) that provides the necessary analog electronics to, among other things, provide: (i) support for multiple potentiostats and connection to oxygen or peroxide based multi- electrode glucose sensors; (ii) connection to a microcontroller to form a micro- power sensor system; and (iii) implementation of EIS diagnostics, fusion algorithms, and other EIS based processes based on measurements of EIS based parameters. More specifically, the ASIC incorporates diagnostic capabilities to measure real and imaginary impedance parameters of the sensor over a wide range of frequencies, and also incorporates digital interface circuitry to communicate bi-directionally with a microprocessor chip. In addition, the ASIC contains power control circuitry and a real time clock and crystal oscillator that enable operation at extremely low standby and operating power, such that the power to an external microprocessor can be turned off.

[0434] FIG. 42B and FIG. 42A A block diagram of the ASIC is shown, and Table 1 below provides a pad signal description (shown on the left side of FIG. 42B and FIG. 42A of the ASIC), with some signals being multiplexed onto a single pad.

[0435]

[0436]

[0437] The ASIC will now be described with reference to FIG. 42B and Power plane and Table 1.

[0438] Bias generator

[0439] The ASIC has one power plane that is powered by the supply pad VBAT (4210) that has an operating input range from 2.0 volts to 4.5 volts. This power plane has a regulator for reducing the voltage of some circuits in this plane. The supply is referred to as VDDBU (4212) and has output pads for testing and bypass. The circuits on the VBAT supply include an RC oscillator, real time clock (RC osc) 4214, battery protection circuit, regulator control, power on reset circuit (POR), and various inputs / outputs. The pads on the VBAT power plane are configured to draw less than 75 nA at 40°C and VBAT = 3.50 V.

[0440] The ASIC also has a VDD power supply to supply logic. The VDD power supply voltage range can be programmed from at least 1.6 volts to 2.4 volts. The circuits on the VDD power plane include most of the digital logic, timers (32 kHz), and real time clock (32 kHz). The VDD power plane includes level shifters connected to another voltage plane as needed. The level shifters in turn have interfaces that are regulated so that if another power plane is not powered, the current of any powered power plane does not increase by more than 10 nA.

[0441] The ASIC includes the option of an on-board regulator (with shutdown control) and an external VDD source. The regulator input is a separate pad, REG_VDD_IN (4216), that has electrostatic discharge (ESD) protection like the other I / Os on VBAT. The on-board regulator has an output pad, REG_VDD_OUT (4217). The ASIC also has an input pad for VDD that is separate from the REG_VDD_OUT pad.

[0442] The ASIC includes an analog power plane, referred to as VDDA (4218), that is powered by either the VDD on-board regulator or an external source, and is typically supplied through a filtered VDD. The circuits supplied by VDDA are configured to operate within 0.1 volts of VDD, so that no level shifting between the VDDA and VDD power planes is needed. The VDDA power supplies the sensor analog circuits, analog measurement circuits, and any other noise sensitive circuitry.

[0443] The ASIC includes pad power for the digital interface signals, VPAD. The pad power has an operating voltage range of at least 1.8 V to 3.3 V. These pads have separate power pads and are powered by an external source. The pads also incorporate level shifters to other on-board circuits to allow a flexible pad power range independent of the VDD logic power voltage. The ASIC can regulate the VPAD pad ring signals so that other power current does not increase by more than 10 nA when the VPAD power is not enabled.

[0444] Voltage reference

[0445] The ASIC has a bias generator circuit BIAS_GEN (4220) that is supplied from the VBAT power supply and generates a bias current that is stable when supplying voltage to the system. The output current has the following specifications: (i) power supply sensitivity: 1.6v to 4.5V power supply voltage < ±2.5%; and (ii) current accuracy: < ±3% after trimming.

[0446] The BIAS_GEN circuit generates switched and unswitched output currents to supply circuits that require bias current to operate. The operating current consumption of the BIAS_GEN circuit is less than 0.3uA at 25°C with VBAT from 2.5V to 4.5V (not including any bias output current). Finally, the temperature coefficient of the bias current is typically between 4,000ppm / °C and 6,000ppm / °C.

[0447] 32 kHz oscillator

[0448] As described herein, the ASIC is configured to have a low power voltage reference that is powered by the VBAT power supply. The voltage reference has an enable input that can accept a signal from logic powered by either VBAT or VDDBU. The ASIC is designed such that when VBAT is powered, the enable signal does not cause an increase in current from any power supply of the interface of more than 10nA.

[0449] The reference voltage has the following specifications: (i) output voltage: 1.220 ±3mV after trimming; (ii) power supply sensitivity: 1.6V to 4.5V input < ±6mV; (iii) temperature sensitivity: 0°C to 60°C, < ±5mV; and (iv) output voltage default accuracy (no trimming): 1.220V ±50mV. Additionally, the supply current should be less than 800nA at 4.5V, 40°C. In this embodiment, when the reference is disabled, the reference output will be forced to VSSA to prevent the VDD voltage regulator from overshooting to a level that exceeds the logic breakdown voltage.

[0450] Timer

[0451] The ASIC contains a low power 32.768 kHz crystal oscillator 4222 that is powered from the VDDA supply and can have the capacitance of the crystal oscillator pads (XTALI, XTALO) trimmed by software. Specifically, the frequency trim range is at least -50 ppm to +100 ppm with a step size of at most 2 ppm across the trim range. Here, it can be assumed that the load capacitance on each crystal terminal is 7 pF, Ls = 6.9512 kH, Cs = 3.3952 fF, Rs = 70 k, shunt capacitance = 1 pF, and the PC board parasitic capacitance is 2 pF.

[0452] The ASIC has a VPAD level output available on pad CLK_32kHZ that can be disabled under software and logic control. The default value is to drive the 32 kHz oscillator. The input pin OSC32K_BYPASS (4224) can disable the 32 kHz oscillator (no power consumption) and allow a digital input to the XTALI pad. The circuit associated with this function is configured to not increase any ASIC current beyond 10 nA in either state of the OSC32K_BYPASS signal beyond the oscillator current when OSC32K_BYPASS is low.

[0453] The 32 kHz oscillator is required to be always operational except for the bypass condition when the VDDA plane is powered. If OSC32K_BYPASS is true, then the 32 KHZ oscillator analog circuitry is put into a low power state and the XTALI pad is configured to accept a digital input with a level of 0 to VDDA. Note that the duty cycle of the 32 kHz oscillator output is between 40% and 60%.

[0454] Real time clock (RTC)

[0455] The ASIC contains a timer 4226 that is clocked from the divide-by-2 32 kHz oscillator. The timer is programmable and has two programmable timeouts. The timer has 24 programmable bits for a total time of 17 minutes 4 seconds. The timer also has a programmable delay for disabling the clocking of the CLK_32KHz pad and setting the microprocessor (uP) interface signals on the VPAD plane to a predetermined state (see section below on microprocessor wake-up control signals). This will allow the microprocessor to enter a suspend mode without an external clock. However, this function can be disabled by software with programmable bits.

[0456] The timer also contains a programmable delay for waking up the microprocessor by enabling the CLK_32KHZ clock output and setting UP_WAKEUP high. The POR2 (VDD POR) transition from power low to power good will enable the 32 kHz oscillator, the CLK_32KHZ clock output, and set UP_WAKEUP high. Power off and power up are configured to be controlled by programmable control bits.

[0457] RC oscillator

[0458] The ASIC also has a 48-bit readable / writable binary counter that is operated by a non-gated, free-running 32 kHz oscillator. Writes to the real-time clock 4228 require a write to the address with a key before the clock can be written. Write access to the clock is configured to terminate between 1 and 20 milliseconds after a write to the key address.

[0459] The real-time clock 4228 is configured to be reset to a half count (MSB = 1, all other bits 0) by POR1 IN (VBAT POR) or POR2 IN (VDD POR) on power up reset. In embodiments of the invention, the real-time clock has programmable interrupt capability and is designed to be robust to single event upset (SEU), which can be achieved by layout techniques or by adding capacitance to appropriate nodes as needed.

[0460] Real time RC clock (based on RC oscillator)

[0461] The ASIC further contains an RC clock that is powered by the VBAT supply or a supply derived from VBAT. The RC oscillator is always running, but it can be bypassed by writing a register bit in the analog test mode (see section on digital testing) and applying a signal to GPIO VBAT from 0 to VBAT levels. The RC oscillator is not trimmable and contains the following specifications: (i) a frequency between 750 Hz and 1500 Hz; (ii) a duty cycle between 50% ± 10%; (iii) a current consumption less than 200 nA at 25°C; (iv) a frequency variation less than ± 2% from 1 V to 4.5 V VBAT supply and better than 1% from 1.8 V to 4.5 V VBAT supply; and (v) a frequency variation less than +2, -2% from a temperature of 15°C to 40°C with VBAT = 3.5 V. The RC frequency can be measured using a 32 kHz crystal oscillator or an external frequency source (see oscillator calibration circuit).

[0462] Battery protection circuit

[0463] The ASIC contains a 48-bit readable / writable binary ripple counter based on an RC oscillator. Writes to the RC real-time clock require a write to the address using a key before the clock can be written. After a write to the key address, write access to the clock terminates between 1 millisecond and 20 milliseconds, with the time of the protection window configured to be generated with the RC clock.

[0464] If the crystal oscillator is turned off, the real-time RC clock allows for a relative time stamp and is configured to reset on POR1 IN (BAT POR) to a half count (MSB = 1, all others 0). The real-time RC clock is designed to be robust to single event upset (SEU) through layout techniques or by adding capacitance to the appropriate nodes when needed. On the falling edge of POR2 IN, or if the ASIC enters a battery low state, the RT real-time clock value, which can be read through the SPI port, can be captured into a register. This register and associated logic is on the VBAT or VDDBU power plane.

[0465] Battery power plane power-on reset

[0466] The ASIC contains a battery protection circuit 4230 that monitors the battery voltage using a comparator and is powered by a supply derived from the VBAT power plane. The battery protection circuit is configured to be always running with a supply to the VBAT power plane. The battery protection circuit can use the RC oscillator for a clock signal and has an average current consumption of less than 30 nA, including a 3 mega-ohm total resistance external voltage divider.

[0467] The battery protection circuit uses an external switched voltage divider with a ratio of.421 for a 2.90V battery threshold. The ASIC also has an internal voltage divider with a ratio of.421 ± 0.5%. This voltage divider is connected between BATT_DIV_EN (4232) and VSSA (4234) and the voltage divider output is a pin called BATT_DIV_INT (4236). To save the pin in the package components, BATT_DIV_INT in this embodiment is connected to BATT_DIV internally inside the package. Also in this configuration, BATT_DIV_EN does not need to be taken out of the package, saving two package pins.

[0468] The battery protection circuit is configured to sample the voltage on the input pin BATT_DIV (4238) at a rate of approximately 2 times per second, with the sampling time generated by the RC oscillator. The ASIC is able to adjust the voltage divider of the RC oscillator to adjust the sampling time interval to.500 seconds ± 5 milliseconds, with the RC oscillator operating within its operating tolerance. In a preferred embodiment, the ASIC has a test mode that allows a more frequent sampling interval during testing.

[0469] The comparator input is configured to accept inputs from 0 to VBAT volts. For inputs from 0 to VBAT volts, the input current to the comparator input BATT_DIV is less than 10 nA. The comparator sampling circuit outputs a positive pulse to the pad BATT_DIV_EN which can be used by external circuitry to enable the off-chip resistor divider only during the sampling time to save power. The voltage high logic level is the VBAT voltage and the low level is the VSS level.

[0470] The output resistance of the BATT_DIV_EN pad should be less than 2 kOhms at VBAT = 3.0 V. This allows driving the divider directly from this output. After a programmable number of consecutive samples indicate a low battery condition, the comparator control circuitry triggers an interrupt to the interrupt output pad UP INT. The default number of samples is 4, but the number of consecutive samples can be programmed from 4 to 120.

[0471] After a programmable number of consecutive samples indicate a low battery after the UP INT above has been generated, the comparator control circuitry is configured to generate a signal that will put the ASIC into a low power mode: the VDD regulator will be disabled and a low signal will be asserted to the pad VPAD_EN. This will be called the battery low state. Again, the number of consecutive samples can be programmed from 4 to 120 samples with a default value of 4 samples.

[0472] The comparator has separate programmable thresholds for falling and rising voltages on BATT_DIV. This is implemented in digital logic to multiplex the two values to the circuit depending on the state of the battery low state. So, if the battery low state is low, the falling threshold applies; if the battery low state is high, the rising threshold applies. Specifically, the comparator has 16 programmable thresholds from 1.22 to 1.645 ± 3% with a DNL set to less than 0.2 LSB of the programmable thresholds.

[0473] The comparator thresholds vary less than + / - 1% from 20 °C to 40 °C. The default threshold for the falling voltage is 1.44 V (3.41 V VBAT threshold for a nominal divider) and the default threshold for the rising voltage is 1.53 V (3.63 V VBAT threshold for a nominal divider). After the ASIC enters the battery low state, if the comparator senses 4 consecutive indications of a good battery, the ASIC will start the microprocessor start-up sequence.

[0474] VDD power-on reset (POR)

[0475] A power-on reset (POR) output is generated on pad nPOR1_OUT (4240) if the input VBAT swings more than 1.2 volts in a 50 microsecond period, or the VBAT voltage is below 1.6 ±.3 volts. This POR is extended to a minimum pulse width of 5 milliseconds. The output of the POR circuit is configured to be active low and goes to the pad nPOR1_OUT on the VBAT power plane.

[0476] The IC has an input pad nPOR1_IN (4242) for the battery power plane POR. This input pad has RC filtering so that pulses shorter than 50 nanoseconds will not cause a logic reset. In this embodiment, nPOR1_OUT is externally connected to nPOR1_IN in normal operation, separating the analog circuitry from the digital circuitry for testing. nPOR1_IN causes a reset of all logic on any power plane and initializes all registers to their default values. Thus, the reset status register POR bit is set and all other reset status register bits are cleared. The POR reset circuitry is configured to consume no more than 0.1 uA from the VBAT power supply for more than 5 seconds after power-up.

[0477] Sensor interface electronics

[0478] The ASIC also has a voltage comparator circuit that generates a VDD voltage plane reset signal upon power-up or when VDD falls below a programmable threshold. The range can be programmed through several voltage thresholds. The default value is 1.8V - 15% (1.53V). POR2 has a programmable threshold for the rising voltage that implements a hysteresis. The rising threshold is also programmable with a default value of 1.60V ± 3%.

[0479] The POR signal is active low and has an output pad nPOR2_OUT (4244) on the VDD power plane. The ASIC also has an active low POR open drain output nPOR2_OUT_OD (4246) on the VBAT power plane. This can be used to apply the POR to other system components.

[0480] The VDD powered logic has a POR that originates from an input pad nPOR2_IN (4248). The nPOR2_IN pad is on the VDD power plane and has RC filtering so that pulses shorter than 50 nanoseconds will not cause a logic reset. In normal use cases, nPOR2_OUT is configured to be externally connected to the nPOR2_IN input pad, separating the analog circuitry from the digital circuitry.

[0481] After VDD exceeds the programmable threshold, the resulting reset is extended to an active time of at least 700 milliseconds to ensure crystal oscillator stabilization. The POR reset circuitry consumes no more than 0.1 uA from the VDD power supply for a time exceeding 5 seconds after power-up, and no more than 0.1 uA from the VBAT power supply for a time exceeding 5 seconds after power-up. The register storing the POR threshold is powered by the VDD power plane.

[0482] FIG. 43

[0483] In embodiments of the application described herein, the sensor circuitry supports up to five sensor WORK electrodes (4310) in any combination of peroxide or oxygen sensors, but in other embodiments, a larger number of such electrodes can also be accommodated. When the peroxide sensor WORK electrodes provide current, the oxygen sensor WORK electrodes draw current. For the present embodiment, the sensors can be configured in a potentiostat configuration as shown in FIG. 44

[0484] The sensor electronics has programmable power control for each electrode interface circuit to minimize current consumption by turning off current to unused sensor electronics. The sensor electronics also contains electronics for driving the COUNTER electrode 4320 which uses feedback from the RE (reference) electrode 4330. When this circuitry is not in use, its current can be programmed to be off to conserve power. The interface electronics contains multiplexers 4250 so that the COUNTER and RE electrodes can be connected to any (redundant) WORK electrode.

[0485] The ASIC is configured to provide the following sensor interfaces: (i) RE: a reference electrode which can establish a reference potential for the electronics solution to set the WORK voltage; (ii) WORK1 to WORK5: working sensor electrodes where the required reduction / oxidation (redox) reactions occur; and (iii) COUNTER: an output from this pad maintains a known voltage on the RE electrode relative to system VSS. In this embodiment, the ASIC is configured to be able to set the WORK voltage for up to 5 WORK electrodes individually, with resolution and accuracy better than or equal to 5 mV.

[0486] ​In oxygen mode, the WORK voltage can be programmed between at least 0 and 1.22V with respect to VSSA. In peroxide mode, the WORK voltage can be programmed between at least 0.6 volts to 2.054 volts with respect to VSSA. If VDDA is less than 2.15V, then the WORK voltage can be operated to VDDA - 0.1V. The ASIC includes a current measurement circuit to measure the WORK electrode current in peroxide sensor mode. This can be implemented, for example, by a current-to-voltage or current-to-frequency converter, which can have the following specifications: (i) current range: 0 to 300nA; (ii) voltage output range: same as WORK electrode in peroxide / oxygen mode; (iii) output offset voltage: maximum ±5mV; and (iv) uncalibrated resolution: ±.25nA.

[0487] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0488] 5pA - 1nA: ±3% ±20pA

[0489] 1nA - 10nA: ±3% ±20pA

[0490] 10nA - 300nA: ±3% ±.2nA

[0491] For current-to-frequency converter (ItoF) only, the frequency range can be between 0 Hz and 50 kHz. In peroxide mode, the current converter must operate within the specified voltage range of VSS of the WORK electrode. Here, the current consumption of the 2.5V power supply is less than 2uA, with the WORK electrode current of each converter less than 10nA, including digital-to-analog (DAC) current.

[0492] The current converter can be enabled or disabled by software control. When disabled, the WORK electrode will exhibit an extremely high impedance value, i.e., greater than 100 mega ohms. Additionally, for ItoF only, the output of the I-F converter will go into a 32-bit counter, which can be read, written, and cleared by the microprocessor and test logic. During counter read, the clocking to the counter is suspended to ensure accurate reading.

[0493] In embodiments of the application described herein, the ASIC also contains a current measurement circuit to measure the WORK electrode current in the oxygen sensor mode. The circuit can be implemented as a current-to-voltage or current-to-frequency converter, and programmable bits can be used to configure the current converter to operate in the oxygen mode. As previously described, in the oxygen mode, the current converter must work in the specified voltage range of the WORK electrode relative to VSS. Here again, the current range is 3.7 pA to 300 nA, the voltage output range is the same as the WORK electrode in the oxygen mode, the output offset voltage is a maximum of ±5 mV, and the uncalibrated resolution is 3.7 pA ± 2 pA.

[0494] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0495] 5 pA - 1 nA: ±3% ±20 pA

[0496] 1 nA - 10 nA: ±3% ±20 pA

[0497] 10 nA - 300 nA: ±3% ±.2 nA

[0498] For current-to-frequency converters (ItoF) only, the frequency range can be between 0 Hz and 50 kHz, and the current consumption of the 2.5 V power supply is less than 2 uA, with the WORK electrode current per converter less than 10 nA, including DAC current. The current converter can be enabled or disabled by software control. When disabled, the WORK electrode will exhibit an extremely high impedance value, i.e., greater than 100 mega-ohms. Also, for ItoF only, the output of the I-F converter will go into a 32-bit counter, which can be read, written, and cleared by the microprocessor and test logic. During counter read, the clocking to the counter is suspended to ensure an accurate read.

[0499] In embodiments of the application described herein, the reference electrode (RE) 4330 has an input bias current of less than.05 nA at 40°C. The COUNTER electrode adjusts its output to maintain the desired voltage on the RE electrode. This is accomplished by an amplifier 4340, which to the output of the COUNTER electrode 4320 attempts to minimize the difference between the actual RE electrode voltage and the target RE voltage, which is set by a DAC.

[0500] The RE set voltage is programmable between at least 0 and 1.80V, and the common mode input range of the COUNTER amplifier includes at least.20 to (VDD-.20)V. If necessary, a register bit can be used to select the common mode input range and enable programming of the operating mode of the COUNTER. The WORK voltage is set to be better than or equal to 5mV in resolution and accuracy. It should be noted that in normal mode, the COUNTER voltage seeks to maintain the RE voltage at the level of the programmed RE target value. However, in the force counter mode, the COUNTER voltage is forced to be the programmed RE target voltage.

[0501] All electrode drive circuits are configured to be able to drive the electrodes to the electrode load, and no oscillation occurs in any use case. FIG. 43 An equivalent ac inter-electrode circuit is shown according to an embodiment having a potentiostat configuration as shown in FIG. 44 Current calibrator The equivalent circuit shown can be located between any of the electrodes, WORK1 to WORK5, COUNTER, and RE, with the value ranges of the respective circuit components as follows:

[0502] Ru = [200-5k] ohms

[0503] Cc = [10-2000] pF

[0504] Rpo = [1-20] kOhms

[0505] Rf = [200-2000] kOhms

[0506] Cf = [2-30] uF

[0507] During initialization, the drive current for the WORK electrodes and the COUNTER electrode requires a higher current to be supplied than for the normal potentiostat operation described previously. Therefore, if additional drive is required, then a programmable register bit can be used to program the electrode drive circuit to a higher power state. Low power operation is important in the normal potentiostat mode, where the electrode current is typically less than 300nA.

[0508] ​In a preferred embodiment, during initialization, the WORK1-5 electrodes can be programmed between 0 and VDD volts in steps equal to or less than 5mV, and the drive or sink current output capability of the electrodes is a minimum of 20uA from.20V to (VDD-.20V). Also during initialization, the ASIC is typically configured to measure the current of one WORK electrode up to 20uA with an accuracy of ±2% ±40nA. Further, during initialization, the RE set voltage is programmable as previously described, the COUNTER DRIVE CIRCUIT output must be able to source or sink a minimum of 50uA using the COUNTER electrodes from.20V to (VDD-.20V), and the supply current (VDDA and VDDB) to the initialization circuitry is required to be less than 50uA over any output current provided.

[0509] High speed RC oscillator

[0510] In embodiments of the application, the ASIC has a current reference that can be directed to any WORK electrode for calibration. In this regard, the calibrator contains programmable bits that cause the current output to sink or source current. Assuming a 0 tolerance external precision resistor, the programmable currents contain at least 10nA, 100nA, and 300nA with an accuracy better than ±1% ±1nA. For the reference resistor, the calibrator uses a 1 mega-ohm precision resistor connected to pad TP_RES (4260). Additionally, for initialization and / or sensor state purposes, the current reference can be directed to the COUNTER or RE electrodes. A constant current can be applied to the COUNTER or RE electrodes, and the electrode voltage can be measured using the ADC.

[0511] Analog to digital converter

[0512] Referring back to FIG. 42, the ASIC further contains a high speed RC oscillator 4262 that supplies the analog-to-digital converter (ADC) 4264, the ADC sequencer 4266, and other digital functions that require a speed clock higher than 32kHz. The high speed RC oscillator is phase locked to the 32kHz clock (32.768kHz) to provide an output frequency programmable between 524.3kHz and 1048kHz. Additionally, the high speed RC oscillator has a duty cycle of 50% ±10%, a phase jitter of less than.5% rms, a current of less than 10uA, and a frequency that is stable over the VDD operating range (voltage range of 1.6 to 2.5V). The default value for the high speed RC oscillator is "off" (i.e., disabled), in which case the current consumption is less than 10nA. However, the ASIC has programmable bits to enable the high speed RC oscillator.

[0513] FIG. 42A

[0514] The ASIC contains a 12-bit ADC (4264) with the following characteristics: (i) capable of converting in less than 1.5 milliseconds when running off a 32 kHz clock; (ii) capable of performing faster conversions when clocked from a high speed RC oscillator; (iii) accuracy of at least 10 bits (12 bits ± 4 counts); (iv) reference voltage input of 1.220 V, and temperature sensitivity of less than 0.2 mV / °C between 20°C and 40°C; (v) full scale input range of 0 to 1.22 V, 0 to 1.774 V, 0 to 2.44 V, and 0 to VDDA, with the 1.774 and 2.44 V ranges having programmable bits to reduce the conversion range to lower values to accommodate lower VDDA voltages; (vi) current drawn from power supply of less than 50 uA; (vi) has a converter capable of operating off a 32 kHz clock or a high speed RC clock; (vii) DNL of less than 1 LSB; and (viii) issues an interrupt at the end of conversion.

[0515] As shown in FIGS. 26A and 26B, the ASIC has an analog multiplexer 4268 at the input of the ADC 4264, both the ADC and the multiplexer are controllable by software. In the preferred embodiment, at least the following signals are connected to the multiplexer: Sensor diagnostics 42B (i) VDD - core voltage and regulator output

[0516] (ii) VBAT - battery power

[0517] (iii) VDDA - analog power

[0518] (iv) RE - reference electrode of sensor

[0519] (v) COUNTER - counter electrode of sensor

[0520] (vi) WORK1 to WORK5 - working electrodes of sensor

[0521] (vii) temperature sensor

[0522] (viii) at least two external pin analog signal inputs

[0523] (ix) EIS integrator output

[0524] (x) ItoV current converter output.

[0525] (x) ItoV current converter output.

[0526] ​The ASIC is configured such that the load on the ADC for the inputs COUNTER, RE, WORK1 to WORK5, temperature sensor, and any other inputs that can be adversely affected by load, does not exceed ±0.01 nA. The multiplexer includes a voltage divider for any input with a voltage above the ADC input voltage range, and a buffer amplifier that reduces the input resistance of the divided input to less than 1 nA for load sensitive inputs. The buffer amplifier in turn has a common mode input range of at least 0.8V to VDDA voltage, and an offset of less than 3mV between the input range of 0.8V to VDDA-.1V.

[0527] In a preferred embodiment, the ASIC has a mode to make ADC measurements in a programmed sequence. Thus, the ASIC includes a programmable sequencer 4266 that supervises the measurement of up to 8 input sources for ADC measurements with the following programmable parameters:

[0528] (i) ADC MUX input

[0529] (ii) ADC range

[0530] (iii) Delay time before measurement, where the delay can be programmed from 0 to 62 milliseconds in steps of.488 milliseconds

[0531] (iv) Number of measurements per input from 0 to 255

[0532] (v) Number of measurement loops: 0 to 255, where a measurement loop refers to repeating a sequence of up to 8 input measurements (e.g., as an outer loop in a program)

[0533] (vi) Delay between measurement loops, where the delay can be programmed from 0 to 62 milliseconds in steps of.488 milliseconds.

[0534] The sequencer 4266 is configured to start upon receiving an automatic measurement start command, and the measurement values can be stored in the ASIC for retrieval through the SPI interface. It should be noted that the sequencer time base can be programmed between a 32 kHz clock and a high speed RC oscillator 4262.

[0535] FIG. 42B

[0536] As previously described in detail, embodiments of the application described herein are directed to the use of impedance and impedance-related parameters, for example, in sensor diagnostic procedures and Isig / SG fusion algorithms. To this end, in preferred embodiments, the ASIC described herein is capable of measuring the impedance magnitude and phase angle of any WORK sensor electrode relative to the RE and COUNTER electrodes when in the potentiostat configuration. This is done, for example, by measuring the amplitude and phase of the current waveform in response to a sinusoidal waveform superimposed on the WORK electrode voltage. See, for example, the diagnostic circuitry 4255 in Calibration voltage

[0537] The ASIC is capable of measuring the resistance and capacitance components of any electrode to any electrode through, for example, the electrode multiplexer 4250. It should be noted that such measurements can disturb the sensor balance and can require a settling time or sensor initialization to record a stable electrode current. As previously described, although the ASIC can be used for impedance measurements over a wide spectrum of frequencies, for the purposes of embodiments of the application, a relatively narrow range of frequencies can be used. Specifically, the sinusoidal wave measurement capability of the ASIC can include test frequencies from about 0.10 Hz to about 8192 Hz. When making such measurements, the minimum frequency resolution according to embodiments of the application can be limited as shown in Table 2 below:

[0538] Table 2

[0539]

[0540] The sinusoidal wave amplitude can be programmed from at least 10 mVp-p to 50 mVp-p in at least 5 mV steps and from 60 mVp-p to 100 mVp-p in at least 10 mV steps. In preferred embodiments, the amplitude accuracy is better than ±5% or ±5 mV, whichever is greater. Additionally, the ASIC can measure electrode impedance with the accuracy specified in Table 3 below:

[0541]

[0542]

[0543] In embodiments of the application, the ASIC can measure the input waveform phase relative to a time base, which can be used for impedance calculations to improve accuracy. The ASIC can also have an on-chip resistor to calibrate the electrode impedance circuit described above. The on-chip resistor, in turn, can be calibrated by comparison to a known 1 mega-ohm off-chip precision resistor.

[0544] ​The waveform data samples can also be used to determine impedance. The data can be transmitted to an external microprocessor using a serial peripheral interface (SPI) for computation and processing. The converted current data is sufficiently buffered to enable 2000 ADC data conversions to be transferred to an external device through the SPI interface without loss of data. This assumes a maximum latency time of 8 milliseconds for servicing data transfer request interrupts.

[0545] In embodiments of the application, instead of or in addition to measuring electrode impedance with a sine wave, the ASIC can also measure electrode current with a step input. Here, the ASIC can provide a programmable amplitude step of 10 to 200 mV to the electrode at a resolution better than 5 mV and sample (measure) the resulting current waveform. The duration of the sample can be programmed in.25 second steps for at least 2 seconds, and the sample interval for measuring current can include at least five programmable binary weighted steps of approximately.5 milliseconds to 8 milliseconds.

[0546] The resolution of the electrode voltage samples is less than 1 mV, with a range of up to ±.25 volts. This measurement can be made with respect to a suitable stable voltage in order to reduce the required data conversion dynamic range. Similarly, the resolution of the electrode current samples is less than.04 uA, with a range of up to 20 uA. If the measurement polarity is programmable, then the current measurement can be unipolar.

[0547] In embodiments of the application, the current measurement can use an I-V converter. In addition, the ASIC can have an on-chip resistor to calibrate the current measurement. The on-chip resistor can in turn be calibrated by comparison to a known 1 mega-ohm off-chip precision resistor. The current measurement sample accuracy is better than ± 3% or ± 10 nA, whichever is greater. As previously mentioned, the converted current data is sufficiently buffered to enable 2000 ADC data conversions to be transferred to an external device through the SPI interface without loss of data. This assumes a maximum latency time of 8 milliseconds for servicing data transfer request interrupts.

[0548] Temperature sensor

[0549] The ASIC includes a precision voltage reference for calibrating the ADC. The output voltage is 1.000 V ± 3%, with a production variation of less than ± 1.5%, and a stability better than ± 3 mV over a temperature range of 20°C to 40°C. This precision calibration voltage can be calibrated by the on-chip ADC by comparison to an external precision voltage during manufacturing. In manufacturing, the calibration factor can be implemented in system non-volatile memory (not on this ASIC) for higher accuracy.

[0550] The current consumption of the calibration voltage circuit is preferably less than 25uA. Furthermore, the calibration voltage circuit can be powered down to less than 10nA when not in use to conserve battery power.

[0551] VDD voltage regulator

[0552] The ASIC has a temperature transducer with a sensitivity of between 9 and 11 mV per degree Celsius between -10°C and 60°C. The output voltage of the temperature sensor is such that the ADC can measure the temperature dependent voltage with a 0 to 1.22V ADC input range. The current consumption of the temperature sensor is preferably less than 25uA and the temperature sensor can be powered down to less than 10nA when not in use to conserve battery power.

[0553] General purpose comparator

[0554] The ASIC has a VDD voltage regulator with the following characteristics:

[0555] (i) Minimum input voltage range: 2.0V to 4.5V.

[0556] (ii) Minimum output voltage: 1.6 to 2.5V ± 5%, default value 2.0V.

[0557] (iii) Differential voltage: Vin-Vout <.15V at Iload = 100uA, Vin = 2.0V.

[0558] (iv) The output voltage is programmable with an accuracy of within 2% of the values shown in Table 4 below:

[0559] Table 4

[0560]

[0561]

[0562] (v) The regulator can supply 1mA output at 2.5V with an input voltage of 2.8V.

[0563] (vi) The regulator also has input and output pads which can be left open if an external regulator is used. In this inoperable mode, the current consumption of the regulator circuit is preferably less than 100nA.

[0564] (vii) The variation in output voltage from 10uA load to 1mA load is preferably less than 25mV.

[0565] (viii) The current consumption of the source excluding the output current at 1mA load is less than 100uA.

[0566] (ix) The current draw of the source not including the output current under a 0.1 mA load is less than 10 uA.

[0567] (x) The current draw of the source not including the output current under a 10 uA load is less than 1 uA.

[0568] Sensor connected sensing circuitry on RE

[0569] The ASIC contains at least two comparators 4270, 4271 powered by VDDA. The comparators use 1.22 V as a reference to produce a threshold. The output of the comparators can be read by the processor and will produce a maskable interrupt on the rising or falling edge determined by configuration registers.

[0570] The comparators have power control to reduce power when not in use, and the current supply of each comparator is less than 50 nA. The response time of the comparators is preferably less than 50 microseconds for a 20 mV overdrive signal, and the offset voltage is less than ±8 mV.

[0571] The comparators also have programmable hysteresis, where the hysteresis options include: threshold on rising input = 1.22 V + Vhyst, threshold on falling input = 1.22 - Vhyst, or no hysteresis (Vhyst = 25 ± 10 mV). The output from either comparator can be used on any GPIO on any power plane. (See the "GPIO" section).

[0572] WAKEUP pad

[0573] The analog switched capacitor circuit monitors the impedance of the RE connection to determine if a sensor has been connected. Specifically, a capacitor of about 20 pF is switched at a frequency of 16 Hz driven by an inverter with an output swing of VSS to VDD. A comparator will sense the voltage swing on the RE pad, and if the swing is less than a threshold, the comparator output will indicate a connection. The above comparison is made on both transitions of the pulse. A swing below threshold on both transitions is required to indicate a connection, and a comparison indicating a high swing on either phase will indicate a disconnection. The connection / disconnection signal is debounced such that a transition in state requires at least 1 1 / 2 second of stabilization in the new state indication.

[0574] The circuit has six thresholds defined by the following resistors in parallel with a 20 pF capacitor: 500 kOhms, 1 MOhms, 2 MOhms, 4 MOhms, 8 MOhms, and 16 MOhms. This parallel equivalent circuit is between the RE pad and a virtual ground, which can be at any voltage between the power rails. The threshold accuracy is better than ±30%.

[0575] If a sensor is connected or disconnected, the output of the sensor connection sensing circuitry can programmably generate an interrupt or processor wake-up. This circuit is active whenever nPOR2_IN is high and VDD and VDDA are present. The average current consumption of this circuit is less than 100 nA.

[0576] UART WAKEUP

[0577] The WAKEUP circuitry is powered by the VDD supply with an input range of 0V to VBAT. The WAKEUP pad 4272 has a weak pull-down of 80 ± 40 nA. This current can be sourced from the output of BIAS_GEN 4220. With a 0v input, the average current consumed by the circuit is less than 50 nA.

[0578] The rising input voltage threshold Vih for the WAKEUP input is 1.22 ± 0.1V and the falling input threshold is -25mV ± 12mV of the rising threshold. In a preferred embodiment, the circuitry associated with the WAKEUP input draws no more than 100 nA of current for any input value between -.2 and VBAT voltage (this current does not include the input pull-down current). The WAKEUP pad is debounced for at least 1 / 2 second.

[0579] The output of the WAKEUP circuit can programmably generate an interrupt or processor wake-up if the WAKEUP pad changes state. (See the Event Handler section). It is important to note that if the battery protection circuit indicates a low battery condition, the WAKEUP pad circuitry is configured to assume a low current of <1 nA.

[0580] Microprocessor wake-up control signal

[0581] The ASIC is configured to monitor the nRX_EXT pad 4274. If the nRX_EXT level is continuously high (UART BREAK) for more than 1 / 2 seconds, a UART WAKEUP event will be generated. Due to sampling, the UART WAKEUP event can be generated with as little as 1 / 4 seconds of consecutive high. The UART WAKEUP event can programmably generate an interrupt, WAKEUP, and / or microprocessor reset (nRESET OD). (See the Event Handler section).

[0582] In a preferred embodiment, the circuit associated with the UART WAKEUP input draws no more than 100 nA, and if the battery protection circuitry indicates a low battery condition, then the UART WAKEUP pad circuitry is configured to assume a low current of <1 nA. The UART wake-up input has a rising input voltage threshold Vih of 1.22 ± 0.1 V. The falling input threshold is -25 mV ± 12 mV of the rising threshold.

[0583] Event handler / watchdog

[0584] The ASIC is capable of generating signals to help control the power management of the microprocessor. Specifically, the ASIC can generate the following signals:

[0585] (i) nSHUTDN - nSHUTDN can control the power enable of an off-chip VDD regulator. The nSHUTDN pad is on the VBAT power rail. If the battery protection circuitry indicates a low battery condition, then nSHUTDN is low, otherwise nSHUTDN is high.

[0586] (ii) VPAD_EN - VPAD_EN can control the power enable of an external regulator that supplies the VPAD power. An internal signal corresponding to this external signal ensures that when the VPAD power is disabled, the input from the VPAD pad does not generate additional current due to a floating input. The VPAD_EN pad is an output on the VBAT power rail. If the battery protection signal indicates a low battery, then the VPAD_EN signal is low. The VPAD_EN signal can be set low by a software command that starts a timer; the terminal count of the timer forces VPAD_EN low. If the battery protection signal indicates a good battery, then the following events can make the VPAD_EN signal go high (see event handler for more details): nPOR2_IN transitions from low to high; SW / timer (programmable); WAKEUP transitions; low to high and / or high to low (programmable); sensor connection transitions; low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable).

[0587] (iii) UP_WAKEUP - The UP_WAKEUP can be connected to the microprocessor wake up pad. Intended to wake the microprocessor from sleep mode or similar power down mode. The UP_WAKEUP pad is an output on the VPAD power rail. The UP_WAKEUP signal can be programmed to be active low, active high, or pulsed. The UP_WAKEUP signal can be set low by a software command to start the timer; the terminal count of the timer forces UP_WAKEUP low. If the battery protection signal indicates good battery level, the following events can make the UP_WAKEUP signal go high (see event handler for more details): nPOR2_IN transitions from low to high; SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable). The WAKEUP signal can be delayed by a programmable amount. If WAKEUP is programmed to be pulsed, the pulse width can be programmed.

[0588] (iv) CLK_32KHZ - The CLK_32KHZ pad can be connected to the microprocessor to supply a low speed clock. The clock is open- closed programmable and turned on programmatically for wake up events. The CLK_32KHZ pad is an output on the VPAD power rail. The CLK_32KHZ signal is low if the battery protection signal indicates low battery level. The CLK_32KHZ output can be programmed off by a programmable bit. Default is on. The CLK_32KHZ signal can be disabled by a software command to start the timer; the terminal count of the timer forces CLK_32KHZ low. If the battery protection signal indicates good battery level, the following events can enable the CLK_32KHZ signal (see event handler for more details): nPOR2_IN transitions from low to high; SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; RTC time event (programmable); and detection of low battery level by the battery protection circuit.

[0589] (v) nRESET OD - nRESET OD can be connected to the microprocessor to cause the microprocessor to reset. nRESET OD is programmable to wake-up events. The nRESET OD pad is an output on the VPAD power rail. This pad is open-drain (nfet output). The nRESET OD signal is low if the battery protection signal indicates low battery. The nRESET OD active time can be programmed from 1 to 200 milliseconds. The default value is 200 ms. The following events can cause the nRESET OD signal to be asserted low (see event handler for more details): nPOR2 IN; SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; and RTC time event (programmable).

[0590] (vi) UP INT - UP INT can be connected to the microprocessor to communicate an interrupt. UP INT is programmable to wake-up events. The UP INT pad is an output on the VPAD power rail. The UP INT signal is low if the battery protection signal indicates low battery. The UP INT signal can be set high by a software command that starts a timer; the terminal count of the timer forces UP INT high. If the battery protection signal indicates good battery, the following events can cause the UP INT signal to be asserted high (see event handler for more details): SW / timer (programmable); WAKEUP transition; low to high and / or high to low (programmable); sensor connection transition; low to high and / or high to low (programmable); UART interrupt; RTC time event (programmable); detection of low battery by the battery protection circuit; and any ASIC interrupt when not masked.

[0591] The ASIC has GPIO1 and GPIO0 pads that can be used as boot mode controls for the microprocessor. The POR2 event will reset a 2-bit counter whose bits are mapped to GPIO1 and GPIO0 (MSB and LSB, respectively). The rising edge of a UART interrupt increments the counter by one, where the counter counts modulo 4 and becomes zero if it was incremented in state 11. The boot mode counter is presettable by SPI.

[0592] Digital to analog (D / A)

[0593] The ASIC incorporates an event handler to define the response to events, which include changes in system state and input signals. Events include all interrupt sources (e.g., UART BRK, WAKE UP, sensor connection, etc.). The software can program the response of the event handler to stimuli through the SPI interface. However, certain responses can be hardwired (not programmable).

[0594] The actions of the event handler include enabling / disabling VPAD_EN, enabling / disabling CLK_32KHZ, asserting nRESET_OD, asserting UP_WAKEUP, and asserting UP_INT. The event watchdog timers 1 through 5 can be individually programmed from 250 milliseconds to 16,384 seconds in 250 millisecond increments. The timeouts of the event watchdog timers 6 through 8 are hard coded. The timeout of Timer 6 and Timer 7 is 1 minute; the timeout of Timer 8 is 5 minutes.

[0595] The ASIC also has a watchdog function to monitor the microprocessor's response upon an event trigger. The event watchdog will be activated when the microprocessor fails to acknowledge the event initiated activity. Once activated, the event watchdog performs a series of programmable actions, i.e., event watchdog timers 1 through 5, and then a series of hardwired actions, i.e., event watchdog timers 6 through 8, to regain the microprocessor's response. The series of actions include interrupts, resets, wakeups, assertion of 32KHz clock, power down and power up of the processor.

[0596] During the series of actions, if the microprocessor regains its ability to acknowledge the recorded activity, the event watchdog resets. If the ASIC fails to get an acknowledgement from the microprocessor, the event watchdog powers down the microprocessor under the condition that will allow the UART_BRK to restart the microprocessor and will activate the alarm. Upon activation, the alarm condition generates a square wave on pad ALARM in a programmable repeating pattern with a frequency of approximately 1 kHz. The programmable pattern has two programmable sequences with programmable burst on and off times. The alarm has another programmable pattern that can be programmed through the SPI port. The pattern will have two programmable sequences with programmable burst on and off times.

[0597] Charger / data downloader interface

[0598] In the preferred embodiment, the ASIC has two 8-bit D / A converters 4276, 4278 with the following characteristics:

[0599] (i) The D / As are stable with less than 50 pF load in less than 1 millisecond.

[0600] (ii) The D / As have at least 8-bit accuracy.

[0601] (iii) The output range is programmable from 0 to 1.22 V or 0 to VDDA.

[0602] (iv) The temperature sensitivity of the D / A voltage reference is less than 1 mV / °C

[0603] (v) DNL is less than 1 LSB.

[0604] (vi) D / A draws less than 2 uA from the VDDA supply.

[0605] (vii) Each D / A has an output I to a pad.

[0606] (viii) D / A outputs are high impedance. Load current must be less than 1 nA.

[0607] (ix) D / A pads can be programmed to output a digital signal from a register. Output swing is VSSA to VDDA.

[0608] Sensor connection switch

[0609] TX_EXT_OD 4280 is an open drain output whose input is the signal on the TX_UP input pad. This will allow the TX_EXT_OD pad to turn on during UART idle conditions. The TX_EXT_OD pad has a comparator that monitors its voltage. If the voltage is above the comparator threshold voltage for a de-bounce period (1 / 4 second), then the output nBAT_CHRG_EN (4281) will go low. This comparator and other associated circuitry with this function are on the VBAT and / or VDDBU planes.

[0610] The circuitry associated with this function must allow a low on the TX_EXT_OD pad due to normal communication with an external device without de-asserting nBAT_CHRG_EN. If POR1 is active, then nBAT_CHRG_EN will be high (not asserted). The threshold voltage of the comparator is between.50V and 1.2V. The comparator will have hysteresis; the falling threshold is about 25mV lower than the rising threshold.

[0611] The nRX_EXT pad inverts the signal on this pad and outputs it to RX_UP. In this way, the nRX_EXT signal will idle low. The nRX_EXT must accept an input of up to the VBAT voltage. The nRX_EXT threshold is 1.22V ± 3%. The output of this comparator will be available over the SPI bus for the microprocessor to read.

[0612] The nRX_EXT pad also incorporates a way to programmably source a current that will be 80 ± 30 nA with a maximum voltage of VBAT. The ASIC layout has shield programmable options to adjust this current from 30 nA to 200 nA in steps of less than 50 nA with a minimum number of shield layer changes. A programmable bit will be available to block UART interrupt detection and force RX_UP high. In normal operation this bit will be set high before enabling the current source to nRX_EXT and then set low after disabling the current source to ensure that no glitch is generated on RX_UP or to ensure that a UART interrupt event is generated. Note that a wet connector detector is implemented, the RX comparator output indicating a low input voltage will indicate a leakage current when the current source to nRX_EXT is active. The ASIC contains a pull down resistor of approximately 100 kOhm on the nRX_EXT pad. This pull down will be disconnected when the current source is active.

[0613] Oscillator calibration circuit

[0614] The ASIC shall have a pad SEN_CONN_SW (4282) that is capable of detecting low resistance to VSS (4284). SEN_CONN_SW provides a current of 5 to 25 uA with a maximum open circuit voltage of.4V with SEN_CONN_SW = 0V. The ASIC layout has shield programmable options to adjust this current from 1 uA to 20 uA in steps of less than 5 uA with a minimum number of shield layer changes. SEN_CONN_SW has associated circuitry that can detect the presence of a resistance between SEN_CONN_SW and VSSA (4234) with a threshold of between 2 to 15 kOhm. The average current consumption of this circuit is a maximum of 50 nA. This low current must be implemented using sampling.

[0615] Oscillator bypass

[0616] The ASIC has counters whose inputs can be directed to internal or external clock sources. One counter generates a programmable gated interval for another counter. The gated interval contains 1 to 15 seconds from a 32 kHz oscillator. The clocks that can be directed to either counter are a 32 kHz RC oscillator, a high speed RC oscillator, and an input from any GPIO pad.

[0617] SPI slave port

[0618] The ASIC can have an external clock replace each output of the oscillator. The ASIC has a register that can only be written to when a particular TEST_MODE is asserted. This register has bits that enable the external input of the RC oscillator and can be shared with other analog test control signals. However, if the TEST_MODE is not active, then this register will not allow any oscillator bypass bits to be active.

[0619] The ASIC also has an input pad for an external clock to bypass the RC oscillator. The pad GPIO_VBAT is on the VBAT power plane. The ASIC further includes a bypass enable pad OSC32K_BYPASS for the 32KHZ oscillator. When high, the 32KHZ oscillator output is supplied by driving the OSC32KHZ_IN pad. Note that typically, the OSC32KHZ_IN pad is connected to a crystal.

[0620] The ASIC has an input for an external clock to bypass the HS_RC_OSC. The bypass is enabled by a programmable register bit. The HS_RC_OSC can be programmably supplied by a GPIO on the VDD plane or a GPIO on the VPAD plane.

[0621] Microprocessor interrupt

[0622] The SPI slave port includes an interface consisting of a chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a serial data input (SPI_MOSI) 4287, and a serial data output (SPI_MISO) 4288. The chip select input (SPI_nCS) is a low active input, asserted by an off-chip SPI master device to initiate and scope an SPI transaction. When SPI_nCS is asserted low, the SPI slave port configures itself as an SPI slave device and performs data transactions based on the clock input (SPI_CK). When SPI_nCS is inactive, the SPI slave port resets itself and remains in reset mode. Since this SPI interface supports block transfers, the master device should keep SPI_nCS low until the transfer is complete.

[0623] The SPI clock input (SPI_CK) will always be asserted by the SPI master device. The SPI slave port latches incoming data on the SPI_MOSI input using the rising edge of SPI_CK and drives outgoing data on the SPI_MISO output using the falling edge of SPI_CK. The serial data input (SPI_MOSI) is used to transfer data from the SPI master device to the SPI slave device. All data bits are asserted after the falling edge of SPI_CK. The serial data output (SPI_MISO) is used to transfer data from the SPI slave device to the SPI master device. All data bits are asserted after the falling edge of SPI_CK.

[0624] SPI_nCS, SPI_CK, and SPI_MOSI are always driven by the SPI master device, unless the SPI master device is powered down. If VPAD_EN is low, these inputs are conditioned such that the current consumption associated with these inputs is less than 10 nA, and the SPI circuitry remains in a reset or inactive state. SPI_MISO is only driven by the SPI slave port when SPI_nCS is active, otherwise SPI_MISO is tri-stated.

[0625] The chip select (SPI_nCS) defines and constitutes a data transfer packet for an SPI data transaction. The data transfer packet consists of three parts. There is a 4-bit command portion, followed by a 12-bit address portion, and then any number of 8-bit data bytes. Command bit 3 is used as a direction bit. "1" indicates a write operation, and "0" indicates a read operation. The combination of command bits 2, 1, 0 has the following definitions. The unused combinations are undefined.

[0626] (i) 0000: Read data and increment address.

[0627] (ii) 0001: Read data, address unchanged

[0628] (iii) 0010: Read data, decrement address

[0629] (iv) 1000: Write data and increment address

[0630] (v) 1001: Write data, address unchanged

[0631] (vi) 1010: Write data, decrement address

[0632] (vii) x011: Test port addressing

[0633] The 12-bit address portion defines the starting byte address. If SPI_nCS remains active after the first data byte to indicate a multi-byte transfer, the address will be incremented by one after each byte is transferred. Bit <11> of the address (address<11:0>) indicates the most significant address bit. The address wraps after reaching the limit.

[0634] Data is in byte format and block transfers can be performed by extending SPI_nCS to allow all bytes to be transferred in one data packet.

[0635] General purpose input / output (GPIO) / parallel test port

[0636] The ASIC has an output at the VPAD logic level UP_INT for sending an interrupt to the host microprocessor. The microprocessor interrupt module consists of an interrupt status register, an interrupt mask register, and a function to logically OR all interrupt status into a microprocessor interrupt. The interrupt is implemented to support edge sensitive and level sensitive styles. The polarity of the interrupt is programmable. The default interrupt polarity is TBD.

[0637] In the preferred embodiment, all interrupt sources on the AFE ASIC will be recorded in the interrupt status register. Writing a "1" to the corresponding interrupt status bit will clear the corresponding pending interrupt. All interrupt sources on the AFE ASIC can be masked by the interrupt mask register. Writing a "1" to the corresponding interrupt mask bit enables masking of the corresponding pending interrupt. Writing a "0" to the corresponding interrupt mask bit disables masking of the corresponding interrupt. The default state of the interrupt mask register is TBD.

[0638] Analog test port

[0639] In an embodiment, the ASIC can have eight GPIOs that operate at the VPAD level signal. The ASIC has one GPIO that operates at the VBAT level signal and one GPIO that operates at the VDD level signal. All GPIOs have at least the following characteristics:

[0640] (i) Register bits control the selection and direction of each GPIO.

[0641] (ii) The ASIC is capable of configuring a GPIO as an input that can be read through the SPI interface.

[0642] (iii) The ASIC is capable of configuring a GPIO as an input for generating an interrupt.

[0643] (iv) The ASIC is capable of configuring each GPIO as an output, controlled by register bits, that can be written through the SPI interface.

[0644] (v) The ASIC is able to programmably output the input signal applied to GPIO_VBAT or GPIO_VDD to a GPIO (on the VPAD power plane). (Level shift function).

[0645] (vi) The ASIC is able to configure each GPIO as an input to the oscillator calibration circuit.

[0646] (vii) The ASIC is able to configure each general purpose comparator output as at least one GPIO on each power plane. The polarity of the comparator output is programmable by a programmable bit.

[0647] (viii) The GPIO has microprocessor interrupt generation capability.

[0648] (ix) The GPIO is programmable as an open drain output.

[0649] (x) The GPIO on the VPAD power plane is configurable to implement the microprocessor's start-up control.

[0650] The parallel test port shares an 8-bit GPIO on the VPAD voltage plane. The test port will be used to observe register contents and various internal signals. In normal mode, the output of this port is controlled by the port configuration register. Writing 8'hFF into the GPIO_O1S_REG and GPIO_O2S_REG registers will direct the test port data on the GPIO output, while writing 8'h00 into the GPIO_ON_REG register will disable the test port data and enable the GPIO data to the GPIO output.

[0651] The registers and pre-packaged internal signals can be observed through this test port by addressing the target register from the port via SPI. The command bit of the SPI data packet is set to 4'b0011, followed by a 12-bit target register address. The parallel test port continues to display the contents of the addressed register until the next test port addressing command is received.

[0652] Chip ID

[0653] The IC has a multiplexer feeding pad TP_ANAMUX (4290) that will provide visibility to an internal analog circuit node for testing. The IC also has a multiplexer feeding pad TP_RES (4260) that will provide visibility to an internal analog circuit node for testing. In a common application, this pad will also accommodate a 1 meg precision resistor to perform various system calibrations.

[0654] Standby test output

[0655] The ASIC contains a 32-bit mask programmable ID. A microprocessor using the SPI interface will be able to read this ID. This ID will be placed in the analog electronic block so that changing the ID does not require a chip reflow. The design should be such that changing the ID requires only one metal or one contact cap.

[0656] Digital test

[0657] The ASIC has 16 spare digital output signals that can be multiplexed to 8-bit GPIOs according to commands sent through the SPI interface. These signals will be organized in two 8-bit bytes and will be connected to VSS when not in use.

[0658] Leakage test pin

[0659] The ASIC has a test mode controller using two input pins TEST_CTL0 (4291) and TEST_CTL1 (4292). The test controller generates a signal according to the combination of test control signals with the following functionality (TEST_CTL<1:0>):

[0660] (i) 0 is the normal operating mode;

[0661] (ii) 1 is the analog test mode;

[0662] (iii) 2 is the scan mode;

[0663] (iv) 3 is the analog test mode where VDD_EN is controlled by an input to GPIO_VBAT.

[0664] The test controller logic is split between the VDD and VDDBU power planes. During the scan mode, the test LT_VBAT should be asserted high to adjust the analog outputs to digital logic. The ASIC has scan chains implemented in as much digital logic as possible for fast digital testing.

[0665] Power requirements

[0666] The ASIC has a pin called LT_VBAT that when high, puts all analog blocks into an inactive mode so that only leakage current will be drawn from the power supply. LT_VBAT puts all digital outputs from the analog blocks in a stable high or low state so as not to affect the interface logic current consumption. The LT_VBAT pad is on the VBAT plane with a pull-down resistance between 10 and 40 kOhms.

[0667] Environment

[0668] In embodiments of the application, the ASIC contains a low power mode in which at a minimum, the microprocessor clock is off, the 32 kHz real time clock is running, and the circuitry is active to detect a sensor connection, a level change on the WAKE UP pin, or a BREAK on the nRX_EXT input. This mode has a total current consumption of VBAT (VDDBU), VDD, and VDDA to a maximum of 4.0 uA. When the battery protection circuit detects a low battery (see battery protection circuit description), the ASIC enters a mode in which only the VBAT and VDDBU power planes are active. This is called the low battery state. The VBAT current in this mode is less than.3 uA.

[0669] In the case where the ASIC is programmed in the potentiostat configuration in which any one of the WORK electrodes is active in the H2O2 (peroxide) mode and its voltage is set to 1.535 V, the COUNTER amplifier is on and VSET_RE is set to 1.00 V, the 20 MEG load resistor is connected between WORK and COUNTER, and COUNTER and RE are connected together, and assuming the WORK electrode current is measured once per minute, the average current consumption of all power supplies is less than 7 uA. The current measured after calibration should be 26.75 nA ± 3%. The addition of another WORK electrode can increase the combined current consumption by less than 2 uA in the case where the WORK electrode current is 25 nA.

[0670] In the case where the ASIC is programmed in the potentiostat configuration in which the diagnostic function is enabled to measure the impedance of one of the WORK electrodes relative to the COUNTER electrode, the ASIC is configured to meet the following requirements:

[0671] (i) Test frequencies: 0.1, 0.2, 0.3, 0.5 Hz, 1.0, 2.0, 5.0, 10, 100, 1000, and 4000 Hz.

[0672] (ii) The measurement at the above frequencies should not exceed 50 seconds.

[0673] (iii) The total charge supplied to the ASIC is less than 8 milliCoulombs.

[0674] FIG. 45

[0675] In preferred embodiments of the application, the ASIC:

[0676] (i) Operates over a commercial temperature range of 0 to 70 °C and meets all specifications.

[0677] (ii) Functionally operates over a temperature range of -20 °C to 80 °C, but doing so can degrade accuracy.

[0678] (iii) expected to operate after storage in the temperature range of -30 to 80°C.

[0679] (iv) expected to operate in the relative humidity range of 1% to 95%.

[0680] (v) ESD protection greater than ±2KV when the human body model package is encapsulated in TBD package unless otherwise specified.

[0681] (vi) configured such that WORK1 to WORK5, COUNTER, RE, TX_EXT_OD, and nRX_EXT pads withstand greater than ±4KV of human body model.

[0682] (vii) configured such that the leakage current of WORK1 to WORK5 and RE pads is less than.05 nA at 40°C.

[0683] In embodiments of the application, the ASIC can be fabricated in a.25 micron CMOS process, and the backup data for the ASIC is on a DVD disk 916-TBD.

[0684] As described in detail above, the ASIC provides the necessary analog electronics to, among other things, provide the following functions: (i) support multiple potentiostats and connect to multi- electrode glucose sensors based on oxygen or peroxide; (ii) connect to a microcontroller to form a micro-power sensor system; and (iii) implement EIS diagnostics based on measurements of EIS-based parameters. Measurement and calculation of EIS-based parameters will now be described in accordance with embodiments of the application.

[0685] As previously described, impedance at frequencies in the range of 0.1 Hz to 8 kHz can provide information about the state of the sensor electrodes. The AFE IC circuitry incorporates circuitry for generating the measurement forcing signals and for making the measurements used to calculate the impedance. Design considerations for this circuitry include current consumption, accuracy, measurement speed, required processing, and on-time required to control the microprocessor.

[0686] In a preferred embodiment of the application, the technique used by the AFE IC to measure electrode impedance is to superimpose a sinusoidal wave voltage on the dc voltage driving the electrode and measure the resulting AC current phase and amplitude. To generate the sinusoidal wave, the AFE IC incorporates a digitally synthesized sinusoidal wave current. This digital technique is used because the frequency and phase can be controlled precisely by a crystal derived time base and frequencies from DC to 8 kHz can be easily generated. The sinusoidal current is added to a resistor in series with the voltage source in order to add an AC component to the electrode voltage. This voltage is the AC forcing voltage. The voltage is then buffered by an amplifier that drives the selected sensor electrode.

[0687] The current driving the electrode contains the resulting AC current component from the for...

Claims

1. A glucose monitoring system, comprising: A glucose sensor device for determining the glucose concentration in a user's body during a total sensor device wearing time, the total sensor device wearing time including a first time window, a subsequent second time window, and a transition period between the first time window and the second time window, the glucose sensor device comprising: First glucose sensor; and A second glucose sensor, wherein the first glucose sensor and the second glucose sensor have different characteristics in at least one aspect of hydration, stability, and durability; and Sensor electronics, the sensor electronics comprising at least one physical microprocessor, the physical microprocessor being configured to: (a) Periodically receive a corresponding first output signal from the first glucose sensor, the first output signal indicating the glucose concentration level in the user's body; (b) During the first time window, the glucose concentration level in the user's body is calculated entirely based on the first output signal; (c) Periodically receive a corresponding second output signal from the second glucose sensor, the second output signal indicating the glucose concentration level in the user's body; (d) During the transition period, the glucose concentration level in the user's body is calculated based on both the first output signal and the second output signal; and (e) During the second time window, the glucose concentration level in the user's body is calculated entirely based on the second output signal.

2. The system of claim 1, wherein each of the first glucose sensor and the second glucose sensor is calibrated using at least one of the following: the output signal (Isig) from the other sensor, the voltage (Vcntr) from the counter electrode of each respective sensor, electrochemical impedance spectroscopy (EIS) related parameters, and the diagnostic output of each respective sensor.

3. The system according to claim 1, wherein, Based on the hydration, stability, and durability characteristics of the first and second glucose sensors, the microprocessor determines the start and end times of each of the first time window, the transition period, and the second time window.

4. The system of claim 3, wherein the microprocessor periodically fuses the first output signal and the second output signal during the transition period to calculate a single fused glucose value.

5. The system according to claim 1, wherein, During the transition period, the microprocessor compares the first output signal with the second output signal to diagnose whether each corresponding glucose sensor is operating properly.

6. The system according to claim 5, wherein, Based on the comparison and diagnosis, the microprocessor assigns corresponding weights to the first output signal and the second output signal to generate corresponding weighted first and second signals.

7. The system of claim 6, wherein the microprocessor periodically calculates a single fusion glucose value based on the weighted first and second signals.

8. The system according to claim 1, wherein both the first glucose sensor and the second glucose sensor are calibration-free sensors.

9. The system of claim 1, wherein the microprocessor uses the first output signal from the first glucose sensor to calibrate the second glucose sensor, or The microprocessor uses the second output signal from the second glucose sensor to calibrate the first glucose sensor.

10. The system of claim 1, wherein at least one of the first glucose sensor and the second glucose sensor is optionally calibrated using a reference blood glucose (BG) value.

11. A method for determining glucose concentration in a user's body during a total sensor device wearing time, the total sensor device wearing time comprising a first time window, a subsequent second time window, and a transition period between the first and second time windows, the glucose sensor device including a first glucose sensor and a second glucose sensor, the first glucose sensor and the second glucose sensor having different characteristics in at least one aspect of hydration, stability, and durability, the method comprising: (a) Periodically receive a corresponding first output signal from the first glucose sensor, the first output signal indicating the glucose concentration level in the user's body; (b) During the first time window, the glucose concentration level in the user's body is calculated entirely based on the first output signal; (c) Periodically receive a corresponding second output signal from the second glucose sensor, the second output signal indicating the glucose concentration level in the user's body; (d) During the transition period, the glucose concentration level in the user's body is calculated based on both the first output signal and the second output signal; and (e) During the second time window, the glucose concentration level in the user's body is calculated entirely based on the second output signal.

12. The method of claim 11, wherein each of the first glucose sensor and the second glucose sensor is calibrated using at least one of the following: the output signal (Isig) from the other sensor, the voltage (Vcntr) from the counter electrode of each respective sensor, electrochemical impedance spectroscopy (EIS) related parameters, and the diagnostic output of each respective sensor.

13. The method of claim 11, further comprising: Based on the hydration, stability, and durability characteristics of the first glucose sensor and the second glucose sensor, the start and end times of each of the first time window, the transition period, and the second time window are determined.

14. The method of claim 13, further comprising: During the transition period, the first output signal and the second output signal are periodically fused to calculate a single fused glucose value.

15. The method of claim 11, further comprising: During the transition period, the first output signal is compared with the second output signal to diagnose whether each corresponding glucose sensor is operating properly.

16. The method of claim 15, further comprising: Based on the comparison and diagnosis, corresponding weights are assigned to the first output signal and the second output signal to generate corresponding weighted first and second signals.

17. The method of claim 16, further comprising: The value of a single fusion glucose is calculated periodically based on the weighted first and second signals.

18. The method of claim 11, further comprising: The second glucose sensor is calibrated using the first output signal from the first glucose sensor, or The first glucose sensor is calibrated using the second output signal from the second glucose sensor.

19. The method of claim 11, wherein at least one of the first glucose sensor and the second glucose sensor is optionally calibrated using a reference blood glucose (BG) value.

20. A non-volatile computer-readable medium comprising instructions, wherein, When the instructions are executed by one or more processors, the one or more processors cause the one or more processors to perform any operation described in any one of claims 11 to 19.

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