Elliptical vector model-based frequency mixing pulse width modulation method and inversion system

Abstract

The invention discloses a mixing pulse width modulation method based on an elliptic vector model and an inverter system, and mainly relates to the technical field of mixing pulse width modulation inverters. Comprising the steps that a switching frequency generation model based on an elliptic vector model is established, and the switching frequency generation model is used for adjusting the switching frequency in real time according to the phase of sinusoidal modulation waves and completing frequency mixing modulation with controllable switching harmonic distribution frequency bands; performing discretization processing on the elliptic vector model by setting a harmonic discretization ratio and a carrier wave ratio to generate a sampling point phase sequence and a sampling period sequence in a sine period; according to the sampling period sequence, a pulse width modulation pulse sequence is generated through a natural sampling method and used for controlling on-off of an inverter switching device. The method has the beneficial effects that the problem of concentrated distribution of harmonic waves of the high-frequency switch is solved, the output efficiency of the converter is improved, and the purpose of enabling the output of the inverter to be more accurate, higher in quality and more efficient is achieved.

Description

TECHNICAL FIELD

[0001] The application relates to the field of mixed-frequency pulse width modulation inverters, in particular to a mixed-frequency pulse width modulation method based on an elliptical vector model and an inverter system. BACKGROUND

[0002] According to the use, the inverter can be roughly divided into a drive inverter and a grid-connected inverter. The drive inverter plays a core role in the motor control field, and the grid-connected inverter is mainly applied to the new energy power generation field. In addition, the power conversion process of an extra-high voltage converter station also needs to rely on the grid-connected inverter technology to be realized.

[0003] The quality of the inverter output waveform and the conversion efficiency are directly affected by the pulse width modulation mode. The sinusoidal pulse width modulation technology (SPWM) is the most basic modulation technology of the inverter, which converts the direct-current voltage into a group of pulse sequences through the on-off control of the switching device to realize the frequency conversion and voltage conversion control from direct current to sinusoidal alternating current. Based on the optimization and improvement of the sinusoidal pulse width modulation technology, the subsequent researches have proposed quasi-optimal PWM, space vector sinusoidal pulse width modulation (SVPWM), multi-level PWM and carrier phase-shifted modulation technology based on the multi-level topology.

[0004] With the increasing growth of the installed capacity of photovoltaic, wind power and other new energy sources and the development and application of extra-high voltage flexible direct current transmission, higher requirements are put forward for the modulation performance of the inverter, and the electromagnetic noise problem caused by the high-frequency harmonic of the inverter is also paid more and more attention. The harmonic spectrum of the output waveform of the constant frequency modulation method is very close to the sinusoidal modulation wave in the low frequency band, but the harmonic presents a concentrated distribution and a large peak value at the high frequency band, and a large number of narrow pulses are generated at the positive and negative amplitudes of the sinusoidal wave at a high modulation ratio, increasing the working stress of the switching device.

[0005] Therefore, an mixed-frequency pulse width modulation method based on an elliptical vector model and an inverter system are needed to solve the above problems. SUMMARY

[0006] The application aims to provide a mixed-frequency pulse width modulation method based on an elliptical vector model and an inverter system, which solves the problem of concentrated distribution of high-frequency switching harmonics, improves the output efficiency of the converter, and achieves the purpose of making the output of the inverter more accurate, better and more efficient.

[0007] In order to achieve the above purpose, the application realizes the following technical scheme: On the one hand, the application provides a mixed-frequency pulse width modulation method based on an elliptical vector model, which comprises the following steps: An elliptical vector model based switching frequency generation model is established, which is used to adjust switching frequency in real time according to the phase of sinusoidal modulation wave, and is used to complete the mixing modulation with controllable switching harmonic distribution frequency band. The expression of the elliptical vector model is as follows: ; Wherein, represents a normalized sampling period vector, which is not a space vector, but a two-dimensional quantity composed of phase and sampling period multiplier, and is used to describe the elliptical trajectory; represents the phase angle of the sinusoidal modulation wave; By setting the harmonic discrete ratio and the carrier ratio , the elliptical vector model is discretized to generate a sampling point phase sequence and a sampling period sequence within a sinusoidal period; According to the sampling period sequence, a pulse width modulation pulse sequence is generated by using the natural sampling method, which is used to control the on-off of the inverter switching device.

[0008] Preferably, the harmonic discrete ratio is defined as the ratio of the sampling period at the amplitude of the sinusoidal wave to the sampling period at the zero point of the sinusoidal wave , that is: whereby setting different values to control the switching frequency distribution bandwidth, and the carrier ratio is defined as the number of sampling periods contained in a sinusoidal modulation wave period, and different values are set to control the center frequency of the switching frequency.

[0009] Preferably, when , the elliptical focus is located on the Y axis, the switching frequency is maximum at the amplitude of the sinusoidal wave, and minimum at the zero point, and the elliptical eccentricity satisfies: ; when , the elliptical focus is located on the X axis, the sampling period is minimum at the amplitude of the sinusoidal wave, and maximum at the zero point, and the elliptical eccentricity satisfies: ; when , the ellipse degenerates into a circle, the switching frequency is constant, and is equivalent to the traditional constant frequency modulation.

[0010] Preferably, the discretization process includes the following steps: According to the given and , the initial phase increment is determined by iterative calculation ; by the recursive relationship: ; ; wherein, is the phase of the nth sampling point; is the phase increment; is the sine modulation wave frequency; is the nth sampling period; is the nth sampling period; is calculated by the elliptical model at each phase , and normalized: ; Finally, the sampling period sequence is obtained.

[0011] Preferably, the process of iterative calculation of the initial phase increment is completed by a microprocessor or an external computer, specifically including: The distance from the intersection point of the ellipse and the sampling period vector to the coordinate origin is solved by a numerical method, and the at each phase is obtained; The initial is adjusted so that the total phase accumulation is , and the number of sampling periods, i.e., the carrier ratio, is .

[0012] On the other hand, the present application also provides a system for implementing the above-mentioned mixed frequency pulse width modulation method based on an elliptical vector model, comprising: a DC bus, an AC bus, a power loop, a current and voltage detection module, a signal conversion module, a control unit, a drive circuit and an auxiliary power supply; The control unit takes a digital signal processor as the core and is configured with a PWM module and an ADC module, which are used to execute the mixed frequency modulation algorithm; The drive circuit adopts an optical coupling isolation technology to convert the PWM signal output by the control unit into an IGBT driving pulse.

[0013] Preferably, the current and voltage detection module adopts a Hall sensor, including a Hall current sensor HBC25C04 and a Hall voltage sensor HBVS025A, the output current of which is converted into a voltage signal through a measurement resistor, and then is conditioned into a single polarity signal in the range of 0-3.3V by the signal conversion module, for collection by the ADC module.

[0014] ​Preferably, the signal conversion module adopts an inverting adder and an inverter composed of operational amplifier NE5532 to convert the bipolar sensor output signal into a unipolar signal and superimpose a 1.5V bias voltage to adapt to the input range of the ADC.

[0015] Preferably, the software flow of the control unit comprises: system initialization, sensor zero point correction, ADC data acquisition and filtering, grid phase-locked loop control, PID closed loop regulation, PWM interrupt service program; In the PWM interrupt, the phase increment is obtained according to the current grid phase and the set H value, the current sampling period is calculated, and the driving pulse is obtained by updating the PWM period and the comparison value register.

[0016] Compared with the prior art, the present application has the following advantages: 1. Disperse harmonic energy and improve EMI performance: The main defect of traditional constant frequency modulation is that the switching harmonic energy is highly concentrated at the switching frequency and its multiples, forming a very high harmonic peak, which is extremely detrimental to electromagnetic compatibility (EMI). By making the switching frequency periodically change according to the elliptical law within a set frequency band (controlled by the harmonic dispersion ratio H), the high amplitude harmonic originally concentrated is dispersed into a continuous frequency band, effectively smoothing the harmonic spectrum and significantly reducing the harmonic peak at the switching frequency. This greatly reduces the design pressure of the output filter, reduces the electromagnetic interference (EMI) of the system, and improves the compliance and reliability of the product; 2. Suppress narrow pulses and improve waveform characteristics: In high-voltage high-modulation-ratio applications, constant frequency modulation produces a large number of extremely narrow pulses near the sine wave peak. These pulses are difficult to be effectively output and increase the stress on the device. The present application, in the mode, realizes the merging of multiple narrow pulses, increases the width of individual pulses, and reduces the number of narrow pulses, fundamentally eliminating the narrow pulse problem and improving the reliability of the inverter, making the output waveform more regular; 3. Effectively reduce switching loss: The switching loss of power devices (such as IGBT) is closely related to the modulation method. In the mode, the system uses a lower switching frequency in the section where the current is large (near the sine wave amplitude). Since the switching loss is closely related to the size of the current instantaneous value, reducing the switching frequency in the large current period can most effectively reduce the total switching loss. This makes the present application have higher conversion efficiency than constant frequency modulation under the same output performance, especially suitable for photovoltaic inverters, UPS and other applications that require high efficiency, which can directly improve energy utilization efficiency and reduce operating temperature rise and cooling cost. BRIEF DESCRIPTION OF DRAWINGS

[0017] Figure 1This is a flowchart of the method of the present invention; Figure 2 This is a schematic diagram of the sampling period vector model of the present invention; Figure 3 This is a schematic diagram of the pulse sequence generation method of the present invention using natural sampling. Figure 4 This is a schematic diagram of the FFT analysis results of the inverter output waveform of the present invention; Figure 5 This is a schematic diagram of the system structure of the present invention; Figure 6 This is a schematic diagram of the system software program flow of the present invention. Detailed Implementation

[0018] The present invention will be further illustrated below with reference to specific embodiments. It should be understood that these embodiments are for illustrative purposes only and are not intended to limit the scope of the invention. Furthermore, it should be understood that after reading the teachings of this invention, those skilled in the art can make various alterations or modifications to the invention, and these equivalent forms also fall within the scope defined in this application.

[0019] In this invention, terms such as "upper," "lower," "left," "right," "front," "back," "vertical," "horizontal," "side," and "bottom" indicate the orientation or positional relationship based on the orientation or positional relationship shown in the accompanying drawings. These terms are used only to facilitate the description of the structural relationships of the various components or elements of this invention and do not specifically refer to any component or element in this invention. They should not be construed as limiting the invention.

[0020] Example: like Figure 1 As shown, this embodiment provides a mixing pulse width modulation method based on an elliptic vector model, including the following steps: A switching frequency generation model based on an elliptic vector model is established. This model is used to adjust the switching frequency in real time according to the phase of the sinusoidal modulation wave, thereby achieving frequency mixing modulation with controllable frequency band of the switching harmonic distribution. The expression for the elliptic vector model is: ; in, This represents the standardized sampling period vector; This represents the phase angle of the sinusoidal modulated wave; the harmonic discrepancy ratio is set accordingly. carrier ratio The elliptical vector model is discretized to generate a sampling point phase sequence and a sampling period sequence within a sinusoidal period. Based on the sampling period sequence, a pulse width modulation pulse sequence is generated using the natural sampling method to control the on / off state of the inverter switching devices; By solving the sampling period using discrete ellipses, the switching frequency can be dynamically varied at different phases of the sine wave, thereby dispersing high-frequency harmonics, reducing total harmonic distortion (THD), and decreasing switching losses.

[0021] Specifically: The main difference between mixed-frequency pulse width modulation (PWM) and constant-frequency pulse width modulation (PWM) schemes lies in the modulation algorithm. The mixed-frequency modulation scheme proposed in this embodiment is derived based on the uniform distribution of sampling points along a sine curve as the basis for generating the pulse sequence. In this case, phase is used... and the length of the sine curve between the two sampling points The constructed vector expression is: ; vector The modulus value at the phase of the modulated wave is 0 and When the minimum value is obtained ,exist and When the maximum value is obtained Using vectors Minimum value of the modulus After standardization as a unit length, such as Figure 2 As shown, the expression for the standardized vector is: ; The trajectory of the vector at its endpoint within one sinusoidal period is an ellipse with its focus on the y-axis. The phase angle still represents the phase of the sine wave, and the magnitude represents the ratio of the current sampling period to the minimum sampling period. The magnitude of the standardized vector reaches its maximum value at the intersection of the ellipse and the y-axis. The minimum value of 1 is obtained at the intersection with the x-axis, that is: During modulation, at phase and The switching period reaches its maximum value (minimum frequency) at this time, and this minimum frequency is also the lower limit of the switching bandwidth, which appears at the amplitude of the sine wave; the phase is 0 and... The switching period reaches its minimum value (maximum frequency) at which point the maximum frequency is also the upper limit of the switching bandwidth, which occurs at the zero point of the sine wave; the harmonic discrepancy ratio is defined. The sampling period at the amplitude of the sine wave Sampling period at the zero point of the sine wave The ratio, that is: In the standardized ellipse trajectory of the vector endpoint, the value of H is... Based on the transformation relationship of the ellipse parameters, the eccentricity of the ellipse trajectory of the vector endpoint at this time can be obtained. .

[0022] For eccentricity Extending the ellipse to a general ellipse yields the unified model for the mixing modulation method proposed in this patent, namely, an elliptical vector model for calculating the switching period under different phases. When using the vector model calculated from the sampling period proposed in this patent for mixing modulation, the following patterns emerge: when At that time, the sampling points were evenly distributed along the sine curve.

[0023] when At this time, the focus of the ellipse is on the X-axis. During the process of the sine wave from zero to the amplitude, the switching frequency increases, and the minimum value is obtained at the zero point of the sine wave and the maximum value is obtained at the amplitude. when At this time, the focus of the ellipse is on the Y-axis. During the process of the sine wave from zero to the amplitude, the switching frequency decreases, and the maximum value is obtained at the zero point of the sine wave and the minimum value is obtained at the amplitude. when When the eccentricity of the ellipse is 0, the trajectory of the vector's endpoint is circular, meaning the switching period remains constant during modulation. The resulting sampling period is identical to that of constant-frequency modulation. Therefore, constant-frequency modulation is the elliptical sampling period calculation vector model proposed in this patent. exist A special form of time.

[0024] The key point of this embodiment is to discretely solve the sequence of sampling period and sampling point phase of the ellipse, thereby realizing the controllable frequency band of the switching harmonic distribution of the mixing pulse width modulation.

[0025] When using the elliptical model proposed in this embodiment for modulation, a harmonic discrepancy ratio needs to be given. The number of modulations within a sinusoidal period (i.e., carrier ratio) Two parameters, harmonic discrepancy ratio Eccentricity of the ellipse exist ( The conversion relationship of the parameter is such that when the parameter is... Once the endpoint of the vector is given, the ellipse that forms the trajectory is uniquely determined, and its analytical form is: ; In phase place, The modulus is the linear value. The distance from the intersection point of the ellipse to the origin of the coordinate system can be used to obtain the ratio of the sampling period for any phase to the minimum sampling period based on this equality relationship. In pulse width modulation, the time increment and phase increment of each sampling period satisfy the following conditions: The phase of the next sampling point is directly proportional to the phase of the current sampling point plus the phase increment. It can also be calculated using a vector model. Based on this recursive relationship, when the phase accumulates from 0° to 360°, the value at a given discrete ratio can be obtained. Given a phase sequence of sampling points within a sinusoidal period, the number of sampling points can be increased or decreased by changing the magnitude of the initial phase increment. Therefore, during the discretization process, it is necessary to find a suitable initial phase increment corresponding to the minimum sampling period to satisfy the second parameter, namely the carrier ratio. This process can be accomplished through iterative calculations by the processor; It is a standardized vector model, so the multiplier calculated during the discretization process is... This does not represent the actual sampling period size; it needs to be determined. Further processing is performed to obtain the sampling period, first for... Normalization is performed over a sinusoidal period: ; at this time This indicates that at a modulation ratio of The magnitude of the p-th sampling period of a 1Hz sinusoidal modulation wave at a frequency of 1Hz is... When, the size of the p-th sampling period is This completes the process of obtaining the sampling point phase sequence and sampling period sequence from the discretization of the ellipse containing the parameter H. The natural sampling method can then be used to convert this sampling period sequence into a sinusoidal pulse width modulated pulse sequence, such as... Figure 3 As shown; The iterative calculation of the sampling period using elliptic discretization involves a large amount of computation, posing a significant challenge to the implementation of this mixing modulation method on a microprocessor (MCU). To address this issue, the elliptic discretization process can be performed using a computer to obtain the phase corresponding to N sampling points within one modulation wave period. In an MCU, this phase sequence can be used to obtain the phase difference between two adjacent sampling points. ,according to and The carrier period can then be calculated. In the PWM interrupt service function, based on the carrier period By modifying the PWM period register, this mixing modulation method can be simplified and implemented on the MCU.

[0026] exist When the switching frequency reaches its minimum value at the zero point of the sine wave and its maximum value at the amplitude, the number of pulses at the amplitude of the sine wave decreases and the width of a single pulse increases in the pulse sequence generated by the mixed-frequency sine pulse width modulation. Therefore, the mixed-frequency modulation can effectively solve the problem of narrow pulses generated at both positive and negative amplitudes under high modulation ratios. When the instantaneous value of the sine wave changes rapidly, the number of pulses increases. The mixed-frequency modulation method can adjust the duty cycle of the pulses in a timely manner according to the rapidly changing sine wave value, so that the inverter output waveform can track the sine modulated wave more accurately. High-frequency harmonics in the inverter are mainly caused by the switching of the IGBTs during the modulation process. The pulse sequences generated by constant-frequency modulation and mixed-frequency modulation are used as drive signals to control the IGBTs in the full-bridge inverter, such as... Figure 4 The figure shows the FFT analysis results of the output pulse voltage waveforms after filtering for the two schemes. The harmonic discrepancy ratio used in the mixing modulation is... The harmonic spectrum shows that the high-frequency harmonics in the FFT spectrum of the mixing modulation method are effectively dispersed within a single frequency band. The THD of the output voltage waveform under the constant frequency modulation method is 1.12%, while that under the mixing modulation method is 1.02%. The THD performance of the mixing modulation method is 8.93% better than that of the constant frequency modulation control method. The test results of THD under other harmonic dispersion ratios are shown in Table 1. Table 1 Test results of THD under different harmonic dispersion ratios

[0027] From the perspective of switching losses, At the same operating conditions, the mixing modulation control method reduces the number of chopping operations at the amplitude level, thus resulting in lower switching losses. Therefore, the mixing modulation method proposed in this embodiment has lower switching losses. It is more efficient than constant frequency modulation method.

[0028] like Figure 5 As shown, this embodiment also provides an inverter system, including: a DC bus, an AC bus, a power circuit, a current and voltage detection section, a signal conversion section, a control unit, a drive circuit, and an auxiliary power supply section.

[0029] The DC bus connects the DC power supply to the DC side of the inverter power circuit. The power circuit adopts a three-phase full-bridge topology composed of IGBTs. The AC bus connects the AC side of the power circuit and the AC output terminal. There are also AC feedback acquisition points on the AC bus.

[0030] The current and voltage detection section uses Hall current element HBC25C04 and Hall voltage element HBVS025A to isolate and transmit the current and voltage on the AC bus. The secondary coils of the Hall elements are both current outputs. The output current I2 of the Hall element is converted into voltage after being grounded through the measuring resistor RL. The voltage Vout on the measuring resistor RL is the transmission value of the Hall sensor. The ±Vcc required by the Hall element is provided by the auxiliary power supply section.

[0031] The signal conversion section connects the Hall sensor's output signal Vout to the processor's ADC acquisition interface. The ADC module's analog voltage signal input range is 0-3.3V. The signal conversion section uses an adder and a proportionalizer built with operational amplifiers to convert the Hall sensor's bipolar output signal Vout to the 0-3.3V ADC acquisition range. The signal conversion process is divided into two stages, using the NE5532 bipolar operational amplifier, which integrates two operational amplifiers. The auxiliary power supply provides power to the signal conversion section. The system uses a bipolar operational amplifier power supply and a 1.5V DC bias power supply. The first-stage operational amplifier, together with resistors R4, R5, R6, and R10, forms an inverting adder. The Vout signal is amplified by R5 / R6 and then added to the 1.5V bias voltage to form the signal im2. The second-stage operational amplifier forms an inverter that converts the negative signal im2 from the inverting adder into a signal within the 0-3.3V ADC acquisition range.

[0032] The control unit uses the DSPF28335 digital processor as its core to realize the mixing and modulation process. The DSP's ADC module completes the acquisition of the feedback quantity and performs closed-loop control-related calculations based on the difference between the feedback quantity and the given quantity. Finally, the frequency and duty cycle of the pulse-type drive signal output by the PWM module are adjusted according to the calculation results.

[0033] The drive circuit connects the control unit and the power circuit, providing electrical isolation between the two. An optocoupler converts the drive signal output from the control unit into drive pulses that control the IGBT's on / off state. The drive signal from the control unit, together with the LED inside the optocoupler, forms a control loop, controlling the on / off state of the phototransistor to generate drive pulses with driving capability. Electrical isolation between the two sides of the optocoupler is achieved through optical transmission. The auxiliary power supply provides DC power between VCC and PGND for the drive circuit.

[0034] The program flowchart of the software control system in this embodiment is as follows: Figure 6As shown, the main function begins by initializing the DSP peripherals used in the system, specifying the operating mode of each peripheral module. After all peripherals are ready, the initial values ​​of the system operating parameters are set. After initial settings are completed, zero-point offset correction is performed on the sensors. If a sensor signal with a large offset is found during correction, a sensor abnormality is indicated and subsequent program execution is stopped. In the main loop, the ADC conversion results are first read from the DMA transfer buffer, and then the read data is filtered. Another part of the main function checks whether parameters such as the inverter output current amplitude, power factor, and discrepancy ratio need to be adjusted.

[0035] In the timer closed-loop interrupt function, phase-locked loop (PLL) is first established with respect to the grid voltage. Modulation is not initiated until stable tracking of the grid frequency and phase is achieved. The interrupt service function exits immediately after executing the PLL-related functions. If the PLL has already achieved stable tracking of the grid frequency and phase, it checks if this is the first time stable tracking has been achieved. If so, a PWM pulse width modulation interrupt is initiated. Based on the frequency and phase of the current grid voltage output by the PLL, the d-axis and q-axis components of the current in the rotating coordinate system are calculated. After normalizing the current using the given d-axis component Id0, it is input to the PID controller for closed-loop feedback control. Finally, the result output by the PID controller is returned as a named value.

[0036] In the PWM modulation interrupt, the phase increment corresponding to this modulation is first obtained by looking up the current phase of the grid voltage output from the phase-locked loop and the set discrete parameter h in a table. The sampling period of this modulation can be calculated from the grid frequency and the phase increment. The count value of the PWM module's period register is then calculated and modified based on the PWM module's counting frequency. The output of the PID controller is transformed between rotating and stationary coordinate systems to obtain the composite vector. The duty cycle of the current sampling period pulse can be calculated using parameters such as the amplitude, phase, and DC bus voltage of the composite vector. Finally, based on the PWM module's period register count value and duty cycle, the PWM module's comparison register is modified accordingly, thus completing the modulation of one sampling period.

[0037] The above is a detailed description of the preferred embodiments of the present invention. However, the present invention is not limited to the embodiments described. Those skilled in the art can make various equivalent modifications or substitutions without departing from the spirit of the present invention. All such equivalent modifications or substitutions are included within the scope defined by the claims of this application.

Claims

1. A mixing pulse width modulation method based on an elliptic vector model, characterized in that, Includes the following steps: A switching frequency generation model based on an elliptic vector model is established. This model is used to adjust the switching frequency in real time according to the phase of the sinusoidal modulation wave, thereby achieving frequency mixing modulation with controllable frequency band of the switching harmonic distribution. The expression for the elliptic vector model is: ； in, The standardized sampling period vector is not a spatial vector, but a two-dimensional quantity composed of phase and sampling period multiplier, used to describe elliptical trajectories. This represents the phase angle of the sinusoidal modulated wave; By setting the harmonic dispersion ratio carrier ratio The elliptical vector model is discretized to generate a sampling point phase sequence and a sampling period sequence within a sinusoidal period. Based on the sampling period sequence, a pulse width modulation pulse sequence is generated using the natural sampling method to control the on / off state of the inverter switching devices.

2. The mixing pulse width modulation method based on an elliptic vector model according to claim 1, characterized in that, The harmonic dispersion ratio Defined as the sampling period at the amplitude of the sine wave Sampling period at the zero point of the sine wave The ratio, that is: Among them, by setting different The value enables control over the bandwidth distribution of the switching frequency, wherein the carrier ratio Defined as the number of sampling periods contained within a sinusoidal modulation wave period, by setting different... The value enables control of the center frequency of the switching frequency.

3. The mixing pulse width modulation method based on an elliptic vector model according to claim 2, characterized in that, when At this time, the ellipse's focus is located on the Y-axis, and the switching frequency is maximum at the sine wave amplitude and minimum at the zero-crossing point. Eccentricity of the ellipse satisfy: ; when At this time, the ellipse's focus is located on the X-axis, and the sampling period is minimum at the sine wave amplitude and maximum at the zero-crossing point. Eccentricity of the ellipse satisfy: ; when At this time, the ellipse degenerates into a circle, the switching frequency is constant, which is equivalent to traditional constant frequency modulation.

4. The mixing pulse width modulation method based on an elliptic vector model according to claim 1, characterized in that, The discretization process includes the following steps: According to the given and The initial phase increment is determined through iterative calculation. ; Based on the recursive relation: ； ； in, For the first The phase of each sampling point; This is the phase increment; The frequency of the sinusoidal modulation wave; For the first One sampling period; The sampling period multiplier at each phase is calculated using an elliptic model. And perform normalization: ； Finally, the sampling period sequence is obtained. .

5. The mixing pulse width modulation method based on an elliptic vector model according to claim 4, characterized in that, The iterative calculation of the initial phase increment is performed by a microprocessor or an external computer, specifically including: The distance from the intersection of the ellipse and the sampling period vector to the origin is calculated using numerical methods, yielding the distance at each phase. ; By adjusting the initial This makes the total phase accumulation to And satisfy the number of sampling periods, i.e., the carrier ratio is .

6. A system for implementing a mixing pulse width modulation method based on an elliptic vector model as described in claims 1-5, characterized in that, include: DC bus, AC bus, power circuit, current and voltage detection module, signal conversion module, control unit, drive circuit and auxiliary power supply; The control unit is based on a digital signal processor and is equipped with a PWM module and an ADC module to execute the mixing modulation algorithm. The drive circuit uses optocoupler isolation technology to convert the PWM signal output by the control unit into IGBT drive pulses.

7. The system according to claim 6, characterized in that, The current and voltage detection module uses Hall sensors, including Hall current sensor HBC25C04 and Hall voltage sensor HBVS025A. Its output current is converted into a voltage signal by a measuring resistor, and then conditioned by a signal conversion module into a unipolar signal in the range of 0–3.3V for acquisition by the ADC module.

8. The inverter system according to claim 6, characterized in that, The signal conversion module uses an inverting adder and an inverter composed of an operational amplifier NE5532 to convert the bipolar sensor output signal into a unipolar signal, and adds a 1.5V bias voltage to adapt to the ADC input range.

9. The inverter system according to claim 6, characterized in that, The software flow of the control unit includes: System initialization, sensor zero-point calibration, ADC data acquisition and filtering, power grid phase-locked loop control, PID closed-loop regulation, PWM interrupt service routine; In the PWM interrupt, the phase increment is obtained by looking up the table based on the current grid phase and the set H value, the current sampling period is calculated, and the PWM period and comparison value register are updated to obtain the drive pulse.

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