Tunneling magnetoresistance sensor system and method for high power density power electronics systems

Abstract

A contactless current sensing system and method for high power density power electronics systems are provided. The contactless current sensing system is a two current sensor system in which two current sensors having substantially the same sensitivity are positioned on opposite sides of a current-carrying conductor.

Description

TUNNELING MAGNETORESISTANCE SENSOR SYSTEM AND METHOD FOR HIGH POWER DENSITY POWER ELECTRONICS SYSTEMSCROSS REFERENCE TO RELATED APPLICATION

[0001] The present disclosure claims the benefit of U.S. Provisional Patent Application No. 63 / 636,311 filed April 19, 2024, the entire content and disclosure of which is incorporated herein by reference.GOVERNMENT SUPPORT

[0002] This invention was made with government support under DE-EE0008707 awarded by the Department of Energy. The government has certain rights in the invention.FIELD OF THE INVENTION

[0003] The present disclosure relates to high power density power electronics systems, and more particularly to a contactless current sensing system for high power density power electronics systems and a method for measuring current using the same.BACKGROUND

[0004] With the emergence of next-generation power semiconductor devices based on Wide Band Gap (WBG) technology, the switching frequency of power electronic converters is advancing into the megahertz (MHz) range. This shift ensures higher power density by reducing the size of passive components and enhances power efficiency due to the fast-switching capability and low on-resistance of WBG devices. To keep pace with these advancements in high-frequency power electronics, significant developments are imperative for other essential elements of power converters, such as controllers, sensors, and thermal management systems.

[0005] Current state variable is one critical parameter in power electronic converters, serving various purposes including closed-loop control, short-circuit and ovcrcurrcnt protection, diagnostics / prognostics, and life prediction. However, traditional current sensing strategies prove inadequate for measuring current in modem power converters due to their requirement for high-bandwidth and accuracy, coupled with the need for a small footprint. Therefore, it is important to research various high-bandwidth current sensing solutions for measuring the current in modern power electronic systems.

[0006] Power converters can be divided into two main groups: (1) power converters based on power modules or (2) power converters based on discrete devices. These two separate groups of power converters have their pros and cons and challenge in developing advanced current sensing approaches.

[0007] Power modules provide various benefits including, for example, housing multiple components like switching devices such as transistors and diodes, and usually thermal management and cooling parts in a single package. Thus, using power modules for developing power converters can simplify the design. However, power modules are not available for all required configurations and ratings, and they are costly.

[0008] Using discrete devices as power converters provides more freedom in choosing the switching devices and configurations based on the application’s requirements. However, this privilege comes at a cost of more complexity and challenges accompanied by thermal management solutions.

[0009] Thus, based on the limits and requirements of the application, power converters can be designed with discrete devices or power modules. In either case however, the necessity of measuring the current is the same, and each of these approaches poses various challenges when implementing high-frequency current measurement solutions.SUMMARY

[0010] A contactless current sensing system and method for high power density power electronics systems are provided. The contactless current sensing system is a two current sensor system in which two current sensors, such as, for example, two tunneling magnetoresistance (TMR) sensors, having substantially the same sensitivity are positioned on opposite sides of a current-carrying conductor. The contactless current sensing system of the present disclosure can be used to provide a high-precision current measurement solution to address the challenges of current measurement in WBG power electronics. In some embodiments, the contactless current sensing system of the present disclosure provides a high-precision current measurement solution for WBG power electronics with high dv / dt, high di / dt and high switching frequencies.

[0011] In embodiments of the present disclosure, a contactless current sensing system is provided that includes a current-carrying conductor having a first side and a second side opposite the first side; a first current sensor present in a first location on the first side of the currentcarrying conductor; and a second current sensor present in a second location on the second side of the current-carrying conductor, where the first current sensor and the second current sensor have a substantially same sensitivity and are spaced apart from the current-carrying conductor, and wherein the first location is substantially identical to the second location.

[0012] In some embodiments, the contactless current sensing system of the present disclosure further includes a first dielectric barrier separating the first current sensor from the currentcarrying conductor, and a second dielectric barrier separating the second current sensor from the current-carrying conductor.

[0013] In some embodiments, the current-carrying conductor is a bus bar or a component of a printed circuit board.

[0014] In some embodiments, the first current sensor and the second current sensor of the present disclosure are both magnetoresistance sensors.

[0015] In embodiments in which magnctorcsistancc sensors arc employed as the first current sensor and the second current sensor, the magnetoresistance sensors include anisotropic magnetoresistance sensors, giant magnetoresistance sensors or tunneling magnetoresistance sensors.

[0016] In some embodiments, the first current sensor and the second current sensor employed in the present disclosure are both Hall sensors, Rogowski sensors or shunt sensors.

[0017] In some embodiments, the first current sensor and the second current sensor employed in the present disclosure are both tunneling magnetoresistance sensors.

[0018] In any of the embodiments mentioned above, the current-carrying conductor can, in some instances, have two openings separating a section of the current-carrying conductor, and the first location of the first current sensor and the second location of the second current sensor are both in an area defined by the section of the current-carrying conductor that is located between the two openings.

[0019] In embodiments of the present disclosure, the contactless current sensing system can further include a first voltage input source electrically connected to a first portion of the first current sensor and a first voltage output source electrically connect to a second portion of the first current sensor, and a second voltage input source electrically connected to a first portion of the second current sensor and a second voltage output source electrically connect to a second portion of the second current sensor.

[0020] In embodiments of the present disclosure, the first voltage input source, the first voltage output source, the second voltage input source and the second voltage output source are spaced apart from the current-carrying conductor.

[0021] In embodiments of the present disclosure, the contactless current sensing system further includes a first wire electrically connecting the first voltage input source to the first portion of the first current sensor, a second wire electrically connecting the first voltage output source to the second portion of the first current sensor, a third wire electrically connecting the second voltage input source to the first portion of the second current sensor, a fourth wire electrically connecting the second voltage output source to the second portion of the second current sensor, where the first wire, the second wire, the third wire and the fourth wire are spaced apart from the currentcarrying conductor.

[0022] In any of the embodiments of the present disclosure, the current-carrying conductor is a component of a power module.

[0023] In any of the embodiments of the present disclosure, the current-carrying conductor is a component of a power converter with discrete devices.

[0024] In one highly preferred embodiment, a contactless current sensing system is provided that includes a current-carrying conductor having a first side and a second side opposite the first side, where the current-carrying conductor has two openings separating a section of the currentcarrying conductor; a first tunneling magnetoresistance (TMR) sensor present in a first location on the first side of the current-carrying conductor; and a second TMR sensor present in a second location on the second side of the current-carrying conductor, where the first TMR sensor and the second TMR sensor have a substantially same sensitivity and are spaced apart from the current-carrying conductor, and wherein the first location is substantially identical to the second location, and wherein the first location of the first TMR sensor and the second location of the second TMR sensor are both in an area defined by the section of the current-carrying conductor that is located between the two openings.

[0025] In some embodiments, the contactless current sensing system including the two TMR sensors further includes a first dielectric barrier separating the first TMR sensor from the current-carrying conductor, and a second dielectric barrier separating the second TMR sensor from the cun’cnt-carrying conductor.

[0026] In embodiments of the present disclosure including the two TMR sensors, the contactless current sensing system can further include a first voltage input source electrically connected to a first portion of the first TMR sensor and a first voltage output source electrically connect to a second portion of the first TMR sensor, and a second voltage input source electrically connected to a first portion of the second TMR sensor and a second voltage output source electrically connect to a second portion of the second TMR sensor.

[0027] In embodiments of the present disclosure including the two TMR sensors, the first voltage input source, the first voltage output source, the second voltage input source and the second voltage output source are spaced apart from the current-carrying conductor.

[0028] In embodiments of the present disclosure including the two TMR sensors, the contactless current sensing system further includes a first wire electrically connecting the first voltage input source to the first portion of the first TMR sensor, a second wire electrically connecting the first voltage output source to the second portion of the first TMR sensor, a third wire electrically connecting the second voltage input source to the first portion of the second TMR sensor, a fourth wire electrically connecting the second voltage output source to the second portion of the second TMR sensor, where the first wire, the second wire, the third wire and the fourth wire are spaced apart from the current-carrying conductor.

[0029] In any of the embodiments of the present disclosure including the two TMR sensors, the current-carrying conductor is a component of a power module.

[0030] In any of the embodiments of the present disclosure including the two TMR sensors, the current-cany ing conductor is a component of a power converter with discrete devices.BRIEF DESCRIPTION OF THE DRAWINGS

[0031] FIG. 1 is cross sectional view of a contactless current sensing system in accordance with an embodiment of the present disclosure.

[0032] FIG. 2 is a circuit diagram for a magnetoresistance current sensor that can be employed in the present application.

[0033] FIG. 3 is a top down view of contactless current sensing system in accordance with an embodiment of the present disclosure in which a slitted current-carrying conductor is employed.

[0034] FIG. 4A is an illustration of a contactless current sensing system in accordance with an embodiment of the present disclosure including a first tunneling magnetoresistance sensor present on a first side of a bus bar, and a second tunneling magnetoresistance sensor present on a second side of the bus bar, in which the second side is opposite the first side.

[0035] FIG. 4B is an illustration of a contactless current sensing system in accordance with an embodiment of the present disclosure including a first tunneling magnetoresistance sensor present on a first side of a printed circuit board, and a second tunneling magnetoresistance sensor present on a second side of the printed circuit board, in which the second side is opposite the first side.

[0036] FIG. 5A is an illustration of a system including a power module with the contactless current sensing system attached to the output AC terminal.

[0037] FIG. 5B is a test circuit schematic used as a test bed for the system illustrated in FIG. 5A.

[0038] FIG. 6 is a schematic showing a test setup for verification of the contactless current sensing system illustrated in FIGS. 5A-5B.

[0039] FIG. 7 is a top down view of a first design, Design I, of the present disclosure for TTS system validation.

[0040] FIG. 8 is the Design I test results for 50 A peak-to-peak 70 kHz current pulses.

[0041] FIG. 9 is a top down view of a second design, Design II, of the present disclosure for investigating the impact of capacitance coupling.

[0042] FIG. 10 is the Design II test results for 30 A peak-to-peak 70 kHz current pulses.

[0043] FIG. 11 is a top down view of a third design, Design III, of the present disclosure for alleviating capacitance coupling and enabling high current-carrying capability.

[0044] FIG. 12A is the Design I results for the Bxfield component.

[0045] FIG. 12B is the Design III results for the Bxfield component.

[0046] FIG. 13 is the Design III test results for 50 A peak-to-peak 70 kHz current pulses.

[0047] FIG. 14A is a top down view of a fourth design, Design IV, of the present disclosure.

[0048] FIG. 14B is a top down view of a developed prototype of the fourth design of the present disclosure.

[0049] FIG. 15 is the Design IV test results for 50 A peak-to-peak 70 kHz current pulses.

[0050] FIG. 16A is a conceptual drawing of a power converter employed in Example 2 of the present disclosure.

[0051] FIG. 16B is a hardware protype of the power converter employed in Example 2 of the present disclosure.

[0052] FIG. 17 illustrates a bus bar with a TTS solution in accordance with the present disclosure.

[0053] FIG. 18 is a circuit schematic for signal conditioning for the bus bar with the TTS solution in accordance with the present disclosure.

[0054] FIG. 19 is a schematic of an experimental set up for use with the for the bus bar with the TTS solution in accordance with the present disclosure.

[0055] FIG. 20 shows the test results comparing the performance of current probe for a shunt resistor and the TTS solution in accordance with the present disclosure.DETAIEED DESCRIPTION

[0056] The present disclosure will now be described in greater detail by referring to the following discussion and drawings that accompany the present disclosure. In the following description, numerous specific details are set forth, such as particular structures, components, materials, dimensions, processing steps and techniques, in order to provide an understanding of the various embodiments of the present disclosure. However, it will be appreciated by one of ordinary skill in the art that the various embodiments of the present disclosure may be practiced without these specific details. As used throughout the present disclosure, the term “about” generally indicates no more than ±10 %, ±5 %, ±2 %, ±1 % or ±0.5 % from a number. When a range is expressed in the present disclosure as being from one number to another number (e.g., 20 to 40), the present disclose contemplates any numerical value that is within the range (i.e., 22, 24, 26, 28.5, 31, 33.5, 35, 37.7, 39 or 40) or any in amount that is bounded by any of the two values that can be present within the range (e.g., 28.5-35).

[0057] The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the present disclosure. As used herein, the singular forms “a”, “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprises” and / or “comprising”, when used in this disclosure, specify the presence of stated features, integers, steps, operations, elements, and / or components, but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and / or groups thereof.

[0058] Within current technologies which implement high power and high-efficiency power converters, the current measurement solution must have different features such as: (1) Wide current measurement bandwidth: Capturing acceptable switching details of a WBG device demands its accompanying current sensor to have a wide bandwidth, ideally from de to MHz or gigahertz (GHz); (2) Wide current range: The current measurement solution should be able to handle high currents, primarily based on the requirements of the application, if it may be subjected to transient currents and spikes; (3) Noninvasive integration with no, or minimum, stray induction addition and design choice: Stray inductance can generate voltage over / undershoots when switching WBG devices. Therefore, it is critical that the current measurement solution adds negligible as possible inductance to the current path. Also, it is generally preferred that the addition of the current measurement solution does not change the design of the power module or add more complexity; (4) Small dimensions: To achieve high power density, the size of the power converters needs to be small. Therefore, the current measurement solution should also be small to make integration feasible yet have high reliability and precision; (5) High noise immunity: Due to harsh dv / dt environment driven by the fast switching of devices, common-mode noise related concerns may appeal-, mainly since growing power density necessities components to be placed near each other, which can exacerbate electromagnetic interference problems, leading to errors in the current measurement. Therefore, selecting a current measurement solution immune to noise or applying techniques to acquire this feature is essential; (6) Low cost: Developing power converters at a lower cost is one main goal in all applications. This aim highlights the need for applying current measurement solutions withlow cost, but prominent in other features; and (7) Wide temperature range: Fast switching of the devices can lead to higher temperature, especially inside the power modules. Therefore, the current measurement solution should have stability and acceptably low drift over a wide temperature range.

[0059] The present disclosure provides a current measurement solution for high power and high- efficiency power converters that satisfies many, if not all, of the features mentioned above. Notably, the present disclosure provides a current measurement solution including a contactless and high-precision current measurement approach employing two current sensors such as, for example, tunneling magnetoresistance (TMR) sensors, which are located on opposing sides of a current-carrying conductor. In embodiments in which TMR sensors are employed, the contactless current sensing system of the present disclosure can be referred to as a Two TMR Sensor (TTS) system, which includes two TMR sensors positioned on opposite sides of the current-carrying conductor. In some embodiments, particularly when TMR sensors are employed, the challenges posed by current measurement in WBG power electronics with high dv / dt, di / dt, and switching frequencies is addressed. Notably and is some embodiments, a contactless current sensing system and method for high power density power electronics systems are provided.

[0060] Considering that power converters can be categorized into two main groups: those based on power modules and those based on discrete devices, the system and method of the present disclosure is applicable to both categories. Therefore, the system and method does not have limitations on its use for either module-level or system-level power electronics systems.

[0061] The system and method of the present disclosure increases the resolution of sensor readings while decreasing or eliminating the impact of noise and temperature fluctuations on sensor performance. In certain embodiments, a specific current-carrying conductor design, known as the slitted design, is incorporated to eliminate the skin effect and capacitive coupling phenomenon. Finally, the system and method of the present disclosure can be implemented onthe slitted current-carrying conductor design to achieve a highly accurate and noise-immune current measurement solution across a wide range of frequencies.

[0062] In certain embodiments, the system and method of the present disclosure is applicable to any industry where monitoring and measuring current is needed. The components of the system can be power modules or system level power converters with discrete devices.

[0063] In further embodiments, the system and method of the present disclosure is applicable to any sensor integrated power electronic system for: component level health monitoring; component level fault monitoring; and component lifetime estimation.

[0064] In certain embodiments, the system and method of the present disclosure provides advantages, including the following: (1) Applicable to both power module and system level converters. (2) Implementing the system and method does not need any change in the layout of the system. (3) Although it is a high-precision solution, it is cost effective, small size, high bandwidth with no power loss, capable of measuring both AC and DC currents and it is a high- resolution solution. (4) The system and method mitigats all weaknesses of MR sensors such as sensitivity to external magnetic fields (noise) and temperature fluctuations, and small output voltage. (5) The system and method is not limited to TMR sensors and it is applicable with different current sensing solutions such as Hall sensors, and other current sensors as further defined discussed herein.

[0065] In some embodiments, a low-cost, noninvasive, precise, and high-bandwidth TTS current measurement solution is provided that is based on TMR current sensors for next-generation WBG-based power modules and power converters. The TTS measurement solution includes TTSs located on two opposite sides of a current-carrying conductor. The TTS measurement solution is compatible with any configuration, power module or power converter. Further, due to the proximity of the two sensors, the TTS solution of the present disclosure is insensitive to temperature variation with twice the sensitivity. The efficacy of the TTS solution of the present disclosure is validated on an in-house developed double-sided cooled power module and a 3-Dpackaged power converter, as will be described in the examples of the present disclosure. A current-carrying conductor design is disclosed or the power module, which alleviates the skin effect and has lower parasitic capacitive coupling due to wire-bonded connections. Likewise, a bus bar-based TTS solution, coupled with a high CMRR-based signal conditioning, is disclosed and benchmarked against a known shunt measurement solution for the power converter. The TTS solution outperforms the shunt measurement system with high bandwidth, enabling the development of advanced health monitoring and fault prognostics and diagnostics for future WBG-based power electronics.

[0066] These and other aspects of the present disclosure will now be described in greater detail by first referring to FIG. 1. FIG. 1 illustrates (in cross sectional view) a contactless current sensing system 10 in accordance with the present disclosure. Notably, the contactless current sensing system 10 illustrated in FIG. 1 includes current-carrying conductor 12 having a first side, SI, and a second side, S2, in which S2 is opposite SI. The contactless current sensing system 10 illustrated in FIG. 1 further includes first current sensor 16A present in a first location on the first side, SI, of the current-carrying conductor 12, and a second current sensor 16B present in a second location on the second side, S2, of the current-carrying conductor 12. The placement of the first current sensor 16A and the second current sensor 16B on opposite sides of the currentcarrying conductor 12 provides noise cancellation for its sensing current.

[0067] In accordance with the present disclosure, the first current sensor 16A and the second current sensor 16B have a substantially same sensitivity. The term “substantially same sensitivity” is used throughout the present disclosure to denote that the sensitivity of the first current sensor 16A is the same as, or is within ±10 %, ±5 %, ±2 %, ±1 %, ±0.5 %, or ±0.1 % of the sensitivity of the second current sensor 16B. Using current sensors having substantially the same sensitivity leads to a more accurate assessment of the current flowing through the currentcarrying conductor 12. In the present disclosure, the sensitivity of the current sensors that are employed can vary depending on the application in which they are employed. A current sensor is a device that detects electric current in current-carrying conductor 12 and generates a signal proportional to that current. The generated signal could be analog voltage or current or a digitaloutput. The generated signal can be then used to display the measured current in an ammeter, or can be stored for further analysis in a data acquisition system, or can be used for the purpose of control.

[0068] The sensed current and the output signal can be:AC (alternating current) input, analog output, which duplicates the wave shape of the sensed current, bipolar output, which duplicates the wave shape of the sensed current, unipolar output, which is proportional to the average or RMS value of the sensed current.(2) DC (direct current) input, unipolar, with a unipolar output, which duplicates the wave shape of the sensed current. digital output, which switches when the sensed current exceeds a certain threshold.

[0069] In the present disclosure, sensor sensitivity refers to the ability of a current sensor to detect and respond to changes in current. In one example, the sensitivity of the current sensors that are employed are from about -50 A (i.e., Amps) to about 50 A.

[0070] In accordance with the present disclosure, the first current sensor 16A and the second current sensor 16B are spaced apart from the current-carrying conductor 12; thus the first current sensor 16A and the second current sensor 16B are not in direct physical contact with an electrically conductive surface of the current-carrying conductor 12. In embodiments of the present disclosure, a dielectric barrier can be used to physical separate, and electrically isolation, the first current sensor 16A and the second current sensor 16B from the current-carrying conductor 12. In the embodiment illustrated in FIG. 1, a first dielectric barrier 14A separates the first current sensor 16A from the current-carrying conductor 12, and a second dielectric barrier 14B separates the second current sensor 16B from the current-cany ing conductor 12. Note that the dielectric bander can be a component (uppemiost or bottommost dielectric material layer of aPCB. The separation of the first current sensor 16 A and the second current sensor 16B from the current-carrying conductor 12 prevents electrical shorting of the contactless current sensing system 10 of the present application. The first dielectric banner 14A and the second dielectric barrier 14B can be composed of a dielectric material that can electrically isolate the sensors from the current-carrying conductor 12. Exemplary dielectric materials include, but are not limited to, silicon dioxide, silicon nitride, silicon oxynitride, and / or organic dielectric materials. In some embodiments, the first dielectric barrier 14A and / or the second dielectric barrier 14B can be an air gap. The thickness of the first dielectric barrier 14A and the second dielectric barrier 14B can vary depending on the application in which the contactless current sensing system 10 is employed. By way of one example, the first dielectric barrier 14A and the second dielectric barrier 14B can have a thickness from about 1 nm to about 50 nm. It is noted that the dielectric barrier (i.e. , the first dielectric barrier 14A and the second dielectric barrier 14B) is not only is used to separate the first current sensor 16A and the second current sensor 16B from the currentcarrying conductor 12, but it also electrically isolates and separates the supply voltage lines / wires and output voltage lines / wires of each sensor from the current-carrying conductor 12.

[0071] In some embodiments, each of the first dielectric barrier 14A and the second dielectric barrier 14B can be a continuous layer that is present across an entirety of the current-carrying conductor 12. In other embodiments, the first dielectric barrier 14A and the second dielectric barrier 14B can be discrete pads that are present between the sensor and the conductive surface of the current-carrying conductor (pads can also be located between the input and output sources that are electrically connected to the sensor). The pads do not cover an entirely of the conductive surface of the current-carrying conductor 12.

[0072] In accordance with the present disclosure, the first location of the first current sensor 16A is substantially the same as the second location of the second current sensor 16B. The term “substantially same” when used in connection with “the first location” and the “second location” denotes that the first current sensor 16A accompanies an area on the first side, SI, that is the same as, or is within ±10 %, ±5 %, ±2 %, ±1 %, ±0.5 %, or ±0.1 % of an area on the second side, S2 that includes the second current sensor 16B. Thus, the area defined by the first current sensor16A can be vertically aligned over an area defined by the second current sensor 16B of there can be some staggering (with substantially some overlapping or slightly no overlapping) between the area defined by the first current sensor 16A and the area defined by the second current sensor 16B. Stated in other terms, the first current sensor 16A and the second current sensor 16B are in substantial vertical proximity to each other. By accompanying a substantially same location on each side of the current-carrying conductor 12, the first current sensor 16A and the second current sensor 16B are exposed to the same temperature and interference from external fields. As such, the noise cancellation effect is maximized by strategically locating the two current sensors in similar positions to experience comparable external fields and noise.

[0073] The current-carrying conductor 12 that can be employed in the present application can be composed of an electrically conductive metal or electrically conductive metal alloy. Exemplary electrically conductive metals that can be used as the current-carrying conductor 12 include, but are not limited to, copper (Cu), aluminum (Al), or tungsten (W). An exemplary electrically conductive metal alloy that can be used as the current-carrying conductor 12 can be a Cu-Al alloy. The current-carrying conductor 12 can be bus bar, a component of a printed circuit board (PCB), or any other current-carrying conductor-containing structure. In some embodiments, the current-carrying conductor 12 can be referred to as a current-carrying trace. A PCB is a rigid structure that contains electrically circuitry made up of electrically conductive metal-containing surfaces called traces, and larger areas of metal called planes. PCBs are made of dielectric (i.e. electrically insulating) materials with a thin layer of an electrically conductive material (i.e., an electrically conductive metal or electrically conductive metal alloy) laminated onto its surface. The conductive layer is then etched to create a pattern of conductive pathways of traces, which connect different components on the PCB. A bus bar’ includes a power rail that is composed of an electrically conductive metal or electrically conductive metal alloy, both as mentioned above.

[0074] Current sensors (or current measurement sensors) that can be used in the present application as the first current sensor 16A and the second current sensor 16B include sensors, such as, for example, shunt resistors that measure voltage drop as a proxy for current, nonmagnetic methods, and sensors such as, for example, Hall effect sensors, Rogowski coil sensors,and magnetoresistance (MR) sensors, that measure magnetic field as a representative of the current in a conductor, magnetic methods. Of the various sensors that can be employed in the present application, those that measure magnetic field as a representative of the current in a conductor, magnetic methods are generally preferred over those that measure voltage drop as a proxy for current, non-magnetic methods. Of the current sensors that measure magnetic field as a representative of the current in a conductor, magnetic methods, MR sensors are preferred over the other types of current sensors that measure magnetic field as a representative of the current in a conductor, magnetic methods. MR sensors are preferred in the present disclosure since they are the most compatible option to be used for integrated current measurement.

[0075] MR sensors work by the linear variation of resistance of MR caused by the magnetic field generated by the current to be measured. In the configuration of MR sensors, resistors, R, are configured as a Wheatstone bridge to enhance the linearity of the sensor and compensate for thermal drift. As exhibited in FIG. 2, the Vout is proportional to the magnetic flux density and the current. There are different type of MR sensors, but all types have the same measuring principle. The different sensors are known as anisotropic magnetoresistive (AMR), giant magnetoresistive (GMR), and tunneling magnetoresistive (TMR). The only difference between the three groups is the selected magnetoresistors. AMR only includes a single layer of ferromagnetic materials.The resistance variation of AMR magnetoresistors is about 2%-4% as the magnetization in the layer turns from parallel to perpendicular to the current direction.

[0076] GMR, which includes multilayers composed of alternating ferromagnetic and nonmagnetic conductive layers, is introduced to enhance the resistance change ratio. The fundamental phenomenon of GMR consequences is the current flowing through two magnetic elements separated by a thin, non-magnetic spacer layer, which enhances the resistance change of the GMR to 12%-20%.

[0077] TMR, which includes a tunnel barrier layer positioned between a magnetic reference layer and a magnetic free layer, can further enhance the resistance change ratio. TMR has the highest resistance variation percentage, which leads to the highest single-to-noise ratio (SNR).Moreover, TMR has the lowest temperature drift, which is essential when used to measure the current inside a power module since the temperature inside a power module can alter considerably for different loads. Therefore, based on the promising features of TMR in high- frequency current measurement, they are preferred options to be used for high-frequency current measurements solution for power modules. In some embodiments, the present disclosure provides a solution for increasing the SNR and alleviating the limitations of MR sensors, which are noise immunity and temperature drift, is disclosed.

[0078] In some embodiments of the present disclosure, the first current sensor 16A and the second current sensor 16B are both MR sensors including, for example, AMR sensors, GM sensors and TMR sensors. In other embodiments of the present disclosure, the first current sensor 16A and the second current sensor 16B are both are both Hall sensors, Rogowski sensors or shunt sensors. In some highly preferred embodiments of the present disclosure in which high- frequency current measurements are required, the first current sensor 16A and the second current sensor 16B are both are both TMR sensors.

[0079] In embodiments of the present disclosure and as further illustrated in FIG. 1, the contactless current sensing system 10 can further include a first voltage input source, VSiul, electrically connected to a first portion of the first current sensor 16A and a first voltage output source VSoutl electrically connect to a second portion of the first current sensor 16A, and a second voltage input source, VSin2, electrically connected to a first portion of the second current sensor 16B and a second voltage output source, VS0Ut2, electrically connect to a second portion of the second current sensor 16B.

[0080] In the present application, the first voltage input source, VSinl, the first voltage output source, VSoutl, the second voltage input source, VSin2, and the second voltage output source VSOut2,are spaced apart from the current-carrying conductor 12. The spacing can be achieved utilizing a dielectric barrier as defined above.

[0081] In embodiments, the contactless current sensing system further includes a first wire 18A electrically connecting the first voltage input source, VSinl, to the first portion of the first current sensor 16A, a second wire 18B electrically connecting the first voltage output source, VSOutl,to the second portion of the first current sensor 16 A, a third wire 18C electrically connecting the second voltage input source, VSin2, to the first portion of the second current sensor 16B, a fourth wire 18D electrically connecting the second voltage output source, VSOut2, to the second portion of the second current sensor 16B, where the first wire 18A, the second wire 18B, the third wire 18C and the fourth wire 18D are spaced apart from the current-carrying conductor 12. In some embodiments the various wires are formed utilizing a wiring bonding process. The wires are composed of an electrically conductive material as mentioned above.

[0082] In the present application, the signals from, the first and second output sources can be used in determining the AC or DC current that flows through the current-carrying conductor 12. These aspect of the present disclosure will be described in greater detail herein below.

[0083] In some embodiments (referred to as Design I), a current-carrying conductor with a dimension that is substantially the same as that of the AC out-put terminal of a power module can be employed. Design I will be described in greater detail in Example 1.

[0084] In some embodiments (referred to as Design II), the current-carrying conductor with a thinned current trace can be employed. Design II will be described in greater detail in Example 1.

[0085] In some embodiments (referred to as Design III), see for example, FIG. 3, the currentcarrying conductor 12 has two openings, 01 and 02, separating a section, X, of the currentcarrying conductor 12. In such an embodiment, the first location of the first current sensor 16A and the second location of the second current sensor 16B are both in ar e ar ea defined by the section, X, of the current-carrying conductor 12 that is located between the two openings. Since FIG. 3 is a top down view only one side of the current-carrying conductor 12 is shown including one of the two current sensors, e.g., the first current sensor 16A. The current-carrying conductor12 illustrated in FIG. 3 can be referred to as a slitted current-carrying conductor. Slitted currentcarry conductors can be used for PCBs, bus bars or any other type of current-carry conductors. The openings can extend completely or partially through the current-carrying conductor 12. The slitted current-carrying conductor is used in the present disclosure to address the impact of the skin effect and provide a best location to place the two current sensors, e.g., two TMR sensors. The skin effect is the tendency of high frequency current to flow from sides of the currentcarrying conductor 12 rather than uniformly across the cross section of it. Findling a location with consistent magnetic field distribution across a wide range of frequencies poses a significant challenge. The slitted current-carrying conductor design illustrated in FIG. 3 enables a consistent location to place the two sensors without variation in magnetic field distribution across different frequencies. Design III will be described in greater detail in Example 1.

[0086] In some embodiments (referred to as Design IV), wiring bonding is used to connect the sensor output pad to the main terminal of the slitted current-carrying conductor. In Design IV, shield pads (i.e., a dielectric barrier as defined above) can also be portioned beneath each of the sensors. Design IV will be described in greater detail in Example 1. This design address parasitic capacitive issues that can arise during current measurement.

[0087] In addition to providing a contactless current sensing system, the present disclosure also provides a method of measuring current (AC or DC) flowing through a current-carrying conductor. The method includes first providing a contactless current sensing system that includes a current-carrying conductor having a first side and a second side opposite the first side; a first current sensor present in a first location on the first side of the current-carrying conductor; and a second current sensor present in a second location on the second side of the currentcarrying conductor, where the first current sensor and the second current sensor have a substantially same sensitivity and are spaced apart from the current-carrying conductor, and wherein the first location is substantially identical to the second location. The method continues by introducing a first input signal (e.g., first input voltage) into the first current sensor and a second input signal (e.g., second input voltage) into the second current sensor; this introducing step occurs simultaneously. The method further continues by collecting a first out signal (e.g.,first output voltage) from the first current sensor and a second output signal (e.g., second output voltage) from the second current sensor. Next, current within the current-carrying conductor is calculated from the first output single and second output signal. The calculation of the current is typically conducted by a computer that is attached to the contactless current sensing system.

[0088] The present disclosure is now described in greater detail in reference to a two TMR Sensor (TTS) system. To enhance the accuracy of current measurement and mitigate noise- induced errors, a two TMR Sensor (TTS) system, is provided in the present disclosure. Considering the importance of the integrated current measurement in power modules and discrete device level power converters, implementation of the two TMR Sensor (TTS) system for both classes is applicable. The concept of using two sensors to augment sensitivity and counter temperature drift for MR technology has been proposed, See, for example, C. Musuroi, et al. “Low field optimization of a non-contacting high- sensitivity GMR-based DC / AC current sensor” Sensors, vol. 21, no. 7, p. 2564, Apr. 2021, and S. Shao et al. Tunnel magnetoresistance-based short-circuit and over-current protection for IGBT module,” IEEE Trans. Power Electron., vol. 35, no. 10, pp. 10930-10944, Oct. 2020. Although, the same approach can be extended to the TMR technology, the sensor layout adopted in these works exhibits performance limitation for WBG-based power modules and power converter. For instance, C. Musuroi et al. discloses a Ilshaped design with sensor placed on the two legs of the trace in parallel. In this way, the magnetic field in the interior of the U-shaped structure is maximized, thereby increasing the sensitivity of the sensor. However, the U-shaped structure potentially forms a loop antenna (H- field). Since WBG devices are capable of switching at high frequencies, the U-shaped loop can become an effective antenna and interfere with other critical control and power circuitries nearby.

[0089] Similarly, the loop is also susceptible to the external H-fields, thereby corrupting the signal output. Lastly, the footprint of the layout is also large, making it infeasible for integration on the terminal or inside the power module. Moreover, a TMR-based two sensor solution for Si IGBT power module is presented in S. Shao, et al. The design provides a reaction time of 530 ns. However, the signal conditioning structure, developed in this research, is not differential andis highly prone to noise. Further, the sensors are placed relatively far from each other, leading the sensors to be exposed to different temperatures and noisy environments. Therefore, using the differential method will not be beneficial to provide a clean reading from the measurement system.

[0090] Besides this, the validation of the proposed solution is only done at 1 kHz, which is substantially less than the nominal switching frequency of WBG devices.

[0091] The disclosed two TMR Sensor (TTS) system overcomes the limitation of the structures in the above-mentioned literature by placing the two current sensors parallelly (i.e., in a parallel manner) with the current-carrying conductor 12. FIGS. 4A and 4B illustrate a TTS system of the present disclosure including a first TMR sensor 22A and a second TMR sensor 22B located on opposite sides of an electrically conductive metal trace (or bus bar) 20A, e.g., Cu trace as current-carrying conductor 12, or a PCB 20B as the current-carrying conductor 12, respectively, which can be used for power module or power converters. In FIG. 4A, a dielectric barrier is present between the first and second TMR sensors (in the top down view only first dielectric barrier 14A is illustrated). PCB 20B includes alternating layers of dielectric material 21B and electrically conductive material layers 23B. The two TMR sensors (i.e., the first TMR sensor 22A and the second TMR sensor 22B illustrated in FIGS. 4A and 4B) have a substantially same sensitivity and are spaced apart from the current-carrying conductor, and are positioned in locations on opposites sides of the current-carrying conductor that are substantially identical to each other. The current-carrying conductor in between the two TMR sensors, generates a magnetic field, which is sensed by two TMR positioned in opposite side of the current-carrying conductor. The two TMR sensors yield output signals denoted as VS1 and VS2, respectively. Because of the arrangement of the two TMR sensors, VS 1 is equal to the negative of VS2. Further, as the two TMR sensors are in substantial proximity to each other, they are exposed to the same temperature and interference from external fields. Therefore, VS1 and VS2 can be modeled as:VS1 = S1 ■ Bl + 1V1 + K1 ■ 1T1 (eq. 1)VS2 = S2 ■ B2 + 1V2 + K2 ■ 1T2 (eq. 2)

[0092] where SI and S2 are the sensitivities of the TMR sensors in V / mT, 1V1 and 1V2 are the errors of the output voltage due to external fields, Bl and B2 are the magnetic fields the two TMR sensors are exposed to, and KI and K2 are the sensitivities of the sensors due to temperature variation, representing the errors in output voltage generated from temperature fluctuations. The difference of VS1 and VS2, 1VS = VS1 - VS2 can be written as:1VS = (S IB 1) - (S2B2) + (K11T1) - (K21T2) (eq. 3)

[0093] Considering the specific arrangement of the two TMR sensors, VS1 = -VS2 and Bl = -B2. Moreover, placing TMR sensors on two sides of the current-carrying conductor grantees, they are exposed to similar temperatures, which means 1T1 = 1T2. Therefore, equation 3 can be simplified as:1VS = 2S1 ■ Bl (eq. 4)

[0094] Therefore, the noise cancellation effect is maximized by strategically locating the two TMR sensors in similar positions to experience comparable magnetic fields and noise. Besides, rather than increasing the resolution or sensitivity of the current measurement system, the two sensor system, partially the two TMR sensor system, reduces the impact of temperature fluctuations on sensor performance, as errors induced by temperature sensitivity are also mitigated in the current measurement system output.

[0095] The two TMR sensor system of the present application can be extended to both WBG- based power modules and converters. The following examples present the case studies of implementing the two TMR sensor system of the present application for both applications. The sensor used for case studies is CT100 from Crocus (See, Crocus Technology. (Sep. 2021).CT100 Current Sensor Datasheet, Rev. 2.0. [Online], Available: https: / / crocus- technology.com / wpcontent / uploads / 2021 / 09 / CT100-Data-Sheet-Rev2.0.pdf). CT100, similar tothe other MR sensors, generates linear output voltage with a changing magnetic field (in the magnetic field in the range of ±50 mT). The characteristics of the sensor is summarized in TableI.

[0096] Table I: Crocus CT 100 TMR sensor Specification

[0097] Examples have been set forth below for the purpose of further illustrating the present disclosure. The scope of the present disclosure is not limited to any of the examples set forth herein.

[0098] EXAMPLE 1. TTS SOLUTION FOR A DOUBLE-SIDED COOLED SIC POWER MODULE

[0099] In this example, a TTS system in accordance with the present disclosure was used in a double-sided cooled SiC power module. The double-sided cooled SiC power module used in this example was developed by A. I. Emon, et al. A double-sided cooled split-phase SiC power module with fuzz button interposer,” IEEE J. Emerg. Sei. Topics Power Electron., vol. 11, no. 5, pp. 4918-4928, Oct. 2023. The TMR sensors used in this example are two Crocus CT100 TMR sensors as defined above. A first step in designing the TTS solution for this example, was to decide the placement of the sensors. The TTS system can either be embedded inside the module on the direct bonded copper (DBC) or on the AC output terminal. Considering the complicated magnetic and electric fields generated by high di / dt and dv / dt transients of SiC switching devices and the small area inside the power module, placing the sensors on the DBC board can cause errors due to coupling and interference. Moreover, a general solution that can implemented onany power module design without changing the layout of the module is more favorable. Therefore, the most suitable option was to place the sensors on two sides of the AC terminal of the power module. In this location, both sensors will be exposed to similar external magnetic fields and temperature variations. Hence, the differential current measurement solution, TTS, can cancel noise and temperature variation errors. FIG. 5A presents a conceptual drawing for the power module with the proposed TTS system, attached to the output AC terminal. The power module was built on split-output configuration and has two AC terminals which are shorted to form a simple half-bridge configuration with a single AC output terminal. The TTS system of the present disclosure can be used for any configuration of power module or converter.Therefore, to ensure that TTS system of the present application is applicable for both unipolar and bipolar currents, bidirectional TMR sensors (i.e., two Crocus CT100 TMR sensors) were selected for this example. For testing the sensor, a full-bridge testbed was developed. The test schematic is depicted in FIG. 5B and includes a full bridge formed by coupling the power module (half-bridge) to another off-the-shelf half-bridge evaluation board KIT-CRD-8FF90P from Wolfspeed.

[0100] The gate drivers for the module and evaluation board were driven by Texas Instrument F28379D DSP. Further, the sensors were supplied from an isolated 5-V power supply. The output voltage of the sensors was measured using Tektronix TDP1500 differential probe. FIG. 6 shows a schematic of the test setup to verify the TTS system. For the test current, a square pulse (resistive load) was chosen over the triangular’ waveform (inductive load). Generally, prior ail processes predominantly employ a triangular’ waveform for testing sensors. However, the di / dt of a triangular waveform is considerably less than that of a square pulse and there is no constant de (B field) interval. As WBG devices can handle high di / dt, it is also imperative to test and benchmark the sensor’s performance with test current having high di / dt transients. The square pulse provides both fast rising and falling di / dt transients and constant de interval and is, therefore, more realistic than a triangular waveform.

[0101] The accurate TTS current measurement system was achieved systematically by improving the layout in four stages (Designs I-IV). Each design stage targets and improves various issues accompanied by high-frequency current measurement.A. Design I

[0102] The first design, Design I, consists of full wide current-carrying PCB with dimensions kept the same as that of the AC output terminal of the in-house designed power module. The two sensors (CT100) were placed on both sides of the PCB, See for example, FIG. 7. The main current-carrying trace included four PCB layers; each 6 oz thick to meet the current-carrying requirement. The sensor supply and output traces were routed on extra layers on both sides and brought out at one corner. FIG. 8 illustrates the output of a single sensor from the TTS system and the current probe (Tektronix TCP0030A) for 50 A peak-to-peak 70 kHz current pulses. This test evaluated each sensor’s performance and impact on the TTS system. While the current measured by the current probe showed some deviations from a perfect square-wave shape due to parasitic inductance, the current sensor output displays a similar’ waveform corrupted with spikes at the switching transition intervals. These spikes are caused by capacitive coupling between the main current-carrying trace and the sensor output traces. The high dv / dt producing during the switching transient causes noise to couple to the sensor output.B . Design II

[0103] To solely analyze and troubleshoot the impact of capacitive coupling in Design I, a PCB terminal with a thinned current trace strategically passing underneath the sensor was developed as depicted in FIG. 9. This modification reduced the coupling between the main current-carrying trace and the sensor output trace, as there was minimum overlap between the main currentcarrying trace and the sensor supply and output traces. The developed prototype was tested with a current pulse of 70 kHz and 30 A peak and the test results are presented in FIG. 10. As can be observed from the test results, thinning the current-carrying trace reduces the coupling betweenthe main trace and sensor pads. However, thinning the trace brings concerns about the heating of the trace and, hence is not a feasible solution.C. Design III

[0104] Although Design II provides better results, but the limited current-carrying capacity overshadows its effectiveness. This limitation can be addressed by slitting Design I in the center to create a thin trace and place a sensor on this thin trace in the center of the terminal. The proposed concept PCB layout is illustrated in FIG. 11. The current-carrying trace is divided into three sections, and the sensor is placed in the middle part. The thin trace carries a portion of the primary current, which will be sensed by the sensors above the trace on the two sides of the board. By slitting the main current-carrying trace, this design approach not only helps decrease coupling with the thin trace beneath the sensor but it also provides freedom in selecting current sensors with low B-field sense ratings, even for high current applications.

[0105] Another approach to implementing a slitted current trace involved dividing the currentcarrying trace into two sections: a thin sensing trace and a thick side trace. However, in this configuration, the proximity effect causes the maximum magnetic field to shift toward one side. Consequently, this phenomenon disrupts the sensor’s exposure to the maximum magnetic field when placed in the middle of the thin trace, thereby compromising its performance. Thus, opting for a design with the thin trace positioned in the middle and two thick traces on the sides emerges as the optimal solution.

[0106] Further, the thin- slitted trace also provided the benefit of having uniform magnetic field distribution over a wide frequency range. For a wide full trace (Design I), one challenge was the concentration of the magnetic field around the corner due to the skin effect at high frequencies. Consequently, if the sensor is placed in the middle, its sensitivity to detecting high-frequency current decreases. Hence, optimal sensor placement is required, which is mainly performed using finite-element analysis (FEA). However, the slitted trace was thin enough such that the variation in magnetic field with frequency is little due to the lower cross section area of the thintrace. To validate this, a simulation was performed in ANSYS Maxwell using eddy current solver to determine and compare the magnetic field component along the x-axis Bx (sensing axis of CT100) for 50 A current for various frequencies for Design I and Design III PCB layouts.

[0107] The results are summarized in FIGS. 12A-12B; FIG. 12A provide the result for Design I, while FIG. 12B provide the result for Design III. As evident from the simulation results in FIGS. 12A-12B, the concentration of magnetic field density along the corners increased with frequency for Design I due to the skin effect. This phenomenon, owing to the skin effect, caused error in sensor reading for high-frequency content and the underlying sensor output signal did not truly represent the frequency spectrum of the excitation current. This limitation was addressed in the thin-slitted trace design, where the magnetic field around the thin trace (highlighted region) stayed constant over the wide frequency range (10 kHz- 100 MHz), minimizing the impact of skin effect. Hence, the sensor placement on the thin trace for Design III is always optimal.

[0108] As one approach to substituting the design involved splitting the current-carrying terminal into two thick traces on both sides, with one side featuring a thicker trace and the thin sensing trace positioned on the other. However, in such a configuration, the proximity effect caused the maximum magnetic field to shift toward one side. Consequently, this phenomenon disrupted the sensor’s exposure to the maximum magnetic field when situated in the middle of the thin trace, thereby compromising its performance. Thus, opting for a design with the thin trace positioned in the middle and two thick traces on the sides emerges as the optimal solution.

[0109] The developed prototype was tested using the experimental setup in FIG. 6. The tests results for 70 kHz, 50 A are presented in FIG. 13. Compared with Design I, the final sensor output was cleaner and captures the pulse waveform shape. However, voltage spikes due to capacitive coupling appear at the switching transition. The root cause of these spikes was the parasitic capacitance between the main trace and the sensor pins / pad and sensor output traces routed to the connection header over the main trace. This conclusion served as the motivation for developing Design IV with lower parasitic capacitance.D. Design IV

[0110] The traces connecting the sensor output to the connection header in Design III were capacitively coupled to the main slitted current-carrying trace, leading to voltage spikes during the switching transient. To lower the coupling, these connections were brought out to the header with sufficient separation between the main and the sensor output traces. For this purpose, wire bonding the connection between the sensor output pad and the traces was feasible. Further, another source of parasitic capacitance was the coupling between the sensor pads and the main current-carrying trace, which can be reduced by adding shield pads beneath the pads of the sensor. These modifications constitute Design IV.

[0111] Further, to investigate the impact of proposed modifications on coupling effect and accuracy of current measurement, parasitic extraction was performed using ANSYS Q3D to estimate the capacitance between the pads and the main trace. Table 2 summarizes the results.

[0112] Table 2: Estimated Parasitic Capacitance using ANSYS Q3D

[0113] Placing shields beneath the sensor pads helped to decrease the capacitance in Design III. However, eliminating the sensor output traces over the main current-carrying trace had a prominent influence on capacitance reduction (99.12%). Based on the limits of the application, the sensor traces can be routed externally through wire bonding. FIG. 14A illustrates a Design IV PCB layout. The developed prototype for Design IV is depicted in FIG. 14B. The sensor pads were connected to the headers using a 12-mil thick wire bond with a height of 1.5 mm above the PCB surface. For practical application, the wire bonds can be encapsulated to protect them against getting detached or displaced. The prototype was tested on the setup in FIG. 5 with the same 70 kHz, 50 A current pulse. The test results are presented in FIG. 15. Compared withDesign III, the extent of voltage spikes is reduced considerably, justifying the efficacy of the proposed TTS solution.

[0114] Despite using the wire bonds, small spikes still appeared during the switching transition. The cause of these spikes was the intrinsic capacitive coupling between the sensor package (SOT-26) pins and the current-carrying trace, which is uncontrollable. The extent of these spikes can be reduced by using a low-pass filter. The low-pass filter needs to be designed to achieve adequate bandwidth and fast response time. Further, although wire bonds proved effective in lowering the capacitive coupling between the main trace and the sensor output traces, a concern related to distortion in the measurement due to the wire bonds’ parasitic inductance arises. Nevertheless, it is worth noting that the wire bonds serve as replacements for traces on the PCB in prior Designs I, II, and III. The traces on the PCB also introduced parasitic inductance similar to wire bonds. Moreover, the output of each TMR sensor was differential (two signal outputs Vout+ and Vout- with respect to the ground). In a differential configuration with the same length of both signal lines, the impact of parasitics was suppressed as both lines have the same distortion or noise, which was eliminated in the differential measurement. The length of wire bonds for each sensor output pair (Vout-i- and Vout-) was kept the same in the developed prototype for Design IV (See, FIG. 14B). As a result, the noise or distortion due to wire bond parasitics was the same in both lines for each TMR sensor and was canceled.

[0115] Lastly, a two- stage signal conditioning circuitry can be used to amplify the output of sensors and their difference to make it compatible with a commercially available ADC having 3.3 or 5 V reference. The schematic and design of the signal conditioning circuitry will be discussed in Example 2 focusing on the TTS application in a 75-kVA SiC-based power converter.

[0116] Referring back to Design III, the slitted current trace needs to be designed properly to prevent saturation of the TMR sensor due to current Isflowing through the thin trace. Theoretically, Is, assuming uniform cross section of the PCB layer, depends on the width of thetrace ws and the width of the thicker trace wc on each side. The widths ws and wc are in millimeters. The expression for Is, based on current division rule, can be written as:Is =(ws / ws + 2wc)I (eq. 5)

[0117] where I represents the total current flowing out of the AC terminal of the module. After computing the value of Is, the next step is to estimate the magnetic field density across the sensing trace. Considering the magnetic axis orientation of FIG. 4A, the relevant magnetic field component is along the x-axis Bx. Bxin mT can be calculated using equation 6 below, derived from the Biot-Savart law, h, in millimeter, represents the vertical distance of the sensor above the sensing trace:

[0118] In the linear range, the output voltage VS of each sensor in millivolt is related to Bxas:Vs = SBx(eq. 7) where S is the sensitivity of the sensor in mV / V / mT, specified in the datasheet for a specific supply voltage. Combining equations 5-7, Vs is related to the total current I as:

[0119] It is pertinent to mention that equation 8 is applicable only when Bxis within the linear range of the sensor. Therefore, from the design perspective, ws can be set to the maximum for the given sensor package, leaving width of thicker trace wc as the only design parameter, wc needs to be adjusted till the maximum value of Bxfor the maximum value of 1 falls within the linear range.

[0120] Finally, assuming symmetry of the two sensors SI and S2, the TTS output 1VS = VS1 - VS2 can be expressed as:[01211 EXAMPLE 2: TTS SOLUTION FOR A 75-KVa SiC-BASED POWER CONVERTER

[0122] Power converters include passive components such as inductors, capacitors, and power semiconductor devices, which can be in discrete or module packages. The components degrade over time due to switching actions. Hence, it is imperative to promptly monitor these components’ health to prevent major system failures. Several approaches have been proposed in the literature that rely on measuring the converter current and voltage state variables to determine the values of health precursors. For instance, a DT-bascd health monitoring approach is developed for monitoring health of dc-dc stage of an in-house designed 75-kVA SiC -based power stage. The performance of the DT approach relies on accurate measurement of current and voltages in the megahertz range. Conventionally, shunt and hall-based sensors are predominantly used for sensing current for high -power applications. However, they suffer from having low bandwidth (hundreds of kilohertz) and conduction losses. For instance, in shunt measurement, the resistor must carry the rated current, generating heat, and if the resistor breaks down, the connection between the converter and the load breaks, potentially causing system instability or breakdown. Similarly, Hall effect sensors suffer from large footprint, magnetic saturation, and high-temperature sensitivity.

[0123] For an in-house developed converter (See, for example, A. B. Mirza et al., Hardware design and implementation of a 75 kVA 3-D integrated intelligent power stage, “in Proc. IEEE Appl. Power Electron. Conf. Expo. (APEC), Mar. 2023, pp. 977-983, a shunt-based current sensor is used for measuring inductor current of the dc-dc stage. The sensor included a thin-film resistor, with lower parasitics, coupled with an isolated AMC1302x series optically isolated differential amplifier from Texas Instruments, based on delta-sigma modulator, with a bandwidth up to 800 kHz. However, due to parasitics of the resistor, the shunt measurement results have distortion and variable time delay. To counter this, digital filtering (Savitzky- Golay) filter and synchronization algorithms are used, which increases the computational complexity and slow the DT response time.

[0124] FIGS. 16A-16B show the structure of the converter, developed in the A. B. Mirza ct al. publication mentioned above. The power stage of the intelligent power stage (IPS) was electro- mechanically-thermally co-designed and 3-D packaged on a cylindrical hole-based heat sink, providing optimal thermal performance and mechanical stability. The dc-dc stage was split into two parts, each comprising two phases, placed on the side of the heat sink. The dc-ac stage was arranged on the top side of the heat sink, with the dc-link capacitor PCB sitting on top of it. The dc-dc stage boost inductors are placed on the top with side bus bars connecting the inductors with the boards.

[0125] The proposed TTS solution in FIG. 4B is low-cost and noninvasive and has the potential to alleviate the performance limitation of the shunt-based measurement system. The sensors can be placed on the top and bottom side of the bus bar and their output can be subtracted to yield the final sensor output. FIG. 17 shows the converter bus bar for channel 1 of the dc-dc stage with TMR sensor PCBs attached on both sides. The TMR sensor employed was CT100 from Crocus (see, Table 1). The sensor pins on the board were shielded using the same approach of Design IV. Further, the PCB was 0.8 mm thick and served as an insulator between the sensor and the current-carrying bus bar. For amplifying the output of the sensor, a signal conditioning board using Texas Instruments INA821 Instrumentation Amplifier (IA) is developed for concept validation. FIG. 18 depicts a schematic of the bus bar with TTS solution. The signal conditioning includes two-stage amplification. In stage 1, the individual output of each of sensor is amplified. As the TMR sensor output and IA input are fully differential, the CM noise was canceled to a greater extent in the first amplification stage. The amplified output in stage 1 was then fed to the final IA, which subtracts the sensor outputs to yield the final single-ended output with an offset of VCC / 2. Any residual unattenuated CM noise after the first amplification stage was further attenuated in the second stage. The final sensor output can be expressed as:VS = G2G1(VS1 - VS2) (eq. 10)

[0126] where G1 and G2 are the gains of the first and second stage, respectively. The peak inductor current for the dc-dc stage was 37.5 A for which the sensor produced an output of 6 mV with 5 V supply for the given thickness (2.5 mm) and width (15 mm) of the bus bar. Therefore, to amplify the output substantially, G1 and G2 were both set to 9, providing a total gain of 81 with 1 MHz bandwidth. The developed bus bar with TTS solution was assembled back to the converter and continuous standalone tests for the dc-dc stage are performed at 500 V de and 3.5 kW (See, FIG. 19 which illustrates a schematic of an experiment setup used in this embodiment of the present disclosure). The test conditions are summarized in Table 3.

[0127] Table 3: DC-DC Stage Test Conditions

[0128] The TMR sensors and the signal condition circuitry was powered from an isolated 5-V power supply on the control card. For benchmarking, the shunt current sensor, originally developed for the converter, was also connected in series with the bus bar. FIG. 20 shows the excerpt of the measured current waveforms. The first trace from top was the current waveform captured by the hall-based current probe (Tektronix TCP0030A). The second trace was the final output of the TTS solution, followed by the output of shunt sensor in the third trace. Both TTS and shunt sensor were able to capture the triangular' shape of the inductor current waveform.

[0129] However, the output of the shunt sensor is superimposed with high-frequency noise. In contrast, the output of TTS solution was clean and follows the current probe results, justifying the efficacy of the proposed approach.

[0130] While the present disclosure has been particularly shown and described with respect to preferred embodiments thereof, it will be understood by those skilled in the art that the foregoingand other changes in forms and details may be made without departing from the spirit and scope of the present disclosure. It is therefore intended that the present disclosure not be limited to the exact forms and details described and illustrated, but fall within the scope of the appended claims.

Claims

CLAIMSWhat is claimed is:

1. A contactless current sensing system comprising: a current-carrying conductor having a first side and a second side opposite the first side; a first current sensor present in a first location on the first side of the current-carrying conductor; and a second current sensor present in a second location on the second side of the currentcarrying conductor, wherein the first current sensor and the second current sensor have a substantially same sensitivity and are spaced apart from the current-carrying conductor, and wherein the first location is substantially identical to the second location.

2. The contactless current sensing system of Claim 1, further comprising a first dielectric barrier separating the first current sensor from the current-carrying conductor, and a second dielectric banner separating the second current sensor from the current-carrying conductor.

3. The contactless current sensing system of Claim 1, wherein the current-carrying conductor is a bus bar or a component of a printed circuit board.

4. The contactless current sensing system of Claim 1, wherein the first current sensor and the second current sensor are both magnetoresistance sensors.

5. The contactless current sensing system of Claim 4, wherein the magnetoresistance sensors comprise anisotropic magnetoresistance sensors, giant magnetoresistance sensors or tunneling magnetoresistance sensors.

6. The contactless current sensing system of Claim 1, wherein the first current sensor and the second current sensor are both Hall sensors, Rogowski sensors or shunt sensors.

7. The contactless current sensing system of Claim 1 , wherein the first current sensor and the second current sensor arc both tunneling magnctorcsistancc sensors.

8. The contactless current sensing system of Claim 1, wherein the current-carrying conductor has two openings separating a section of the current-carrying conductor, and the first location of the first current sensor and the second location of the second current sensor are both in an area defined by the section of the current-carrying conductor that is located between the two openings.

9. The contactless current sensing system of Claim 1, further comprising a first voltage input source electrically connected to a first portion of the first current sensor and a first voltage output source electrically connect to a second portion of the first current sensor, and a second voltage input source electrically connected to a first portion of the second current sensor and a second voltage output source electrically connect to a second portion of the second current sensor.

10. The contactless current sensing system of Claim 9, wherein the first voltage input source, the first voltage output source, the second voltage input source and the second voltage output source are spaced apart from the current-carrying conductor.

11. The contactless current sensing system of Claim 9, further comprising a first wire electrically connecting the first voltage input source to the first portion of the first current sensor, a second wire electrically connecting the first voltage output source to the second portion of the first current sensor, a third wire electrically connecting the second voltage input source to the first portion of the second current sensor, a fourth wire electrically connecting the second voltage output source to the second portion of the second current sensor, wherein the first wire, the second wire, the third wire and the fourth wire are spaced apart from the current-carrying conductor.

12. The contactless current sensing system of Claim 1, wherein the current-carrying conductor is a component of a power module.

13. The contactless current sensing system of Claim 1, wherein the current-carrying conductor is a component of a power converter with discrete devices.

14. A contactless current sensing system comprising: a current-carrying conductor having a first side and a second side opposite the first side, wherein the current-carrying conductor has two openings separating a section of the currentcarrying conductor; a first tunneling magnetoresistance (TMR) sensor present in a first location on the first side of the current-carrying conductor; and a second TMR sensor present in a second location on the second side of the current-carrying conductor, wherein the first TMR sensor and the second TMR sensor have a substantially same sensitivity and are spaced apart from the current-carrying conductor, and wherein the first location is substantially identical to the second location, and wherein the first location of the first TMR sensor and the second location of the second TMR sensor are both in an area defined by the section of the current-carrying conductor that is located between the two openings.

15. The contactless current sensing system of Claim 14, further comprising a first dielectric barrier separating the first TMR sensor from the current-carrying conductor, and a second dielectric barrier separating the second TMR sensor from the current-carrying conductor.

16. The contactless current sensing system of Claim 14, further comprising a first voltage input source electrically connected to a first portion of the first TMR sensor and a first voltage output source electrically connect to a second portion of the first TMR sensor, and a second voltage input source electrically connected to a first portion of the second TMR sensor and a second voltage output source electrically connect to a second portion of the second TMR sensor.

17. The contactless current sensing system of Claim 16, wherein the first voltage input source, the first voltage output source, the second voltage input source and the second voltage output source are spaced apart from the current-carrying conductor.

18. The contactless current sensing system of Claim 16, further comprising a first wire electrically connecting the first voltage input source to the first portion of the first TMR sensor, a second wire electrically connecting the first voltage output source to the second portion of the first TMR sensor, a third wire electrically connecting the second voltage input source to the first portion of the second TMR sensor, a fourth wire electrically connecting the second voltage output source to the second portion of the second TMR sensor, wherein the first wire, the second wire, the third wire and the fourth wire are spaced apart from the current-carrying conductor.

19. The contactless current sensing system of Claim 14, wherein the current-carrying conductor is a component of a power module.

20. The contactless current sensing system of Claim 14, wherein the current-carrying conductor is a component of a power converter with discrete devices.

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