A switching power supply circuit

Abstract

The present invention discloses a switching power supply circuit, comprising a high-frequency switching network, a transformer, a rectifier module, a DC-DC converter, and a control circuit. The high-frequency switching network directly converts a low-frequency AC input voltage into a high-frequency AC signal. The high-frequency AC signal is then converted into a DC signal via the transformer and rectifier module and output to the DC-DC converter. The DC-DC converter then steps up or down the DC signal and outputs it to charge the battery. The control circuit controls the operating state of the DC-DC converter based on charging requirements to adjust the charging current so that the average charging current meets the charging requirements. The switching power supply circuit of the present invention eliminates the need for a rectifier bridge and directly converts the AC signal into a DC signal. This reduces the number of network stages and energy loss, improving efficiency, and charges the battery with an intermittent current, thereby increasing the service life and safety of the switching power supply circuit.

Description

Technical Field

[0001] The present invention relates to the field of power electronics, and more particularly to a switching power supply circuit. Background Art

[0002] In daily life, charging of mobile terminals (such as smart phones) is usually done through a switching power supply circuit. Many semiconductor electronic devices require a switching power supply circuit that converts AC power into DC power so that the DC power required for operation can be obtained from the AC mains network. In most cases, in order to meet safety requirements, it is necessary to isolate the load end and the power end. The switching power supply circuit in the prior art is as follows: Figure 1 As shown, it includes a rectifier bridge, a DC-DC converter, and a transformer. The rectifier bridge receives an AC input voltage, and its output is connected to the input of the DC-DC converter. The output of the DC-DC converter is coupled to the primary winding of the transformer, and the secondary winding of the transformer is coupled to the load. The rectifier bridge converts the received AC signal into a DC signal, and then transfers the energy to the output through the DC-DC converter and transformer. In this way, the input AC power is converted into a stable low-voltage DC power suitable for the needs of the mobile terminal, which is provided to the power management device and battery of the mobile terminal, thereby achieving charging of the mobile terminal.

[0003] Due to the presence of a rectifier bridge in the switching power supply circuit of the prior art, a large number of switching tubes are required, resulting in large energy losses and low efficiency, which is more obvious when the AC input voltage is low. Summary of the Invention

[0004] In view of this, the present invention proposes a switching power supply circuit that converts AC signals into DC signals without the need for a rectifier bridge, solving the technical problems in the prior art caused by the presence of a rectifier bridge, such as the need for a large number of switching tubes, large energy loss, and low efficiency.

[0005] The present invention provides a switching power supply circuit for charging a battery, comprising: a high-frequency switching network, a transformer, a rectifier module, a DC-DC converter and a control circuit.

[0006] The high-frequency switching network includes an AC input terminal for receiving a low-frequency AC input voltage, a first switching tube, a second switching tube, a third switching tube, a fourth switching tube, and an energy storage module, wherein the first end of the first switching tube and the first end of the second switching tube are respectively connected to two ports of the AC input terminal, the second end of the first switching tube is connected to the first end of the third switching tube, the second end of the second switching tube is connected to the first end of the fourth switching tube, and the second end of the third switching tube is connected to the second end of the fourth switching tube; the energy storage module and the primary winding of the transformer are connected between a first node and a second node, wherein the first node is a common end of the first switching tube and the third switching tube, and the second node is a common end of the second switching tube and the fourth switching tube;

[0007] The secondary winding of the transformer is connected to the input end of the rectifier module, the output end of the rectifier module is connected to the DC-DC converter, and the DC-DC converter outputs a charging current;

[0008] The control circuit is configured to control the working state of the DC-DC converter according to the charging requirement to adjust the charging current so that the average value of the charging current meets the charging requirement.

[0009] Preferably, the energy storage module includes a first inductor and a first capacitor, and the first inductor, the first capacitor and the primary winding are connected in series and connected between the first node and the second node.

[0010] Preferably, the high-frequency switching network operates in an LLC resonance state.

[0011] Preferably, during the positive half cycle of the low-frequency AC input voltage, the first switch tube and the third switch tube are always turned on, and the second switch tube and the fourth switch tube are alternately turned on in a high-frequency PWM manner; during the negative half cycle of the low-frequency AC input voltage, the second switch tube and the fourth switch tube are always turned on, and the first switch tube and the third switch tube are alternately turned on in a high-frequency PWM manner.

[0012] Preferably, the control circuit includes a second control signal generating circuit, which is used to generate a second control signal according to the charging requirement and the charging current; and the control circuit is configured to adjust the charging current according to the second control signal.

[0013] Preferably, the control circuit is configured to control the working state of the DC-DC converter according to the charging requirements to adjust the charging current, so that the charging current is intermittent, and at least when the absolute value of the low-frequency AC input voltage is lower than a predetermined threshold, the value of the charging current is zero.

[0014] Preferably, the frequency of the charging current is configured to be approximately twice the frequency of the low-frequency AC input voltage.

[0015] Preferably, the value of the interval in which the charging current is not zero is fixed.

[0016] Preferably, the value of the interval in which the charging current is not zero varies.

[0017] Preferably, the DC-DC converter is a resonant converter, and the control circuit is configured to adjust the operating frequency of the DC-DC converter to adjust the charging current.

[0018] Preferably, the DC-DC converter is configured to operate with a fixed duty cycle at the operating frequency.

[0019] Preferably, the control circuit includes:

[0020] a first control signal generating circuit, configured to generate a first control signal according to the low-frequency AC input voltage and a predetermined threshold;

[0021] a second control signal generating circuit, configured to generate a second control signal according to a charging requirement and the charging current; and

[0022] a logic circuit, configured to adjust the charging current according to the first control signal and the second control signal;

[0023] The logic circuit is configured to control the DC-DC converter to operate at a fixed duty cycle according to the second control signal in response to the first control signal being valid, and to control the DC-DC converter not to operate in response to the first control signal being invalid.

[0024] Preferably, the first control signal generating circuit includes:

[0025] The comparison circuit is configured to compare the absolute value of the sampling signal of the low-frequency AC input voltage with a predetermined threshold value to obtain the first control signal.

[0026] Preferably, the second control signal generating circuit includes:

[0027] a first error amplifier, configured to compare the sampling signal of the charging current with a current reference signal to obtain a first error signal, wherein the current reference signal is used to represent a charging requirement;

[0028] a second error amplifier, configured to compare the sampling signal of the charging current with the first error signal to obtain a second error signal;

[0029] a compensation circuit, configured to generate a compensation signal according to the second error signal; and

[0030] A driving circuit is configured to generate the second control signal according to the compensation signal.

[0031] Preferably, the DC-DC converter is a resonant converter, and the driving circuit generates the second control signal according to the compensation signal and the resonant current sampling signal.

[0032] Preferably, the driving circuit includes:

[0033] a second comparator, configured to obtain a set signal according to the resonant current sampling signal and the compensation signal;

[0034] a third comparator, configured to obtain a reset signal according to the resonant current sampling signal and the compensation signal; and

[0035] A trigger is configured to generate the second control signal according to the set signal and the reset signal.

[0036] Preferably, the DC-DC converter is a resonant converter, and the driving circuit generates the second control signal according to the compensation signal and the resonant voltage sampling signal.

[0037] Preferably, the driving circuit includes:

[0038] a second comparator, configured to obtain a set signal according to the resonant voltage sampling signal and the compensation signal;

[0039] a third comparator, configured to obtain a reset signal according to the resonant voltage sampling signal and the compensation signal; and

[0040] A trigger is configured to generate the second control signal according to the set signal and the reset signal.

[0041] Preferably, the rectifier module is a full-wave rectifier circuit.

[0042] Compared with the prior art, the technical solution of the present invention has the following advantages: The switching power supply circuit of the present invention includes a high-frequency switching network, a transformer, a rectifier module, a DC-DC converter, and a control circuit. The high-frequency switching network directly converts a low-frequency AC input signal into a high-frequency AC signal. The transformer and rectifier module then convert the high-frequency AC signal into a DC signal and output it to the DC-DC converter. The DC-DC converter boosts or bucks the DC signal and outputs it to charge the battery. The control circuit controls the operating state of the DC-DC converter according to charging requirements to adjust the charging current so that the average charging current meets the charging requirements. The switching power supply circuit of the present invention does not require a rectifier bridge and directly converts the AC signal into a DC signal. This reduces the number of switching transistors, reduces the number of network stages and energy loss, and improves efficiency. Furthermore, the switching power supply circuit of the present invention can charge the battery using intermittent current, eliminates the electrolytic capacitor in the switching power supply circuit, reduces the size of the switching power supply circuit, and improves the service life and safety of the switching power supply circuit. BRIEF DESCRIPTION OF THE DRAWINGS

[0043] The above and other objects, features and advantages of the present invention will become more apparent through the following description of the embodiments of the present invention with reference to the accompanying drawings, in which:

[0044] Figure 1 A circuit diagram of a switching power supply circuit in the prior art;

[0045] Figure 2 This is a circuit diagram of a first embodiment of a switching power supply circuit of the present invention;

[0046] Figure 3 This is a working waveform diagram of the high-frequency switching network of the present invention;

[0047] Figure 4 This is a signal waveform diagram of the high-frequency switching network of the present invention;

[0048] Figure 5 2 is a circuit diagram of a second embodiment of a switching power supply circuit of the present invention;

[0049] Figure 6 Schematic diagram of a circuit of a DC-DC converter and a control circuit according to a first embodiment of the present invention;

[0050] Figure 7 This is a working waveform diagram of the switching power supply circuit of the present invention under the control of the control circuit embodiment 1;

[0051] Figure 8 This is a working waveform diagram of the driving circuit of the present invention;

[0052] Figure 9 This is a circuit diagram of a second embodiment of the control circuit of the present invention;

[0053] Figure 10 Schematic diagram of a driving circuit in a second embodiment of a control circuit of the present invention;

[0054] Figure 11 This is a working waveform diagram of the switching power supply circuit of the present invention under the second control circuit embodiment;

[0055] Figure 12 This is a circuit diagram of a third embodiment of the control circuit of the present invention;

[0056] Figure 13 Schematic diagram of a driving circuit in a third embodiment of a control circuit of the present invention;

[0057] Figure 14 This is a working waveform diagram of the switching power supply circuit of the present invention under the control circuit embodiment 3. DETAILED DESCRIPTION

[0058] The present invention is described below based on the following embodiments, but the present invention is not limited to these embodiments. In the detailed description of the present invention below, certain specific details are described in detail. Those skilled in the art can fully understand the present invention without these details. To avoid obscuring the essence of the present invention, well-known methods, processes, procedures, components, and circuits are not described in detail.

[0059] Furthermore, persons of ordinary skill in the art will appreciate that the figures provided herein are for illustration purposes only and are not necessarily drawn to scale.

[0060] At the same time, it should be understood that in the following description, "circuit" refers to a conductive loop composed of at least one element or subcircuit connected electrically or electromagnetically. When an element or circuit is said to be "connected to" another element or an element / circuit is said to be "connected" between two nodes, it can be directly coupled or connected to the other element or there can be intermediate elements. The connection between the elements can be physical, logical, or a combination thereof. Conversely, when an element is said to be "directly coupled to" or "directly connected to" another element, it means that there are no intermediate elements between the two.

[0061] Figure 2 This is a circuit diagram of a first embodiment of a switching power supply circuit. The switching power supply circuit is used to charge a battery and includes: a high-frequency switching network 1, a transformer 2, a rectifier module 3, a DC-DC converter 4, and a control circuit 5.

[0062] The high-frequency switch network 1 includes an AC input terminal for receiving a low-frequency AC input voltage Vin, a first switch tube Q1, a second switch tube Q2, a third switch tube Q3, a fourth switch tube Q4, and an energy storage module. The first end of the first switch tube Q1 and the first end of the second switch tube Q2 are respectively connected to two ports N and L of the AC input terminal, the second end of the first switch tube Q1 is connected to the first end of the third switch tube Q3, the second end of the second switch tube Q2 is connected to the first end of the fourth switch tube Q4, and the second end of the third switch tube Q3 is connected to the second end of the fourth switch tube Q4. The energy storage module and the primary winding L2 of the transformer 2 are connected between a first node and a second node, wherein the first node is a common end of the first switch tube Q1 and the third switch tube Q3, and the second node is a common end of the second switch tube Q2 and the fourth switch tube Q4.

[0063] The secondary winding of transformer 2 is connected to the input of rectifier module 3, and the output of rectifier module 3 is connected to the input of DC-DC converter 4. DC-DC converter 4 is configured to receive the output voltage Vin1 of the rectifier module and output a charging current Io to charge the battery. DC-DC converter 4 is a resonant converter.

[0064] The control circuit 5 is configured to control the working state of the DC-DC converter according to the charging requirement Iref to adjust the charging current Io so that the average value of the charging current meets the charging requirement.

[0065] Furthermore, the energy storage module includes a first inductor L1 and a first capacitor C1. The first inductor L1, the first capacitor C1 and the primary winding L2 are connected in series and connected between the first node and the second node.

[0066] Furthermore, the transformer 2 includes two secondary windings L3 and L4, which are connected in series. The common end of the secondary winding L3 and the secondary winding L4 is the middle end of the secondary winding. The other end of the secondary winding L3 is the first end of the secondary winding, and the other end of the secondary winding L4 is the second end of the secondary winding. In other embodiments, the transformer 2 includes a secondary winding with a center tap, wherein the center tap is the middle end of the secondary winding, and the other two ends are the first end and the second end of the secondary winding, respectively.

[0067] Furthermore, the rectifier module 3 is a full-wave rectifier circuit, which includes a first rectifier tube S1 and a second rectifier tube S2. The first end of the first rectifier tube S1 is connected to the first end of the secondary winding L3, and the second end of the first rectifier tube S1 serves as the high potential end of the output voltage. The first end of the second rectifier tube S2 is connected to the second end of the secondary winding, and the second end of the second rectifier tube S2 is connected to the second end of the first rectifier tube S1. The middle end of the secondary winding serves as the low potential end of the output voltage. The rectifier tube in the present invention can be various existing electrically controllable switching devices, such as metal oxide semiconductor field effect transistors (MOSFETs), bipolar junction transistors (BJTs), or insulated gate bipolar transistors (IGBTs), without limitation. The primary side is used to control the opening or closing of the first and second rectifier tubes.

[0068] Furthermore, the control circuit 5 controls the working state of the DC-DC converter 4 according to the sampling signal Vfb of the charging current Io and the charging requirement Iref, so that the charging current Io meets the charging requirement.

[0069] Figure 3 : This is a working waveform diagram of the high-frequency switching network of the present invention. When the low-frequency AC input voltage Vin is in a positive half-cycle, that is, the voltage at the L terminal is greater than that at the N terminal, the first switch Q1 and the third switch Q3 are always on, and the second switch Q2 and the fourth switch Q4 are alternately turned on in a high-frequency PWM manner. At this time, the first capacitor C1, the first inductor L1, and the primary winding L2 in the second embodiment form an LLC resonant circuit, and the first switch Q1, the second switch Q2, the third switch Q3, and the fourth switch Q4 operate in a resonant state. When the low-frequency AC input voltage Vin is in a negative half-cycle, that is, the voltage at the N terminal is greater than that at the L terminal, the second switch Q2 and the fourth switch Q4 are always on, and the first switch Q1 and the third switch Q3 are alternately turned on in a high-frequency PWM manner. At this time, the first capacitor C1, the first inductor L1, and the primary winding L2 in the second embodiment form an LLC resonant circuit, and the first switch Q1, the second switch Q2, the third switch Q3, and the fourth switch Q4 operate in a resonant state, thereby achieving high-frequency chopping. The frequency of the high-frequency PWM is much greater than the frequency of the low-frequency AC input voltage. The high-frequency switch network of this embodiment operates in a fixed frequency state.

[0070] When the second switch tube Q2 is turned on and the fourth switch tube Q4 is turned off, the low-frequency AC input voltage Vin charges the energy storage module and the primary winding. The polarity of the voltage of the primary winding and the low-frequency AC input voltage Vin do not necessarily correspond, and may be the same or opposite. When the second switch tube Q2 is turned off and the fourth switch tube Q4 is turned on, the energy storage module discharges the primary winding. When the first switch tube Q1 is turned on and the third switch tube Q3 is turned off, the low-frequency AC input voltage Vin charges the energy storage module and the primary winding. The polarity of the voltage of the primary winding and the low-frequency AC input voltage Vin do not necessarily correspond, and may be the same or opposite. When the first switch tube Q1 is turned off and the third switch tube Q3 is turned on, the energy storage module discharges the primary winding.

[0071] Figure 4 is a signal waveform diagram of the high-frequency switching network of the present invention; Figure 4 As shown, when the low-frequency AC input voltage Vin is in the positive half cycle, the second switch tube Q2 and the fourth switch tube Q4 are alternately turned on in a high-frequency PWM manner; when the low-frequency AC input voltage Vin is in the negative half cycle, the first switch tube Q1 and the third switch tube Q3 are alternately turned on in a high-frequency PWM manner.

[0072] It should be noted that complementary conduction is a special case of alternating conduction. When complementary conduction is performed, the efficiency is the highest. However, in order to prevent a transient short circuit caused by a delay in turning off the switch tube, a dead time is inserted between the state switching of the second switch tube Q2 and the fourth switch tube Q4 or the first switch tube Q1 and the third switch tube Q3.

[0073] The switch tube described in the invention can adopt various existing electrically controllable switch devices, such as metal oxide semiconductor field effect transistor (MOSFET), bipolar junction transistor (BJT), or insulated gate bipolar transistor (IGBT), without limitation.

[0074] Figure 2 In the embodiment, the switching power supply circuit further includes a capacitor C2 and a capacitor C3, the capacitor C2 is connected to the output end of the rectifier module 3, and is used to filter the output signal of the rectifier module 3, and the capacitor C3 is connected to the output end of the DC-DC converter 4, and is used to filter the output signal of the DC-DC converter 4.

[0075] The high-frequency switching network 1, transformer 2, and rectifier module 3 in this embodiment directly convert AC signals into DC signals, eliminating the need for a rectifier bridge and utilizing fewer switching transistors. This reduces the number of network stages and energy loss, improving efficiency and resolving the technical issues inherent in prior art systems, which often require a large number of switching transistors, experience significant energy losses, and exhibit low efficiency due to the presence of a rectifier bridge. The switching power supply circuit in this embodiment is used to charge a battery. Based on the charging requirement Iref, the operating state of the DC-DC converter is controlled to adjust the charging current Io so that its average value meets the required charging current. The output current of the DC-DC converter 4 in this embodiment is continuous.

[0076] Under normal circumstances, when AC power is supplied, most devices cannot work directly with AC power. This is because AC power (such as 50Hz 220V mains power) outputs power intermittently. In order to ensure uninterrupted power output, an electrolytic capacitor (such as Figure 2 The capacitor C3 in the switching circuit stores energy, so when the power supply is at a trough, the continuity of the power supply depends on the energy stored in the electrolytic capacitor to maintain a stable power supply. However, electrolytic capacitors are generally large and easily damaged, making the switching power supply circuit relatively large and having a short lifespan. When the switching power supply circuit charges the battery of the mobile terminal, the battery powers the mobile terminal, and the continuity of the power supply is guaranteed by the battery. At the same time, the battery can be regarded as a capacitive load. This means that the switching power supply circuit does not need to continuously output stable DC power when charging the battery, so the output current of the switching power supply circuit can be controlled to be intermittent.

[0077] Compared to traditional constant voltage and constant current charging, pulsating current charging can reduce lithium plating in lithium batteries, extending battery life. It also reduces the probability and intensity of arcing at the charging interface contacts, extending the life of the charging interface. It also helps reduce battery polarization, speed up charging, and reduce battery heat generation, ensuring safe and reliable charging. Furthermore, because the output is pulsating current, there's no need for an electrolytic capacitor in the battery charging circuit, simplifying and miniaturizing the charging circuit while significantly reducing costs.

[0078] Therefore, the present invention provides the case where the output current of the DC-DC converter is discontinuous, such as Figure 5 As shown in the circuit diagram of the second embodiment of the switching power supply circuit, the control circuit 5 is configured to control the operating state of the DC-DC converter 4 according to the charging requirement to adjust the charging current, so that the charging current is intermittent, and at least when the absolute value of the low-frequency AC input voltage Vin is lower than the predetermined threshold value Vth, the value of the charging current is zero, and the average value of the charging current meets the charging requirement.

[0079] Specifically, when the absolute value of the low-frequency AC input voltage Vin is lower than a predetermined threshold, the control circuit 5 controls the DC-DC converter 4 to not operate, so that the charging current Io is zero. When the absolute value of the low-frequency AC input voltage Vin is higher than a predetermined threshold, the control circuit 5 controls the DC-DC converter 4 to operate, so that the charging current Io is non-zero. In this way, a discontinuous charging current Io can be obtained.

[0080] Furthermore, since the output voltage of the rectifier module 3 is positively correlated with the low-frequency AC input voltage Vin, in other embodiments, the output voltage of the rectifier module 3 may be compared with another predetermined threshold to obtain an intermittent charging current Io. Since the output voltage of the rectifier module 3 is DC, when the output voltage of the rectifier module 3 is lower than the other predetermined threshold, the control circuit 5 controls the DC-DC converter 4 to not operate, so that the charging current Io is zero. When the output voltage of the rectifier module 3 is higher than the other predetermined threshold, the control circuit 5 controls the DC-DC converter 4 to operate, so that the charging current Io is non-zero.

[0081] Furthermore, the value of the interval in which the charging current Io is not zero is fixed, and the waveform of the charging current Io may be a square wave.

[0082] Furthermore, the value of the interval in which the charging current Io is not zero is variable, and the waveform of the charging current Io can be a triangular wave or a sine wave.

[0083] Furthermore, when the absolute value of the low-frequency AC input voltage Vin is higher than a predetermined threshold, the control circuit 5 is configured to adjust the operating frequency of the DC-DC converter 4 to adjust the charging current Io, and the DC-DC converter 4 is configured to operate with a fixed duty cycle at the operating frequency.

[0084] Furthermore, the frequency of the charging current Io is twice the frequency of the low-frequency AC input voltage Vin, or the frequency of the charging current Io is approximately twice the frequency of the low-frequency AC input voltage Vin.

[0085] Furthermore, the DC-DC converter 4 may be a resonant converter. The present invention is described using a half-bridge LLC resonant circuit as an example. It should be understood that the DC-DC converter 4 may also be other resonant circuits, such as an LCC circuit.

[0086] The control circuit 5 controls the operating state of the DC-DC converter 4 based on the sampling signal Vs of the low-frequency AC input voltage Vin, so that the charging current Io is intermittent. Specifically, at least when the absolute value of the low-frequency AC input voltage Vin is lower than a predetermined threshold, the DC-DC converter 4 is controlled to be inoperative, so that the value of the charging current Io is zero. When the absolute value of the low-frequency AC input voltage Vac is higher than the predetermined threshold, the DC-DC converter 4 is controlled to be operational, so that the value of the charging current Io is non-zero. This allows the charging current Io to be a pulsating current.

[0087] Figure 5 The high frequency switch network 1, transformer 2, rectifier module 3 and DC-DC converter 4 in the second embodiment of the switching power supply circuit are all connected to the Figure 2 The high-frequency switch network 1 , transformer 2 , rectifier module 3 and DC-DC converter 4 in the first embodiment of the switching power supply circuit are similar in structure and are not described in detail here.

[0088] Figure 6 Schematic diagram of a circuit of a first embodiment of a DC-DC converter and a control circuit according to the present invention; the DC-DC converter 4 adopts a half-bridge LLC circuit, including a switching circuit 41, a transformer 42, a secondary rectifier circuit 43, a resonant inductor L5 and a resonant capacitor C4.

[0089] In this embodiment, the switch circuit 41 includes a first switch Q21 and a second switch Q22, wherein the first and second switches Q21 and Q22 share a common terminal a. The first switch Q21 is connected between the output terminal of the rectifier module 3 and the common terminal a. The second switch Q22 is connected between the common terminal a and ground. The first and second switches Q21 and Q22 are controlled to turn on or off by control signals G1 and G2, respectively.

[0090] Specifically, when the first switch Q21 is on and the second switch Q22 is off, the resonant circuit is connected to the output terminal of the rectifier module 3, and the input voltage Vin1 is used to power the rechargeable battery while simultaneously charging the energy storage element in the resonant circuit. When the first switch Q21 is off and the second switch Q22 is on, the energy storage element in the resonant circuit discharges to power the rechargeable battery.

[0091] The capacitor C2 is connected to the output end of the rectifier module 3 and is used to filter the rectified voltage signal output by the rectifier module 3 .

[0092] Transformer 42 includes a primary winding L6 and a secondary winding L7. A resonant inductor L5 is connected between the common terminal a and one end of the primary winding L6, and a resonant capacitor C4 is connected between the other end of the primary winding L6 and ground. Thus, the resonant inductor L5, primary winding L6, and resonant capacitor C4 are connected in series to form an LLC resonant circuit. A secondary rectifier circuit 43 is connected to the secondary winding L7. The induced AC current on the secondary side is rectified by the secondary rectifier circuit 43 to provide a charging current Io to the output terminal, thereby charging the battery.

[0093] Furthermore, the LLC resonant circuit uses a fixed duty cycle control, adjusting the operating frequency (switching frequency) to adjust the output charging current Io. That is, under different input voltages, the duty cycle remains unchanged, and the LLC resonant converter's switching frequency is adjusted to adjust the average value of the charging current Io to meet charging requirements.

[0094] The control circuit 5 is configured to adjust the operating frequency of the DC-DC converter 4 to regulate the charging current Io. Specifically, the control circuit 5 includes a first control signal generating circuit 51, a second control signal generating circuit 52, and a logic circuit 53. The first control signal generating circuit 51 is configured to generate a first control signal En based on the AC input voltage Vin and a predetermined threshold value Vth. The second control signal generating circuit 52 is configured to generate second control signals PWMH and PWML based on charging requirements and the charging current Io output by the DC-DC converter 4. The logic circuit 53 is configured to adjust the charging current Io based on the first control signal En and the second control signals PWMH and PWML.

[0095] Furthermore, the logic circuit 53 is configured to, in response to the first control signal En being valid, control the DC-DC converter 4 to operate at a fixed duty cycle at an operating frequency according to the second control signals PWMH and PWML; and, in response to the first control signal En being invalid, control the DC-DC converter 4 to not operate.

[0096] The first control signal generating circuit 51 includes an input signal sampling circuit and a comparison circuit. The input signal sampling circuit samples the AC input signal Vin to obtain an input voltage sampling signal Vs. The comparison circuit receives the input voltage sampling signal Vs and a predetermined threshold value Vth at its input terminal and compares the absolute value Vs′ of the input voltage sampling signal Vs with the predetermined threshold value Vth to obtain the first control signal En.

[0097] Furthermore, the first control signal En is an enable signal. In response to the absolute value Vs' of the comparison input voltage sampling signal Vs being greater than a predetermined threshold value Vth, the first control signal En is at a high level. In response to the absolute value Vs' of the comparison input voltage sampling signal Vs being less than the predetermined threshold value Vth, the first control signal En is at a low level. It should be understood that the first control signal generating circuit 51 is not limited to the method described herein, and those skilled in the art may employ other equivalent or existing methods to generate the first control signal.

[0098] The second control signal generation circuit 52 includes a first error amplifier Gm1, a second error amplifier Gm2, a compensation circuit, and a drive circuit. The first error amplifier Gm1 is configured to compare the output current sampling signal Vfb with the current reference signal Iref to generate a first error signal Vref. The second error amplifier Gm2 is configured to compare the output current sampling signal Vfb with the first error signal Vref to generate a second error signal Vcs. The compensation circuit is configured to generate a compensation signal Vcp based on the second error signal Vcs. The drive circuit is configured to generate the second control signals PWMH and PWML based on the compensation signal Vcp.

[0099] The first error amplifier Gm1 receives a current reference signal Iref at its non-inverting input and an output current sampling signal Vfb at its inverting input. The output current sampling signal is a voltage signal representing the output current. The current reference signal Iref is used to represent the charging requirement, specifically the average current required for battery charging. Consequently, the output signal of the first error amplifier Gm1 represents the difference between the actual charging current Io and the required charging current Io. This forms an average current loop to control the average value of the charging current Io to approach the current reference signal Iref.

[0100] Furthermore, the second control signal generating circuit 52 also includes a capacitor C5 connected between the output terminal of the first error amplifier Gm1 and the ground terminal, which converts the current signal output by the first error amplifier Gm1 into a voltage signal, namely, the first error signal Vref. The first error signal Vref serves as a reference signal and is input to the non-inverting input terminal of the second error amplifier Gm2. The output current sampling signal Vfb is input to the inverting input terminal of the second error amplifier Gm2. The second error signal Vcs is obtained by comparing the output current sampling signal Vfb with the first error signal Vref. This forms an instantaneous current loop that controls the instantaneous value of the charging current Io by adjusting the operating frequency of the DC-DC converter 4.

[0101] Furthermore, the second control signal generating circuit 52 further includes a capacitor C6 connected between the output terminal of the second error amplifier Gm2 and the ground terminal, for converting the current signal output by the second error amplifier into a corresponding voltage signal Vcs.

[0102] exist Figure 6 In an embodiment, the compensation circuit includes a resistor R1, a resistor R2 and an optocoupler, and is configured to generate a compensation signal Vcp according to the second error signal Vcs.

[0103] The driving circuit is configured to generate a PWM (pulse width modulation) signal according to the compensation signal Vcp, that is, the second control signals are PWMH and PWML. In one embodiment, the driving circuit generates the second control signals PWMH and PWML by comparing the compensation signal Vcp with the ramp signal Vsaw.

[0104] The logic circuit 53 is configured to adjust the charging current Io according to the first control signal En and the second control signals PWMH and PWML.

[0105] Furthermore, in response to the first control signal En being at a low level, the logic circuit 53 does not output the control signals G1 and G2. In response to the first control signal En being at a high level, the logic circuit 53 generates the control signals G1 and G2 based on the second control signals PWMH and PWML to control the switching circuit to operate at a fixed duty cycle at the operating frequency. It should be understood that the operation of the logic circuit is not limited to the one described above. For example, in response to the first control signal En being at a low level, the control signals G1 and G2 may both be output at a low level. In response to the first control signal En being at a high level, the second control signals PWMH and PWML may be output as the control signals G1 and G2.

[0106] It should be noted that if Figure 6 In the embodiment, the control circuit 5 removes the first control signal generating circuit 51 and only generates the second control signal through the second control signal generating circuit 52 to control the switching state of the DC-DC converter 4 and thus control the output current to meet the charging requirement. At this time, the charging current is continuous, that is, Figure 3 The situation in the embodiment is obvious to those skilled in the art and is described here without being further described as a separate embodiment.

[0107] Figure 7 This is a working waveform diagram of the switching power supply circuit of the present invention under the control of the control circuit embodiment 1. Figure 7The waveform diagram shows the absolute value Vs' of the input voltage sampling signal Vs, the predetermined threshold Vth, the first control signal En, the output current sampling signal Vfb and the second control signals G1 and G2 (PWM signals), where the abscissa is time t and the ordinate is each component.

[0108] Specifically, at time t1, the absolute value Vs' of the input voltage sampling signal Vs drops to equal the predetermined threshold Vth, the first control signal En switches to a low level, the logic circuit 53 does not output the control signals G1 and G2, and controls the switch circuit 41 to be inoperative, that is, the first switch Q21 and the second switch Q22 are disconnected, causing the charging current Io to be zero (or close to zero), thereby causing the output current sampling signal Vfb to be 0. At time t2, the absolute value Vs' of the input voltage sampling signal Vs rises to equal the predetermined threshold Vth, the first control signal En switches to a high level, and the logic circuit 53 outputs the control signals G1 and G2, controlling the switch circuit 41 to operate, causing the first switch Q21 and the second switch Q22 to be alternately turned on and off, resulting in a high charging current Io and a high output current sampling signal Vfb. At time t3, the absolute value Vs' of the input voltage sampling signal Vs drops again to equal the predetermined threshold Vth, and the cycle repeats. The first switch Q21 and the second switch Q22 are both in the off state for a period of time within each cycle, making the charging current Io zero; and are both in the alternate on or off state for another period of time within each cycle, making the charging current Io higher. In this way, the charging current Io can be a pulsed current.

[0109] Figure 8 This is a working waveform diagram of the driving circuit in Example 1 of the control circuit of the present invention. Figure 8 The waveform diagram of the compensation signal Vcp, the ramp signal Vsaw and the PWM (PWMH and PWML) signals during the time period when the first switch Q21 and the second switch Q22 are in the alternate on or off state is shown, wherein the abscissa is time t and the ordinate is each component.

[0110] Specifically, during the period when the first switch Q21 and the second switch Q22 are alternately on or off, at time t4, the ramp signal Vsaw begins to rise, PWMH switches to a high level, and PWML remains at a low level. At time t5, the ramp signal Vsaw rises to be equal to the compensation signal Vcp, PWMH switches to a low level, and PWML remains at a low level. After the reading time Δt (dead time), that is, at time t6, the ramp signal Vsaw begins to rise, PWMH remains at a low level, and PWML switches to a high level. This cycle continues so that the duty cycle of the PWM signal is always 50%, and the DC-DC converter 4 is controlled to operate at a fixed duty cycle at the operating frequency.

[0111] Therefore, the operating frequency of the DC-DC converter 4 can be adjusted according to the difference between the current reference signal Iref and the output current sampling signal Vfb, and the DC-DC converter 4 can be operated at a fixed duty cycle at the operating frequency, so as to adjust the average value of the charging current Io to meet the charging requirements to charge the battery.

[0112] Figure 9 A circuit diagram of a second embodiment of the control circuit of the present invention is provided; Figure 9 The structure of the DC-DC converter 4 and the control circuit 5 is similar to Figure 6 The structure of the control circuit 5 of the DC-DC converter 4 is similar, and the similarities are not repeated here. The difference is that the drive circuit obtains the resonant current sampling signal Ichg and generates the second control signals PWMH and PWML according to the resonant current sampling signal Ichg and the compensation signal Vcp.

[0113] Figure 10 FIG. 1 is a circuit diagram of a driving circuit in a second embodiment of a control circuit of the present invention; FIG. Figure 10 As shown, the drive circuit includes a second comparator CMP2, a third comparator CMP3, and an RS trigger. Specifically, the drive circuit also includes a capacitor Cchg, which converts the resonant current sampling signal Ichg into a corresponding voltage signal Vchg, which is input to the inverting input of the second comparator CMP2 and the non-inverting input of the third comparator CMP3. Capacitor Cchg is not an electrolytic capacitor. It should be understood that the drive circuit can directly sample the voltage of the resonant capacitor C3 as the voltage signal Vchg, without the need for capacitor Cchg. Simultaneously, the drive circuit acquires a fixed common-mode voltage signal Vcm. The non-inverting input of the second comparator CMP2 receives the difference between the signal Vcm and the compensation signal Vcp, while the inverting input of the third comparator CMP3 receives the sum of the signal Vcm and the compensation signal Vcp. The logic circuit 53 is configured to adjust the charging current Io based on the first control signal En and the second control signals PWMH and PWML.

[0114] In response to the first control signal En being at a low level, the logic circuit 53 does not output the control signals G1 and G2. In response to the first control signal En being at a high level, the logic circuit 53 outputs the control signals G1 and G2 to control the switching circuit to operate at a fixed duty cycle at the operating frequency.

[0115] It should be noted that if Figure 9In the embodiment, the control circuit 6 removes the first control signal generating circuit 51 and only generates the second control signal through the second control signal generating circuit 52 to control the switching state of the DC-DC converter 4 and thus control the output current to meet the charging requirement. At this time, the charging current is continuous, that is, Figure 3 The situation in the embodiment is obvious to those skilled in the art and is described here without being further described as a separate embodiment.

[0116] Figure 11 This is a working waveform diagram of the switching power supply circuit of the present invention under the control of the second embodiment of the control circuit. Figure 11 The waveforms of the resonant current sampling signal Ichg, the voltage signal Vchg, and the PWM (PWMH and PWML) signals during the time periods when the first switch Q21 and the second switch Q22 are alternately turned on or off are shown, wherein the abscissa is time t and the ordinate is each component.

[0117] Specifically, at time t1, the voltage signal Vchg rises to equal Vcm + Vcp, and the output of the third comparator CMP3 is high. This means that the reset signal of the RS flip-flop is active, PWMH switches to a low level, and PWML switches to a high level. At time t2, the voltage signal Vchg falls to equal Vcm - Vcp, and the output of the second comparator CMP2 is high. This means that the set signal of the RS flip-flop is active, PWMH switches to a high level, and PWML switches to a low level. This cycle ensures that the duty cycle of the PWM signal remains at 50%, controlling the DC-DC converter 4 to operate at a fixed duty cycle at the operating frequency.

[0118] Figure 12 A circuit diagram of a third embodiment of the control circuit of the present invention is provided; Figure 12 The structure of the DC-DC converter 4 and the control circuit 5 is similar to Figure 9 The structure of the control circuit 5 of the DC-DC converter 4 is similar, and the similarities are not repeated here. The difference is that the drive circuit directly obtains the resonant capacitor voltage as the voltage signal Vchg, and generates the second control signals PWMH and PWML according to the voltage signal Vchg and the compensation signal Vcp. Similarly, it should be understood that the drive circuit is as follows Figure 12 As shown, the voltage signal Vchg is obtained by sampling the resonant capacitor current and integrating it through the capacitor Cchg.

[0119] Furthermore, Figure 13 FIG. 1 is a circuit diagram of a driving circuit according to a third embodiment of the present invention. Figure 13As shown, a second comparator CMP2, a third comparator CMP3, and an RS flip-flop are provided. Specifically, the voltage signal Vchg is input to the inverting input of the second comparator CMP2 and the non-inverting input of the third comparator CMP3. Simultaneously, the drive circuit obtains a fixed common-mode voltage signal Vcm. The non-inverting input of the second comparator CMP2 receives the difference between the signal Vcm and the compensation signal Vcp, while the inverting input of the third comparator CMP3 receives the sum of the signal Vcm and the compensation signal Vcp.

[0120] The logic circuit 53 is configured to adjust the charging current Io according to the first control signal En and the second control signals PWMH and PWML.

[0121] Furthermore, in response to the first control signal En being at a low level, the logic circuit 53 does not output the control signals G1 and G2. In response to the first control signal En being at a high level, the logic circuit 53 outputs the control signals G1 and G2 to control the switching circuit to operate at a fixed duty cycle at the operating frequency.

[0122] It should be noted that if Figure 12 The control circuit 6 in the embodiment removes the first control signal generating circuit 51, and only generates the second control signal through the second control signal generating circuit 52 to control the switching state of the DC-DC converter 4 and thus control the output current to meet the charging requirement. At this time, the charging current is continuous, that is, Figure 3 The situation in the embodiment is obvious to those skilled in the art and is described here without being further described as a separate embodiment.

[0123] Figure 14 This is a working waveform diagram of the switching power supply circuit of the present invention under the control of the control circuit embodiment 3. Figure 14 The waveform diagram of the voltage signal Vchg and the PWM (PWMH and PWML) signal during the time period when the first switch Q21 and the second switch Q22 are in the alternate on or off state is shown, wherein the abscissa is time t and the ordinate is each component.

[0124] Specifically, at time t1, the voltage signal Vchg rises to equal Vcm + Vcp, and the output of the third comparator CMP3 is high. This means that the reset signal of the RS flip-flop is active, PWMH switches to a low level, and PWML switches to a high level. At time t2, the voltage signal Vchg falls to equal Vcm - Vcp, and the output of the second comparator CMP2 is high. This means that the set signal of the RS flip-flop is active, PWMH switches to a high level, and PWML switches to a low level. This cycle ensures that the duty cycle of the PWM signal remains at 50%, controlling the DC-DC converter 4 to operate at a fixed duty cycle at the operating frequency.

[0125] While embodiments of the present invention have been described above, these embodiments do not exhaustively describe all details and do not limit the invention to the specific embodiments described. Obviously, many modifications and variations are possible based on the above description. These embodiments are selected and described in detail in this specification in order to better explain the principles and practical applications of the present invention, thereby enabling those skilled in the art to better utilize the present invention and its modifications. The present invention is limited only by the claims and their full scope and equivalents.

Claims

1. A switching power supply circuit for charging a battery, characterized in that: Including: high-frequency switching network, transformer, rectifier module, DC-DC converter and control circuit, The high-frequency switching network includes an AC input terminal for receiving a low-frequency AC input voltage, a first switching transistor, a second switching transistor, a third switching transistor, a fourth switching transistor, and an energy storage module. The first end of the first switching transistor and the first end of the second switching transistor are respectively connected to two ports of the AC input terminal, the second end of the first switching transistor is connected to the first end of the third switching transistor, the second end of the second switching transistor is connected to the first end of the fourth switching transistor, and the second end of the third switching transistor is connected to the second end of the fourth switching transistor. The energy storage module and the primary winding of the transformer are connected between a first node and a second node, wherein the first node is a common terminal of the first and third switching transistors, and the second node is a common terminal of the second and fourth switching transistors. During a positive half-cycle of the low-frequency AC input voltage, the first and third switching transistors are always turned on, and the second and fourth switching transistors are alternately turned on in a high-frequency PWM manner. During a negative half-cycle of the low-frequency AC input voltage, the second and fourth switching transistors are always turned on, and the first and third switching transistors are alternately turned on in a high-frequency PWM manner. The secondary winding of the transformer is connected to the input end of the rectifier module, the output end of the rectifier module is connected to the DC-DC converter, and the DC-DC converter outputs a charging current; The control circuit is configured to control the working state of the DC-DC converter according to the charging requirement to adjust the charging current so that the average value of the charging current meets the charging requirement.

2. The switching power supply circuit according to claim 1, wherein: The energy storage module includes a first inductor and a first capacitor. The first inductor, the first capacitor, and the primary winding are connected in series and connected between the first node and the second node.

3. The switching power supply circuit according to claim 2, wherein: The high-frequency switching network operates in an LLC resonance state.

4. The switching power supply circuit according to claim 1, wherein: The control circuit includes a second control signal generating circuit, which is used to generate a second control signal according to a charging requirement and the charging current; the control circuit is configured to adjust the charging current according to the second control signal.

5. The switching power supply circuit according to claim 1 or 2, wherein: The control circuit is configured to control the operating state of the DC-DC converter according to charging requirements to adjust the charging current so that the charging current is intermittent and the value of the charging current is zero at least when the absolute value of the low-frequency AC input voltage is lower than a predetermined threshold.

6. The switching power supply circuit according to claim 5, wherein: The frequency of the charging current is configured to be twice the frequency of the low-frequency AC input voltage.

7. The switching power supply circuit according to claim 5, wherein: The value of the interval in which the charging current is not zero is fixed.

8. The switching power supply circuit according to claim 5, wherein: The value of the interval in which the charging current is not zero varies.

9. The switching power supply circuit according to claim 5, wherein: The DC-DC converter is a resonant converter, and the control circuit is configured to adjust the operating frequency of the DC-DC converter to adjust the charging current.

10. The switching power supply circuit according to claim 9, wherein: The DC-DC converter is configured to operate at a fixed duty cycle at the operating frequency.

11. The switching power supply circuit according to claim 5, wherein: The control circuit comprises: a first control signal generating circuit, configured to generate a first control signal according to the low-frequency AC input voltage and a predetermined threshold; a second control signal generating circuit, configured to generate a second control signal according to a charging requirement and the charging current; and a logic circuit, configured to adjust the charging current according to the first control signal and the second control signal; The logic circuit is configured to control the DC-DC converter to operate at a fixed duty cycle according to the second control signal in response to the first control signal being valid, and to control the DC-DC converter not to operate in response to the first control signal being invalid.

12. The switching power supply circuit according to claim 11, wherein: The first control signal generating circuit includes: The comparison circuit is configured to compare the absolute value of the sampling signal of the low-frequency AC input voltage with a predetermined threshold value to obtain the first control signal.

13. The switching power supply circuit according to claim 4 or 11, characterized in that: The second control signal generating circuit includes: a first error amplifier, configured to compare the sampling signal of the charging current with a current reference signal to obtain a first error signal, wherein the current reference signal is used to represent a charging requirement; a second error amplifier, configured to compare the sampling signal of the charging current with the first error signal to obtain a second error signal; a compensation circuit, configured to generate a compensation signal according to the second error signal; and A driving circuit is configured to generate the second control signal according to the compensation signal.

14. The switching power supply circuit according to claim 13, wherein: The DC-DC converter is a resonant converter, and the driving circuit generates the second control signal according to the compensation signal and the resonant current sampling signal.

15. The switching power supply circuit according to claim 14, wherein: The driving circuit includes: a second comparator, configured to obtain a set signal according to the resonant current sampling signal and the compensation signal; a third comparator, configured to obtain a reset signal according to the resonant current sampling signal and the compensation signal; and A trigger is configured to generate the second control signal according to the set signal and the reset signal.

16. The switching power supply circuit according to claim 13, wherein: The DC-DC converter is a resonant converter, and the driving circuit generates the second control signal according to the compensation signal and the resonant voltage sampling signal.

17. The switching power supply circuit according to claim 16, wherein: The driving circuit includes: a second comparator, configured to obtain a set signal according to the resonant voltage sampling signal and the compensation signal; a third comparator, configured to obtain a reset signal according to the resonant voltage sampling signal and the compensation signal; and A trigger is configured to generate the second control signal according to the set signal and the reset signal.

18. The switching power supply circuit according to claim 1, wherein: The rectifier module is a full-wave rectifier circuit.

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