Power conversion device

Abstract

The present invention provides a power conversion device capable of miniaturizing a DC capacitor connected between an AC / DC converter and a DC / DC converter without increasing the size of components constituting the DC / DC converter. The device comprises a DC capacitor (5) connected between an AC / DC converter (101) and a DC / DC converter (102), and a control circuit (30) for controlling the AC / DC converter (101) and the DC / DC converter (102). The control circuit (30) generates an output current command for the DC / DC converter (102) by superimposing an AC current command on a DC current command, and determines the amplitude of the AC current command based on temperature information obtained from a DC capacitor temperature acquisition unit (20).

Description

Technical Field

[0001] The present application relates to a power conversion device in which an AC / DC converter and a DC / DC converter are connected via a DC capacitor. Background Art

[0002] In a conventional power conversion device that connects an AC / DC converter and a DC / DC converter in series, the AC power supply is rectified by a rectifier circuit, and the input DC voltage is boosted and output by a high power factor converter circuit. This high power factor converter circuit improves the power factor by controlling the on / off time ratio of the switching transistor to a sinusoidal waveform, thereby controlling the DC voltage. The boosted DC voltage is stabilized by a DC capacitor, and the DC / DC converter converts the DC voltage supplied from the DC capacitor to the desired DC voltage to drive the load.

[0003] When the DC capacitor is reduced in capacitance, the ripple voltage of the DC bus voltage, which fluctuates at twice the power supply frequency, increases, particularly when the commercial AC power supply is single-phase. If the ripple voltage of the DC bus voltage increases, the upper limit of the ripple voltage exceeds the withstand voltage of the DC capacitor, or the lower limit falls below the commercial AC power supply voltage, causing an inrush current to flow from the commercial AC power supply, deteriorating the power factor. To suppress the ripple voltage of the DC bus voltage, an AC current command is determined that is minimum at the zero-crossing phase of the commercial AC power supply and maximum at the peak phase. This suppresses the ripple current of the DC capacitor. The AC current command is superimposed on the DC current command to generate an output current command for the DC / DC converter, and the output current command is used to control the output of the DC / DC converter (for example, see Patent Document 1).

[0004] Prior art literature

[0005] Patent Literature

[0006] Patent Document 1: Japanese Patent No. 6026049 Summary of the Invention

[0007] Technical problem to be solved by the invention

[0008] In the power conversion device of Patent Document 1, an AC current command is superimposed on the DC current command of the DC / DC converter to perform output control. As a result, a current ripple with a frequency twice the power supply frequency is generated in the DC output current of the DC / DC converter. As a result, the effective value of the current flowing through the switching elements and reactors, which are components of the DC / DC converter, increases, and losses increase, so it is necessary to increase the size to achieve heat generation. In the case of an isolated DC / DC converter, the losses of the insulation transformer also increase, so it is necessary to increase the size, which hinders the miniaturization and cost reduction of the power converter. In addition, in the reactor, the required DC superposition characteristics increase due to the increase in the current flowing through it. Therefore, especially in applications where a core with a drastically reduced DC superposition characteristic, such as ferrite, is used as the core, the core needs to be enlarged to ensure the DC superposition characteristics, which hinders the miniaturization and cost reduction of the power converter.

[0009] The present application is made to solve the above-mentioned problems, and its object is to obtain a power conversion device that can miniaturize a DC capacitor without increasing the size of components constituting a DC / DC converter.

[0010] Technical means for solving technical problems

[0011] The power conversion device disclosed in the present application includes: an AC / DC converter that converts AC power from an AC power supply into DC power; a DC / DC converter that is connected to the DC side of the AC / DC converter and performs voltage conversion on the DC power; a DC capacitor that is connected between the AC / DC converter and the DC / DC converter and filters the power; and a control circuit that controls the AC / DC converter and the DC / DC converter. The control circuit superimposes an AC current instruction that is minimum at the zero-crossing phase of the AC power supply and maximum at the peak phase on the DC current instruction to generate an output current instruction for the DC / DC converter, and uses the output current instruction to control the output of the DC / DC converter. The power conversion device has a DC capacitor temperature acquisition unit that acquires the ambient temperature of the DC capacitor. The control circuit determines the amplitude of the AC current instruction based on the temperature information obtained from the DC capacitor temperature acquisition unit.

[0012] Effects of the Invention

[0013] The power conversion device disclosed in this application includes a DC capacitor for power filtering between an AC / DC converter and a DC / DC converter. Control is performed by superimposing an AC current command on a DC current command. By varying the AC current command based on the temperature of the DC capacitor, the DC capacitor can be miniaturized without increasing the size of the semiconductor switching elements that make up the DC / DC converter. This enables a more compact and cost-effective power conversion device. BRIEF DESCRIPTION OF THE DRAWINGS

[0014] Figure 1 This is a configuration diagram of a power conversion device according to the first embodiment.

[0015] Figure 2 This is a current path diagram illustrating the operation of the AC / DC converter of the power conversion device according to the first embodiment.

[0016] Figure 3 This is a current path diagram illustrating the operation of the AC / DC converter of the power conversion device according to the first embodiment.

[0017] Figure 4 This is a current path diagram illustrating the operation of the AC / DC converter of the power conversion device according to the first embodiment.

[0018] Figure 5 This is a current path diagram illustrating the operation of the AC / DC converter of the power conversion device according to the first embodiment.

[0019] Figure 6 This is a current path diagram illustrating the operation of the DC / DC converter of the power conversion device according to the first embodiment.

[0020] Figure 7 This is a current path diagram illustrating the operation of the DC / DC converter of the power conversion device according to the first embodiment.

[0021] Figure 8 This is a waveform diagram showing the voltage and current of the AC power supply and the ripple voltage of the DC capacitor according to the reference example of the first embodiment.

[0022] Figure 9 1 is a waveform diagram showing each component included in the output current of the DC capacitor according to the first embodiment.

[0023] Figure 10 1 is a waveform diagram showing input and output currents of the DC capacitor according to the first embodiment.

[0024] Figure 11 This is a control block diagram showing a configuration for generating a gate signal of the AC / DC converter according to the first embodiment.

[0025] Figure 12 This is a control block diagram showing a configuration for generating a gate signal of the AC / DC converter according to the first embodiment.

[0026] Figure 13 This is a control block diagram showing a configuration for generating a gate signal of the DC / DC converter according to the first embodiment.

[0027] Figure 14 This is a waveform diagram showing current waveforms of components constituting the DC / DC converter according to the first embodiment.

[0028] Figure 15 This is a waveform diagram showing current waveforms of components constituting the DC / DC converter according to the first embodiment.

[0029] Figure 16 This is a diagram showing DC superposition characteristics of a filter reactor constituting the DC / DC converter according to the first embodiment.

[0030] Figure 17 This is a diagram showing the temperature characteristics of the capacitance of the DC capacitor constituting the power conversion device according to the first embodiment.

[0031] Figure 18 This is a diagram showing changes in the ripple current amplitude command value depending on the temperature in the power conversion device according to the first embodiment. DETAILED DESCRIPTION

[0032] Implementation method 1.

[0033] Hereinafter, embodiment 1 will be described.

[0034] Figure 1 1 is a diagram showing a circuit configuration of a power conversion device according to Embodiment 1. Figure 1 As shown, the power conversion device converts the AC voltage Vin of the single-phase AC power supply 1 into a primary-side DC voltage Vdc, further converts the primary-side DC voltage Vdc into a secondary-side DC voltage insulated by a transformer 7, and outputs the DC voltage VL to a load 11 such as a battery.

[0035] The power conversion device includes: an AC / DC converter 101 that takes an AC power supply 1 as input and converts an AC voltage Vin into a primary-side DC voltage Vdc; a DC capacitor 5 that filters the output of the AC / DC converter 101; and a DC / DC converter 102 that takes the DC capacitor 5 as input and converts the primary-side DC voltage Vdc into a DC voltage VL for a load 11.

[0036] The AC / DC converter 101 includes: a PFC (Power Factor Collection) inductor 2 for improving the power factor; and a rectifier circuit 3. The rectifier circuit 3 is a totem pole type, which rectifies the AC voltage Vin of the AC power supply 1 into a DC power supply Vdc for the DC capacitor 5. In this totem pole type, semiconductor switching elements 3a and 3b composed of MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) with built-in diodes between the source and the drain form a half-bridge structure, and a series circuit is formed by connecting diodes 4a and 4b in series in parallel with the half-bridge structure.

[0037] DC / DC converter 102 includes an insulating transformer (insulating transformer) 7; a single-phase inverter 6 connected to primary winding 7a of transformer 7. This inverter 6 forms a full-bridge structure with semiconductor elements 6a and 6b, each consisting of MOSFETs with built-in diodes between their source and drain electrodes, to convert DC voltage Vdc from DC capacitor 5 into AC voltage; and a rectifier circuit 8 connected to secondary winding 7b of transformer 7. This rectifier circuit 8 forms a full-bridge structure with diodes 8a to 8d, serving as rectifier elements (semiconductor elements). The output of rectifier circuit 8 is connected to an output filtering reactor 9 and an output capacitor 10, and outputs a DC voltage VL to a load 11.

[0038] Furthermore, a control circuit 30 is configured outside the main circuit and monitors the input voltage Vin and output voltage VL, respectively, and inputs them to the control circuit 30. Furthermore, the input current Iin is monitored by a current sensor 33, and the output current IL is monitored by an output current sensor 34, and both are input to the control circuit 30. The control circuit 30 outputs gate signals 31 to the semiconductor switching elements 3a and 3b so that the DC voltage Vdc reaches the target voltage and the input current has a high power factor, thereby controlling the on-duty ratio (on-period) of the semiconductor switching elements 3a and 3b. Furthermore, the control circuit 30 outputs gate signals 32 to the semiconductor switching elements 6a to 6d so that the output current IL reaches the target current, thereby controlling the on-duty ratio (on-period) of the semiconductor switching elements 6a to 6d.

[0039] The ambient temperature Tcon of the DC capacitor 5 is acquired by the DC capacitor temperature acquisition unit 20 via a thermistor, for example, and input to the control circuit 30. The control circuit 30 determines a target output current value based on the ambient temperature Tcon of the DC capacitor 5.

[0040] The semiconductor switching elements 3a, 3b, and 6a to 6d are not limited to MOSFETs, and may be self-extinguishing semiconductor switching elements such as IGBTs (Insulated Gate Bipolar Transistors) in which diodes are connected in antiparallel.

[0041] The operation of the power conversion device configured in this manner will be described below. Figures 2 to 5 1 is a current path diagram illustrating the operation of the AC / DC converter 101. When the AC voltage Vin is positive, if the semiconductor switch element 3b is turned on, the input current is short-circuited through the reactor 2, the reactor 2 is excited, and the current increases in positive polarity ( Figure 2 If the semiconductor switching element 3b is turned off, the excitation energy accumulated in the reactor 2 is output to the DC capacitor 5 side via the semiconductor switching element 3a. At this time, the current of the reactor 2 decreases ( Figure 3 When the AC voltage Vin is negative, if the semiconductor switch element 3a is turned on, the input current is short-circuited through the reactor 2, the reactor 2 is excited, and the current increases in negative polarity ( Figure 4 When the semiconductor switching element 3a is turned off, the excitation energy accumulated in the reactor 2 is output to the DC capacitor 5 side via the diode 4a. At this time, the current of the reactor 2 decreases ( Figure 5 ).

[0042] The control circuit 30 controls the semiconductor switching elements 3a and 3b to conduct / cut off as described above to perform high power factor control on the input AC current iac. In addition, the semiconductor switching element 3a and the semiconductor switching element 3b are ideally driven with the same duty cycle. Here, the theoretical duty cycle D3a of the semiconductor switching element 3a when the current iac is controlled to have a high power factor (an example in which the input voltage Vin is positive) is shown in the following formula (1). At this time, the duty cycle D3b of the semiconductor switching element 3b is expressed by formula (2) based on formula (1). Among them, the voltage vac of the AC power supply 1 is defined in formula (3). Therefore, the current iin flowing into the DC capacitor 5 is calculated using formula (4).

[0043] Furthermore, it is assumed that no loss occurs between the AC power supply 1 and the DC capacitor 5. Furthermore, the duty ratios of D3a and D3b include a dead time period during which both semiconductor switching elements 3a and 3b are turned off to prevent the arms from short-circuiting. Furthermore, the duty ratio of D3a can be used to turn on semiconductor switching element 3b.

[0044] D3a=(Vdc-vac) / Vdc…(1)

[0045] D3b=vac / Vdc…(2)

[0046]

[0047] iin=(vac / Vdc)iac

[0048] =(2Vac·Iac / Vdc)sin2ωt…(4)

[0049] Next, the operation of DC / DC converter 102 will be described. DC / DC converter 102 is an example of an isolated full-bridge converter circuit. Control circuit 30 controls the on / off state of semiconductor switching elements 6a-6a, causing DC capacitor 5 to output DC power and controlling current IL and voltage VL to load 11 to desired values. Figure 6 、 Figure 7 : This is a current path diagram illustrating the operation of the DC / DC converter 102. While the semiconductor switching elements 6a and 6b are on, the semiconductor switching elements 6b and 6c are off, and current flows from the DC capacitor 5 to the semiconductor switching element 6a, the primary winding 7a of the transformer 7, and the semiconductor switching element 6d. At the same time, on the secondary side, current flows to the secondary winding 7b of the transformer 7, the diode 8a, the reactor 9, the load 11, and the diode 8d ( Figure 6 Then, the semiconductor switching elements 6a to 6d are turned off, and the current does not flow to the primary side. On the secondary side, the current flows to the diode 8a, diode 8b, reactor 9, load 11 or the diode 8c, diode 8d, reactor 9, load 11 ( Figure 7 ). Then, similar to the period in which semiconductor switching elements 6a and 6d are on and semiconductor switching elements 6b and 6c are off, semiconductor switching elements 6a to 6d are turned off after a period in which semiconductor switching elements 6b and 6c are on and semiconductor switching elements 6a and 6d are off. Furthermore, the period in which semiconductor switching elements 6a and 6d are on and semiconductor switching elements 6b and 6c are off is the same length as the period in which semiconductor switching elements 6b and 6c are on and semiconductor switching elements 6a and 6d are off.

[0050] In the DC / DC converter 102 operating in this manner, the control circuit 30 adjusts the power supplied to the load 11 by adjusting the duty ratio between the period when the semiconductor switching elements 6a and 6d are on (or the period when the semiconductor switching elements 6b and 6c are on) and the period when the semiconductor switching elements 6a to 6d are off, thereby adjusting the load current IL. The DC / DC converter 102 then supplies the DC voltage VL to the load 11 as the current IL, thereby supplying DC power to the load 11. The output current iout outputted from the DC capacitor 5 is discontinuous with respect to the switching cycle of the semiconductor switching elements 6a to 6d, but can be considered as an average continuous current with respect to the cycle of the AC power supply 1. Assume that the output current iout of the DC capacitor 5 is the DC current idc. In this case, the voltage relationship of the DC capacitor 5 can be expressed by the following equation (5). Here, the electrostatic capacitance of the DC capacitor 5 is Cdc, and the AC voltage component (ripple voltage) of the DC capacitor 5 is vc2. The input AC current iac is expressed by equation (6) under the premise of being controlled at a high power factor. If equation (5) is solved for the AC voltage component (ripple voltage) vc2 of the DC capacitor 5, equation (7) is derived.

[0051] Cdc(dvc2 / dt)

[0052] =iin-idc

[0053] =(2Vac·Iac / Vdc)sin2ωt-idc…(5)

[0054]

[0055] vc2=(2Vac·Iac / 2ωCdc·Vdc)sin(2ωt)…(7)

[0056] Formula (7) shows the following situation: if the output current iout of the DC capacitor 5 is a DC current, then due to the high power factor control performed by the AC / DC converter 101 connected to the AC power supply 1, a ripple voltage vc2 with a frequency twice that of the AC power supply 1 will inevitably be generated in the DC capacitor 5. As a reference example of this embodiment, Figure 8 The waveforms of the ripple voltage vc2, the voltage vac and the current iac of the AC power supply 1 are shown in FIG. Figure 8 As shown, the ripple voltage vc2 fluctuates greatly at a frequency twice that of the AC power supply 1.

[0057] In this embodiment, the control circuit 30 intentionally superimposes an AC current component (ripple current) irp on the output current iout of the DC capacitor 5 in an attempt to suppress the ripple voltage vc2 generated in the DC capacitor 5. Specifically, the DC / DC converter 102 is controlled so that an AC current component (ripple current) is generated in the current IL output to the load 11, thereby generating the ripple current irp in the output current iout of the DC capacitor 5.

[0058] Figure 9 This is a waveform diagram showing the components included in the voltage vac and current iac of the AC power supply 1 and the output current iout of the DC capacitor 5. The output current iout of the DC capacitor 5 is a current obtained by superimposing the ripple current irp on the DC current component idc. The superimposed ripple current irp is set to a sinusoidal current with a frequency twice that of the voltage vac of the AC power supply 1. The initial phase of the ripple current irp is set so that it reaches the minimum value at the zero-crossing phase of the AC power supply 1 and the maximum value at the peak phase. If it is defined as in the above formula (3), Figure 9 The voltage vac of the AC power supply 1 is shown, and the AC current iac controlled to a power factor of 1 is defined as in the above equation (6). The ripple current irp of the DC capacitor 5 is expressed by the following equation (8), and the output current iout is expressed by the following equation (9). Here, the effective value of the ripple current irp is denoted as IRP.

[0059]

[0060]

[0061] Figure 10 : is a waveform diagram showing the input and output currents of the DC capacitor 5 in this embodiment. Figure 10 As shown, at the zero-crossing phase of the AC power supply 1, the output current iout reaches its minimum value, matching the fact that the input current iin of the DC capacitor 5 reaches almost zero. Furthermore, at the peak phase, the output current iout reaches its maximum value, matching the fact that the input current iin reaches its maximum value. This suppresses the ripple current compensated by the DC capacitor 5, i.e., the charge and discharge current (iin - iout), and also suppresses the ripple voltage vc2 and the effective value of the ripple current of the DC capacitor 5.

[0062] The DC capacitor 5 outputs the output current iout shown in the above equation (9). Therefore, the voltage relationship of the DC capacitor 5 can be expressed by the following equation (10). If the equation (10) is solved for the ripple voltage vc2 of the DC capacitor 5, the equation (11) is derived.

[0063]

[0064]

[0065] As shown in the above formula (11), if the peak value of the ripple current irp superimposed on the output current iout of the DC capacitor 5 is As ΔVdc increases, the ripple voltage vc2 generated in the DC capacitor 5 decreases. Based on the above equation (11), the amplitude ΔVdc of the ripple voltage vc2, which is the AC voltage component of the DC capacitor 5, is defined by the following equation (12).

[0066]

[0067] Furthermore, the input current iin flowing into the DC capacitor 5 can be expressed by the following equation (13). The output current iout flowing out of the DC capacitor 5 is expressed by the above equation (9). Therefore, the charge and discharge current (iin - iout) of the DC capacitor 5 is expressed by the following equation (14). Furthermore, equations (13) and (14) can be expressed by equations (13a) and (14a) using the load voltage VL and the load current IL. Here, the command value of the load current IL is IL*, and the effective value of the ripple current generated in the load current IL is ILrp.

[0068] iin=(Vac·Iac / Vdc)·(1+cos(2ωt-π))

[0069] …(13)

[0070] =(VL·IL / Vdc)·(1+cos(2ωt-π))

[0071] …(13a)

[0072]

[0073]

[0074] As shown in the above formula (14a), the charge and discharge current (iin-iout) of the DC capacitor 5 becomes a frequency component twice the voltage vac of the AC power supply 1. In addition, if the peak value of the ripple current irp superimposed on the output current iout is Or the peak ripple current generated in the load current IL Based on the above equation (14a), the amplitude ΔIrp of the charge and discharge current (iin-iout) of the DC capacitor 5 is defined by the following equation (15).

[0075]

[0076] Furthermore, the current components of the carrier frequency of the AC / DC converter 101 and the DC / DC converter 102 flow into and out of the DC capacitor 5. The charge and discharge current of the DC capacitor 5 is not only expressed by the above-mentioned formula (14a), but is actually also defined by the total sum of other frequency components such as the carrier frequency component. In particular, the carrier frequency is dominant and is much larger than the power supply frequency of the AC power supply 1. The current components of the carrier frequency flowing into and out of the DC capacitor 5 are constant and do not depend on the frequency component twice that of the AC power supply 1. That is, in the charge and discharge current of the DC capacitor 5, the current component shown in formula (14a) fluctuates, but the current component of the carrier frequency is constant. Therefore, in this embodiment, the current component shown in formula (14a) is suppressed, and the ripple current converges to the current component of the carrier frequency.

[0077] As described above, the DC capacitor 5 outputs the ripple current irp shown in the above equation (8), thereby suppressing the ripple voltage vc2 generated in the DC capacitor 5 based on the above equation (11), and suppressing the charge and discharge current (iin - iout) compensated by the DC capacitor 5 based on the above equation (14a). The ripple current irp output by the DC capacitor 5 is a sinusoidal wave with a frequency twice that of the voltage vac of the AC power supply 1. The ripple current irp is a current whose initial phase is set so that it reaches its minimum value at the zero-crossing phase and reaches its maximum value at the peak phase. The ripple current irp is output from the DC capacitor 5 to the DC / DC converter 102.

[0078] Next, control of the AC / DC converter 101 and the DC / DC converter 102 by the control circuit 30 will be described. Figure 11 This is a control block diagram illustrating the generation of gate signals for the AC / DC converter 101 in the control circuit 30. In controlling the AC / DC converter 101, the control circuit 30 controls the current iac input from the AC power supply 1 to a power factor of 1 relative to the voltage vac of the AC power supply 1. Furthermore, the control circuit 30 selectively controls the voltage of the DC capacitor 5. When the control circuit 30 implements constant voltage control to maintain a constant voltage vc1 across the DC capacitor 5, a PI control unit 37 performs PI control on the difference 36 between the DC voltage command value Vdc* and the detected voltage vc1, obtained by the addition and subtraction unit 35, to calculate a current command amplitude 38. A multiplication unit 39 then multiplies the current command amplitude 38 by a sinusoidal signal sinωt in phase with the voltage vac of the AC power supply 1 to calculate a current command 40.

[0079] On the other hand, when controlling AC / DC converter 101, if control circuit 30 does not perform constant voltage control of voltage vc1 across DC capacitor 5 but only implements high power factor control of current iac, a current command Iac* is prepared. A switch 41 selects a current command 42, either current command 40 or current command Iac*, depending on whether constant voltage control of DC capacitor 5 is being implemented. A PI control unit 45 then performs PI control on a deviation 44 between current command 42 and detected current iac, obtained by an adder / subtractor 43, to calculate a voltage command value 46. A divider 47 then divides this by the DC voltage component Vdc across DC capacitor 5 to calculate a duty cycle 48. Based on duty cycle 48, a PWM circuit 49 generates a gate signal 50 for PWM control of AC / DC converter 101. In PWM circuit 49, the carrier frequency can be arbitrarily adjusted, and a triangular wave, sawtooth wave, or the like is used as the carrier.

[0080] Figure 12 This is a control block diagram illustrating the generation of gate signals for each semiconductor switching element 3a and 3b within the AC / DC converter 101 in the control circuit 30. Gate signal 50 is input to selectors 53 for semiconductor switching element 3b and 60 for semiconductor switching element 3a, respectively. Polarity determiner 54 determines the polarity of voltage vac from AC power source 1 and outputs signal 55, which is 1 if voltage vac is positive and 0 if voltage vac is negative. Based on signal 55 from polarity determiner 54, selector 53 selects gate signal 50 if voltage vac is positive, and selects signal 50, which is the polarity-inverted gate signal 50 by polarity inverter 51, as the gate signal for semiconductor switching element 3b if voltage vac is negative. Furthermore, selector 60 selects gate signal 50 or signal 59, which is the polarity-inverted gate signal 50 by polarity inverter 58, based on signal 57 obtained by inverting signal 55 by inverter 56. Specifically, when voltage vac is negative, gate signal 50 is selected, and when voltage vac is positive, signal 59, obtained by inverting the polarity of gate signal 50 by polarity inverter 58, is selected as the gate signal for semiconductor switch element 3a. Furthermore, a dead time period can be provided for the gate signals of semiconductor switch elements 3a and 3b, during which both semiconductor switch elements 3a and 3b are turned off to prevent an arm short circuit. Furthermore, when voltage vac is positive, semiconductor switch element 3a can be fixed off, and when voltage vac is negative, semiconductor switch element 3b can be fixed off.

[0081] Therefore, the control circuit 30 follows Figure 11The control shown in FIG. 1 performs high power factor control of the current iac and generates a duty cycle 48 for constant voltage control of the DC capacitor 5 as needed to generate a gate signal 50 for the AC / DC converter 101. Then, the control circuit 30 performs high power factor control of the current iac and generates a duty cycle 48 for constant voltage control of the DC capacitor 5 as needed to generate a gate signal 50 for the AC / DC converter 101. Figure 12 In the control shown, the gate signals of the semiconductor switching elements 3 a and 3 b are controlled according to the polarity of the voltage vac of the AC power supply 1 .

[0082] Figure 13 1 is a control block diagram showing the generation of gate signals of the DC / DC converter 102 in the control circuit 30. The command value IL* of the load current IL output to the load 11 is a DC current command of only the DC component. Figure 13 In FIG, a constant current control is provided to the load 11 for providing a constant DC current IL. Figure 13 As shown, the amplitude calculator 61 calculates the ripple current peak value 61a based on the command value IL* and the ambient temperature Tcon of the DC capacitor 5. The above equation (11) is a theoretical formula for reducing the ripple voltage vc2 of the DC capacitor 5. The ripple current peak value 61a is calculated using the equation (11). If the above equation (12) obtained from the equation (11) is transformed, the ripple current peak value It can be calculated based on the amplitude ΔVdc of the ripple voltage vc2 of the DC capacitor 5. The command value The target value △Vdc* of the amplitude of the ripple voltage vc2 can be calculated according to the following formula (16), and Let this be a ripple current peak value 61 a . In this case, the ripple current peak value 61 a is theoretically a target peak value of the ripple current irp superimposed on the output current iout of the DC capacitor 5 .

[0083]

[0084] A limit value 62a is set for the ripple current peak value 61a. Comparator 61 compares the command value IL* of the load current IL with the limit value Lim previously set for load 11 and outputs the lower value as limit value 62a. Limiting with command value IL* prevents the instantaneous current supplied to load 11 from falling below 0A, thereby entering discontinuous mode. For example, assuming a battery is load 11, limit value Lim set for load 11 is determined based on the reduction in battery life caused by heat generation due to an increase in the AC component.

[0085] Comparator 63 compares ripple current peak value 61a with limit value 62a, and comparator 63 outputs the lower value as AC current command amplitude 64. Control circuit 30 uses multiplication unit 65 to multiply amplitude 64 by a function P represented by equation (17) below, where the angular frequency of AC voltage vac shown in equation (3) is ω, to calculate ripple current command 66, which is the AC current command. For example, based on the temperature characteristics of DC capacitor 5, control circuit 30 sets the AC current command amplitude to an amount that is greater than the amount by which the ripple voltage of DC capacitor 5 increases while the capacitance decreases to the maximum within the range of ambient temperature Tcon of DC capacitor 5.

[0086] P=cos(2ωt-π)…(17)

[0087] The control circuit 30 adds the calculated ripple current command 66 to the command value IL* of the load current IL using the adding unit 67 to generate a current command value 68 including the ripple current as an output current command for the DC / DC converter 102 .

[0088] Next, the current command value 68 is compared with the detected load current IL, and the PI control unit 71 performs PI control on the difference 70 between the two values obtained by the addition and subtraction unit 69 to calculate the voltage command value 72. The voltage command value 72 is then divided by the DC voltage VL of the load 11 by the division unit 73 to calculate the duty cycle 74. In the PWM circuit 75, a gate signal 76 for PWM control of the DC / DC converter 102 is generated using a carrier signal based on the duty cycle 74. Figure 12 In the same processing as that of the control block shown, gate signals are generated for the semiconductor switching elements 6 a to 6 d in the DC / DC converter 102 .

[0089] Next, a method for calculating the ripple current peak value 61 a based on the ambient temperature Tcon of the DC capacitor 5 by the amplitude calculator 61 and its effects will be described.

[0090] Figure 14 、 Figure 15 The currents flowing through the semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9 constituting the DC / DC converter 102 are shown when control is performed in which the ripple current command is not added to the output current IL to the load 11 and when control is performed in which the ripple current command is added. Figure 14 In the embodiment, the envelope of the current flowing through the semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9 is constant. In contrast, the output current IL to the load 11 is controlled by adding the ripple current instruction. Figure 15In the envelope of the current flowing through the semiconductor switching elements 6 a to 6 d , the transformer 7 , and the reactor 9 , a ripple component superimposed on the output current IL to the load 11 appears.

[0091] This ripple component increases the effective value Irms of the current flowing through semiconductor switching elements 6a-6d, transformer 7, and reactor 9. If the resistance value is R, the conduction loss calculated as R·Irms² increases. The temperature increases of semiconductor switching elements 6a-6d, transformer 7, and reactor 9 due to the ripple component superimposed on output current IL are denoted by ΔTSW, ΔTTR, and ΔTRe, respectively.

[0092] also, Figure 16 In the figure, for example, the DC superposition characteristics are shown when a ferrite core is used as the core of the reactor 9. The ferrite core is rapidly saturated due to the change in magnetic flux, so the inductance of the reactor using the ferrite core decreases sharply with respect to DC. The current at which the inductance decreases is set to I1. Generally, the higher the temperature, the lower the saturation magnetic flux density of the ferrite core. Therefore, the higher the temperature, the smaller the current value I1 at which the inductance begins to decrease. For example, if the current at which the inductance decreases at the design maximum temperature Tmax of the reactor 9 is set to I1h, and the maximum value of the ripple superposition amount is added, The load current value after the load current is set as I1r, the temperature of the reactor 9 that can ensure the load current value I1r is set as T1, and the difference from the design maximum temperature Tmax is set as the reactor DC superposition ensuring temperature ΔTRe′ (=Tmax-T1).

[0093] On the other hand, when an aluminum electrolytic capacitor is used as the DC capacitor 5, for example, generally, Figure 17 As shown in the temperature characteristics, the capacitance tends to decrease as the ambient temperature of the capacitor decreases. In addition, the representative capacitance value Ctyp is often defined by the representative ambient temperature Tcon1 (e.g., room temperature), and the capacitance decreases below the representative ambient temperature Tcon1.

[0094] Figure 18 An example of a method for calculating the ripple current peak value 61a based on the ambient temperature Tcon of the DC capacitor 5 is shown in FIG. When the ambient temperature Tcon of the DC capacitor 5 is less than the threshold value Tth, the value shown in equation (16) is set as the ripple current peak value. Furthermore, when the ambient temperature Tcon of the DC capacitor 5 is greater than the threshold value Tth, the ripple current peak value 61a is set to zero.

[0095] Here, the threshold value Tth is set to be above the representative ambient temperature Tcon1 that defines the representative capacitance value Ctyp of the DC capacitor 5. Furthermore, the threshold value Tth is set to be below T1 (=Tcon_max - ΔTmax), which is obtained by subtracting ΔTmax from the maximum ambient temperature Tcon_max of the DC capacitor 5. ΔTmax is defined as the maximum of any one of the temperature increases ΔTSW, ΔTTR, ΔTRe, and the reactor DC superimposition ensuring temperature ΔTRe' caused by the ripple component superimposed on the load current IL. By setting the threshold value Tth in this manner and superimposing the ripple component Irp (ripple amount) on the load current IL at or below the representative ambient temperature Tcon1 that defines the representative capacitance value Ctyp of the DC capacitor 5, the DC capacitor 5 can be made smaller and less expensive without increasing the capacitance of the DC capacitor 5 due to capacitance drop below the representative ambient temperature Tcon1.

[0096] Furthermore, the threshold value Tth is set to be equal to or less than T1. When the temperature increase ΔTSW, ΔTTR, ΔTRe of the semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9 due to the ripple superimposition on the load current IL is subtracted from the maximum ambient temperature of the DC capacitor 5 by the amount of the temperature increase ΔTSW, ΔTTR, ΔTRe, and the reactor DC superimposition ensuring temperature ΔTRe', the ripple component Irp superimposed on the load current IL is set to zero. This eliminates the temperature increase of the semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9 at high temperatures and the insufficient DC superimposition inductance of the reactor 9. Therefore, there is no need to increase the size of the semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9 or to enhance the cooling capacity. The semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9 can be miniaturized and reduced in cost.

[0097] As described above, in this embodiment, the control circuit 30 uses a current command value 68 obtained by superimposing a ripple current command 66 having a frequency twice that of the AC power supply 1, which is minimum at the zero-crossing phase and maximum at the peak phase, on the DC current command (command value IL*) supplied to the load 11. If the ambient temperature Tcon of the DC capacitor 5 is less than the threshold value Tth, the amplitude 64 of the ripple current command 66 uses the amplitude calculated by equation (16). If the ambient temperature Tcon of the DC capacitor 5 is greater than the threshold value Tth, the amplitude 64 is set to zero, thereby controlling the current of the DC / DC converter 102. This suppresses the increase in high-temperature losses of the semiconductor switching elements 6a to 6d, transformer 7, and reactor 9 that constitute the DC / DC converter 102, which is caused by the superimposition of the ripple current on the DC current supplied to the load 11. By suppressing the increase in the DC superimposition inductance required for the reactor 9 at high temperatures, the required capacitance of the DC capacitor, whose capacitance decreases at low temperatures, can be reduced without increasing the size of the semiconductor switching elements 6a to 6d, transformer 7, and reactor 9. Therefore, the power conversion device can be made smaller and less expensive.

[0098] This embodiment shows an example in which the amplitude 64 of the ripple current command 66 is calculated using equation (16) when the ambient temperature Tcon of the DC capacitor 5 is less than the threshold value Tth. If the ambient temperature Tcon of the DC capacitor 5 is greater than the threshold value Tth, the amplitude 64 is set to zero, thereby controlling the current of the DC / DC converter 102. However, the amplitude 64 may also be set to be proportional to the amount of capacitor capacitance drop below the threshold value Tth. As shown in equation (16), the required ripple amount Irp and the capacitor capacitance Cdc are proportional to each other for the required ΔVdc. By performing such control, the ripple current superimposed on the DC current supplied to the load 11 can be kept to the minimum required, thereby suppressing the deterioration in efficiency caused by increased losses in the semiconductor switching elements 6a to 6d, the transformer 7, and the reactor 9.

[0099] In this embodiment, when determining the upper limit T1 of the threshold value Tth for the ambient temperature Tcon of the DC capacitor 5, the value is determined by subtracting ΔTmax from the maximum ambient temperature Tcon_max of the DC capacitor 5. ΔTmax is the maximum value of any one of the temperature increases ΔTSW, ΔTTR, ΔTRe, and the reactor DC superimposition guaranteed temperature ΔTRe' due to the ripple component superimposed on the load current IL. However, if the power conversion device is cooled by cooling water, the value may be determined by subtracting ΔTmax from the maximum cooling water temperature Twater_max. In a power conversion device cooled by cooling water for high power density applications, the temperature inside the power conversion device (=the ambient temperature of the DC capacitor 5) becomes higher than the cooling water temperature due to the influence of heat-generating components such as the semiconductor switching elements 6a-6d, the transformer 7, and the reactor 9. Furthermore, the temperatures of heat-generating components such as the semiconductor switching elements 6a-6d, transformer 7, and reactor 9 are cooled by the cooling water and are therefore highly correlated with the cooling water temperature. Therefore, by setting the upper limit T1 of the threshold value Tth for the ambient temperature Tcon of the DC capacitor 5 to the value obtained by subtracting ΔTmax from the maximum cooling water temperature Twater_max, it is possible to more reliably suppress increases in losses in the semiconductor switching elements 6a-6d, transformer 7, and reactor 9 at high temperatures, and to suppress increases in the DC superposition inductance required for the reactor 9 at high temperatures.

[0100] This embodiment illustrates an example in which a thermistor is provided around the DC capacitor 5 as the DC capacitor temperature acquisition unit 20 for measuring the ambient temperature Tcon of the DC capacitor 5. However, a temperature sensor related to the ambient temperature Tcon of the DC capacitor 5, such as a substrate temperature sensor, a temperature sensor for measuring the internal temperature of the power converter, or a temperature sensor for measuring the cooling water temperature, may also be used. This eliminates the need for a separate temperature sensor for measuring the ambient temperature Tcon of the DC capacitor 5, enabling a more compact and cost-effective power converter.

[0101] This embodiment shows an example using MOSFETs as semiconductor switching elements 6a-6d. However, in power conversion devices using wide-bandgap semiconductors such as GaN (Gallium Nitride), high-frequency switching results in miniaturization and higher power density. Consequently, the temperature increases ΔTSW, ΔTTR, and ΔTRe of the semiconductor switching elements 6a-6d, transformer 7, and reactor 9 caused by the ripple component of load current IL are significant. Therefore, as shown in this embodiment, the amplitude 64 of the ripple current command 66 is set to zero when the ambient temperature Tcon of the DC capacitor 5 is above a threshold value Tth, and current control is performed on the DC / DC converter 102. This reduces the increase in losses at high temperatures in the semiconductor switching elements 6a-6d, transformer 7, and reactor 9 that constitute the DC / DC converter 102, which is caused by the ripple current superimposed on the DC current supplied to the load 11. This significantly reduces the size and cost of the power conversion device.

[0102] In this embodiment, a totem-pole circuit is shown as an example of AC / DC converter 101, but the present invention is not limited to this. Circuits such as a monolithic, interleaved, or semi-bridgeless type may also be used. Furthermore, an isolated full-bridge converter is shown as an example of DC / DC converter 102, but the present invention is not limited to this. A half-bridge or non-isolated DC / DC converter may also be used.

[0103] This application describes exemplary embodiments, but the various features, aspects, and functions described in the embodiments are not limited to application to specific embodiments and can be applied to the embodiments alone or in various combinations.

[0104] Therefore, numerous modifications not shown in the examples are considered to be within the technical scope disclosed in the present specification, including, for example, modifications, additions, or omissions of at least one component.

[0105] Description of labels

[0106] 1 AC power supply

[0107] 3a, 3b, 6a to 6d semiconductor switching elements

[0108] 5 DC capacitors

[0109] 20 DC capacitor temperature acquisition unit

[0110] 101 AC / DC Converter

[0111] 102 DC / DC converter

[0112] 30 Control circuit

[0113] 31, 32, 50, 76 Gate signals.

Claims

1. A power conversion device, include: AC / DC converters that convert AC power from an AC power source into DC power; a DC / DC converter connected to the DC side of the AC / DC converter and performing voltage conversion on the DC power output to the load; a DC capacitor connected between the AC / DC converter and the DC / DC converter for filtering power; and a control circuit for controlling the AC / DC converter and the DC / DC converter, wherein the control circuit superimposes an AC current instruction that is a minimum value at the zero-crossing phase of the AC power supply and a maximum value at the peak phase onto a DC current instruction to generate an output current instruction for the DC / DC converter, and uses the output current instruction to control the output of the DC / DC converter. The power conversion device is characterized in that: The control circuit includes a DC capacitor temperature acquisition unit for acquiring the ambient temperature of the DC capacitor. The control circuit calculates a ripple current value based on the amplitude of the ripple voltage of the DC capacitor according to a load current command to be output to the load and the ambient temperature of the DC capacitor to obtain the amplitude of the AC current command. If the ambient temperature is above a threshold, the control circuit sets the amplitude of the AC current command to be less than a predetermined value. If the ambient temperature is below the threshold, the control circuit sets the amplitude of the AC current command to be greater than the predetermined value.

2. The power conversion device according to claim 1, wherein: When the ambient temperature is equal to or higher than a threshold value, the control circuit sets the amplitude of the AC current command to zero.

3. The power conversion device according to claim 1, wherein: The control circuit sets the amplitude of the AC current command to an amplitude proportional to a reduction in capacitance of the DC capacitor.

4. The power conversion device according to claim 1, wherein: The DC / DC converter includes a reactor formed of a ferrite core.

5. The power conversion device according to claim 1, wherein: The DC / DC converter includes an isolation transformer for isolating input and output.

6. The power conversion device according to claim 1, wherein: The DC capacitor is composed of an aluminum electrolytic capacitor.

7. The power conversion device according to any one of claims 1 to 6, characterized in that: The DC / DC converter includes a switching element using a wide bandgap semiconductor as a switching element for power conversion.

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