A control method for a dual-input boost inverter

Abstract

The present invention discloses a dual-input boost inverter and its control method, belonging to the technical field of converters. In one solution of the present invention, one end of the first input side is connected to the anode of D1, the cathode of D1 is connected to one end of L1, the other end of L1 is connected to terminal 1 of S1, one end of L3, and the anode of D3, the cathode of D3 is connected to terminal 2 of S3 and S4, the cathode of D4, and the positive terminal of C1, the anode of D4 is connected to terminal 1 of S2, one end of L4, and one end of L2, the other end of L2 is connected to the cathode of D2, the anode of D2 is connected to one end of the second input side; the other ends of the first input side and the second input side, terminal 2 of S1 and S2, and the negative terminal of C1 are all grounded; the other end of L3 is connected to terminal 1 of S3 at point a, and the other end of L4 is connected to terminal 1 of S4 at point b; nodes a and b form the output side. The present invention uses dual power supplies, can achieve seamless switching of the two power supplies, ensure the normal operation of the inverter, improve the stability of the system, and obtain a higher voltage gain.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic converters, and particularly to a dual-input boost inverter and a control method thereof. Background Art

[0002] With the gradual depletion of fossil energy, clean and renewable energy power generation technologies such as photovoltaic, wind energy, and hydrogen energy have received great attention worldwide. Inverters are extremely important components in the field of new energy power generation. Therefore, the research on high-performance inverters has important practical significance. For renewable energy power generation systems with low-voltage inputs, due to the advantages of small volume, low cost, and high power generation efficiency of non-isolated boost photovoltaic inverters, they have become a research hotspot for current scholars and the industry. Commonly used boost inverter circuits usually adopt a two-stage structure. If photovoltaic power generation is carried out through a cascaded structure composed of two-stage converters, it is easy to form the "barrel effect", reducing the reliability of the system. At the same time, the number of required power devices increases, and the two-stage power processing will also affect the transmission efficiency of the system.

[0003] The boost inverter is an inverter topology that has been studied and applied more in recent years. This topology usually multiplexes the boost and inverter units, reducing the number of switching tubes. At the same time, it realizes the boost and inverter functions for distributed energy, improving the power density and efficiency of the system. However, these topologies are all single-input topologies. If a single-input boost converter is applied in occasions such as photovoltaic power generation, since the photovoltaic cells can hardly provide power in rainy weather or at night, the system is in an idle state during this time range, greatly reducing the utilization rate of power generation equipment and the reliability of the power generation system. Therefore, dual-input or multi-input inverters have received attention.

[0004] Proceedings of the CSEE, Vol. 34, No. 6, publication date: February 25, 2014, published a paper "A Novel Single-Stage Non-Isolated Dual-Cuk Inverter", authors: Wang Liqiao, Wang Xin, Qiu Lei. This inverter is composed of two improved Cuk DC converters through input series and output parallel combination. This inverter can adapt to a wide range of input DC voltages, has 2 independent input power supplies, and also has buck-boost capabilities, suitable for application occasions with wide-range fluctuations in input voltage. And in this circuit, the AC output side is grounded with one of the input power supplies, solving part of the leakage current problem. However, this circuit uses more devices, increasing the system loss, and this inverter cannot operate in single-power supply mode, greatly reducing the reliability of the system.

[0005] In the prior art, multiple documents are related to the research of dual-input boost inverters. For example: 1) The title of the paper is "Research on the Half-Cycle Modulation Strategy of Dual Boost Inverters", and the authors are Xu Fei, Tang Yu, etc.; it was published in the "Proceedings of the Chinese Society for Electrical Engineering" in December 2014, Vol. 34, No. 36. 2) The title of the paper is "Research on a Novel Dual Boost Inverter Power Supply", and the authors are Chen Zhe, Cui Yulong; it was published in "Power Electronics" in August 2013, Vol. 47, No. 8. The application scenarios of the above paper structures are limited and cannot meet the wide range of requirements. Summary of the Invention

[0006] 1. Technical problems to be solved by the invention

[0007] Aiming at the problem of low boost ratio in the boost inverter of the prior art, the present invention provides a dual-input boost inverter and its control method. The present invention uses dual power supplies, can achieve seamless switching between dual power supply and single power supply, ensure the normal operation of the inverter, improve the stability of the system, and can obtain a higher voltage gain.

[0008] 2. Technical solutions

[0009] To achieve the above object, the technical solutions provided by the present invention are as follows:

[0010] In the first aspect, a dual-input boost inverter includes power switching tubes S1, S2, S3, and S4, diodes D1, D2, D3, and D4, a capacitor C1, and inductors L1, L2, L3, and L4; one end of the first input side of a dual-input boost inverter is connected to the anode of diode D1, the cathode of diode D1 is connected to one end of inductor L1, the other end of inductor L1 is connected to terminal 1 of power switching tube S1, one end of inductor L3, and the anode of diode D3, the cathode of D3 is connected to terminal 2 of power switching tubes S3 and S4, the cathode of D4, and the positive end of capacitor C1, the anode of diode D4 is connected to terminal 1 of power switching tube S2, one end of inductor L4, and one end of L2, the other end of L2 is connected to the cathode of diode D2, and the anode of diode D2 is connected to one end of the second input side; the other ends of the first input side and the second input side, terminal 2 of power switching tubes S1 and S2, and the negative end of capacitor C1 are all grounded; the other end of inductor L3 is connected to terminal 1 of power switching tube S3 at point a, and the other end of inductor L4 is connected to terminal 1 of power switching tube S4 at point b; nodes a and b form the output side.

[0011] Further, the power switching tubes S1, S2, S3, and S4 are IGBTs or MOSFETs.

[0012] Further, the first input side and the second input side are respectively connected to the positive and negative poles of a DC power supply.

[0013] Further, the output side formed by the inverter nodes a and b is connected to the input side of the filter, and the output side of the filter is connected to a load or the power grid, as Figure 2 shown.

[0014] Further, the filter is of type I, as Figure 3 shown. One end of the inductor L3 is connected to the node a, the node a is connected to one end of the load R, the other end of the load R is connected to the node b, the node b is connected to one end of the inductor L4, and the nodes a and b form the output side of the filter.

[0015] Further, the filter is of type II, as Figure 4 shown. One end of the inductor L 30 is connected to the node a, the node a is connected to one end of the filter capacitor C0 and one end of the load R, the other ends of the filter capacitor C0 and the load R are connected to the node b, the node b is connected to one end of the inductor L 40 and the two ends of the filter capacitor C0 and the load R form the output side of the filter.

[0016] Further, the filter is of type III, as Figure 5 shown. One end of the inductor L 301 is connected to the node a, the node a is connected to one end of the filter inductor L 501 and one end of the filter capacitor C 01 , the other end of the filter inductor L 501 is connected to one end of the load R, the other ends of the filter capacitor C 01 and the load R are connected to the node b, the node b is connected to one end of the inductor L 401 , and the other end of the filter inductor L 501 and the node b form the output side of the filter.

[0017] In a second aspect, the present invention further provides a control circuit for a dual-input boost inverter, as Figure 19 shown. The voltage on the output side of the filter is used as a feedback voltage, which is compared with a given voltage U ref to obtain an error value. The error value is adjusted by a regulator and then compared with a triangular wave to generate a pulse signal, which controls the terminal 3 of the power switch tubes S1, S2, S3, and S4.

[0018] In a third aspect, the present invention further provides a control method for a dual-input boost inverter. When powered by dual power supplies, the working principle is the same in the positive and negative half-cycles of the sinusoidal modulation wave, and it includes the following working modes:

[0019] In the positive half-cycle of the sinusoidal modulation wave, when the power switch tubes S1 and S4 are controlled to be turned on and S2 and S3 are turned off, the input side one charges the inductor L1; the voltage U C1Power is supplied to the load or the power grid through power switching tubes S1 and S4; when controlling power switching tube S4 to conduct and S1, S2, and S3 to turn off, the first input side and inductor L1 charge capacitor C1 through diode D1, and filter inductor L3 supplies power to the load or the power grid through power switching tube S4 and diode D3. At this time, the current i of inductor L1 L1 linearly decreases until the current i of inductor L1 L1 linearly decreases to zero, and the first input side stops charging capacitor C1.

[0020] In the negative half cycle of the sinusoidal modulation wave, when controlling power switching tubes S2 and S3 to conduct and S1 and S4 to turn off, the second input side charges inductor L2; the voltage U across capacitor C1 C1 supplies power to the load or the power grid through power switching tubes S2 and S3; when controlling power switching tube S3 to conduct and S1, S2, and S4 to turn off, the second input side and inductor L2 charge capacitor C1 through diode D2, and filter inductor L4 supplies power to the load or the power grid through power switching tube S3 and diode D4. At this time, the current i of inductor L2 L2 linearly decreases until the current i of inductor L2 L2 linearly decreases to zero, and the second input side stops charging capacitor C1.

[0021] Furthermore, the present invention also proposes a control method for a dual-input boost inverter. When powered by a single power supply, it includes the following working modes:

[0022] In the positive half cycle of the sinusoidal modulation wave, when controlling power switching tubes S1 and S4 to conduct and S2 and S3 to turn off, the first input side charges inductor L1; the voltage U across capacitor C1 C1 supplies power to the load or the power grid through power switching tubes S1 and S4; when controlling power switching tube S4 to conduct and S1, S2, and S3 to turn off, the first input side and inductor L1 charge capacitor C1 through diode D1, and filter inductor L3 supplies power to the load or the power grid through power switching tube S4 and diode D3. At this time, the current i of inductor L1 L1 linearly decreases until the current i of inductor L1 L1 linearly decreases to zero, and the first input side stops charging capacitor C1.

[0023] In the negative half cycle of the sinusoidal modulation wave, when controlling power switching tubes S2 and S3 to conduct and S1 and S4 to turn off, the voltage U across capacitor C1 C1 supplies power to the load or the power grid through power switching tubes S2 and S3; when controlling power switching tube S3 to conduct and S1, S2, and S4 to turn off, filter inductor L4 supplies power to the load or the power grid through power switching tube S3 and diode D4.

[0024] Furthermore, when powered by a dual power supply, the voltage ratio G between the output side and the input side:

[0025]

[0026] Among them, U om is the voltage amplitude on the output side; the voltage amplitudes of the first input side and the second input side are equal and both are U in ; m is the modulation ratio; f s is the switching frequency; R is the equivalent impedance of the load or power grid connected to the output side of the filter.

[0027] 3. Beneficial effects

[0028] Adopting the technical solution provided by the present invention, compared with the existing well-known technologies, it has the following remarkable effects:

[0029] (1) For a dual-input boost inverter of the present invention, the two input power supplies can supply power jointly or separately, with good flexibility and reliability; and seamless switching between dual-power supply and single-power supply can be achieved, greatly improving the stability of the system, further enhancing the boost ability, and a higher voltage gain can be obtained;

[0030] (2) For a dual-input boost inverter of the present invention, boost inversion can be realized, the value range of the input voltage can vary greatly, and there is a stable AC output; there is no need to set a delay time between drive signals, and at any moment during operation, only one high-frequency switch and one power-frequency switch are working, reducing the conduction loss of the switching tubes; in addition, since the capacitor voltage cannot change suddenly, the capacitor C1 has an inhibitory effect on the input voltage disturbance, and the output is AC, so the value of the capacitor C1 is also relatively flexible;

[0031] (3) For a dual-input boost inverter of the present invention, which belongs to an integrated inverter, significantly reduces the number of components, reduces the system cost and improves the integration degree, the circuit occupies a small space, and overcomes the disadvantage of the complex circuit of the traditional two-stage inverter;

[0032] (4) For a control method of a dual-input boost inverter of the present invention, it has a high boost ability, and realizes the functions of boost and inversion by controlling the on and off of the power switching tubes S1, S2, S3, and S4;

[0033] (5) For a dual-input boost inverter of the present invention, it can largely suppress the generation of leakage current, and has a better effect of suppressing leakage current than the traditional dual-input inverter;

[0034] (6) In a traditional two - stage cascaded boost inverter, filters need to be set at the output end of the front - stage boost converter and also at the output end of the rear - stage inverter. The filters take up a large amount of space and are cumbersome to design, which will undoubtedly increase the volume of the entire circuit and the circuit design cost. The single - stage boost inverter of the present invention creatively overcomes the above - mentioned disadvantages and occupies less space;

[0035] (7) For a dual - input boost - type inverter of the present invention, when the inverter operates in the free - wheeling state, the current flows through the independent diode. Therefore, the switching tubes and diodes in the circuit can be optimally designed respectively, which is beneficial to further improving the performance of the system. BRIEF DESCRIPTION OF THE DRAWINGS

[0036] Figure 1 is a schematic diagram of the circuit structure of Embodiment 1 of the present invention.

[0037] Figure 2 is Figure 1 the preferred implementation.

[0038] Figure 3 is Figure 2 one of the preferred implementations.

[0039] Figure 4 is Figure 2 two of the preferred implementations.

[0040] Figure 5 is Figure 2 three of the preferred implementations.

[0041] Figure 6 is the input waveform diagram of the power switch tube terminal 3 in the circuit structure of Embodiment 1.

[0042] Figure 7 is the schematic diagram of the working state of Mode 1 of the circuit structure of Embodiment 1.

[0043] Figure 8 is the schematic diagram of the working state of Mode 2 of the circuit structure of Embodiment 1.

[0044] Figure 9 is the schematic diagram of the working state of Mode 3 of the circuit structure of Embodiment 1.

[0045] Figure 10 is the schematic diagram of the working state of Mode 4 of the circuit structure of Embodiment 1.

[0046] Figure 11 is the schematic diagram of the working state of Mode 5 of the circuit structure of Embodiment 1.

[0047] Figure 12 is the schematic diagram of the working state of Mode 6 of the circuit structure of Embodiment 1.

[0048] Figure 13 It is a schematic diagram of the working state of Mode 7 of the circuit structure in Embodiment 1.

[0049] Figure 14 It is a schematic diagram of the working state of Mode 8 of the circuit structure in Embodiment 1.

[0050] Figure 15 It is a working timing diagram of the circuit structure in Embodiment 1.

[0051] Figure 16 It is a simulation waveform diagram of the inductor currents when the circuit structure in Embodiment 1 selects to use Filter II.

[0052] Figure 17 It is the voltage U across capacitor C1 when the circuit structure in Embodiment 1 selects to use Filter II. C1 of the simulation waveform diagram.

[0053] Figure 18 It is a dynamic experimental waveform diagram when the circuit structure in Embodiment 1 switches from the dual-power-supply mode to the single-power-supply mode when selecting to use Filter II.

[0054] Figure 19 It is a schematic diagram of the control circuit of the circuit structure in Embodiment 1. Detailed implementation manners

[0055] To further understand the content of the present invention, the present invention will be described in detail in combination with the accompanying drawings and embodiments.

[0056] In the present invention, the output-side voltage is denoted as output voltage U o , and the corresponding output voltage amplitude is U om ; when the power switching transistors S1, S2, S3, and S4 use IGBTs, terminals 1, 3, and 2 of the power switching transistors S1, S2, S3, and S4 respectively correspond to the collector, base, and emitter, and when using MOSFETs, terminals 1, 3, and 2 of the power switching transistors S1, S2, S3, and S4 respectively correspond to the drain, gate, and source.

[0057] Both the first input side and the second input side of the inverter in the present invention are DC power supplies. There are many types of DC power supplies, which can be determined according to specific application methods, scenarios, or fields, and are not limited to the several situations listed in the present invention.

[0058] The present invention only lists several limited implementation manners. According to the needs of actual applications, the technical solutions of the present invention can be widely promoted and applied, such as in the photovoltaic field, energy storage battery field, and household appliance fields such as air conditioners, power tools, sewing machines, TVs, washing machines, range hoods, refrigerators, fans, lighting, or other implementable scenario fields.

[0059] Due to the high popularity of automobiles, when going out to work or travel, the boost inverter proposed by the present invention can be used to connect a storage battery to drive electrical appliances and various tools to work. The in-vehicle inverter output through the cigarette lighter has power specifications of 20W, 40W, 80W, 120W to 150W. The boost-type inverter of the present invention is made into a power converter. The input side of the power converter is connected to the output side of the cigarette lighter, and household electrical appliances are connected to the output side of the power converter, so that various electrical appliances can be used in the car. The electrical appliances that can be used include electric tools, in-vehicle refrigerators, and various tourism, camping, and first-aid medical appliances, etc. At this time, the cigarette lighter serves as the DC power supply.

[0060] Embodiment 1

[0061] As Figure 1 shown, this embodiment proposes a dual-input boost-type inverter, which includes power switching tubes S1, S2, S3, and S4, diodes D1, D2, D3, and D4, capacitor C1, inductors L1, L2, L3, and L4. One end of the first input side of a dual-input boost-type inverter is connected to the anode of diode D1, the cathode of diode D1 is connected to one end of inductor L1, the other end of inductor L1 is connected to terminal 1 of power switching tube S1, one end of inductor L3, and the anode of diode D3. The cathode of D3 is connected to terminal 2 of power switching tubes S3 and S4, the cathode of D4, and the positive end of capacitor C1. The anode of diode D4 is connected to terminal 1 of power switching tube S2, one end of inductor L4, and one end of L2. The other end of L2 is connected to the cathode of diode D2, and the anode of diode D2 is connected to one end of the second input side. The other ends of the first input side and the second input side, terminal 2 of power switching tubes S1 and S2, and the negative end of capacitor C1 are all grounded. The other end of inductor L3 is connected to terminal 1 of power switching tube S3 at point a, and the other end of inductor L4 is connected to terminal 1 of power switching tube S4 at point b. Nodes a and b form the output side.

[0062] Different from the boost inverter composed of the series combination of a boost converter and an inverter in the prior art, the inventor of this embodiment creatively proposes a dual-input boost-type inverter, which overcomes the disadvantage of the complex structure of the traditional two-stage inverter. At the same time, the design of the dual power supply greatly increases the stability of the system. During inversion, the boosting process is also completed, reducing the number of components, thereby reducing the switching loss and cost. Moreover, it has a high integration degree, a small volume, a high boost ratio, and an increased system safety and stability.

[0063] Embodiment 2

[0064] This embodiment relates to a control circuit of a dual-input boost-type inverter, which is applicable to a dual-input boost-type inverter in each of the technical solutions described in Embodiment 1. The voltage of the output side of the inverter is used as the feedback voltage, which is compared with the given voltage U refThe error value is obtained through comparison. After being adjusted by a regulator (the type of regulator can be selected as needed, and the example of a PID regulator is given in the appendix. In actual application, PI, PD, etc. can be selected, which are not limited by the examples listed in this embodiment and the attached drawings), it is compared with a triangular wave to generate a pulse signal, which is input to terminal 3 of power switching transistors S1, S2, S3, and S4. Figure 19 As shown in the appendix, the example of a PID regulator is given. In actual application, PI, PD, etc. can be selected, which are not limited by the examples listed in this embodiment and the attached drawings. After being adjusted, it is compared with a triangular wave to generate a pulse signal, which is input to terminal 3 of power switching transistors S1, S2, S3, and S4.

[0065] Embodiment 3

[0066] This embodiment proposes a control method for a dual-input boost inverter. Control signals are input to the gates of power switching transistors S1, S2, S3, and S4, and the waveforms are as Figure 6 shown. From top to bottom, they are the gate input signals of power switching transistors S1, S2, S3, and S4. At most two switching transistors work simultaneously in any mode. Compared with the literature "ANew High-EfficiencySingle-Phase Transformerless PV Inverter Topology", there is no need to increase the dead time, which improves the quality of the output waveform. In addition, the number of power switching transistors in the comparative literature is two more than that in this embodiment, which will undoubtedly increase the volume of the boost inverter, increase the switching loss, and then reduce the conversion efficiency and service life of the entire boost inverter. However, this embodiment creatively solves the above problems, and its working mode is divided into a dual-power supply mode and a single-power supply mode. The working modes of the dual-power supply mode include Mode 1, Mode 2, Mode 3, Mode 4, Mode 5, and Mode 6, while the working modes of the single-power supply mode include Mode 1, Mode 2, Mode 3, Mode 7, and Mode 8. The details are as follows: The working modes are as Figures 7 - 14 shown, and the timing diagram is as Figure 15 shown, and the filter II is used in the appendix Figures 7 - 14 as shown.

[0067] Mode 1

[0068] As Figure 7 shown, power switching transistors S1 and S4 are turned on, S2 and S3 are turned off, diode D1 is turned on, and diodes D2, D3, and D4 are turned off. The first input side charges inductor L1 through power switching transistor S1 and diode D1, and the inductor current i L1 increases linearly. Capacitor C1 supplies power to the output side through power switching transistors S1 and S4, and the inductor current i L3 increases. The output bridge arm voltage of the inverter is the voltage U C1 across capacitor C1. At this time, the output voltage amplitude U om = +mU C1 , where m is the modulation ratio. In this mode, the second input side does not work.

[0069] Mode 2

[0070] As shown Figure 8 in the figure, power switch tube S4 conducts, S1, S2, and S3 turn off, diodes D1 and D3 conduct, and diodes D2 and D4 turn off; the first input side and inductor L1 charge capacitor C1 through diodes D1 and D3, and the inductor current i L1 decreases linearly. Inductor L3 supplies continuous current to the output side through switch tube S4 and diode D3 until the inductor current i L1 decreases linearly to zero, and Mode 2 ends. In this mode, the second input side does not work.

[0071] Mode 3

[0072] As shown Figure 9 in the figure, power switch tube S4 conducts, S1, S2, and S3 turn off, diode D3 conducts, and diodes D1, D2, and D4 turn off; capacitor C1 maintains the voltage across its terminals unchanged, and inductor L3 continues to supply continuous current to the output side through switch tube S4 and diode D3. The inductor current i L3 decreases. Then, in the positive half-cycle of the sinusoidal modulation wave, Modes 1 - 3 cycle. In this mode, neither the first input side nor the second input side works.

[0073] Mode 4

[0074] As shown Figure 10 in the figure, power switch tubes S2 and S3 conduct, S1 and S4 turn off, diode D2 conducts, and diodes D1, D3, and D4 turn off; the second input side charges inductor L2 through power switch tubes S2 and diode D2, and the inductor current i L2 increases linearly. Capacitor C1 supplies power to the output side through power switch tubes S2 and S3, and the inductor current i L4 increases. The voltage of the output bridge arm of the inverter is the voltage U C1 across capacitor C1. At this time, the output voltage amplitude U om = -mU C1 , where m is the modulation ratio. In this mode, the first input side does not work.

[0075] Mode 5

[0076] As shown Figure 11 in the figure, power switch tube S3 conducts, S1, S2, and S4 turn off, diodes D2 and D4 conduct, and diodes D1 and D3 turn off; the second input side and inductor L2 charge capacitor C1 through diodes D2 and D4, and the inductor current i L2 decreases linearly; inductor L4 supplies continuous current to the output side through switch tube S3 and diode D4 until the inductor current i L2 decreases linearly to zero, and Mode 5 ends. In this mode, the first input side does not work.

[0077] Mode 6

[0078] As Figure 12 shown, the power switch tube S3 is turned on, S1, S2, and S4 are turned off, the diode D4 is turned on, and the diodes D1, D2, and D3 are turned off; the capacitor C1 keeps the voltage across its terminals unchanged, and the inductor L4 supplies power to the load through the switch tube S3 and the diode D4 for freewheeling, and the inductor current i L4 decreases. Then, in the negative half-cycle of the sinusoidal modulation wave, the cycle modes 4-6 are cycled. In this mode, neither the first input side nor the second input side works.

[0079] Mode 7

[0080] As Figure 13 shown, the power switch tubes S2 and S3 are turned on, S1 and S4 are turned off, and the diodes D1, D2, D3, and D4 are turned off; the capacitor C1 supplies power to the output side through the power switch tubes S2 and S3, and the inductor current i L4 increases. In this mode, neither the first input side nor the second input side works.

[0081] Mode 8

[0082] As Figure 14 shown, the power switch tube S3 is turned on, S1, S2, and S4 are turned off, the diode D4 is turned on, and the diodes D1, D2, and D3 are turned off; the capacitor C1 keeps the voltage across its terminals unchanged, and the inductor L4 supplies power to the output side through the switch tube S4 and the diode D4 for freewheeling, and the inductor current i L4 continues to decrease, and Mode 8 ends. Then, in the negative half-cycle of the sinusoidal modulation wave, the cycle modes 7-8 are cycled. In this mode, neither the first input side nor the second input side works.

[0083] The following assumptions are made for simplified analysis: (1) All components in the circuit are ideal devices, that is, the influence of parasitic parameters is not considered; (2) The parameters of the two Boost units are completely symmetrical; (3) The capacitor C1 is large enough so that its terminal voltage U s remains basically unchanged within one switching period T C1 ; (4) The DC power supply U in1 of the first input side and the DC power supply U in2 of the second input side are exactly equal.

[0084] The duty cycle D i of the power switch tube S1 changes sinusoidally in each carrier cycle. That is, in the i-th carrier cycle, the duty cycle of S1 is D i , and m is the modulation ratio. According to the regular symmetric sampling rule, the duty cycle can be obtained as:

[0085] D i = msinωt i (1)

[0086] According to the volt - second balance of inductor L1, we have:

[0087] U in1 D i T S =(U C1 -U in1 )D i ′T S (2)

[0088] The duty cycle d of the power switch tube S2 in each carrier period changes according to a sine law. That is, in the i - th carrier period, the duty cycle of S2 is d i , and m is the modulation ratio. According to the regular - type symmetric sampling rule, the duty cycle can be obtained as: i

[0089] d i =msinωt i (3)

[0090] According to the volt - second balance of inductor L2, we have:

[0091] U in2 d i T S =(U C2 -U in2 )d i ′T S (4)

[0092] For this inverter circuit, if all device losses in the circuit are ignored, the input power is equal to the output power. Thus, we can obtain:

[0093]

[0094] Among them, assuming U in1 =U in2 =U in , U om is the amplitude of the output voltage of the inverter, and U om =mU C1 ;

[0095] In the positive half - cycle of the sinusoidal modulation wave, the average value of the current on inductor L1 is:

[0096]

[0097] Among them, I L1P =U in1 D i T S / L1, which is the change in the current of inductor L1;

[0098] Similarly, in the negative half - cycle of the sinusoidal modulation wave, the average value of the current on inductor L2 is: ​

[0099]

[0100] where I L2P = U in2 d i T S / L2, which is the change in current of inductor L2;

[0101] I in1 is the average current of input side one, and I in2 is the average current of input side two, that is

[0102] I in1 = I L1 / 2 (8)

[0103] I in2 = I L2 / 2 (9)

[0104] According to equations (1) to (9), it can be solved that the relationship between the input voltage and the DC bus voltage is:

[0105]

[0106] where D i , d i take the effective values, then Then the voltage ratio G between the output side and the input side is:

[0107]

[0108] where U om is the amplitude of the output side voltage; the voltage amplitudes of input side one and input side two are equal, both being U in ; m is the modulation ratio; T S is the modulation period; R is the equivalent impedance of the load or power grid connected to the output side of the filter.

[0109] After comparison, the working principle of a dual-input boost inverter in this embodiment is different from that of a traditional dual-input inverter, mainly including the following points:

[0110] 1. The two input power supplies of a dual-input boost inverter in this embodiment can supply power jointly or separately, with good flexibility and reliability;

[0111] 2. A dual-input boost inverter in this embodiment can achieve seamless switching between dual-power supply and single-power supply, and the system operates more reliably;

[0112] 3. A dual-input boost inverter in this embodiment is suitable for the application scenario of combined power generation of two low-voltage renewable energy sources, and the amplitudes of the two low-voltage DC input power supplies can be different;

[0113] 4. The power semiconductor devices used in the dual-input boost inverter of this embodiment are few, and only one power switch operates at a high frequency during the entire working process, which is easy to improve the power density and reduce the system cost;

[0114] 5. The dual-input boost inverter of this embodiment can largely suppress the generation of leakage current, and has a better effect of suppressing leakage current than traditional dual-input inverters;

[0115] 6. The dual-input boost inverter of this embodiment realizes the boost effect while inverting.

[0116] The two-stage combination of traditional DC-DC converters and inverters can also achieve the boost and inversion effects. For example, the front stage is a BOOST converter and the rear stage is a full-bridge inverter. The following technical problems exist in comparing this solution with the technical solution of this embodiment:

[0117] 1. The dual-input boost inverter of this embodiment is integrated and single-stage, and does not need to consider the mutual matching problem between the output end of the boost converter and the input end of the inverter, with high integration;

[0118] 2. The dual-input boost inverter of this embodiment does not have the problem that the traditional two-stage series boost inverter still needs to consider the matching between the front-stage output and the rear-stage input, reducing the design cost;

[0119] 3. A filter needs to be set at the output end of the front-stage boost converter of the traditional two-stage series boost inverter, and a filter also needs to be set at the output end of the rear-stage inverter. The filter occupies a large space and the design is cumbersome, which will undoubtedly increase the volume of the entire circuit and the circuit design cost. The dual-input boost inverter of this embodiment overcomes the above disadvantages and only requires one filter, occupying a small space;

[0120] 4. For the two-stage combination of traditional DC-DC converters and inverters, in the control circuit, two control loops are often adopted to control the front-stage DC-DC converter and the rear-stage inverter respectively. In addition, when designing and controlling the two control circuits, the problem of matching between the front stage and the rear stage in terms of control effect needs to be considered, resulting in a complex control circuit structure, high design difficulty, high design cost, time-consuming control process, and inconvenient operation; while the proposed dual-input boost inverter of this embodiment creatively solves the above technical problems: In view of the characteristics of the circuit structure of this embodiment, only one control loop is required to control the circuit, and there is no technical problem of matching between control loops, simplifying the control circuit structure, reducing the design cost, and making the control process convenient.

[0121] To verify the technical effects of a dual-input boost inverter in this embodiment, parameter selection for each component was carried out as shown in Table 1, and the circuit topology was constructed on Matalab for circuit simulation. Figure 16 It is the simulation waveform diagram of each inductor current. Figure 17 It is the voltage U across capacitor C1 C1 of the simulation waveform diagram. Figure 18 It is the dynamic experimental waveform diagram when switching from the dual-input mode to the single-input mode. From this, it can be seen that a dual-input boost inverter in this embodiment realizes the functions of boosting and inverting, the sinusoidality of the output waveform is good, and at the same time, seamless switching between dual-power supply and single-power supply can be achieved, with good dynamic performance.

[0122] Table 1 Parameter Selection of Circuit Components

[0123] Parameter Value Parameter Value <![CDATA[Input voltage U in / V]]> 45-60 <![CDATA[Filter inductor L3 / mH]]> 3 <![CDATA[Output voltage U o / V]]> 110 <![CDATA[Capacitance C1 / μF]]> 220 <![CDATA[Rated power P o / W]]> 200 <![CDATA[Output capacitor C o / μF]]> 5 <![CDATA[Input inductors L1, L2 / μH]]> 150 <![CDATA[Switching frequency f s / kHz]]> 20

[0124] The above schematically describes the present invention and its embodiments. This description is not restrictive. What is shown in the drawings is only one of the embodiments of the present invention, and the actual structure is not limited thereto. Therefore, if those of ordinary skill in the art are inspired by it and design similar structural ways and embodiments to this technical solution without creative work without departing from the purpose of the present invention, they shall fall within the protection scope of the present invention.

Claims

1. A control method for a dual-input boost inverter, characterized in that: The dual-input boost inverter includes power switch tubes S1, S2, S3, and S4, diodes D1, D2, D3, and D4, a capacitor C1, and inductors L1, L2, L3, and L4; One end of the first input side is connected to the anode of diode D1, and the cathode of diode D1 is connected to one end of inductor L1; The other end of inductor L1 is connected to terminal 1 of power switch tube S1, one end of inductor L3, and the anode of diode D3; The cathode of diode D3 is connected to terminal 2 of power switch tubes S3 and S4, the cathode of diode D4, and the positive terminal of capacitor C1; The anode of diode D4 is connected to terminal 1 of power switch tube S2, one end of inductor L4, and one end of inductor L2. The other end of inductor L2 is connected to the cathode of diode D2; The anode of diode D2 is connected to one end of the second input side; the other ends of the first input side and the second input side, terminal 2 of power switch tubes S1 and S2, and the negative terminal of capacitor C1 are all grounded; The other end of inductor L3 is connected to terminal 1 of power switch tube S3 at point a, and the other end of inductor L4 is connected to terminal 1 of power switch tube S4 at point b; Nodes a and b form the output side; When powered by dual power supplies, the operating principle is the same during the positive and negative half-cycles of the sinusoidal modulation wave, including: During the positive half-cycle of the sinusoidal modulation wave, control the power switch tubes S1 and S4 to conduct, and S2 and S3 to turn off. The input side charges the inductor L1; the voltage U across the capacitor C1 C1 supplies power to the load or the grid through the power switch tubes S1 and S4; control the power switch tube S4 to conduct, and S1, S2, and S3 to turn off. The input side 1 and the inductor L1 charge the capacitor C1 through the diode D1, and the filter inductor L3 supplies power to the load or the grid through the power switch tube S4 and the diode D3. At this time, the current i of the inductor L1 L1 decreases linearly until the current i of the inductor L1 L1 decreases linearly to zero, and the input side 1 stops charging the capacitor C1; During the negative half-cycle of the sinusoidal modulation wave, control the power switch tubes S2 and S3 to conduct, and S1 and S4 to turn off, so that the input-side two inductors L2 are charged; the voltage U across the capacitor C1 C1 powers the load or the power grid through the power switch tubes S2 and S3; control the power switch tube S3 to conduct, and S1, S2, and S4 to turn off, so that the input-side two and the inductor L2 charge the capacitor C1 through the diode D2, and the filter inductor L4 powers the load or the power grid through the power switch tube S3 and the diode D4. At this time, the current i of the inductor L2 L2 decreases linearly until the current i of the inductor L2 L2 decreases linearly to zero, and the input-side two stops charging the capacitor C1.

2. The control method of a dual-input boost inverter according to claim 1, characterized in that: When powered by a single power supply, it includes: During the positive half-cycle of the sinusoidal modulation wave, control the power switch tubes S1 and S4 to conduct, and S2 and S3 to turn off. The input side charges the inductor L1; the voltage U across the capacitor C1 C1 supplies power to the load or the power grid through the power switch tubes S1 and S4; when controlling the power switch tube S4 to conduct and S1, S2, and S3 to turn off, the input side one and the inductor L1 charge the capacitor C1 through the diode D1, and the filter inductor L3 supplies power to the load or the power grid through the power switch tube S4 and the diode D3. At this time, the current i of the inductor L1 L1 decreases linearly until the current i of the inductor L1 L1 decreases linearly to zero, and the input side one stops charging the capacitor C1; During the negative half-cycle of the sinusoidal modulation wave, when the power switch tubes S2 and S3 are controlled to conduct and S1 and S4 are turned off, the voltage U across the capacitor C1 C1 supplies power to the load or the power grid through the power switch tubes S2 and S3; when the power switch tube S3 is controlled to conduct and S1, S2, and S4 are turned off, the filter inductor L4 supplies power to the load or the power grid through the power switch tube S3 and the diode D4.

3. The control method of a dual-input boost inverter according to claim 2, wherein: When powered by dual power supplies, the voltage ratio G between the output side and the input side: Among them, U om is the voltage amplitude on the output side; the voltage amplitudes of the first input side and the second input side are equal, both being U in ; m is the modulation ratio; f s is the switching frequency; R is the equivalent impedance of the load or power grid connected to the output side of the filter.

4. The control method of a dual-input boost inverter according to claim 1, characterized in that: The power switch tubes S1, S2, S3, and S4 are IGBTs or MOSFETs; The first input side and the second input side are respectively connected to the positive and negative poles of the DC power supply.

5. The control method of a dual-input boost inverter according to claim 4, characterized in that: The output side formed by the inverter nodes a and b is connected to the input side of the filter, and the output side of the filter is connected to the load or the power grid.

6. The control method of a dual-input boost inverter according to claim 5, characterized in that: The filter is of type I. One end of inductor L3 is connected to node a. Node a is connected to one end of load R. The other end of load R is connected to node b. Node b is connected to one end of inductor L4. Nodes a and b form the output side of the filter.

7. The control method of a dual-input boost inverter according to claim 5, characterized in that: The filter is of type II. One end of inductor L3 is connected to node a. Node a is connected to one end of filter capacitor C0 and one end of load R. The other ends of filter capacitor C0 and load R are connected at node b. Node b is connected to one end of inductor L4. The two ends of filter capacitor C0 and load R form the output side of the filter.

8. A control method for a dual-input boost inverter according to claim 5, characterized in that: The filter described is of type III. One end of the inductor L3 is connected to node a, and node a is connected to one end of the filtering inductor L 501 , one end of the filtering capacitor C 01 . The other end of the filtering inductor L 501 is connected to one end of the load R. The other ends of the filtering capacitor C 01 and the load R are connected to node b. Node b is connected to one end of the inductor L 40 . The other end of the filtering inductor L 501 and node b form the output side of the filter.

9. A control method for a dual-input boost-type inverter according to any one of claims 6-8, characterized in that: In the control circuit of the inverter, the voltage on the output side of the filter is used as the feedback voltage, which is compared with the given voltage U ref to obtain an error value. After being adjusted by a regulator, the error value is compared with a triangular wave to generate a pulse signal, which controls the terminal 3 of the power switching transistors S1, S2, S3, and S4.

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