Power conversion device

Abstract

A power conversion device (20, 90, 100) has input-side terminals (CH1, CL1), output-side terminals (CH1, CL1), and a switching section (Q5-Q8, Q11) that is turned on and off in the case of power transmission from an electric power storage section (10) connected to the input-side terminals to a power supply object (11) connected to the output-side terminals. The switching section has a characteristic in which the greater the gate voltage of the switching section on the negative side in the case where a reverse direction conduction current flows through the switching section, the greater the conduction loss generated in the case where the reverse direction conduction current flows. In addition, the power conversion device judges whether or not there is a request for an increase in the amount of heat generated in association with power transmission, and the power conversion device includes a control section (70) that increases the gate voltage of the switching section on the negative side at the time of turning off the switching section through which the reverse direction conduction current flows at the time of power transmission in the case where it is judged that there is the request for the increase, compared to the case where it is judged that there is not the request for the increase.

Description

[0001] Citation of relevant applications

[0002] This application is based on Japanese Patent Application No. 2019-197371, filed on October 30, 2019, the contents of which are incorporated herein by reference. Technical Field

[0003] This disclosure relates to an electric power conversion device. Background Technology

[0004] Previously, control was performed by converting the electrical energy of a storage battery into heat energy to raise the temperature of the component to be heated. Patent Document 1 discloses a power conversion device that uses a heat exchange fluid heated by an electric heater to raise the temperature of a storage battery, which is the component to be heated.

[0005] Existing technical documents

[0006] Patent documents

[0007] Patent Document 1: Japanese Patent No. 3451141 Summary of the Invention

[0008] If an electric heater is added to heat the battery, it may be possible to increase the size of the power conversion device.

[0009] This disclosure is made in view of the above-mentioned technical problems, and its main objective is to provide a power conversion device that can be miniaturized.

[0010] To solve the above-mentioned technical problems, this disclosure provides a power conversion device having an input terminal, an output terminal, and a switching unit. When power is transmitted from a storage unit connected to the input terminal to a power supply object connected to the output terminal, the switching unit is switched on and off. The switching unit has the characteristic that when a reverse current flows through it, its gate voltage increases towards the negative side, and the conduction loss generated when a reverse current flows through it increases. The power conversion device includes a judgment unit and a control unit. The judgment unit determines whether there is a request for an increase in heat generation accompanying the power transmission. Compared to a case where the increase request is not determined, the control unit, when determining that the increase request exists, increases the gate voltage of the switching unit when the reverse current flows through it during the power transmission, which is then switched off, towards the negative side.

[0011] In this disclosure, when power is transmitted from the energy storage unit connected to the input terminal to the power supply object connected to the output terminal, the switch is disconnected. At this time, a reverse conduction current flows through the switch, resulting in conduction losses, and heat is generated due to these conduction losses. By utilizing this heat, a heating device for heating the object to be heated can be eliminated, or the device can be miniaturized even if a heating device is provided. Therefore, according to this disclosure, miniaturization of the power conversion device can be achieved.

[0012] Furthermore, the switching section of this disclosure has the characteristic that, when a reverse conduction current flows, the gate voltage becomes increasingly negative, and the conduction loss generated under the same conditions increases. Considering this characteristic, in this disclosure, when an increase in heat generation associated with power transmission is determined to exist, compared to a situation where no increase is expected, the gate voltage of the switching section when the reverse conduction current flows during power transmission is increased towards the negative side. This allows the conduction loss generated in the switching section to be greater when the aforementioned increase is expected than when it is not. As a result, the heat generated in the power conversion device can be increased. Attached Figure Description

[0013] The above-mentioned objects, other objects, features, and advantages of this disclosure will become clearer with reference to the accompanying drawings and the following detailed description. The accompanying drawings are described below.

[0014] Figure 1 This is a structural diagram of the power conversion device according to the first embodiment.

[0015] Figure 2 This is a functional block diagram of the control unit in normal mode and heating mode.

[0016] Figure 3 This is a graph showing the characteristics of the reverse conduction current flowing through the Gan-HEMT.

[0017] Figure 4 It is a timing diagram showing the transition of the operating states of each switch in normal mode and heating mode.

[0018] Figure 5 This is a flowchart illustrating the processing steps of the control unit.

[0019] Figure 6 It is a timing diagram that shows the current waveforms, etc., in normal mode and heating mode.

[0020] Figure 7 It is a diagram showing the current path in normal mode and heating mode.

[0021] Figure 8 This is a structural diagram of the power conversion device of Modification 2 of the first embodiment.

[0022] Figure 9 is a timing chart showing the transition of the operation state of each switch in the normal mode and the heating mode.

[0023] Figure 10 is a diagram showing the current path in the normal mode and the heating mode.

[0024] Figure 11 is a structural diagram of the power conversion device of the second embodiment.

[0025] Figure 12 is a functional block diagram of the control section in the normal mode and the heating mode.

[0026] Figure 13 is a flowchart showing the processing steps of the control section.

[0027] Figure 14 is a timing chart showing the current waveform and the like in the normal mode and the heating mode.

[0028] Figure 15 is a diagram showing the voltage control method of the gate voltage setting section in the heating mode.

[0029] Figure 16 is a functional block diagram of the gate voltage setting section in the heating mode.

[0030] Figure 17 is a timing chart showing the current waveform and the like in the heating mode of the fourth embodiment.

[0031] Figure 18 is a timing chart showing the transition of the operation state of each switch in the heating mode of the fifth embodiment.

[0032] Figure 19 is a timing chart showing the current path in the heating mode.

[0033] Figure 20 is a diagram showing the characteristics of the amount of current and the amount of voltage drop when the gate voltage at the time of turn-on is reduced.

[0034] Figure 21 is a functional block diagram of the control section in the normal mode of the sixth embodiment.

[0035] Figure 22 is a timing chart showing the transition of the operation state of each switch in the normal mode.

[0036] Figure 23 is a structural diagram of the power conversion device of the seventh embodiment. DETAILED DESCRIPTION

[0037] <First Embodiment>

[0038] Hereinafter, a first embodiment in which the power conversion device of the present disclosure is embodied will be described with reference to the drawings. The power conversion device of the present embodiment is installed in an electric vehicle such as a plug-in hybrid vehicle or an electric vehicle.

[0039] As shown in FIG. 1, the power conversion system includes a storage battery 10 as a power storage portion, a power supply target 11, a first capacitor 12, a second capacitor 13, and a power conversion device 20. Figure 1

[0040] The storage battery 10 supplies electric power to the power supply target 11 via the power conversion device 20. The storage battery 10 is a secondary battery that can be charged and discharged, and is, for example, a lithium ion battery. In addition, the power supply target 11 includes at least one of a low-voltage storage battery and an electric load, the output voltage of which is lower than that of the storage battery 10. The low-voltage storage battery is, for example, a lead storage battery. The electric load is, for example, a headlight or the like.

[0041] The power conversion device 20 includes a first high-potential side terminal CH1, a first low-potential side terminal CL1, a first full-bridge circuit 30, a second high-potential side terminal CH2, a second low-potential side terminal CL2, a second full-bridge circuit 40, and a transformer 50. In the present embodiment, the first high-potential side terminal CH1 and the first low-potential side terminal CL1 correspond to a pair of input side terminals, and the second high-potential side terminal CH2 and the second low-potential side terminal CL2 correspond to a pair of output side terminals. In addition, the first full-bridge circuit 30 corresponds to a conversion circuit.

[0042] The first full-bridge circuit 30 includes first to fourth switches Q1 to Q4. In the present embodiment, the first to fourth switches Q1 to Q4 are N-channel MOSFETs. In addition, the second full-bridge circuit 40 includes fifth to eighth switches Q5 to Q8. In the present embodiment, the fifth to eighth switches Q5 to Q8 are GaN-HEMTs. Here, GaN means gallium nitride, and HEMT means high electron mobility transistor. The GaN-HEMT is a switching device having a characteristic that the voltage drop increases when reverse conduction occurs when a negative voltage is applied to the gate. In the present embodiment, the fifth to eighth switches Q5 to Q8 correspond to a switching portion. Furthermore, in the present embodiment, the drain corresponds to the high-potential side terminal, and the source corresponds to the low-potential side terminal.

[0043] ​In the first full-bridge circuit 30, the first high-potential side terminal CH1 is connected to the drains of the first switch Q1 and the third switch Q3. The source of the first switch Q1 is connected to the drain of the second switch Q2, and the source of the third switch Q3 is connected to the drain of the fourth switch Q4. The sources of the second switch Q2 and the fourth switch Q4 are connected to the first low-potential side terminal CL1. The first high-potential side terminal CH1 is connected to the first low-potential side terminal CL1 via the first capacitor 12. In addition, the first high-potential side terminal CH1 is connected to the positive terminal of the storage battery 10, and the negative terminal of the storage battery 10 is connected to the first low-potential side terminal CL1.

[0044] In the second full-bridge circuit 40, the drains of the fifth switch Q5 and the seventh switch Q7 are connected to the second high-potential side terminal CH2. The source of the fifth switch Q5 is connected to the drain of the sixth switch Q6, and the source of the seventh switch Q7 is connected to the drain of the eighth switch Q8. The sources of the sixth switch Q6 and the eighth switch Q8 are connected to the second low-potential side terminal CL2. The second high-potential side terminal CH2 is connected to the second low-potential side terminal CL2 via the second capacitor 13. In addition, the second high-potential side terminal CH2 is connected to the second low-potential side terminal CL2 via the power supply object 11.

[0045] The transformer 50 has a first coil 50a and a second coil 50b. The first end of the first coil 50a is connected to the source of the first switch Q1 and the drain of the second switch Q2, and the second end of the first coil 50a is connected to the source of the third switch Q3 and the drain of the fourth switch Q4. The first end of the second coil 50b is connected to the source of the fifth switch Q5 and the drain of the sixth switch Q6, and the second end of the second coil 50b is connected to the source of the seventh switch Q7 and the drain of the eighth switch Q8.

[0046] The first coil 50a and the second coil 50b are magnetically coupled to each other. In a case where the potential at the first end of the first coil 50a becomes higher than the potential at the second end, an induced voltage that makes the potential at the first end higher than the potential at the second end is generated at the second coil 50b. On the other hand, in a case where the potential at the second end of the first coil 50a becomes higher than the potential at the first end, an induced voltage that makes the potential at the second end higher than the potential at the first end is generated at the second coil 50b.

[0047] The power conversion system includes a first current sensor 60, a first voltage sensor 61, a second current sensor 62, a second voltage sensor 63, and a temperature sensor 64. The first current sensor 60 detects a first current II flowing through the first high-potential side terminal CH1, and the first voltage sensor 61 detects a first voltage VI that is a terminal-to-terminal voltage of the first high-potential side terminal CH1 and the first low-potential side terminal CL1. Here, the first current II is positive when a discharge current flows in the storage battery 10, and is negative when a charge current flows.

[0048] The second current sensor 62 detects the second current I2 flowing through the second high-potential side terminal CH2, and the second voltage sensor 63 detects the second voltage V2 as the terminal-to-terminal voltage between the second high-potential side terminal CH2 and the second low-potential side terminal CL2. Here, the second current I2 is positive when the current flows in the direction from the second high-potential side terminal CH2 to the drain of the fifth switch Q5 and the seventh switch Q7, and is negative when the current flows in the opposite direction.

[0049] The temperature sensor 64 detects the ambient temperature T. Here, in the present embodiment, the ambient temperature T is the temperature of the element to be warmed up. In the present embodiment, the element to be warmed up is, for example, the power conversion device 20.

[0050] Each of the detected values I1, V1, I2, V2, and T is input to a control section 70 included in the power conversion device 20. The control section 70 outputs a drive signal to the gate of each of the switches Q1 to Q8 on the basis of each of the detected values I1, V1, I2, V2, and T, thereby causing each of the switches Q1 to Q8 to turn on and off.

[0051] Next, the warming-up control implemented in the present embodiment will be described. In the present embodiment, in a case where the ambient temperature T is higher than a set temperature, which is a case where there is no request for an increase in the amount of heat generation, a normal mode is set. On the other hand, in a case where the ambient temperature T is equal to or lower than the set temperature, which is a case where there is a request for an increase in the amount of heat generation, a heat generation mode is set.

[0052] Figure 2 A control block diagram of the control implemented by the control section 70 in the normal mode and the heat generation mode is shown in FIG. 6.

[0053] The control section 70 includes a command current setting section 71. The command current setting section 71 includes a current calculation section 72 and a minimum value selection section 73.

[0054] The current calculation section 72 divides the power command value P2* by the second voltage V2 as the detected voltage of the second voltage sensor 63, thereby calculating a command current I2f. The definition of the sign of the command current I2f is the same as that of the second current I2.

[0055] The minimum value selection section 73 selects the smaller one of the command current I2f calculated by the current calculation section 72 and a current limit value I2L as a final command current Iref2. The current limit value I2L is set so as to protect the power conversion system from an overcurrent.

[0056] The command current Iref2 output from the minimum value selection section 73 is limited by a first limiter 74 to an upper limit value or a lower limit value.

[0057] The control section 70 includes a current controller 75. The current controller 75 includes a current deviation calculation section 76, a feedback control section 77, and a second limiter 78. The current deviation calculation section 76 subtracts the second current I2, which is a detected current of the second current sensor 62, from the command current Iref2 output from the first limiter 74, thereby calculating a current deviation ΔI2.

[0058] The feedback control section 77 calculates the inter-bridge phase φ1 as an operation amount for feedback-controlling the calculated current deviation ΔI2 to 0. In the present embodiment, proportional integral control is used as this feedback control. Note that the feedback control used in the feedback control section 77 is not limited to proportional integral control, and may, for example, be proportional integral derivative control.

[0059] The inter-bridge phase φ1 calculated by the feedback control section 77 is limited by the second limiter 78 to an upper limit value or a lower limit value. In the present embodiment, the inter-bridge phase φ1 is limited to a range from 0° to a prescribed phase, in which range the larger the value, the larger the transmission power transmitted from the storage battery 10 to the power receiving object 11.

[0060] The control section 70 includes a PWM generation section 79, a gate voltage correction section 80, and a gate voltage setting section 81.

[0061] The PWM generation section 79 generates drive signals for the respective switches Q1 to Q8 based on the inter-bridge phase φ1 received from the second limiter 78, and outputs them to the gate voltage correction section 80. In the present embodiment, the PWM generation section 79 generates drive signals that turn on and off the first switch Q1 to the fourth switch Q4 and turn off the fifth switch Q5 to the eighth switch Q8.

[0062] The gate voltage correction section 80 performs correction of the gate voltage Voff at the time of turning off the fifth switch Q5 to the eighth switch Q8 based on a command received from the gate voltage setting section 81. Thereafter, the gate voltage correction section 80 outputs the drive signals that reflect the above-described correction to the gates of the respective switches Q1 to Q8.

[0063] The gate voltage setting section 81 sets the gate voltage Voff at the time of turning off the fifth switch Q5 to the eighth switch Q8 to the first off voltage Vα (for example, 0 V) in the case of the normal mode, and to the second off voltage Vβ in the case of the heat generation mode. Here, the second off voltage Vβ is a value smaller than the first off voltage Vα, and is a negative voltage in the present embodiment. Setting the gate voltage Voff at the time of turning off to the second off voltage Vβ corresponds to increasing the gate voltage Voff at the time of turning off to the negative side.

[0064] Figure 3 is a graph showing the characteristics of the reverse conduction current flowing through the GaN-HEMT. The vertical axis shows the reverse conduction current amount, and the horizontal axis shows the voltage drop amount.Figure 3 The arrows indicate the direction in which the gate voltage Voff increases towards the negative side when it is off. Therefore, compared to the case where Voff = Vα, in the case where Voff = Vβ, the voltage drop during reverse conduction is greater when the gate voltage Voff increases towards the negative side when it is off, and thus the conduction loss also increases.

[0065] Figure 4 This indicates the shift in the operating state of each switch Q1 to Q8 in normal mode and heating mode. Figure 4 (a) represents the transition of the operating states of the first switch Q1 to the fourth switch Q4. Figure 4 (b) indicates the shift in the operating states of the fifth switch Q5 to the eighth switch Q8.

[0066] Figure 4 The solid line in (a) represents the shift in the operating state of the first switch Q1. The operating state of the first switch Q1 after being flipped becomes the operating state of the second switch Q2. Figure 4 The dashed line in (a) represents the shift in the operating state of the third switch Q3. The operating state of the third switch Q3 after being flipped becomes the operating state of the fourth switch Q4. For example, in Figure 4 In (a), the period during which the first switch Q1 is turned on is the period during which the second switch Q2 is turned off.

[0067] Figure 4 The solid line in (b) represents the operating state of switches Q5 through Q8. In this embodiment, switches Q5 through Q8 are open.

[0068] One switching cycle Tsw of the first switch Q1 to the fourth switch Q4 is the same for each other. The phase difference between the moment when the first switch Q1 switches to open and the moment when the third switch Q3 switches to open is the bridge phase φ1. In this embodiment, the bridge phase φ1 is positive when the moment when the third switch Q3 switches to open is earlier than the moment when the first switch Q1 switches to open. For example, in Figure 5 In (a), the phase φ1 between the bridges is positive.

[0069] Figure 6 This indicates the steps of the processing performed by the control unit 70. This processing is performed repeatedly, for example, at a predetermined control cycle. Furthermore, in this embodiment, it is assumed that the product of the first voltage V1 and the number of turns of the first coil 50a is greater than the product of the second voltage V2 and the number of turns of the second coil 50b. That is, through this processing, power is transferred from the battery 10 to the power supply object 11.

[0070] In step S10, it is determined whether the ambient temperature T is below the set temperature. In this embodiment, step S10 is equivalent to the determination unit.

[0071] In the case where the negative determination is made in step S10, as a case where there is no request for an increase in the amount of heat generation, the processing proceeds to step Sll, and the control mode is set to the normal mode. In the normal mode, in the next step S12, the gate voltage Voff at the time of turning off the fifth switch Q5 to the eighth switch Q8 is set to the first off voltage Vα.

[0072] In the case where the positive determination is made in step S10, as a case where there is a request for an increase in the amount of heat generation, the processing proceeds to step S13, and the control mode is set to the heat generation mode. In the heat generation mode, in the next step S14, the gate voltage Voff at the time of turning off the fifth switch Q5 to the eighth switch Q8 is set to the second off voltage Vβ.

[0073] In step S15, the command current Iref2 is set by the command current setting section 71.

[0074] In step S16, the first switch Ql to the fourth switch Q4 are turned on and off, and the fifth switch Q5 to the eighth switch Q8 are turned off, so that the second current I2 is controlled to the command current Iref2.

[0075] Figure 6 The transition of the operation state and the like of each of the switches Ql to Q8 in the normal mode and the heat generation mode is shown. Figure 6 (a) of FIG. 9 shows the transition of the operation state of the first switch Ql to the fourth switch Q4. Figure 6 (b) of FIG. 9 shows the transition of the operation state of the fifth switch Q5 to the eighth switch Q8.

[0076] Figure 6 (c) of FIG. 9 shows the transition of the first current Iland the second current I2. Figure 6 (d) of FIG. 9 shows the first electric power PI, the second electric power P2, and the loss electric power PI+P2. Here, the first electric power PI is the time average of the electric power supplied from the storage battery 10 to the first full-bridge circuit 30, and the second electric power P2 is the time average of the electric power supplied from the second full-bridge circuit 40 to the power supply object 11. If the first electric power PI is positive, the storage battery 10 is discharged. In addition, if the second electric power P2 is negative, the power supply object 11 is supplied with electric power. Therefore, the sum PI+P2 of the first electric power PI and the second electric power P2 becomes the loss electric power. When the loss electric power PI+P2 is positive, it indicates that electric energy is converted into heat energy and heat generation occurs. Figure 6 (e) of FIG. 9 is a graph in which the loss electric power PI+P2 in (d) of FIG. 9 is plotted against the time. Figure 6 (d) of FIG. 9 is a graph in which the loss electric power PI+P2 in (d) of FIG. 9 is enlarged. Figure 6 The scale of the vertical axis in the normal mode and the heat generation mode in (e) of FIG. 9 is the same, and is used to compare the magnitude of the loss electric power PI+P2 in the normal mode and the heat generation mode.

[0077] Hereinafter, the operation of the control device 70 will be described with reference to FIGS. 10 to 14.Figure 7 and Figure 6 The temperature increase control in the normal mode and the heat generation mode of the present embodiment will be described. In the heat generation mode of the present embodiment, the inter-bridge phase φ1 is a value in the range from 0° to a prescribed phase so that the second current I2 is feedback-controlled to the command current Iref2. Since the inter-bridge phase φ1 is limited to the range from 0° to the prescribed phase, it is possible to prevent the electric power that can be transmitted from the storage battery 10 to the power receiving object 11 from becoming small when the inter-bridge phase φ1 is too large.

[0078] As shown in (a) of FIG. 10, in the normal mode and the heat generation mode of the present embodiment, in one switching period Tsw, the first period T1 to the fourth period T4 occur. In the present embodiment, the switching pattern of the normal mode and the heat generation mode is the same. The first period T1 is a period in which the second switch Q2 and the third switch Q3 are on and the first switch Q1, the fourth switch Q4, the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, and the eighth switch Q8 are off. The second period T2 is a period in which the first switch Q1 and the third switch Q3 are on and the second switch Q2, the fourth switch Q4, the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, and the eighth switch Q8 are off. Figure 7 The third period T3 is a period in which the first switch Q1 and the fourth switch Q4 are on and the second switch Q2, the third switch Q3, the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, and the eighth switch Q8 are off. The fourth period T4 is a period in which the second switch Q2 and the fourth switch Q4 are on and the first switch Q1, the third switch Q3, the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, and the eighth switch Q8 are off.

[0079]

[0080] Figure 6 (a) of FIG. 10 indicates the current path in the first period T1. In the first full bridge circuit 30, a current path including the first high-potential side terminal CH1, the third switch Q3, the first coil 50a, the second switch Q2, and the first low-potential side terminal CL1 is formed. On the other hand, in the second full bridge circuit 40, a current path including the second low-potential side terminal CL2, the sixth switch Q6, the second coil 50b, the seventh switch Q7, and the second high-potential side terminal CH2 is formed. In this case, the sixth switch Q6 and the seventh switch Q7 are off, but due to the characteristics of GaN-HEMT, a reverse conduction current flows through the sixth switch Q6 and the seventh switch Q7.

[0081] Therefore, as shown in (b) of FIG. 10, in the first period T1, the first current I1 flows through the first full bridge circuit 30, and the second current I2 flows through the second full bridge circuit 40. In the second period T2, the first current I1 flows through the first full bridge circuit 30, and the second current I2 flows through the second full bridge circuit 40. In the third period T3, the first current I1 flows through the first full bridge circuit 30, and the second current I2 flows through the second full bridge circuit 40. In the fourth period T4, the first current I1 flows through the first full bridge circuit 30, and the second current I2 flows through the second full bridge circuit 40. Figure 7 ​As shown in (c), during the first period T1, the first current I1 gradually increases in the positive direction, and the second current I2 gradually increases in the negative direction. During the first period T1, compared with the normal mode, in the heating mode, the gate voltage Voff when disconnected increases to the negative side. Therefore, due to the characteristics of GaN-HEMT, the conduction losses of the sixth switch Q6 and the seventh switch Q7 caused by the reverse conduction current are greater than in the normal mode.

[0082] Figure 6 (b) represents the current path during the second period T2. In the first full-bridge circuit 30, a current path including the first switch Q1, the third switch Q3, and the first coil 50a is formed. On the other hand, in the second full-bridge circuit 40, the same current path as during the first period T1 is formed.

[0083] During the second period T2, since the first coil 50a is not connected to the first low-potential side terminal CL1, therefore, as Figure 7 As shown in (c), the first current I1 is 0. In the first full-bridge circuit 30, an induced voltage is temporarily generated in the first coil 50a, causing the potential at the first terminal to be higher than that at the second terminal. Consequently, an induced voltage is temporarily generated in the second coil 50b, causing the potential at the second terminal to be higher than that at the first terminal. Therefore, the sign of the second current I2 remains negative, and its absolute value gradually decreases. In this case, due to the characteristics of GaN-HEMT, the conduction loss caused by the reverse conduction current is greater in the heating mode compared to the normal mode.

[0084] Figure 6 (c) indicates the current path during the third period T3. In the first full-bridge circuit 30, a current path is formed including the first high-potential side terminal CH1, the first switch Q1, the first coil 50a, the fourth switch Q4, and the first low-potential side terminal CL1. On the other hand, in the second full-bridge circuit 40, a current path is formed including the second low-potential side terminal CL2, the eighth switch Q8, the second coil 50b, the fifth switch Q5, and the second high-potential side terminal CH2. In this case, the fifth switch Q5 and the eighth switch Q8 are open, but due to the characteristics of GaN-HEMT, reverse current flows through the fifth switch Q5 and the eighth switch Q8.

[0085] Therefore, as Figure 7 As shown in (c), during the third period T3, the first current I1 gradually increases in the positive direction, and the second current I2 gradually increases in the negative direction. During the third period T3, compared with the normal mode, in the heating mode, the gate voltage Voff when disconnected increases to the negative side. Therefore, due to the characteristics of GaN-HEMT, the conduction losses of the fifth switch Q5 and the eighth switch Q8 caused by the reverse conduction current are greater than in the normal mode.

[0086] Figure 6 (d) indicates a current path during the fourth period T4. In the first full bridge circuit 30, a current path including the second switch Q2, the first coil 50a, and the fourth switch Q4 is formed. On the other hand, in the second full bridge circuit 40, the same current path as that during the third period T3 is formed.

[0087] In the fourth period T4, since the first coil 50a is not connected to the first high potential side terminal CH1, as shown in (c) of FIG. 6, the first current I1 is 0. In the first coil 50a in the first full bridge circuit 30, an induced voltage that makes the potential at the second end higher than the potential at the first end is temporarily generated. Thereby, in the second coil 50b, an induced voltage that makes the potential at the first end higher than the potential at the second end is temporarily generated. Therefore, the sign of the second current I2 remains negative, and its absolute value gradually decreases. In this case, since the characteristics of GaN-HEMT, the on loss caused by the reverse conduction current is larger in the heat mode than in the normal mode. Figure 6

[0088] In (c) of FIG. 6, the time average value I1ave of the first current I1 and the time average value I2ave of the second current I2 in the first period T1 to the fourth period T4 are indicated by a broken line. The first current average value I1ave is a positive value, and the second current average value I2ave is a negative value. Figure 6

[0089] As shown in (d) of FIG. 6, the first power P1 is a positive value, and the second power P2 is a negative value. This indicates that in the first period T1 to the fourth period T4, electric power is transmitted from the storage battery 10 to the power supply object 11. Due to this electric power transmission, a loss of electric power occurs in each of the switches Q1 to Q8, and the like, and thus the loss electric power P1+P2 is a positive value. Figure 6

[0090] As shown in (e) of FIG. 6, it is known that the loss electric power P1+P2 in the heat mode is larger than the value of the loss electric power P1+P2 in the normal mode. This is because, in the heat mode, the gate voltage Voff at the time of turning off is increased to the negative side compared to the normal mode, and thus, due to the characteristics of GaN-HEMT, the on loss caused by the reverse conduction current in the fifth switch Q5 to the eighth switch Q8 is larger. Figure 8

[0091] According to the above-described embodiment, the following effects can be obtained.

[0092] ​​​​In the present embodiment, in a case where electric power is transmitted from the storage battery 10 to the power supply object 11, the fifth switch Q5 to the eighth switch Q8 are turned off. At this time, conduction loss occurs in association with the reverse conduction current flowing through the fifth switch Q5 to the eighth switch Q8, and heat is generated due to the conduction loss. By utilizing the heat, it is possible to omit a heating device for warming up the warming-up target element, or to downsize the device even in a case where the heating device is provided. Thus, according to the present disclosure, it is possible to achieve downsizing of the power conversion device 20.

[0093] In addition, the fifth switch Q5 to the eighth switch Q8 of the present embodiment are configured of GaN-HEMT, and thus have a characteristic that the greater the gate voltage Voff at the time of turning off in a case where a reverse conduction current flows, the greater the conduction loss generated. In view of this characteristic, in the present embodiment, the gate voltage Voff at the time of turning off of the fifth switch Q5 to the eighth switch Q8 is increased to the negative side in a case where the heating mode is set, as compared with a case where the normal mode is set. Due to this, it is possible to make the conduction loss generated in the fifth switch Q5 to the eighth switch Q8 greater in the heating mode than in the normal mode. As a result, it is possible to increase the heat generated in the power conversion device 20.

[0094] <Variant 1 of the first embodiment>

[0095] Hereinafter, Variant 1 of the first embodiment will be described, focusing on the points different from the first embodiment.

[0096] As the first switch Q1 to the fourth switch Q4, not only N-channel MOSFET but also IGBT can be used. In this case, the high-potential side terminal is the collector, and the low-potential side terminal is the emitter. In addition, in a case where IGBT is used, the first full-bridge circuit 30 includes freewheeling diodes connected in reverse parallel to the first switch Q1 to the fourth switch Q4, respectively. In this case, the group of the switch and the freewheeling diode corresponds to the switching section.

[0097] <Variant 2 of the first embodiment>

[0098] Hereinafter, Variant 2 of the first embodiment will be described, focusing on the points different from the first embodiment. In the first embodiment, the first full-bridge circuit 30 was used as the conversion circuit, but it can be changed. In the present embodiment, a half-bridge circuit is used as the conversion circuit.

[0099] Figure 8 is a configuration diagram of the present embodiment. In Figure 1 , for convenience, the points different from the previous Figure 9Structures shown in the same structure, the same symbol. The power conversion system includes a power conversion device 90.

[0100] The power conversion device 90 includes a third capacitor 91 and a half bridge circuit 92. In the present embodiment, the half bridge circuit 92 corresponds to the conversion circuit.

[0101] The half bridge circuit 92 includes a ninth switch Q9 and a tenth switch Q10. In the present embodiment, the ninth switch Q9 and the tenth switch Q10 are N-channel MOSFETs. The drain of the ninth switch Q9 is connected to the first high potential side terminal CH1. The source of the ninth switch Q9 is connected to the drain of the tenth switch Q10. The source of the tenth switch Q10 is connected to the first low potential side terminal CL1 and the second end of the first coil 50a. The first end of the first coil 50a is connected to the source of the ninth switch Q9 and the drain of the tenth switch Q10 via the third capacitor 91.

[0102] In the present embodiment, the control section 70 alternately turns on the ninth switch Q9 and the tenth switch Q10 that constitute the half bridge circuit 92. In this case, the duty ratio of the ninth switch Q9 is 0.5. Here, the duty ratio is the ratio of the on time Ton of the ninth switch Q9 with respect to one switching period Tsw (Ton / Tsw).

[0103] Figure 9 The transition of the operating state of the fifth switch Q5 to the tenth switch Q10 and the like in the normal mode and the heat generation mode of the present embodiment is shown. Figure 9 The solid line of (a) of FIG. 9 shows the transition of the operating state of the ninth switch Q9, and the dashed line shows the transition of the tenth switch Q10. Figure 9 (b) of FIG. 9 shows the operating states of the fifth switch Q5 to the eighth switch Q8. Figure 9 (c) of FIG. 9 shows the transition of the first current I1 and the second current I2.

[0104] Hereinafter, the use of Figure 10 and Figure 9 The temperature increase control in the normal mode and the heat generation mode of the present embodiment is described.

[0105] As Figure 10As shown in (a), in the normal mode and heating mode of this embodiment, the fifth period T5 to the eighth period T8 occur within one switching cycle Tsw. The fifth period T5 and the eighth period T8 are the periods when the ninth switch Q9 is turned on and the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, the eighth switch Q8, and the tenth switch Q10 are turned off. The sixth period T6 and the seventh period T7 are the periods when the tenth switch Q10 is turned on and the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, the eighth switch Q8, and the ninth switch Q9 are turned off. Thus, in this embodiment, the fifth switches Q5 to the eighth switches Q8 constituting the second full-bridge circuit 40 are also fixed to be turned off, therefore, conduction losses caused by reverse conduction current are generated.

[0106] Figure 9 (a) indicates the current path during the fifth period T5. In the half-bridge circuit 92, a current path is formed including the first high-potential side terminal CH1, the ninth switch Q9, the third capacitor 91, the first coil 50a, and the first low-potential side terminal CL1. On the other hand, in the second full-bridge circuit 40, the same current path as the third period T3 is formed.

[0107] Therefore, as Figure 10 As shown in (c), during the fifth period T5, the first current I1 gradually increases in the positive direction. On the other hand, the second current I2 gradually increases in the negative direction.

[0108] Figure 9 (b) represents the current path during the sixth period T6. In the half-bridge circuit 92, a current path is formed including the tenth switch Q10, the third capacitor 91, and the first coil 50a. In the second full-bridge circuit 40, the same current path as during the fifth period T5 is formed.

[0109] During the sixth period T6, since the first coil 50a is not connected to the first high-potential side terminal CH1, therefore, as Figure 10 As shown in (c), the first current I1 is 0. In the half-bridge circuit 92, an induced voltage is temporarily generated in the first coil 50a, causing the potential at the second terminal to be higher than that at the first terminal. Consequently, an induced voltage is temporarily generated in the second coil 50b, causing the potential at the first terminal to be higher than that at the second terminal. Therefore, the sign of the second current I2 remains negative, and its absolute value gradually decreases.

[0110] Figure 9 (c) indicates the current path during the seventh period T7. In the half-bridge circuit 92, the current in the opposite direction flows in the same current path as during the sixth period T6. In the second full-bridge circuit 40, a current path is formed including the second low-potential side terminal CL2, the sixth switch Q6, the second coil 50b, the seventh switch Q7, and the second high-potential side terminal CH2.

[0111] In the seventh period T7, since the first coil 50a is not connected to the first high potential side terminal CH1, as shown in (c) of FIG. 7, the first current I1 is 0. Since the charge stored in the third capacitor 91 is discharged in the half-bridge circuit 92, the current flows in the direction opposite to that in the sixth period T6. Therefore, in the second coil 50b, an induced voltage is generated which makes the potential of the second terminal higher than that of the first terminal. Thus, the second current I2 gradually increases in the negative direction. Figure 10

[0112] Figure 9 (d) of FIG. 7 shows the current path in the eighth period T8. In the half-bridge circuit 92, a current path including the first low potential side terminal CL1, the second coil 50b, the third capacitor 91, the ninth switch Q9, and the first high potential side terminal CH1 is formed. In the second full-bridge circuit 40, the same current path as that in the seventh period T7 is formed.

[0113] In the eighth period T8, an induced voltage is temporarily generated in the first coil 50a which makes the potential of the first terminal higher than that of the second terminal. Thus, as shown in (c) of FIG. 7, the first current I1 is negative in sign and its absolute value gradually decreases. On the other hand, in the second coil 50b, an induced voltage is temporarily generated which makes the potential of the second terminal higher than that of the first terminal. Thus, the second current I2 remains negative in sign and its absolute value gradually decreases. Figure 11

[0114] Thus, even in the case where the half-bridge circuit 92 is used as a configuration of the conversion circuit, it is possible to convert the direct current voltage supplied from the storage battery 10 into an alternating current voltage. Therefore, it is possible to flow the reverse conduction current through the fifth switch Q5 to the eighth switch Q8 configuring the second full-bridge circuit 40 using the alternating current voltage.

[0115] Therefore, in the above-described embodiment, the same effects as those of the first embodiment can be obtained.

[0116] In addition, in the present embodiment, as the tenth switch Q10, not only an N-channel MOSFET but also an IGBT can be used.

[0117] <Second Embodiment>

[0118] Hereinafter, with reference to the drawings, the second embodiment will be described focusing on the difference from the first embodiment. In the first embodiment, the full-bridge circuit is used in the structure of the power conversion device 20, but it can be changed. In the present embodiment, the structure of the power conversion device is set to a boost chopper circuit.

[0119] Figure 11 shows a block diagram of the present embodiment. In the present embodiment, the power conversion device 20 is configured as a boost chopper circuit.​​Figure 1 In the following description, for convenience, the same reference signs are assigned to the same structures as those shown in the first embodiment. Figure 12 The power conversion system includes a storage battery 10, a power supply object 11, a capacitor 12, and a power conversion device 100.

[0120] The power conversion device 100 includes a first high-potential side terminal CH1, a first low-potential side terminal CL1, a second high-potential side terminal CH2, a second low-potential side terminal CL2, a half-bridge circuit 101, and a reactor 102.

[0121] The half-bridge circuit 101 includes an eleventh switch Q11 and a twelfth switch Q12. In the present embodiment, the eleventh switch Q11 is a GaN-HEMT, and the twelfth switch Q12 is an IGBT. The drain of the eleventh switch Q11 is connected to the second high-potential side terminal CH2. The source of the eleventh switch Q11 is connected to the collector of the twelfth switch Q12. The emitter of the twelfth switch Q12 is connected to the first low-potential side terminal CL1 and the second low-potential side terminal CL2. The second high-potential side terminal CH2 is connected to the second low-potential side terminal CL2 via the capacitor 12. In addition, the second high-potential side terminal CH2 is connected to the second low-potential side terminal CL2 via the power supply object 11.

[0122] Further, the twelfth switch Q12 is connected in antiparallel with a diode D as a freewheeling diode. As the twelfth switch Q12, not only an IGBT but also an N-channel MOSFET can be used. In this case, the diode D can not be included.

[0123] The first end of the reactor 102 is connected to the first high-potential side terminal CH1. The second end of the reactor 102 is connected to the source of the eleventh switch Q11 and the collector of the twelfth switch Q12. The first high-potential side terminal CH1 is connected to the positive terminal of the storage battery 10, and the negative terminal of the storage battery 10 is connected to the first low-potential side terminal CL1.

[0124] The power conversion system includes a first current sensor 60, a first voltage sensor 61, a second current sensor 62, a second voltage sensor 63, and a temperature sensor 64. The detection objects of the above-described sensors are the same as those of the first embodiment.

[0125] The control section 70 outputs drive signals to the gates of the eleventh switch Q11 and the twelfth switch Q12 on the basis of the respective detection values I1, V1, I2, V2, T, thereby causing the eleventh switch Q11 and the twelfth switch Q12 to turn on and off. Here, in the present embodiment, the definition of the sign of the first current I1 is the same as that of the first embodiment, but the second current I2 is positive when the current flows from the second high-potential side terminal CH2 to the power supply object 11, and is negative when the current flows in the opposite direction.

[0126] Figure 13 is a control block diagram of the control section 70 in the present embodiment. The control section 70 includes a voltage controller 82. The voltage controller 82 includes a voltage deviation calculation section 83, a feedback control section 77, and a duty calculation section 84.

[0127] The voltage deviation calculation section 83 subtracts the second voltage V2, which is the detected voltage of the second voltage sensor 63, from the command voltage V2* to thereby calculate a voltage deviation ΔV2.

[0128] The feedback control section 77 calculates a duty of the twelfth switch Q12 as an operation amount for feedback-controlling the calculated voltage deviation ΔV2 to 0.

[0129] In the duty calculation section 84, the duty calculated by the feedback control section 77 is added to a feedforward term (V2* - V1) / V2* and is used as the final duty of the twelfth switch Q12. The duty of the twelfth switch Q12 is output to the PWM generation section 79.

[0130] The PWM generation section 79 generates drive signals of the eleventh switch Q11 and the twelfth switch Q12 and outputs them to the gate voltage correction section 85. In the present embodiment, the eleventh switch Q11 is turned off. Also, the twelfth switch Q12 is turned on and off based on the duty of the twelfth switch Q12.

[0131] The gate voltage correction section 85 performs correction of the gate voltage Voff at the time of turning off the eleventh switch Q11 based on a command received from the gate voltage setting section 81. Thereafter, the gate voltage correction section 85 outputs the drive signals to the eleventh switch Q11 and the twelfth switch Q12.

[0132] As in the first embodiment, the gate voltage setting section 81 sets the gate voltage Voff at the time of turning off the eleventh switch Q11.

[0133] Figure 13 The steps represent processes performed by the control section 70. The processes are repeatedly performed, for example, at a prescribed control cycle. Also, in the Figure 5 , the same symbols are attached to the same processes as those shown in the Figure 14 .

[0134] In the case where the negative determination is made in step S10, as a case where there is no request for an increase in the amount of heat generation, the processing proceeds to step S17, and the control mode is set to the normal mode. In the normal mode, in the next step S18, the gate voltage Voff at the time of turning off the eleventh switch Q11 is set to the first off voltage Vα.

[0135] If a positive judgment is made in step S10, and an increase in heat generation is requested, the process proceeds to step S19, where the control mode is set to heat generation mode. In heat generation mode, in the next step S20, the gate voltage Voff when the eleventh switch Q11 is turned off is set to the second off voltage Vβ. Here, the second off voltage Vβ is a value that increases further towards the negative side than the first off voltage Vα.

[0136] In step S21, the value of the command voltage V2* is set.

[0137] In step S22, the eleventh switch Q11 is turned off, and the duty cycle of the twelfth switch Q12 is controlled so that the second voltage V2 is controlled to be the command voltage V2*.

[0138] Figure 14 This indicates the progression of the operating states of the eleventh switch Q11 and the twelfth switch Q12 in normal mode and heating mode. Figure 14 The solid line in (a) represents the change in the operating state of the eleventh switch Q11, and the dashed line represents the change in the operating state of the twelfth switch Q12. Figure 14 (b) represents the first voltage V1 and the second voltage V2. Figure 14 (c) represents the shift of the first current I1 and the second current I2. Figure 14 (d) represents the first power P1, the second power P2, and the loss power P1+P2. Here, in this embodiment, the second power P2 is set to positive when it is supplied from the power conversion device 100 to the power supply object 11. Therefore, the value obtained by subtracting the second power P2 from the first power P1 is the loss power P1-P2.

[0139] like Figure 15 As shown in (d), the power loss P1-P2 in the heating mode is greater than that in the normal mode. This is because, in this embodiment, in the heating mode, the gate voltage Voff increases to the negative side compared to the normal mode. Therefore, due to the characteristics of GaN-HEMT, the conduction loss caused by the reverse conduction current in the eleventh switch Q11 increases.

[0140] In the embodiment described above, the same effect as in the first embodiment can also be achieved.

[0141] <Third Implementation Method>

[0142] Hereinafter, with reference to the accompanying drawings, the third embodiment will be described focusing on the differences from the first embodiment. In this embodiment, the gate voltage setting unit 81 sets the gate voltage Voff during the off-state in the heating mode.

[0143] In the heating mode of the present embodiment, the gate voltage setting section 81 controls the gate voltage Voff at the time of turning off of the fifth to eighth switches Q5 to Q8 based on the ambient temperature T.

[0144] Figure 16 A method of voltage control performed by the gate voltage setting section 81 is shown. The upper limit temperature TL is a temperature set in accordance with the heat resistance of the element to be warmed and the like, and is set to, for example, the allowable upper limit temperature of the element to be warmed. When the ambient temperature T reaches a limit start temperature TC set to a temperature lower than the upper limit temperature TL, the gate voltage setting section 81 sets the gate voltage Voff at the time of turning off of the fifth to eighth switches Q5 to Q8 to a value corresponding to the ambient temperature T. In detail, in a case where the ambient temperature T exceeds the limit start temperature TC, the closer the ambient temperature T is to the upper limit temperature TL, the closer the gate voltage Voff at the time of turning off of the fifth to eighth switches Q5 to Q8 is made by the gate voltage setting section 81 to the first off voltage Va. In a case where the ambient temperature T reaches the limit start temperature TC, the gate voltage setting section 81 sets the gate voltage Voff at the time of turning off of the fifth to eighth switches Q5 to Q8 to the first off voltage Va. The gate voltage Voff at the time of turning off of the fifth to eighth switches Q5 to Q8 thus set is output to the gate voltage correction section 80.

[0145] Further, in a case where the ambient temperature T is the upper limit temperature TL or more, the gate voltage Voff at the time of turning off is set to the first off voltage Va. Also, in a case where the ambient temperature T is the limit start temperature TC or less, the gate voltage Voff at the time of turning off is set to the second off voltage Vb.

[0146] Thus, in a case where the ambient temperature T exceeds the limit start temperature TC, the gate voltage Voff of the fifth to eighth switches Q5 to Q8 is gradually lowered based on the ambient temperature T, and thus it is possible to reduce the heat generation due to the reverse conduction current. As a result, it is possible to prevent the ambient temperature T from exceeding the upper limit temperature TL.

[0147] <Third Embodiment Variation>

[0148] Hereinafter, with reference to the drawings, the third embodiment variation will be described focusing on the points different from the third embodiment. In the present embodiment, as a method of setting the gate voltage Voff at the time of turning off in the heating mode by the gate voltage setting section 81, feedback control is used.

[0149] Figure 17 is a control block diagram of the gate voltage setting section 81 in the heating mode of the present embodiment. The gate voltage setting section 81 includes a temperature controller 86. The temperature controller 86 includes a temperature deviation calculation section 87, a feedback control section 88, and a limiter 89.

[0150] The temperature deviation calculation section 87 calculates the temperature deviation ΔT by subtracting the ambient temperature T from the target temperature T*. In the present embodiment, the target temperature T* corresponds to the command temperature.

[0151] The feedback control section 88 calculates the gate voltage Voff at the time of turning off the fifth to eighth switches Q5 to Q8 as an operation amount for feedback-controlling the calculated temperature deviation ΔT to 0. For example, in the case where the ambient temperature T is to be increased, the gate voltage Voff at the time of turning off is increased to the negative side. On the other hand, in the case where the ambient temperature T is to be decreased, the absolute value of the gate voltage Voff at the time of turning off is decreased. In the present embodiment, proportional-integral control is used as the feedback control in the feedback control section 88. The feedback control used in the feedback control section 88 is not limited to proportional-integral control, and may, for example, be proportional-integral-derivative control.

[0152] The gate voltage Voff at the time of turning off the fifth to eighth switches Q5 to Q8 calculated by the feedback control section 88 is limited by the limiter 89 to an upper limit value or a lower limit value, and is output to the gate voltage correction section 80. In the present embodiment, the upper limit value is the first off voltage Vα, and the lower limit value is a voltage determined in accordance with the characteristics of the fifth to eighth switches Q5 to Q8.

[0153] In this way, by feedback-controlling the gate voltage Voff of the fifth to eighth switches Q5 to Q8 on the basis of the ambient temperature T, it is possible to control the ambient temperature T to the target temperature T*.

[0154] <Fourth Embodiment>

[0155] Hereinafter, the fourth embodiment will be described with reference to the drawings, focusing on the points of difference from the first embodiment. In the present embodiment, the control section 70 temporarily switches the gate voltage Voff at the time of turning off the fifth and sixth switches Q5 and Q6 from the second off voltage Vβ to the first off voltage Vα.

[0156] Figure 17 The graphs show the transition of the operation states of the respective switches Q1 to Q8 in the case where the gate voltage Voff at the time of turning off is fixed, and in the case where the gate voltage Voff at the time of turning off is temporarily switched. Figure 17 (a) of FIG. 10 shows the transition of the operation states of the first and fourth switches Q1 and Q4. The operation states of the first and fourth switches Q1 and Q4 after being flipped become the operation states of the second and third switches Q2 and Q3. Figure 17 (b) of FIG. 10 shows the transition of the operation states of the fifth to eighth switches Q5 to Q8. In the present embodiment, the fifth to eighth switches Q5 to Q8 are also turned off. Figure 17(c) represents the shift of the gate voltage Voff when the fifth switch Q5 and the sixth switch Q6 are turned off. Figure 17 (d) represents the shift of the first current I1 and the second current I2.

[0157] like Figure 17 As shown in (c), when Voff is fixed, the control unit 70 fixes the gate voltage Voff when the fifth switch Q5 and the sixth switch Q6 are turned off to the second turn-off voltage Vβ. On the other hand, when Voff switching occurs, the control unit 70 temporarily switches the gate voltage Voff when turned off from the second turn-off voltage Vβ to the first turn-off voltage Vα. For example, in Figure 17 In the ninth period T9 and the tenth period T10 shown in (a), the control unit 70 controls the gate voltage Voff when the sixth switch Q6 is turned off to the first turn-off voltage Vα.

[0158] During the ninth period T9, the operating states of each switch Q1 to Q8 are the same as those during the first period T1.

[0159] The current path during the ninth period T9 is the same as that during the first period T1. In this case, the gate voltage Voff when the sixth switch Q6 is turned off is switched from the second turn-off voltage Vβ to the first turn-off voltage Vα, thereby reducing the voltage drop of the sixth switch Q6. Consequently, since the conduction loss of the sixth switch Q6 is reduced, therefore, as... Figure 18 As shown in (d), compared to the case where Voff is fixed, the second current I2 increases towards the negative side faster when Voff is switched.

[0160] During the tenth period T10, the operating states of each switch Q1 to Q8 are the same as those during the third period T3.

[0161] The current path during the tenth period T10 is the same as that during the third period T3. In this case, the gate voltage Voff when the fifth switch Q5 is turned off is switched from the second turn-off voltage Vβ to the first turn-off voltage Vα, thereby reducing the voltage drop of the fifth switch Q5. In this case, since the conduction loss of the fifth switch Q5 is reduced, therefore, as Figure 18 As shown in (d), compared to the case where Voff is fixed, the second current I2 increases towards the negative side faster when Voff is switched.

[0162] Thus, in this embodiment, during the period when the fifth switch Q5 to the eighth switch Q8 are turned off, the control unit 70 provides a period during which the gate voltage Voff when the switch with reverse conduction current flowing through it is temporarily switched to the first disconnection voltage Vα. Therefore, even if the difference between the first voltage V1 and the second voltage V2 cannot be sufficiently guaranteed, the reverse conduction current increases during this period, so even if the gate voltage Voff when it is turned off returns to the second disconnection voltage Vβ, the reverse conduction current can still flow through the fifth switch Q5 to the eighth switch Q8. As a result, the conduction losses caused by the reverse conduction current can be increased, and therefore, the heat generated in the power conversion device 20 can be increased.

[0163] Furthermore, in this embodiment, the value of the gate voltage Voff when temporarily switched off is set as the first off voltage Vα. However, this value is not limited to the first off voltage Vα, and can also be adjusted in the direction of reducing conduction loss, so as to adjust the magnitude of the reverse conduction current.

[0164] Furthermore, increasing the duration of the gate voltage Voff during temporary switching-off increases the reverse conduction current, but reduces the heat generated by this current. Therefore, increasing the duration of the gate voltage Voff during temporary switching-off causes the heat generated per switching cycle to decrease after reaching its maximum. By pre-designing this characteristic or implementing feedback control corresponding to the heat generation duration, the duration of the gate voltage Voff during temporary switching-off can be set to achieve the desired heat generation.

[0165] <Fifth Implementation>

[0166] Hereinafter, with reference to the accompanying drawings, the fifth embodiment will be described focusing on the differences from the first embodiment. In the fifth embodiment, the switching mode in the heating mode is changed. In the first embodiment, in the heating mode, the fifth switch Q5 to the eighth switch Q8 are disconnected, but in this embodiment, the fifth switch Q5 to the eighth switch Q8 are switched on and off.

[0167] Figure 18 This indicates the progression of the operating states of each switch Q1 to Q8 in the heating mode of this embodiment. Figure 18 The solid line in (a) represents the shift in the operating state of the first switch Q1. The operating state of the first switch Q1 after being flipped becomes the operating state of the second switch Q2. Figure 18 The dashed line in (a) represents the operating state of the fourth switch Q4. The operating state of the fourth switch Q4 after being flipped becomes the operating state of the third switch Q3. Figure 18The solid line in (b) represents the shift in the operating state of the fifth switch Q5. The operating state of the fifth switch Q5 after being flipped becomes the operating state of the sixth switch Q6. Figure 18 The dashed line in (b) represents the shift in the operating state of the eighth switch Q8. The operating state of the eighth switch Q8 after being flipped becomes the operating state of the seventh switch Q7.

[0168] The power transmission between the first full-bridge circuit 30 and the second full-bridge circuit 40 is adjusted by controlling the phase φ21 between the circuits. Here, the phase φ21 between the circuits is the phase difference between the moment when the fourth switch Q4 switches to the ON position and the moment when the eighth switch Q8 switches to the ON position.

[0169] Figure 19 (c) represents the shift in gate voltage between the fifth switch Q5 and the eighth switch Q8. Von1 represents the gate voltage during normal mode turn-on, and Von2 represents the gate voltage during heat mode turn-on. The gate voltage Von2 during heat mode turn-on is lower than Von1 during normal mode turn-on. Additionally, Voff1 represents the gate voltage during normal mode turn-off, and Voff2 represents the gate voltage during heat mode turn-off. The gate voltage Voff2 during heat mode turn-off increases towards the negative side compared to Voff1 during normal mode turn-off. For example, in... Figure 19 In the eleventh period T11 shown in (a), the gate voltage when the fifth switch Q5 and the eighth switch Q8 are turned on is set to Von2. In addition, in the twelfth period T12, the gate voltage when the fifth switch Q5 is turned on is set to Von2, and the gate voltage when the eighth switch Q8 is turned off is set to Voff2.

[0170] During the eleventh period T11, the first switch Q1, the fourth switch Q4, the fifth switch Q5, and the eighth switch Q8 are turned on, while the second switch Q2, the third switch Q3, the sixth switch Q6, and the seventh switch Q7 are turned off.

[0171] like Figure 3 As shown in (a), the current path in the eleventh period T11 is the same as in the third period T3. However, unlike the third period T3, the fifth switch Q5 and the eighth switch Q8 are turned on. Therefore, the control unit 70 can increase the conduction loss during reverse conduction by setting the gate voltage of the fifth switch Q5 and the eighth switch Q8 when they are turned on to a value Von2 that is smaller than the gate voltage Von1 when they are turned on in the normal mode.

[0172] During the twelfth period T12, the second switch Q2, the third switch Q3, the fifth switch Q5, and the seventh switch Q7 are turned on, while the first switch Q1, the fourth switch Q4, the sixth switch Q6, and the eighth switch Q8 are turned off.

[0173] Figure 20 (b) indicates a current path during the twelfth period T12. In the second full bridge circuit 40, a circuit including the second coil 50b, the seventh switch Q7, and the fifth switch Q5 is formed. In this case, the control section 70 is able to increase the on-loss at the time of forward conduction by setting the gate voltage at the time of turn-on of the fifth switch Q5 flowing the forward conduction current to Von2 which is smaller than the gate voltage Vonl at the time of turn-on in the normal mode.

[0174] The on-loss at the time of reverse conduction is able to be increased by reducing the gate voltage at the time of turn-on due to the characteristics of the reverse conduction current flowing the GaN-HEMT as shown in FIG. 8. Figure 20 FIG. 9 indicates the amount of voltage drop and the amount of reverse conduction current when the gate voltage at the time of turn-on of the fifth switch Q5 to the eighth switch Q8 is reduced. Figure 21 FIG. 9 indicates the amount of voltage drop and the amount of reverse conduction current when the gate voltage at the time of turn-on of the fifth switch Q5 to the eighth switch Q8 is reduced. Figure 21 The arrow in FIG. 9 indicates that the gate voltage at the time of turn-on is reduced. When the gate voltage at the time of turn-on is reduced, it is known that the amount of voltage drop increases with respect to the same amount of forward conduction current.

[0175] Thus, in the present embodiment, by reducing the gate voltage at the time of turn-on in the heat generation mode than in the normal mode, the amount of voltage drop of the fifth switch Q5 to the eighth switch Q8 is able to be increased, and the on-loss is able to be increased.

[0176] In addition, by reducing the size of the gate voltage at the time of turn-on, the difference between the gate voltage at the time of turn-on and the gate voltage at the time of turn-off which increases to the negative side is able to be reduced, and thus, the stress applied to the fifth switch Q5 to the eighth switch Q8 is also able to be reduced. As a result, the fifth switch Q5 to the eighth switch Q8 is able to be suppressed from deteriorating.

[0177] < Sixth Embodiment >

[0178] Hereinafter, with reference to the drawings, the sixth embodiment will be described focusing on the difference from the first embodiment. In the first embodiment, the fifth switch Q5 to the eighth switch Q8 is turned off in the normal mode, but it can be changed. In the present embodiment, the fifth switch Q5 to the eighth switch Q8 is turned on and off in the normal mode.

[0179] Figure 2 FIG. 10 indicates a control block diagram of the control section 70 in the normal mode. In FIG. 10, for convenience, the same reference numerals as in FIG. 5 are given to the same parts as in the previous Figure 22 Figure 22 ​The same structure is denoted by the same symbol. In the present embodiment, the feedback control section 77 calculates the intercircuit phase φ21 as an operation amount for feedback-controlling the current deviation ΔI2 to 0. In the present embodiment, the intercircuit phase φ21 is limited to a range from 0° to a prescribed phase, and the greater the value is in this range, the greater the transmission power from the storage battery 10 to the power supply object 11 is.

[0180] The PWM generation section 79 generates the drive signals of the respective switches Q1 to Q8 on the basis of the intercircuit phase φ21.

[0181] The gate voltage correction section 80 receives an instruction to set the gate voltage at the time of disconnection to Voff = Vα from the gate voltage setting section 81 and corrects the gate voltage Voff of the fifth switch Q5 to the eighth switch Q8. Thereafter, the gate voltage correction section 80 outputs the drive signals to the gates of the respective switches Q1 to Q8.

[0182] Figure 22 The solid line of (a) of FIG. 18 shows the transition of the operation state of the first switch Q1. The operation state of the first switch Q1 after the operation state is inverted becomes the operation state of the second switch Q2. Figure 22 The solid line of (b) of FIG. 18 shows the transition of the operation state of the fifth switch Q5. The operation state of the fifth switch Q5 after the operation state is inverted becomes the operation state of the sixth switch Q6. Figure 22 The dashed line of (a) of FIG. 18 shows the transition of the operation state of the third switch Q3. The operation state of the third switch Q3 after the operation state is inverted becomes the operation state of the fourth switch Q4. Figure 23 The dashed line of (b) of FIG. 18 shows the transition of the operation state of the seventh switch Q7. The operation state of the seventh switch Q7 after the operation state is inverted becomes the operation state of the eighth switch Q8. The phase difference between the timing at which the first switch Q1 switches to ON and the timing at which the fifth switch Q5 switches to ON is the intercircuit phase φ21. Figure 23

[0183] In the present embodiment, the same effects as those of the first embodiment are also obtained.

[0184] <Seventh Embodiment>

[0185] Hereinafter, the seventh embodiment will be described with reference to the drawings, focusing on the points of difference from the first embodiment. In the seventh embodiment, a heat transfer section is added to the structure of the first embodiment.

[0186] Figure 1 A block diagram of the present embodiment is shown. In ​ In the present embodiment, for convenience, the same reference numerals are used for the same components as those of the previous ​ ​The same structure is shown by the same reference numeral. The power conversion device 20 includes a heat transfer section 110. In the present embodiment, the heat transfer section 110 is configured to absorb heat generated in each of the switches Q1 to Q8, the transformer 50, and the power supply target 11 as heat exchange target elements. The heat transfer section 110 transfers the absorbed heat to a temperature increase target element to increase the temperature of the temperature increase target element. The temperature increase target element is, for example, the power supply target 11.

[0187] As the heat transfer section 110, for example, a circulation path that circulates cooling water between the heat exchange target elements and the temperature increase target element can be included, and the temperature increase target element can be increased in temperature via the cooling water. In addition, as the heat transfer section 110, for example, a gas (air) can be used as a cooling fluid in addition to the above. In addition, as the heat transfer section 110, for example, a cooling fluid can not be used, and a heat sink or the like that abuts against the heat exchange target elements and the temperature increase target element can be used.

[0188] Thus, heat generated in association with power conversion can be recovered and transferred to the temperature increase target element, and therefore, the temperature of the temperature increase target element can be efficiently increased.

[0189] <Other Embodiments>

[0190] In addition, each of the above embodiments can be implemented by the following modifications.

[0191] • In the third embodiment, the modified example of the third embodiment, the fourth embodiment, and the fifth embodiment, the circuit structure of the power conversion device 20 of the first embodiment is used, but the same effects can be obtained even if the circuit structure of the power conversion device 90 shown in the modified example 2 of the first embodiment is used.

[0192] • The control section and the method thereof according to the present disclosure can be implemented by a special-purpose computer provided by a processor programmed to execute one or more functions embodied by a computer program, or a special-purpose computer provided by a processor composed of one or more special-purpose hardware logic circuits. Alternatively, the control section and the method thereof according to the present disclosure can be implemented by a special-purpose computer composed of a processor programmed to execute one or more functions and a processor composed of one or more hardware logic circuits. In addition, the computer program can be stored in a computer-readable non-transitory tangible storage medium as instructions to be executed by a computer.

[0193] While the present disclosure is described based on the embodiments, it should be understood that the present disclosure is not limited to the above-described embodiments, structures. The present disclosure also includes various modifications, modifications within the equivalent scope. In addition to this, various combinations, modes, further including only a single element, above or below other combinations, modes, also belong to the scope, the range of ideas of the present disclosure.

Claims

1. A power conversion device having an input terminal, an output terminal, and a switching unit, wherein the switching unit is switched on and off when power is transmitted from a power storage unit connected to the input terminal to a power supply object connected to the output terminal. The switching section has the characteristic that when a reverse conduction current flows through it, the gate voltage of the switching section becomes more negative, and the conduction loss generated when a reverse conduction current flows through it becomes greater. The power conversion device includes a judgment unit and a control unit. The determination unit determines whether there is a request for an increase in heat generation accompanying the power transmission. Compared to the case where the increase request is determined not to exist, when the increase request is determined to exist, the control unit increases the gate voltage of the switch unit when it is turned off during the power transmission, which carries the reverse conduction current, towards the negative side.

2. The power conversion device as described in claim 1, characterized in that, If the increased request is determined to exist, the control unit sets the gate voltage when the switch is turned off based on the temperature of the heating target element that is heated by the heat generated along with the power transmission.

3. The power conversion device as described in claim 2, characterized in that, When the temperature of the element to be heated is above the upper limit temperature, the control unit sets the gate voltage of the switch unit when it is turned off to a first disconnect voltage. When the temperature of the element to be heated is below the limit start temperature which is lower than the upper limit temperature, the control unit sets the gate voltage of the switch unit when it is turned off to a second disconnect voltage which is increased to the negative side of the first disconnect voltage.

4. The power conversion device as described in claim 3, characterized in that, When the temperature of the element to be heated is higher than the limiting start temperature but lower than the upper limit temperature, the control unit makes the gate voltage when the switch is turned off closer to the first disconnection voltage as the temperature of the element to be heated gets closer to the upper limit temperature.

5. The power conversion device as described in claim 2, characterized in that, In order to control the temperature feedback of the heated element to the target temperature of the heated element, the control unit sets the gate voltage when the switch is turned off.

6. The power conversion device according to any one of claims 1 to 5, characterized in that, The power conversion device includes: A transformer having a first coil and a second coil magnetically coupled to each other; A conversion circuit, connected to the input terminal and the first coil, converts the DC voltage supplied from the energy storage unit into AC voltage and applies it to the first coil; and A full-bridge circuit, wherein the full-bridge circuit is connected to the output side terminal and the second coil, and includes the switching section. If the increase request is determined to exist, the control unit sets a period during which the gate voltage of the switch unit, which flows through the reverse conduction current, is temporarily increased to the positive side when it is turned off.

7. The power conversion device according to any one of claims 1 to 5, characterized in that, The power conversion device includes: A transformer having a first coil and a second coil magnetically coupled to each other; A conversion circuit, connected to the input terminal and the first coil, converts the DC voltage supplied from the energy storage unit into AC voltage and applies it to the first coil; and A full-bridge circuit, wherein the full-bridge circuit is connected to the output side terminal and the second coil, and includes the switching section. If the control unit determines that the increase request exists, the control unit lowers the gate voltage of the switch unit when it is turned on compared to when the increase request does not exist.

8. The power conversion device according to any one of claims 1 to 7, characterized in that, If it is determined that there is no increase request, the control unit turns on the switch unit that carries the reverse conduction current during the power transmission.

9. The power conversion device according to any one of claims 1 to 8, characterized in that, The power conversion device includes a heat transfer section that absorbs the heat generated during the power transmission and transfers the heat to the element to be heated.

Images