Motor control device, motor control method, hybrid system, boost converter system, electric power steering system

Abstract

The present application effectively suppresses vibration and noise generated in a permanent magnet synchronous motor. A motor control device (1) of the present application is provided with: a triangular wave generation section (17) that generates a triangular wave signal (Tr) as a carrier wave; a carrier wave frequency adjustment section (16) that adjusts a carrier wave frequency (fc) that indicates a frequency of the triangular wave signal (Tr); and a gate signal generation section (18) that pulse width-modulates three-phase voltage commands (Vu*, Vv*, Vw*) corresponding to a torque command (T*) using the triangular wave signal (Tr), and generates a gate signal for controlling the operation of an inverter. The carrier wave frequency adjustment section (16) adjusts the carrier wave frequency (fc) in a manner that changes a voltage phase error (Δθv) that indicates a phase difference between the three-phase voltage commands (Vu*, Vv*, Vw*) and the triangular wave signal (Tr), in accordance with the torque command (T*) and a rotational speed (ωr) of the motor.

Description

TECHNICAL FIELD

[0001] The present application relates to a motor control device, a motor control method, a hybrid system, a step-up converter system, and an electric power steering system. BACKGROUND

[0002] A permanent magnet synchronous motor does not need a mechanical current rectifying mechanism such as a brush and a commutator, is easy to maintain, and is small and light, and has high efficiency and power factor, so it is widely used for driving and power generation of an electric vehicle. A permanent magnet synchronous motor is generally composed of a stator composed of an armature coil and the like and a rotor composed of a permanent magnet and an iron core and the like. A direct current voltage supplied from a direct current power source such as a battery is converted into an alternating current voltage by an inverter, and an alternating current flows to the armature coil of the permanent magnet synchronous motor, thereby generating an armature magnetic flux. The permanent magnet synchronous motor is driven by a magnet torque generated by an attractive force and a repulsive force between the armature magnetic flux and a magnet magnetic flux of the permanent magnet and a reluctance torque generated to minimize a magnetic reluctance of the armature magnetic flux that permeates the rotor.

[0003] In the permanent magnet synchronous motor, electromagnetic forces caused by the armature magnetic flux and the magnet magnetic flux are generated in a rotational direction (circumferential direction) of the motor and a direction (radial direction) perpendicular to a rotation axis of the motor, respectively. The above-mentioned torque is obtained by integrating the electromagnetic force in the circumferential direction, and includes a fluctuation (torque ripple) of the torque caused by a magnetic circuit structure of the motor. On the other hand, the electromagnetic force generated in the radial direction of the motor functions as an exciting force (electromagnetic exciting force) that deforms and vibrates the stator and a housing of the motor.

[0004] At a low rotation speed of the motor, other vibration and noise factors are small, so vibration and noise caused by the torque ripple become noticeable. In particular, in an environmentally friendly vehicle such as an electric vehicle and a hybrid vehicle that uses the permanent magnet synchronous motor, there is a case where the vibration and the noise become noticeable due to a body resonance caused by two inertial systems of a rotor of the motor and a tire at a low rotation speed. On the other hand, in a rotation speed region of the motor except at a low rotation speed, the electromagnetic force in the radial direction (electromagnetic exciting force) reaches a size of about 5 to 10 times as large as the electromagnetic force in the circumferential direction (torque ripple). Therefore, the vibration and the noise caused by the electromagnetic exciting force dominate.

[0005] In addition, the alternating current flowing to the motor includes a fundamental current component such as a sine wave used for driving control of the motor and a frequency conversion according to the rotation speed of the motor and a high-order harmonic current component caused by a switching operation of the inverter. The frequency of the high-order harmonic current is determined by the frequency of the fundamental current and the frequency of a carrier used in PWM modulation. Therefore, at certain motor rotation speeds, there is a case where the electromagnetic exciting force or the torque ripple generated in the motor due to the fundamental current and the electromagnetic exciting force or the torque ripple generated in the motor due to the high-order harmonic current overlap, and large vibration and noise are generated.

[0006] As a related technology of the present application, the technology disclosed in Patent Literature 1 is known. Patent Literature 1 discloses a method of controlling a second phase in which an electromagnetic exciting force is periodically generated in a motor due to a higher harmonic current caused by a switching operation, in a manner that the second phase overlaps with a first phase in which the electromagnetic exciting force is periodically generated in the motor due to a fundamental wave current corresponding to a rotational speed of a permanent magnet motor, at a prescribed rotational speed of the motor.

[0007] Prior Art Documents

[0008] Patent Literature

[0009] Patent Literature 1: International Publication No. 2018 / 139295 SUMMARY

[0010] Problems to be Solved by the Invention

[0011] As described above, at a low rotational speed of the motor, other vibration and noise factors are less, so vibration and noise caused by torque pulsation become conspicuous. On the other hand, in a rotational speed region of the motor other than the low rotational speed, vibration and noise caused by the electromagnetic exciting force are dominant. Thus, in an environmentally friendly vehicle such as an electric vehicle or a hybrid vehicle that uses a permanent magnet synchronous motor, vibration and noise become a problem in a wide range of rotational speeds. However, the method disclosed in Patent Literature 1 cannot effectively improve such a problem in a wide range of rotational speeds.

[0012] The present application has been made in view of the above problems, and aims to effectively suppress vibration and noise generated in a permanent magnet synchronous motor.

[0013] Technical Means for Solving the Problems

[0014] A motor control device according to the present application controls driving of an alternating-current motor that is driven using alternating current by being connected to a power converter that performs power conversion from direct current to the alternating current, the motor control device including: a carrier generation section that generates a carrier; a carrier frequency adjustment section that adjusts a frequency of the carrier; and a gate signal generation section that pulse-width-modulates a voltage command corresponding to a torque command using the carrier to generate a gate signal for controlling an operation of the power converter; the carrier frequency adjustment section adjusts the frequency of the carrier in a manner that changes a phase difference between the voltage command and the carrier, in accordance with the torque command and a rotational speed of the alternating-current motor.

[0015] The motor control method of the present application controls driving of an alternating-current motor that is driven using alternating current generated by a power converter that performs power conversion from direct current, and adjusts a frequency of a carrier wave in a manner that changes a phase difference between a voltage command corresponding to a torque command of the alternating-current motor and the carrier wave, generates the carrier wave at the adjusted frequency, and performs pulse width modulation of the voltage command using the carrier wave to generate a gate signal for controlling an operation of the power converter.

[0016] The hybrid system of the present application includes the motor control device, a power converter that performs power conversion from direct current to alternating current in accordance with the gate signal output from the motor control device, an alternating-current motor that is driven using the alternating current, and an engine system that is connected to the alternating-current motor.

[0017] The step-up converter system of the present application includes the motor control device, a step-up converter that is connected to a direct-current power supply and generates direct current obtained by stepping up the direct-current power supply in accordance with control by the motor control device, and a power converter that performs power conversion from the direct current stepped up by the step-up converter to alternating current in accordance with the gate signal output from the motor control device.

[0018] The electric power steering system of the present application includes the motor control device, a plurality of power converters that perform power conversion from direct current to alternating current in accordance with the gate signal output from the motor control device, and an alternating-current motor that has a plurality of winding systems and is driven by causing the plurality of winding systems to flow the alternating current generated by the plurality of power converters, respectively, and uses the alternating-current motor to control steering of a vehicle.

[0019] Effects of the Invention

[0020] According to the present application, it is possible to effectively suppress vibration and noise generated in a permanent magnet synchronous motor. BRIEF DESCRIPTION OF DRAWINGS

[0021] Figure 1 FIG. 1 is a block diagram showing the functional configuration of the motor control device of the first embodiment of the present application.

[0022] Figure 2 FIG. 2 is a block diagram showing the functional configuration of the motor control device of the first embodiment of the present application.

[0023] Figure 3 FIG. 3 is a block diagram showing the functional configuration of the carrier wave frequency adjustment section of the first embodiment of the present application.

[0024] Figure 4 A graph showing an example of measured data of the spectrum intensity of sound and vibration when the carrier frequency is fixed.

[0025] Figure 5 A block diagram showing the functional configuration of the voltage phase error operation section of the first embodiment of the present application.

[0026] Figure 6 A conceptual diagram of the reference voltage phase operation.

[0027] Figure 7 A graph showing an example of the noise level caused by the electromagnetic exciting force and the torque ripple of each frequency generated in the motor.

[0028] Figure 8 A graph showing an example of the phase difference data of the torque ripple and the electromagnetic exciting force with respect to the reference voltage phase θvb for each rotational speed ωr of the reference voltage phase θvb.

[0029] Figure 9 A graph showing an example of the phase difference data of the torque ripple and the electromagnetic exciting force with respect to the reference voltage phase θvb for each modulation ratio H of the reference voltage phase θvb.

[0030] Figure 10 A graph showing the reduction effect of the electromagnetic exciting force obtained by the present application.

[0031] Figure 11 A graph showing the reduction effect of the torque ripple obtained by the present application.

[0032] Figure 12 A graph showing the configuration of the hybrid system of the second embodiment of the present application.

[0033] Figure 13 A graph showing the configuration of the step-up converter system of the third embodiment of the present application.

[0034] Figure 14 A graph showing the configuration of the electric power steering system of the fourth embodiment of the present application.

[0035] Figure 15 A graph showing the configuration of the drive control system in the electric power steering system of the fourth embodiment of the present application. DETAILED DESCRIPTION

[0036] Hereinafter, a specific embodiment of the present application will be described in detail with reference to the accompanying drawings. In the present embodiment, a use case in a motor drive system used in an electric vehicle or a hybrid vehicle will be described.

[0037] (First Embodiment)

[0038] Figure 1 FIG. 1 is a diagram showing the overall configuration of a motor drive system equipped with a motor control device according to an embodiment of the present application. Figure 1 In the embodiment, the motor drive system 100 is provided with a motor control device 1, a permanent magnet synchronous motor (hereinafter referred to as "motor") 2, an inverter 3, a rotational position detector 41, and a high voltage battery 5.

[0039] The motor control device 1 generates a gate signal for controlling the drive of the motor 2 in accordance with a torque command T* corresponding to a target torque required by the vehicle to the motor 2, and outputs it to the inverter 3. Further, the details of the motor control device 1 will be described later.

[0040] The inverter 3 has an inverter circuit 31, a PWM signal drive circuit 32, and a smoothing capacitor 33. The PWM signal drive circuit 32 generates a PWM signal for controlling each switching element possessed by the inverter circuit 31 in accordance with a gate signal input from the motor control device 1, and outputs it to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper and lower arms of the U-phase, V-phase, and W-phase. By controlling these switching elements respectively in accordance with the PWM signal input from the PWM signal drive circuit 32, the direct current supplied from the high voltage battery 5 is converted into alternating current and output to the motor 2. The smoothing capacitor 33 smooths the direct current supplied from the high voltage battery 5 to the inverter circuit 31.

[0041] The motor 2 is a synchronous motor that is rotationally driven by the alternating current supplied from the inverter 3, and has a stator and a rotor. When the alternating current input from the inverter 3 is applied to the armature coils Lu, Lv, Lw provided in the stator, three-phase alternating currents Iu, Iv, Iw are conducted in the motor 2, and armature magnetic fluxes are generated in each armature coil. Attraction and repulsion are generated between the armature magnetic fluxes of each armature coil and the magnetic fluxes of the permanent magnets arranged in the rotor, and thus a torque is generated in the rotor to rotationally drive the rotor.

[0042] The motor 2 is provided with a rotational position sensor 4 for detecting the rotational position θ of the rotor. The rotational position detector 41 calculates the rotational position θ in accordance with the input signal of the rotational position sensor 4. The calculation result of the rotational position θ obtained by the rotational position detector 41 is input to the motor control device 1, and is utilized in the phase control of the alternating current by generating the gate signal in accordance with the phase of the induced voltage of the motor 2 by the motor control device 1.

[0043] Here, the rotary position sensor 4 is preferably a resolver composed of a core and a winding, but a sensor using a magnetic resistance element or a Hall element such as a GMR sensor is also acceptable. In addition, the rotary position detector 41 can also infer the rotary position θ without using the input signal from the rotary position sensor 4 but using the three-phase alternating current Iu, Iv, Iw flowing to the motor 2 or the three-phase alternating voltage Vu, Vw, Vw applied to the motor 2 from the inverter 3.

[0044] A current detection unit 7 is disposed between the inverter 3 and the motor 2. The current detection unit 7 detects the three-phase alternating current Iu, Iv, Iw (U-phase alternating current Iu, V-phase alternating current Iv, and W-phase alternating current Iw) flowing in the motor 2. The current detection unit 7 is composed of, for example, a Hall current sensor or the like. The detection results of the three-phase alternating current Iu, Iv, Iw obtained by the current detection unit 7 are input to the motor control device 1 and utilized in the generation of the gate signal performed by the motor control device 1. Further, Figure 1 The example in which the current detection unit 7 is composed of three current detectors is shown, but the current detectors can also be provided as two, and the alternating current of the remaining one phase is calculated from the fact that the sum of the three-phase alternating current Iu, Iv, Iw is zero. In addition, the pulse-shaped direct current flowing from the high-voltage battery 5 to the inverter 3 can also be detected by a shunt resistor or the like inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase alternating current Iu, Iv, Iw is obtained from the direct current and the three-phase alternating voltage Vu, Vw, Vw applied to the motor 2 from the inverter 3.

[0045] Next, the details of the motor control device 1 will be described. Figure 2 A block diagram showing the functional configuration of the motor control device 1 of the first embodiment of the present application. Figure 2 In the example, the motor control device 1 has each functional block of a current command generation section 11, a speed calculation section 12, a three-phase / dq conversion current control section 13, a current control section 14, a dq / three-phase voltage command conversion section 15, a carrier frequency adjustment section 16, a triangular wave generation section 17, a gate signal generation section 18, and the like. The motor control device 1 is composed of, for example, a microcomputer, and these functional blocks can be realized by executing a prescribed program in the microcomputer. Alternatively, a part or all of these functional blocks can also be realized using a hardware circuit such as a logic IC or an FPGA.

[0046] The current command generation section 11 calculates the d-axis current command Id* and the q-axis current command Iq* from the input torque T* command and the power supply voltage Hvdc. Here, the d-axis current command Id* and the q-axis current command Iq* corresponding to the torque command T* are obtained, for example, using a current command map and a formula or the like set in advance.

[0047] The speed calculation unit 12 calculates a motor rotational speed ωr that represents the rotational speed (number of revolutions) of the motor 2, based on the time change in the rotational position θ. Further, the motor rotational speed ωr can be a value expressed in either angular velocity (rad / s) or number of revolutions (rpm). Furthermore, these values can be converted into each other and used.

[0048] The three-phase / dq conversion current control unit 13 performs dq conversion based on the rotational position θ calculated by the rotational position detector 41 on the three-phase alternating currents Iu, Iv, Iw detected by the current detection unit 7, and calculates a d-axis current value Id and a q-axis current value Iq.

[0049] The current control unit 14 calculates a d-axis voltage command Vd* and a q-axis voltage command Vq* in such a manner that the d-axis current command Id* and the q-axis current command Iq* output from the current command generation unit 11 and the d-axis current value Id and the q-axis current value Iq output from the three-phase / dq conversion current control unit 13 coincide with each other, based on the difference between these values. Here, the d-axis voltage command Vd* corresponding to the difference between the d-axis current command Id* and the d-axis current value Id and the q-axis voltage command Vq* corresponding to the difference between the q-axis current command Iq* and the q-axis current value Iq are calculated, for example, by a control method such as PI control.

[0050] The dq / three-phase voltage command conversion unit 15 performs three-phase conversion based on the rotational position θ calculated by the rotational position detector 41 on the d-axis voltage command Vd* and the q-axis voltage command Vq* calculated by the current control unit 14, and calculates three-phase voltage commands Vu*, Vv*, Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). Thus, the three-phase voltage commands Vu*, Vv*, Vw* corresponding to the torque command T* are generated.

[0051] The carrier frequency adjustment unit 16 calculates a carrier frequency fc that represents the frequency of the carrier used in the generation of the gate signal, based on the d-axis voltage command Vd* and the q-axis voltage command Vq* generated by the current command generation unit 11, the rotational position θ calculated by the rotational position detector 41, the rotational speed ωr calculated by the speed calculation unit 12, the torque command T*, and the power supply voltage Hvdc. The triangular wave generation unit 17 generates a carrier based on the carrier frequency fc, and thus adjusts the frequency of the carrier in such a manner that the vibration and noise generated in the motor 2 can be suppressed. Further, the details of the method of calculation of the carrier frequency fc by the carrier frequency adjustment unit 16 will be described later.

[0052] The triangular wave generation unit 17 generates a triangular wave signal (carrier signal) Tr based on the carrier frequency fc calculated by the carrier frequency adjustment unit 16.

[0053] The gate signal generation section 18 performs pulse width modulation on the three-phase voltage commands Vu*, Vv*, Vw* output from the dq / three-phase voltage command conversion section 15 using the triangular wave signal Tr output from the triangular wave generation section 17, and generates gate signals for controlling the operation of the inverter 3. Specifically, a pulse-shaped voltage is generated for each of the U-phase, V-phase, and W-phase based on the comparison result of the three-phase voltage commands Vu*, Vv*, Vw* output from the dq / three-phase voltage command conversion section 15 and the triangular wave signal Tr output from the triangular wave generation section 17. Then, the gate signals for the switching elements of each phase of the inverter 3 are generated based on the generated pulse-shaped voltage. At this time, the gate signals Gup, Gvp, Gwp of the upper arms of each phase are respectively logically inverted to generate the gate signals Gun, Gvn, Gwn of the lower arms. The gate signals generated by the gate signal generation section 18 are output from the motor control device 1 to the PWM signal drive circuit 32 of the inverter 3, and are converted to PWM signals by the PWM signal drive circuit 32. As a result, each switching element of the inverter circuit 31 is subjected to on / off control, and thus the output voltage of the inverter 3 is adjusted.

[0054] Next, the operation of the carrier frequency adjustment section 16 in the motor control device 1 will be described. As described above, the carrier frequency adjustment section 16 calculates the carrier frequency fc based on the d-axis voltage command Vd*, the q-axis voltage command Vq*, the rotational position θ, the rotational speed ωr, the torque command T*, and the power supply voltage Hvdc. By successively controlling the frequency of the triangular wave signal Tr generated by the triangular wave generation section 17 based on the carrier frequency fc, the voltage waveform of the three-phase voltage commands Vu*, Vv*, Vw* corresponding to the torque command T* is adjusted in such a manner that the period and phase of the triangular wave signal Tr serving as a carrier become a desired relationship. Further, the so-called desired relationship here means that the electromagnetic exciting force or torque ripple generated in the motor 2 due to the high-order harmonic current caused by the switching operation of the inverter 3 under the PWM signal and the electromagnetic exciting force or torque ripple generated due to the fundamental wave current corresponding to the voltage command become the same period and opposite phase.

[0055] Figure 3 A block diagram of the carrier frequency adjustment section 16 of the first embodiment of the present application. The carrier frequency adjustment section 16 has a synchronous PWM carrier number selection section 161, a voltage phase calculation section 162, a modulation rate calculation section 163, a voltage phase error calculation section 164, a synchronous carrier frequency calculation section 165, and a carrier frequency setting section 166.

[0056] The synchronous PWM carrier number selection section 161 selects a synchronous PWM carrier number Nc in accordance with the rotational speed ωr, the synchronous PWM carrier number Nc indicating the number of carriers corresponding to one period of a voltage waveform in synchronous PWM control. The synchronous PWM carrier number selection section 161 selects the synchronous PWM carrier number Nc in such a manner that the value of Nc ± 3 or Nc x 2 coincides with the number of times (a multiple of 6) of electromagnetic excitation force or torque pulsation generated in the motor 2 due to the fundamental current corresponding to the voltage command, for example. Specifically, for example, if the rotational speed ωr is less than a prescribed threshold value, Nc = 15 is set, and if it is equal to or more than the threshold value, Nc = 9 is set. Thus, the synchronous PWM carrier number Nc corresponding to the number of times of electromagnetic excitation force or torque pulsation generated in the motor 2 due to the fundamental current can be set to an optimum value in accordance with the rotational speed ωr.

[0057] Further, the reason for setting the synchronous PWM carrier number Nc as described above will be described below. The number of pulsation of higher harmonic currents (sideband components) due to pulse width modulation can be expressed as Nc ± 2, Nc ± 4, Nc x 2 ± 1 using the synchronous PWM carrier number Nc. The number of times of electromagnetic excitation force and torque pulsation generated in the motor 2 due to these sideband components is Nc ± 3, Nc x 2. Therefore, to suppress the electromagnetic excitation force or torque pulsation due to the fundamental current in the motor 2, it is preferable to adjust the triangular wave signal Tr serving as a carrier by setting the synchronous PWM carrier number Nc as described above so as to satisfy the desired relationship described above. Thus, the electromagnetic excitation force or torque pulsation due to the carrier used in pulse width modulation cancels the electromagnetic excitation force or torque pulsation due to the fundamental current, so that vibration and noise generated in the motor 2 can be suppressed.

[0058] Figure 4 A graph showing an example of measured data of the frequency spectrum intensity of sound and vibration generated in the motor 2 when the frequency of the triangular wave signal Tr serving as a carrier is fixed and the rotational speed of the motor 2 is raised from 0 rpm to an arbitrary rotational speed. Figure 4 In FIG. 6, an example of measured data of the frequency spectrum intensity of sound and vibration generated in the motor 2 when the frequency of the triangular wave signal Tr serving as a carrier is fixed and the rotational speed of the motor 2 is raised from 0 rpm to an arbitrary rotational speed is shown. Figure 4 In FIG. 6, the horizontal axis indicates elapsed time from the start of rotation of the motor 2, and the vertical axis indicates frequency. Further, Figure 4 In FIG. 6, the intensity of sound and vibration is indicated by the darkness of the line.

[0059] Observation Figure 4 It is found that the electromagnetic excitation force and torque pulsation generated in the motor 2 due to the carrier used in pulse width modulation is strong in the frequency spectrum at fc ± 3fl. Further, Figure 4 In FIG. 6, the frequency of the carrier fc is fixed. Further, fl indicates the frequency of the fundamental current corresponding to the voltage command, which is proportional to the rotational speed (rotational velocity ωr) of the motor 2.

[0060] The synchronous PWM carrier number selection section 161 selects the synchronous PWM carrier number Nc in accordance with the rotational speed ωr on the basis of the above fact, with a value as described above. For example, if the synchronous PWM carrier number is set to Nc = 9, the number of times of the electromagnetic exciting force and the torque ripple generated in the motor 2 due to the high-order harmonic currents caused by the carriers is 9 - 3 = 6, 9 + 3 = 12, 9 x 2 = 18, in accordance with the above-described formula. These calculated numbers are all consistent with the number of times of the electromagnetic exciting force and the torque ripple generated due to the fundamental current, i.e., a multiple of 6. Also in the case where the synchronous PWM carrier number is set to Nc = 15, the same is true, and the number of times of the electromagnetic exciting force and the torque ripple generated due to the fundamental current, i.e., a multiple of 6, is consistent. Therefore, the electromagnetic exciting force or the torque ripple caused by the carriers used in the pulse width modulation can cancel the electromagnetic exciting force or the torque ripple caused by the fundamental current.

[0061] Further, the synchronous PWM carrier number selection section 161 can select the synchronous PWM carrier number Nc not only in accordance with the rotational speed ωr but also in accordance with the torque command T*. Also, for example, a hysteresis or the like can be provided to change the selection reference of the synchronous PWM carrier number Nc when the rotational speed ωr is rising and when it is falling.

[0062] The voltage phase operation section 162 operates the voltage phase θv in accordance with the d-axis voltage command Vd* and the q-axis voltage command Vq*, the rotational position θ, the rotational speed ωr, and the carrier frequency fc, by the following formulas (1) to (4).

[0063] θv = θ + φv + φdqv + 0.5π... (1)

[0064] φv = ωr - 1.5Tc... (2)

[0065] Tc = 1 / fc... (3)

[0066] φdqv = atan(Vq / Vd)... (4)

[0067] Here, φv indicates an operation lag compensation value of the voltage phase, Tc indicates a carrier period, and φdqv indicates a voltage phase from the d-axis. The operation lag compensation value φv is a value for compensating for an operation lag of 1.5 control periods that occurs during a period from when the rotational position θ is acquired from the rotational position detector 41 to when the motor control device 1 outputs the gate signals to the inverter 3. Further, in the present embodiment, 0.5π is added to the 4th term on the right side of formula (1). The purpose of this operation is to transform the point of view of the voltage phase operated by the 1st to 3rd terms on the right side of formula (1) to a sin wave, since the voltage phase is a cos wave.

[0068] The modulation factor operation section 163 operates the modulation factor H in accordance with the d-axis voltage command Vd* and the q-axis voltage command Vq*, the power supply voltage Hvdc, in accordance with the following equation (5). Further, the modulation factor H indicates the voltage amplitude ratio of the direct current supplied from the high voltage battery 5 to the inverter 3 and the alternating current output from the inverter 3 to the motor 2.

[0069] H = Vd2+ Vq2 / (Hvdc / 2)... (5)

[0070] The voltage phase error operation section 164 operates the voltage phase error Δθv in accordance with the synchronous PWM carrier number Nc selected by the synchronous PWM carrier number selection section 161, the voltage phase θv operated by the voltage phase operation section 162, the modulation factor H operated by the modulation factor operation section 163, the rotational speed ωr, and the torque command T*. The voltage phase error Δθv indicates the phase difference between the voltage command of the inverter 3, i.e., the three-phase voltage commands Vu*, Vv*, Vw*, and the carrier used in the pulse width modulation, i.e., the triangular wave signal Tr. The voltage phase error operation section 164 operates the voltage phase error Δθv at every predetermined operation period, whereby the frequency adjustment of the triangular wave signal Tr can be performed in the carrier frequency adjustment section 16 in such a manner that the phase difference between the voltage command of the inverter 3 and the carrier used in the pulse width modulation is changed.

[0071] The synchronous carrier frequency operation section 165 operates the synchronous carrier frequency fcs in accordance with the voltage phase error Δθv operated by the voltage phase error operation section 164, the rotational speed ωr, and the synchronous PWM carrier number Nc selected by the synchronous PWM carrier number selection section 161, in accordance with the following equation (6).

[0072] fcs = ωr- Nc- (1 + Δθv- K) / (2π)... (6)

[0073] The synchronous carrier frequency operation section 165 can operate the synchronous carrier frequency fcs based on equation (6), for example, by PLL (Phase Locked Loop) control. Further, in equation (6), the gain K can be set to a fixed value or can be changed depending on the conditions.

[0074] The carrier frequency setting section 166 selects either the synchronous carrier frequency fcs operated by the synchronous carrier frequency operation section 165 or the nonsynchronous carrier frequency fcns in accordance with the rotational speed ωr, and outputs the same as the carrier frequency fc. The nonsynchronous carrier frequency fcns is a fixed value set in advance in the carrier frequency setting section 166. Further, a plurality of nonsynchronous carrier frequencies fcns can be prepared in advance, and any one of them can be selected in accordance with the rotational speed ωr. For example, the nonsynchronous carrier frequency fcns can be selected in the carrier frequency setting section 166 in such a manner that the greater the value of the rotational speed ωr, the greater the value of the nonsynchronous carrier frequency fcns, and the same is output as the carrier frequency fc.

[0075] Next, details of the operation method of the voltage phase error Δθv in the voltage phase error operation section 164 in the carrier frequency adjustment section 16 will be described.

[0076] Figure 5 A block diagram of the voltage phase error operation section 164 of the first embodiment of the present application. The voltage phase error operation section 164 has a reference voltage phase operation section 1641, a pulsation frequency conversion section 1642, a pulsation contribution degree selection section 1643, an electromagnetic excitation force reduction phase difference table 1644a, a torque pulsation reduction phase difference table 1644b, a voltage phase difference conversion section 1645, an addition section 1646, and a subtraction section 1647.

[0077] The reference voltage phase operation section 1641 operates a reference voltage phase θvb for fixing the phase of the carrier in the synchronous PWM control, based on the synchronous PWM carrier number Nc and the voltage phase θv. By the operation of the reference voltage phase θvb by the reference voltage phase operation section 1641, the period of the carrier with respect to the voltage phase θv and the period of the electromagnetic excitation force or the torque pulsation in the motor 2 due to the fundamental current are made to coincide with each other.

[0078] Figure 6 A conceptual diagram of the reference voltage phase operation performed by the reference voltage phase operation section 1641. The reference voltage phase operation section 1641, for example, operates the reference voltage phase θvb that changes in a stepped manner between 0 and 2π at a number of steps corresponding to the synchronous PWM carrier number Nc as shown in Figure 6 Again, in order to make the explanation easy to understand, an example in which the synchronous PWM carrier number Nc is 3 is shown in Figure 6 However, in reality, the synchronous PWM carrier number Nc is preferably set to Nc = 9 or Nc = 15 as described above.

[0079] In the present embodiment, in order to reduce the processing load, for example, as shown in Figure 6 The carrier frequency adjustment section 16 is able to adjust the frequency of the carrier only in the interval in which the triangular carrier rises from the minimum value (trough) to the maximum value (peak), that is, the trough division interval. In this case, the synchronous carrier frequency operation section 165 operates the synchronous carrier frequency fcs using the voltage phase error Δθv in the trough division interval of the carrier, as described later, thereby performing the synchronous PWM control. The reference voltage phase operation section 1641 calculates the reference voltage phase θvb used in the operation of this voltage phase error Δθv in the form of discrete values that change at intervals of π / 3 as shown in Figure 6 Again, the interval of this reference voltage phase θvb changes according to the synchronous PWM carrier number Nc. The larger the synchronous PWM carrier number Nc, the smaller the interval of the reference voltage phase θvb.

[0080] Specifically, the reference voltage phase calculation section 1641 calculates the reference voltage phase θvb from the voltage phase θv and the number of synchronous PWM carriers Nc in accordance with the following equations (7) to (8).

[0081] θvb = int(θv / θs) · θs + 0.5θs... (7)

[0082] θs = 2π / Nc... (8)

[0083] Here, θs represents the width of change in the voltage phase θv per 1 carrier, and int represents a rounding-off operation below the decimal point.

[0084] Further, in the present embodiment, the reference voltage phase θvb is calculated in the reference voltage phase calculation section 1641 in accordance with equations (7) to (8) with the reference voltage phase θvb being 0 rad in the peak-dividing interval in which the triangular carrier decreases from the maximum value (peak) to the minimum value (trough). However, the period in which the reference voltage phase θvb is 0 rad is not limited to the peak-dividing interval. As long as the voltage phase θv can be used to calculate the reference voltage phase θvb that changes in a stepped manner between 0 and 2π in a number corresponding to the number of synchronous PWM carriers Nc, the reference voltage phase calculation section 1641 can also perform the calculation of the reference voltage phase θvb by an operation method other than equations (7) to (8).

[0085] The pulsation frequency conversion section 1642 converts the rotational speed ωr into the pulsation frequency fr in accordance with the following equation (9).

[0086] fr = ωr · 4 · Nr / (2π)... (9)

[0087] Here, Nr represents the number of times of the electromagnetic excitation force or the torque pulsation due to the fundamental current, and is a multiple of 6 (6, 12, 18, 24,...) as described above. In the pulsation frequency conversion section 1642, the value of Nr can be set in accordance with the number of times of the electromagnetic excitation force or the torque pulsation to be suppressed as an object.

[0088] The pulsation contribution degree selection section 1643 selects the one of the electromagnetic excitation force generated in the radial direction of the motor 2 and the torque pulsation generated in the circumferential direction of the motor 2 that has a large contribution degree to the vibration and noise generated in the motor 2, in accordance with the pulsation frequency fr calculated by the pulsation frequency conversion section 1642.

[0089] Figure 7 A graph showing an example of the noise level due to the electromagnetic excitation force and the torque pulsation of each frequency generated in the motor 2. Figure 7 Here, the horizontal axis represents the frequency, and the vertical axis represents the magnitude of the noise level. According to the graph, the noise level due to the electromagnetic excitation force is higher than the noise level due to the torque pulsation in the frequency range of 0 to 100 Hz, and the noise level due to the torque pulsation is higher than the noise level due to the electromagnetic excitation force in the frequency range of 100 to 200 Hz. Figure 7It is known that the noise level caused by the torque pulsation in the circumferential direction in the intervals a and c is high, and the noise level caused by the electromagnetic exciting force in the radial direction in the interval b is high. The frequency characteristics of such noise levels are determined by the structure of the motor 2, and thus are inherent to each specification of the motor 2. Therefore, the relationship between the noise levels at each frequency shown in FIG. 8 is obtained in advance by simulation or actual measurement, and the relationship is stored in the memory 1630. Figure 7 The relationship between the noise levels at each frequency shown in FIG. 8 can be selected by the pulsation contribution degree selection section 1643 as the one of the electromagnetic exciting force and the torque pulsation that has a large contribution degree to the vibration and noise generated in the motor 2, according to the pulsation frequency fr corresponding to the rotational speed ωr.

[0090] The electromagnetic exciting force reduction phase difference table 1644a is a table indicating the phase difference for reducing the electromagnetic exciting force of the motor 2, and the torque pulsation reduction phase difference table 1644b is a table indicating the phase difference for reducing the torque pulsation of the motor 2. The so-called phase difference here means the phase difference with respect to the reference voltage phase θvb. These tables are set for a plurality of values of the rotational speed ωr, the torque command T*, and the modulation ratio H, respectively. In the voltage phase error operation section 164, these tables are referred to according to the rotational speed ωr, the torque command T*, and the modulation ratio H, respectively, whereby the phase difference suitable for the reduction of the electromagnetic exciting force and the phase difference suitable for the reduction of the torque pulsation can be determined, respectively.

[0091] The phase difference data with respect to the reference voltage phase θvb at which the electromagnetic exciting force or the torque pulsation is reduced is obtained in advance for each of the rotational speed ωr, the torque command T*, and the modulation ratio H, for example, by simulation or actual measurement. The electromagnetic exciting force reduction phase difference table 1644a and the torque pulsation reduction phase difference table 1644b are set according to these phase difference data obtained in advance, respectively. Here, the reason why the electromagnetic exciting force reduction phase difference table 1644a and the torque pulsation reduction phase difference table 1644b are set for each modulation ratio H is to compensate for the fact that the dominant order of the electromagnetic exciting force or the torque pulsation generated by the high-order harmonic current changes according to the modulation ratio H. Further, the phase difference output according to these tables can be either of the current phase difference and the voltage phase difference. In the present embodiment, the phase difference output from the electromagnetic exciting force reduction phase difference table 1644a and the torque pulsation reduction phase difference table 1644b is the current phase difference, and the conversion from the current phase difference to the voltage phase difference is performed in the voltage phase difference conversion section 1645 at the subsequent stage.

[0092] Figure 8 A graph showing an example of the phase difference data of the torque pulsation and the electromagnetic exciting force with respect to the reference voltage phase θvb for each rotational speed ωr. Figure 9This is a graph showing the phase difference data of the reference voltage phase θvb, torque ripple, and electromagnetic excitation force relative to the reference voltage phase θvb for each modulation rate H. Furthermore, the torque command T* also changes according to the rotational speed ωr, therefore... Figure 8 The data also includes the reference voltage phase θvb for each torque command T*, as well as the phase difference data of the torque ripple and electromagnetic excitation force relative to the reference voltage phase θvb.

[0093] Figure 8 , Figure 9 In this paper, for a phase difference of -180° to +180° relative to the reference voltage phase θvb, the magnitudes of the torque pulsation and electromagnetic excitation force generated in the motor 2 due to the fundamental current corresponding to the voltage command are displayed in the form of multiples of 6, namely 6, 12, 18, and 24 times, as well as the sum of the torque pulsation and the sum of the electromagnetic excitation force. In the voltage phase error calculation unit 164, based on these phase difference data, the phase difference that minimizes the sum of the torque pulsation and the phase difference that minimizes the sum of the electromagnetic excitation force are respectively tabulated, thereby setting the torque pulsation reduction phase difference table 1644b and the electromagnetic excitation force reduction phase difference table 1644a. Alternatively, the phase difference that can effectively reduce the torque pulsation or electromagnetic excitation force by a specific number, such as 12 times, can also be tabulated to set the torque pulsation reduction phase difference table 1644b and the electromagnetic excitation force reduction phase difference table 1644a to avoid resonance frequencies, etc.

[0094] When the electromagnetic excitation force is selected as the factor contributing more to the vibration and noise generated in the motor 2 by the pulsation contribution selection unit 1643, the current phase difference determined in the electromagnetic excitation force reduction phase difference table 1644a based on the rotational speed ωr, torque command T*, and modulation rate H is input to the voltage phase difference conversion unit 1645. On the other hand, when the torque pulsation is selected as the factor contributing more to the vibration and noise generated in the motor 2 by the pulsation contribution selection unit 1643, the current phase difference determined in the torque pulsation reduction phase difference table 1644b based on the rotational speed ωr, torque command T*, and modulation rate H is input to the voltage phase difference conversion unit 1645.

[0095] The voltage phase difference conversion section 1645 adds 0.5π to the current phase difference input from the electromagnetic vibration force reduction phase difference table 1644a or the torque ripple reduction phase difference table 1644b, thereby converting the current phase difference into a voltage phase difference. The reason for adding 0.5π here is that the high-order harmonic current is less affected by the resistance than the fundamental current, so the differential value (leading by 0.5π) of the high-order harmonic current flowing in the inductance component of the motor 2 mainly affects the voltage of the motor 2. Further, in the case where the phase difference output from the electromagnetic vibration force reduction phase difference table 1644a and the torque ripple reduction phase difference table 1644b is set as the voltage phase difference as described above, the voltage phase difference conversion section 1645 need not be provided.

[0096] The addition section 1646 adds the voltage phase difference calculated in the voltage phase difference conversion section 1645 to the reference voltage phase θvb calculated in the reference voltage phase calculation section 1641, thereby calculating a modified reference voltage phase θvb2 for reducing the electromagnetic vibration force or the torque ripple due to the high-order harmonic current.

[0097] The subtraction section 1647 subtracts the modified reference voltage phase θvb2 from the voltage phase θv, thereby calculating a voltage phase error Δθv.

[0098] In the voltage phase error calculation section 164, the voltage phase error Δθv is calculated as described above. Thus, the voltage phase error Δθv can be determined in such a manner that the torque ripple or the electromagnetic vibration force due to the fundamental current corresponding to the three-phase voltage command Vu*, Vv*, Vw* and the torque ripple or the electromagnetic vibration force of each of the high-order harmonic components of the fundamental current having a multiple of 6 as the order among the high-order harmonic components of the fundamental current are canceled by the torque ripple or the electromagnetic vibration force due to the carrier used in the pulse width modulation, in accordance with the rotation speed ωr, the torque command T*, and the modulation rate H. As a result, the phase difference between the voltage command to the inverter 3 and the carrier used in the pulse width modulation can be set to change the carrier frequency fc in such a manner that the torque ripple or the electromagnetic vibration force generated in the motor 2 is reduced.

[0099] Figure 10 A graph showing the reduction effect of the electromagnetic vibration force obtained by the present application. Figure 10In the drawing, an example of the electromagnetic vibration force generated by the fundamental current in the case where the present application is not used (prior art) is indicated by a broken line, and an example of the electromagnetic vibration force generated by the fundamental current in the case where the present application is used (the present application) is indicated by a solid line. Further, the case where the present application is not used corresponds to a case where the voltage phase error Δθv is calculated using the difference between the voltage phase θv and the reference voltage phase θvb and the synchronous PWM control is performed using the voltage phase error Δθv. On the other hand, the case where the present application is used is a case where the phase difference obtained using the electromagnetic vibration force reduction phase difference table 1644a, that is, the phase difference for reducing the electromagnetic vibration force of the motor 2, is added to the reference voltage phase θvb to obtain the corrected reference voltage phase θvb2, the voltage phase error Δθv is calculated using the corrected reference voltage phase θvb2, and the synchronous PWM control is performed using the voltage phase error Δθv. According to the present application, the electromagnetic vibration force can be reduced in the case where the present application is used as compared with the case where the present application is not used, and the present application is effective. Figure 10 It can be confirmed that the electromagnetic vibration force can be reduced in the case where the present application is used as compared with the case where the present application is not used, and the present application is effective.

[0100] Figure 11 A drawing showing the reduction effect of the torque ripple obtained by the present application. Figure 11 In the drawing, an example of the torque ripple generated by the fundamental current in the case where the present application is not used (prior art) is indicated by a broken line, and an example of the torque ripple generated by the fundamental current in the case where the present application is used (the present application) is indicated by a solid line. Further, the case where the present application is not used corresponds to a case where the voltage phase error Δθv is calculated using the difference between the voltage phase θv and the reference voltage phase θvb and the synchronous PWM control is performed using the voltage phase error Δθv. On the other hand, the case where the present application is used is a case where the phase difference obtained using the torque ripple reduction phase difference table 1644b, that is, the phase difference for reducing the torque ripple of the motor 2, is added to the reference voltage phase θvb to obtain the corrected reference voltage phase θvb2, the voltage phase error Δθv is calculated using the corrected reference voltage phase θvb2, and the synchronous PWM control is performed using the voltage phase error Δθv. According to the present application, the torque ripple can be reduced in the case where the present application is used as compared with the case where the present application is not used, and the present application is effective. Figure 11 It can be confirmed that the torque ripple can be reduced in the case where the present application is used as compared with the case where the present application is not used, and the present application is effective.

[0101] Further, in the carrier frequency adjusting section 16, the above-described processing can be performed at any time during the traction drive of the motor 2 or the regenerative drive. During the traction drive, the torque command T* is a positive value, and during the regenerative drive, the torque command T* is a negative value. Thus, in the carrier frequency adjusting section 16, it is determined which one of the traction drive and the regenerative drive the motor 2 is based on the value of the torque command T*, and the above-described calculation processing is performed in the voltage phase error calculating section 164 based on the result of the determination, whereby the voltage phase error Δθv can be changed to set the carrier frequency fc in such a manner that the torque ripple or the electromagnetic vibration force generated in the motor 2 is reduced.

[0102] According to the embodiment of the application described above, the following effects are obtained.

[0103] (1) A motor control device 1 that controls driving of a motor 2 that is driven using alternating current that is generated by an inverter 3 that performs power conversion from direct current, the motor control device 1 including: a triangular wave generation section 17 that generates a triangular wave signal Tr as a carrier; a carrier frequency adjustment section 16 that adjusts a carrier frequency fc that indicates a frequency of the triangular wave signal Tr; and a gate signal generation section 18 that pulse width modulates three-phase voltage commands Vu*, Vv*, Vw* corresponding to a torque command T* using the triangular wave signal Tr, and generates a gate signal for controlling operation of the inverter 3. The carrier frequency adjustment section 16 adjusts the carrier frequency fc in a manner that changes a voltage phase error Δθv that indicates a phase difference between the three-phase voltage commands Vu*, Vv*, Vw* and the triangular wave signal Tr, in accordance with the torque command T* and a rotational speed ωr of the motor 2. Therefore, it is possible to effectively suppress vibration and noise generated in the motor 2.

[0104] (2) The carrier frequency adjustment section 16 selects a synchronous PWM carrier number Nc as a prescribed integer value by a synchronous PWM carrier number selection section 161, and thereby adjusts the carrier frequency fc in a manner that the carrier frequency fc becomes an integer multiple of a frequency of the three-phase voltage commands Vu*, Vv*, Vw*. Therefore, it is possible to adjust voltage waveforms of the three-phase voltage commands Vu*, Vv*, Vw* in a manner that a period and a phase of the triangular wave signal Tr as a carrier each become a desired relationship, and thereby it is possible to reliably perform synchronous PWM control.

[0105] (3) The carrier frequency adjustment section 16 selects one of torque ripple generated in a circumferential direction of the motor 2 and electromagnetic excitation force generated in a radial direction of the motor 2 in accordance with the rotational speed ωr of the motor 2 by a ripple contribution degree selection section 1643. Also, the voltage phase error Δθv is changed in a manner that reduces the selected torque ripple or electromagnetic excitation force by a voltage phase error operation section 164. Therefore, it is possible to effectively suppress vibration and noise at an arbitrary rotational speed for the motor 2 that has a frequency characteristic of noise level due to torque ripple and noise level due to electromagnetic excitation force as shown in FIG. 8. Figure 7

[0106] ​(4) The carrier frequency adjusting section 16 changes the voltage phase error Δθv in accordance with the torque command T*, the rotational frequency ωr, and a modulation ratio H, which is calculated by the modulation ratio calculating section 163 and indicates the ratio of the voltage amplitudes of the direct current supplied to the inverter 3 and the alternating current output from the inverter 3. Thus, even in the case where the dominant order of the electromagnetic exciting force or the torque ripple due to the high-order harmonic current changes in accordance with the modulation ratio H, thereby changing the vibration and noise of the motor 2 in accordance with the modulation ratio H, the change can be reliably compensated for, and the vibration and noise generated in the motor 2 can be effectively suppressed.

[0107] (5) The carrier frequency adjusting section 16 refers to the electromagnetic exciting force reduction phase difference table 1644a and the torque ripple reduction phase difference table 1644b, and thereby changes the voltage phase error Δθv in accordance with each of the high-order harmonic components of the fundamental current corresponding to the three-phase voltage commands Vu*, Vv*, and Vw* that have an order of a multiple of 6. Thus, the electromagnetic exciting force or the torque ripple due to the carrier used in the pulse width modulation can be used to cancel the electromagnetic exciting force or the torque ripple due to the fundamental current, and the vibration and noise generated in the motor 2 can be effectively suppressed.

[0108] (6) The carrier frequency adjusting section 16 can determine which of the traction drive and the regenerative drive the motor 2 is in in accordance with the torque command T*, and change the voltage phase error Δθv in accordance with the result of the determination. In this way, optimal control can be achieved in accordance with the drive state of the motor 2.

[0109] (2nd Embodiment) (Series Hybrid System)

[0110] Next, the 2nd embodiment of the present application will be described. In this embodiment, a use case in a hybrid system in which a motor and an engine are combined will be described.

[0111] Figure 12 A diagram showing the configuration of the hybrid system of the 2nd embodiment of the present application. The hybrid system 72 has the motor control device 1, the motor 2, the inverter 3, the rotational position detector 41, and the high-voltage battery 5 described in the 1st embodiment, and has a motor 2a, an inverter 3a, and a rotational position detector 41a corresponding to the motor 2, the inverter 3, and the rotational position detector 41, respectively.

[0112] The rotational position sensor 4a for detecting the rotational position θa of the rotor is attached to the motor 2a. The rotational position detector 41a calculates the rotational position θa from the input signal of the rotational position sensor 4a and outputs it to the motor control device 1. The current detecting unit 7a is disposed between the inverter 3a and the motor 2a.

[0113] The inverter 3a has an inverter circuit 31a, a PWM signal drive circuit 32a, and a smoothing capacitor 33a. The PWM signal drive circuit 32a is connected to the motor control device 1 common to the PWM signal drive circuit 32 of the inverter 3, generates a PWM signal for controlling each switching element possessed by the inverter circuit 31a according to a gate signal input from the motor control device 1, and outputs to the inverter circuit 31a. The inverter circuit 31a and the smoothing capacitor 33a are connected to the high voltage battery 5 common to the inverter circuit 31 and the smoothing capacitor 33.

[0114] The torque command T* for the motor 2 and the torque command Ta* for the motor 2a are input to the motor control device 1. The motor control device 1 generates a gate signal for controlling the drive of the motor 2, 2a according to these torque commands by the method explained in the first embodiment, and outputs to the inverter 3, 3a, respectively. That is, the frequency of the carrier is adjusted by the voltage phase error operation section 164 operating the voltage phase error in a manner that can suppress the vibration and noise generated in the motor 2, 2a, respectively. Further, as for the electromagnetic excitation force reduction phase difference table 1644a and the torque pulsation reduction phase difference table 1644b referred to in this operation, for example, values that can most effectively reduce the electromagnetic excitation force or the torque pulsation in the motor 2, 2a, respectively, are set. Alternatively, values that are not the values that can most effectively reduce the electromagnetic excitation force or the torque pulsation in the motor 2, 2a, respectively, but can most effectively reduce the pulsation of the motor 2, 2a in total can be set.

[0115] The engine system 721 and the control unit 722 are connected to the motor 2. The engine system 721 is driven according to the control of the control unit 722 to rotate drive the motor 2. The motor 2 functions as a generator by being rotationally driven by the engine system 721, and generates alternating current. The alternating current generated by the motor 2 is converted into direct current by the inverter 3 and charged into the high voltage battery 5. Thus, the hybrid power system 72 can function as a series type hybrid power system. Further, the engine system 721 and the control unit 722 can be connected to the motor 2a.

[0116] (Third Embodiment) (Boost Converter System)

[0117] Next, the third embodiment of the present application will be explained. In this embodiment, a use case in a boost converter system will be explained.

[0118] Figure 13 A diagram showing the configuration of the boost converter system of the third embodiment of the present application. The boost converter system 73 has the motor control device 1, the motor 2, the inverter 3, the rotational position detector 41, and the high voltage battery 5 explained in the first embodiment, and has a boost converter 74.

[0119] The step-up converter 74 connects the switching elements 743, 744 in series, and the high-voltage battery 5 is connected to the intermediate connection point of the switching elements 743, 744 connected in series via the reactor 742. In addition, the capacitor 741 is connected in parallel with the high-voltage battery 5.

[0120] The step-up converter 74 is instructed by the motor control device 1 to cause the switching elements 743, 744 to perform switching operations, respectively, thereby stepping up the direct-current voltage supplied from the high-voltage battery 5 to a direct-current voltage optimal for the efficiency of the step-up converter system 73. Thus, direct current obtained by stepping up the high-voltage battery 5 is generated and supplied to the inverter 3. The inverter 3 operates in accordance with the gate signals output from the motor control device 1, and performs power conversion from the direct current stepped up by the step-up converter 74 to alternating current.

[0121] In the present embodiment, the direct-current voltage is stepped up by the step-up converter 74, so in the motor control device 1, the modulation factor calculation section 163 calculates the post-stepping-up modulation factor H' in accordance with the d-axis voltage command Vd* and the q-axis voltage command Vq*, the post-stepping-up direct-current voltage Hvdc', in accordance with the following equation (10).

[0122] H' = Vd2+ Vq2 / (Hvdc' / 2) ··· (10)

[0123] In the present embodiment, in the voltage phase error calculation section 164, the post-stepping-up modulation factor H' is referred to in accordance with the rotational frequency ωr and the torque command T* to determine the phase difference suitable for reduction of the electromagnetic excitation force and the phase difference suitable for reduction of the torque ripple, respectively, from the electromagnetic excitation force reduction phase difference table 1644a and the torque ripple reduction phase difference table 1644b.

[0124] (4th Embodiment)

[0125] Next, the 4th embodiment of the present application will be described. In the present embodiment, a case of application to an electric power steering system will be described.

[0126] Figure 14 A diagram showing the configuration of the electric power steering system of the 4th embodiment of the present application. The electric power steering system 61 has a drive control system 75 including the motor control device 1 described in the 1st embodiment and the redundant drive systems 102A, 102B. The electric power steering system 61 detects the torque of the steering wheel 62 with the torque sensor 63, and causes the drive control system 75 to operate in accordance with the torque. Thus, an assist torque corresponding to the input of the steering wheel 62 is generated and output to the steering mechanism 65 via the steering assist mechanism 64, thereby assisting the steering force. As a result, the steering mechanism 65 turns the tires 66, thereby controlling the traveling direction of the vehicle.

[0127] The electric power steering system of a vehicle is directly linked to the driver via the steering wheel, so vibrations and noise are easily transmitted to the driver, and the requirements for vibrations and noise are high. In particular, in a state where the driver is rotating the steering wheel at high speed, the operation of the motor is dominant in the cause of vibrations and noise compared to other factors. In view of this, the electric power steering system 61 of the present embodiment is able to effectively reduce vibrations in a state where the driver is rotating the steering wheel 62 at high speed, so a low-vibration and low-noise electric power steering system can be realized.

[0128] Figure 15 A diagram showing the configuration of the drive control system 75 in the electric power steering system 61 of the fourth embodiment of the present application. In the drive control system 75, the motor control device 1, the motor 2, and the high-voltage battery 5 are commonly connected to the redundant drive systems 102A, 102B. In the present embodiment, the motor 2 has two winding systems 21, 22, one winding system 21 constituting the drive system 102A and the other winding system 22 constituting the drive system 102B.

[0129] The drive system 102A has the inverter 3 and the rotational position detector 41, and the motor 2 is provided with the rotational position sensor 4 for detecting the rotational position θ of the rotor corresponding to the winding system 21. The motor 2 is rotationally driven by the alternating current generated by the inverter 3 to the winding system 21. In the drive system 102A, the current detection unit 7 is disposed between the inverter 3 and the motor 2.

[0130] The drive system 102B has the inverter 3a and the rotational position detector 41a, and the motor 2 is provided with the rotational position sensor 4a for detecting the rotational position θa of the rotor corresponding to the winding system 22. The motor 2 is rotationally driven by the alternating current generated by the inverter 3a to the winding system 22. In the drive system 102B, the current detection unit 7a is disposed between the inverter 3a and the motor 2. Further, the inverter 3a, the rotational position detector 41a, the rotational position sensor 4a, and the current detection unit 7a are respectively the same as the units described in the second embodiment. Figure 12

[0131] ​The torque command T* for the motor 2 is input to the motor control device 1. The motor control device 1 generates the gate signals for controlling the driving of the motor 2 according to the input torque command T* by the method explained in the first embodiment, and outputs them to the inverters 3, 3a, respectively. That is, the frequency of the carrier is adjusted by the voltage phase error operation section 164 operating the voltage phase error in a manner that can suppress the vibration and noise generated in the drive systems 102A, 102B, respectively. Further, as for the electromagnetic vibration force reduction phase difference table 1644a and the torque ripple reduction phase difference table 1644b referred to in this operation, for example, values that can most effectively reduce the electromagnetic vibration force or the torque ripple in each of the drive systems 102A, 102B are set. Alternatively, values that are not the values that can most effectively reduce the electromagnetic vibration force or the torque ripple in each of the drive systems 102A, 102B, but can most effectively reduce the sum of the ripples of the drive systems 102A, 102B can also be set.

[0132] Further, the above-described embodiments and various modifications are only examples, and the present application is not limited to these. In addition, the above-described embodiments and various modifications are explained, but the present application is not limited to these. Other modes thought within the scope of the technical idea of the present application are also included in the scope of the present application.

[0133] Symbol explanation

[0134] 1 motor control device, 2, 2a permanent magnet synchronous motor (motor), 3, 3a inverter, 4, 4a rotary position sensor, 5 high voltage battery, 7, 7a current detection unit, 11 current command generation section, 12 speed calculation section, 13 three phase / dq conversion current control section, 14 current control section, 15 dq / three phase voltage command conversion section, 16 carrier frequency adjustment section, 17 triangular wave generation section, 18 gate signal generation section, 31, 31a inverter circuit, 32, 32a PWM signal drive circuit, 33, 33a smoothing capacitor, 41, 41a rotary position detector, 61 electric power steering system, 72 series hybrid system, 73 step-up converter system, 74 step-up converter, 75 drive control system, 100 motor drive system, 102A, 102B drive system, 161 synchronous PWM carrier number selection section, 162 voltage phase operation section, 163 modulation rate operation section, 164 voltage phase error operation section, 165 synchronous carrier frequency operation section, 166 carrier frequency setting section, 721 engine system, 722 control unit, 1641 reference voltage phase operation section, 1642 pulsation frequency conversion section, 1643 pulsation contribution degree selection section, 1644a electromagnetic excitation force reduction phase difference table, 1644b torque pulsation reduction phase difference table, 1645 voltage phase difference conversion section, 1646 addition section, 1647 subtraction section.

Claims

1. A motor control device that controls driving of an alternating-current motor that is driven using alternating current by being connected to a power converter that performs power conversion from direct current to the alternating current, the motor control device characterized by comprising: a carrier generation section that generates a carrier; a carrier frequency adjustment section that adjusts a frequency of the carrier; and a gate signal generation section that pulse-width-modulates a voltage command corresponding to a torque command using the carrier to generate a gate signal for controlling an operation of the power converter, the carrier frequency adjustment section adjusts the frequency of the carrier in a manner that changes a phase difference of the voltage command from the carrier in accordance with the torque command and a rotational speed of the alternating-current motor, the carrier frequency adjustment section selects one of a torque ripple generated in a circumferential direction of the alternating-current motor and an electromagnetic exciting force generated in a radial direction of the alternating-current motor in accordance with the rotational speed of the alternating-current motor, and changes the phase difference in a manner that reduces the selected torque ripple or the electromagnetic exciting force.

2. The motor control device according to claim 1, characterized in that the carrier frequency adjustment section adjusts the frequency of the carrier in a manner that the frequency of the carrier becomes an integral multiple of a frequency of the voltage command.

3. The motor control device according to claim 1, characterized in that the carrier frequency adjustment section changes the phase difference in accordance with the torque command, the rotational speed, and a voltage amplitude ratio of the direct current to the alternating current.

4. The motor control device according to claim 1, characterized in that the carrier frequency adjustment section changes the phase difference in accordance with each of higher harmonic components of a fundamental wave current corresponding to the voltage command, the each of the higher harmonic components having a multiple of 6 as a frequency.

5. The motor control device according to claim 1, characterized in that the carrier frequency adjustment section judges which one of traction driving and regenerative driving the alternating-current motor is in accordance with the torque command, and changes the phase difference in accordance with a result of the judgment.

6. A motor control method that controls driving of an alternating-current motor that is driven using alternating current by being connected to a power converter that performs power conversion from direct current to the alternating current, the motor control method characterized by adjusting a frequency of a carrier in a manner that changes a phase difference of a voltage command corresponding to a torque command of the alternating-current motor from the carrier in accordance with the torque command and a rotational speed of the alternating-current motor, generating the carrier at the adjusted frequency, pulse-width-modulating the voltage command using the carrier to generate a gate signal for controlling an operation of the power converter, and selecting one of a torque ripple generated in a circumferential direction of the alternating-current motor and an electromagnetic exciting force generated in a radial direction of the alternating-current motor in accordance with the rotational speed of the alternating-current motor, and changing the phase difference in a manner that reduces the selected torque ripple or the electromagnetic exciting force.

7. The motor control method according to claim 6, characterized in that the frequency of the carrier is adjusted in a manner that the frequency of the carrier becomes an integral multiple of a frequency of the voltage command. ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ 8. The motor control method according to claim 6, characterized by changing the phase difference in accordance with the torque command, the rotational speed, and the voltage amplitude ratio of the direct current to the alternating current.

9. The motor control method according to claim 6, characterized by changing the phase difference in accordance with each of the high-order harmonic components of the fundamental wave current corresponding to the voltage command, which has a multiple of 6 as a frequency.

10. The motor control method according to claim 6, characterized by determining which of traction drive or regenerative drive the alternating-current motor is in accordance with the torque command, and changing the phase difference in accordance with a result of the determination.

11. A hybrid system characterized by, provided with: the motor control device according to any one of claims 1 to 5; a power converter that operates in accordance with the gate signal output from the motor control device, and performs power conversion from direct current to alternating current; an alternating-current motor that is driven using the alternating current; and an engine system that is connected to the alternating-current motor.

12. A boost converter system characterized by, provided with: the motor control device according to any one of claims 1 to 5; a step-up converter that is connected to a direct-current power supply, and generates direct current obtained by stepping up the direct-current power supply in accordance with control by the motor control device; and a power converter that operates in accordance with the gate signal output from the motor control device, and performs power conversion from the direct current stepped up by the step-up converter to alternating current. provided with:

13. An electric power assisted steering system characterised in that, the motor control device according to any one of claims 1 to 5; a plurality of power converters that operate in accordance with the gate signal output from the motor control device, and respectively perform power conversion from direct current to alternating current; and an alternating-current motor that has a plurality of winding systems, and is driven by causing the plurality of winding systems to flow the alternating current respectively generated by the plurality of power converters, a vehicle is controlled using the alternating-current motor. ​ ​

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