Wireless energy transfer system

Abstract

The present invention relates to a wireless energy transfer system. Described herein are a source resonator coupled to an energy source having a Q-factor Q<subgt;1< / subgt; > 100 and a characteristic dimension x<subgt;1< / subgt>, and a resonator having a Q-factor Q<subgt;2< / subgt; > 100 and a characteristic dimension x

Description

[0001] This application is a divisional application of a patent application with an application date of September 25, 2009, a divisional application number of 201611115306.7, and an invention title of "Wireless Energy Transfer System".

[0002] Cross - reference to related applications

[0003] This application claims priority to the following U.S. patent applications, each of which is incorporated herein by reference in its entirety:

[0004] U.S. Application No. 61 / 100,721, filed September 27, 2008; U.S. Application No. 61 / 108,743, filed October 27, 2008; U.S. Application No. 61 / 147,386, filed January 26, 2009; U.S. Application No. 61 / 152,086, filed February 12, 2009; U.S. Application No. 61 / 178,508, filed May 15, 2009; U.S. Application No. 61 / 182,768, filed June 1, 2009; U.S. Application No. 61 / 121,159, filed December 9, 2008; U.S. Application No. 61 / 142,977, filed January 7, 2009; U.S. Application No. 61 / 142,885, filed January 6, 2009; U.S. Application No. 61 / 142,796, filed January 6, 2009; U.S. Application No. 61 / 142,889, filed January 6, 2009; U.S. Application No. 61 / 142,880, filed January 6, 2009; U.S. Application No. 61 / 142,818, filed January 6, 2009; U.S. Application No. 61 / 142,887, filed January 6, 2009; U.S. Application No. 61 / 156,764, filed March 2, 2009; U.S. Application No. 61 / 143,058, filed January 7, 2009; U.S. Application No. 61 / 152,390, filed February 13, 2009; U.S. Application No. 61 / 163,695, filed March 26, 2009; U.S. Application No. 61 / 172,633, filed April 24, 2009; U.S. Application No. 61 / 169,240, filed April 14, 2009; and U.S. Application No. 61 / 173,747, filed April 29, 2009.

[0005] Background Technical Field

[0006] This disclosure relates to wireless energy transfer, also known as wireless power transfer. Background Art

[0007] A variety of known radiative or far-field and non-radiative or near-field techniques can be used to wirelessly transfer energy or power. For example, radiative wireless information transfer using low-directivity antennas (such as those used in radio and cellular communication systems and home computer networks) can be considered wireless energy transfer. However, such radiative transfer is very inefficient because only a small fraction of the supplied or radiated power is captured, namely the fraction that is directed along and overlaps with the receiver. Most of the power is radiated away in all other directions and lost in free space. Such inefficient power transfer is acceptable for data transmission, but is not practical for transferring useful amounts of electrical energy for work purposes (such as powering or charging electrical devices). One way to improve the transfer efficiency of some energy transfer schemes is to use directional antennas to confine and preferably direct the radiated energy towards the receiver. However, these directional radiative schemes may require an unobstructed line of sight and potentially complex tracking and steering mechanisms in the case of mobile transmitters and / or receivers. Additionally, such schemes may pose a danger to objects or people passing through or across the beam when moderate or large amounts of power are being transmitted. Known non-radiative or near-field wireless energy transfer schemes, often referred to as inductive or traditional induction, do not (intentionally) radiate power, but instead use an oscillating current flowing through a primary coil to generate an oscillating magnetic near-field that induces a current in a nearby receiving or secondary coil. Traditional induction schemes have demonstrated the transfer of moderate to large amounts of power, however only over very short distances and with a very small offset tolerance between the main power unit and the auxiliary receiver unit. Electrical transformers and proximity chargers are examples of devices that utilize this known short-range, near-field energy transfer scheme.

[0008] Accordingly, there is a need for a wireless power transfer scheme that can transfer useful amounts of electrical power over mid-range distances or alignment offsets. Such a wireless power transfer scheme should be able to achieve useful energy transfer over greater distances and alignment offsets than those achieved by traditional induction schemes, but without the limitations and risks inherent in radiative transmission schemes. SUMMARY OF THE INVENTION

[0009] This disclosure presents a non-radiative or near-field wireless energy transfer scheme capable of delivering useful amounts of power over mid-range distances and alignment offsets. The techniques of the present invention use coupled electromagnetic resonators with long-lived oscillatory resonant modes to transfer power from a power source to a power drain. The techniques are comprehensive and can be applied to a wide range of resonators, even in the context of the specific examples of electromagnetic resonators disclosed herein. If a resonator is designed such that the energy stored by the electric field is predominantly confined within the structure and the energy stored by the magnetic field is predominantly in the region surrounding the resonator, then the energy exchange is predominantly mediated by the resonant magnetic near-field. These types of resonators can be referred to as magnetic resonators. If a resonator is designed such that the energy stored by the magnetic field is predominantly confined within the structure and the energy stored by the electric field is predominantly in the region surrounding the resonator, then the energy exchange is predominantly mediated by the resonant electric near-field. These types of resonators can be referred to as electric resonators. Either type of resonator can also be referred to as an electromagnetic resonator. Both types of resonators are disclosed herein.

[0010] The omnidirectional but fixed (lossless) nature of the near-field of the resonators we disclose enables efficient wireless energy transfer over mid-range distances over a wide range of directions and resonator orientations, suitable for charging, powering, or simultaneously powering and charging multiple electronic devices. As a result, a system can have a variety of possible applications, where a first resonator connected to a power source is in one location and a second resonator potentially connected to electrical / electronic devices, batteries, powering and charging circuits, etc. is in a second location, and where the distance from the first resonator to the second resonator is on the order of a few centimeters to several meters. For example, a first resonator connected to a wired power grid can be located on the ceiling of a room, while other resonators connected to devices such as robots, vehicles, computers, communication devices, medical devices, etc. move about within the room, and where these devices constantly or intermittently receive power wirelessly from the source resonator. For this one example, one can envision many applications where the systems and methods disclosed herein can provide wireless power over mid-range distances, including consumer electronics, industrial applications, infrastructure powering and lighting, transportation vehicles, electronic gaming, military applications, etc.

[0011] When the resonators are tuned to substantially the same frequency and the losses in the system are minimized, the energy exchange between two electromagnetic resonators can be optimized. A wireless energy transfer system can be designed such that the "coupling time" between resonators is much shorter than the "loss time" of the resonators. Thus, the systems and methods described herein can utilize high quality factor (high-Q) resonators with low intrinsic loss rates. Additionally, the systems and methods described herein can use sub-wavelength resonators with near fields that extend significantly longer than the characteristic dimensions of the resonators, such that the near fields of the resonators exchanging energy overlap at mid-range distances. This is an operating region not previously practiced and is significantly different from conventional inductive designs.

[0012] It is important to recognize the difference between the high-Q magnetic resonator scheme disclosed herein and known short-range or near-inductive schemes, namely that those known schemes do not conventionally utilize high-Q resonators. Using coupled mode theory (CMT) (see, for example, Waves and Fields in Optoelectronics , H.A. Haus, Prentice Hall, 1984), it can be shown that the high-Q resonator coupling mechanism can achieve efficient power delivery that is several orders of magnitude higher than the power delivery between resonators separated by mid-range distances achieved by conventional inductive schemes. Coupled high-Q resonators have demonstrated efficient energy transfer at mid-range distances and improved efficiency and offset tolerance in short-range energy transfer applications.

[0013] The systems and methods described herein can provide near-field wireless energy transfer via strongly coupled high-Q resonators, a technology with the potential to safely transfer power levels from picowatts to kilowatts at distances much greater than those achievable using conventional inductive techniques. For various general systems of strongly coupled resonators, efficient energy transfer can be achieved, such as systems of strongly coupled acoustic resonators, atomic resonators, mechanical resonators, etc., as originally described by M.I.T. in its publications "Efficient wireless non-radiative mid-range energy transfer", Annals of Physics, vol. 323, Issue 1, p. 34 (2008) and "Wireless Power Transfer via Strongly Coupled Magnetic Resonances", Science, vol. 317, no. 5834, p. 83, (2007). Disclosed herein are electromagnetic resonators and systems of coupled electromagnetic resonators, more specifically also referred to as coupled magnetic resonators and coupled electric resonators, having an operating frequency below 10 GHz.

[0014] The present disclosure describes wireless energy transfer techniques, also known as wireless power transfer techniques. Throughout the present disclosure, we may interchangeably use the terms wireless energy transfer, wireless power transfer, wireless power transmission, etc. We may refer to supplying energy or power from a source, an AC or DC source, a battery, a source resonator, a power supply, a generator, a solar panel, and a collector, etc. to a device, a remote device, multiple remote devices, one or more device resonators, etc. We may describe an intermediate resonator that extends the range of a wireless energy transfer system by allowing energy to jump, transfer through, be temporarily stored, be partially dissipated, or allow transfer from a source resonator to any combination of other devices and intermediate resonators in any manner, so that an energy transfer network or string or extended path can be realized. A device resonator can receive energy from a source resonator, convert a part of the energy into electric power for powering and charging the device, and simultaneously transfer a part of the received energy to other devices or mobile device resonators. Energy can be transferred from a source resonator to multiple device resonators, significantly extending the distance over which energy can be wirelessly transferred. Multiple system architectures and resonator designs can be used to implement a wireless power transfer system. The system can include a single source or multiple sources for transferring power to a single device or multiple devices. A resonator can be designed as a source or a device resonator, or it can be designed as a repeater. In some cases, a resonator can be both a device and a source resonator, or it can be switched from operating as a source to operating as a device or a repeater. Those skilled in the art will understand that multiple system architectures can be supported by a wide range of resonator designs and functions described in this application.

[0015] In the wireless energy transfer system we describe, wirelessly supplied power or energy can be used to directly power a remote device, or a device can be coupled to an energy storage unit (or other types of power-consuming devices) such as a battery, a farad capacitor, a supercapacitor, etc., where the energy storage element can be wirelessly charged or recharged, and / or where the wireless power transfer mechanism is merely a supplement to the device's main power source. A device can be powered by a hybrid battery / energy storage device such as one having an integrated storage capacitor. In addition, new batteries and energy storage devices can be designed to take advantage of the operational improvements enabled by a wireless power transfer system.

[0016] Other power management schemes include using wirelessly supplied power to recharge a battery or charge a storage unit while the device being powered is turned off, in an idle state, in a sleep mode, etc. The battery or storage unit can be charged or recharged at a high (fast) or low (slow) rate. The battery or storage unit can be trickle charged or float charged. Multiple devices can be charged or powered simultaneously in parallel, or power delivery to multiple devices can be serialized such that one or more devices receive power for a period of time after power delivery to other devices has been switched to other devices. Multiple devices can share power from one or more sources simultaneously, or in a time-division multiplexed manner, or in a frequency-division multiplexed manner, or in a space-division multiplexed manner, or in an orientation-division multiplexed manner, or in any combination of time and frequency and space and orientation multiplexing. Multiple devices can share power with each other, with at least one device being continuously, intermittently, periodically, occasionally, or temporarily reconfigured to operate as a wireless power source. Those skilled in the art will understand that there are multiple ways to power and / or charge a device, and the multiple ways can be applied to the technologies and applications described herein.

[0017] Wireless energy transfer has a variety of possible applications, including, for example, placing a source (e.g., one connected to a wired power grid) on the ceiling of a room, under the floor, or in a wall, while placing devices such as robots, vehicles, computers, PDAs, etc. indoors or allowing them to move freely indoors. Other applications can include powering or recharging electric engine vehicles such as buses and / or hybrid cars and medical devices such as wearable or implantable devices. Additional exemplary applications include the ability to power or recharge stand-alone electronic devices (e.g., laptop computers, cellular phones, portable music players, household robots, GPS navigation systems, displays, etc.), sensors, industrial and manufacturing equipment, medical devices and monitors, household appliances and tools (e.g., lights, fans, drills, saws, heaters, displays, televisions, countertop appliances, etc.), military equipment, warm or lighting clothing, communication and navigation devices, including devices embedded in vehicles, clothing, and protective clothing such as helmets, bulletproof vests, and vests, etc., and the ability to transfer power to physically isolated devices such as implanted medical devices, to hidden, buried, implanted, or embedded sensors or tags, and / or from rooftop solar panels to indoor distribution panels.

[0018] In one aspect, the systems disclosed herein include a source resonator having a quality factor Q1 and a characteristic dimension x1 and coupled to a generator, and a second resonator having a quality factor Q2 and a characteristic dimension x2 and coupled to a load located at a distance D from the source resonator, wherein the source resonator and the second resonator are coupled to wirelessly exchange energy between the source resonator and the second resonator, and wherein

[0019] Q1 can be less than 100. Q2 can be less than 100. The system can include a third resonator having a quality factor Q3 configured to non-radiatively transfer energy with a source and a second resonator, where, and Q3 can be less than 100.

[0020] The source resonator can be coupled to a generator with a direct electrical connection. The system can include an impedance matching network, where the source resonator is coupled and impedance matched to the generator by a direct electrical connection. The system can include a tunable circuit, where the source resonator is coupled to the generator by a tunable circuit having a direct electrical connection. The tunable circuit can include a variable capacitor. The tunable circuit can include a variable inductor. At least one direct electrical connection can be configured to substantially maintain the resonant mode of the source resonator. The source resonator can have a first terminal, a second terminal, and a center terminal, and the impedance between the first terminal and the center terminal and between the second terminal and the center terminal can be substantially equal. The source resonator can include a capacitively loaded loop having a first terminal, a second terminal, and a center terminal, where the impedance between the first terminal and the center terminal and between the second terminal and the center terminal is substantially equal. The source resonator can be coupled to an impedance matching network, and the impedance matching network can also include a first terminal, a second terminal, and a center terminal, where the impedance between the first terminal and the center terminal and between the second terminal and the center terminal is substantially equal.

[0021] The first and second terminals can be directly coupled to a generator and driven with an oscillating signal having a phase difference of approximately 180 degrees. The source resonator can have a resonant frequency ω1, and the first and second terminals can be directly coupled to a generator and driven with an oscillating signal having a frequency substantially equal to the resonant frequency ω1. The center terminal can be connected to electrical ground. The source resonator can have a resonant frequency ω1, and the first and second terminals can be directly coupled to a generator and driven with a frequency substantially equal to the resonant frequency. The system can include a plurality of capacitors coupled to the generator and the load. Each of the source resonator and the second resonator can be enclosed in a low-loss tangent material. The system can include a power conversion circuit, wherein the second resonator is coupled to the power conversion circuit to deliver DC power to the load. The system can include a power conversion circuit, wherein the second resonator is coupled to the power conversion circuit to deliver AC power to the load. The system can include a power conversion circuit, wherein the second resonator is coupled to the power conversion circuit to deliver both AC and DC power to the load. The system can include a power conversion circuit and a plurality of loads, wherein the second resonator is coupled to the power conversion circuit, and the power conversion circuit is coupled to the plurality of loads. The impedance matching network can include capacitors. The impedance matching network can include inductors.

[0022] Throughout this disclosure, we can refer to certain circuit components such as capacitors, inductors, resistors, diodes, switches, etc. as circuit components or elements. We can also refer to series and parallel combinations of these components as elements, networks, topologies, circuits, etc. We can describe combinations of capacitors, diodes, varactors, transistors, and / or switches as adjustable impedance networks, tuning networks, matching networks, tuning elements, etc. We can also refer to "self-resonant" objects that have both capacitance and inductance distributed throughout the object (or partially distributed, as opposed to being lumped separately). Those skilled in the art will understand that adjusting and controlling variable components within a circuit or network can adjust the performance of that circuit or network, and those adjustments can generally be described as tuning, adjusting, matching, correcting, etc. In addition to adjusting tunable components such as inductors and capacitors or groups of inductors and capacitors, other methods of tuning or adjusting a wireless power transfer system can be used alone.

[0023] Unless otherwise defined, all technical and / or scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs. In case of conflict with publications, patent applications, patents, and other references mentioned or incorporated herein by reference, this specification (including definitions) will prevail.

[0024] Any of the above features may be used alone or in combination without departing from the scope of the present disclosure. Other features, objects, and advantages of the systems and methods disclosed herein will be apparent from the following detailed description and the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS

[0025] Figure 1 Figures (a) and (b) therein depict an exemplary wireless power system including a source resonator 1 and a device resonator 2 separated by a distance D.

[0026] Figure 2 Shows exemplary resonators labeled according to the tagging convention described in the present disclosure. Note that no extraneous objects or additional resonators are shown in the vicinity of resonator 1.

[0027] Figure 3 Shows exemplary resonators labeled according to the tagging convention described in the present disclosure in the presence of a "loading" object.

[0028] Figure 4 Shows exemplary resonators labeled according to the tagging convention described in the present disclosure in the presence of a "perturbing" object.

[0029] Figure 5 Shows a plot of efficiency η versus strong coupling coefficient of.

[0030] Figure 6 Figure (a) therein shows a circuit diagram of an example of a resonator, Figure 6 Figure (b) therein shows an illustration of an example of a capacitively loaded inductor loop magnetic resonator, Figure 6 Figure (c) therein shows a diagram of a self-resonant coil with distributed capacitance and inductance, Figure 6 Figure (d) therein shows a simplified diagram of the electric and magnetic field lines associated with an exemplary magnetic resonator of the present disclosure, and Figure 6 Figure (e) therein shows an illustration of an example of an electric resonator.

[0031] Figure 7 Shows a plot of the "quality factor" Q (solid line) as a function of frequency for an exemplary resonator that can be used for wireless power transfer at MHz frequencies. The absorptive Q (dash line) increases with frequency, while the radiative Q (dot line) decreases with frequency, thus causing the total Q to peak at a specific frequency.

[0032] Figure 8 Shows a diagram of a resonator structure, with its characteristic dimensions, thickness, and width all indicated.

[0033] Figure 9 900A and 900B therein show diagrams of exemplary inductive loop elements.

[0034] Figure 10 Examples (a) and (b) in [reference] show two examples of trace structures formed on a printed circuit board and used to implement inductive elements in a magnetic resonator structure.

[0035] Figure 11 Example (a) in [reference] shows a perspective view of a planar magnetic resonator, Figure 11 example (b) in [reference] shows a perspective view of two planar magnetic resonators with various geometries, and Figure 11 example (c) in [reference] shows a perspective view of two planar magnetic resonators separated by a distance D.

[0036] Figure 12 is a perspective view of an example of a planar magnetic resonator.

[0037] Figure 13 is a perspective view of a planar magnetic resonator arrangement with a circular resonator coil.

[0038] Figure 14 is a perspective view of the active area of a planar magnetic resonator.

[0039] Figure 15 is a perspective view of an application of a wireless power transfer system, where a source at the center of a table powers multiple devices placed around the source.

[0040] Figure 16 Example (a) in [reference] shows a 3D finite element model of a copper and magnetic material structure driven by a square loop of current around a blocking point at its center. In this example, the structure can consist of two boxes made of a conductive material such as copper, covered with a layer of magnetic material and connected by a piece of magnetic material. The interiors of the two conductive boxes in this example will be shielded from the AC electromagnetic field generated outside the boxes and can accommodate lossy objects that may reduce the Q of the resonator or sensitive components that may be negatively affected by the AC electromagnetic field. Also shown are the calculated magnetic field streamlines generated by this structure, indicating that the magnetic field lines tend to follow the lower magnetic resistance paths in the magnetic material. Figure 16 Example (b) in [reference] shows the interaction between two identical structures as indicated by the calculated magnetic field streamlines as shown in (a). Due to symmetry and to reduce computational complexity, only half of the system is modeled (however, the calculation assumes a symmetric arrangement of the other half).

[0041] Figure 17 shows an equivalent circuit representation of a magnetic resonator including a wire wound N times around a structure, which may contain a magnetically permeable material. The inductance is implemented using a conductive loop wound around a structure including a magnetic material, and the resistor represents the loss mechanism in the system (R wireFor resistive losses in a loop, R μ represents the equivalent series resistance of the structure surrounded by the loop). The losses can be minimized to achieve a high-Q resonator.

[0042] Figure 18 Shows a finite element method (FEM) simulation of two highly conductive surfaces above and below a disk composed of a lossy dielectric material in an external magnetic field at a frequency of 6.78 MHz. Note that the magnetic field is uniform before the disk, and conductive materials are introduced into the simulation environment. This simulation is performed in a cylindrical coordinate system. The image is azimuthally symmetric about the r = 0 axis. The lossy electrolyte disk has ∈ r = 1 and σ = 10 S / m.

[0043] Figure 19 Shows a diagram of a magnetic resonator with a lossy object in its vicinity that is completely covered by a highly conductive surface.

[0044] Figure 20 Shows a diagram of a magnetic resonator with a lossy object in its vicinity that is partially covered by a highly conductive surface.

[0045] Figure 21 Shows a diagram of a magnetic resonator with a lossy object in its vicinity that is placed above a highly conductive surface.

[0046] Figure 22 Shows a diagram of a fully wireless projector.

[0047] Figure 23 Shows the magnitudes of the electric and magnetic fields along the diameter of a circular loop inductor and along the axis of the loop inductor.

[0048] Figure 24 Shows a diagram of a magnetic resonator and its enclosure and the necessary but lossy objects placed (a) in the corners of the enclosure, as far as possible from the resonator structure or (b) at the center of the surface enclosed by the inductive elements in the magnetic resonator.

[0049] Figure 25 Shows a diagram of a magnetic resonator with a highly conductive surface above it and a lossy object that can be brought near the resonator but above the highly conductive sheet.

[0050] Figure 26 In (a) shows an axisymmetric FEM simulation of a thin conductive (copper) cylinder or disk (20 cm in diameter and 2 cm in height) exposed to an initially uniform externally applied magnetic field (gray magnetic flux lines) along the z-axis. The axis of symmetry is at r = 0. The magnetic streamlines shown originate at z = -∞, where they are spaced 1 cm apart from r = 3 cm to r = 10 cm. The axis scale is in meters. Figure 26(b) in shows the same structure and applied field as in (a), except that the conducting cylinder has been modified to include a 0.25 mm thick magnetic material layer (invisible) with μ r ' = 40 on its outer surface. Note that the degree of magnetic flux line deflection away from the cylinder is significantly smaller than in (a).

[0051] Figure 27 shows an axially symmetric view based on Figure 26 a variation of the system shown. Only one surface of the lossy material is covered by a layered structure of copper and magnetic material. As shown, the inductor loop is placed on one side of the copper and magnetic material structure opposite the lossy material.

[0052] Figure 28 (a) in depicts the general topology of a matching circuit including an indirect coupling to a high-Q inductive element.

[0053] Figure 28 (b) in shows a block diagram of a magnetic resonator including a conductor loop inductor and a tunable impedance network. The physical electrical connection to this resonator can be made to terminal connections.

[0054] Figure 28 (c) in depicts the general topology of a matching circuit directly coupled to a high-Q inductive element.

[0055] Figure 28 (d) in depicts the general topology of a symmetric matching circuit directly coupled to a high-Q inductive element and driven antisymmetrically (balanced drive).

[0056] Figure 28 (e) in depicts the general topology of a matching circuit directly coupled to a high-Q inductive element and grounded at the symmetric point of the main resonator (unbalanced drive).

[0057] Figure 29 (a) in and Figure 29 (b) in depict two topologies of a matching circuit transformer coupled (i.e., indirectly or inductively) to a high-Q inductive element. Figure 29 (c) in the highlighted part of the Smith chart depicts the complex impedance (resulting from the L and R of the inductive element) that can be matched to an arbitrary real impedance Z0 in the case of ωL2 = 1 / ωC2 from the topology of (b). Figure 31 (b) in

[0058] Figure 30 (a), (b), (c), (d), (e), (f) in depict six topologies of a matching circuit directly coupled to a high-Q inductive element and including a capacitor in series with Z0. Driven with a common-mode signal at the input terminals Figure 30the topological structures shown in (a), (b), and (c) therein, while Figure 30 the topological structures shown in (d), (e), and (f) therein are symmetric and receive balanced driving. Figure 30 The highlighted part of the Smith chart in (g) therein depicts the complex impedance that can be matched by these topological structures. Figure 30 (h), (i), (j), (k), (l), and (m) therein depict six topological structures of a matching circuit that is directly coupled to a high-Q inductive element and includes an inductor in series with Z0.

[0059] Figure 31 (a), (b), and (c) therein depict three topological structures of a matching circuit that is directly coupled to a high-Q inductive element and includes a capacitor in series with Z0, which is grounded at the center point of the capacitor and receives unbalanced driving. Figure 31 The highlighted part of the Smith chart in (d) therein depicts the complex impedance that can be matched by these topological structures. Figure 31 (e), (f), and (g) therein depict three topological structures of a matching circuit that is directly coupled to a high-Q inductive element and includes an inductor in series with Z0.

[0060] Figure 32 (a), (b), and (c) therein depict three topological structures of a matching circuit that is directly coupled to a high-Q inductive element and includes a capacitor in series with Z0. It is grounded through a tapping at the center point of the inductor loop and receives unbalanced driving. Figure 32 The highlighted part of the Smith chart in (d) therein depicts the complex impedance that can be matched by these topological structures, Figure 32 (e), (f), and (g) therein depict three topological structures of a matching circuit that is directly coupled to a high-Q inductive element and includes an inductor in series with Z0.

[0061] Figure 33 (a), (b), (c), (d), (e), and (f) therein depict six topological structures of a matching circuit that is directly coupled to a high-Q inductive element and includes a capacitor in parallel with Z0. Driven by the common-mode signal at the input terminals Figure 33 the topological structures shown in (a), (b), and (c) therein, while Figure 33 the topological structures shown in (d), (e), and (f) therein are symmetric and receive balanced driving. Figure 33 The highlighted part of the Smith chart in (g) therein depicts the complex impedance that can be matched by these topological structures. Figure 33(h), (i), (j), (k), (l), (m) therein depict six topologies of a matching circuit that is directly coupled to a high-Q inductive element and includes an inductor in parallel with Z0.

[0062] Figure 34 (a), (b), (c) therein depict three topologies of a matching circuit that is directly coupled to a high-Q inductive element and includes a capacitor in parallel with Z0. It is grounded at the center point of the capacitor and receives an unbalanced drive. Figure 34 The highlighted portion of the Smith chart in (d) therein depicts the complex impedances that can be matched by these topologies. Figure 34 (e), (f), (g) therein depict three topologies of a matching circuit that is directly coupled to a high-Q inductive element and includes an inductor in parallel with Z0.

[0063] Figure 35 (a), (b), (c) therein depict three topologies of a matching circuit that is directly coupled to a high-Q inductive element and includes a capacitor in parallel with Z0. It is grounded through a tap at the center point of the inductor loop and receives an unbalanced drive. Figure 35 The highlighted portions of the Smith charts in (d), (e), and (f) therein depict the complex impedances that can be matched by these topologies.

[0064] Figure 36 (a), (b), (c), (d) therein depict four network topologies that are designed to produce a total variable capacitance with a finer tuning resolution on a variable capacitor and certain fixed and variable capacitors with reduced voltages.

[0065] Figure 37 (a) and (b) therein depict two network topologies of a fixed capacitor and a variable inductor that are designed to produce a total variable capacitance.

[0066] Figure 38 Depicts a high-level block diagram of a wireless power transfer system.

[0067] Figure 39 Depicts a block diagram of an exemplary wirelessly powered device.

[0068] Figure 40 Depicts a block diagram of a source of an exemplary wireless power transfer system.

[0069] Figure 41 Shows an equivalent circuit diagram of a magnetic resonator. The short slashes through the capacitor symbols indicate that the capacitors represented can be fixed or variable. A port parameter measurement circuit can be configured to measure certain electrical signals, and the amplitude and phase of the signals can be measured.

[0070] Figure 42 A circuit diagram of a magnetic resonator is shown in which a voltage - controlled capacitor is used to implement an adjustable impedance network. Such embodiments can be adjusted, tuned, or controlled by a circuit including a programmable or controllable voltage source and / or a computer processor. The voltage - controlled capacitor can be adjusted in response to data measured by a port parameter measurement circuit and processed by a measurement analysis and control algorithm and hardware. The voltage - controlled capacitor can be a switched capacitor bank.

[0071] Figure 43 An end - to - end wireless power transfer system is shown. In this example, both the source and the device contain port measurement circuits and processors. The box labeled "coupler / switch" indicates that the port measurement circuit can be connected to the resonator by a directional coupler or a switch, enabling measurement, adjustment, and control of the source and device resonators either in combination with the power transfer function or separately.

[0072] Figure 44 An end - to - end wireless power transfer system is shown. In this example, only the source contains a port measurement circuit and a processor. In this case, the operating characteristics of the device resonator can be fixed or can be adjusted by an analog control circuit and do not require a control signal generated by a processor.

[0073] Figure 45 An end - to - end wireless power transfer system is shown. In this example, both the source and the device contain port measurement circuits, but only the source contains a processor. Data from the device is transmitted via a wireless communication channel that can be implemented with a separate antenna or by some modulation of the source drive signal.

[0074] Figure 46 An end - to - end wireless power transfer system is shown. In this example, only the source contains a port measurement circuit and a processor. Data from the device is transmitted via a wireless communication channel that can be implemented with a separate antenna or by some modulation of the source drive signal.

[0075] Figure 47 A coupled magnetic resonator is shown that can automatically adjust its frequency and impedance using an algorithm implemented with a processor or computer.

[0076] Figure 48 A varactor array is shown.

[0077] Figure 49 A device (laptop computer) wirelessly powered or charged by a source is shown, where the source and device resonators are physically separated from the source and device but are electrically connected to them.

[0078] Figure 50 a) in shows an illustration of a laptop computer application being wirelessly powered or charged, where the device resonator is inside the laptop computer case and not visible.

[0079] Figure 50 b) in it is an illustration of a laptop computer application that is wirelessly powered and charged, where the resonator is under the laptop computer base and is electrically connected to the laptop computer power input through a cable.

[0080] Figure 50 c) in it is an illustration of a laptop computer application that is wirelessly powered or charged, where the resonator is attached to the laptop computer base.

[0081] Figure 50 d) in it is an illustration of a laptop computer application that is wirelessly powered and charged, where the resonator is attached to the laptop computer display.

[0082] Figure 51 is an illustration of a rooftop PV panel with wireless power transfer. Detailed Description

[0083] As described above, the present disclosure relates to coupled electromagnetic resonators having a long - lived oscillating resonance mode that can wirelessly transfer power from a power source to a power - consuming device. However, the technology is not limited to electromagnetic resonators, but is general and can be applied to a variety of resonators and resonant objects. Therefore, we first describe the general technology and then disclose electromagnetic examples for wireless energy transfer.

[0084] Resonator

[0085] A resonator can be defined as a system capable of storing energy in at least two different forms, and the stored energy oscillates between the two forms. The resonance has a specific oscillation mode, which has a resonance (modal) frequency f and a resonance (modal) field. The angular resonance frequency ω can be defined as ω = 2πf, the resonance wavelength λ can be defined as λ = c / f, where c is the speed of light, and the resonance period T can be defined as T = 1 / f = 2π / ω. In the absence of loss mechanisms, coupling mechanisms, or external energy supply or consumption mechanisms, the total resonator stored energy W will remain fixed, and the two energy forms will oscillate, where one will be at a maximum when the other is at a minimum, and vice versa.

[0086] In the absence of extraneous materials or objects, Figure 1 the energy in the illustrated resonator 102 may decay or be lost to intrinsic losses. The resonator field then obeys the following linear equation:

[0087]

[0088] where the variable a(t) is the resonator field amplitude, which is defined such that by |a(t)| 2To give the energy contained within the resonator. Γ is the natural energy decay or loss rate (e.g., due to absorption and radiation losses).

[0089] The quality factor or Q-factor or Q of the resonator that characterizes the energy decay is inversely proportional to these energy losses. It can be defined as Q = ω*W / P, where P is the time-averaged power dissipated in the steady state. That is, a resonator 102 with a high Q has relatively low natural losses and can store energy for a relatively long time. Since the resonator loses energy at its natural decay rate 2Γ, its Q, also known as its natural Q value, is given by Q = ω / 2Γ. The quality factor also represents the number of oscillation periods T that cause the energy in the resonator to decay by a factor of e.

[0090] As described above, we define the quality factor or Q of the resonator to be due only to natural loss mechanisms. A subscript such as Q1 indicates the resonator to which the Q refers (in this case, resonator 1). Figure 2 Shows an electromagnetic resonator 102 labeled according to this convention. Note that in this figure, there are no extraneous objects or additional resonators near resonator 1.

[0091] Depending on various factors such as the distance between the resonator and an object or other resonator, the material composition of the object or other resonator, the structure of the first resonator, the power in the first resonator, etc., extraneous objects and / or additional resonators near the first resonator can perturb or load the first resonator, thereby perturbing or loading the Q of the first resonator. The unintentional external energy losses or coupling mechanisms to extraneous materials and objects near the resonator can be referred to as causing a "perturbation" to the Q of the resonator and can be indicated by a subscript within parentheses (). The expected external energy losses associated with energy transfer via coupling to other resonators and generators and loads in a wireless energy transfer system can be referred to as "loading" the Q of the resonator and can be indicated by a subscript within square brackets [].

[0092] The Q of resonator 102 that is connected or coupled to generator g or load 302l can be referred to as the "loaded quality factor" or "loaded Q", and as Figure 3 shown, can be denoted by Q [g] or Q[l]. Typically, more than one generator or load 302 can be connected to resonator 102. However, instead of listing those generators or loads separately, we use "g" and "l" to refer to the equivalent circuit loading imposed by the combination of the generator and load. In a general description, we can use the subscript "l" to refer to the generator or load connected to the resonator.

[0093] In some discussions herein, we define the "loaded quality factor" or "loaded Q", due to a generator or load connected to a resonator, as δQ [l] , where 1 / δQ [l] ≡1 / Q [l] - 1 / Q. Note that the larger the loaded Q of the generator or load, i.e., δQ [l] , the less the loaded Q, i.e., Q [l] deviates from the unloaded Q of the resonator.

[0094] In the presence of an extraneous object 402p that is not intended to be part of the energy transfer system, the Q of the resonator can be referred to as the "perturbed quality factor" or "perturbed Q", and as Figure 4 shown can be denoted by Q (p) . Typically, there can be many extraneous objects represented as p1, p2, etc. or a set of extraneous objects {p} that perturb the Q of the resonator 102. In such a case, the perturbed Q can be denoted as Q (p1+p2+...) or Q ({p}) . For example, Q 1(brick+wood) can represent the perturbed quality factor of the first resonator in a system for wireless power transfer in the presence of a brick or a piece of wood, and Q 2({office}) can represent the perturbed quality factor of the second resonator in a system for wireless power transfer in an office environment.

[0095] In some discussions herein, we define the "perturbation quality factor" or "perturbation Q", due to an extraneous object p, as δQ (p) , where 1 / δQ (p) ≡1 / Q (p) - 1 / Q. As described above, the perturbation quality factor can be due to multiple extraneous objects p1, p2, etc. or a set of extraneous objects {p}. The larger the perturbation Q of the object, i.e., δQ (p) , the less the perturbed Q, i.e., Q (p) deviates from the unperturbed Q of the resonator.

[0096] In some discussions herein, we also define Θ (p) ≡Q (p) / Q and, in the presence of an extraneous object, refer to it as the "quality factor insensitivity" or "Q insensitivity" of the resonator. Subscripts such as Θ 1(p) indicate the resonator to which the perturbed or unperturbed quality factor refers, i.e., Θ 1(p) ≡Q 1(p) / Q1.

[0097] Note that the quality factor Q can also be characterized as "unperturbed" when necessary to distinguish it from the perturbed quality factor Q (p)Distinguish it and, if necessary, characterize it as "unloaded" to distinguish it from the loaded quality factor Q [l] Similarly, the perturbed quality factor Q (p) can also be characterized as "unloaded" if necessary to distinguish it from the loaded and perturbed quality factor Q (p)[l] Distinguish it.

[0098] Coupled resonators

[0099] Resonators that are coupled through any part of their near fields and have substantially the same resonant frequency can interact and exchange energy. There are various physical pictures and models that can be used to understand, design, optimize, and characterize this energy exchange. One way to describe two coupled resonators and model the energy exchange between them is to use coupled-mode theory (CMT).

[0100] In coupled-mode theory, the resonator fields obey the following system of linear equations:

[0101]

[0102] where the indices denote different resonators, and κ mn is the coupling coefficient between the resonators. For reciprocal systems, the coupling coefficient can obey the relation κ mn = κ nm . Note that for the purposes of this specification, far-field radiation interference effects will be ignored, and thus the coupling coefficient will be considered a real value. Additionally, since the coupling coefficient appears almost always squared in all subsequent calculations of the system performance in this specification, we use κ mn to denote the absolute value of the real coupling coefficient. [[ID=3,3]]

[0103] Note that the coupling coefficient κ mn from the above CMT is related to the so-called coupling factor k between resonators m and n through mn . We define the "strong coupling coefficient" U as the ratio of the coupling and loss rates between resonators m and n. mn

[0104] In a similar manner to the loading of a resonator by a power generating or consuming device connected to it, the quality factor of resonator m in the presence of resonator n or an additional resonator with a similar frequency can be loaded by that resonator n or an additional resonator. The fact that resonator m can be loaded by resonator n and vice versa is simply a different way of looking at how the resonators are coupled.

[0105] The loaded Q of the resonator in these cases can be expressed as Q m[n] and Qn[m] For multiple resonators or loading sources or devices, the total loading of a resonator can be determined by modeling each load as a resistive loss and summing the multiple loads in appropriate parallel and / or series combinations to determine the equivalent load of the ensemble.

[0106] In some discussions herein, we define the "loaded quality factor" or "loaded Q" of resonator m due to resonator n as δQ m where 1 / δQ m[n] ≡1 / Q m[n] - 1 / Q m[n] Note that resonator m also loads resonator n, and its "loaded Q" is given by 1 / δQ m ≡1 / Q n[m] - 1 / Q n[m] n n .

[0107] When one or more resonators are connected to a generator or a load, the linear equations are modified to:

[0108]

[0109] where s +m (t) and S -m (t) are the amplitudes of the fields from the generator into resonator m and from resonator m back towards the generator or into the load, respectively, which are defined such that the power carried by them is given by |s +m (t)| 2 and |s -m (t)| 2 . The loading coefficient κ m is related to the rate of energy exchange between resonator m and the connected generator or load.

[0110] Note that the loading coefficient κ from the above CMT is related to the earlier defined loaded quality factor δQ m through δQ m[l] = ω m / 2κ m . m[l]

[0111] We define the "strong loading factor" U m[l] as the ratio of the loading to the loss rate of resonator m, U m[l] = κ m / Γ m = Q m / δQ m[l] .

[0112] Figure 1 ​​​(a) in [the figure] shows an example of two coupled resonators 1000, namely a first resonator 102S configured as a source resonator and a second resonator 102D configured as a device resonator. Energy can be transferred over the distance D between the resonators. The source resonator 102S can be driven by a power supply or a generator (not shown). Work can be extracted from the device resonator 102D by a power-consuming device or a load (e.g., a load resistor, not shown). Let us use the subscript “s” to denote the source, “d” to denote the device, “g” to denote the generator, and “l” to denote the load, and since there are only two resonators in this example and κ sd =K ds , let us drop the subscripts on κ sd , k sd and U sd and denote them as κ, k, and U respectively.

[0113] The generator can constantly drive the source resonator at a constant drive frequency f corresponding to the angular drive frequency ω, where ω = 2πf.

[0114] In this case, the efficiency η of power transfer from the generator to the load (via the source and device resonators) is η = |s -d | 2 / |s +s | 2 and is maximized under the following conditions: The source resonance frequency, the device resonance frequency, and the generator drive frequency must be matched, i.e.,

[0115] ω s = ω d = ω.

[0116] In addition, the loading QδQ of the source resonator due to the generator must be matched (equal) to the loaded QQ of the source resonator due to the device resonator and the load, and conversely, the loading QδQ of the device resonator due to the load must be matched (equal) to the loaded QQ of the device resonator due to the source resonator and the generator, i.e., s[g] s[dl] d[l] d[sg] s[g] s[dl] d[l] d[sg]

[0117] δQ s[g] = Q s[dl] and δQ d[l] = Q d[sg] .

[0118] These equations determine the optimal loading rates of the source resonator by the generator and of the device resonator by the load as

[0119]

[0120] Note that the above frequency matching and Q matching conditions are together referred to as "impedance matching" in electrical engineering.

[0121] Under the above conditions, maximizing efficiency is a monotonically increasing function of only the strong coupling factor between the source and the device resonator, and is given by as Figure 5 shown. Note that the coupling efficiency η is greater than 1% when U is greater than 0.2, greater than 10% when U is greater than 0.7, greater than 17% when U is greater than 1, greater than 52% when U is greater than 3, greater than 80% when U is greater than 9, greater than 90% when U is greater than 19, and greater than 95% when U is greater than 45. In some applications, the operating region where U > 1 can be referred to as the "strong coupling" region.

[0122] Since in some cases a large is desired, resonators with high Q can be used. The Q of each resonator can be high. The geometric mean of the Qs of the resonators can also be or alternatively be high.

[0123] The coupling factor k is a number between 0 ≤ k ≤ 1, and it can be independent of (or nearly independent of) the resonant frequencies of the source and the device resonator. Specifically, it can be mainly determined by their relative geometric structures and the physical decay law of the field that mediates their coupling. In contrast, the coupling coefficient can be a strong function of the resonant frequency. The resonant frequency of the resonator can preferably be selected to achieve high Q rather than to achieve low Γ, because these two goals are achievable at two separate resonant frequency regions.

[0124] A high-Q resonator can be defined as a resonator with Q > 100. Two coupled resonators can be called a system of high-Q resonators when each of them has a Q greater than 100, Q s >100 and Q d >100. In other embodiments, when the geometric mean of the Qs of the resonators is greater than 100, two coupled resonators can be called a system of high-Q resonators.

[0125] The resonators can be named or numbered. They can be called the source resonator, the device resonator, the first resonator, the second resonator, the repeater resonator, etc. It should be understood that although in Figure 1Two resonators are shown, but in many of the examples below, other embodiments may include three (3) or more resonators. For example, a single source resonator 102S can transfer energy to multiple device resonators 102D or multiple devices. Energy can be transferred from a first device to a second, then from the second device to a third, and so on. Multiple sources can transfer energy to a single device or multiple devices connected to a single device resonator or multiple devices connected to multiple device resonators. The resonator 102 can alternately or simultaneously act as a source, a device, or it can be used to relay power from a source at one location to a device at another location. Intermediate electromagnetic resonators 102 can be used to extend the distance range of a wireless energy transfer system. Multiple resonators 102 can be daisy chained together to exchange energy with a wide range of sources and devices over an extended distance. High power levels can be split between multiple sources 102S, transferred to multiple devices, and recombined at a remote location.

[0126] The analysis of a single source and a single device resonator can be extended to multiple source resonators and / or multiple device resonators and / or multiple intermediate resonators. In such an analysis, the conclusion can be a large strong coupling factor U between at least some or all of the multiple resonators mn which is preferred for high system efficiency in wireless energy transfer. Again, embodiments can use source, device, and intermediate resonators with high Q. The Q of each resonator can be high. The geometric mean of the Qs for resonator pairs m and n (for which a large U mn ) is desired can also or alternatively be high.

[0127] Note that since the strong coupling factor between two resonators can be determined by the relative magnitudes of the coupling mechanism between the two resonators and the loss mechanism of each resonator, the strength of any or all of these mechanisms can be perturbed in the presence of foreign objects near the resonators as described above.

[0128] Continuing the convention for labeling from the previous section, we will describe k as the coupling factor in the absence of foreign objects or materials. We will denote the coupling factor p in the presence of foreign objects as k (p) and refer to it as the "perturbed coupling factor" or "perturbed k". Note that when necessary, the coupling factor k can also be characterized as "unperturbed" to distinguish it from the perturbed coupling factor k (p) distinguish.

[0129] We define δk (p) ≡k (p) -k, and we refer to it as the "perturbation of the coupling factor" or "perturbation of k" due to the foreign object p.

[0130] We also define β (p) ≡ k (p) / k, and we refer to it as the "coupling factor insensitivity" or "k insensitivity". Subscripts such as β 12(p) indicate the resonators involved in the perturbed and unperturbed coupling factors, i.e., β 12(p) ≡ k 12(p) / k 12 .

[0131] Similarly, we describe U as the strong coupling factor in the absence of extraneous objects. We denote the strong coupling factor p in the presence of extraneous objects as U (p) , and we refer to it as the "perturbed strong coupling factor" or "perturbed U". Note that the strong coupling factor U can also be characterized as "unperturbed" when necessary, to distinguish it from the perturbed strong coupling factor U (p) . Note that the strong coupling factor U can also be characterized as "unperturbed" when necessary, to distinguish it from the perturbed strong coupling factor U (p) .

[0132] We define δU (p) ≡ U (p) - U and refer to it as the "perturbation of the strong coupling factor" or "perturbation of U" due to the extraneous object p.

[0133] We also define Ξ(p) ≡ U(p) / U and refer to it as the "strong coupling factor insensitivity" or "U insensitivity". Subscripts such as Ξ 12(p) indicate the resonators involved in the perturbed and unperturbed coupling factors, i.e., Ξ 12(p) ≡ U 12(p) / U 12 .

[0134] The efficiency of the energy exchange in the perturbed system can be given by the same formula that gives the efficiency of the unperturbed system, where all parameters such as the strong coupling factor, coupling factor, and quality factor are replaced by their perturbed equivalent parameters. For example, in a wireless energy transfer system including a source and a device resonator, the optimal efficiency can be calculated as Thus, in a wireless energy exchange system perturbed by extraneous objects, for high system efficiency in wireless energy transfer, it may be desirable to have a large perturbed strong coupling factor U mn(p) between at least some or all of the multiple resonators. The source, device, and / or intermediate resonators can have a high Q (p) .

[0135] Some spurious perturbations are sometimes detrimental to the perturbed strong coupling factor (via large perturbations of the coupling factor or quality factor). Thus, techniques can be used to reduce the impact of spurious perturbations on the system and maintain insensitivity to large strong coupling factors.

[0136] Energy exchange efficiency

[0137] The so-called "useful energy" in a useful energy transfer is the energy or power that must be transferred to one or more devices to power or charge the devices. The transfer efficiency corresponding to the useful energy transfer can be system- or application-dependent. For example, a high-power vehicle charging application that transfers several kilowatts of power may need to have an efficiency of at least 80% to supply a useful amount of power, resulting in a useful energy transfer sufficient to recharge the vehicle battery without significantly heating the various components of the transfer system. In some consumer electronics applications, the useful energy transfer can include any energy transfer efficiency greater than 10%, or any other acceptable amount that keeps a rechargeable battery "topped off" and operating for a long time. For some wireless sensor applications, a transfer efficiency much less than 1% may be suitable for powering multiple low-power sensors from a single source located at a significant distance from the sensors. For other applications where wired power transfer is not possible or impractical, a wide range of transfer efficiencies can be acceptable for useful energy transfer and can be considered to supply useful power to the devices in those applications. Generally, the operating distance is any distance at which useful power transfer is maintained or can be maintained according to the principles described herein.

[0138] Useful energy transfer for wireless energy transfer in powering or recharging applications can be efficient, highly efficient, or sufficiently efficient as long as the wasted energy level, heat dissipation, and associated field strength are within admissible limits. The admissible limits can depend on the application, environment, and system location. Useful energy transfer for wireless energy transfer in powering or recharging applications can be efficient, highly efficient, or sufficiently efficient as long as the desired system performance can be achieved for reasonable cost limitations, weight limitations, size limitations, etc. Efficient energy transfer can be determined relative to what can be achieved using traditional inductive techniques with non-high-Q systems. Then, if more energy is delivered than can be delivered by a coil structure of similar size within a similar distance or alignment offset in a traditional inductive scheme, the energy transfer can be defined as efficient, highly efficient, or sufficiently efficient.

[0139] Note that even though certain frequency and Q matching conditions can optimize the system efficiency of energy transfer, it is not necessary to fully satisfy these conditions to have a sufficiently efficient energy transfer for useful energy transfer. As long as the relative offset of the resonant frequency is less than 1 / Qm(p) ,1 / Q n(p) and k mn(p) Among the approximate maximum values, efficient energy exchange can be achieved. For efficient energy exchange, the Q matching condition can be less strict than the frequency matching condition. The strong loading factor U of the resonator caused by the generator and / or load m[l] can deviate from its optimal value but still have a sufficient degree of efficient energy exchange depending on the specific system, whether all or some of the generators and / or loads are Q-mismatched, etc.

[0140] Therefore, the resonant frequency of the resonator may not be perfectly matched, but is matched within the above tolerances. The strong loading factors of at least some of the resonators caused by the generator and / or load may not be perfectly matched to their optimal values. The voltage level, current level, impedance value, material parameters, etc. may not be at the exact values described in this disclosure, but will be within some acceptable tolerances of those values. In addition to considerations such as efficiency, Q, frequency, strong coupling factor, etc., system optimization can include considerations such as cost, size, weight, complexity, etc. Some system performance parameters, specifications, and designs may be far from optimal in order to optimize other system performance parameters, specifications, and designs.

[0141] In some applications, at least some of the system parameters can change over time, for example because components such as sources or devices may move or age, or because the load may be variable, or because disturbances or environmental conditions are changing, etc. In these cases, in order to achieve acceptable matching conditions, at least some of the system parameters may need to be dynamically adjusted or tuned. All system parameters can be dynamically adjusted or tuned to achieve approximate optimal operating conditions. However, based on the above discussion, even if some of the system parameters are not variable, sufficient efficient energy exchange can be achieved. In some examples, at least some of the devices may not be dynamically adjusted. In some examples, at least some of the sources may not be dynamically adjusted. In some examples, at least some of the intermediate resonators may not be dynamically adjusted. In some examples, none of the system parameters can be dynamically adjusted.

[0142] Electromagnetic resonator

[0143] The resonator used to exchange energy can be an electromagnetic resonator. In such resonators, the inherent energy decay rate Γ is given by the absorption (or resistance) loss and radiation loss of the resonator m .

[0144] The resonator can be constructed such that the energy stored by the electric field is mainly confined within the structure and the energy stored by the magnetic field is mainly in the region around the resonator. Then, the energy exchange is mainly mediated by the resonant magnetic near field. These types of resonators can be called magnetic resonators.

[0145] The resonator can be constructed such that the energy stored by the magnetic field is mainly confined within the structure and the energy stored by the electric field is mainly in the region around the resonator. Then the energy exchange is mainly mediated by the resonant electric near-field. These types of resonators can be referred to as electric resonators.

[0146] Note that the total electric and magnetic energy stored by the resonator must be equal, but their localization can be quite different. In some cases, the ratio of the average electric field energy to the average magnetic field energy specified at a certain distance from the resonator can be used to characterize or describe the resonator.

[0147] The electromagnetic resonator can include inductive elements, distributed inductance, or a combination of inductances with an inductance L, and capacitive elements, distributed capacitance, or a combination of capacitances with a capacitance C. In Figure 6 The minimum circuit model of the electromagnetic resonator 102 is shown in (a) of. The resonator can include an inductive element 108 and a capacitive element 104. Given an initial energy such as the electric field energy stored in the capacitor 104, the system will oscillate as the capacitor discharges, transferring the energy to the magnetic field energy stored in the inductor 108, which in turn transfers the energy back to the electric field energy stored in the capacitor 104.

[0148] The Figure 6 The resonators 102 shown in (b), (c), and (d) of can be referred to as magnetic resonators. For wireless energy transfer applications in a living environment, magnetic resonators may be preferred because most everyday materials including animals, plants, and humans are non-magnetic (i.e., μ r ≈1), so their interaction with the magnetic field is minimal and is mainly due to eddy currents induced by the time-varying magnetic field (which is a second-order effect). This property is important for safety reasons and because it reduces the possibility of interaction with unrelated environmental objects and materials that may change the system performance.

[0149] Figure 6 (d) of shows a simplified diagram of some of the electric and magnetic field lines associated with the exemplary magnetic resonator 102B. The magnetic resonator 102B can include a conductor loop that acts as an inductive element 108 and a capacitive element 104 (at the ends of the conductor loop). Note that this figure depicts most of the energy stored in the region around the resonator in the magnetic field and most of the energy stored in the resonator (between the capacitor plates) in the electric field. Some electric fields due to fringe fields, free charges, and time-varying magnetic fields can be stored in the region around the resonator, but the magnetic resonator can be designed to confine the electric field as close as possible to the resonator or within the resonator itself.

[0150] The inductor 108 and capacitor 104 of the electromagnetic resonator 102 can be bulk circuit elements, or the inductance and capacitance can be distributed and can be generated by the way of forming, shaping, and positioning conductors in the structure. For example, as Figure 6 shown in (b), (c), and (d) of [reference], the inductor 108 can be realized by shaping the conductor into a closed surface area. Such resonators 102 can be referred to as capacitively loaded loop inductors. Note that we can use the terms "loop" or "coil" to generally refer to a conductive structure (wire, tube, strip, etc.) that encloses a surface of any shape and size with any number of turns. In Figure 6 the (b) of [reference], the enclosed surface area is circular, but the surface can be any of a variety of other shapes and sizes and can be designed to achieve certain system performance specifications. As an example indicating how the inductance scales with physical dimensions, the inductance of a section of circular conductor arranged to form a circular single-turn loop is approximately

[0151]

[0152] where μ0 is the magnetic permeability of free space, x is the radius of the enclosed circular surface area, and a is the radius of the conductor used to form the inductor loop. More accurate inductance values of the loop can be calculated analytically or numerically.

[0153] The inductance of other cross-sectional conductors arranged to form other closed surface shapes, areas, sizes, etc. with any number of turns can be calculated analytically or numerically, or the inductance can be determined by measurement. Inductance can be realized using inductor elements, distributed inductance, networks, arrays, series and parallel combinations of inductors and inductance, etc. The inductance can be fixed or variable and can be used to change the impedance matching and resonant frequency operating conditions.

[0154] There are various ways to achieve the capacitance required to reach the desired resonant frequency of the resonator structure. The capacitor plates 110 can be formed and utilized as shown in (b) of [reference], or as shown in (c) of [reference], the capacitance can be distributed and realized between adjacent windings of the multi-loop conductor 114. Capacitance can be realized using capacitor elements, distributed capacitance, networks, arrays, series and parallel combinations of capacitance, etc. The capacitance can be fixed or variable and can be used to change the impedance matching and resonant frequency operating conditions. Figure 6 shown in (b) of [reference], or as shown in (c) of [reference], the capacitance can be distributed and realized between adjacent windings of the multi-loop conductor 114. Capacitance can be realized using capacitor elements, distributed capacitance, networks, arrays, series and parallel combinations of capacitance, etc. The capacitance can be fixed or variable and can be used to change the impedance matching and resonant frequency operating conditions. Figure 6 shown in (c) of [reference], the capacitance can be distributed and realized between adjacent windings of the multi-loop conductor 114. Capacitance can be realized using capacitor elements, distributed capacitance, networks, arrays, series and parallel combinations of capacitance, etc. The capacitance can be fixed or variable and can be used to change the impedance matching and resonant frequency operating conditions.

[0155] It should be understood that the inductance and capacitance in the electromagnetic resonator 102 can be lumped, distributed, or a combination of lumped and distributed inductance and capacitance, and the electromagnetic resonator can be realized by a combination of various elements, techniques, and effects described herein.

[0156] The electromagnetic resonator 102 may include inductors, inductance, capacitors, capacitance, and additional circuit elements such as resistors, diodes, switches, amplifiers, diodes, transistors, transformers, conductors, connectors, etc.

[0157] Resonant frequency of the electromagnetic resonator

[0158] The electromagnetic resonator 102 may have characteristics, natural or resonant frequencies determined by its physical properties. This resonant frequency is the frequency at which the energy stored in the resonator oscillates between the energy W E (W E = q 2 / 2C, where q is the charge on the capacitor C) and the energy W B (W B = Li 2 / 2, where i is the current through the inductor L) stored in the magnetic field of the resonator. In the absence of any losses in the system, energy will continuously be exchanged between the electric field in the capacitor 104 and the magnetic field in the inductor 108. The frequency at which this energy is exchanged can be referred to as the characteristic frequency, natural frequency, or resonant frequency of the resonator, and is given by ω,

[0159]

[0160] The resonant frequency of the resonator can be changed by tuning the inductance L and / or capacitance C of the resonator. The resonant frequency can be designed to operate at so-called ISM (Industrial, Scientific, and Medical) frequencies specified by the FCC. The resonator frequency can be selected to meet certain field limit specifications, specific absorption rate (SAR) limit specifications, electromagnetic compatibility (EMC) specifications, electromagnetic interference (EMI) specifications, component size, cost, or performance specifications, etc.

[0161] Quality factor of the electromagnetic resonator

[0162] Figure 6 The energy in the resonator 102 shown can decay or be lost due to inherent losses including absorptive losses (also known as ohmic or resistive losses) and / or radiative losses. The quality factor or Q of the resonator that characterizes the energy decay is inversely proportional to these losses. Absorptive losses can be caused by the finite conductivity of the conductors used to form the inductor and losses in other elements, components, connectors, etc. in the resonator. An inductor formed of a low-loss material can be referred to as a "high-Q inductive element", and elements, components, connectors, etc. with low losses can be said to have "high-resistance Q". Generally, the total absorptive losses of the resonator can be calculated as the appropriate series and / or parallel combination of the resistive losses of the various elements and components that make up the resonator. That is, in the absence of any significant radiative or component / connection losses, it can be given by Q absto give the Q of the resonator,

[0163]

[0164] where ω is the resonance frequency, L is the total inductance of the resonator, and, for example, the resistance of the conductor used to form the inductor can be given by R abs = lρ / A (l is the length of the wire, ρ is the resistivity of the conductor material, and A is the cross-sectional area through which the current flows in the wire). For alternating current, the cross-sectional area through which the current flows can be less than the physical cross-sectional area of the conductor due to the skin effect. Thus, a high-Q magnetic resonator can be composed of a conductor with high conductivity, a relatively large surface area, and / or a specifically designed profile (such as Litz wire) to minimize the proximity effect and reduce the AC resistance.

[0165] The magnetic resonator structure can include a high-Q inductive element composed of a high-conductivity wire, coated wire, Litz wire, tape, strip, or plate, tube, coating, gel, trace, etc. The magnetic resonator can be self-resonant, or it can include external coupling elements such as capacitors, inductors, switches, diodes, transistors, transformers, etc. The magnetic resonator can include distributed and lumped capacitances and inductances. Generally, the Q of the resonator is determined by the Q of all the individual components of the resonator.

[0166] Since Q is proportional to the inductance L, the resonator can be designed to increase L within certain other constraints. For example, one way to increase L is to use more than one turn of the conductor to form the inductor in the resonator. The design techniques and trade-offs can depend on the application, and a variety of structures, conductors, components, and resonance frequencies can be selected in the design of a high-Q magnetic resonator.

[0167] In the absence of significant absorption losses, the Q of the resonator can be mainly determined by radiation losses, and is given by Q rad = ωL / R rad where RE rad is the radiation loss of the resonator and can depend on the size of the resonator relative to the operating frequency ω or wavelength λ. For the magnetic resonator discussed above, the radiation loss can scale with R rad ~(x / λ) 4 (characteristic of magnetic dipole radiation), where x is a characteristic dimension of the resonator, such as Figure 6 the radius of the inductive element as shown in (b) therein, and here λ = c / f, where c is the speed of light and f is as defined above. The size of the magnetic resonator can be much smaller than the operating wavelength, so the radiation loss can be very small. Such structures can be called sub-wavelength resonators. Radiation can be a loss mechanism for non-radiative wireless energy transfer systems, and the design can be chosen to reduce R rador minimize it. Note that for non-radiative wireless energy transfer schemes, high Q rad may be desirable.

[0168] Also note that the design of resonators for non-radiative wireless energy transfer is different from that of antennas designed for communication or far-field energy transfer purposes. Specifically, a capacitively loaded conductive loop can be used as a resonant antenna (e.g., in a cellular phone), but for those operating in the far-field region, where the radiative Q is deliberately designed to be small so that the antenna is efficient at radiating energy. Such designs are not suitable for the efficient near-field wireless energy transfer technology disclosed in this application.

[0169] The quality factor of a resonator that includes both radiative and absorption losses is Q = ωL / (R abs +R rad ). Note that there may be a maximum Q value for a particular resonator, and the resonator can be designed with special considerations for the size of the resonator, the materials and components used to construct the resonator, the operating frequency, the connection mechanism, etc., in order to achieve a high-Q resonator. Figure 7 A plot showing Q of an exemplary magnetic resonator that can be used for wireless power transfer at MHz frequencies (in this case, a coil with a diameter of 60 cm made of a copper tube with an outer diameter (OD) of 4 cm). The absorptive Q (dashed line) 702 increases with frequency, while the radiative Q (dotted line) 704 decreases with frequency, thus causing the total Q to peak at 708 at a particular frequency. Note that the Q of this exemplary resonator is greater than 100 over a large frequency range. The magnetic resonator can be designed to have a high Q over a certain frequency range, and the system operating frequency can be set to any frequency within that range.

[0170] When describing a resonator in terms of the loss rate, as previously mentioned, the intrinsic decay rate 2Γ can be used to define Q. The intrinsic decay rate is the rate at which an uncoupled and undriven resonator loses energy. For the above magnetic resonator, the intrinsic loss can be given by Γ = (R abs +R rad ) / 2L, and the quality factor Q of the resonator is given by Q = ω / 2Γ.

[0171] Note that the quality factor related only to a specific loss mechanism can be denoted as Q mechanism (if the resonator is not specified) or Q 1,mechanism (if the resonator is specified (e.g., resonator 1)). For example, Q 1，rad is the quality factor for resonator 1 and is related to its radiative losses.

[0172] Near field of the electromagnetic resonator

[0173] The high-Q electromagnetic resonators used in the near-field wireless energy transfer systems disclosed herein can be sub-wavelength objects. That is, the physical size of the resonator can be much smaller than the wavelength corresponding to the resonance frequency. The sub-wavelength magnetic resonator can store most of the energy in the region around the resonator in its magnetic near-field, and can also describe these fields as stationary or non-propagating, because it does not radiate far from the resonator. Generally, the range of the near-field in the region around the resonator is set by the wavelength. Therefore, for a sub-wavelength resonator, the range of the near-field can extend far beyond the resonator itself. The limiting surface where the field characteristics change from near-field characteristics to far-field characteristics can be called the "radiation caustic".

[0174] When moving further away from the resonator, the intensity of the near-field decreases. Although the field intensity of the resonator near-field decays as it moves away from the resonator, the field can still interact with objects entering the general vicinity of the resonator. The degree of interaction of the field depends on various factors, some of which can be controlled and designed, while some cannot. When the distance between the coupled resonators is such that one resonator is within the radiation caustic surface of the other, the wireless energy transfer scheme described herein can be achieved.

[0175] The near-field distribution of the electromagnetic resonator can be similar to those field distributions generally associated with dipole resonators or oscillators. Such field distributions can be described as omnidirectional, meaning that the amplitude of the field is non-zero in all directions away from the object.

[0176] Characteristic dimension of the electromagnetic resonator

[0177] Magnetic resonators with sufficient Q and spatial separation and / or offset can achieve efficient wireless energy transfer at much larger distances than seen in the prior art, even if the size and shape of the resonator structures are different. Such resonators can also be operated to achieve more efficient energy transfer at shorter range distances than achievable by the aforementioned techniques. We describe such resonators as being capable of achieving mid-range energy transfer.

[0178] The mid-range distance can be defined as a distance larger than the characteristic size of the smallest one of the resonators involved in the transfer, where the distance is measured from the center of one resonator structure to the center of the second spatially separated resonator structure. In this definition, two-dimensional resonators are spatially separated when the regions restricted by their inductive elements do not intersect, and three-dimensional resonators are spatially separated when their volumes do not intersect. When the region restricted by a two-dimensional resonator is outside the volume of a three-dimensional resonator, the former is spatially separated from the latter.

[0179] Figure 8Show certain exemplary resonators with their characteristic dimensions marked. It should be understood that the characteristic dimensions 802 of the resonator 102 can be defined in terms of the dimensions of the conductors and the area bounded or enclosed by the inductive elements in the magnetic resonator and the length of the conductors forming the capacitive elements of the electric resonator. Then, the characteristic dimension 802x of the resonator 102 char can be equal to the radius of the smallest sphere that can fit around the inductive or capacitive elements of the magnetic or electric resonator respectively, and the center of the resonator structure is the center of that sphere. The characteristic thickness 804t of the resonator 102 char can be the minimum possible height of the highest point of the inductive or capacitive elements of the magnetic or capacitive resonator measured from the flat surface on which it lies. The characteristic width 808w of the resonator 102 char can be the radius of the smallest possible circle through which the inductive or capacitive elements of the magnetic or electric resonator can pass while traveling along a straight line. For example, the characteristic width 808 of a cylindrical resonator can be the radius of the cylinder.

[0180] In the wireless energy transfer technology of the present invention, energy can be efficiently exchanged over a large distance range, but this technology is well-known for its ability to exchange useful energy at mid-range distances and between resonators with different physical sizes, components, and orientations for powering or recharging devices. Note that although k may be small in these cases, strong coupling and efficient energy transfer can be achieved by using high-Q resonators to achieve high U, that is, an increase in Q can be used to at least partially overcome the decrease in k to maintain useful energy transfer efficiency.

[0181] Also note that although the near field of a single resonator can be described as omnidirectional, the efficiency of energy exchange between two resonators can depend on the relative position and orientation of the resonators. That is, the efficiency of energy exchange can be maximized for a specific relative orientation of the resonators. The sensitivity of the transfer efficiency to the relative position and orientation of two uncompensated resonators can be captured in the calculation of k or κ. Although coupling can be achieved between resonators that are offset and / or rotated relative to each other, the efficiency of the exchange can depend on the details of the positioning and any feedback, tuning, and compensation techniques implemented during operation.

[0182] High-Q magnetic resonator

[0183] In the near field region of a subwavelength capacitively loaded loop magnetic resonator (x << λ), the resistance associated with a circular conductive loop inductor composed of N turns of wire (whose radius is greater than the skin depth) is approximately and R rad = π / 6·η o N 2 (ωx / c) 4, where ρ is the resistivity of the conductor material and η o ≈120πΩ is the impedance of free space. The inductance L of such an N-turn loop is approximately N times the inductance of the single-turn loop given previously. 2 The quality factor Q of such a resonator is Q = ωL / (R abs +R rad ) and is highest for a specific frequency determined by the system parameters ( Figure 4 ). As mentioned previously, at lower frequencies, Q is mainly determined by absorption losses, and at higher frequencies, Q is mainly determined by radiation losses.

[0184] Note that the formulas given above are approximate and are intended to illustrate the functional dependencies of R abs , R rad and L on the physical parameters of the structure. For the precise design of the resonator structure, more accurate numerical calculations of these parameters that take into account the deviations from the strict quasistatic limit (e.g., non-uniform current / charge distribution along the conductor) may be useful.

[0185] Note that absorption losses can be minimized by forming the inductive element using a low-loss conductor. For example, absorption losses can be minimized by using a large surface area conductor such as an inductive tube, bar, strip, machined object, plate, etc., by using a specially designed conductor such as Litz wire, braided wire, wire of any cross-section, and other conductors with low proximity losses (in which case the above frequency scaling properties may also be different), and by using a low resistivity material such as high purity copper and silver. An advantage of using a conductive tube as a conductor at higher operating frequencies is that it can be cheaper and lighter than a solid conductor of similar diameter and can have a similar resistance since most of the current travels along the outer surface of the conductor due to the skin effect.

[0186] To obtain a rough estimate of an achievable resonator design made of copper wire or copper tube and suitable for operation in the microwave region, the optimal Q and resonant frequency can be calculated for a resonator consisting of a circular inductive element (N = 1) made of copper wire (ρ = 1.69·10-8 Ωm) with various cross-sections. Then, for an inductive element with characteristic dimension x = 1 cm and conductor diameter a = 1 mm (e.g., suitable for a cellular phone), the quality factor peaks at Q = 1225 when f = 380 MHz. For x = 30 cm and a = 2 mm, dimensions of an inductive element that might be suitable for a laptop computer or a home robot, at f = 17 MHz, Q = 1103. For a larger source inductive element that might be located in a ceiling, for example, x = 1 m and a = 4 mm, at f = 5 MHz, Q can be as high as Q = 1315. Note that many practical examples provide an expected quality factor of Q ≈ 1000 - 1500 at λ / x ≈ 50 - 80. Measurements of more kinds of coil shapes, dimensions, materials, and operating frequencies than those above show that using generally available materials, Q > 100 can be achieved for a variety of magnetic resonator structures.

[0187] As described above, the rate of energy transfer between two resonators with characteristic dimensions x1 and x2 and separated by a distance D between their centers can be given by κ. To give an example of how the defining parameters scale, at three (3) distances, consider the cellular phone, laptop computer, and ceiling resonator examples from above; D / x = 10, 8, 6. In the examples considered here, the source and device resonators are the same size (x1 = x2) and shape, and are oriented as shown in (b) of Figure 1 In the cellular phone example, respectively, ω / 2κ = 3033, 1553, 655. In the laptop computer example, respectively, ω / 2κ = 7131, 3651, 1540, and for the ceiling resonator example, ω / 2κ = 6481, 3318, 1400. The corresponding coupling loss ratios peak at the frequencies where the inductive element Q peaks, and for the above three inductive element sizes and distances, κ / Г = 0.4, 0.79, 1.97 and 0.15, 0.3, 0.72 and 0.2, 0.4, 0.94. An example using inductive elements of different sizes is an x1 = 1 m inductor (e.g., a source in the ceiling) and an x2 = 30 cm inductor (e.g., a home robot on the floor) separated by a distance D = 3 m (e.g., the height of a room). In this example, for an efficiency of approximately 14%, at the optimal operating frequency of f = 6.4 MHz, the strong coupling figure of merit, Here, the optimal system operating frequency is between the peaks of the individual resonator Qs.

[0188] Inductive elements can be formed for use in high-Q magnetic resonators. We have demonstrated various high-Q magnetic resonators based on copper conductors of inductive elements formed as closed surfaces. Inductive elements can be formed using various conductors arranged in various shapes (enclosing regions of any size or shape), and they can be single-turn or multi-turn elements. In Figure 9 FIG. shows exemplary inductive elements 900A-B. The inductive elements can be formed as closed circles, rectangles, squares, triangles, shapes with rounded corners, shapes following the contours of a particular structure and device, shapes following, filling, or taking advantage of dedicated spaces within a structure or device, etc. The design can be optimized for size, cost, weight, appearance, performance, etc.

[0189] These conductors can be bent or formed into desired sizes, shapes, and number of turns. However, it may be difficult to accurately reproduce the conductor shape and size using manual techniques. Additionally, it may be difficult to maintain a uniform or desired center spacing between conductor segments in adjacent turns of the inductive element. For example, accurate or uniform spacing may be important in determining the self-capacitance of the structure and the increase in AC resistance induced by any proximity effects.

[0190] Molds can be used to replicate inductor elements for high-Q resonator designs. Additionally, molds can be used to accurately form conductors into any kind of shape without creating kinks, buckles, or other potentially harmful effects in the conductors. Molds can be used to form inductor elements, and then the inductor elements can be removed from these forms. Once removed, these inductive elements can be incorporated into a housing or device that can accommodate a high-Q magnetic resonator. The formed elements can also or alternatively be retained in the molds used to form them.

[0191] Standard CNC (Computer Numerical Control) routing or milling tools or any other known techniques for cutting or forming grooves in blocks can be used to form the molds. Machining techniques, injection molding techniques, casting techniques, pouring techniques, vacuum techniques, thermoforming techniques, in-situ cutting techniques, compression molding techniques, etc. can also or alternatively be used to form the molds.

[0192] The formed elements can be removed from the mold, or they can be retained in the mold. The mold can be modified with internal inductive elements. The mold can be covered, machined, attached, painted, etc. The mold and conductor combination can be integrated into another housing, structure, or device. The grooves cut into the mold can be of any size and can be designed to form conductive tubes, wires, strips, bands, blocks, etc. into the desired inductor shape and size.

[0193] The inductive element used in a magnetic resonator can include more than one loop and can combine spirals inwardly or outwardly or upwardly or downwardly or along certain directions. Generally, the magnetic resonator can have various shapes, sizes, and number of turns, and it can be composed of various conductive materials.

[0194] The magnetic resonator can be freestanding, or it can be enclosed in a housing, container, sleeve, or casing. The magnetic resonator can include a template for fabricating the inductive element. These different templates and housings can be composed of almost any kind of material. For certain applications, low-loss materials such as Teflon, REXOLITE, styrene, etc. may be preferred. These housings can include fixtures for holding the inductive element.

[0195] The magnetic resonator can be composed of a self-resonant coil of copper wire or copper tube. The magnetic resonator composed of a self-resonant wire coil can include a wire of length l and a cross-section of radius a, wound into a helical coil of radius x, height h, and number of turns N, which can be characterized, for example, as

[0196] The magnetic resonator structure can be configured such that x is about 30 cm, h is about 20 cm, a is about 3 mm, and N is about 5.25, and during operation, the power source coupled to the magnetic resonator can drive the resonator at a resonant frequency f, where f is about 10.6 MHz. In the case where x is about 30 cm, h is about 20 cm, a is about 1 cm, and N is about 4, the resonator can be driven at a frequency f, where f is about 13.4 MHz. In the case where x is about 10 cm, h is about 3 cm, a is about 2 cm, and N is about 6, the resonator can be driven at a frequency f, where f is about 21.4 MHz.

[0197] Printed circuit board traces can be used to design high-Q inductive elements. Compared with inductive elements formed mechanically, printed circuit board traces can have various advantages, including that they can be accurately reproduced and easily integrated using established printed circuit board manufacturing techniques, custom-designed conductor traces can be used to reduce their AC resistance, and the cost of mass-producing them can be significantly reduced.

[0198] High-Q inductive elements can be fabricated using standard PCB techniques on any PCB material such as FR-4 (epoxy E-glass), multifunctional epoxy, high-performance epoxy, bismaleimide triazine resin / epoxy, polyimide, cyanate ester, polytetrafluoroethylene (Teflon), FR-2, FR-3, CEM-1, CEM-2, Rogers, Resolute, etc. Conductor traces can be formed on printed circuit board materials with a lower loss tangent.

[0199] The conductive traces can be composed of copper, silver, gold, aluminum, nickel, etc., and they can be composed of paint, ink, or other curable materials. The circuit board can be flexible, and it can be a flexible circuit. The conductive traces can be formed by chemical deposition, etching, lithography, spray deposition, cutting, etc. The conductive traces can be applied to form a desired pattern, and they can be formed using crystal and structure growth techniques.

[0200] The dimensions of the conductive traces, the number of layers containing the conductive traces, the positions, sizes, and shapes of those traces, and the architecture for interconnecting them can be designed to achieve certain system specifications or to optimize them, such as resonator Q, Q (p) , resonator size, resonator material, and manufacturing cost, U, U (p) etc.

[0201] As an example, as shown in (a) of Figure 10 , a three-turn high-Q inductor element 1001A is fabricated on a four-layer printed circuit board using a rectangular copper trace pattern. The copper traces are shown in black and the PCB is shown in white. The width and thickness of the copper traces in this example are approximately 1 cm (400 mils) and 43 μm (1.7 mils), respectively. The edge spacing between the turns of the conductive traces on a single layer is approximately 0.75 cm (300 mils), and the thickness of each board layer is approximately 100 μm (4 mils). The pattern shown in (a) of Figure 10 is repeated on each layer of the board, and the conductors are connected in parallel. The external dimensions of the 3-loop structure are approximately 30 cm by 20 cm. The measured inductance of this PCB loop is 5.3 μH. The quality factor Q of the magnetic resonator using this inductor element and a tunable capacitor is 550 at its designed resonance frequency of 6.78 MHz. The resonance frequency can be tuned by changing the inductance and capacitance values in the magnetic resonator.

[0202] As another example, as shown in (b) of Figure 10 , a two-turn inductor 1001B is fabricated on a four-layer printed circuit board using a rectangular copper trace pattern. The copper traces are shown in black and the PCB is shown in white. The width and height of the copper traces in this example are approximately 0.75 cm (300 mils) and 43 μm (1.7 mils), respectively. The edge spacing between the turns of the conductive traces on a single layer is approximately 0.635 cm (250 mils), and the thickness of each board layer is approximately 100 μm (4 mils). The pattern shown in Figure 10the pattern shown in (b) of, and connecting the conductors in parallel. The outer dimensions of the double-loop structure are approximately 7.62 cm by 26.7 cm. The measured inductance of this PCB loop is 1.3 μH. Stacking two boards together with a vertical spacing of approximately 0.635 cm (250 mils) and connecting the two boards in series produces a PCB inductor with an inductance of approximately 3.4 μH. The magnetic resonator using this stacked inductor loop and a tunable capacitor has a quality factor Q of 390 at its designed resonance frequency of 6.78 MHz. The resonance frequency can be tuned by changing the inductance and capacitance values in the magnetic resonator.

[0203] Inductive elements can be formed using magnetic materials of any size, shape, thickness, etc. and materials with a wide range of permeability and loss values. These magnetic materials can be solid blocks, which can enclose a hollow volume, which can be formed by many small pieces of magnetic material tiled or stacked together, and can be integrated with conductive sheets or enclosures made of highly conductive materials. Wires can be wound around the magnetic materials to generate a magnetic near field. These wires can be wound around one or more axes of the structure. Multiple wires can be wound around the magnetic material and combined in parallel or in series or via switches to form a custom near-field pattern.

[0204] The magnetic resonator can include 15 turns of Litz wire wound around a 19.2 cm × 10 cm × 5 mm tiled block of 3F3 ferrite material. The Litz wire can be wound around the ferrite material in any direction or combination of directions to achieve the desired resonator performance. The number of turns of the wire, the spacing between the turns, the type of wire, the size and shape of the magnetic material, and the type of magnetic material are all design parameters that can be changed or optimized for different application scenarios.

[0205] High-Q magnetic resonator using a magnetic material structure

[0206] Assembled magnetic materials can be used to form an open magnetic circuit (although one with a void of approximately the size of the entire structure) to achieve a magnetic resonator structure. In these structures, a highly conductive material is wound around the structure made of magnetic material to form the inductive element of the magnetic resonator. A capacitive element can be connected to the highly conductive material, and then the resonance frequency can be determined as described above. These magnetic resonators have their dipole moments in the plane of the two-dimensional resonator structure (rather than perpendicular to it as in the case of a capacitively loaded inductor loop resonator).

[0207] In Figure 11The figure in (a) shows a single planar resonator structure. The planar resonator structure consists of a core of magnetic material 1121, such as a ferrite having one or more loops of conductive material 1122 wound around the core 1121. This structure can be used as a source resonator for transferring power and a device resonator for capturing energy. When used as a source, the ends of the conductor can be coupled to a power source. An alternating current flowing through the conductor loops excites an alternating magnetic field. When this structure is used to receive power, the ends of the conductor can be coupled to a power-consuming device or load. The changing magnetic field induces an electromotive force in one or more loops of the conductor wound around the core magnetic material. The dipole moment of these types of structures is in the plane of the structure and is oriented, for example, along the Y-axis of the structure in (a) as shown. When substantially placed in the same plane (i.e., the X, Y plane of Figure 11 ), two such structures have strong coupling. Figure 11 The structure in (a) as shown has an optimal orientation when the resonators are aligned in the same plane along their Y-axis. Figure 11 When substantially placed in the same plane (i.e., the X, Y plane of Figure 11 ), two such structures have strong coupling. Figure 11 The structure in (a) as shown has an optimal orientation when the resonators are aligned in the same plane along their Y-axis.

[0208] For some applications, the geometry and coupling orientation of the planar resonator may be preferred. A planar or flat resonator shape may be more easily integrated into many electronic devices that are relatively flat and planar. The planar resonator can be integrated into the entire back or side of the device without requiring a change in the geometry of the device. Due to the flat shape of many devices, the natural position of the device when placed on a plane is lying flat on its largest dimension parallel to the surface on which it is placed. The planar resonator integrated into the flat device is naturally parallel to the plane of the surface and is in a suitable coupling orientation relative to the resonators of other devices or planar resonator sources placed on the flat surface.

[0209] As described, the geometry of the planar resonator can allow for easier integration into the device. Its low profile can allow the resonator to be integrated into the entire side of the device or as part of its entire side. When the entire side of the device is covered by the resonator, the magnetic flux can flow through the resonator core without being impeded by lossy materials that can be part of the device or the device circuitry.

[0210] The core of the planar resonator structure can have various shapes and thicknesses, and it can be flat or planar such that the minimum dimension does not exceed 30% of the maximum dimension of the structure. The core can have a complex geometry and can have notches, indentations, ridges, etc. Geometric enhancements can be used to reduce the coupling dependence on orientation, and they can be used to facilitate integration into devices, packages, enclosures, housings, covers, skins, etc. In Figure 11Two exemplary variations of the core geometry are shown in (b) of []. For example, the planar core 1131 can be shaped such that the ends are much wider than the middle of the structure to create a notch for conductor windings. The core material can have a varying thickness, where the ends are thicker and wider than the middle. The core material 1132 can have any number of notches or cuts 1133 of various depths, widths, and shapes to accommodate conductor loops, housings, packages, etc.

[0211] The shape and size of the core can also be dictated by the size and characteristics of the devices integrated into it. The core material can be bent to follow the contour of the device, or asymmetric notches or cuts may be required to allow clearance for various parts of the device. The core structure can be a monolithic piece of magnetic material, or it can be composed of multiple tiles, blocks, or sheets arranged together to form a larger structure. The different layers, tiles, blocks, or sheets of the structure can be of similar materials, or they can be of different materials. It may be desirable to use materials with different permeabilities at different locations in the structure. Core structures with different permeabilities can be useful for guiding magnetic flux, improving coupling, and affecting the shape or extent of the effective area of the system.

[0212] The conductor of the planar resonator structure can be wound around the core at least once. In some cases, it may be preferred to wind at least three turns. The conductor can be any good conductor, including wires, litz wires, conductive tubes, sheets, strips, gels, inks, traces, etc.

[0213] The size, shape, or dimensions of the effective area of the source can also be further enhanced, altered, or modified by using materials that block, shield, or guide magnetic fields. To create an asymmetric effective area around the source, the sides of the source can be covered with a magnetic shield to reduce the intensity of the magnetic field in a specific direction. The shield can be a conductor or a layered combination of a conductor and a magnetic material that can be used to direct the magnetic field away from a specific direction. A structure composed of a conductor and magnetic material layers can be used to reduce the energy losses that may occur due to the shielding of the source.

[0214] The multiple planar resonators can be integrated or combined into a single planar resonator structure. One or more conductors can be wound around the core structure such that the loops formed by two conductors are not coaxial. Examples of such structures are shown in Figure 12 where two conductors 1201, 1202 are wound around a planar rectangular core 1203 at an orthogonal angle. The core can be rectangular, or it can be of various geometries with multiple extensions or protrusions. The protrusions can be useful for winding the conductors, reducing the weight, size, or mass of the core, or they can be used to enhance the directivity or omnidirectionality of the resonator. In Figure 13The multi-wound planar resonator with four protruding parts is shown in the internal structure 1310 in Chinese, where four conductors 1301, 1302, 1303, 1304 are wound around the core. The core may include extension parts 1305, 1306, 1307, 1308 having one or more conductor loops. A single conductor can be wound around the core to form non-coaxial loops. For example, four conductor loops can be formed with a continuous conductor sheet or using two conductors where a single conductor is used to implement all coaxial loops. Figure 13 of the four conductor loops.

[0215] An uneven or asymmetric field distribution around the resonator including multiple conductor loops can be generated by driving some of the conductor loops with different parameters. Some of the conductor loops of the source resonator having multiple conductor loops can be driven by power supplies with different frequencies, voltages, power levels, duty cycles, etc., all of which can be used to affect the intensity of the magnetic field generated by each conductor.

[0216] The planar resonator structure can be combined with a capacitively loaded inductor resonator coil to provide a comprehensive omnidirectional effective region, including above and below the source, while maintaining the flat resonator structure. As Figure 13 shown, an additional resonator loop coil 1309 including one or more conductor loops can be placed in the plane common with the planar resonator structure 1310. The external resonator coil provides an effective region substantially above and below the source. The resonator coil can be arranged with any number of planar resonator structures and the arrangements described herein.

[0217] The planar resonator structure can be enclosed in a magnetically permeable package or integrated into other devices. The planar profile of the resonator in a single common plane allows for packaging and integration into flat devices. In Figure 14 a diagram is shown illustrating an application of the resonator. A flat source 1411 including one or more planar resonators 1414 (each having one or more conductor loops) can transfer power to devices 1412, 1413 integrated with other planar resonators 1415, 1416 and placed within the effective region 1417 of the source. The devices can include multiple planar resonators such that the effective region of the source remains unchanged regardless of the orientation of the devices relative to the source. In addition to the invariance for rotational misalignment, the flat device including the planar resonator can be completely flipped with substantially no effect on the effective region because the planar resonator remains in the plane of the source.

[0218] In Figure 15Another illustration showing a possible use of a power transfer system using a planar resonator structure is shown. A planar source 1521 placed on a surface 1525 can generate an effective area covering a substantial surface area, which generates an "energized surface" area. Devices such as a computer 1524, a mobile phone 1522, a gaming console, and other electronic devices 1523 coupled to their respective planar device resonators can receive energy from the source when placed within the effective area of the source (which can be anywhere above the surface). Multiple devices of different sizes can be placed in the effective area without strict placement or alignment constraints and used normally while being charged or powered from the source. The source can be placed under the surface of a table, counter, desk, cabinet, etc., allowing it to be completely hidden while energizing the top surface of the table, counter, desk, cabinet, etc., creating an effective area much larger than the source on the surface.

[0219] The source can include a display or other visual, auditory, or vibration indicator to show the direction of the charging device or what device is being charged, charging errors or problems, power level, charging time, etc.

[0220] The source resonator and circuitry can be integrated into any number of other devices. The source can be integrated into devices such as a clock, keyboard, monitor, photo frame, etc. For example, a keyboard integrated with a planar resonator and appropriate power and control circuitry can be used as a source for devices placed around the keyboard, such as a computer mouse, web camera, mobile phone, etc., without occupying any additional desk space.

[0221] Although the planar resonator structure has been described in the context of mobile devices, it should be clear to those skilled in the art that a flat planar source for wireless power transfer with an effective area extending beyond its physical dimensions has many other consumer and industrial applications. The structure and configuration can be useful for many applications where electronic or electrical devices and power sources are typically positioned, set up, or manipulated substantially in the same plane and alignment. Some possible application scenarios include devices on walls, floors, ceilings, or any other substantially planar surface.

[0222] The flat source resonator can be integrated into a photo frame or hung on a wall, providing an effective area within the plane of the wall, where other electronic devices such as a digital photo frame, television, lamp, etc. can be installed and powered without wires. The planar resonator can be integrated into the floor, resulting in an energized floor or an effective area on the floor where devices can be placed to receive power. Audio speakers, lamps, heaters, etc. can be placed within the effective area and receive power wirelessly.

[0223] A planar resonator can have additional components coupled to a conductor. Components such as capacitors, inductors, resistors, diodes, etc. can be coupled to the conductor and can be used to adjust or tune the resonant frequency and impedance matching for the resonator.

[0224] A planar resonator structure of the type described above and shown in (a) of can be produced, for example, with a quality factor Q of 100 or higher and even 1000 or higher. As shown in (c) of, energy can be wirelessly transferred from one planar resonator structure to another over a distance greater than the characteristic dimension of the resonator. Figure 11 in Figure 11 in

[0225] In addition to using magnetic materials to achieve properties similar to the inductive elements in a magnetic resonator, a combination of a good conductor material and a magnetic material can also be used to implement such an inductive structure. Figure 16 (a) of shows a magnetic resonator structure 1602, which can include one or more enclosures made of a high conductivity material (the interior of which can be shielded from the externally generated AC electromagnetic field) surrounded by at least one layer of magnetic material and linked by a magnetic material block 1604.

[0226] The structure can include a sheet of high conductivity material covered with a layer of magnetic material on one side. Alternatively, the layered structure can be conformally applied to an electronic device such that parts of the device can be covered with layers of high conductivity and magnetic materials, while other parts that need to be easily accessible (such as buttons or screens) are not covered. The structure can also or alternatively include only layers or blocks of magnetic material sheets. Thus, the magnetic resonator can be incorporated into an existing device without significantly interfering with its existing functions and with little need for extensive re-design. In addition, the layers of good conductor and / or magnetic material can be made thin enough (about one millimeter or less) such that they will add almost no additional weight and volume to the finished device. An oscillating current applied to a section of conductor wound around the structure, as shown by the square loop at the center of the structure of, can be used to excite the electromagnetic field associated with this structure. Figure 16 in

[0227] Quality factor of the structure

[0228] Structures of the above type can be produced with a quality factor Q of about 1000 or higher. Such a high Q is possible even if the losses in the magnetic material are high, provided that the fraction of the magnetic energy within the magnetic material is small compared to the total magnetic energy associated with the object. For structures consisting of conductive and magnetic material layers, the losses in the conductive material can be reduced by the presence of the magnetic material as described above. In a structure where the thickness of the magnetic material layer is about 1 / 100 of the maximum dimension of the system (e.g., the magnetic material is about 1 mm thick and the area of the structure is about 10 cm × 10 cm) and the relative permeability is about 1000, the fraction of the magnetic energy contained within the magnetic material can be made only a few percent of the total magnetic energy associated with the object or resonator. To see how that occurs, note that the expression for the magnetic energy contained within a volume is U m = ∫ V dr B(r) 2 / (2μ r μ0), provided that B (rather than H) is the dominant field that is maintained across the magnetic material - air interface (which is usually the case in an open magnetic circuit), the fraction of the magnetic energy contained within the high - μ r region can be significantly reduced compared to that in air.

[0229] If the fraction of the magnetic energy in the magnetic material is represented by frac and the loss tangent of the material is tanδ, then the Q of the resonator is Q = 1 / (frac x tanδ), assuming that the magnetic material is the only source of loss. Thus, even for a loss tangent as high as 0.1, a Q of about 1000 can be achieved for these types of resonator structures.

[0230] If the structure is driven with N turns of wire wound around it, then if N is high enough, the losses in the driving inductor loop can be neglected. Figure 17 FIG. 1700 shows a schematic of an equivalent circuit for these structures and the loss mechanisms and the scaling of the inductance with the number of turns N wound around a structure made of conductive and magnetic materials. If the proximity effect can be neglected (by using appropriate windings, or wire designed to minimize the proximity effect, such as Litz wire, etc.), then the resistance 1702 due to the wire in the loop conductor scales linearly with the length of the loop, which in turn is proportional to the number of turns. On the other hand, both the equivalent resistance 1708 and the equivalent inductance 1704 of these particular structures are proportional to the square of the magnetic field within the structure. Since this magnetic field is proportional to N, both the equivalent resistance 1708 and the equivalent inductance 1704 are proportional to N 2 squared. Thus, for large enough N, the resistance 1702 of the wire is much smaller than the equivalent resistance 1708 of the magnetic structure, and the Q of the resonator asymptotes to Q max = ωL μ / Rμ .

[0231] Figure 16 Figure (a) in [reference] shows a diagram of a copper and magnetic material structure 1602 driven by a square current loop around a narrowed section at the center of structure 1604 and the magnetic field streamlines 1608 generated by this structure. This exemplary structure includes two 20 cm × 8 cm × 2 cm hollow regions that are enclosed by copper and then completely covered by a 2 mm magnetic material layer with properties μ′ r = 1,400, μ″ r = 5 and σ = 0.5 S / m. The two parallelepipeds are spaced 4 cm apart and are connected by a 2 cm × 4 cm × 2 cm block of the same magnetic material. The excitation loop is wound around the center of this block. At a frequency of 300 kHz, this structure has a calculated Q of 890. The conductor and magnetic material structure can be shaped to optimize certain system parameters. For example, the size of the structure enclosed by the excitation loop can be small to reduce the resistance of the excitation loop, or it can be large to reduce the losses in the magnetic material associated with a large magnetic field. Note that the magnetic streamlines and Q associated with the same structure composed of magnetic material will only be similar to the layer conductor and magnetic material design shown here.

[0232] Interaction of the electromagnetic resonator with other objects

[0233] For an electromagnetic resonator, the non-intrinsic loss mechanisms that perturb the intrinsic Q value can include absorption losses within the material of nearby unrelated objects and radiation losses related to the scattering of the resonant field from nearby unrelated materials. Absorption losses can be associated with materials that have a non-zero but finite conductivity σ (or equivalently a non-zero and finite imaginary part of the dielectric permittivity) within the frequency range of interest, such that the electromagnetic field can penetrate it and induce currents therein, which then dissipate energy through resistive losses. If an object at least partially includes a lossy material, it can be described as lossy.

[0234] Consider an object that consists of a homogeneous isotropic material with conductivity σ and permeability μ. The penetration depth of the electromagnetic field within this object is given by the skin depth The power P d dissipated within the object can be determined according to P V = ∫ 2 dr σ|E| V = ∫ 2 dr |J| d , where we

[0235] use Ohm's law J = σE, and where E is the electric field and J is the current density.

[0236] If, within the frequency range of interest, the conductivity σ of the material constituting the object is low enough such that the skin depth δ of the material can be considered long (i.e., δ is longer than the characteristic dimension of the object, or δ is longer than the characteristic dimension of the lossy part of the object), then the electromagnetic fields E and H (where H is the magnetic field) can penetrate significantly into the object. Then, these fields with finite values can generate a dissipated power that scales with P d ~σV ol <|E| 2 > where, within the volume under consideration, V ol is the volume of the lossy object and <|E| 2 > is the spatial average of the square of the electric field. Thus, at the lower limit of conductivity, the dissipated power scales proportionally with the conductivity and goes to zero at the limit of a non-conducting (pure dielectric) material.

[0237] If, within the frequency range of interest, the conductivity σ of the material constituting the object is high enough such that the skin depth of the material can be considered short, then the electromagnetic fields E and H can penetrate only a short distance into the object (i.e., they stay near the'mantle' of the material, where δ is less than the characteristic thickness of the lossy part of the object). In this case, the currents induced within the material can be concentrated very close to the material surface, approximately within the skin depth, and can be approximated in magnitude by the product of the surface current density K(x, y) (which is mainly determined by the shape of the incident electromagnetic field and, as long as the thickness of the conductor is much larger than the skin depth, is independent of the frequency and conductivity to the first order) (where x and y are the coordinates parameterizing the surface) and a function that decays exponentially into the surface: exp(−z / δ) / δ (where z represents the coordinate locally perpendicular to the surface): J(x, y, z) = K(x, y) exp(−z / δ) / δ. Then, the dissipated power P d ,

[0238]

[0239] Thus, at the high-conductivity limit, the dissipated power scales inversely with the square root of the conductivity and goes to zero at the limit of a perfectly conducting material.

[0240] If, within the frequency range of interest, the conductivity σ of the material constituting the object is finite, then the skin depth δ of the material can penetrate a certain distance into the object and a certain amount of power may be dissipated within the object, also depending on the size of the object and the strength of the electromagnetic field. This can be generalized to also describe the general case of an object including multiple different materials with different properties and conductivities, such as an object with an arbitrary inhomogeneous and anisotropic distribution of conductivity within the object.

[0241] Note that the magnitude of the above loss mechanism can depend on the position and orientation of the extraneous object relative to the resonator field and the material composition of the extraneous object. For example, a material with high electrical conductivity can shift the resonant frequency of the resonator and detune it from other resonant objects. This frequency shift can be fixed by applying a feedback mechanism to the resonator that corrects its frequency, such as by varying the inductance and / or capacitance of the resonator. These variations can be achieved using variable capacitors and inductors, and in some cases by varying the geometry of the components in the resonator. Other novel tuning mechanisms described below can also be used to vary the resonator frequency.

[0242] In cases where the external losses are high, the perturbed Q can be low, and steps can be taken to limit the absorption of resonator energy within such extraneous objects and materials. Due to the functional dependence of the dissipated power on the electromagnetic field intensity, the system performance can be optimized by designing the system such that the evanescent resonant field tails are shorter at the source resonator and longer at the device resonator to achieve the desired coupling, such that the perturbed Q of the source in the presence of other objects is optimized (or vice versa if the perturbed Q of the device is to be optimized).

[0243] Note that many common extraneous materials and objects, such as people, animals, plants, building materials, etc., can have low electrical conductivity and can thus have little effect on the wireless energy transfer scheme disclosed herein. An important fact related to the magnetic resonator design we describe is that its electric field can be mainly confined within the resonator structure itself, and thus it should be able to operate within the generally accepted guidelines for human safety while providing wireless power exchange at mid-range distances.

[0244] Electromagnetic resonator with reduced interaction

[0245] An interesting frequency range for near-field wireless power transfer is between 10 kHz and 100 MHz. Within this frequency range, many common non-metallic materials, such as multiple types of wood and plastics, can have relatively low electrical conductivity, such that only a small amount of power can be dissipated within them. Additionally, materials with a low loss tangent tanΔ (where tanΔ = ε″ / ε′, and ε″ and ε′ are the imaginary and real parts of the permittivity, respectively) can also cause only a small amount of power to be dissipated within them. Metallic materials with relatively high electrical conductivity, such as copper, silver, and gold, can also have little power dissipated within them because, as discussed previously, the electromagnetic field cannot penetrate these materials significantly. These materials and objects with very low and very high electrical conductivity and low loss tangent can have a negligible effect on the losses of the magnetic resonator.

[0246] However, within the frequency range of interest, there are materials and objects such as certain electronic circuits and certain low-conductivity metals that can have moderate (often inhomogeneous and anisotropic) conductivity and / or moderate to high loss tangent, and that can have relatively high dissipation losses. Relatively large amounts of power can be dissipated within them. These materials and objects can dissipate enough energy to reduce Q to (p) The reduction is significant, and can be called a "lossy object".

[0247] Reducing the Q of the resonator by lossy materials (p) One way to influence the lossy object is to use a high conductivity material to shape the resonator field so that it avoids the lossy object. The process of using a high conductivity material to shape the electromagnetic fields so that they avoid the lossy object in its vicinity can be understood by thinking of the high conductivity material as a material that deflects or reshapes the field. This conception is qualitatively correct as long as the thickness of the conductor is greater than the skin depth, because the boundary conditions for the electromagnetic field at the surface of a good conductor force the electric field to be close to completely perpendicular to the plane of the conductor and the magnetic field to be close to completely tangential to the plane of the conductor. Therefore, the perpendicular magnetic field or tangential electric field will be "deflected away" from the conductive surface. Moreover, the tangential magnetic field or perpendicular electric field can even be forced to be reduced in amplitude on one side and / or at a location in particular of the conductive surface, depending on the relative positions of the source of the field and the conductive surface.

[0248] As an example, Figure 18 A finite element method (FEM) simulation of two high-conductivity surfaces 1802 above and below a lossy dielectric material 1804 in an external, initially uniform magnetic field at a frequency of f = 6.78 MHz is shown. The system is azimuthally symmetric about the r = 0 axis. In this simulation, the lossy dielectric material 1804 is sandwiched between two conductors 1802 (shown as white lines at approximately z = ±0.01 m). In the absence of conductive surfaces above and below the dielectric disk, the magnetic field (represented by the depicted magnetic field lines) would still be essentially uniform (straight and parallel to the z-axis), indicating that the magnetic field would pass straight through the lossy dielectric material. In this case, power would have been dissipated in the lossy dielectric disk. However, in the presence of conductive surfaces, this simulation shows that the magnetic field is reshaped. The magnetic field is forced tangential to the surfaces of the conductors and is therefore deflected around those conductive surfaces 1802, minimizing the amount of power that could be dissipated in the lossy dielectric material 1804 behind or between the conductive surfaces. As used herein, an axis of electrical symmetry refers to any axis about which a fixed or time-varying electric or magnetic field is substantially symmetric during energy exchange as disclosed herein.

[0249] Similar effects are observed even with only one conductive surface above or below the dielectric disk. If the dielectric disk is thin, the fact that the electric field is essentially zero at the surface and approaches it continuously and smoothly means that the electric field is very low anywhere near the surface (i.e., within the dielectric disk). A single-surface implementation for deflecting the resonator field away from lossy objects may be preferred for applications where covering the lossy material or both sides of the surface is not allowed (such as an LCD screen). Note that in the presence of lossy material, even a very thin surface of conductive material on the order of a few skin depths can be sufficient (the skin depth in pure copper at 6.78 MHz is ~20 μm and at 250 kHz is ~100 μm) to significantly improve the Q of the resonator. (p) 。

[0250] Lossy-unrelated materials and objects can be part of the device in which the high-Q resonator will be integrated. Many techniques can be used to reduce the energy dissipation in these lossy materials and objects, including:

[0251] By positioning the lossy materials and objects away from the resonator, or in special positions and orientations relative to the resonator.

[0252] By using high-conductivity materials or structures to partially or completely cover the lossy materials and objects near the resonator.

[0253] By placing a closed surface of high-conductivity material (such as a sheet or mesh) around the lossy object to completely cover the lossy object and shape the resonator field such that it avoids the lossy object.

[0254] By placing a surface of high-conductivity material (such as a sheet or mesh) around only a portion of the lossy object, such as along the top, bottom, along the sides, etc. of the object or material.

[0255] By placing even a single surface of high-conductivity material (such as a sheet or mesh) above, below, or on one side of the lossy object to reduce the intensity of the field at the location of the lossy object.

[0256] Figure 19 Shows the capacitive-loaded loop inductor forming the magnetic resonator 102 and the disk-shaped surface of the high-conductivity material 1802 placed inside the loop inductor and completely surrounding the lossy object 1804. Note that some lossy objects can be components such as electronic circuits that may need to interact, communicate, or be connected to the outside environment and thus cannot be completely electromagnetically isolated. Partially covering the lossy material with high-conductivity material can still reduce the additional losses while allowing the lossy material or object to function properly.

[0257] Figure 20Shows the surface of a capacitively loaded loop inductor used as resonator 102 and a highly conductive material 1802 that is placed inside the inductor loop and surrounds only a portion of the lossy object 1804.

[0258] By placing a single surface of a highly conductive material above, below, or alongside a lossy object or material, etc., additional losses can be reduced but not completely eliminated. In Figure 21 an example is shown where a capacitively loaded loop inductor is used as resonator 102 and the surface of the highly conductive material 1802 is placed inside the inductor loop below the lossy object 1804 to reduce the intensity of the field at the location of the lossy object. Due to considerations such as cost, weight, assembly complexity, air flow, visual accessibility, physical accessibility, etc., it may be preferable to cover only one side of the material or object.

[0259] A single surface of a highly conductive material can be used to avoid objects (such as LCD or plasma screens) that cannot or should not be covered from both sides. An optically transparent conductor can be used to avoid such lossy objects. As an alternative to or in addition to an optically transparent conductor, a highly conductive optically opaque material can alternatively be placed on only a portion of the lossy object. The suitability of single-sided coverage relative to multi-sided coverage embodiments and the design trade-offs inherent therein can depend on the details of the wireless energy transfer scenario and the nature of the lossy materials and objects.

[0260] Below, we describe an example of using a highly conductive surface to improve the Q-insensitivity Θ of an integrated magnetic resonator used in a wireless energy transfer system (p) of. Figure 22 Shows a wireless projector 2200. The wireless projector can include a device resonator 102C, a projector 2202, a wireless network / video adapter 2204, and a power conversion circuit 2208 arranged as shown. The device resonator 102C can include a three-turn conductor loop arranged as a closed surface, and a capacitor network 2210. The conductor loop can be set such that the device resonator 102C has a high Q (e.g., >100) at its operating resonance frequency. Before integration in a fully wireless projector 2200, this device resonator 102C had a Q of approximately 477 at a designed operating resonance frequency of 6.78 MHz. When integrated and the wireless network / video adapter card 2204 is placed at the center of the resonator loop inductor, the resonator Q (integrated) is reduced to approximately 347. From Q to Q (integrated)At least some of the reduction in Q is attributed to perturbing the losses in the wireless network / video adapter card. As described above, the electromagnetic field associated with the magnetic resonator 102C can induce currents in and on the wireless network / video adapter card 2204, which can be dissipated in resistive losses in the lossy materials comprising the card. We have observed that the Q of a resonator can be affected differently depending on the composition, position, and orientation of objects and materials placed near the resonator. (integrated) .

[0261] In the completely wireless projector example, covering the network / video adapter card with a thin copper bag (a folded copper sheet covering the top and bottom of the wireless network / video adapter card, but not the communication antenna) will reduce the Q of the magnetic resonator. (integrated) Improved to Q of about 444 (integrated+copper pocket) In other words, using a copper bag to deflect the resonator field away from the lossy material eliminates most of the Q distortion caused by the perturbations introduced by the extraneous network / video adapter card. (integrated) of the reduction.

[0262] In another example of a completely wireless projector, covering the network / video adapter card with a single copper sheet placed under the card provides approximately equal Q (integrated+copperpocket) Q (integrated+copper sheet) In this example, the perturbed Q of the system can be kept high with a single high conductivity sheet used to keep the resonator field away from the lossy adapter card.

[0263] It may be advantageous to position or orient lossy materials and objects that are part of a device that includes a high-Q electromagnetic resonator at locations where the field generated by the resonator is relatively weak, so that little power is dissipated in these objects, and thus the Q is insensitive Θ. (p) can be large. As shown earlier, materials of different conductivity can respond differently to electric fields versus magnetic fields. Therefore, depending on the conductivity of the extraneous object, the positioning technology can be specific to one field or the other.

[0264] Figure 23Shows the amplitudes of the electric field 2312 and magnetic field 2314 along the line of the diameter of a circular loop inductor resonating at 10 MHz and the electric field 2318 and magnetic field 2320 along the axis of the loop inductor of a capacitively loaded circular loop inductor for a wire with a radius of 30 cm. It can be seen that the amplitude of the resonant near field reaches its maximum value near the wire and decays away from the loops 2312, 2314. In the plane of the loop inductors 2318, 2320, the field reaches a local minimum at the center of the loop. Thus, given the finite size of the device, it is possible that the field is weakest at the extremities of the device and it is possible that the field amplitude has local minima at some places within the device. This argument applies to any other type of electromagnetic resonator 102 and any type of device. In Figure 24 Examples are shown in (a) and (b) therein, where a capacitively loaded inductor loop forms a magnetic resonator 102 and an irrelevant lossy object 1804 is located at the position where the electromagnetic field has the minimum amplitude.

[0265] In a demonstration example, a three-turn conductor loop and a capacitor network arranged to enclose a square surface (with rounded corners) are used to form a magnetic resonator. The Q of the resonator is approximately 619 at the designed operating resonance frequency of 6.78 MHz. The perturbed Q of this resonator depends on the placement of the perturbing object (in this case a pocket projector) relative to the resonator. When the perturbing projector is placed inside the inductor loop and at its center or on the inductor turns, Q (projector) is approximately 96, which is lower than when the perturbing projector is placed outside the resonator (in this case, Q (projector) is approximately 513). These measurement results support the analysis showing that the fields inside the inductor loop can be larger than those outside it, so a lossy object placed inside such a loop inductor can provide a lower perturbed Q for the system compared to when the lossy object is placed outside the loop inductor. Depending on the resonator design and the material composition and orientation of the lossy object, Figure 24 the arrangement shown in (b) in Figure 24 can provide a higher Q insensitivity Θ than the arrangement shown in (a) in (projector) .

[0266] A high-Q resonator can be integrated inside a device. Irrelevant materials and objects with high dielectric permittivity, magnetic permeability, or conductivity can be part of the device into which the high-Q resonator will be integrated. For these irrelevant materials and objects near a high-Q electromagnetic resonator, depending on their size, position, and orientation relative to the resonator, the resonator field distribution can be deformed and deviate significantly from the original unperturbed field distribution of the resonator. Such deformation of the unperturbed field of the resonator can significantly reduce the Q to a lower Q (p) , even if the irrelevant objects and materials are lossless.

[0267] It may be advantageous to place high - conductivity objects, which are part of a device including a high - Q electromagnetic resonator, in an orientation such that the surfaces of these objects are as perpendicular as possible to the electric field lines generated by the undisturbed resonator and parallel to the magnetic field lines generated by the undisturbed resonator, thus deforming the resonant - field distribution by the smallest possible amount. Other common objects that can be set perpendicular to the plane of the magnetic resonator loop include screens (LCD, plasma, etc.), batteries, enclosures, connectors, radiating antennas, etc. The Q - insensitivity Θ (p) of the resonator can be much larger than in cases where the objects are set in different orientations relative to the resonator fields.

[0268] Loss - independent materials and objects that are not part of an integrated device including a high - Q resonator can be located or brought near the resonator, e.g., during the use of the device. In some cases, high - conductivity materials are used to adjust the resonator field so that it avoids the regions where the loss - independent objects are located or introduced to reduce the power dissipation in these materials and objects and increase the Q - insensitivity Θ (p) may be advantageous. In Figure 25 an example is shown where a capacitively - loaded loop inductor and capacitor are used as resonator 102, and the surface of a high - conductivity material 1802 is placed above the inductor loop to reduce the magnitude of the field in the region above the resonator, where the loss - independent object 1804 can be located or introduced.

[0269] Note that a high - conductivity surface brought near the resonator to reshape the field can also cause Q (cond . surface) <Q. The reduction in the perturbed Q can be due to energy dissipation within a lossy conductor or the deformation of the undisturbed resonator field associated with the field - boundary conditions at the surface of a matching conductor. Thus, while a high - conductivity surface can be used to reduce the additional losses due to dissipation within the irrelevant lossy objects, in some cases, especially in some cases where this is achieved by significantly reshaping the electromagnetic field, using such a high - conductivity surface to make the field avoid the lossy objects can effectively result in Q (p+cond.surface) <Q (p) instead of the desired result Q (p+cond.surface) >Q (p) .

[0270] As described above, in the presence of objects with induced losses, if the electromagnetic field associated with the magnetic resonator is reshaped to avoid the objects with induced losses, the perturbed quality factor of the magnetic resonator can be improved. Another way to reshape the undisturbed resonator field is to use a high - permeability material to completely or partially enclose or cover the objects with induced losses, thus reducing the interaction of the magnetic field with the objects with induced losses.

[0271] Magnetic field shielding has been previously described, for example, in Electrodtnamics 3rd Ed., Jackson, pp. 201 - 203. Therein, a spherical shell of a permeable material that shields its interior from an external magnetic field is shown. For example, if a shell with an inner diameter a, an outer diameter b, and a relative permeability μ r is placed in an initially uniform magnetic field H0, the field inside the shell will have a constant magnitude of 9μ r H0 / [(2μ r + 1)(μ r + 2) - 2(a / b) 3 (μ r - 1) 2 , and if μ r >> 1, it tends to 9H0 / 2μ r (1 - (a / b) 3 ). This result shows that the incident magnetic field (but not necessarily the incident electric field) can be greatly attenuated inside the shell, even if the shell is quite thin, provided that the permeability is high enough. In some cases, it may be advantageous to use a high - permeability material to partially or completely cover lossy materials and objects such that they are avoided by the resonator magnetic field and thus little power is dissipated in these materials and objects. In this method, the Q - insensitivity Θ (p) can be larger than in the case where the materials and objects are not covered, possibly greater than 1.

[0272] It is desirable to keep both the electric and magnetic fields away from loss - inducing objects. As described above, one way to shape the fields in this manner is to use a high - conductivity surface to completely or partially enclose or cover the loss - inducing objects. A layer of permeable material, also known as a magnetic material (any material or metamaterial with a significant permeability), can be placed on or around the high - conductivity surface. This additional magnetic material layer can present a lower - magnetic - resistance path (compared to free space) for the deflected magnetic field to follow and can partially shield the underlying electrical conductor from the incident magnetic flux. This setup can reduce the losses due to the induced currents in the high - conductivity surface. In some cases, the lower magnetic resistance presented by the magnetic material can improve the perturbed Q of the structure.

[0273] Figure 26 Figure (a) in [reference] shows an axisymmetric FEM simulation of a thin conducting 2604 (copper) disk (20 cm in diameter and 2 cm in height) exposed to an initially uniform, externally applied magnetic field (gray magnetic flux lines) along the z - axis. The axis of symmetry is at r = 0. The shown magnetic streamlines originate from z = -∞, where these magnetic streamlines are spaced 1 cm apart from r = 3 cm to r = 10 cm. The axis scale is in meters. For example, imagine that this conducting cylinder encloses a loss - inducing object within a region defined by Figure 19within the region defined by the magnetic resonator in the wireless energy transfer system shown.

[0274] This highly conductive enclosure can increase the perturbed Q of a lossy object and thus increase the total perturbed Q of the system, but the perturbed Q may still be less than the unperturbed Q due to induced losses in the conductive surface and changes in the electromagnetic field distribution. The reduction in perturbed Q associated with the highly conductive enclosure can be at least partially restored by including a layer of magnetic material along one or more outer surfaces of the highly conductive enclosure. Figure 26 In (b) shows the axisymmetric FEM simulation of the thin conductive 2604A (copper) disk (20 cm in diameter and 2 cm in height) from Figure 26 in (a), but with an additional layer of magnetic material directly placed on the outer surface of the highly conductive enclosure. Note that the presence of the magnetic material can provide a lower magnetic resistance path for the magnetic field, thereby at least partially shielding the underlying conductor and reducing the losses due to induced eddy currents in the conductor.

[0275] Figure 27 depicts a modification (in an axisymmetric view) of the system shown in Figure 26 where not all of the lossy material 2708 may be covered by the highly conductive surface 2706. In some cases, it may be useful to cover only one side of the material or object, such as due to considerations of cost, weight, assembly complexity, air flow, visual accessibility, physical accessibility, etc. In the Figure 27 exemplary arrangement shown, only one surface of the lossy material 2708 is covered, and the resonator inductor loop is placed on the opposite side of the highly conductive surface.

[0276] A mathematical model is used to simulate a highly conductive enclosure made of copper and shaped like a cylindrical disk 20 cm in diameter by 2 cm in height placed within the region defined by a magnetic resonator, the inductive element of which is a single-turn wire loop with a loop radius r = 11 cm and a wire radius a = 1 mm. Simulation of the 6.78 MHz electromagnetic field applied shows that the perturbed quality factor δQ (enclosure) of this highly conductive enclosure r is 1,870. When the highly conductive enclosure is modified to include a 0.25 cm thick layer of magnetic material with a real relative magnetic permeability μ′ r = 40 and an imaginary relative magnetic permeability μ″ -2 = 10 <0, (enclosure+magnetic material) = 5,060.

[0277] If a large portion of the region defined by the loop inductor 2704 of the resonator is filled with a high-conductivity outer shell, the performance improvement due to the addition of a thin layer of magnetic material 2702 can be even more significant. In the above example, if the radius of the inductor loop 2704 is reduced such that it is only 3 mm away from the surface of the high-conductivity outer shell, the perturbation quality factor can be improved from 670 (for the conductive outer shell only) to 2,730 (for the conductive outer shell with a thin layer of magnetic material) by adding a thin layer of magnetic material 2702 outside the outer shell.

[0278] The resonator structure can be designed to have a highly restricted electric field using, for example, shielding or distributed capacitors (e.g., which yield high), even when the resonator is very close to a material that would normally induce losses.

[0279] Coupled electromagnetic resonator

[0280] The efficiency of energy transfer between two resonators can be determined by the strong coupling quality factor. In a magnetic resonator implementation, The coupling factor between two resonators can be made related to the inductances L1 and L2 of the inductive elements in each of the resonators and the mutual inductance M between them. Note that this expression assumes negligible coupling through electric dipoles. For a capacitively loaded inductor loop resonator where the inductor loop is formed by circular conductive loops with N turns, separated by a distance D and oriented as shown in (b) of Figure 1 the mutual inductance is M = π / 4·μ o N1N2(x1x2) 2 / D 3 , where x1, N1 and x2, N2 are the characteristic dimensions and number of turns of the conductor loops of the first and second resonators respectively. Note that this is a quasi-static result and thus assumes that the dimensions of the resonators are much smaller than the wavelength, and the distance between the resonators is much smaller than the wavelength and is at least several times their dimensions. For these circular resonators operating at the quasi-static limit and at mid-range distances, as described above, When the quality factor of the resonators is large enough to compensate for the small k at mid-range distances, strong coupling (large U) can be established between the resonators at mid-range distances.

[0281] For electromagnetic resonators, if the two resonators include conductive parts, the coupling mechanism can be the induction of currents on one resonator due to the electric and magnetic fields generated from the other resonator. The coupling factor can be proportional to the flux of the magnetic field generated by the high-Q inductive element in one resonator across the enclosed region of the high-Q inductive element of the second resonator.

[0282] Coupled electromagnetic resonator with reduced interaction

[0283] As described above, a high conductivity material surface can be used to shape the resonator field such that it avoids the lossy object p near the resonator, thereby reducing the total additional loss and maintaining the high Q insensitivity Θ of the resonator. (p+cond..surface) However, such surfaces may cause a perturbed coupling factor k between resonators. (p+cond.surface) which is less than the perturbed coupling factor k (p) and depends on the size, position, and orientation of the high conductivity material relative to the resonator. For example, if the high conductivity material is placed in and within the region of the plane defined by the inductive element of at least one of the magnetic resonators in a wireless energy transfer system, it can block a portion of the magnetic flux passing through the region of the resonator (regulating the coupling), and k can be reduced.

[0284] Consider again Figure 19 the example. In the absence of a high conductivity disk enclosure, a certain amount of external magnetic flux can pass through the defined region of the loop. In the presence of a high conductivity disk enclosure, some of this magnetic flux may be deflected or blocked and may no longer pass through this region of the loop, thus resulting in a smaller perturbed coupling factor k. 12(p+cond.surfaces) However, since the deflected magnetic field lines may closely follow the edges of the high conductivity surface, the reduction in the flux through the loop defining the disk may be less than the ratio of the area of the disk's face to the area of the loop.

[0285] High conductivity material structures can be used alone or in combination with magnetic materials to optimize the perturbed quality factor, the perturbed coupling factor, or the perturbed efficiency.

[0286] Consider Figure 21 the example. Let the lossy object have dimensions equal to those of a capacitively loaded inductor loop resonator, thereby filling its region A 2102. A high conductivity surface 1802 can be placed beneath the lossy object 1804. Let this resonator 1 be in a system of two coupled resonators 1 and 2, and let us consider how U scales as the area A S 2104 of the conductive surface increases. 12(object+cond.surface) versus U 12 Compared. In the absence of the conductive surface 1802 beneath the lossy object 1804, the k insensitivity β 12(object) can be approximately 1, but the Q insensitivity Θ 1(object) can be small, and thus the U insensitivity Ξ 12(object) can be small.

[0287] When the high conductivity surface beneath the lossy object covers the entire area of the inductor loop resonator (A S = A), k 12(object+cond.surface)can be close to zero because little flux is allowed to pass through the inductor loop, and thus U 12(object+cond.surface) can be close to zero. For intermediate sizes of highly conductive surfaces, the suppression of non-intrinsic losses and the associated Q-insensitivity Θ 1(object+cond.surface) compared to Θ 1(object) can be large enough, while the reduction in coupling may not be significant, and the associated k-insensitivity β 12(object+cond.surface) may not be much smaller than β 12(object) such that compared to U 12(object) the total U can be increased 12(object+cond.surface) . The optimal degree of avoiding unwanted lossy objects via a highly conductive surface in a wireless energy transfer system can depend on the details of the system configuration and application.

[0288] We describe a potential method for achieving a high perturbed Q for a system by using a highly conductive material to completely or partially enclose or cover lossy objects near a high-Q resonator. However, using a good conductor alone to cover the object can reduce the coupling of the resonator as described above, thus reducing the efficiency of wireless power transfer. As the area of the conductive surface approaches the area of the magnetic resonator, for example, the perturbed coupling factor k (p) can be close to zero, making the use of the conductive surface incompatible with efficient wireless power transfer.

[0289] One way to address the above issues is to place a layer of magnetic material around the highly conductive material, as this additional layer of permeable magnetic material can present a lower magnetic resistance path (compared to free space) for the deflected magnetic field to follow and can partially shield the underlying electrical conductor from the incident magnetic flux. In some cases, the lower magnetic resistance path presented by the magnetic material can improve the electromagnetic coupling of the resonator to other resonators. The reduction in the perturbed coupling factor associated with using a conductive material to shape the resonator field to avoid lossy objects in and around a high-Q magnetic resonator can be at least partially restored by including a layer of magnetic material along one or more outer surfaces of the conductive material. The magnetic material can increase the perturbed coupling factor relative to its initial unperturbed value.

[0290] Note that Figure 26 the simulation results in Figure 26 show that a layered magnetic material and a conductive structure can deflect the incident magnetic field less compared to a conductive structure alone. If a magnetic resonator ring with a radius only slightly larger than the disk shown in (a) of Figure 26 and (b) of Figure 26 defines the disk, it is clear that more magnetic flux lines will be trapped in the case shown in (b) of Figure 26 compared to the case shown in (a) of Figure 26For the case shown in (b) therein, k (disk) will be larger. Thus, including a layer of magnetic material on the conductive material can improve the overall system performance. System analysis can be performed to determine whether these materials should be partially, fully, or minimally integrated into the resonator.

[0291] As described above, Figure 27 illustrates a layered conductor 2706 and magnetic material 2702 structure that can be suitable for use when not all lossy materials 2708 can be covered by conductor and / or magnetic material structures. Previously shown for a copper conductor disk with a 20 cm diameter and 2 cm height defined by a resonator with an inductor loop radius of 11 cm and a wire radius of a = 1 mm, the perturbation Q calculated for a copper cylinder was 1.870. If the resonator and the conductive disk shell are placed in a uniform magnetic field (aligned along the axis of symmetry of the inductor loop), we calculated a coupling factor insensitivity of 0.34 for the copper conductor. For comparison, we modeled the same arrangement but included a 0.25 cm thick magnetic material layer with a real relative permeability μ r ′ = 40 and an imaginary relative permeability μ r ″ = 10 -2 . Using the same model and parameters as above, we found that by adding magnetic material to the surface of the conductor, the coupling factor insensitivity was improved to 0.64.

[0292] The magnetic material can be placed within the region defined by the magnetic resonator to increase the coupling in a wireless energy transfer system. Consider a solid sphere of magnetic material with a relative permeability μ r placed in an initially uniform magnetic field. In this example, the lower magnetic resistance path provided by the magnetic material can cause the magnetic field to concentrate within the volume of the sphere. We found that by adding magnetic material, the magnetic flux through the region defined by the mid-latitude of the sphere is increased to 3μ r / (μ r + 2) times. If μ r >> 1, this enhancement factor can be close to 3.

[0293] It can also be shown that the dipole moment of a system including a magnetic sphere defined by an inductive element in a magnetic resonator will enhance its magnetic dipole by the same factor. Thus, a magnetic sphere with a high magnetic permeability effectively triples the dipole magnetic coupling of the resonator. If we use a spherical shell of magnetic material with an inner diameter of a and an outer diameter of b, most of this increase in coupling can be maintained even if this shell is on top of a block or enclosure made of a highly conductive material. In this case, the enhancement of the flux through the mid-plane is

[0294]

[0295] For μ r = 1,000 and (a / b) = 0.99, this enhancement factor is still 2.73, so that even with a thin layer of magnetic material, the coupling can be significantly improved.

[0296] As described above, a structure including a magnetic material can be used to implement a magnetic resonator. Figure 16 (a) in shows a three-dimensional model of a copper and magnetic material structure 1600 driven by a square current loop around a blocking point at its center. Figure 16 (b) in shows the interaction indicated by magnetic field streamlines between two identical structures 1600A - B having the same properties as one shown in Figure 16 (a) in. Due to symmetry and to reduce computational complexity, only half of the system is modeled. If we fix the relative orientation between the two objects and change their center distance (the images shown are at a relative separation of 50 cm), we find that at 300 kHz, the coupling efficiency changes from 87% to 55% as the separation between the structures changes from 30 cm to 60 cm. Each exemplary structure 1600A - B shown includes two 20 cm × 8 cm × 2 cm parallelepipeds made of copper joined by a 4 cm × 4 cm × 2 cm magnetic material block and completely covered by a 2 mm thick layer of the same magnetic material (assumed to have μ r = 1,400 + j5). Resistive losses in the drive loop are ignored. Each structure has a calculated Q of 815.

[0297] Electromagnetic resonator and impedance matching

[0298] Impedance matching architecture for low-loss inductive elements

[0299] For the purposes of this discussion, an inductive element can be any coil or loop structure ('loop') of any conductive material with or without (with or without a gap) a core made of a magnetic material, which can also be inductively or in any other contactless manner coupled to other systems. The element is inductive because its impedance (including both the impedance of the loop and the so-called'reflected' impedance of any potentially coupled system) has a positive reactance X and a resistance R.

[0300] Consider external circuits such as a drive circuit or a driven load or a transmission line to which the inductive element can be connected. The external circuit (e.g., the drive circuit) can deliver power to the inductive element, and the inductive element can deliver power to the external circuit (e.g., the driven load). The efficiency and amount of power delivered between the inductive element and the external circuit at a desired frequency can depend on the impedance of the inductive element relative to the nature of the external circuit. Impedance matching networks and external circuit control techniques can be used to regulate the power delivery between the external circuit and the inductive element at the desired frequency f.

[0301] The external circuit can be a drive circuit configured to form amplifiers of classes A, B, C, D, DE, E, F, etc., and can deliver power with maximum efficiency (i.e., with minimum loss within the drive circuit) when driving a resonant network having a specific impedance , where Z o can be a complex number, and * denotes the complex conjugate. The external circuit can be a driven load configured to form rectifiers of classes A, B, C, D, DE, E, F, etc., and can receive power with maximum efficiency (i.e., with minimum loss within the driven load) when being driven by a resonant network having a specific impedance , where Z o can be a complex number. The external circuit can be a transmission line having a characteristic impedance Z o and can exchange power with maximum efficiency (i.e., with zero reflection) when connected to an impedance . We will refer to the characteristic impedance Z o of the external circuit as the complex conjugate of the impedance to which it can be connected for maximum efficiency power exchange.

[0302] Typically, the impedance R + jX of an inductive element can be very different from . For example, if the inductive element has low loss (high X / R), its resistance R can be much lower than the real part of the characteristic impedance Z0 of the external circuit. Additionally, the inductive element itself may not be a resonant network. The impedance matching network connected to the inductive element can typically create a resonant network and can adjust its impedance.

[0303] Therefore, the impedance matching network can be designed to maximize the efficiency of the power delivered between the external circuit and the inductive element (including the reflected impedance of any coupling system). The efficiency of the delivered power can be maximized by matching the impedance of the combination of the impedance matching network and the inductive element to the characteristic impedance of the external circuit (or transmission line) at the desired frequency.

[0304] The impedance matching network can be designed to deliver a specified amount of power between the external circuit and the inductive element (including the reflected impedance of any coupling system). The delivered power can be determined by adjusting the complex ratio of the impedance of the combination of the impedance matching network and the inductive element to the impedance of the external circuit (or transmission line) at the desired frequency.

[0305] The impedance matching network connected to the inductive element can create a magnetic resonator. For some applications such as wireless power transfer using strongly coupled magnetic resonators, a high Q may be desired for the resonator. Therefore, the inductive element can be selected to have low loss (high X / R).

[0306] Since matching circuits typically can include additional loss sources within the resonator, components of the matching circuit can also be selected to have low loss. Additionally, in high power applications and / or due to high resonator Q, large currents travel in a portion of the resonator circuit and there are large voltages across certain circuit elements within the resonator. Such currents and voltages can exceed the specified tolerances for particular circuit elements and may be too high for a particular component to withstand. In some cases, it may be difficult to find or implement components (e.g., such as tunable capacitors) with dimensions, cost, and performance (loss and current / voltage ratings) specifications sufficient to achieve a high Q and high power resonator for certain applications. We disclose designs, methods, embodiments, and techniques for matching circuits that can maintain a high Q for magnetic resonators while reducing the requirements for low loss and / or high current / voltage rated components.

[0307] Matching circuit topologies can be designed that minimize the loss and current rating requirements for certain elements of the matching circuit. The topology of a circuit that matches a low-loss inductive element to an impedance Z0 can be selected such that some of its components are external to the associated high-Q resonator in series with the external circuit. The requirements for low series loss or high current rating for these components can be reduced. Mitigating the low series loss and / or high current rating requirements on circuit elements is particularly useful when the elements need to be variable and / or have large voltage ratings and / or low parallel loss.

[0308] Matching circuit topologies can be designed that minimize the voltage rating requirements for certain elements of the matching circuit. The topology of a circuit that matches a low-loss inductive element to an impedance Z0 can be selected such that some of its components are external to the associated high-Q resonator in parallel with Z0. The requirements for low parallel loss or high voltage rating for these components can be reduced. Mitigating the low parallel loss and / or high voltage requirements on circuit elements is particularly useful when the elements need to be variable and / or have large current ratings and / or low series loss.

[0309] The topology of a circuit that matches a low-loss inductive element to an external characteristic impedance Z0 can be selected such that the field pattern of the associated resonant mode and thus its high Q is maintained when coupling the resonator to the external impedance. Otherwise, inefficient coupling to the desired resonant mode (possibly due to coupling to other undesired resonant modes) can occur, resulting in an effective reduction in resonator Q.

[0310] For applications where low-loss inductive elements or external circuits may exhibit variations, it may be necessary to dynamically adjust a matching circuit to match the impedance of the inductive element to an external circuit impedance Z0 at a desired frequency f. Since there can typically be two tuning objectives, which match or control the real and imaginary parts of the impedance level Z0 at the desired frequency f, there can be two variable elements in the matching circuit. For an inductive element, the matching circuit may need to include at least one variable capacitive element.

[0311] A low-loss inductive element can be matched by using the topology of two variable capacitors or a network of two variable capacitors. For example, the variable capacitor can be a tunable butterfly capacitor having, for example, a center terminal for connection to a ground or other lead for a power supply or load and at least one other terminal across which the capacitance of the tunable butterfly capacitor can be changed or tuned, or can be any other capacitor having a user-configurable, variable capacitance.

[0312] A low-loss inductive element can be matched by using the topology of a network of one variable capacitor or variable capacitors and a network of one variable inductor or variable inductors.

[0313] A low-loss inductive element can be matched by using the topology of a network of one variable capacitor or variable capacitors and a network of one variable mutual inductance or variable mutual inductances, the variable mutual inductance transformer-coupling the inductive element to an external circuit or other system.

[0314] In some cases, it may be difficult to find or implement tunable lumped elements having the size, cost, and performance specifications sufficient to achieve a high-Q, high-power, and potentially high-speed, tunable resonator design. The topology of the circuit that matches the variable inductive element to the external circuit can be designed such that certain variability is imparted to the external circuit by changing the frequency, amplitude, phase, waveform, duty factor, etc. of a drive signal applied to transistors, diodes, switches, etc. in the external circuit.

[0315] Variations in the resistance R and inductance L of the inductive element at the resonant frequency can be compensated for only partially or not at all. Thus, proper system performance can be maintained by tolerances designed into other system components or specifications. Partial tuning achieved with fewer tunable components or less capable tunable components may be sufficient.

[0316] A matching circuit architecture can be designed that achieves the desired variability of an impedance matching circuit under high power conditions while minimizing the voltage / current rating requirements on its tunable elements and achieving finer (i.e., more precise, with higher resolution) overall tunability. The topology of a circuit that matches a variable inductive element to an impedance Z0 can include an appropriate combination and placement of fixed and variable elements such that the voltage / current requirements on the variable components can be reduced and the desired tuning range can be covered with finer tuning resolution. The voltage / current requirements on the non-variable components can be reduced.

[0317] The disclosed impedance matching architectures and techniques can be used to achieve the following:

[0318] Maximize the power delivered from a power driving generator to a source low-loss inductive element (and any other system wirelessly coupled thereto), or minimize the impedance mismatch therebetween.

[0319] Maximize the power delivered from a device low-loss inductive element (and any other system wirelessly coupled thereto) to a power driving load, or minimize the impedance mismatch therebetween.

[0320] Deliver a controlled amount of power from a power driving generator to a source low-loss inductive element (and any other system wirelessly coupled thereto), or achieve a certain impedance relationship therebetween.

[0321] Deliver a controlled amount of power from a device low-loss inductive element (and any other system wirelessly coupled thereto) to a power driving load, or achieve a certain impedance relationship therebetween.

[0322] Topology for mode distribution maintenance (high-Q)

[0323] A resonator structure can be designed to be connected wirelessly (indirectly) or hard-wired (directly) to a generator or a load.

[0324] Consider a general indirect coupling matching topology such as shown by the block diagram in (a) of Figure 28 wherein the inductive element 2802 labeled (R, L) and represented by the circuit symbol for an inductor can be any inductive element discussed in the present disclosure or in the references provided herein, and wherein the impedance matching circuit 2402 includes or consists of parts A and B. B can be a part of the matching circuit that connects the impedances 2804, Z0 to the rest of the circuit (a combination of A and the inductive element (A+(R, L))) via a wireless connection (inductive or capacitive coupling mechanism).

[0325] The combination of A and inductive element 2802 can form a resonator 102 with associated current and charge distributions, which can individually support high-Q resonator electromagnetic modes. The absence of a wired connection between the external circuit Z0 and B and the resonator A+(R, L) can ensure that the high-Q resonator electromagnetic mode and its current / charge distribution can take the form of their inherent (isolated) distribution, as long as the degree of wireless coupling is not too large. That is, using an indirect coupling matching topology, the electromagnetic mode, current / charge distribution, and thus the high-Q of the resonator can be automatically maintained.

[0326] In cases where inductive coupling is used between an external circuit and an inductor loop, this matching topology can be referred to as indirect coupling or transformer coupling or inductive coupling. In the demonstration of wireless energy transfer at mid-range distances described in the referenced Science article, this type of coupling is used to couple a power source to a source resonator and a device resonator to a light bulb.

[0327] Next, consider an example of a system where the inductive element can include an inductive element and any indirect coupling. In this case, as disclosed above, and again due to the absence of a wired connection between the external circuit or coupling system and the resonator, the coupling system can, to a good approximation for a not-too-large degree of indirect coupling, not affect the resonator electromagnetic mode distribution and current / charge distribution of the resonator. Thus, as defined herein, an indirect coupling matching circuit can function equally well for any general inductive element that is part of a resonator and for an inductive element that is wirelessly coupled to other systems. Throughout this disclosure, the matching topologies we disclose refer to the matching topologies for such general inductive elements, that is, where any additional system can be indirectly coupled to a low-loss inductive element, and it should be understood that those additional systems do not significantly affect the resonator electromagnetic mode distribution and current / charge distribution of the resonator.

[0328] Based on the above discussion, in a wireless power transfer system with any number of coupled source resonators, device resonators, and intermediate resonators, the wireless magnetic (inductive) coupling between the resonators does not affect the electromagnetic mode distribution and current / charge distribution of each resonator. Thus, when these resonators have high (unloaded and undisturbed) Q, their (unloaded and undisturbed) Q can be maintained in the presence of wireless coupling. (Note that the loaded Q of a resonator can be reduced in the presence of wireless coupling to another resonator, but we may be interested in maintaining the unloaded Q, which only involves loss mechanisms and not coupling / loading mechanisms).

[0329] Consider a matching topology such as Figure 28 shown in (b) of Figure 28The capacitor shown in (b) therein can represent a capacitor circuit or network. The shown capacitor can be used to form a resonator 102 and adjust the frequency and / or impedance of the source and device resonators. This resonator 102 can be directly coupled to an impedance Z0 using the port labeled "Terminal Connection" 2808. Figure 28 (c) therein shows a generalized direct-coupling matching topology, where the impedance matching circuit 2602 includes or consists of parts A, B, and C. Here, the circuit elements in A, B, and C can be regarded as part of the resonator 102 and part of the impedance matching 2402 (and frequency tuning) topology. B and C can be parts of the matching circuit 2402 that connect the impedance Z0 2804 (or network terminal) to the rest of the circuit (A and inductive elements) via each single-wire connection. Note that B and C can be empty (shorted). If we disconnect or open the parts B and C (i.e., those single-wire connections), the combination of A and the inductive elements (R, L) can form a resonator.

[0330] The electromagnetic mode of a high-Q resonator can cause the distribution of the voltage along the inductive element to have nodes, i.e., positions where the voltage is zero. One node can be approximately at the center of the length of the inductive element, such as the center of the conductor used to form the inductive element (with or without magnetic material), and at least one other node can be within A. The voltage distribution can be approximately antisymmetric along the inductive element with respect to its voltage nodes. High Q can be maintained by designing the matching topology (A, B, C) and / or the terminal voltages (V1, V2) such that this high-Q resonator electromagnetic mode distribution can be approximately maintained on the inductive element. This high-Q resonator electromagnetic mode distribution can be approximately maintained on the inductive element by keeping the voltage nodes of the inductive element (approximately at the center). Examples of achieving these design goals are provided herein.

[0331] A, B, and C can be arbitrary (i.e., without any special symmetry), and V1 and V2 can be chosen such that the voltage across the inductive element is symmetric (voltage node at the center inductance). These results can be achieved using a simple matching circuit but potentially complex terminal voltages, as a topology-dependent common-mode signal (V1 + V2) / 2 may be required on the two terminals.

[0332] Consider an 'axis' that connects all the voltage nodes of the resonator, where again, one node is approximately at the center of the length of the inductive element and the others are within A. (Note that the 'axis' is actually a set of points (voltage nodes) within the circuit topology and may not necessarily correspond to a straight axis of the actual physical structure. The 'axis' can align with the physical axis when the physical structure has symmetry.) If the impedance seen between each of two points and a point on the 'axis' (i.e., a voltage node of the resonator) is the same, then the two points of the resonator are electrically symmetric with respect to the 'axis'.

[0333] B and C can be the same (C = B), and as Figure 28 shown in (d) of [], two terminals can be connected to any two points of a resonator (A+(R,L)) that are electrically symmetric with respect to the 'axis' defined above and driven by opposite voltages (V2 = -V1). Two electrically symmetric points of resonator 102 can be two electrically symmetric points on the inductor loop. Two electrically symmetric points of the resonator can be two electrically symmetric points within A. If the two electrically symmetric points (to each of which an equal part B and C is connected) are within A, then it may be necessary to design A such that these electrically symmetric points are accessible as connection points within the circuit. This topology can be referred to as a 'balanced drive' topology. These balanced drive examples can have the advantage that, for example, any common mode signals that may appear on the ground line (and which may not reach the resonator) can be automatically rejected due to perturbations at the external circuit or power grid. In some balanced drive examples, this topology may require more components than other topologies.

[0334] In other examples, C can be chosen to be shorted, and the corresponding terminal is connected to ground (V = 0) and any point on the electrically symmetric (zero voltage) 'axis' of the resonator, and B is connected to any other point of the resonator that is not on the electrically symmetric 'axis', as Figure 28 shown in (e) of []. The ground point on the electrically symmetric 'axis' can be a voltage node on the inductive element, approximately at the center of its conductor length. The ground point on the electrically symmetric 'axis' can be within circuit A. In the case where the ground point on the electrically symmetric 'axis' is within A, it may be necessary to design A to include one such electrically accessible point on the electrically symmetric 'axis', i.e., a place where a connection can be made.

[0335] This topology can be referred to as an 'unbalanced drive' topology. An approximately antisymmetric voltage distribution of the electromagnetic modes along the inductive element can be approximately maintained, even if the resonator may not be driven completely symmetrically. The reason is that the high Q and large associated R compared to the Z0 mismatch require that a much smaller current can flow through B and ground compared to the much larger current that can flow inside the resonator (A+(R,L)). In this case, the perturbation on the resonator modes can be weak, and the position of the voltage node can be approximately maintained at the center position of the inductive element. These unbalanced drive examples can have the advantage that they can be implemented using a simple matching circuit, and there is no limitation on the drive voltage at the V1 terminal. In some unbalanced drive examples, additional design may be required to reduce the common-mode signals that may appear at the ground terminal.

[0336] The direct-coupled impedance matching circuit generally including parts A, B, and C or consisting of them as shown in (c) of Figure 28 can be designed such that the wires and components of the circuit do not perturb the electric and magnetic field distributions of the electromagnetic modes of the inductive element and / or resonator and thus maintain a high resonator Q . The wires and metal components of the circuit can be oriented perpendicular to the electric field lines of the electromagnetic mode. The wires and components of the circuit can be placed in a region where the electric and magnetic fields of the electromagnetic mode are weak.

[0337] Topology for reducing low series loss and high current rating requirements on components

[0338] If the matching circuit used to match the small resistance R of the low-loss inductive element to the larger characteristic impedance Z0 of the external circuit can be considered lossless, then and the current flowing through the terminal is much smaller than the current flowing through the inductive element. Therefore, the components directly connected in series with the terminal (such as in B, C ( Figure 28 in (c) of)) may not carry a high current. Then, even if the matching circuit has lossy elements, the resistive losses in the elements appearing in series with the terminal may not cause a significant reduction in the high Q of the resonator. That is, the resistive losses in those series elements may not significantly reduce the efficiency of power transfer from Z0 to the inductive element or vice versa. Therefore, for these components, there may be no need for strict requirements on low series losses and / or high current ratings. Generally, such reduced requirements can result in a wider selection of components that can be designed into high Q and / or high power impedance matching and resonator topologies. These reduced requirements are particularly helpful in expanding the variety of variable and / or high voltage and / or low parallel loss components that can be used in these high Q and / or high power impedance matching circuits.

[0339] Topology for reducing low parallel loss and high voltage rating requirements on components

[0340] If the matching circuit used to match the small resistance R of the low-loss inductive element to the larger characteristic impedance Z0 of the external circuit as described above is lossless, then using the foregoing analysis,

[0341]

[0342] Moreover, for a low-loss (high X / R) inductive element, the voltage across the terminals can typically be much smaller than the voltage across the inductive element. Thus, the element directly shunted to the terminals may not need to withstand a high voltage. Then, even if the matching circuit has lossy elements, the resistive losses in the elements shunted to the terminals may not result in a significant reduction in the high Q of the resonator. That is, the resistive losses in those shunted elements may not significantly reduce the efficiency of power transfer from Z0 to the inductive element or vice versa. Thus, for these components, there may be no need for strict requirements on low shunt losses and / or high voltage ratings. Generally, such reduced requirements can result in a wider selection of components that can be designed into high Q and / or high power impedance matching and resonator topologies. These reduced requirements are particularly helpful in expanding the variety of variable and / or high current and / or low series loss components that can be used in these high Q and / or high power impedance matching and resonator circuits.

[0343] Note that the above design principles can differently reduce the current and voltage on various elements, as they suggest using a network in series with Z0 (such as directly coupled B, C) or using a network in parallel with Z0 in different ways. The preferred topology for a given application may depend on the availability of low series loss / high current rating or low shunt loss / high voltage rating elements.

[0344] Group of fixed and variable elements for achieving fine tunability and reducing high rating requirements on variable elements Combination

[0345] Circuit topology

[0346] Obtaining variable circuit elements with satisfactory low losses and high voltage or current ratings may be difficult or too costly. In the present disclosure, we describe impedance matching topologies that can combine a combination of fixed and variable elements, such that a large voltage or current can be assigned to the fixed elements in the circuit (which are more likely to have appropriate voltage and current ratings) and the voltage and current rating requirements on the variable elements in the circuit can be alleviated.

[0347] A variable circuit element can have a tuning range that is greater than those required for a given impedance matching application, and, in those cases, it may be difficult to obtain a fine tuning resolution using only such a wide-range element. In the present disclosure, we describe impedance matching topologies that combine both fixed and variable elements such that a finer tuning resolution can be achieved with the same variable element.

[0348] Thus, a topology that uses a combination of both fixed and variable elements can produce two advantages simultaneously: a reduced voltage across or current through a sensitive tuning component in the circuit and a finer tuning resolution. Note that the maximum achievable tuning range can be related to the maximum reduction in voltage across or current through a tunable component in the circuit design.

[0349] Element topology

[0350] A single variable circuit element (as opposed to a network of elements discussed above) can be implemented by a topology that uses a combination of fixed and variable components connected in series or in parallel to achieve a reduced rating requirement for the variable component and a finer tuning resolution. This can be proven mathematically by the fact that:

[0351] If x |total| -x |fixed| +x |variable| ,

[0352] then Δx |total| / x |total| = Δx |variable| / (x |fixed| +x |variable| ),

[0353] and X variable / X total = X variable / (X fixed +X variable ), where x |subscript| is any element value (e.g., capacitance, inductance), X is voltage or current, and the “+ sign” represents the appropriate combination of elements (series addition or parallel addition). Note that the subscript format for x |subscript| is chosen to easily distinguish it from the radius of the area enclosed by a circular inductive element (e.g., x, x1, etc.).

[0354] Furthermore, by using different types of variable elements, this principle can be used to implement a certain type of variable electrical element (e.g., capacitance or inductance) if different types of variable elements are appropriately combined with other fixed elements.

[0355] In summary, a topology optimization algorithm can be applied that determines the required number, location, placement, type, and value of fixed and variable elements having a required tunable range as optimization constraints and determines the minimization of current and / or voltage on the variable elements as an optimization goal.

[0356] Examples

[0357] In the following schematic diagrams, we show different specific topology implementations for impedance matching of low-loss inductive elements and resonator designs for low-loss inductive elements. Additionally, for each topology, we indicate which of the above principles are used, give equations for the values of the variable elements that can be used to achieve the match, and the range of complex impedances that can be matched (described using inequalities and Smith charts). For these examples, we assume that Z0 is real, but the extension to characteristic impedances with non-zero imaginary parts is straightforward as it only implies small adjustments to the required values of the components of the matching network. We will use the convention of subscripting quantities with n to imply normalization (division by Z0) to Z0.

[0358] Figure 29 Two examples of transformer-coupled impedance matching circuits are shown, where the two tunable elements are the capacitance and the mutual inductance between two inductive elements. If we define X2 = ωL2 for (a) in Figure 29 and X2 = ωL2 - 1 / ωC2 for (b) in Figure 29 , and X ≡ ωL, then the required values of the tunable elements are:

[0359]

[0360] For Figure 29 the topology of (b), a particularly simple design can be to choose X2 = 0. In this case, these topologies can match impedances that satisfy the following inequalities:

[0361] R n > 0, X n > 0,

[0362] which is shown by the region enclosed by the thick line on the Smith chart of (c) in Figure 29 .

[0363] Given a preselected fixed M, the matching topology with tunable C2 mentioned above can be used instead.

[0364] Figure 30 Six examples (a) - (f) of direct-coupled impedance matching circuits (where the two tunable elements are capacitors) and six examples (h) - (m) of direct-coupled impedance matching circuits (where the two tunable elements are a capacitor and an inductor) are shown. ForFigure 30 The topologies of (a), (b), (c), (h), (i), (i) in [reference] may require common - mode signals at two terminals to maintain the voltage node of the resonator at the center of the inductive element and thus maintain a high Q. Note that these examples can be described as Figure 28 implementations of the general topology shown in (c) of [reference]. For Figure 30 the symmetric topologies of (d), (e), (f), (k), (1), (m) in [reference], it may be necessary to drive the two terminals anti - symmetrically (balanced drive) to maintain the voltage node of the resonator at the center of the inductive element and thus maintain a high Q. Note that these examples can be described as Figure 28 implementations of the general topology shown in (d) of [reference]. It will be recognized that the network of capacitors used herein generally can refer to any circuit topology including one or more capacitors, including but not limited to any circuit using capacitors specifically disclosed herein or any other equivalent or different circuit structure(s), unless explicitly specified or a different meaning is clear from the context.

[0365] Let us define separately for Figure 30 (a), (d), (h), (k) in [reference] that \(Z = R + j\omega L\), for Figure 30 (b), (e), (i), (1) in [reference] that \(Z = R + j\omega L+\frac{1}{j\omega C_3}\) and for Figure 30 (c), (f), (i), (m) in [reference] that \(Z=(R + j\omega L)\parallel(\frac{1}{j\omega C_3})\), where the symbol “‖” means “parallel combination of... ”, \(R_E\equiv Re\{Z\}\), \(X\equiv Im\{Z\}\). For Figure 30 (a) - (f) in [reference], the required value of the tunable element can be given by the following formula:

[0366]

[0367] And these topologies can match impedances that satisfy the following inequality:

[0368]

[0369] which is shown by the region enclosed by the thick line on the Smith chart of (g) in [reference]. Figure 30

[0370] For Figure 30 (h) - (m) in [reference], the required value of the tunable element can be given by the following formula:

[0371]

[0372] Figure 31 ​Shows three examples (a) - (c) of a direct - coupled impedance - matching circuit (where the two tunable elements are capacitors) and three examples (e) - (g) of a direct - coupled impedance - matching circuit (where the two tunable elements are a capacitor and an inductor). For Figure 31 Regarding the topologies of (a), (b), (c), (e), (f), (g) in Figure 28 the ground terminal is connected between two equal - value capacitors 2C1 (i.e., on the axis of symmetry of the main resonator) to maintain the voltage node of the resonator at the center of the inductive element and thus maintain a high Q. Note that these examples can be described as

[0373] Let's define respectively for Figure 31 (a) and (e) in Figure 31 Z = R + jωL, for Figure 31 (b) and (f) in Figure 31 Z = R + jωL+1 / jωC3, and for

[0374]

[0375] (c) and (g) in

[0376]

[0377] Z=(R + jωL)||(1 / jωC3), and then R≡Re{Z}, X≡Im{z}. Then, for Figure 31 (a) - (c) in

[0378] the required values of the tunable elements can be given by the following formula: Figure 31

[0379]

[0380] Figure 32 And these topologies can match impedances that satisfy the following inequality: Shows three examples (a) - (c) of a direct - coupled impedance - matching circuit (where the two tunable elements are capacitors) and three examples (e) - (g) of a direct - coupled impedance - matching circuit (where the two tunable elements are a capacitor and an inductor). For Figure 32 Regarding the topologies of (a), (b), (c), (e), (f), (g) inFigure 28 The implementation of the general topology shown in (e) in

[0381] Let us respectively define Z = R + jωL for (a) in Figure 32 in Figure 32 Z = R + jωL + 1 / jωC3 for (b) in Figure 32 in Figure 32 Z = (R + jωL) || (1 / jωC3) for (c) in

[0382]

[0383] where k is defined by M′ = -kL′, where L’ is the inductance of each half inductor loop and M’ is the mutual inductance between the two halves. These topologies can match impedances that satisfy the following inequality:

[0384]

[0385] which is shown by the area enclosed by the thick line on the Smith chart of (d) in Figure 32 in

[0386] For Figure 32 in (e) - (g), the required values of the tunable elements can be given by the following formula:

[0387]

[0388] In Figure 30 and 31 In the circuit of 32, the capacitor C2 or the inductor L2 (or two capacitors 2C2 or two inductors L2 / 2) is connected in series with the terminals and may not need to have very low series losses or withstand large currents.

[0389] Figure 33 Six examples (a) - (f) of a direct - coupled impedance - matching circuit (where the two tunable elements are capacitors) and six examples (h) - (m) of a direct - coupled impedance - matching circuit (where the two tunable elements are a capacitor and an inductor) are shown. For Figure 33 the topologies of (a), (b), (c), (h), (i), (i) in Figure 28 in Figure 33For the symmetric topologies of (d), (e), (f), (k), (l), (m) in [ ], it may be necessary to drive two terminals anti-symmetrically (balanced drive) to maintain the voltage node of the resonator at the center of the inductive element and thus maintain a high Q. Note that these examples can be described as Figure 28 Implementations of the general topology shown in (d) in [ ], where B and C are short-circuited and A is balanced.

[0390] Let us define separately for Figure 33 Z = R + jωL for (a), (d), (h), (k) in [ ], for Figure 33 Z = R + jωL + 1 / jωC3 for (b), (e), (i), (l) in [ ] and for Figure 33 Z = (R + jωL) || (1 / jωC3) for (c), (f), (j), (m) in [ ], and then R ≡ Re{Z}, X ≡ Im{Z}. Then, for Figure 33 (a) to (f) in [ ], the required values of the tunable elements can be given by the following formula:

[0391]

[0392] And these topologies can match impedances that satisfy the following inequalities:

[0393]

[0394] Which is shown by the region enclosed by the thick line on the Smith chart of (g) in [ ]. Figure 33 in [ ].

[0395] For Figure 35 (h) to (m) in [ ], the required values of the tunable elements can be given by the following formula:

[0396]

[0397] Figure 34 Shows three examples (a) to (c) of a direct-coupled impedance matching circuit (where two tunable elements are capacitors) and three examples (e) to (g) of a direct-coupled impedance matching circuit (where two tunable elements are a capacitor and an inductor). For the topologies of (a), (b), (c), (e), (f), (g) in [ ], the ground terminal is connected between two equal-value capacitors 2C2 (i.e., on the axis of symmetry of the main resonator) to maintain the voltage node of the resonator at the center of the inductive element and thus maintain a high Q. Note that these examples can be described as Figure 34 Implementations of the general topology shown in (e) in [ ]. Figure 28 in [ ].

[0398] Let us respectively define Z = R + jωL for (a) and (e) in Figure 34 , Z = R + jωL + 1 / jωC3 for (b) and (f) in Figure 34 , and Z = (R + jωL)||(1 / jωC3) for (c) and (g) in Figure 34 . Then, let R ≡ Re{Z} and X ≡ Im{Z}. For (a) to (c) in Figure 34 , the required values of the tunable elements can be given by the following equations:

[0399]

[0400] And these topologies can match impedances that satisfy the following inequalities:

[0401]

[0402] which is shown by the region enclosed by the thick line on the Smith chart of (d) in Figure 34 .

[0403] For (e) to (g) in Figure 34 , the required values of the tunable elements can be given by the following equations:

[0404]

[0405] Figure 35 Three examples of direct-coupled impedance matching circuits are shown, where two of the tunable elements are capacitors. For the Figure 35 topology, a ground terminal can be connected at the center of the inductive element to maintain the voltage node of the resonator at that point and thus maintain a high Q. Note that these examples can be described as Figure 28 implementations of the general topology shown in (e) of

[0406] Let us respectively define Z = R + jωL for (a) in Figure 35 , Z = R + jωL + 1 + jωC3 for (b) in Figure 35 , and Z = (R + jωL)||(1 / jωC3) for (c) in Figure 35 . Then, let R ≡ Re{Z} and X ≡ Im{Z}. Then, the required values of the tunable elements can be given by the following equations:

[0407]

[0408] where M′ = -kL′ is defined for and k, where L’ is the inductance of each half of the inductive element and M’ is the mutual inductance between the two halves. These topologies can match impedances that satisfy the following inequalities:

[0409]

[0410] wherein

[0411]

[0412] by the regions enclosed by the thick lines on three Smith charts shown in (d) in Figure 35 for k = 0, (e) in Figure 35 for k = 0.05, and (f) in Figure 35 for k = 1. Note that for 0 < k < 1, this topology can match two unconnected regions of the Smith chart.

[0413] In the Figure 33 , 34 , 35 circuit, the capacitor C2 or the inductor L2 (or one of two capacitors 2C2 or one of two inductors 2L2) is connected in parallel with the terminal, so it may not be necessary to have a high voltage rating. For two capacitors 2C2 or two inductors 2L2, both may not need to have a high voltage rating because approximately the same current flows through them, so they experience approximately the same voltage across their terminals.

[0414] For the topology shown using the capacitor C3 in Figures 30 - 35 , the use of the capacitor C3 can result in finer tuning of the frequency and impedance. For the Figures 30 - 35 topology, the use of a fixed capacitor C3 in series with an inductive element can ensure that most of the high inductive element voltage will be across this fixed capacitor C3, thus potentially reducing the voltage rating requirements for other elements of the impedance matching circuit (some of which can be variable). Whether such a topology is preferred depends on the availability, cost, and specifications of appropriately fixed, tunable components.

[0415] In all the above examples, a pair of equal - value variable capacitors without a common terminal can be implemented using a bank of capacitors or a varactor or diode bank or array that is biased and controlled to tune its value as an ensemble. A pair of equal - value variable capacitors with a common terminal can be implemented using a tunable butterfly capacitor or any other tunable or variable capacitor or a varactor or diode bank or array that is biased and controlled to tune its capacitance value as an ensemble.

[0416] Another criterion that can be considered when selecting an impedance matching network is the response of the network to frequencies different from the desired operating frequency. The signal generated in the external circuit to which the inductive element is coupled may not be monochromatic at the desired frequency, but rather periodic with the desired frequency, such as the drive signal of a switching amplifier or the reflected signal of a switching rectifier. In some such cases, it may be desirable to suppress the amount of higher-order harmonics entering the inductive element (e.g., to reduce the radiation of these harmonics from this element). Then, the selection of the impedance matching network can be one that sufficiently suppresses the amount of such harmonics entering the inductive element.

[0417] The impedance matching network can cause the impedance experienced by the external circuit at frequencies higher than the fundamental frequency to be high when the external periodic signal is a signal that can be considered to act as a voltage source signal (such as the drive signal of a class D amplifier with a series resonant load), so that almost no current flows through the inductive element at higher frequencies. Among Figures 30 - 35 the topologies, those using inductor L2 may be preferred because this inductor presents a high impedance at high frequencies.

[0418] The impedance matching network can cause the impedance experienced by the external circuit at frequencies higher than the fundamental frequency to be low when the external periodic signal is a signal that can be considered to act as a current source signal, so that almost no voltage is induced across the inductive element at higher frequencies. Among Figures 30 - 35 the topologies, those using capacitor C2 are preferred because this capacitor presents a low impedance at high frequencies.

[0419] Figure 36 Four examples of variable capacitances of networks using one variable capacitor and the remaining fixed capacitors are shown. Using these network topologies, fine tunability of the total capacitor value can be achieved. In addition, Figure 36 the topologies of (a), (c), and (d) in

[0420] Figure 37 can be used to reduce the voltage across the variable capacitor because most of the voltage can be distributed across the fixed capacitors.

[0421] Tunable elements such as tunable capacitors and tunable inductors can be mechanically tunable, electrically tunable, thermally tunable, etc. The tunable element can be a variable capacitor or inductor, varactor, diode, Schottky diode, reverse-biased PN diode, varactor array, diode array, Schottky diode array, etc. The diode can be a Si diode, GaN diode, SiC diode, etc. GaN and SiC diodes may be particularly attractive for high-power applications. The tunable element can be a switched capacitor bank, a switched mechanically tunable capacitor bank, a switched varactor array bank, a switched transformer-coupled inductor bank, etc. The tunable element can be a combination of the above-listed elements.

[0422] As described above, the efficiency of power transfer between coupled high-Q magnetic resonators can be affected by how closely the resonators are matched at the resonant frequency and how well their impedance matches the power source and power-consuming devices in the system. Since various external factors including the relative position of extraneous objects or other resonators in the system or changes in those relative positions can change the resonant frequency and / or input impedance of the high-Q magnetic resonators, it may be required that a tunable impedance network maintain a sufficient power transfer level in various environments or operating scenarios.

[0423] The capacitance value of the capacitor shown can be adjusted to adjust the resonant frequency and / or impedance of the magnetic resonator. The capacitor can be adjusted electrically, mechanically, thermally, or by any other known method. It can be adjusted manually or automatically, such as in response to a feedback signal. It can be adjusted to achieve certain power transfer efficiency or other operating characteristics between the power source and the power-consuming device.

[0424] The inductance value of the inductor and inductive elements in the resonator can be adjusted to adjust the frequency and / or impedance of the magnetic resonator. The inductance can be adjusted using a coupled circuit including tunable components such as tunable capacitors, inductors, and switches. The inductance can be adjusted using a transformer-coupled tuning circuit. The inductance can be adjusted by turning on and off different sections of the conductor in the inductive element and / or using ferromagnetic tuning and / or moving-iron tuning, etc.

[0425] The resonant frequency of the resonator can be adjusted to, or allowed to become, a lower or higher frequency. The input impedance of the resonator can be adjusted to, or allowed to become, a lower or higher impedance value. The amount of power delivered by the source and / or received by the device can be adjusted to, or allowed to become, a lower or higher power level. The amount of power delivered to the source and / or received by the device from the device resonator can be adjusted to, or allowed to become, a lower or higher power level. The resonator input impedance, resonant frequency, and power level can be adjusted based on power-consuming devices in the system and based on objects or materials near the resonator. The resonator input impedance, frequency, and power level can be adjusted manually or automatically and can be adjusted in response to feedback or control signals or algorithms.

[0426] A circuit element can be connected directly (i.e., by physical electrical contact) to the resonator, such as to the end of a conductor forming an inductive element and / or terminal connector. The circuit element can be soldered, fused, curled, adhered, clamped, or positioned closely to the conductor, or attached using a variety of electrical components, connectors, or connection techniques. A power source and power-consuming device can be connected to the magnetic resonator directly, indirectly, or inductively. Electrical signals can be supplied to or obtained from the resonator through terminal connections.

[0427] Those skilled in the art will understand that in the actual implementation of the principles described herein, there may be associated tolerances or acceptable variations in the values of actual components (capacitors, inductors, resistors, etc.) from the values calculated via the above equations, the values of actual signals (voltages, currents, etc.) from the values proposed by symmetry or antisymmetry, and the true geometric positions of points (such as connection points or 'axis' points of ground terminals near the center of inductive elements) from the positions proposed by symmetry or antisymmetry.

[0428] Embodiment

[0429] System block diagram

[0430] We disclose embodiments of high-Q resonators for wireless power transfer systems that can wirelessly power or charge devices at mid-range distances. The high-Q resonator wireless power transfer system can also be used to wirelessly power or charge devices with magnetic resonators that are different from any source resonator in the system in terms of size, shape, composition, arrangement, etc.

[0431] Figure 1 Figures (a), (b) therein show high-level diagrams of two exemplary dual-resonator systems. Each of these exemplary systems has a single source resonator 102S or 104S and a single device resonator 102D or 104D. Figure 38A high-level block diagram showing a system with some features highlighted. A device 2310 that is wirelessly powered or charged can include a device resonator 102D, device power and control circuitry 2304, etc., and one or more devices 2308 to which power, either DC or AC or both AC and DC, is transferred or of which it is composed. An energy or power source for the system can include source power and control circuitry 2302, a source resonator 102S, etc. One or more devices 2308 that receive power from the device resonator 102D and power and control circuitry 2304 can be any kind of device 2308 as described previously. The device resonator 102D and circuitry 2304 deliver power to one or more devices 2308 when in the vicinity of the source resonator 102S, which can be used to recharge the batteries of the one or more devices, directly power the one or more devices, or both simultaneously.

[0432] The source and device resonators can be separated by many meters, or they can be very close to each other, or they can be separated by any distance in between. The source and device resonators can be laterally or axially offset from each other. The source and device resonators can be directly aligned (no lateral offset), or they can be offset by several meters, or anything in between. The source and device resonators can be oriented such that the surface areas enclosed by their inductive elements are approximately parallel to each other. The source and device resonators can be oriented such that the surface areas enclosed by their inductive elements are approximately perpendicular to each other, or they can be oriented at any relative angle (0 to 360 degrees) between them.

[0433] The source and device resonators can be freestanding, or they can be enclosed in a housing, container, sleeve, or case. These different enclosures can be composed of almost any kind of material. For some applications, low loss tangent materials such as Teflon, REXOLITE, styrene, etc. may be preferred. The source and device resonators can be integrated into power sources and power consuming devices. For example, the source and device resonators can be integrated into a keyboard, computer mouse, display, cellular phone, etc. such that they are not visible outside of these devices. The source and device resonators can be separate from the power sources and power consuming devices in the system and can be connected to them by standard or custom wires, cables, connectors, or plugs.

[0434] The source 102S can be powered by many DC or AC voltage, current, and power sources including the USB port of a computer. The source 102S can be powered from the power grid, from a wall plug, from a battery, from a power supply, from an engine, from a solar cell, from a generator, from another source resonator, etc. The source power and control circuit 2302 can include circuits and components that isolate the source electronics from the power supply so that any reflected power or signal is not coupled out through the source input terminals. The source power and control circuit 2302 can include a power factor correction circuit and can be configured to monitor power usage for purposes of monitoring accounts, billing, control, and similar functions.

[0435] The system can operate bidirectionally. That is, the energy or power generated or stored in the device resonator can be fed back to a power supply including the power grid, a battery, any type of energy storage unit, etc. The source power and control circuit can include a power factor correction circuit and can be configured to monitor power usage for purposes of monitoring accounts, billing, control, and similar functions for the bidirectional energy flow. The wireless energy transfer system can enable or facilitate vehicle-to-grid (V2G) applications.

[0436] The source and the device can have a tuning capability that allows adjustment of the operating point to compensate for varying environmental conditions, perturbations, and load conditions that can affect the operation of the source and device resonators and the efficiency of the energy exchange. The tuning capability can also be used to multiplex power delivery to multiple devices, from multiple sources to multiple systems, to multiple repeaters or relays, etc. The tuning capability can be controlled manually or automatically and can be performed continuously, periodically, intermittently, or at a predefined time or interval.

[0437] For example, the device resonator and the device power and control circuit can be integrated into any part of the device, such as a battery compartment or a device cover or sleeve or motherboard, and can be integrated next to a standard rechargeable battery or other energy storage element. The device resonator can include a device field reformer that can shield any combination of the device resonator elements and the device power and control electronics from the electromagnetic field for power transfer and that can deflect the resonator field away from lossy device resonator elements and the device power and control electronics. Magnetic materials and / or high conductivity field reformers can be used to increase the perturbed quality factor Q of the resonator and the perturbed coupling factor of the source and device resonators.

[0438] The source resonator, source power, and control circuitry can be integrated into any type of furniture, structure, cushion, rug, picture frame (including digital picture frames, electronic frames), plug-in, electronic device, vehicle, etc. The source resonator can include a source field reformer that can shield any combination of source resonator elements and source power and control electronics from the electromagnetic field for power transfer and that can deflect the resonator field away from lossy source resonator elements and source power and control electronics. Magnetic materials and / or high conductivity field reformers can be used to increase the perturbed quality factor Q of the resonator and the perturbed coupling factor between the source and device resonators.

[0439] In Figure 39 Figure 5 shows a block diagram of a subsystem in an example of a wireless power transfer device. The power and control circuitry can be designed to convert alternating current power from device resonator 102D and convert it into steady direct current power suitable for powering or charging the device. The power and control circuitry can be designed to convert alternating current power at one frequency from the device resonator into alternating current power at a different frequency suitable for powering or charging the device. The power and control circuitry can include, or consist of, impedance matching circuit 2402D, rectifier circuit 2404, voltage limiting circuit (not shown), current limiting circuit (not shown), AC to DC converter 2408 circuitry, DC to DC converter 2408 circuitry, DC to AC converter 2408 circuitry, AC to AC converter 2408 circuitry, battery charging control circuit (not shown), etc.

[0440] The impedance matching 2402D network can be designed to maximize the power delivered between device resonator 102D and device power and control circuitry 2304 at a desired frequency. Impedance matching elements can be selected and connected such that a high Q of the resonator is maintained. The impedance matching circuit 2402D can be varied or tuned according to operating conditions to control the power delivered from the source to the device, from the source to the device resonator, between the device resonator and device power and control circuitry, etc. Power, current, and voltage signals can be monitored at any point in the device circuitry, and feedback algorithm circuits and techniques can be used to control components to achieve desired signal levels and system operation. Feedback algorithms can be implemented using analog or digital circuit techniques, and the circuitry can include a microprocessor, digital signal processor, field programmable gate array processor, etc.

[0441] Figure 39A third-party box shows a rectifier circuit 2404 that can rectify the AC voltage power from the device resonator into a DC voltage. In this structure, the output power of the rectifier 2404 can be the input to the voltage clamping circuit. The voltage clamping circuit (not shown) can limit the maximum voltage at the input to the DC-DC converter 2408D or the DC-AC converter 2408D. Generally, it may be desirable to use a DC-DC / AC converter with a large input voltage dynamic range so that large variations in device position and operation can be tolerated while sufficient power is delivered to the device. For example, the voltage level at the output of the rectifier can fluctuate and reach high levels as the power input to the device and the load characteristics change. When the device performs different tasks, it can have varying power requirements. The varying power requirements can cause high voltages at the output of the rectifier as the load characteristics change. Similarly, when the device and the device resonator are moved closer to and further away from the source, the power delivered to the device resonator can change, causing a change in the voltage level at the output of the rectifier. The voltage clamping circuit can prevent the voltage output from the rectifier circuit from exceeding a predetermined value within the operating range of the DC-DC / AC converter. The voltage clamping circuit can be used to extend the operating mode and range of the wireless energy transfer system.

[0442] The next box of the device's power and control circuit is the DC-DC converter 2408D that can generate a stable DC output voltage. The DC-DC converter can be a boost converter, a buck converter, a buck-boost converter, a single-ended primary inductor converter (SEPIC), or any other DC-DC topology that meets the requirements of a specific application. If the device requires AC power, the DC-DC converter can be replaced with a DC-AC converter, or a DC-AC converter can follow the DC-DC converter. If the device includes a rechargeable battery, the last box of the device power and control circuit can be a battery charging control unit that can manage the charging and maintenance of the battery in the battery-powered device.

[0443] The device power and control circuit 2304 can include a processor 2410D, such as a microcontroller, a digital signal processor, a field programmable gate array processor, a microprocessor, or any other type of processor. The processor can be used to read or detect the status or operating point of the power and control circuit and the device resonator. The processor can implement algorithms to interpret and adjust the operating point of the circuit, components, subsystems, and resonator. The processor can be used to adjust the impedance matching, resonator, DC-DC converter, DC-AC converter, battery charging unit, rectifier, etc. of the wirelessly powered device.

[0444] The processor may have a wireless or wired data communication link to other devices or sources and may transmit or receive data that can be used to adjust the operating point of the system. Any combination of power, voltage, and current signals at a single frequency or within a frequency range may be monitored at any point in the device circuitry. These signals may be monitored using analog or digital or a combination of analog and digital techniques. These monitored signals may be used in a feedback loop, or may be reported to the user in a variety of ways, or may be stored and retrieved at a later time. These signals may be used to warn the user of system failures, to indicate performance, or to provide feedback such as audio, visual, vibration, etc. to the user of the system.

[0445] Figure 40 Components of a source power and control circuit 2302 of an exemplary wireless power transfer system configured to supply power to a single or multiple devices are shown. The source power and control circuit 2302 of the exemplary system may be powered from an AC voltage source 2502 such as a household outlet, a DC voltage source such as a battery, a USB port of a computer, a solar cell, another wireless power source, etc. The source power and control circuit 2302 may drive the source resonator 102S with alternating current (such as with a frequency greater than 10 kHz and less than 100 MHz). The source power and control circuit 2302 may drive the source resonator 102S with alternating current at a frequency less than 10 GHz. The source power and control circuit 2302 may include a DC-to-DC converter 2408S, an AC-to-DC converter 2408S, or both an AC-to-DC converter 2408S and a DC-to-DC 2408S converter, an oscillator 2508, a power amplifier 2504, an impedance matching network 2402S, etc.

[0446] The source power and control circuit 2302 may be powered from multiple AC-to-DC voltage sources 2502 and may include AC-to-DC and DC-to-DC converters 2408S to provide the required voltage levels for the circuit components and the DC voltage for the power amplifier that may be used to drive the source resonator. The DC voltage may be adjusted and may be used to control the output power level of the power amplifier. The source may include a power factor correction circuit.

[0447] The output of the oscillator 2508 may be used as an input to the power amplifier 2504 that drives the source resonator 102S. The oscillator frequency may be tunable and the amplitude of the oscillator signal may be varied as a means of controlling the output power level from the power amplifier. The frequency, amplitude, phase, waveform, and duty cycle of the oscillator signal may be controlled by an analog circuit, by a digital circuit, or by a combination of analog and digital circuits. The control circuit may include a processor 2410S such as a microprocessor, a digital signal processor, a field programmable gate array processor, etc.

[0448] The impedance matching block 2402 of the source and the device tuner can be used to tune the source and the control circuit and the source and the device resonator. For example, the tuning of these circuits can be adjusted for the perturbation of the quality factor Q of the source or the device resonator caused by the change of the distance between the source and the device in the irrelevant object or system. The tuning of these circuits can also be used to sense the working environment, control the power flow to one or more devices, control the power to the wireless power network, reduce the power when an unsafe or fault mode condition is detected, etc.

[0449] Any combination of power, voltage, and current signals can be monitored at any point in the source circuit. Analog or digital or combined analog and digital techniques can be used to monitor these signals. These monitored signals can be used in the feedback circuit, or can be reported to the user in various ways, or can be stored and retrieved at a later time. These signals can be used to warn the user of system failures, to warn the user of exceeded safety thresholds, to indicate performance, or to provide feedback such as audio, visual, vibration, etc. to the user of the system.

[0450] The source power and control circuit can include a processor. The processor can be used to read the state or operating point of the power and control circuit and the source resonator. The processor can implement algorithms to interpret and adjust the operating points of the circuit, components, assemblies, subsystems, and resonators. The processor can be used to adjust the impedance matching, resonator, DC-DC converter, AC-DC converter, oscillator, power amplifier of the source, etc. The processor and adjustable components of the system can be used to implement a frequency and / or time power delivery multiplexing scheme. The processor can have a wireless or wired data communication link to the device and other sources, and can transmit or receive data that can be used to adjust the operating point of the system.

[0451] Although detailed and specific designs are shown in these block diagrams, it should be clear to those skilled in the art that many different modifications and rearrangements of the components and building blocks can be made within the spirit of the exemplary system. The partitioning of the circuit is outlined for illustrative purposes, and it should be clear to those skilled in the art that the components of each block can be further partitioned into smaller blocks or merged or shared. In an equivalent example, the power and control circuit can be composed of separate discrete components or a larger integrated circuit. For example, the rectifier circuit can be composed of discrete diodes, or diodes integrated on a single chip can be used. Many other circuits and integrated devices can be replaced in the design according to design criteria such as power or size or cost or application. Any part of the entire power and control circuit or the source or device circuit can be integrated into a single chip.

[0452] The impedance matching network of the device and / or source can include a capacitor or a network of capacitors, an inductor or a network of inductors, or any combination of capacitors, inductors, diodes, switches, resistors, etc. The components of the impedance matching network can be adjustable and variable, and can be controlled to affect the efficiency and operating point of the system. Impedance matching can be performed by controlling the connection points of the resonator, adjusting the permeability of the magnetic material, controlling the bias field, adjusting the excitation frequency, etc. Impedance matching can use or include any number of varactors, varactor arrays, switching elements, capacitor banks, switches and tunable elements, reverse-biased diodes, air-gap capacitors, compression capacitors, BZT electrically tunable capacitors, MEMS tunable capacitors, voltage-variable dielectrics, transformer-coupled tuning circuits, etc. or combinations thereof. The variable components can be mechanically tuned, thermally tuned, electrically tuned, piezoelectrically tuned, etc. The elements of the impedance matching can be silicon devices, gallium nitride devices, silicon carbide devices, etc. The elements can be selected to withstand high current, high voltage, high power, or any combination of current, voltage, and power. The elements can be selected as high-Q elements.

[0453] Source matching and tuning calculations can be performed on an external device through the USB port that powers the device. The device can be a computer, PDA, or other computing platform.

[0454] A demonstration system uses a source resonator coupled to a device resonator to wirelessly power / recharge a plurality of electronic consuming devices, including but not limited to laptop computers, DVD players, projectors, cellular phones, monitors, TVs, projectors, digital photo frames, lights, TV / DVD players, portable music players, circuit breakers, hand-held tools, personal digital assistants, external battery chargers, mice, keyboards, cameras, active loads, etc. Multiple devices can be powered simultaneously from a single device resonator. The device resonator can operate as a source resonator simultaneously. The power supplied to the device resonator can pass through additional resonators before being transferred to its intended device resonator.

[0455] Monitoring, Feedback and Control

[0456] The so-called port parameter measurement circuit can measure or monitor certain power, voltage, and current signals in a system, and a processor or control circuit can adjust certain settings or operating parameters based on those measurements. In addition to these port parameter measurement results, the amplitudes and phases of the voltage and current signals passing through the system and the amplitude of the power signal can also be accessed to measure or monitor system performance. The measured signals mentioned throughout this disclosure can be port parameter signals and any combination of voltage signals, current signals, power signals, etc. These parameters can be measured using analog or digital signals, can be sampled and processed, and can be digitized or converted using many known analog and digital processing techniques. The measured or monitored signals can be used in a feedback circuit or system to control the operation of the resonator and / or the system. Generally, we can refer to these monitored or measured signals as reference signals or port parameter measurements or signals, although sometimes they are also referred to as error signals, monitor signals, feedback signals, etc. We will refer to the signals used to control circuit elements (e.g., the voltage used to drive a voltage-controlled capacitor) as control signals.

[0457] In some cases, circuit elements can be adjusted to achieve a specified or predetermined impedance value for the source and device resonators. In other cases, the impedance can be adjusted to achieve a desired impedance value for the source and device resonators when the device resonator is connected to one or more power-consuming devices. In other cases, the impedance can be adjusted to reduce variations in the resonant frequency, variations in impedance or power levels due to movement of the source and / or device resonators, or changes in the environment near the resonator (such as movement of interacting materials or objects). In other cases, the impedance of the source and device resonators can be adjusted to different impedance values.

[0458] Coupled resonators can be made of different materials and can include different circuit, component, and structural designs, or they can be the same. Coupled resonators can include performance monitoring and measurement circuits, signal processing and control circuits, or a combination of measurement and control circuits. Some or all of the high-Q magnetic resonators can include tunable impedance circuits. Some or all of the high-Q magnetic resonators can include automatically controlled tunable impedance circuits.

[0459] Figure 41A magnetic resonator is shown having a port parameter measurement circuit 3802 configured to measure certain parameters of the resonator. The port parameter measurement circuit can measure the input impedance or reflected power of the structure. The port parameter measurement circuit can be included in source and / or device resonator designs and can be used to measure two-port circuit parameters such as S-parameters (scattering parameters), Z-parameters (impedance parameters), Y-parameters (admittance parameters), T-parameters (transmission parameters), H-parameters (hybrid parameters), ABCD-parameters (chain, cascade, or transmission parameters), etc. These parameters can be used to describe the electrical performance of a linear electrical network when applying various types of signals.

[0460] Different parameters can be used to characterize the electrical network in different operating or coupling schemes. For example, S-parameters can be used to measure matched and unmatched loads. Additionally, the amplitudes and phases of voltage and current signals within the magnetic resonator and / or within the source and device itself can be monitored at multiple points to provide system performance information. This information can be presented to the user of the system via a user interface such as a lamp, readout, horn, noise, vibration, etc., or it can be presented as a digital signal, or it can be provided to a processor in the system and used in the automatic control of the system. This information can be recorded, stored, or it can be used by an advanced monitoring and control system.

[0461] Figure 42 A circuit diagram of a magnetic resonator is shown in which a tunable impedance network can be implemented with a voltage-controlled capacitor 3902 or a capacitor network. Such an implementation can be adjusted, tuned, or controlled by a circuit such as a programmable voltage source 3908 and / or a computer processor. For example, the voltage-controlled capacitor can be adjusted in response to data acquired by the port parameter measurement circuit 3802 and processed by the measurement analysis and control algorithm subsystem 3904. A reference signal can be derived from the port parameter measurement circuit or other monitoring circuits designed to measure the degree of deviation from a desired system operating point. The measured reference signal can include voltage, current, complex impedance, reflection coefficient, power level, etc. at one or more points in the system and at one or more frequencies.

[0462] The reference signal can be provided to the measurement results analysis and control algorithm subsystem module, which can generate control signals to change the values of various components in the tunable impedance matching network. The control signals can change the resonant frequency and / or input impedance of the magnetic resonator or the power level supplied by the source or the power level absorbed by the device to achieve a desired power exchange between the power source / generator and the power-consuming device / load.

[0463] An adjustment algorithm can be used to adjust the frequency and / or impedance of a magnetic resonator. The algorithm can accept a reference signal regarding the degree of deviation from a desired operating point for the system and output a correction or control signal for a variable or tunable element of the control system related to the deviation such that the system returns towards one or more desired operating points. The reference signal for the magnetic resonator can be acquired while the resonator is exchanging power in a wireless power transfer system, or it can be switched out of the circuit during system operation. The corrections to the system can be applied or executed continuously, periodically, upon an over-limit condition, digitally, using analog methods, etc.

[0464] Figure 43 An end-to-end wireless power transfer system is shown. Both the source and the device can include a port measurement circuit 3802 and a processor 2410. The box labeled "Coupler / Switch" 4002 indicates that the port measurement circuit 3802 can be connected to the resonator 102 by a directional coupler or a switch, enabling measurement, adjustment, and control of the source and device resonators either in combination with the power transfer function or separately.

[0465] The port parameter measurement and / or processing circuit can be associated with some, any, or all of the resonators in the system. The port parameter measurement circuit can measure the source / device resonator response (i.e., the transmission and reflection between any two ports in the system) using a portion of the power transfer signal or an excitation signal within a certain frequency range, and can include amplitude and / or phase information. Such measurements can be implemented using a swept single-frequency signal or a multi-frequency signal. The signals used to measure and monitor the resonator and the wireless power transfer system can be generated by one or more processors and standard input / output (I / O) circuits including a digital-to-analog converter (DAC), an analog-to-digital converter (ADC), an amplifier, a signal generation chip, passive components, etc. The measurements can be implemented using a test device such as a network analyzer or using a custom circuit. The measured reference signal can be digitized by an ADC and processed using a custom algorithm running on a computer, a microprocessor, a DSP chip, an ASIC, etc. The measured reference signal can be processed in an analog control loop.

[0466] The measurement circuit can measure any set of two port parameters, such as S-parameters, Y-parameters, Z-parameters, H-parameters, G-parameters, T-parameters, ABCD-parameters, etc. The measurement circuit can be used to characterize the current and voltage signals at various points in the drive and resonator circuits, the impedance and / or admittance of the source and device resonators at opposite ends of the system (i.e., looking into the source resonator matching network towards the device ( Figure 43 "Port 1" in) and vice versa).

[0467] The device can measure relevant signals and / or port parameters, interpret the measurement data, and adjust its matching network to optimize the impedance seen into the coupled system independent of the actions of the source. The source can measure relevant port parameters, interpret the measurement data, and adjust its matching network to optimize the impedance seen into the coupled system independent of the actions of the device.

[0468] Figure 43 A block diagram of a source and a device in a wireless power transfer system is shown. The system can be configured to execute a control algorithm that actively adjusts the tuning / matching network in either or both of the source and device tuners to optimize performance in the coupled system. The port measurement circuit 3802S can measure signals in the source and convey those signals to the processor 2410. The processor 2410 can use the measured signals in a performance optimization or stabilization algorithm and generate control signals based on the outputs of those algorithms. The control signals can be applied to variable circuit elements in the tuning / impedance matching circuit 2402S to adjust operating characteristics of the source, such as power in the resonator and coupling to the device. The control signals can be applied to a power supply or generator to turn the power supply on or off, increase or decrease the power level, modulate the supply signal, etc.

[0469] The power exchanged between the source and the device can depend on a variety of factors. These factors can include the effective impedances of the source and device, the Qs of the source and device, the resonant frequencies of the source and device, the distance between the source and device, interactions of materials and objects near the source and device, etc. The port measurement circuit and processing algorithms can work concurrently to adjust resonator parameters to maximize power transfer, keep power transfer constant, controllably adjust power transfer, etc. under dynamic and steady-state operating conditions.

[0470] Some, all, or none of the source and device in a system implementation can include the port measurement circuit 3802S and the processing 2410 capabilities. Figure 44 An end-to-end wireless power transfer system is shown in which only the source 102S includes the port measurement circuit 3802 and the processor 2410S. In this case, the operating characteristics of the device resonator 102D can be fixed or can be adjusted by an analog control circuit and do not require control signals generated by a processor.

[0471] Figure 45 An end-to-end wireless power transfer system is shown. Both the source and the device can include the port measurement circuit 3802, but Figure 45In the system, only the source includes the processor 2410S. The source and the device can communicate with each other, and the adjustment of certain system parameters can respond to control signals that have been wirelessly transmitted between the source and the device, such as via the wireless communication circuit 4202. The wireless communication channel 4204 can be separated from the wireless power transfer channel 4208, or it can be the same. That is, the resonator 102 for power exchange can also be used to exchange information. In some cases, information can be exchanged by modulating components of the source or device circuit and sensing changes in port parameters or other monitoring devices.

[0472] Implementations where the source only includes the processor 2410 can be beneficial for multi-device systems where the source can handle all tuning and adjustment "decisions" and simply transmit control signals back to the device(s). This implementation can make the device smaller and cheaper because it can eliminate the need for a processor in the device or reduce the required functionality. A portion of the entire data set of each port measurement at each device can be sent back to the source microprocessor for analysis, and control instructions can be sent back to the device. These communications can be wireless communications.

[0473] Figure 46 A system for end-to-end wireless power transfer is shown. In this example, the source only includes the port measurement circuit 3802 and the processor 2410S. The source and the device can communicate with each other, such as via the wireless communication circuit 4202, and the adjustment of certain system parameters can respond to control signals that have been wirelessly transmitted between the source and the device.

[0474] Figure 47 A coupled electromagnetic resonator 102 is shown that can automatically adjust its frequency and impedance using a processor or computer. It can be implemented with reverse-biased diodes, Schottky diodes, and / or varactor elements included in a capacitor network shown as C1, C2, and C3. The circuit topologies that have been constructed, illustrated, and described here are exemplary and are not intended to limit the discussion of automatic system tuning and control in any way. Other circuit topologies can be utilized in conjunction with the measurement and control architectures discussed in this disclosure. Figure 47 The resonant frequency tuning and continuous impedance adjustment of the source and device resonators can be achieved using reverse-biased diodes, Schottky diodes, and / or varactor elements included in a capacitor network shown as C1, C2, and C3. The circuit topologies that have been constructed, illustrated, and described here are exemplary and are not intended to limit the discussion of automatic system tuning and control in any way. Other circuit topologies can be utilized in conjunction with the measurement and control architectures discussed in this disclosure.

[0475] The impedance and resonant frequency of the device and source resonators can be measured using a network analyzer 4402A - B or by other means described above and implemented using a controller (such as LabView 4404). The measurement circuit or device can output data to a computer or processor, which implements a feedback algorithm and dynamically adjusts the frequency and impedance via a programmable DC voltage source.

[0476] In one arrangement, a reverse-biased diode (Schottky, semiconductor junction, etc.) used to implement a tunable capacitor draws little DC current and can be reverse-biased by an amplifier with a large series output resistance. This implementation can enable the application of a DC control signal directly to a controllable circuit element in a resonator circuit while maintaining a very high Q in the magnetic resonator.

[0477] If different DC bias voltages are required, a DC-blocking capacitor as shown in Figure 47 can isolate the C2 bias signal from the C1 and / or C3 bias signals. The output of the bias amplifier can be bypassed to circuit ground to isolate the RF voltage from the bias amplifier and prevent non-fundamental RF voltages from being injected into the resonator. Alternatively, the reverse bias voltage for some capacitors can be applied through an inductive element in the resonator itself, since the inductive element acts as a short circuit at DC.

[0478] A port parameter measurement circuit can exchange signals with a processor (including any required ADC and DAC) that is part of a feedback or control system used to automatically adjust the resonant frequency, input impedance, energy stored or captured by the resonator, or power delivered from a source to a device load. The processor can also send control signals to tuning or adjustment circuits in or attached to the magnetic resonator.

[0479] When using a varactor or diode as a tunable capacitor, it may be beneficial to place fixed capacitors in parallel and in series with the tunable capacitor operating at a high reverse bias in the tuning / matching circuit. This arrangement can provide improvements in circuit and system stability and power handling capabilities by optimizing the operating voltage on the tunable capacitor.

[0480] A varactor or other reverse-biased diode can be used as a voltage-controlled capacitor. When higher voltage uniformity or a different capacitance than a single varactor component is required, a varactor array can be used. The varactors can be arranged as an N by M array connected in series and in parallel and treated as a single two-terminal component with characteristics different from the individual varactors in the array. For example, an N by N array of equal varactors (where the components in each row are connected in parallel and the components in each column are connected in series) can be used as a two-terminal device with the same capacitance as any single varactor in the array but with a voltage uniformity that is N times that of a single varactor in the array. Depending on the variability and differences in the parameters of the individual varactors in the array, additional bias circuits consisting of resistors, inductors, etc. may be required. Figure 48 A schematic diagram of a four by four array of unbiased varactors 4502 that can be suitable for magnetic resonator applications is shown in

[0481] Further improvement in system performance can be achieved through the judicious selection of fixed-value capacitor(s) that are set in parallel and / or series with the tunable (varactor / diode / capacitor) element. Multiple fixed capacitors switched into or out of the circuit can be able to compensate for variations in resonator Q, impedance, resonance frequency, power level, coupling strength, etc. that may be encountered in testing, developing, and operating a wireless power transfer system. A switched capacitor bank and other switched element groups can be used to ensure convergence to the operating frequency and impedance values required by the system design.

[0482] An exemplary control algorithm for isolating and coupling magnetic resonators can be described for the Figure 47 circuits and system elements shown. A control algorithm first tunes each source and device resonator loop "in isolation", i.e., the other resonators in the system are "shorted out" and "removed" from the system. In practice, a resonator can be "shorted out" by making it resonate at a much lower frequency (such as by maximizing the value of C1 and / or C3). This step effectively reduces the coupling between resonators, thus effectively reducing the system to a single resonator at a particular frequency and impedance.

[0483] Tuning a magnetic resonator in isolation involves changing the tunable elements in the tuning and matching circuit until the value measured by the port parameter measurement circuit is at its predetermined, calculated, or measured relative value. The expected value of the quantity measured by the port parameter measurement circuit can be selected based on the desired matching impedance, frequency, strong coupling parameter, etc. For the exemplary algorithms discussed below, the port parameter measurement circuit measures the S-parameters over a certain frequency range. The frequency range used to characterize the resonator can be a trade-off between the system performance information obtained and the calculation / measurement speed. For the algorithms described below, the frequency range can be approximately + / - 20% of the operating resonance frequency.

[0484] Each isolated resonator can be tuned as follows. First, short out the resonators that are not being adjusted. Next, minimize C1, C2, and C3 in the resonator that is being characterized and adjusted. In most cases, there will be fixed circuit elements in parallel with C1, C2, and C3, so this step does not reduce the capacitance value to zero. Next, start increasing C2 until the resonator impedance matches the "target" real impedance at any frequency within the above-measured frequency range. The initial "target" impedance can be less than the expected operating impedance for the coupled system.

[0485] C2 can be adjusted until the initial "target" impedance is achieved for the frequencies within the measurement range. Then, C1 and / or C3 can be adjusted until the loop resonates at the desired operating frequency.

[0486] Each resonator can be adjusted according to the above algorithm. After tuning each resonator in isolation, a second feedback algorithm can be applied to optimize the resonant frequency and / or input impedance for wireless power transfer in a coupled system.

[0487] The required adjustments to C1 and / or C2 and / or C3 in each resonator in the coupled system can be determined by measuring and processing the real and imaginary parts of the input impedance values from any one and / or both of the "ports" shown. For coupled resonators, changing the input impedance of one resonator can change the input impedance of the other resonator. The control and tracking algorithm can adjust one port to a desired operating point based on the measurement results at one port, and then adjust the other port based on the measurement results at the other port. These steps can be repeated until both sides converge to the desired operating point. Figure 43 The required adjustments to C1 and / or C2 and / or C3 in each resonator in the coupled system can be determined by measuring and processing the real and imaginary parts of the input impedance values from any one and / or both of the "ports" shown. For coupled resonators, changing the input impedance of one resonator can change the input impedance of the other resonator. The control and tracking algorithm can adjust one port to a desired operating point based on the measurement results at one port, and then adjust the other port based on the measurement results at the other port. These steps can be repeated until both sides converge to the desired operating point.

[0488] The S-parameters can be measured at both the source and device ports, and the following series of measurements and adjustments can be made. In the following description, Z0 is the input impedance and can be the target impedance. In some cases, Z0 is 50 ohms or close to 50 ohms. Z1 and Z2 are intermediate impedance values that can be the same as or different from Z0. Re{value} means the real part of the value and Im{value} means the imaginary part of the value.

[0489] The algorithms that can be used to adjust the input impedance and resonant frequency of two coupled resonators are described below:

[0490] 1) "Isolate" and adjust each resonator as described above.

[0491] 2) Adjust the source C1 / C3 until, at Re the following holds: Re{S11} = (Z1 + / - ε

[0492] - If Re{S11@ω o} > (Z1 + ε Re ), then decrease C1 / C3. If Re{S11@ω o} < (Zo - ε Re ), then increase C1 / C3.

[0493] 3) Adjust the source C2 until, at ω o the following holds: Im{S11} = (+ / - ε Im ):

[0494] - If Im{S11@ω o} > ε Im , then decrease C2. If Im{S11@ω o} < -ε Im , then increase C2.

[0495] 4) Adjust the devices C1 / C3 until, at ω o , Re{S22} = (Z2 + / - ε Re ) as follows:

[0496] - If Re{S22@ω o}} > (Z2 + ε Re ), then decrease C1 / C3. If Re{S22@ω o}} < (Zo - ε Re ), then increase C1 / C3.

[0497] 5) Adjust the device C2 until, at ω o , Im{S22} = 0 as follows:

[0498] - If Im{S22@ω o}} > ε Im , then decrease C2. If Im{S22@ω o}} < -ε Im , then increase C2.

[0499] We have achieved the operating system by repeating steps 1 to 4 until both (Re{S11}, Im{S11}) and (Re{S22}, Im{S22}) converge to ((Z0 + / - ε o ), (+ / - ε Re )) at ω Im ), where Z0 is the desired matching impedance and ω is the desired operating frequency. Here, ε Im represents the maximum deviation of the imaginary part from the desired value of 0 at ω o , and ε Re represents the maximum deviation of the real part from the desired value of Z0. It should be understood that ε Im and ε Re can be adjusted in order to increase or decrease the number of steps to convergence at the potential cost of system performance (efficiency). It should also be understood that steps 1 to 4 can be performed in a variety of sequences and in a variety of ways other than those described above (i.e., first adjust the source imaginary part and then the source real part; or first adjust the device real part and then the device imaginary part, etc.). The intermediate impedances Z1 and Z2 can be adjusted during steps 1 to 4 to reduce the number of steps required for convergence. The desired or target impedance value can be complex and can change over time or in different operating scenarios.

[0500] Steps 1 - 4 may be performed in any order, in any combination, and any number of times. Having described the above algorithm, modifications to the steps or the described embodiments may be apparent to those skilled in the art. In the same manner that impedance or admittance can be alternatively used to analyze a linear circuit to derive the same result, the above algorithm can be implemented with any equivalent linear network port parameter measurement (i.e., Z-parameters, Y-parameters, T-parameters, H-parameters, ABCD-parameters, etc.) or the other monitoring signals described above.

[0501] Due to the changes in the "loading" resistors Rs and Rd caused by the change in the mutual inductance M (coupling) between the source and the device resonators, it may be necessary to retune the resonators. The changes in the inductances Ls and Ld of the inductive elements themselves can be caused by the influence of external objects (as previously discussed) and may also require compensation. Such changes can be mitigated with the above adjustment algorithm.

[0502] A directional coupler or a switch can be used to connect the port parameter measurement circuit to the source resonator and the tuning / adjustment circuit. The port parameter measurement circuit can measure the properties of the magnetic resonator while it exchanges power in the wireless power transfer system, or it can be switched out of the circuit during system operation. The port parameter measurement circuit can measure the parameters and the processor can control certain tunable elements of the magnetic resonator at startup or at certain intervals or in response to changes in certain system operating parameters.

[0503] The wireless power transfer system can include a circuit that changes or tunes the impedance and / or resonant frequency of the source and device resonators. Note that although tuning circuits are shown in both the source and device resonators, the circuit can alternatively be included only in the source or the device resonator, or can be included only in certain source and / or device resonators. Also note that while we refer to the circuit as "tuning" the impedance and / or resonant frequency of the resonator, this tuning operation simply means that various electrical parameters such as the inductance or capacitance of the structure are being changed. In some cases, these parameters can be changed to achieve a specific predetermined value, in other cases, they can be changed in response to a control algorithm or to stabilize a changing target performance value. In some cases, the parameters can vary according to temperature, other sources or devices in the area, the environment, etc.

[0504] Application

[0505] For each of the listed applications, those skilled in the art will recognize that there are various ways to connect or integrate the resonator structures used to achieve wireless power transfer with the object being powered or receiving power. The resonator can be physically separated from the source and the device object. The resonator can use traditional inductance techniques or supply power to or remove power from the object through a direct electrical connection such as a wire or cable. The electrical connection can be an AC or DC power input port from the resonator output to the object. The electrical connection can be from the output power port of the object to the resonator input.

[0506] Figure 49 Shown is a source resonator 4904 physically separated from the power source and a device resonator 4902 physically separated from the device 4900 (a laptop computer in this illustration). Through an electrical connection, power can be supplied to the source resonator, and power can be directly obtained from the device resonator. Those skilled in the art will understand from the materials incorporated by reference that the shape, size, material composition, arrangement, position, and orientation of the above-mentioned resonators are provided by way of non-limiting examples, and a wide variety of changes to any and all of these parameters can be supported by the techniques disclosed for various applications.

[0507] Continuing with the example of the laptop computer, and without limitation, the device resonator can be physically connected to the device it is powering or charging. For example, as shown in a) of Figure 50 and b) of Figure 50 , the device resonator 5002 can be a) integrated into the housing of the device 5000 or b) attached by an adapter. The resonator 5002 can be visible on the device ((b) - (d) of Figure 50 ) or may not be ((a) of Figure 50 ). The resonator can be attached to the device, integrated into the device, plugged into the device, and so on.

[0508] The source resonator can be physically connected to the source supplying power to the system. As described above for the device and the device resonator, there are various ways to attach, connect to, or integrate the resonator with the power source. Those skilled in the art will understand that there are various ways to integrate the resonator in a wireless power transfer system, and the source and the device can utilize similar or different integration techniques.

[0509] Continuing again with the example of the laptop computer, and without limitation, the laptop computer can be powered, charged, or recharged by a wireless power transfer system. The source resonator can be used to supply wireless power, and the device resonator can be used to capture wireless power. The device resonator 5002 can be integrated into the edge of the screen (display), as shown in d) of Figure 50 and / or asFigure 50 As shown in c) in is integrated into the base of the laptop. The source resonator 5002 can be integrated into the base of the laptop, and the device resonator can be integrated into the edge of the screen. The resonator can also or alternatively be attached to the power supply and / or the laptop. The source and device resonators can also or alternatively be physically separated from the power supply and the laptop and can be electrically connected with cables. The source and device resonators can also or alternatively be physically separated from the power supply and the laptop and can be electrically coupled using conventional inductance techniques. Those skilled in the art will understand that although the previous examples relate to wireless power transfer for laptops, the methods and systems disclosed in this application can be suitably adapted for use with other electrical or electronic devices. Generally, the source resonator can be external to the source and supply power to the device resonator, which in turn supplies power to the device, or the source resonator can be connected to the source and supply power to the device resonator, which in turn supplies power to a part of the device, or the source resonator can be inside the source and supply power to the device resonator, which in turn supplies power to a part of the device, and any combination of these.

[0510] The systems or methods disclosed herein can provide power to electrical or electronic devices such as, but not limited to, telephones, cellular phones, cordless phones, smart phones, PDAs, audio devices, music players, MP3 players, radios, portable radios and players, wireless headsets, wireless earphones, computers, laptops, wireless keyboards, wireless mice, televisions, monitors, flat screen monitors, computer monitors, monitors embedded in furniture, digital photo frames, e-books (such as Kindle, e-ink books, magazines, etc.), remote control units (also known as controllers, game controllers, commanders, vote counters, etc. and used for remote control of multiple electronic devices such as televisions, video games, monitors, computers, audio-visual devices, lights, etc.), lighting devices, cooling devices, air circulation devices, purification devices, personal hearing aids, power tools, security systems, alarms, clocks, flashlights, sirens, sensors, loudspeakers, electronic locks, electronic keypads, lighting switches, other electrical switches, etc. Here, the term electronic lock is used to indicate a door lock that is electronically operated (e.g., having an electronic combination key, magnetic card, RFID card, etc.) rather than a mechanical key lock. Such locks are often battery-operated and have the risk of potentially stopping working and locking the user out when the battery runs out. This can be avoided in the case of charging or fully replacing the battery with a wireless power transfer implementation as described herein.

[0511] Here, the term lighting switch (or other electronic switch) is intended to indicate any switch (e.g., on the wall of a room) in one part of a room that turns on / off a device (e.g., a lighting fixture at the center of the ceiling) in another part of the room. To install such a switch with a direct connection, wires would have to be run all the way from the device to the switch. Once such a switch is installed at a particular location, it may be very difficult to move. Alternatively, a 'wireless switch' can be envisioned, where "wireless" means that the switch (on / off) command is transmitted wirelessly, but such switches traditionally require batteries to operate. Typically, having too many battery-operated switches around a house may be impractical because so many batteries would need to be replaced periodically. Thus, a wireless communication switch may be more convenient provided it is powered wirelessly. For example, there already exist battery-powered communication wireless doorbells, but the batteries in them still have to be replaced periodically. It is possible to make the remote doorbell button completely wireless, where the need to constantly replace the batteries may no longer be required. Note that here, the terms 'cordless' or 'wireless' or 'communication wireless' are used to indicate that there is a cordless or wireless communication device between the device and another electrical component, such as a base station for a cordless phone, a computer for a wireless keyboard, etc. Those skilled in the art will recognize that any electrical or electronic device can include a wireless communication device, and the systems and methods described herein can be used to add wireless power transfer to a device. As described herein, power to an electrical or electronic device can be delivered resonantly from an external or internal source resonator to the device or a part of the device. Wireless power transfer can significantly reduce the need to charge and / or replace the batteries of a device that enters the vicinity of the source resonator and thus can reduce the downtime, cost, and handling issues often associated with batteries.

[0512] The systems and methods described herein can power a lamp without the need for wired power or a battery. That is, the systems and methods described herein can power a lamp without a wired connection to any power source and can non-radiatively provide energy to the lamp over mid-range distances (such as distances of a quarter meter, one meter, three meters, etc.). The 'lamp' as used herein can refer to the light source itself, such as an incandescent bulb, a fluorescent bulb, a halogen lamp, a gas discharge lamp, a fluorescent lamp, a neon lamp, a high-intensity discharge lamp, a sodium vapor lamp, a mercury vapor lamp, an electroluminescent lamp, a light-emitting diode (LED) lamp, etc.; a lamp as part of a lamp fixture, such as a table lamp, a floor lamp, a chandelier, a track light, a recessed lamp fixture, etc.; a lamp fixture integrated with other functions, such as a luminaire / ceiling fan fixture and an illuminated picture frame, etc. Similarly, the systems and methods described herein can reduce the complexity of installing a lamp, such as by minimizing the installation of electrical wiring, and allow a user to place or install a lamp with minimal attention to a wired power source. For example, the lamp can be placed anywhere near the source resonator, where the source resonator can be mounted in a number of different positions relative to the position of the lamp, such as on the floor of an upper room (e.g., in the case of a chandelier, and especially when the upper room is an attic); on the wall of an adjacent room, on the ceiling of a lower room (e.g., in the case of a floor lamp); in a component within the room or in the infrastructure of the room as described herein; etc. For example, a lamp / ceiling fan combination is often installed in a master bedroom, and the master bedroom often has an attic above it. In such a case, a user can more easily install the luminaire / ceiling fan combination in the master bedroom, such as by simply mounting the lamp / ceiling fan combination to the ceiling and placing the source coil (plugged into the house wired AC power) in the attic above the installed fixture. In another example, the lamp can be an external lamp, such as a floodlight or a security lamp, and the source resonator is installed inside the structure. The way of installing lighting can be particularly beneficial to users of rented houses, as now they may be able to install lamps and such other electrical components without the need to install new electrical wiring. Control for the lamp can also be transmitted by near-field communication as described herein or by traditional wireless communication methods.

[0513] The systems and methods described herein can provide power from a source resonator to a device resonator embedded within or outside of a device component, such that the device component can be a conventional electrical component or fixture. For example, a chandelier can be designed or retrofitted with a device resonator integrated into the fixture, or the chandelier can be a conventional wired fixture plugged into a separate electrical device equipped with a device resonator. In an example, the electrical mechanism can be a wireless junction box designed to have a device resonator for receiving wireless power, for example, from a source resonator placed on the floor of an above room (e.g., attic), and which includes a number of conventional power outlets (outlets) powered from the device resonator. A wireless junction box mounted on the ceiling can now provide power to conventional wired electrical components on the ceiling (e.g., chandeliers, track lights, ceiling fans). Thus, a chandelier can now be installed on the ceiling without the need to run wires through the building's infrastructure. Such device resonators for conventional power outlet junction boxes can be used in a number of applications, including those designed for the interior or exterior of a building, made portable, made for use in a vehicle, etc. Wireless power can be transferred through common building materials such as wood, siding, insulation, glass, brick, stone, concrete, etc. The benefits of reduced installation costs, reconfigurability, and increased application flexibility can provide significant benefits to users as compared to conventional wired installations. Device resonators for conventional power outlet junction boxes can include a number of electrical components for facilitating power transfer from the device resonator to the conventional power outlet, such as power electronics for converting the specific frequency required for efficient power transfer into line voltage, power capture electronics for converting high frequency AC into a usable voltage and frequency (AC and / or DC), control for synchronizing the capture device and power output and ensuring consistent, safe, and maximally efficient power transfer, etc.

[0514] The systems and methods described herein can provide advantages for lights or electrical components operating in harsh environments such as wet, severe, controlled, etc. (which are outside and exposed to rain), in a pool / sauna / shower, in marine applications, in enclosed components, in explosion-proof enclosures, on external signs, in volatile environments such as harsh industrial environments from volatile vapors or airborne organics (such as in a grain silo or bakery), etc. For example, lights installed below the water level of a pool are normally difficult to wire up and need to be waterproof, although external wires are required. However, a pool light using the principles disclosed herein can be more easily made waterproof because external wires may not be required. In another example, an explosion-proof enclosure such as one containing volatile vapors may not only need to be airtight, but may also need to have all electrical contacts (which can generate sparks) sealed. Again, the principles disclosed herein can provide a convenient way to supply sealed electrical components for such applications.

[0515] The systems and methods disclosed herein can provide power to a game controller application, such as to a remote handheld game controller. These game controllers may traditionally have been powered individually by batteries, where the use and power distribution of the game controller has led to frequent replacement of the batteries, battery packs, rechargeable batteries, etc., which may be undesirable for consistent use of the game controller, such as during extended play sessions. A device resonator can be placed into the game controller, and a source resonator connected to a power source can be placed nearby. Additionally, the device resonator in the game controller can directly provide power to the game controller electronics without a battery; provide power to a battery, battery pack, rechargeable battery, etc., which then provides power to the game controller electronics; and so on. The game controller can utilize multiple battery packs, where each battery pack is equipped with a device resonator, such that the battery packs can be continuously recharged when in the vicinity of the source resonator, whether or not the game controller is plugged into a power source. The source resonator can be located in the main game controller device for the game, where power can be supplied to the main game controller device and the source resonator from an AC 'house' power source; in an AC power source in the form of an extension device, such as a source resonator integrated into an 'extension cord'; in a gaming chair, which is plugged into a wall AC, plugged into the main game controller device, powered by a battery pack in the gaming chair, or a combination of these; and so on. The source resonator can be set up and implemented in any configuration described herein.

[0516] The systems and methods disclosed herein can integrate a device resonator into a battery pack, such as a battery pack that can be interchanged with other battery packs. For example, certain portable devices may deplete electrical energy at a high rate, such that a user may need to have multiple interchangeable battery packs on hand for use, or the user may operate the device outside the range of a source resonator and require an additional battery pack to continue operation, such as power tools, portable lights, remote-controlled vehicles, etc. Using the principles disclosed herein provides not only a way for a device resonator-enabled battery pack to be recharged while in use and within range, but also a way for a battery pack that is not in use and is placed within the range of a source resonator to be recharged. In this way, the battery pack can be ready for use at any time when the user depletes the battery pack that is being used. For example, a user may be working with a cordless power tool where the demand at that time may be greater than what can be achieved by directly powering from a source resonator. In such a case, although the systems and methods described herein can provide charging power to a battery pack in use that is within range, the battery pack may still deplete because the power usage may have exceeded the recharge rate. Also, when using the device, the user may simply move in and out of range, or be completely outside the range. However, the user can place an additional battery pack near the source resonator, which has been recharged while not in use and is now fully charged for use. In another example, a user may be working with a power tool away from near a source resonator, but leave a supplementary battery pack charging near the source resonator, such as in a room with a portable source resonator or an extension cord source resonator, in the user's vehicle, in the user's toolbox, etc. In this way, the user may not have to worry about spending time and / or remembering to plug in their battery pack for future use. The user may simply replace the used battery pack with a charged one and place the used one near the source resonator to be recharged. The device resonator can be built into an enclosure having a known battery form factor and coverage area, and can replace traditional chemical batteries in known devices and applications. For example, the device resonator can be built into an enclosure having a mechanical size equivalent to that of an AA battery, AAA battery, D battery, 9V battery, laptop computer battery, cellular phone battery, etc. In addition to the device resonator, the enclosure can include a smaller "button battery" to store electrical energy and provide extended operation in terms of time or distance. In addition to or in place of the button battery, other energy storage devices can be integrated with the device resonator or any associated power conversion circuitry. These new energy packs can provide similar voltage and current levels as those provided by traditional batteries, but can consist of a device resonator, energy conversion electronics, small batteries, etc. These new energy packs can be more durable than traditional batteries because they can be more easily recharged and can be continuously recharged when they are located in a wireless power zone.Additionally, such energy packs can be lighter than conventional batteries, safer to use and store, operable over a wider range of temperatures and humidity, less damaging to the environment when discarded, etc. As described herein, when used in the wireless power regions as described herein, they can outlast the product's lifespan.

[0517] The systems and methods described herein can be used to power visual displays, such as in the case of a laptop computer screen, but more generally, will include a wide variety of displays utilized in current electrical and electronic components, such as in televisions, computer monitors, desktop computer monitors, laptop computer monitors, digital photo frames, e-books, mobile device displays (e.g., on phones, PDAs, gaming devices, navigation devices, DVD players), etc. Displays that can be powered by one or more of the wireless power transfer systems described herein can also include embedded displays, such as those embedded in electronic components (e.g., audio devices, household appliances, automotive displays, entertainment devices, cash registers, remote controls), in furniture, in building infrastructure, in vehicles, on the surface of an object (e.g., on the surface of a vehicle, building, clothing, sign, conveyance), etc. The display can be very small, having tiny resonator devices, such as in the smart cards as described herein, or very large, such as in a billboard. Displays powered using the principles disclosed herein can also be any of a variety of imaging technologies, such as liquid crystal displays (LCDs), thin film transistor LCDs, passive LCDs, cathode ray tubes (CRTs), plasma displays, projector displays (e.g., LCD, DLP, LCOS), surface conduction electron emission displays (SEDs), organic light emitting diodes (OLEDs), etc. The source coil configuration can include attachment to a main power source, such as building power, vehicle power, etc., via a wireless extension cord as described herein; attachment to a component power source, such as the base of an electrical component (e.g., the base of a computer, the cable box of a TV); an intermediate relay source coil, etc. For example, hanging a digital display on a wall can be very appealing, such as in the case of a digital photo frame that receives its information signal wirelessly or via a portable storage device, but can be made unsightly due to the need for an unseemly power cord. However, using a device coil embedded in the digital photo frame (such as being wound within a frame section) can allow the digital photo frame to be hung without any wires at all. Then the source resonator can be placed near the digital photo frame, such as in an adjacent room on the other side of the wall, and plugged directly into a conventional power outlet via, for example, a wireless extension cord as described herein, via a room-centered source resonator, etc.

[0518] The systems and methods described herein can provide wireless power transfer between different portions of an electronic device. Continuing with the example of a laptop computer, and without limitation, the screen of the laptop computer may require power from the base of the laptop computer. In such a case, electrical power has conventionally been transferred directly from the base of the laptop computer to the screen through a hinged connection portion of the laptop computer between the screen and the base. When using a wired connection, the wired connection may be subject to wear and tear and breakage, the design functionality of the laptop computer may be limited by the required direct electrical connection, the design aesthetics of the laptop computer may be limited by the required direct electrical connection, and so on. However, a wireless connection can be made between the base and the screen. In such a case, a device resonator can be placed in the screen portion to power the display, and the base can be powered by a second device resonator, by a conventional wired connection, by a hybrid of resonator-battery-direct electrical connection, and so on. In this way, not only can the reliability of the power connection be improved due to the removal of the physical wired connection, but also, since there are no physical wires associated with the hinge, the designer can be allowed to improve the functionality and / or aesthetic design of the hinge portion of the laptop computer. The laptop computer has again been used here to illustrate how the principles disclosed herein can improve the design of electrical or electronic devices, and should not be considered limiting in any way. For example, many other electrical devices having separate physical parts can benefit from the systems and methods described herein, such as a refrigerator having electrical features on a door, including an ice maker, a sensor system, a light, etc.; a robot having moving parts separated by joints; components in a vehicle's power system and vehicle doors, and so on. Those skilled in the art will recognize that the ability to provide power to a device from an external source resonator via a device resonator or to a portion of a device from an external or internal source resonator via a device resonator is widely applicable across the range of electrical and electronic devices.

[0519] The systems and methods disclosed herein can provide sharing of electrical power between devices (such as between a charged device and an uncharged device). For example, a charged device or appliance can act as a source and send a predetermined amount of energy, a dialed amount of energy, a requested and approved amount of energy, etc. to a nearby device or appliance. For example, a user may have a cellular phone and a digital camera, both of which are capable of transferring and receiving power through an embedded source and device resonator, and find that the battery level in one of the devices (e.g., the cellular phone) is low. The user can then transfer battery power from the digital camera to the cellular phone. The source and device resonators in these devices can use the same physical resonator for transmission and reception, use separate source and device resonators, one device can be designed to receive and transmit and the other designed to only receive, one device can be designed to only transmit and the other designed to only receive, and so on.

[0520] To prevent completely draining the device's battery, there can be a setting that allows the user to specify how much power the receiving device is entitled to receive. For example, it may be useful to impose a limit on the amount of power available to an external device and have the ability to turn off power transfer when the battery power drops below a threshold.

[0521] The systems and methods described herein can provide wireless power transfer to nearby electrical or electronic components associated with an electrical device, where a source resonator is in the electrical device and a device resonator is in the electronic component. The source resonator can also be connected to, plugged into, or attached to the electrical device, such as through a universal interface of the electrical device (e.g., USB interface, PC card interface), an additional power outlet, a universal attachment point, etc. For example, the source resonator can be inside the structure of a computer on a desk or integrated into an object, mat, etc., which is connected to the computer, such as into one of the computer's USB interfaces. In an example of a source resonator embedded in an object, mat, etc. and powered through a USB interface, the source resonator can be easily added to a user's desktop computer without the need to be integrated into any other electronic device, thus conveniently providing a wireless energy zone around which multiple electrical and / or electronic devices can be powered. The electrical device can be a computer, a lighting fixture, a dedicated source resonator electrical device, etc., and the nearby components can be those of computer peripherals, peripheral electronic components, infrastructure devices, etc., such as a computer keyboard, a computer mouse, a fax machine, a printer, a speaker system, a cellular phone, an audio device, an intercom, a music player, a PDA, a lamp, an electric pencil sharpener, a fan, a digital photo frame, a calculator, an electronic game, etc. For example, a computer system can be an electrical device with an integrated source resonator that utilizes a 'wireless keyboard' and a 'wireless mouse', where the term wireless is used herein to indicate that there is a wireless communication device between each device and the computer, and where each device must still contain a separate battery power source. As a result, the batteries will need to be replaced periodically, and in a large company, it may impose a significant burden on support staff for battery replacement, battery cost, and proper disposal of the batteries. Alternatively, the systems and methods described herein can provide wireless power transfer from the body of the computer to each of these peripheral devices, as described herein, including power not only to the keyboard and mouse but also to other peripheral components such as fax machines, printers, speaker systems, etc. The source resonator integrated into the electrical device can provide wireless power transfer to multiple peripheral devices, user devices, etc., such that the need to charge and / or replace the batteries for devices in the vicinity of the source resonator integrated electrical device is significantly reduced. The electrical device can also provide tuning or auto-tuning software, algorithms, devices, etc. for adjusting the power transfer parameters between the electrical device and the wirelessly powered device. For example, the electrical device can be a computer on a user's desktop, and the source resonator can be integrated into or plugged into the computer (e.g., through a USB connection), where the computer provides means for providing a tuning algorithm (e.g., through a software program running on the computer).

[0522] The systems and methods disclosed herein can provide wireless power transfer to nearby electrical or electronic components associated with institutional infrastructure components, where a source resonator is mounted in or on the institutional infrastructure component and a device resonator is in the electronic component. For example, the institutional infrastructure component can be a piece of furniture, a fixed wall, a movable wall or partition, a ceiling, a floor, and a source resonator attached or integrated into a table or desk (e.g., just below / above the surface, on the side, integrated into the tabletop or table legs), a mat placed on the floor (e.g., under a desk, placed on a desk), a mat on a garage floor (e.g., that will charge a car and / or devices in the car), in a parking lot / garage (e.g., on a post near a parking spot), a television (e.g., for charging a remote control), a computer monitor (e.g., that will power / charge a wireless keyboard, a wireless mouse, a cellular phone), a chair (e.g., for powering an electric blanket, a medical device, a personal health monitor), a picture, office furniture, common household appliances, etc. For example, the institutional infrastructure component can be a lighting fixture in an office cubicle, where both the source resonator and the lamp within the lighting fixture can be directly connected to the institutional wired power supply. However, with the source resonator now provided in the lighting fixture, those nearby electrical or electronic components connected to or integrated with the device resonator will not need to have any additional wired connections. Additionally, as described herein, the need to replace the battery of a device having a device resonator can be reduced.

[0523] Using the systems and methods described herein to supply power from a central location (such as from a source resonator in an electrical device, from an institutional infrastructure component, etc.) to electrical and electronic devices can minimize the electrical wiring infrastructure in the surrounding work area. For example, in a corporate office space, there are typically a large number of electrical and electronic devices that need to be powered by a wired connection. In the case of utilizing the systems and methods described herein, most of this wiring can be eliminated, saving the corporation installation costs, reducing the physical constraints associated with office walls having electrical wiring, minimizing the need for power outlets and power strips, etc. The systems and methods described herein can save the corporation money by reducing the electrical infrastructure associated with installation, reinstallation (e.g., reconfiguring an office space), maintenance, etc. In another example, the principles disclosed herein can allow for a power outlet to be wirelessly placed in the middle of a room. Here, the source can be placed on the ceiling of a basement below the location on the floor where it is desired to place the power outlet. The device resonator can be placed on the floor of the room directly above it. For the same reason, it is now substantially easier to install a new lighting fixture (or any other electrical device in this regard, such as a camera, a sensor, etc.) at the center of the ceiling.

[0524] In another example, the systems and methods described herein can provide power "through" a wall. For example, assume there is a power outlet in one room (e.g., on a wall), but one wants a power outlet in the adjacent room without having to call an electrician or drill through the wall or run wires around the wall, etc. Then a source resonator can be placed on the wall in one room, and a device resonator power outlet / pickup can be placed on the other side of the wall. This can power, for example, a flat screen TV or a stereo system (e.g., a person may not want ugly wires running up the wall in the living room, but does not mind having similar wires running along the wall in an adjacent room, such as a storage room or closet, or a room with furniture that obscures the wires running along the wall). The systems and methods described herein can be used to transfer power from an indoor source to various electrical devices outside the home or building without having to drill through the exterior wall or install conduits therein. In such a case, the device can be wirelessly powered outside the building without aesthetic or structural damage or the risks associated with drilling through walls and wallboards. Additionally, the systems and methods described herein can provide placement sensors to assist in placing an internal source resonator for an electrical component equipped with an external device resonator. For example, the homeowner can place a security light outside their home that includes a wireless device resonator and now needs to appropriately or optimally place the source resonator inside the home. A placement sensor that functions between the source and device resonators can make the placement better by indicating, such as with a visual indication, an audio indication, a display indication, etc., when the placement is good or how good it is. In another example, and in a similar manner, the systems and methods described herein can provide for the installation of devices on the roof of a home or building, such as radio transmitters and receivers, solar panels, etc. In the case of a solar cell, the source resonator can be associated with the panel, and power can be wirelessly transferred to a distribution board inside the building without having to drill through the roof. The systems and methods described herein can allow for the installation of electrical or electronic devices across the wall of a vehicle (such as through the ceiling) without having to drill, such as for cars, ships, airplanes, trains, etc. Thus, the integrity of the vehicle's wall can be maintained without drilling, thereby preserving the value of the vehicle, maintaining watertightness, eliminating the need for wiring, etc. For example, mounting a siren or light on the ceiling of a police car reduces the future resale value of the car, but with the systems and methods described herein, any light, horn, siren, etc. can be attached to the ceiling without having to drill.

[0525] The systems and methods described herein can be used to wirelessly transfer power from solar photovoltaic (PV) panels. PV panels with wireless power transfer capabilities can have several benefits, including simpler installation, greater flexibility, reliability, and weatherproof design. Wireless power transfer can be used to transfer power from PV panels to devices, houses, vehicles, etc. Solar PV panels can have a wireless source resonator that allows the PV panel to directly power a device enabled to receive wireless power. For example, the PV panel can be directly mounted on the roof of a vehicle, building, etc. The energy captured by the PV panel can be directly transferred wirelessly to devices inside the vehicle or under the roof of a building. Devices with resonators can wirelessly receive power from the PV panel. Wireless power from the PV panel can be used to transfer energy to a resonator coupled to the wired electrical system of a house, vehicle, etc., thus allowing conventional power distribution and supply to conventional devices without requiring any direct contact between the external PV panel and the internal electrical system.

[0526] With wireless power transfer, significantly simpler installation of rooftop PV panels can be achieved because power can be transferred wirelessly from the panel to a capture resonator in the house, eliminating all outdoor wiring, connectors, conduits, and any holes through the roof or walls of the structure. Wireless power transfer used with solar cells can have the benefit of reducing rooftop hazards because it eliminates the need for electrical workers to work on the roof to interconnect the panels, wires, and junction boxes. Installing solar panels integrated with wireless power transfer can require less skilled workers because fewer electrical connections need to be made. With wireless power transfer, less site-specific design may be required because the technology provides installers with the ability to optimize and position each solar PV panel individually, significantly reducing the need for expensive engineering and panel layout services. Careful balancing of the solar load on each panel may be required, and dedicated DC wiring layouts and interconnections are not needed.

[0527] For rooftop or wall mounting of PV panels, the capture resonator can be installed on the underside of the roof, inside the wall, or in any other easily accessible internal space within one or both of the bottoms of the solar PV panel. Figure 51FIG. shows a diagram showing a possible general rooftop PV panel installation. A variety of PV solar collectors can be installed on top of the roof, where a wireless power capture coil is installed inside the building under the roof. The resonator coil in the PV panel can wirelessly transfer its energy through the roof to the wireless capture coil. The captured energy from the PV cells can be collected and coupled to the electrical system of the house to power electrical and electronic devices or, when more power is generated than needed, coupled to the power grid. Energy is captured from the PV cells without requiring holes or wires to penetrate the roof or walls of the building. Each PV panel can have a resonator coupled to a corresponding resonator on a vehicle or inside a building. Multiple panels can transfer or collect power to one or more designated panels with resonators coupled to a vehicle or the interior of a house using wireless power transfer between them. The panels can have wireless power resonators on their sides or in their perimeters, which can couple to resonators in other similar panels, allowing power to be transferred from panel to panel. An additional bus or connection structure can be provided that wirelessly couples power from multiple panels on the outside of a building or vehicle and transfers the power to one or more resonators on the inside of the building or vehicle.

[0528] For example, as Figure 51 shown, a source resonator 5102 can be coupled to a PV cell 5100 mounted above a roof 5104 of a building. A corresponding capture resonator 5106 is placed inside the building. Then, the solar energy captured by the PV cell can be transferred from the source resonator 5102 outside to the device resonator 5106 inside the building without having direct holes and connections through the building.

[0529] Each solar PV panel with wireless power transfer can have its own inverter, significantly improving the economic efficiency of these solar systems by individually optimizing the power generation efficiency of each panel, supporting a mix of panel sizes and types in a single installation, including single-panel "pay-as-you-go" system expansion. The reduction in installation costs can make single panels economical for installation. The need for panel string design and the careful positioning and orientation of multiple panels is eliminated, as well as the single point of failure of the system.

[0530] Wireless power transfer in PV solar panels enables more solar deployment scenarios, as weatherproof solar PV panels eliminate the need to drill holes for wiring through sealed surfaces such as car roofs and boat decks, and eliminate the requirement to mount the panels in fixed positions. With wireless power transfer, PV panels can be deployed temporarily and then moved or removed without leaving permanent alterations to the surrounding structure. They can be placed in the yard on sunny days and moved back and forth with the sun, or ...

Claims

1. A wireless power transmitter, comprising: A first inductive element, the first inductive element including a planar coil occupying a first plane and having first and second terminals; A first capacitive element, the first capacitive element being connected to the first terminal; An inverter configured to provide an oscillating voltage signal to the first inductive element through the first capacitive element such that the first inductive element will generate an oscillating magnetic field to transmit power; And A magnetic material layer, the magnetic material layer centered on the planar coil in a second plane parallel to the first plane, wherein the magnetic material layer is a shield.

2. The wireless power transmitter according to claim 1, further comprising: A second capacitive element, the second capacitive element being connected to the second terminal; A third capacitive element, the third capacitive element being connected to the first and second capacitive elements to form a first conduction path between the first and third capacitive elements and to form a second conduction path between the second and third capacitive elements; A second inductive element, the second inductive element being connected between a first output terminal of the inverter and the first conduction path; And A third inductive element, the third inductive element being connected between a second output terminal of the inverter and the second conduction path.

3. The wireless power transmitter according to claim 1, wherein the inverter includes a class-D inverter.

4. The wireless power transmitter according to claim 3, wherein the inverter includes a full-bridge inverter.

5. The wireless power transmitter according to claim 3, wherein the inverter includes a half-bridge inverter.

6. The wireless power transmitter according to claim 1, wherein the shield is separated from the planar coil by a gap in a direction perpendicular to the first plane.

7. The wireless power transmitter according to claim 6, wherein the planar coil is a first planar coil, and the transmitter further includes: A second planar coil, and A third planar coil, wherein The second and third planar coils are not coplanar with the first coil, and The second and third planar coils are coplanar with each other in a third plane parallel to the first plane.

8. The wireless power transmitter according to claim 7, wherein the second and third planar coils are located between the first planar coil and the shield.

9. The wireless power transmitter according to claim 7, wherein the shield extends to the edges of the second and third coils in a direction parallel to the first plane.

10. The wireless power transmitter according to claim 1, wherein the shield extends beyond the edge of the coil in a direction parallel to the first plane.

11. The wireless power transmitter according to claim 10, wherein the portion of the shield extending beyond the edge of the coil further extends into the first plane.

12. The wireless power transmitter according to claim 1, wherein: The inverter includes four switches, The transmitter further includes a control input, In a first operating mode, the control input causes the inverter to operate two switches in a half-bridge mode; And In the second operating mode, the control input causes the inverter to operate all four switches in a full-bridge mode.

13. A wireless power receiver, comprising: A first inductive element, the first inductive element including a planar coil occupying a first plane and having first and second terminals; A first capacitive element, the first capacitive element being connected to the first terminal; A second capacitive element, the second capacitive element being connected to the first capacitive element at a third terminal and the second capacitive element being connected to the second terminal; A rectifier, the rectifier being connected to the second terminal and the third terminal such that an oscillating magnetic field will induce a current in the first inductive element, resulting in a voltage at the input of the rectifier; A communication modulation circuit; And A magnetic material layer, the magnetic material layer centered on the planar coil in a second plane parallel to the first plane.

14. The receiver according to claim 13, wherein the rectifier includes a class-D rectifier.

15. The receiver according to claim 14, wherein the rectifier includes a full-bridge rectifier of four diodes.

16. The receiver according to claim 13, further comprising a buck converter connected to the output of the rectifier.

17. The receiver according to claim 13, further comprising a linear regulator connected to the output of the rectifier.

18. The receiver according to claim 13, wherein the communication modulation circuit includes a first switch coupled between the second terminal and ground and a second switch coupled between the third terminal and ground.

19. The receiver according to claim 18, wherein the first and second switches are connected to each other at a node, and a third node is connected to ground through a resistor.

20. The receiver according to claim 13, wherein the communication modulation circuit includes a switch coupled to the output of the rectifier.

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