A unipolar modulation reactive zero-crossing current distortion control device and control method

Abstract

The present invention provides a unipolar modulated reactive zero-crossing current distortion control device and control method. The control device includes a totem pole topology, a control module, and a drive module. The control method includes: the control module receives a sampling signal for the totem pole topology and, based on the sampling signal, outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency; the drive module outputs a power-frequency transistor drive signal based on the first high-frequency PWM signal and a high-frequency transistor drive signal based on the second high-frequency PWM signal; the high-frequency transistor in the totem pole topology is driven according to the high-frequency transistor drive signal, and the power-frequency transistor is driven according to the power-frequency transistor drive signal. The present invention outputs a high-frequency PWM signal of the same frequency based on the same sampling signal, achieves synchronous commutation of the power-frequency transistor and the high-frequency transistor, and improves current distortion at the zero-crossing point, particularly current distortion during reactive power output by a grid-connected inverter.

Description

Technical Field

[0001] The present invention belongs to the field of circuit electronics, and in particular relates to a unipolar modulation reactive zero-crossing current distortion control device and a control method. Background Art

[0002] The commonly used structure of the totem pole topology is a pair of high-frequency switching tubes (referred to as high-frequency tubes) and a pair of power-frequency switching tubes (referred to as power-frequency tubes). Its basic topology is as follows: Figure 1 As shown, Q1 and Q2 are high-frequency tubes, Q3 and Q4 are power-frequency tubes, and their basic working principle is: the grid voltage polarity is as follows: Figure 2 and 3 As shown, the power frequency tube Q3 is turned on, and the circuit has two main working modes, mode 1 ( Figure 2 ) and Mode 2 ( Figure 3 ), unipolar modulation is achieved through high-frequency switching of Q1 and Q2. When the grid voltage reverses, Q4 turns on, and the other principles remain the same. In the inverter mode, a DC power supply is connected to the Vbus capacitor. The switching mode is the same as for rectification, but the current is reversed. This is a simple unipolar modulation method and will not be further explained here.

[0003] For the control of switching tubes, the commonly used method is to use DSP digital control to emit high-frequency PWM waves for high-frequency tubes Q1 and Q2. The high-frequency PWM wave is generated by comparing the modulation wave generated by the system loop control with the carrier, that is, unipolar modulation. The industrial frequency PWM wave is flipped by detecting the actual voltage crossing zero.

[0004] The disadvantage of unipolar modulation is that the current will be distorted at the zero crossing point, which is usually caused by the wave generation of the power frequency tube at the zero crossing point. Since the wave generation of the high-frequency tube and the power frequency tube are not synchronously controlled, the control of the power frequency tube is generally controlled according to the actual grid voltage polarity, turning on within half a cycle and turning off when the grid voltage crosses zero; the high-frequency tube will be controlled according to the modulation wave calculated by the control module, and there will always be a delay between the two. Under unipolar modulation, the PWM duty cycle of the high-frequency tube will have a sudden change from 0 to 1 or 1 to 0 near the zero crossing point, and this zero crossing is the zero crossing point of the modulation wave, not the zero crossing point of the grid voltage. This requires the power frequency tube to follow the sudden change in the duty cycle of the high-frequency tube with the minimum delay and make a corresponding flip. For example Figure 4 As shown, if there is a slight delay, the switching mode is changed by Figure 5 Convert to Figure 6 As shown in the figure, the positive and negative marks indicate the positive direction of voltage and current. The actual voltage and current polarity shall be based on the waveform diagram, and the same shall apply to the following text. Q1 duty cycle becomes 1, Q2 duty cycle becomes 0, and since Q1 has a large duty cycle, it will cause the bus voltage to be applied to the inductor, the current increases in the reverse direction, and the current flow direction is converted to Figure 7 As shown in the figure, there may be distortion of zero-crossing current, such as Figure 4 shown. Summary of the Invention

[0005] The technical problem to be solved by the present invention is to provide a unipolar modulation reactive zero-crossing current distortion control device and control method, aiming to solve the problem of current distortion at the voltage zero-crossing point when the grid-connected inverter outputs reactive power.

[0006] To solve the above technical problems, the present invention is implemented as follows: In a first aspect, the present invention provides a unipolar modulation reactive zero-crossing current distortion control device, comprising: a totem pole topology, a control module, and a drive module; wherein:

[0007] The control module includes a loop calculation unit, a modulation wave calculation unit, and a comparator; the loop calculation unit is used to receive a sampling signal of the totem pole topology and output a quasi-sine wave according to the sampling signal; the modulation wave calculation unit is used to calculate a power frequency tube modulation wave and a high-frequency tube modulation wave according to the quasi-sine wave; the comparator is used to compare the power frequency tube modulation wave with a carrier to generate a first high-frequency PWM signal, and compare the high-frequency tube modulation wave with the same carrier to generate a second high-frequency PWM signal;

[0008] The driving module includes a high-frequency driving filter circuit and an industrial frequency driving filter circuit. The high-frequency driving filter circuit is used to receive the second high-frequency PWM signal and output a high-frequency tube PWM signal to the driving circuit connected to the high-frequency driving filter circuit, and the driving circuit outputs the high-frequency tube driving signal. The industrial frequency driving filter circuit is used to receive the first high-frequency PWM signal and output a industrial frequency tube PWM signal to the driving circuit connected to the industrial frequency driving filter circuit, and the driving circuit outputs the industrial frequency tube driving signal. The totem pole topology includes an AC input / output, a high-frequency tube, an industrial frequency tube, a PFC inductor, and a DC input / output. The high-frequency tube is used to be driven according to the high-frequency tube driving signal, and the industrial frequency upper tube is used to be driven according to the industrial frequency tube driving signal.

[0009] A second aspect of the present invention provides a unipolar modulation reactive zero-crossing current distortion control method, which is applied to the unipolar modulation reactive zero-crossing current distortion control device as described above. The control method includes:

[0010] The control module receives a sampling signal of the totem pole topology and outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal;

[0011] The driving module outputs a power-frequency tube driving signal according to the first high-frequency PWM signal, and outputs a high-frequency tube driving signal according to the second high-frequency PWM signal;

[0012] The high-frequency tube in the totem pole topology is driven according to the high-frequency tube driving signal, and the power-frequency tube in the totem pole topology is driven according to the power-frequency tube driving signal.

[0013] The unipolar modulated reactive power zero-crossing current distortion control device and control method provided in the present invention have the following advantages over the prior art: the control device includes a totem pole topology, a control module, and a drive module; the control method provided in the present invention includes the control module receiving a sampling signal for the totem pole topology and outputting a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal; the drive module outputting a power-frequency transistor drive signal based on the first high-frequency PWM signal and a high-frequency transistor drive signal based on the second high-frequency PWM signal; the high-frequency transistors in the totem pole topology are driven according to the high-frequency transistor drive signal, and the power-frequency transistors in the totem pole topology are driven according to the power-frequency transistor drive signal. The present invention outputs a high-frequency PWM signal of the same frequency based on the same sampling signal, and ultimately outputs a high-frequency transistor drive signal and a power-frequency transistor drive signal to drive the high-frequency transistor and the power-frequency transistor, thereby achieving synchronous commutation of the high-frequency transistor and the power-frequency transistor, and improving current distortion at the zero-crossing point, particularly current distortion when the grid-connected inverter outputs reactive power. The technical solution proposed by the present invention does not increase the control difficulty and does not occupy computing resources. At the same time, it can cope with relatively extreme input conditions and ensure that no large current spikes will occur during synchronous switching. BRIEF DESCRIPTION OF THE DRAWINGS

[0014] Figure 1 This is a schematic diagram of a standard totem pole topology;

[0015] Figure 2 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 1 ;

[0016] Figure 3 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 2 ;

[0017] Figure 4 It is a schematic diagram of the zero-crossing current distortion structure;

[0018] Figure 5 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 3 ;

[0019] Figure 6 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 4 ;

[0020] Figure 7 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 5 ;

[0021] Figure 8 It is a schematic diagram of the switch state when the inverter outputs reactive power;

[0022] Figure 9 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 6 ;

[0023] Figure 10 This is a schematic diagram of the working mode of the totem pole topology circuit Figure 7 ;

[0024] Figure 11 This is a waveform diagram of the actual circuit test;

[0025] Figure 12 A schematic diagram of a framework of a control device provided in a first embodiment of the present invention;

[0026] Figure 13 A schematic diagram showing the connection between the control module and the drive module provided in the first embodiment of the present invention;

[0027] Figure 14 A schematic flow chart of a control method provided in a second embodiment of the present invention;

[0028] Figure 15 Schematic diagram of the waveform of the modulation wave output by the control module in the second embodiment of the present invention;

[0029] Figure 16 This is a schematic diagram of the power frequency tube control waveform in the second embodiment of the present invention;

[0030] Figure 17 Schematic diagram of the switch mode in the second embodiment of the present invention Figure 1 ;

[0031] Figure 18 Schematic diagram of the switch mode in the second embodiment of the present invention Figure 2 ;

[0032] Figure 19 Schematic diagram of the switch mode in the second embodiment of the present invention Figure 3 ;

[0033] Figure 20 Schematic diagram of actual output waveform in the second embodiment of the present invention;

[0034] Figure 21 This is the actual test waveform diagram of the present invention when the power factor is 1;

[0035] Figure 22 This is the actual test waveform of the present invention under a power factor of -0.8;

[0036] Figure 23 This is the actual test waveform diagram of the present invention when the power factor is 0.8. DETAILED DESCRIPTION

[0037] In order to make the purpose, technical solutions and advantages of the present invention more clearly understood, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not intended to limit the present invention.

[0038] In related technologies, in order to solve the problem of current distortion, the following methods are adopted:

[0039] (1) The power frequency tube sets a dead time and increases the dead zone at the zero crossing point. Disadvantages: When there is no reactive power, the commutation of the power frequency tube and the high frequency tube is not synchronized, resulting in current distortion.

[0040] (2) Use a phase-locked loop (PLL) to extract the grid phase and use the phase to determine zero crossings. Disadvantages: In poor grid conditions, especially those with high harmonic content, the zero crossing point oscillates widely, and the voltage may cross zero multiple times. However, the PLL phase is fixed, meaning the power frequency tube does not commutate according to the actual voltage, resulting in high current.

[0041] (3) Sampling the grid voltage and inputting it into the zero-crossing comparator as the driving square wave signal for the power-frequency diode. Disadvantages: A very accurate sampling circuit is required, and the power-frequency diode and the high-frequency diode are controlled separately. The high-frequency diode is based on the modulation wave, while the power-frequency diode is based on the grid voltage, which can cause the two to be out of sync at the zero-crossing point.

[0042] (4) Preset the switching sequence at the zero crossing. That is, the switching tubes are controlled according to the preset switching sequence instead of the loop calculation value at the zero crossing. Disadvantages: The control at the zero crossing is not flexible and it is difficult to cope with different input conditions and load conditions.

[0043] (5) Detect the current signal at the voltage zero-crossing point and add a third control loop, in addition to the voltage outer loop and the current inner loop, to compensate for the zero-crossing current. Disadvantages: The control is more complex and the synchronization of the power frequency tube and the high frequency tube is not processed, which will lead to current distortion when transmitting reactive power.

[0044] The above methods for solving zero-crossing distortion all control the power-frequency tube and the high-frequency tube separately. The principle is to control the power-frequency tube as accurately as possible at the zero-crossing point of the grid voltage. However, it is impossible to guarantee the synchronous flipping of the two. Under rectification conditions, due to the presence of the body diode of the switching tube, the current has a loop flow. Under inverter conditions, it will be cut off, resulting in a current drop. This phenomenon is especially obvious when outputting reactive power to the grid: Figure 8 It shows a possible switching state when the inverter outputs reactive power. When Q3 and Q2 are turned on, they transmit energy to the grid and output a certain amount of reactive power. The switching mode is as follows: Figure 9 As shown, when Figure 8 When the high frequency and power frequency of the middle switch tube are inconsistent, the current direction remains unchanged. The only freewheeling circuit is as follows Figure 10 As shown, at this time the energy of the inductor is sent to the capacitor until the current is 0.

[0045] From the inductor current formula we know that

[0046]

[0047] Among them, U is the voltage applied to the two ends of the inductor, ΔT is the voltage maintenance time, and L is the inductance. Take the values ​​in an actual switching power supply application, the DC power supply voltage is 400V, the inductance is 200uH, and ΔT is 10us. In a very short time, the current change can reach 20A, so the synchronous commutation of the power frequency tube and the high frequency tube plays a critical role in the zero-crossing current distortion, especially the current distortion when transmitting reactive power.

[0048] Example 1

[0049] like Figure 12 The figure shows a schematic diagram of the framework structure of the control device provided in this embodiment. In order to solve the problem of current distortion when transmitting reactive power, the present invention provides a unipolar modulation reactive zero-crossing current distortion control device in the first embodiment, including: a totem pole topology, a control module and a drive module.

[0050] In this embodiment, the totem pole topology shown is a standard totem pole topology infrastructure, such as Figure 1 The totem pole topology shown in the figure includes AC input / output terminals, a PFC inductor, a high-frequency transistor, a power-frequency transistor, and a current input / output terminal. The high-frequency transistor includes a high-frequency upper transistor and a high-frequency lower transistor, and the power-frequency transistor includes a power-frequency upper transistor and a power-frequency lower transistor. These high-frequency upper transistor, high-frequency lower transistor, power-frequency upper transistor, and power-frequency lower transistor form a half-bridge arm circuit. Preferably, the high-frequency upper transistor, high-frequency lower transistor, power-frequency upper transistor, and power-frequency lower transistor are all fully controlled MOSFETs or IGBTs.

[0051] The replacement of traditional topology by totem pole topology is an inevitable event in the development of power supply. This new topology can provide higher efficiency and minimize conduction loss.

[0052] In some embodiments, the control module is a DSP digital control unit. A DSP control unit, also known as a digital signal processor, is a microprocessor specifically suited for performing digital signal processing operations. Its primary application is the real-time and rapid implementation of various digital signal processing algorithms. Based on this DSP digital control unit, classic dual-loop control is employed to control the totem pole topology (the specific structural schematic is not shown in the accompanying drawings).

[0053] Specifically, the control module includes a loop calculation unit, a modulation wave calculation unit, and a comparator, connected in sequence. The loop calculation unit can implement a classic dual-loop control system, comprising an outer voltage regulation unit and an inner current regulation unit. Specifically, a totem pole topology input voltage sampling unit, an output voltage sampling unit, and an input current sampling unit can be provided to obtain input and output voltage information as well as inductor current information. The loop calculation unit is configured to output a sine wave based on the received sampling signal.

[0054] The modulation wave calculation unit is connected to the output end of the loop calculation unit, wherein the modulation wave calculation unit is used to calculate a power frequency tube modulation wave and a high frequency tube modulation wave according to the received quasi-sine wave.

[0055] A comparator is connected to the output end of the modulation wave calculation unit and is connected to a carrier. The comparator is used to compare the received power frequency tube modulation wave with the carrier to generate a first high-frequency PWM signal, and to compare the received high-frequency tube modulation wave with the carrier to generate a second high-frequency PWM signal. The first high-frequency PWM signal and the second high-frequency PWM signal are at the same frequency, thereby achieving the purpose of using the modulation wave to control the high-frequency tube and the power frequency tube, rather than controlling the high-frequency tube and the power frequency tube separately.

[0056] The driving module includes a high-frequency driving filter circuit and an industrial frequency driving filter circuit. The high-frequency driving filter circuit is used to receive a second high-frequency PWM signal, output a high-frequency tube PWM signal to a driving circuit connected to the high-frequency driving filter circuit, and the driving circuit outputs the high-frequency tube driving signal; the industrial frequency driving filter circuit is used to receive a first high-frequency PWM signal, output an industrial frequency tube PWM signal to a driving circuit connected to the industrial frequency driving filter circuit, and the driving circuit outputs the industrial frequency tube driving signal.

[0057] In some embodiments, the driving circuit is a half-bridge driving chip.

[0058] In fact, in order to control the drive of the high-frequency tube and the low-frequency tube in the totem pole topology, after the high-frequency tube modulation wave and the power-frequency tube modulation wave are output by the control module, they are divided into a high-frequency tube control branch and a power-frequency tube control branch. Among them, the high-frequency tube control branch is used to support the high-frequency tube modulation wave to output a second high-frequency PWM signal after entering the comparator, and the second high-frequency signal outputs a high-frequency tube PWM signal after passing through the high-frequency drive filter circuit, and finally the high-frequency tube drive signal is output by the drive circuit; the power-frequency tube control branch is used to support the power-frequency tube modulation wave to output a first high-frequency PWM signal after entering the comparator, and the first high-frequency signal outputs the power-frequency tube PWM signal after passing through the power-frequency drive filter circuit, and finally the power-frequency tube drive signal is output by the drive circuit.

[0059] In some embodiments, the high-frequency filtering circuit includes a first resistor R1, a second resistor R2, a first capacitor C1, and a second capacitor C2; wherein, the first end of the first resistor R1 is connected to the output port of the control module, the second end of the first resistor R1 is connected to the first end of the first capacitor C1 and the input interface of the drive circuit, the first end of the second resistor R2 is connected to the port of the control module, the second end of the second resistor R2 is connected to the first end of the second capacitor C2 and the input interface of the drive circuit, and the second end of the first capacitor C1 and the second end of the second capacitor C2 are both grounded.

[0060] The power frequency filter circuit includes a third resistor R3, a fourth resistor R4, a third capacitor C3 and a fourth capacitor C4; wherein, the first end of the third resistor R3 is connected to the output port of the control module, the second end of the third resistor R3 is connected to the first end of the third capacitor C3 and the input interface of the drive circuit, the first end of the fourth resistor R4 is connected to the port of the control module, the second end of the fourth resistor R4 is connected to the first end of the fourth capacitor C4 and the input interface of the drive circuit, and the second end of the third capacitor C3 and the second end of the fourth capacitor C4 are both grounded.

[0061] The first resistor R1 and the second resistor R2, the first capacitor C1 and the second capacitor C2 in the high-frequency filter circuit set the high-frequency PWM filter parameters, and the third resistor R3 and the fourth resistor R4, the third capacitor C3 and the fourth capacitor C4 in the power frequency filter circuit set the power frequency PWM filter parameters. The high-frequency filter circuit and the power frequency filter circuit can be uniformly selected when selecting resistors, usually several hundred ohms, with the preferred resistance range being 100-500Ω. However, when selecting capacitors, it is necessary to distinguish between high frequency and power frequency. The first capacitor C1 and the second capacitor C2 in the high-frequency filter circuit are generally used to filter out interference signals in some circuits, and capacitors in the pF range are generally selected; while the third capacitor C3 and the fourth capacitor C4 in the power frequency filter circuit need to attenuate the PWM control signal, and capacitors in the nF range are generally selected.

[0062] As one implementation of this embodiment, filter parameters can be altered by changing the capacitor and resistor values ​​in the high-frequency filter circuit and the power-frequency filter circuit. For example, by increasing the power-frequency RC parameters, the first PWM output of the control module can be attenuated, filtering out the high-frequency portion. Considering the characteristics of a half-bridge driver chip, a typical driver chip has a low-level threshold that must be reached before it can convert the output to a low level. The present invention uses reasonable RC circuit parameters to control the low-level portion of the high-frequency portion to above the threshold, ultimately outputting a power-frequency drive signal through the driver chip without a high-frequency signal.

[0063] Compared with the existing technology, through the setting of the above-mentioned control device, it is possible to control the power frequency tube and the high frequency tube with the high-frequency PWM signal output by the control module, complete the synchronous commutation of the power frequency tube and the high frequency tube, and improve the current distortion at the zero point, especially the current distortion when the grid-connected inverter outputs reactive power.

[0064] Example 2

[0065] A second embodiment of the present invention provides a unipolar modulation reactive zero-crossing current distortion control method, which is applied to the unipolar modulation reactive zero-crossing current distortion control device provided in the first embodiment, and will not be described in detail here.

[0066] like Figure 14 As shown, the control method includes:

[0067] In step 140 , the control module receives a sampling signal of the totem pole topology and outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal.

[0068] Step 141 specifically includes:

[0069] The loop calculation unit receives the sampling signal of the totem pole topology and outputs a quasi-sine wave according to the sampling signal. Through the control loop, a quasi-sine wave calculation modulation wave close to the grid voltage is finally output. Part of the schematic diagram is shown as follows Figure 15 As shown, the loop calculation outputs a quasi-sine wave for unipolar modulation. This sine wave is hereinafter referred to as Uloop. Both the power frequency tube and the high frequency tube use this sine wave as the basis for judgment and commutate synchronously when it crosses zero.

[0070] The modulation wave calculation unit further calculates one power frequency tube modulation wave and one high frequency tube modulation wave according to the quasi-sine wave.

[0071] Among them, when Uloop>0, the modulation wave calculation unit outputs the high-frequency tube modulation wave = Vpk-Vmin-Uloop, and outputs the power frequency tube modulation wave = Vpk-Vmin;

[0072] When Uloop<0, the modulation wave calculation unit outputs the high-frequency tube modulation wave = -Uloop, and outputs the power frequency tube modulation wave = Vmin;

[0073] Uloop is the instantaneous value of the quasi-sine wave, Vpk is the carrier peak value, and Vmin is the maximum or minimum duty cycle limit. Vmin can be slightly larger or smaller, but it cannot be 0. This prevents the zero criterion from causing the power-frequency diode to repeatedly switch on and off near zero, affecting EMC.

[0074] The comparator compares the power frequency tube modulation wave with a carrier to generate a first high frequency PWM signal, and compares the high frequency tube modulation wave with the same carrier to generate a second high frequency PWM signal.

[0075] When the comparison value jumps from 0 to Vpk, the comparator will not set the output value low immediately, but will wait until the carrier counter goes from 0 to Vpk, that is, after a delay of half a carrier cycle, before setting the output value low. This will cause a half-cycle delay, as shown in the diagram. Figure 20 .

[0076] The final control module outputs the power frequency tube PWM wave as follows Figure 15 As shown, in actual applications, if the Vmin value is very small, a small duty cycle cannot be output because the DSP sets the dead zone module to filter out waveforms with very small pulse widths.

[0077] In step 142 , the driving module outputs a power-frequency tube driving signal according to the first high-frequency PWM signal, and outputs a high-frequency tube driving signal according to the second high-frequency PWM signal.

[0078] Among them, the first high-frequency PWM signal and the second high-frequency PWM signal are transmitted through the high-frequency tube control branch and the power-frequency tube control branch respectively. A high-frequency drive filter circuit is provided on the high-frequency tube control branch, and a power-frequency drive filter circuit is provided on the power-frequency tube control branch; the high-frequency drive filter circuit receives the second high-frequency PWM signal and outputs a high-frequency tube PWM signal, and the power-frequency drive filter circuit receives the first high-frequency PWM signal and outputs a power-frequency tube PWM signal; according to the high-frequency tube PWM signal and the power-frequency tube PWM signal, the drive circuit outputs the high-frequency tube drive signal and the power-frequency tube drive signal respectively.

[0079] The high-frequency drive filter circuit sets high-frequency PWM filter parameters, and after receiving the second high-frequency PWM signal, filters out the interference signal in the second high-frequency PWM signal to obtain a high-frequency tube PWM signal;

[0080] The power frequency drive filter circuit sets the power frequency PWM filter parameters, and after receiving the first high frequency PWM signal, performs attenuation control on the first high frequency PWM signal to obtain a power frequency tube PWM signal.

[0081] In step 143 , the high-frequency transistors in the totem pole topology are driven according to the high-frequency transistor driving signal, and the power-frequency transistors in the totem pole topology are driven according to the power-frequency transistor driving signal.

[0082] This application ultimately achieves the effect of synchronous commutation of the high-frequency tube and the power-frequency tube. After synchronization is ensured, the power-frequency tube is no longer controlled by the actual voltage. If the sampling is inaccurate, the commutation phenomenon will occur before the actual voltage crosses zero. The schematic diagram is as follows Figure 16 In this case, there will be no current distortion, and the switching mode before commutation is as follows. Figure 17As shown, when commutation occurs, the power frequency upper tube Q3 is turned on, and the duty cycle of the high frequency upper tube Q1 changes from 0 to 1. At this time, it will be maintained for a long time. Figure 18 In the switching mode shown, since the grid voltage is low at this time and in the same direction as the inductor current, the change in current is very small and can be ignored; then the high-frequency lower tube Q2 is on with a very small duty cycle, which cannot meet the conduction conditions in practice. It is usually manifested as the driving voltage does not reach the Miller platform and the switch tube cannot be turned on, so its duty cycle can be considered to be 0. After the high-frequency lower tube Q2 is turned on, it will be in Figure 19 In the mode shown, the voltage of the bus capacitor will be superimposed, causing the inductor current to rise sharply. However, since the power frequency tube and the high frequency tube are synchronized, the power frequency upper tube Q3 will be turned on, and the high frequency lower tube Q2 must have a small duty cycle, so the current rise is very small.

[0083] The present invention has been tested in practice and has shown that when no reactive power is output, the current has almost no distortion at the zero-crossing point. When reactive power is output, there is basically no distortion at the voltage zero-crossing point. The only delay between the two is caused by the hardware, which is a significant improvement compared to the current drop before the improvement.

[0084] At the same time, in response to poor input conditions, such as the Anshan waveform, which has high harmonic content and large harmonic amplitude, the power frequency tube and the high frequency tube are guaranteed to be commutated synchronously, so that there will be no current spikes near zero crossing. The principle is the same as above.

[0085] By implementing the control method of the embodiment, it is possible to control the power frequency tube and the high frequency tube with the quasi-sine wave output by the control module, complete the synchronous commutation of the power frequency tube and the high frequency tube, and improve the current distortion at the zero-crossing point, especially the current distortion when the grid-connected inverter outputs reactive power. Figure 21 The actual test waveform shown in the figure is when the power factor is 1. Figure 22 The actual test waveform shown in the figure is at a power factor of -0.8. Figure 23 The actual test waveform shown is when the power factor is 0.8. When no reactive power is output, the current has almost no distortion at the zero point. When reactive power is output, there is basically no distortion at the voltage zero point. Compared with the current drop before improvement, there is a great improvement.

[0086] Here, first, second... only represent the distinction in their names, and do not represent any difference in their importance and position.

[0087] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A unipolar modulation reactive zero-crossing current distortion control device, characterized in that: include: Totem pole topology, control module and driver module; where: The control module includes a loop calculation unit, a modulation wave calculation unit, and a comparator; the loop calculation unit is used to receive a sampling signal of the totem pole topology and output a quasi-sine wave according to the sampling signal; the modulation wave calculation unit is used to calculate a power frequency tube modulation wave and a high-frequency tube modulation wave according to the quasi-sine wave; the comparator is used to compare the power frequency tube modulation wave with a carrier to generate a first high-frequency PWM signal, and compare the high-frequency tube modulation wave with the same carrier to generate a second high-frequency PWM signal; The driving module includes a high-frequency driving filter circuit and an industrial frequency driving filter circuit. The high-frequency driving filter circuit is used to receive the second high-frequency PWM signal and output a high-frequency tube PWM signal to the driving circuit connected to the high-frequency driving filter circuit, and the driving circuit outputs the high-frequency tube driving signal; the industrial frequency driving filter circuit is used to receive the first high-frequency PWM signal and output the industrial frequency tube PWM signal to the driving circuit connected to the industrial frequency driving filter circuit, and the driving circuit outputs the industrial frequency tube driving signal; The totem pole topology includes an AC input / output terminal, a high-frequency tube, a power-frequency tube, a PFC inductor, and a DC input / output terminal, wherein the high-frequency tube is driven according to the high-frequency tube drive signal, and the power-frequency tube is driven according to the power-frequency tube drive signal; The control module receives a sampling signal of the totem pole topology and outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal; The driving module outputs a power-frequency tube driving signal according to the first high-frequency PWM signal, and outputs a high-frequency tube driving signal according to the second high-frequency PWM signal; The high-frequency tube in the totem pole topology is driven according to the high-frequency tube driving signal, and the power-frequency tube in the totem pole topology is driven according to the power-frequency tube driving signal; The control module receives a sampling signal of the totem pole topology and outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal, including: The loop calculation unit receives a sampling signal of the totem pole topology and outputs a quasi-sine wave according to the sampling signal; The modulation wave calculation unit calculates a power frequency tube modulation wave and a high frequency tube modulation wave according to the quasi-sine wave; The comparator compares the power frequency tube modulated wave with a carrier wave to generate a first high frequency PWM signal, and compares the high frequency tube modulated wave with the same carrier wave to generate a second high frequency PWM signal; The modulation wave calculation unit calculates a power frequency tube modulation wave and a high frequency tube modulation wave according to the sine-like wave, including: When Uloop>0, the modulation wave calculation unit outputs a high-frequency tube modulation wave = Vpk-Vmin-Uloop, and outputs a power-frequency tube modulation wave = Vpk-Vmin; When Uloop<0, the modulation wave calculation unit outputs the high-frequency tube modulation wave = -Uloop, and outputs the power-frequency tube modulation wave = Vmin; Wherein, the Uloop is the instantaneous value of the quasi-sine wave, the Vpk is the carrier peak value, and the Vmin is the maximum duty cycle or minimum duty cycle value.

2. The unipolar modulation reactive zero-crossing current distortion control device according to claim 1, characterized in that: The high-frequency tube includes a high-frequency upper tube and a high-frequency lower tube, and the power-frequency tube includes a power-frequency upper tube and a power-frequency lower tube. The high-frequency upper tube, the high-frequency lower tube, the power-frequency upper tube and the power-frequency lower tube form a half-bridge arm circuit.

3. The unipolar modulation reactive zero-crossing current distortion control device according to claim 2, characterized in that: The high-frequency upper tube, the high-frequency lower tube, the power-frequency upper tube and the power-frequency lower tube are all MOSFET switch tubes or IGBT fully-controlled devices.

4. The unipolar modulation reactive zero-crossing current distortion control device according to claim 1, characterized in that: The high-frequency tube driving signal includes a set of complementary high-frequency upper tube driving signals and high-frequency lower tube driving signals, and the power-frequency tube starting signal includes a set of complementary power-frequency upper tube driving signals and power-frequency lower tube driving signals.

5. The unipolar modulation reactive zero-crossing current distortion control device according to claim 1, characterized in that: The driving circuit is used to convert the high-frequency tube PWM signal and the industrial frequency tube PWM signal into a driving signal capable of driving a switching device.

6. A method for controlling unipolar modulation reactive zero-crossing current distortion, characterized in that: Applied to the unipolar modulation reactive zero-crossing current distortion control device according to any one of claims 1 to 5, the control method includes: The control module receives a sampling signal of the totem pole topology and outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal; The driving module outputs a power-frequency tube driving signal according to the first high-frequency PWM signal, and outputs a high-frequency tube driving signal according to the second high-frequency PWM signal; The high-frequency tube in the totem pole topology is driven according to the high-frequency tube driving signal, and the power-frequency tube in the totem pole topology is driven according to the power-frequency tube driving signal.

7. The unipolar modulation reactive zero-crossing current distortion control method according to claim 6, characterized in that: The control module receives a sampling signal of the totem pole topology and outputs a first high-frequency PWM signal and a second high-frequency PWM signal of the same frequency based on the sampling signal, including: The loop calculation unit receives a sampling signal of the totem pole topology and outputs a quasi-sine wave according to the sampling signal; The modulation wave calculation unit calculates a power frequency tube modulation wave and a high frequency tube modulation wave according to the quasi-sine wave; The comparator compares the power frequency tube modulated wave with a carrier to generate a first high frequency PWM signal, and compares the high frequency tube modulated wave with the same carrier to generate a second high frequency PWM signal.

8. The unipolar modulation reactive zero-crossing current distortion control method according to claim 7, characterized in that: The modulation wave calculation unit calculates a power frequency tube modulation wave and a high frequency tube modulation wave according to the sine-like wave, including: When Uloop>0, the modulation wave calculation unit outputs a high-frequency tube modulation wave = Vpk-Vmin-Uloop, and outputs a power-frequency tube modulation wave = Vpk-Vmin; When Uloop<0, the modulation wave calculation unit outputs the high-frequency tube modulation wave = -Uloop, and outputs the power-frequency tube modulation wave = Vmin; Wherein, the Uloop is the instantaneous value of the quasi-sine wave, the Vpk is the carrier peak value, and the Vmin is the maximum duty cycle or minimum duty cycle value.

9. The unipolar modulation reactive zero-crossing current distortion control method according to claim 6, characterized in that: The driving module outputs a power-frequency tube driving signal according to the first high-frequency PWM signal, and outputs a high-frequency tube driving signal according to the second high-frequency PWM signal, including: The high-frequency drive filter circuit receives the second high-frequency PWM signal and outputs a high-frequency tube PWM signal, and the industrial frequency drive filter circuit receives the first high-frequency PWM signal and outputs an industrial frequency tube PWM signal; According to the high-frequency tube PWM signal and the industrial frequency tube PWM signal, the driving circuit outputs a high-frequency tube driving signal and an industrial frequency tube driving signal respectively.

10. The unipolar modulation reactive zero-crossing current distortion control method according to claim 9, characterized in that: The high-frequency drive filter circuit receives the second high-frequency PWM signal and outputs a high-frequency tube PWM signal, and the industrial frequency drive filter circuit receives the first high-frequency PWM signal and outputs an industrial frequency tube PWM signal, including: The high-frequency drive filter circuit sets high-frequency PWM filter parameters, and after receiving the second high-frequency PWM signal, filters out interference signals in the second high-frequency PWM signal to obtain the high-frequency tube PWM signal; The industrial frequency drive filter circuit sets industrial frequency PWM filter parameters, and after receiving the first high frequency PWM signal, performs attenuation control on the first high frequency PWM signal to obtain the industrial frequency tube PWM signal.

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