Virtual Inductance Self-Calibration Method and System for Spread Spectrum Control of Permanent Magnet Motor Current Ripple

Abstract

The present invention discloses a method and system for spread spectrum control of current ripple of a permanent magnet motor with virtual inductor self - correction, belonging to the technical field of motors. According to the set value of the virtual inductor parameter, the present invention takes the peak value of the three - phase current ripple as the control object, updates the switching frequency of the next switching period through the three - phase current ripple spread spectrum control module and calculates the predicted value of the characteristic inflection point of the three - phase current ripple; obtains the sampled value of the characteristic inflection point of the three - phase current ripple through sampling; inputs the sampled value and the predicted value of the characteristic inflection point of the three - phase ripple current into the virtual inductor self - correction module, and obtains the virtual inductor correction value through decoupling function calculation and proportional - integral control; according to the virtual inductor correction value, updates the set value of the virtual inductor parameter, thereby realizing high - accuracy control of the peak value of the three - phase current ripple. The present invention comprehensively considers the self - inductance and mutual - inductance parameters of the motor, reduces the influence of inductance change on the prediction accuracy of current ripple during the operation of the motor, improves the control accuracy of the system, and realizes a better current ripple control effect.

Description

Technical Field

[0001] The present invention belongs to the technical field of motors, and more specifically, relates to a method and system for spreading spectrum control of current ripple of a permanent magnet motor with virtual inductor self - correction. Background Art

[0002] Among many types of motors, permanent magnet synchronous motors are widely used in electric vehicles, wind turbines, machine tools, etc. due to their advantages such as high efficiency, high power density, simplicity and reliability. In recent years, due to the emergence and development of new technologies, new processes and new devices, the performance of permanent magnet synchronous motors has been further improved, and their application scope has been continuously broadened.

[0003] In order to reduce the operating loss of permanent magnet motors and improve the safety and stability of the entire electromagnetic system, the country has control requirements for current ripple in the control system. Traditional systems use the method of increasing the switching frequency to reduce current ripple to meet this requirement. However, this not only causes the current ripple to be far lower than the application requirements at some moments, resulting in additional switching losses, but also the single switching frequency will exacerbate the electromagnetic interference of the switching frequency. According to the existing scheme of calculating the current ripple through given parameters and then adjusting the switching frequency, although a good control effect can be achieved when the parameters are accurate, in most cases, due to the bus voltage fluctuation, the inductor changing with temperature, or the saturation of the magnetic circuit caused by the change of current and internal power factor angle due to loading and unloading, the actual system parameters will deviate from the set values. At this time, it is difficult to achieve the desired ripple control effect using this regulation scheme, and its ripple control stability cannot be guaranteed. Summary of the Invention

[0004] Aiming at the defects of the existing technology, the present invention provides a method and system for spreading spectrum control of current ripple of a permanent magnet motor with virtual inductor self - correction, establishing a simplified model for predicting the current ripple of a permanent magnet motor to simplify the control difficulty on the basis of ensuring the control accuracy, and establishing a virtual inductor self - correction model, aiming at reducing the current ripple control error with the virtual inductor as the control object based on the self - correction model.

[0005] To achieve the above object, on the one hand, the present invention provides a method for spreading spectrum control of current ripple of a permanent magnet motor with virtual inductor self - correction, including the following steps:

[0006] (1) Calculate the expression of the three - phase current change rate in the next switching period according to the set value of the virtual inductor;

[0007] (2) Update the switching frequency of the next switching period according to the expected value of the current ripple and the expression of the three - phase current change rate obtained in step (1), and calculate the predicted value of the characteristic inflection point of the three - phase current ripple in the next switching period;

[0008] (3) sampling in the next switching cycle to obtain a sampled value of the inflection point of the three-phase current ripple characteristic, and obtaining a virtual inductance correction value based on the sampled value and the ripple prediction value of the three-phase current characteristic inflection point obtained in step (2); updating the three-phase current change rate expression in step (1) according to the virtual inductance correction value;

[0009] (4) Repeat steps (1)-(3) to achieve virtual inductance self-correction and current ripple spread spectrum control.

[0010] Furthermore, the step (1) specifically includes:

[0011] Simplify the mathematical model of the permanent magnet motor, ignore the stator winding resistance and the stator winding leakage self-inductance, then the average value of the stator winding mutual inductance M s0 is the average value of stator winding self-inductance L s0 Half of the stator winding mutual inductance second harmonic amplitude M s2 The second harmonic amplitude of the stator winding self-inductance L s2 Equal, that is, M s0 =L s0 / 2;M s2 =L s2 ; The motion back EMF caused by the change of permanent magnet flux and the back EMF caused by the change of stator winding self-inductance and mutual inductance are taken as steady-state back EMF, and the relationship between the upper arm switch duty cycle and the steady-state back EMF is obtained according to the modulation principle;

[0012] According to the original inductance parameter L s0 , L s2 and real-time inductance parameter L s0 ', L s2 'Define virtual inductance L V1 , L V2 ;

[0013] According to the virtual inductance, rotor position, steady-state back EMF, DC bus voltage and upper bridge arm switch state, the current equation (KCL) and voltage equation (KVL) are written to obtain the expression of the three-phase current change rate corresponding to different switch states in the next switching cycle.

[0014] Furthermore, the step (3) specifically includes:

[0015] (3.1) Sampling the next switching cycle to obtain the sampling value of the inflection point of the three-phase current ripple characteristics;

[0016] (3.2) According to the sampling value of the inflection point of the three-phase current ripple characteristic obtained in step (3.1) and the predicted value of the inflection point of the three-phase current ripple characteristic obtained in step (2), the feedback decoupling function value F is obtained respectively through the virtual inductance decoupling function v1m 、F v2m Decoupled from the predicted function value Fv1c , F v2c ;

[0017] (3.3) Subtract the feedback decoupling function values F v1m , F v2m from the predicted decoupling function values F v1c , F v2c to obtain the virtual inductor correction value through a proportional-integral controller. Update the virtual inductor setting value according to the virtual inductor correction value, and further update the three-phase current change rate expression in step (1);

[0018] (3.4) Repeat steps (1) to (3) until the virtual inductor finally stabilizes to the true value.

[0019] Further, step (3.1) specifically includes:

[0020] During one switching period, sample the A, B, and C phase currents when the upper-bridge-arm switch of each phase of the converter operates;

[0021] Consider the fundamental current within the switching period as linearly varying, and obtain the linear expression of the fundamental current within this switching period through the sampled values of the three-phase currents;

[0022] Subtract each three-phase current sampled value from the fundamental current value at the corresponding moment to obtain the sampled values of the inflection points of the three-phase current ripple characteristics.

[0023] Further, step (3.2) specifically includes:

[0024] According to the relationship between the obtained three-phase current ripple and the virtual inductor, obtain the virtual inductor decoupling function, and the calculation result of the equation expression is positively correlated with the magnitude of the virtual inductor.

[0025] Respectively take the sampled values of the inflection points of the three-phase current ripple characteristics obtained in step (3.1) and the predicted values of the inflection points of the three-phase current ripple obtained in step (2) as inputs, and solve the feedback decoupling function values F v1m , F v2m and the predicted decoupling function values F v1c , F v2c .

[0026] The output voltage of the inverter is a stepped wave of the switching frequency, and the back electromotive force of the motor is a sine wave of the electrical cycle frequency. The voltage difference between the two is the back electromotive force generated due to the current fluctuation flowing through the three-phase coupled inductor. Through a simplified motor model, the self-inductance and mutual-inductance matrices of the permanent magnet motor can be equivalently transformed into a matrix containing L s0 , L s2 , and the rotor position. According to the KCL and KVL equations, the three-phase currents can be decoupled to obtain independent current change rate expressions.

[0027] Based on the virtual inductor parameter self - correction technology, the present invention can obtain the switching period of the next moment through conditions such as the expected value of current ripple and the expression of current change rate, so as to perform spread - spectrum control on the inverter, effectively reducing the switching loss of the main circuit and the peak value of EMI noise of the motor system.

[0028] On the other hand, the present invention provides a spread - spectrum control system for current ripple of a permanent - magnet motor with virtual inductor self - correction, including:

[0029] A controller, a three - phase current ripple spread - spectrum control module, and a virtual inductor self - correction module. The controller is used to send the on - bridge arm switch duty ratio of the k - th switching period to the three - phase current ripple spread - spectrum control module at the (k - 1) - th switching period;

[0030] The three - phase current ripple spread - spectrum control module is used to calculate the expression of the change rate of three - phase current within the k - th switching period according to the virtual inductor set value at the (k - 1) - th switching period, calculate and change the switching frequency of the k - th switching period according to the expected value of current ripple, and send the predicted value of the inflection point of the three - phase current ripple characteristics of the k - th switching period to the virtual inductor self - correction module;

[0031] The virtual inductor self - correction module is used to obtain the three - phase current sampling signal of the k - th switching period at the k - th switching period, calculate the sampled value of the inflection point of the three - phase current ripple characteristics of the k - th switching period, and substitute them into the virtual inductor decoupling function respectively with the predicted value of the inflection point of the three - phase current ripple characteristics. After proportional - integral regulation, the virtual inductor correction value is obtained and sent to the three - phase current ripple spread - spectrum control module, which is used to correct the virtual inductor set value used by the three - phase current ripple spread - spectrum control module to regulate the switching frequency of the (k + 2) - th switching period at the (k + 1) - th switching period, so as to achieve a better spread - spectrum control effect of current ripple.

[0032] Furthermore, in the three - phase current ripple spread - spectrum control module, by simplifying the mathematical model of the permanent - magnet motor, ignoring the stator winding resistance and the stator winding leakage self - inductance, the motion back - electromotive force caused by the change of the permanent - magnet flux linkage and the back - electromotive force caused by the change of the self - inductance and mutual inductance of the stator winding are regarded as the steady - state back - electromotive force, and the relationship between the on - bridge arm switch duty ratio and the steady - state back - electromotive force is obtained according to the modulation principle;

[0033] According to the original inductance parameters L s0 、L s2 and the real - time inductance parameters L s0 '、L s2 ', the virtual inductors L V1 、 L V2 are defined;

[0034] Based on the virtual inductor, rotor position, steady-state back electromotive force, DC bus voltage, and the switching state of the upper bridge arm, write the current equation and voltage equation to obtain the expression of the three-phase current change rate corresponding to different switching states in the next switching period.

[0035] Further, in the virtual inductor self-calibration module, the sampling values of the three-phase current ripple characteristic inflection points sampled in the k-th switching period and the predicted values of the three-phase current ripple characteristic inflection points are respectively used to obtain the feedback decoupling function values F v1m 、F v2m and the predicted decoupling function values F v1c 、F v2c through the virtual inductor decoupling function; the feedback decoupling function values F v1m 、F v2m are subtracted from the predicted decoupling function values F v1c 、F v2c , and the virtual inductor correction value is obtained through a proportional-integral controller. The virtual inductor setting value is updated according to the virtual inductor correction value, and further the three-phase current change rate expression is updated.

[0036] Further, the sampling values of the three-phase current ripple characteristic inflection points are obtained by sampling the A, B, and C phase currents when the upper bridge arm switch of each phase of the converter operates within the k-th switching period; the fundamental current in the switching period is regarded as linearly changing, and the linear expression of the fundamental current in this switching period is obtained through the three-phase current sampling values; the current ripple characteristic inflection point sampling values are obtained by subtracting the fundamental current values at the corresponding moments from the three-phase current sampling values.

[0037] Through the above technical solutions conceived by the present invention, compared with the prior art, the following beneficial effects can be achieved:

[0038] (1) The permanent magnet motor model used by the present invention to calculate the current change rate expression is a reasonably simplified model, and the concept of virtual inductor is proposed. The calculation process of the current change rate expression uses fewer parameters, which is convenient for mathematical simplification and law summary.

[0039] (2) By introducing the virtual inductor self-calibration technology, the present invention can, when the motor parameters change, ensure the correctness of the calculation of the current change rate expression through virtual inductor self-calibration, thereby ensuring the accuracy of current ripple spread spectrum control.

[0040] (3) Due to the introduction of the virtual inductor self-calibration technology, the present invention can reduce the accuracy requirement for the initial parameter setting of the motor model, and make up for the initial parameter setting error through virtual inductor self-calibration to ensure the accuracy of current ripple spread spectrum control. BRIEF DESCRIPTION OF THE DRAWINGS

[0041] Figure 1 is a two-level permanent magnet motor circuit model;

[0042] Figure 2 It is a schematic diagram of the equivalent circuit of the two-level permanent magnet motor implemented in the present invention;

[0043] Figure 3 It is a schematic diagram for predicting the single-phase current ripple within a single switching period;

[0044] Figure 4 It is a schematic diagram for measuring the single-phase current ripple within a single switching period implemented in the present invention;

[0045] Figure 5 It is a flow chart of the virtual inductor self-calibration implemented in the present invention;

[0046] Figure 6 It is a block diagram for implementing the current ripple spread spectrum control of the virtual inductor self-calibration permanent magnet motor implemented in the present invention;

[0047] Figure 7 It is an operation diagram of the control system implemented in the present invention;

[0048] Figure 8 It is the current ripple control effect diagram according to the traditional current ripple control scheme under the condition of motor parameter variation;

[0049] Figure 9 It is the current ripple control effect diagram of the virtual inductor self-calibration permanent magnet motor current ripple spread spectrum control scheme implemented according to the present invention under the condition of motor parameter variation. Detailed implementation manners

[0050] In order to make the objectives, technical solutions and advantages of the present invention clearer and more understandable, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not used to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0051] Aiming at the blank of the self-calibration technology in the application of current ripple, the present invention establishes a two-level permanent magnet motor current ripple spread spectrum control model and a virtual inductor self-calibration model based on the concept of virtual inductor. Based on the relationship between the sampling value of the current ripple characteristic inflection point, the predicted value of the current ripple characteristic inflection point and the virtual inductor, the virtual inductor parameters used in the current ripple spread spectrum control are adjusted to improve the accuracy of the current ripple spread spectrum regulation.

[0052] Compared with the traditional current ripple spread spectrum control technology, the current ripple spread spectrum control technology with virtual inductor self-correction has a higher tolerance for the input error of motor parameters and has a better current ripple spread spectrum regulation effect under the condition that the motor parameters change due to various reasons.

[0053] Therefore, the present invention first proposes the control application of the self-calibration technology to the current ripple, effectively improving the stability and reliability of the current ripple spread spectrum control.

[0054] As Figure 1 shown, for the two-level permanent magnet motor mentioned above, the three-phase windings are mutually coupled due to the mutual inductance. Among them, L a , L b , L c are the self-inductances of the three phases, M ab , M bc , M ac are the mutual inductances of the three phases, V la , V lb , V lc are the back electromotive forces generated due to the changes in mutual inductance and self-inductance, and V fa , V fb , V fc are the back electromotive forces generated due to the change in rotor magnetic flux. Simplifying the mathematical model of the permanent magnet motor and ignoring the stator winding resistance and the leakage self-inductance of the stator winding, the average value M s0 of the mutual inductance of the stator winding is half of the average value L s0 of the self-inductance of the stator winding, and the amplitude M s2 of the second harmonic of the mutual inductance of the stator winding is equal to the amplitude L s2 of the second harmonic of the self-inductance of the stator winding, that is, M s0 = L s0 / 2; M s2 = L s2 .

[0055] As Figure 2 shown, V ad , V bd , V cd are the phase output voltages, and there are two voltage states (V dc / 2, -V dc / 2) relative to the midpoint of the DC bus. Among them, V dc is the DC bus voltage, and V ab , V bc , V ca , V ac , V ba , V cb are the voltages generated due to the mutual coupling of the three-phase currents through the mutual inductance, and V aj , V bj , V cj are the equivalent steady-state voltages.

[0056] Among them, V aj , V bj , V cj can be regarded as the sum of the back electromotive force generated due to the changes in mutual inductance and self-inductance, the back electromotive force generated due to the change in rotor magnetic flux, and the common-mode back electromotive force.ki =(2*d i -1)*V dc / 2; k = a, b, c, where d i is the duty cycle of the upper bridge arm, V ab 、V bc 、V ca 、V ac 、V ba 、V cb satisfy:

[0057] V jk =-M jk *di j / dt; j, k = a, b, c; j ≠ k;

[0058] Through the equations:

[0059] i a +i b +i c =0,

[0060] L a *di a / dt = V ad -V ba -V ca -V cj ,

[0061] L b *di b / dt = V bd -V ab -V cb -V cj ,

[0062] L c *di c / dt = V cd -V ac -V bc -V cj ,

[0063] The current change rate expressions for different voltage vector action segments within a single switching period can be calculated as follows (taking phase A as an example):

[0064]

[0065] Introduce the concept of virtual inductance and convert the current change rate expression to (taking phase A as an example):

[0066]

[0067] where i = 0, …, 7; j = a, b, c are the switching states and current phases respectively, L v1 、Lv2 is the virtual inductor, d a , d b , d c are the duty ratios of the upper arms of the three-phase switches. is the motor coefficient, which is always a constant value in this modulation system.

[0068] As Figure 3 shown, according to the current change rate expression and the duty ratios of the upper arms of the three-phase switches, the switching frequency at the next moment is calculated based on the expected value of the current ripple: f = max(x i , y i ) / i rippleref . Among them, x i , y i are the current ripple different characteristic inflection point values of the three-phase currents respectively, and i rippleref is the set value of the current ripple.

[0069] Through the above steps, the switching frequency at the next moment can be calculated according to the existing parameters.

[0070] Secondly, based on the virtual inductor self-correction model, accurate control of the current ripple can be achieved even when the motor parameters change or the initial values are set inaccurately.

[0071] As Figure 4 shown, within a switching period, when the upper-arm switches of each phase of the converter act, sample the A, B, and C phase currents; regard the fundamental current within the switching period as linearly changing, and calculate the linear expression of the fundamental current within this switching period; (Take the midpoint of the switching period as the time zero point), subtract the sampled values of the three-phase currents from the fundamental current values to obtain the sampled values of the current ripple characteristic inflection points of the three-phase currents.

[0072] Substitute the sampled values of the current ripple characteristic inflection points x am , x bm obtained by sampling and the predicted values of the current ripple characteristic inflection points x ac , x bc into the virtual inductor decoupling function:

[0073]

[0074]

[0075]

[0076] Among them, x a , x b are the current ripple values at the first inflection points of the a and b phase current ripples respectively. Obtain the feedback decoupling function values F v1m , F v2m and the predicted decoupling function values Fv1c , F v2c , where V dc is the DC bus voltage, T1 is the switching period, θ is the rotor electrical angle, and d a , d b , d c are the duty ratios of the three-phase upper bridge arms. Subtract the feedback decoupling function values F v1m , F v2m from the predicted decoupling function values F v1c , F v2c and obtain the virtual inductor correction value through proportional-integral regulation.

[0077] Through the above calculation method, the value of the virtual inductor can be corrected in real time. The specific correction flowchart is as Figure 5 shown.

[0078] Based on the above virtual inductor self-correction permanent magnet motor current ripple spread spectrum control model, closed-loop control of the permanent magnet motor current ripple can be achieved.

[0079] The implementation block diagram of the controller is as Figure 6 shown. Input the duty ratio of the upper bridge arm switch into the three-phase current ripple spread spectrum control module. The three-phase current ripple spread spectrum control module takes the peak value of the current ripple as the control object, regulates the switching frequency according to parameters such as the virtual inductor setting value, and outputs the predicted value of the current ripple characteristic inflection point to the virtual inductor self-correction module. The virtual inductor self-correction module calculates the virtual inductor correction value based on the predicted value of the current ripple characteristic inflection point and the sampled value of the current ripple characteristic inflection point obtained by sampling, and outputs it to the three-phase current ripple spread spectrum control module to correct the virtual inductor setting value, forming a closed loop.

[0080] This control system includes a control core, a three-phase current ripple spread spectrum control module, and a virtual inductor self-correction module. The controller receives information such as three-phase current and rotor position at the k-1 switching period, and transmits the duty ratio of the upper bridge arm switch and rotor position information to the three-phase current ripple spread spectrum control module. The three-phase current ripple spread spectrum control module calculates the expression of the three-phase current change rate in each switch state section at the k switching period based on the above information and the virtual inductor setting value at the k-1 switching period, obtains the switching frequency at the k switching period according to the expected value of the current ripple, and transmits the predicted value of the current ripple characteristic inflection point at the k switching period to the virtual inductor self-correction module. The virtual inductor self-correction module samples the current sampling signal at the k switching period at the k switching period, calculates the sampled value of the three-phase current ripple characteristic inflection point at the k switching period, and substitutes it and the predicted value of the three-phase current ripple characteristic inflection point into the virtual inductor decoupling function respectively. After proportional-integral regulation, the virtual inductor correction value is obtained and sent to the three-phase current ripple spread spectrum control module to correct the virtual inductor setting value used by the three-phase current ripple spread spectrum modulation module to regulate the switching frequency at the k+2 switching period at the k+1 switching period. The specific process is asFigure 7 as shown

[0081] As Figure 8 shown, it is the current ripple spread spectrum control effect of the traditional current ripple spread spectrum control scheme under the condition that the motor parameters change, where the per-unit value is obtained relative to the current ripple set value. As Figure 9 shown, it is the control effect of the current ripple spread spectrum control scheme with the virtual inductor self-correction module added under the condition that the motor parameters change, where the per-unit value is obtained relative to the current ripple set value. After adding the virtual inductor self-correction module, the current ripple control effect has been significantly improved.

[0082] It is easy for those skilled in the art to understand that the above are only the preferred embodiments of the present invention and are not intended to limit the present invention. Any modifications, equivalent replacements, and improvements made within the spirit and principle of the present invention shall be included in the protection scope of the present invention.

Claims

1. A method for spread spectrum control of current ripple in a permanent magnet motor with virtual inductor self - calibration, characterized in that, The method includes the following steps: (1) Calculate the expression of the three-phase current change rate in the next switching period according to the virtual inductor setting value. Specifically, it includes: Simplify the mathematical model of the permanent magnet motor, neglect the stator winding resistance and the stator winding leakage self-inductance, take the motional back electromotive force caused by the change of the permanent magnet flux linkage and the back electromotive force caused by the change of the self-inductance and mutual inductance of the stator winding as the steady-state back electromotive force, and obtain the relationship between the upper-bridge-arm switch duty ratio and the steady-state back electromotive force according to the modulation principle; According to the original inductance parameters L s0 、L s2 and the real-time inductance parameters L s0 ', L s2 ' to define the virtual inductances L V1 、L V2 ; Write the current equation and voltage equation according to the virtual inductor, rotor position, steady-state back electromotive force, DC bus voltage and upper-bridge-arm switch state, and obtain the expression of the three-phase current change rate corresponding to different switch states in the next switching period; (2) Update the switching frequency of the next switching period according to the current ripple expectation value and the expression of the three-phase current change rate in the next switching period obtained in step (1), and calculate the predicted value of the three-phase current ripple characteristic inflection point in the next switching period; (3) Sample the sampled value of the three-phase current ripple characteristic inflection point in the next switching period, and obtain the virtual inductor correction value according to the predicted value of the three-phase current ripple characteristic inflection point obtained in step (2); update the virtual inductor setting value according to the virtual inductor correction value. Specifically, it includes: (3.1) Sample the sampled value of the three-phase current ripple characteristic inflection point in the next switching period. Specifically, it includes: During a switching period, sample the ABC three-phase currents when the upper-bridge-arm switch of each phase of the converter acts; Regard the fundamental wave current in the switching period as linearly changing, and obtain the linear expression of the fundamental wave current in the switching period through the three-phase current sampled values; Subtract the fundamental wave current value at the corresponding moment from each three-phase current sampled value to obtain the sampled value of the three-phase current ripple characteristic inflection point; (3.2) According to the sampled values of the three-phase current ripple characteristic inflection points and the predicted values of the three-phase current ripple characteristic inflection points obtained in the step (3.1), the feedback decoupling function values F v1m , F v2m and the predicted decoupling function values F v1c , F v2c are respectively obtained through the virtual inductor decoupling function; specifically, it includes: Derive the virtual inductor decoupling function according to the relationship between the obtained three-phase current ripple and the virtual inductor; Taking the sampled values of the inflection points of the three-phase current ripple characteristics and the predicted values of the inflection points of the three-phase current ripple characteristics as inputs respectively, the feedback decoupling function value F is solved through the virtual inductor decoupling function v1m , F v2m and the predicted decoupling function value F v1c , F v2c ; (3.3) Decouple the feedback function values F v1m and F v2m from the predicted decoupling function values F v1c and F v2c to obtain the virtual inductor correction value through a proportional-integral controller, and update the three-phase current change rate expression in the step (1) according to the virtual inductor correction value; (4) Repeat steps (1)-(3) above to realize virtual inductor self-correction and current ripple spread spectrum control.

2. A control system for spread spectrum control of current ripple in a permanent magnet motor with virtual inductor self - calibration, characterized in that, It includes a controller, a three-phase current ripple spread spectrum control module, and a virtual inductor self-correction module; The controller is used to send the upper-bridge-arm switch duty ratio of the kth switching period to the three-phase current ripple spread spectrum control module in the (k - 1)th switching period; The three-phase current ripple spread spectrum control module is used to calculate the expression of the three-phase current change rate in the kth switching period according to the virtual inductor setting value in the (k - 1)th switching period, calculate and change the switching frequency of the kth switching period according to the current ripple expectation value, and send the predicted value of the three-phase current ripple characteristic inflection point of the kth switching period to the virtual inductor self-correction module; in the three-phase current ripple spread spectrum control module, the mathematical model of the permanent magnet motor is simplified, the stator winding resistance and the stator winding leakage self-inductance are neglected, the motional back electromotive force caused by the change of the permanent magnet flux linkage and the back electromotive force caused by the change of the self-inductance and mutual inductance of the stator winding are taken as the steady-state back electromotive force, and the relationship between the upper-bridge-arm switch duty ratio and the steady-state back electromotive force is obtained according to the modulation principle; According to the original inductance parameters L s0 、L s2 and the real-time inductance parameters L s0 '、L s2 ', define the virtual inductances L V1 、L V2 ; Write the current equation and voltage equation according to the virtual inductor, rotor position, steady-state back electromotive force, DC bus voltage and upper-bridge-arm switch state, and obtain the expression of the three-phase current change rate corresponding to different switch states in the next switching period; The virtual inductor self - calibration module is used to obtain the three - phase current sampling signals in the k - th switching period, calculate the sampling values of the three - phase current ripple characteristic inflection points in the k - th switching period, and substitute them and the predicted values of the three - phase current ripple characteristic inflection points into the virtual inductor decoupling function respectively. Through proportional - integral regulation, the virtual inductor correction value is obtained and sent to the three - phase current ripple spread - spectrum control module to update the virtual inductor setting value, so as to realize the current ripple spread - spectrum control; In the virtual inductor self - calibration module, the sampling values of the three - phase current ripple characteristic inflection points sampled in the k - th switching period and the predicted values of the three - phase current ripple characteristic inflection points are used to obtain the feedback decoupling function values F v1m , F v2m and the predicted decoupling function values F v1c , F v2c ; The difference between the feedback decoupling function values F v1m , F v2m and the predicted decoupling function values F v1c , F v2c is taken, and the virtual inductor correction value is obtained through a proportional - integral controller and sent to the three - phase current ripple spread - spectrum control module. The three - phase current ripple spread - spectrum control module updates the virtual inductor setting value accordingly, and then calculates the three - phase current change rate expression; The sampling values of the three - phase current ripple characteristic inflection points are obtained by sampling the A, B, and C phase currents when the upper - arm switch of each phase of the converter acts within the k - th switching period; The fundamental current within the switching period is regarded as linearly changing, and the linear expression of the fundamental current within the switching period is obtained through the three - phase current sampling values; The difference between the three - phase current sampling values and the fundamental current values at the corresponding moments is used to obtain the sampling values of the three - phase current ripple characteristic inflection points; The virtual inductor self-calibration module corrects the set value of the virtual inductor in the k-th switching period, and through the three-phase current ripple spread spectrum control module, it will be used to regulate the switching frequency in the (k + 2)-th switching period in the (k + 1)-th switching period.

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