Control device for an alternating current rotating electric machine and electric power steering device

Abstract

The present application provides an alternating current rotating electric machine control device and an electric power steering device capable of reducing an error component of a current detection value, the error component being close to a mechanical resonance period of the alternating current rotating electric machine. The alternating current rotating electric machine control device (10) of the present application detects a current flowing through a three-phase winding at a period that is a first natural number (A) times a carrier period (Tc), that is, a current detection period (TIdt), performs current addition processing by adding a current detection value detected this time to a current detection value detected before a period that is a second natural number (B) times the current detection period (TIdt), that is, an addition period (Tadd), calculates a current detection value after the current addition processing, and calculates a current command value for the three phases based on the current detection value after the current addition processing, the second natural number (B) being set to a natural number for which the addition period (Tadd) is closest to a half period (Tr / 2) of the mechanical resonance period of the alternating current rotating electric machine.

Description

TECHNICAL FIELD

[0001] The present application relates to a control device for an alternating-current rotary electric machine and an electric power steering device. BACKGROUND

[0002] As a control device for an alternating-current rotary electric machine and an electric power steering device, the technology of Patent Literature 1 is known. In Patent Literature 1, a current is detected based on an output signal of a current detection resistor element connected in series to a switching element on the negative side of an inverter. In the technology of Patent Literature 1, for a carrier cycle, a current is detected at two or more fixed timings, and the current is acquired so as not to include an error caused by switching noise, and the three-phase voltage command value is offset so as to coincide with a half value Vdc / 2 of a power supply voltage, as shown in FIGS. 23, 24, 25, and 26. Figure 12 、 Figure 13 、 Figure 14 、 Figure 15 、 Figure 16

[0003] PRIOR ART DOCUMENTS

[0004] PATENT LITERATURE

[0005] Patent Literature 1: Japanese Patent No. 6266161 (Claim 1, Claim 2, Figure 12 ) SUMMARY

[0006] PROBLEMS TO BE SOLVED BY THE INVENTION

[0007] However, in the technology of Patent Literature 1, although it is possible to reduce an error in current detection, in the inverter in which the current detection resistor element is connected in series to the switching element on the negative side, if the three-phase voltage command value is offset so as to coincide with the half value Vdc / 2 of the power supply voltage, there is a problem that a third harmonic component superimposed on an output torque of the alternating-current rotary electric machine becomes large. Therefore, in order to reduce the error in current detection while suppressing an increase in the third harmonic torque component, it is necessary to study a method different from that of Patent Literature 1.

[0008] However, among various frequencies included in the error component of the current detection value, an error component of a frequency close to a mechanical resonance frequency of the alternating-current rotary electric machine easily becomes noise of the alternating-current rotary electric machine. Therefore, it is particularly desirable to reduce an error component of a resonance period included in the current detection value.

[0009] Therefore, there is a need for a control device for an alternating-current rotary electric machine and an electric power steering device capable of reducing an error component of a current detection value close to a mechanical resonance period of the alternating-current rotary electric machine.​

[0010] Technical means for solving the technical problem

[0011] The control device for an alternating-current rotary electric machine according to the present application is a control device for an alternating-current rotary electric machine that controls an alternating-current rotary electric machine having a three-phase winding, and includes:

[0012] an inverter provided with three sets of series circuits corresponding to each phase of the three phases, the series circuits connecting in series a positive-side switching element connected to a positive side of a direct-current power supply and a negative-side switching element connected to a negative side of the direct-current power supply, and connecting a connection point of the series connection to the winding of the corresponding phase;

[0013] a current detection circuit having a resistance connected in series to the negative-side switching element of at least two phases; and

[0014] a controller that calculates three-phase voltage command values, and performs on-off control of the switching elements by comparing each of the three-phase voltage command values with a carrier that vibrates at a carrier period,

[0015] the controller, at a current detection period that is a first natural number multiple of the carrier period during which the negative-side switching element is on, detects a current flowing through the three-phase winding based on an output signal of the current detection circuit, performs current addition processing by adding a current detection value detected this time to a current detection value detected before a second natural number multiple of the current detection period, that is, an addition period, calculates a current detection value after the current addition processing, and calculates the three-phase current command values based on the current detection value after the current addition processing,

[0016] the second natural number is set to a natural number at which the addition period is closest to a half period of a mechanical resonance period of the alternating-current rotary electric machine.

[0017] Further, the electric power assisted steering device according to the present application includes:

[0018] the above-described control device for an alternating-current rotary electric machine;

[0019] the alternating-current rotary electric machine; and

[0020] a device that transmits a driving force of the alternating-current rotary electric machine to a driving force transmission mechanism of a steering device of a vehicle,

[0021] the carrier period is set to less than 60 μs,

[0022] the mechanical resonance period of the alternating-current rotary electric machine is in a range of 200 μs to 500 μs.

[0023] Effects of the Invention

[0024] The phase of the resonance period component of the current detection value is reversed at the half period of the resonance period. Therefore, by adding the current detection value detected this time and the current detection value detected before the half period of the resonance period, the resonance period component of the current detection value can be canceled. According to the control device of the alternating-current rotary electric machine and the electric power steering device according to the present application, the second natural number is set to a natural number closest to the half period of the mechanical resonance period of the alternating-current rotary electric machine. Therefore, by adding the current detection value detected this time and the current detection value detected before the addition period, the resonance period component of the current detection value can be canceled. As a result, the noise of the alternating-current rotary electric machine can be suppressed. BRIEF DESCRIPTION OF DRAWINGS

[0025] Figure 1 is a brief configuration diagram of the alternating-current rotary electric machine and the control device of the alternating-current rotary electric machine according to Embodiment 1.

[0026] Figure 2 is a hardware configuration diagram of the controller according to Embodiment 1.

[0027] Figure 3 is a timing chart showing the operation of the three-phase voltage command value obtained by amplitude-reduction modulation according to Embodiment 1.

[0028] Figure 4 is a timing chart showing the operation of the output torque obtained by amplitude-reduction modulation according to the comparative example.

[0029] Figure 5 is a timing chart showing the operation of the output torque obtained by amplitude-reduction modulation according to Embodiment 1.

[0030] Figure 6 is a timing chart for explaining the PWM control operation and the current detection timing according to Embodiment 1.

[0031] Figure 7 is a timing chart for explaining the PWM control operation and the current detection timing according to Embodiment 1.

[0032] Figure 8 is a timing chart for explaining the influence of ringing according to Embodiment 1.

[0033] Figure 9 is a timing chart for explaining the PWM control operation and the current detection timing in a case where the influence of ringing occurs according to Embodiment 1.

[0034] Figure 10 is a timing chart showing the control operation in a case where the current addition processing according to the comparative example is not performed.

[0035] Figure 11 FIG. 1 is a graph showing a frequency characteristic of noise of an alternating-current rotary electric machine according to Embodiment 1.

[0036] Figure 12 FIG. 2 is a timing chart for explaining a component of a resonance period and an addition period included in a current detection value according to Embodiment 1.

[0037] Figure 13 FIG. 3 is a block diagram of a current addition processing section according to Embodiment 1.

[0038] Figure 14 FIG. 4 is a flowchart for explaining processing of the current addition processing section according to Embodiment 1.

[0039] Figure 15 FIG. 5 is a block diagram of the current addition processing section according to Embodiment 1.

[0040] Figure 16 FIG. 6 is a timing chart for explaining a component of a resonance period and an addition period included in a current detection value according to Embodiment 3.

[0041] Figure 17 FIG. 7 is a block diagram of a current addition processing section according to Embodiment 3.

[0042] Figure 18 FIG. 8 is a graph showing a transfer characteristic corresponding to a noise component of a current detection value according to Comparative Example.

[0043] Figure 19 FIG. 9 is a graph showing a transfer characteristic corresponding to a noise component of a current detection value according to Embodiment 3.

[0044] Figure 20 FIG. 10 is a schematic structural diagram of an electric power steering apparatus according to Embodiment 4.

[0045] Figure 21 FIG. 11 is a timing chart showing an operation of a three-phase voltage command value obtained by amplitude reduction modulation according to other embodiments. DETAILED DESCRIPTION

[0046] 1. Embodiment 1

[0047] A control device 10 (hereinafter simply referred to as control device 10) according to Embodiment 1 will be described with reference to the accompanying drawings. Figure 1 FIG. 1 is a schematic structural diagram of an alternating-current rotary electric machine 1 and the control device 10 according to the present embodiment.

[0048] 1-1. Alternating-current rotary electric machine 1

[0049] The alternating-current rotary electric machine 1 has three-phase windings Cu, Cv, Cw of U phase, V phase, and W phase. The alternating-current rotary electric machine 1 includes a stator and a rotor arranged on a radially inner side of the stator. The stator is provided with the three-phase windings Cu, Cv, Cw. In the present embodiment, the rotor is provided with a permanent magnet, and is a synchronous rotary electric machine of the permanent magnet type. Alternatively, the alternating-current rotary electric machine 1 can be a magnetic-field winding type synchronous rotary electric machine provided with an electromagnet on the rotor, or an induction electric machine provided with no permanent magnet on the rotor. The three-phase windings can be connected in a star configuration or in a delta configuration.

[0050] The rotor includes a rotation detection circuit 2 for detecting a rotation angle of the rotor. The rotation detection circuit 2 uses a resolver, an encoder, an MR sensor, or the like. An output signal of the rotation detection circuit 2 is input to the controller 6.

[0051] 1-2. Inverter 4

[0052] The inverter 4 is provided with three sets of series circuits (legs) corresponding to the three phases, respectively. The series circuits are connected in series with positive-side switching elements SP connected to a positive side of the direct-current power supply 3 and negative-side switching elements SN connected to a negative side of the direct-current power supply 3. Then, connection points of two switching elements in each phase series circuit are connected to the winding of the corresponding phase.

[0053] Specifically, in the series circuit of the U phase, the U phase positive-side switching element SPu and the U phase negative-side switching element SNu are connected in series, and the connection point of the two switching elements is connected to the U phase winding Cu. In the series circuit of the V phase, the V phase positive-side switching element SPv and the V phase negative-side switching element SNv are connected in series, and the connection point of the two switching elements is connected to the V phase winding Cv. In the series circuit of the W phase, the W phase positive-side switching element SPw and the W phase negative-side switching element SNw are connected in series, and the connection point of the two switching elements is connected to the W phase winding Cw.

[0054] As the switching elements, IGBTs (Insulated Gate Bipolar Transistors) connected in antiparallel with diodes, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors), bipolar transistors connected in antiparallel with diodes, or the like are used. The gate terminals of the switching elements are connected to the controller 6 via a gate drive circuit or the like. The switching elements are turned on or off by switching signals Gpu to GNw output from the controller 6.

[0055] The DC power supply 3 outputs a DC voltage Vdc to the inverter 4. The DC power supply 3 can be any device that outputs a power supply voltage Vdc, such as a battery, DC-DC converter, diode rectifier, or PWM rectifier. The DC power supply 3 is equipped with a voltage sensor to detect the power supply voltage Vdc, and the output signal of the voltage sensor can be input to the control device 10. The control device 10 can use the detected power supply voltage Vdc for control purposes.

[0056] 1-3. Current Detection Circuit 5

[0057] The system includes a current detection circuit 5, which has resistors connected in series with the negative-side switching elements of at least two phases. In this embodiment, the current detection circuit 5 is configured to detect the current flowing through the negative-side switching elements SNu, SNv, and SNw of the three phases. The current detection circuit 5 has shunt resistors 5u, 5v, and 5w connected in series with the negative-side switching elements of each phase. The U-phase shunt resistor 5u is connected in series with the negative side of the U-phase negative-side switching element SNu, the V-phase shunt resistor 5v is connected in series with the negative side of the V-phase negative-side switching element SNv, and the W-phase shunt resistor 5w is connected in series with the negative side of the W-phase negative-side switching element SNw. The potential differences VRu, VRv, and VRw across the shunt resistors 5u, 4v, and 5w of each phase are input to the controller 6.

[0058] Alternatively, each shunt resistor can be connected in series on the positive side of the switching element on the negative side. Furthermore, the current detection circuit 5 can be configured to detect the current flowing through the negative side switching elements of any two phases. In this case, since the sum of the three-phase winding currents is zero, the controller 6 can also calculate the current of the remaining phase based on the detected values ​​of two phase currents. For example, if the current detection circuit 5 detects the currents Iur and Ivr of phases U and V, the controller 6 can also calculate the current Iwr of phase W using Iwr = -Iur - Ivr.

[0059] 1-4. Controller 6

[0060] Controller 6 controls AC rotating motor 1 via inverter 4. For example... Figure 1 As shown, the controller 6 includes a rotation detection unit 31, a current detection unit 32, a current coordinate conversion unit 33, a current addition processing unit 34, a current command value calculation unit 35, a voltage command value calculation unit 36, and a PWM control unit 37. Each function of the controller 6 is implemented by its processing circuitry. Specifically, the controller 6... Figure 2As shown, the processing circuit includes: an arithmetic processing unit 90 (computer) such as a CPU (Central Processing Unit); a storage device 91 that exchanges data with the arithmetic processing unit 90; an input circuit 92 that inputs external signals to the arithmetic processing unit 90; and an output circuit 93 that outputs signals from the arithmetic processing unit 90 to the outside.

[0061] The arithmetic processing unit 90 can include ASIC (Application Specific Integrated Circuit), IC (Integrated Circuit), DSP (Digital Signal Processor), FPGA (Field Programmable Gate Array), various logic circuits, and various signal processing circuits. Furthermore, the arithmetic processing unit 90 can also include multiple arithmetic processing units of the same or different types to share the execution of various processes. The storage device 91 can include RAM (Random Access Memory) configured to read and write data from the arithmetic processing unit 90, ROM (Read Only Memory) configured to read data from the arithmetic processing unit 90, etc. The input circuit 92 is connected to various sensors and switches such as the rotation detection circuit 2 and the current detection circuit 5, and includes an A / D converter that inputs the output signals of these sensors and switches to the arithmetic processing unit 90. The output circuit 93 is connected to electrical loads such as gate drive circuits that drive the switching elements to turn on and off, and includes drive circuits that output control signals from the arithmetic processing unit 90 to these electrical loads.

[0062] Then, the controller 6 has the following features Figure 1 Each function of the control units 31 to 37, etc., is implemented by the arithmetic processing unit 90 executing the software (program) stored in the storage device 91 such as ROM, and cooperating with other hardware of the controller 6 such as the storage device 91, the input circuit 92, and the output circuit 93. Furthermore, the gain, threshold, and other setting data used by each control unit 31 to 37, etc., are stored as part of the software (program) in the storage device 91 such as ROM. The functions of the controller 6 will be described in detail below.

[0063] <Rotation Detection Unit 31>

[0064] The rotation detection section 31 detects the magnetic pole position θ (rotational angle θ) of the rotor at the electrical angle and the rotational angular velocity ω. In the present embodiment, the rotation detection section 31 detects the magnetic pole position θ (rotational angle θ) of the rotor and the rotational angular velocity ω based on the output signal of the rotation detection circuit 2. The magnetic pole position is set to the direction of the N pole of the permanent magnet provided on the rotor. In addition, the rotation detection section 31 can be configured to estimate the rotational angle (magnetic pole position) based on current information obtained by superimposing a high-order harmonic component on a current command value or the like without using a rotation sensor (so-called sensorless system).

[0065] <Current detection section 32>

[0066] The current detection section 32 detects the currents Iur, Ivr, and Iwr flowing through the three-phase windings based on the output signal of the current detection circuit 5. The current detection section 32 detects the currents Iur, Ivr, and Iwr of the respective phase windings by dividing the potential difference across the shunt resistor of each phase by the resistance value of the shunt resistor.

[0067] Since the current detection circuit 5 detects the current flowing through the negative-side switching element, the current detection section 32 detects the current at the timing when the negative-side switching element is turned on. Further, in the PWM control section 37 described later, the negative-side switching element is turned on every carrier cycle Tc. Thus, the current detection section 32 detects the currents Iur, Ivr, and Iwr of the three-phase windings based on the output signal of the current detection circuit 5 at the current detection period TIdt (= A x Tc) that is the period of the first natural number A times of the carrier cycle Tc in which the negative-side switching element is turned on. In the present embodiment, the current detection circuit 5 detects the current at the timing of every current detection period TIdt in which the carrier CA is at the top of the peak.

[0068] <Current coordinate conversion section 33>

[0069] The current coordinate conversion section 33 converts the current detection values Iur, Ivr, and Iwr of the three-phase windings into the d-axis current detection value Idr and the q-axis current detection value Iqr on the d-axis and q-axis coordinate system every time the current is detected. The d-axis and q-axis coordinate system is a two-axis rotating coordinate system that rotates in synchronization with the magnetic pole position of the rotor. The d-axis is decided in the direction of the magnetic pole position θ (N pole), and the q-axis is decided in the direction that advances by 90 degrees on the electrical angle from the d-axis. Specifically, the current coordinate conversion section 33 performs three-phase to two-phase conversion and rotational coordinate conversion on the three-phase current detection values Iur, Ivr, and Iwr based on the magnetic pole position θ, and converts them into the d-axis current detection value Idr and the q-axis current detection value Iqr.

[0070] <Current addition processing section 34>

[0071] The details will be described later, but the current addition processing section 34 performs current addition processing on the current detection values at each current detection, and calculates the current detection values after the current addition processing. In the present embodiment, the current addition processing section 34 performs current addition processing on the d-axis and q-axis current detection values Idr, Iqr, and calculates the d-axis and q-axis current detection values Idr*, Iqr* after the current addition processing.

[0072] <Current command value calculation section 35>

[0073] The current command value calculation section 35 calculates the d-axis current command value Ido and the q-axis current command value Iqo. In the present embodiment, the current command value calculation section 35 calculates the d-axis and q-axis current command values Ido and Iqo based on the torque command value Tref, the power supply voltage Vdc, the rotational angular velocity ω, and the like. The d-axis and q-axis current command values Ido and Iqo are calculated according to known current vector control methods such as maximum torque current control, maximum torque voltage control, flux-weakening control, and Id = 0 control. For example, when Id = 0 control is performed, the current command value Ido on the d-axis is set to 0 (Ido = 0), and the current command value Iqo on the q-axis is set to a value obtained by multiplying the torque command value Tref by a conversion coefficient. The torque command value Tref can be calculated within the controller 6 or can be transmitted from an external control device.

[0074] <Voltage command value calculation section 36>

[0075] The voltage command value calculation section 36 changes the three-phase voltage command values Vuo, Vvo, and Vwo at each current detection so that the current detection values approach the current command values. In the present embodiment, the voltage command value calculation section 36 includes a dq-axis voltage command value calculation section 361, a voltage coordinate conversion section 362, and a modulation section 363.

[0076] The dq-axis voltage command value calculation section 361 performs current feedback control that changes the d-axis voltage command value Vdo and the q-axis voltage command value Vqo through PI control or the like so that the d-axis current detection value Idr* after the current addition processing approaches the d-axis current command value Ido, and so that the q-axis current detection value Iqr* after the current addition processing approaches the q-axis current command value Iqo. In addition, feedforward control can be performed so that the d-axis current and the q-axis current are not disturbed, or the like.

[0077] The voltage coordinate conversion section 362 performs fixed coordinate conversion and two-phase / three-phase conversion on the d-axis and q-axis voltage command values Vdo, Vqo based on the magnetic pole position θ, and converts them into the three-phase voltage command values Vuoc, Vvoc, and Vwoc after the coordinate conversion. The three-phase voltage command values Vuoc, Vvoc, and Vwoc after the coordinate conversion are sine waves.

[0078] The modulation section 363 applies modulation to the coordinate-transformed three-phase voltage command values Vuoc, Vvoc, Vwoc to calculate final three-phase voltage command values Vuo, Vvo, Vwo. The modulation section 363 applies amplitude-reducing modulation to the coordinate-transformed three-phase voltage command values, which reduces the amplitude of the three-phase voltage command values while maintaining the line-to-line voltage of the three-phase voltage command values.

[0079] In the present embodiment, the modulation section 363 applies amplitude-reducing modulation so that the average of the three-phase voltage command values is smaller than the center of oscillation of the carrier CA, so that the three-phase voltage command values are offset equally. The modulation section 363 calculates a bias voltage Voff by determining the minimum value Vmin of the coordinate-transformed three-phase voltage command values Vuoc, Vvoc, Vwoc, adding half the value of the power supply voltage Vdc to the minimum value Vmin, and subtracting the bias voltage Voff from the coordinate-transformed three-phase voltage command values Vuoc, Vvoc, Vwoc to calculate the three-phase voltage command values Vuo, Vvo, Vwo, as shown in the following equations.

[0080] Vmin = MIN(Vuoc, Vvoc, Vwoc)

[0081] Voff = 0.5 x Vdc + Vmin

[0082] Vuo = Vuoc - Voff (1)

[0083] Vvo = Vvoc - Voff

[0084] Vwo = Vwoc - Voff

[0085] The operation of the amplitude-reducing modulation of equation (1) is shown in FIG. 6. In the graph of the upper section, the coordinate-transformed three-phase voltage command values Vuoc, Vvoc, Vwoc are shown. The coordinate-transformed three-phase voltage command values Vuoc, Vvoc, Vwoc exceed the range from -Vdc / 2 to +Vdc / 2, resulting in voltage saturation. On the other hand, in the three-phase voltage command values Vuo, Vvo, Vwo after the amplitude-reducing modulation, the voltage command value of the phase with the minimum voltage is offset to coincide with -Vdc / 2, thereby preventing the occurrence of voltage saturation. Figure 3

[0086] <Difference in Torque Variation According to Modulation Method>

[0087] As in Patent Document 1, Figure 4 the operation of the output torque involved in the comparative example in which the voltage command value of the phase with the maximum voltage is offset to coincide with +Vdc / 2 is shown. On the other hand, Figure 5 ​The operation of the output torque when the voltage command value of the phase for which the offset is made to minimize the voltage in this embodiment coincides with -Vdc / 2 is shown in Figure 4 and Figure 5 The modulated three-phase voltage command values Vuo, Vvo, Vwo are shown in the upper graph of

[0088] In Figure 4 the comparative example and Figure 5 the output torque also has a third harmonic ripple component superimposed. However, in the case of the comparative example of Figure 4 the amplitude of the third harmonic ripple component is about three times larger than in the case of the present embodiment. This is because, in the present embodiment, the voltage drop in the inverter when the negative-side switching element is on is larger than the voltage drop in the inverter when the positive-side switching element is on, making them unbalanced. Although detailed explanation is omitted, as a result of this unbalance, the amplitude of the third harmonic ripple component becomes large in the case of the comparative example like Patent Document 1, so the modulation method of the present embodiment is desirable. However, as will be described later, in the modulation method of the present embodiment, the on period of the negative-side switching element is sometimes shortened, and a current detection error occurs due to ringing, so this countermeasure is taken by the current addition process.

[0089] <PWM control section 37>

[0090] The PWM control section 37 performs on-off control of the switching elements by comparing each of the three-phase voltage command values Vuo, Vvo, Vwo with a carrier CA that oscillates with a carrier period Tc. The carrier CA is set to a triangular wave that oscillates with an amplitude of half the power supply voltage VDC / 2 centered on 0 in the carrier period Tc.

[0091] As Figure 6As shown, for each phase, when the carrier CA is lower than the voltage command value, the PWM control unit 37 turns on the switching signal GP (1 in this example) of the positive-side switching element, turning on the positive-side switching element; when the carrier CA exceeds the voltage command value, it turns off the switching signal GP (0 in this example) of the positive-side switching element, turning off the positive-side switching element. On the other hand, when the carrier CA is lower than the voltage command value, the PWM control unit 37 turns off the switching signal GN (0 in this example) of the negative-side switching element, turning off the negative-side switching element; when the carrier CA exceeds the voltage command value, it turns on the switching signal GN (1 in this example) of the negative-side switching element, turning on the negative-side switching element. Furthermore, in each phase, a short-circuit prevention period (dead time) is provided between the on-time of the positive-side switching element and the on-time of the negative-side switching element to turn off both the positive-side and negative-side switching elements.

[0092] like Figure 6 As shown, in the interval D centered at the peak of the carrier wave CA, all three-phase negative-side switching signals GNU, GNv, and GNw are turned on. In this interval D, the current detection circuit 5 can detect the current flowing through the three-phase windings. In this embodiment, as described above, the current detection unit 32 is configured to detect the current at a timing point at the peak of the carrier wave CA. Figure 6 In the example, the first natural number A is set to 1, the current detection period TIdt is set to one time the period of the carrier period Tc, and the current is detected at the apex of the peaks of all carrier CA.

[0093] Or it could be like Figure 7 As shown, the first natural number A is set to 2, the current detection period TIdt is set to twice the period of the carrier period Tc, and the current is detected at the peak of the peak of every carrier CA.

[0094] <Current detection error caused by ringing>

[0095] like Figure 8 As shown, when a switching element is turned on, current oscillation occurs immediately. Ringing in one phase can also affect the current in other phases. For example, ringing occurs within a few μs after the switching element is turned on. As the voltage command value increases, the on-time of the switching element on the negative side shortens. When the on-time is less than twice the ringing period, the current detection timing overlaps with the ringing, causing detection errors in the current detection values ​​of each phase.

[0096] For example, in Figure 9In the example of FIG. 6, the U-phase voltage command value Vuo becomes a large value close to half the value Vdc / 2 of the power supply voltage. As a result, the on period of the negative-side switch signal GNu of the U phase becomes short, and the U-phase current detection value Iur generates a detection error due to ringing. In addition, due to the influence of the U-phase current ringing, the other V-phase and W-phase current detection values Ivr, Iwr also generate detection errors.

[0097] In addition, the current detection section 32 can also calculate the current detection value of the phase whose on period of the negative-side switching element is shorter than the threshold value (for example, 5 μs) based on the current detection values in the other two phases, in the case where the on period of the negative-side switching element of an arbitrary phase is shorter than the threshold value. For example, in the case where the on period of the negative-side switch signal GNu of the U phase is shorter than the threshold value, the U-phase current detection value Iur can also be calculated by Iur = -Ivr - Iwr. Even in this case, since the current detection values of the other two phases also generate detection errors due to ringing, the detection error caused by ringing cannot be made zero.

[0098] Figure 10 The control operation in a comparative example in which the current addition processing section 34 does not perform current addition processing is shown. As the rotational angular velocity increases, the amplitude of the three-phase voltage command value also increases. Until time t01, the amplitude maximum value of the three-phase voltage command value is lower than half the value Vdc / 2 of the power supply voltage, and no current detection error is generated due to ringing, and the q-axis current detection value Iqr is less likely to generate a noise component. From time t01 to time t02, as the amplitude maximum value of the three-phase voltage command value approaches half the value Vdc / 2 of the power supply voltage, the current detection error caused by the influence of ringing gradually increases, and the noise component of the q-axis current detection value Iqr gradually increases.

[0099] After time t03, in order to observe the operation in which noise is generated, the increase in the rotational angular velocity is stopped, and it is set to a constant condition. The lower side shows a graph in which a prescribed period after time t03 is enlarged. As shown in the graph, the maximum value of the modulated three-phase voltage command value reaches half the value Vdc / 2 of the power supply voltage, and a state in which the on period of the negative-side switching element becomes short and a current detection error caused by ringing is generated is generated. As a result, it is known that the noise component of the q-axis current detection value Iqr obtained by coordinate conversion of the three-phase current detection value becomes large. In addition, although not shown, similarly, the noise component of the d-axis current detection value Idr also becomes large.

[0100] Then, when current feedback control is performed using the d-axis and q-axis current detection values Idr, Iqr to which noise components are added, the noise components are added to the three-phase voltage command values Vuo, Vvo, Vwo, and the ripple component is added to the output torque of the alternating-current rotating electrical machine 1.

[0101] As a countermeasure against the noise component of the current detection value caused by ringing, the following (A) and (B) are considered.

[0102] (A) The response speed of the current feedback control on the dq axes is reduced, the sensitivity of the current detection values Idr, Iqr on the d and q axes to the noise component is reduced, and the noise component is prevented from being superimposed on the three-phase voltage command values.

[0103] (B) As shown in FIG. 6, contrary to the present embodiment, modulation is applied in which the voltage command value of the maximum phase of the three-phase voltage command values Vuoc, Vvoc, Vwoc after coordinate transformation coincides with the half value Vdc / 2 of the power supply voltage. Figure 4

[0104] In the countermeasure (A), if the response speed of the current feedback control is reduced, the follow-up performance of the current feedback control will deteriorate, and thus cannot be selected. In the countermeasure (B), since the negative-side switching element is always turned off during the period in which the three-phase voltage command values coincide with Vdc / 2, although ringing is not generated, as described above, in the inverter in which the shunt resistor is connected in series to the negative-side switching element, since the third-order torque ripple component becomes large, thus cannot be selected.

[0105] <Noise component of the current detection value with respect to mechanical resonance period of the AC rotary electric machine>

[0106] Therefore, in the present embodiment, the current addition processing of the current addition processing section 34 is performed instead of the countermeasures (A), (B). Figure 11 is a result of measurement of the sensitivity characteristic of the noise of the AC rotary electric machine 1 with respect to the noise component of the current. The larger the gain of the vertical axis, the larger the noise of the AC rotary electric machine 1 with respect to the amplitude of the noise component of the current. In this AC rotary electric machine 1, a peak exists near 2500 Hz. This is because the mechanical resonance frequency of the frame of the AC rotary electric machine 1 is near 2500 Hz (the resonance period Tr is 400 μs).

[0107] In addition, the mechanical resonance of the AC rotary electric machine 1 occurs in the frame of the AC rotary electric machine 1, or in the power pack in which the AC rotary electric machine 1, the inverter 4, and the controller 6 are integrated, or in a power plant or the like including the frame or the power pack and a gear mechanism.

[0108] Among the various frequencies included in the noise component of the current detection value, the noise component of the frequency close to the mechanical resonance frequency of the AC rotary electric machine 1 easily becomes the noise of the AC rotary electric machine 1. Thus, it is desirable to reduce the component of the mechanical resonance period Tr of the AC rotary electric machine 1 included in the current detection value.

[0109] <Current addition processing>

[0110] The principle of the current addition processing will be described.​Figure 12 A schematic waveform of the component of the resonance period Tr included in the current detection value is shown. The component of the resonance period Tr of the current detection value is phase-inverted at the half period Tr / 2 of the resonance period. Thus, by adding the current detection value detected this time and the current detection value detected before the half period of the resonance period, the component of the resonance period Tr of the current detection value can be cancelled.

[0111] Therefore, the current addition processing section 34 performs the current addition processing by adding the current detection value detected this time and the current detection value detected before the addition period Tadd (= B x TIdt) which is the period of the second natural number B times the current detection period TIdt, and thereby calculates the current detection value after the current addition processing. The second natural number B is set to a natural number which makes the addition period Tadd closest to the half period Tr / 2 of the mechanical resonance period of the alternating-current rotating electric machine 1.

[0112] Ideally, the second natural number B can be set to a natural number which makes the addition period Tadd coincide with the half period Tr / 2 of the resonance period, but if they do not coincide, the carrier period Tc needs to be changed. As described above, if the second natural number B is set to a natural number which makes the addition period Tadd closest to the half period Tr / 2 of the resonance period, the component of the resonance period Tr of the current detection value can be reduced depending on the closeness of the addition period Tadd to the half period Tr / 2 of the resonance period.

[0113] To reduce the effect, the addition period Tadd can be set to between the ¼ period Tr / 4 of the resonance period and the ¾ period Tr x 3 / 4 of the resonance period, as shown in the following equation.

[0114] Tr / 4 < Tadd < Tr x 3 / 4... (2)

[0115] For example, if the resonance period Tr is 400 μs, the carrier period Tc is 50 μs, and the first natural number A is 2, the current detection period TIdt is 100 μs, the second natural number B is set to 2, and the addition period Tadd is set to 200 μs which coincides with the half period 200 μs of the resonance period, as shown in the following equation. In addition, the addition period Tadd can not coincide with the half period Tr / 2 of the resonance period.

[0116] Tr = 400 μs

[0117] Tc = 50 μs... (3)

[0118] TIdt = A x Tc = 2 x 50 μs = 100 μs

[0119] Tadd = B x TIdt = 2 x 100 μs = 200 μs

[0120] In this embodiment, the current addition processing unit 34 performs current addition processing on the d-axis and q-axis current detection values ​​Idr and Iqr calculated in the current current detection and the current detection values ​​Idr and Iqr calculated in the current detection before the addition period Tadd, and calculates the d-axis and q-axis current detection values ​​Idr* and Iqr* after current addition processing.

[0121] In this embodiment, as shown in the following formula, the current addition processing unit 34 adds the d-axis current detection value Idr calculated in the current current detection to the d-axis current detection value Idr calculated in the current detection before the addition period Tadd, calculates the d-axis current addition value SIdr, and multiplies the d-axis current addition value SIdr by 0.5 to calculate the d-axis current detection value Idr* after current addition processing. As shown in the following formula, the current addition processing unit 34 adds the q-axis current detection value Iqr calculated in the current current detection to the q-axis current detection value Iqr calculated in the current detection before the addition period Tadd, calculates the q-axis current addition value SIqr, and multiplies the q-axis current addition value SIqr by 0.5 to calculate the q-axis current detection value Iqr* after current addition processing.

[0122] SIdr(t) = Idr(t) + Idr(t - Tadd)

[0123] Idr*(t)=0.5×SIdr(t)…(4)

[0124] SIqr(t) = Iqr(t) + Iqr(t - Tadd)

[0125] Iqr*(t) = 0.5 × SIqr(t)

[0126] The current addition process cancels out the component that is twice the period of the addition period Tadd, but doubles the components outside twice the period of the addition period Tadd. Therefore, by multiplying the addition value by 0.5, the d-axis and q-axis current detection values ​​Idr* and Iqr* after reducing the component that is twice the period of the addition period Tadd set corresponding to the resonant period Tr can be calculated.

[0127] In this embodiment, such as Figure 13 As shown, the current addition processing unit 34 includes a d-axis current delay unit 34a, a d-axis current adder 34b, a d-axis current gain multiplier 34c, a q-axis current delay unit 34d, a q-axis current adder 34e, and a q-axis current gain multiplier 34f. The current addition processing unit 34 is a discrete controller that operates according to each current detection cycle Tidt, and its operation cycle is the current detection cycle TIdt.

[0128] The d-axis current detection value Idr, detected in the current operation cycle (current detection cycle TIdt), is input into the d-axis current delay unit 34a and the d-axis current adder 34b. The d-axis current delay unit 34a delays the input d-axis current detection value Idr by the addition cycle Tadd and outputs it. The d-axis current delay unit 34a has a second natural number B delay units and outputs the d-axis current detection value Idr input before the second natural number B operation cycle (current detection cycle TIdt) in the current operation cycle. The d-axis current adder 34b adds the d-axis current detection value Idr detected in the current operation cycle to the d-axis current detection value Idr detected before the addition cycle Tadd output from the d-axis current delay unit 34a, outputting the d-axis current addition value SIdr. The d-axis current gain multiplier 34c multiplies the d-axis current addition value SIdr by a gain of 0.5 to output the current addition-processed d-axis current detection value Idr*.

[0129] The q-axis current delay unit 34d and the q-axis current adder 34e are input to the q-axis current detection value Iqr detected in the current operation cycle (current detection cycle TIdt). The q-axis current delay unit 34d delays the input q-axis current detection value Iqr by an addition cycle Tadd and outputs it. The q-axis current delay unit 34d has a second natural number B delay units and outputs the q-axis current detection value Iqr input before the second natural number B operation cycle (current detection cycle TIdt) in the current operation cycle. The q-axis current adder 34e adds the q-axis current detection value Iqr detected in the current operation cycle to the q-axis current detection value Iqr detected before the addition cycle Tadd, outputting the q-axis current addition value SIqr. The q-axis current gain multiplier 34f multiplies the q-axis current addition value SIqr by a gain of 0.5 to output the current-added q-axis current detection value Iqr*.

[0130] 2. Implementation Method 2

[0131] The control device 10 according to Embodiment 2 will be described. Descriptions of structural parts identical to those in Embodiment 1 are omitted. The basic structure of the AC rotary motor 1 and control device 10 according to this embodiment is the same as in Embodiment 1, but the current addition processing differs from that in Embodiment 1.

[0132] In Implementation 1, the current addition process is always performed. However, in this embodiment, the current addition process is performed only when necessary.

[0133] If the conduction period of the negative-side switching element of any phase is lower than a preset threshold T_th, the current addition processing unit 34 calculates the three-phase voltage command value based on the current detection value after current addition processing. If the conduction period of the negative-side switching element of any phase is not lower than the threshold T_th, the current addition processing unit 34 calculates the three-phase voltage command value based on the detected current detection value.

[0134] <Flowchart>

[0135] For example, the current addition processing unit 34, such as Figure 14 The flowchart shown is configured as follows. In step S21, as shown in the following formula, the current addition processing unit 34 determines the minimum value Ton_min among T_Gnu, T_Gnv, and T_Gnw during the carrier period Tc of the switching element on the negative side of each of the three phases.

[0136] Ton_min=MIN(T_Gnu,T_Gnv,T_Gnw)…(5)

[0137] For example, the current addition processing unit 34 calculates the conduction periods T_Gnu, T_Gnv, and T_Gnw of the negative side switching elements of each phase based on the three-phase voltage command values ​​Vuo, Vvo, Vwo and the power supply voltage Vdc.

[0138] Then, in step S22, the current addition processing unit 34 determines whether the minimum value Ton_min is lower than the threshold T_th. If the minimum value Ton_min is lower than the threshold T_th, it proceeds to step S23; otherwise, it proceeds to step S24. In step S23, the current addition processing unit 34 outputs the current detection values ​​Idr* and Iqr* of the d-axis and q-axis, calculated in the same manner as in Embodiment 1, to the voltage command value calculation unit 36. On the other hand, in step S24, the current addition processing unit 34 outputs the current detection values ​​Idr and Iqr of the d-axis and q-axis, which have not undergone current addition processing, to the voltage command value calculation unit 36.

[0139] For example, when the carrier period Tc = 50 μs, the threshold T_th is set in the range of 1.5 μs to 9.5 μs.

[0140] Thus, when the conduction period of the negative-side switching element of any phase is shorter than the threshold value T_th, the possibility that ringing occurs at the current detection timing is high, the current addition processing is performed, the component of the resonance period Tr is reduced, and thus the noise is reduced. On the other hand, when the conduction period of the negative-side switching element of all phases is longer than the threshold value T_th, the possibility that ringing occurs at the current detection timing is low, and thus the disturbance of the current detection value is small. Thus, the current addition processing of adding the past current detection value can not be performed, and thus the phase of the current detection value is not delayed.

[0141] <block diagram>

[0142] The current addition processing section 34 is expressed in a block diagram as shown in Figure 15 . The current addition processing section 34 has a switching determiner 34g, a d-axis current switch 34h, and a q-axis current switch 34i in addition to the d-axis current Figure 13 delay 34a to the q-axis current gain multiplier 34f as shown. The d-axis current delay 34a to the q-axis current gain multiplier 34f are the same as those of Embodiment 1, and thus the description is omitted.

[0143] The switching determiner 34g performs the same processing as the step S21 and the step S22 of Figure 14 , and determines whether the conduction period within the carrier period Tc of the negative-side switching element of any phase is lower than the threshold value T_th.

[0144] In a case where it is determined in the switching determiner 34g that it is lower than the threshold value T_th, the d-axis current switch 34h outputs the current addition-processed d-axis current detection value Idr* output from the d-axis current gain multiplier 34c, and in a case where it is determined in the switching determiner 34g that it is not lower than the threshold value T_th, the d-axis current switch 34h outputs the d-axis current detection value Idr detected in the present operation period. The output value of the d-axis current switch 34h is input to the voltage command value calculation section 36.

[0145] In a case where it is determined in the switching determiner 34g that it is lower than the threshold value T_th, the q-axis current switch 34i outputs the current addition-processed q-axis current detection value Iqr* output from the q-axis current gain multiplier 34f, and in a case where it is determined in the switching determiner 34g that it is not lower than the threshold value T_th, the q-axis current switch 34i outputs the q-axis current detection value Iqr detected in the present operation period. The output value of the q-axis current switch 34i is input to the voltage command value calculation section 36.

[0146] 3. Embodiment 3

[0147] The control device 10 according to Embodiment 3 will be described. Descriptions of structural parts identical to those in Embodiment 1 are omitted. The basic structure of the AC rotary motor 1 and control device 10 according to this embodiment is the same as in Embodiment 1, but the current addition processing differs from that in Embodiment 1.

[0148] When the addition period Tadd is inconsistent with the half-period Tr / 2 of the resonant period, in the method of Implementation Method 1, it is sometimes impossible to sufficiently reduce the component of the resonant period Tr from the current detection value. For example, Figure 16 The diagram shows a schematic waveform of the resonant period Tr component included in the current detection value. In this example, the mechanical resonant period Tr of the AC rotating motor 1 is 330 μs, and the current detection period TIdt is 100 μs. The second natural number B is set to 2 so that the addition period Tadd is closest to the half-period of the resonant period, 165 μs, and the addition period Tadd is set to twice the current detection period TIdt, i.e., 200 μs. The addition period Tadd does not match the half-period of the resonant period Tr / 2, with a difference of 35 μs.

[0149] Therefore, in this embodiment, the current addition processing unit 34 adds the current detection value detected this time, the current detection value detected before the addition period Tadd, and the current detection value detected before the additional addition period Taddad (which is a multiple of the third natural number C of the current detection period TIdt). The unit then calculates the current detection value after the current addition processing. The third natural number C is set to a natural number that makes the additional addition period Taddad close to half the mechanical resonance period Tr / 2 of the AC rotating motor.

[0150] exist Figure 16 In the example, the third natural number C is set to 1 so that the second half-cycle of the addition period Taddad, which is close to the resonant period, is 165μs. The addition period Taddad is set to 1 times the current detection period TIdt, which is 100μs.

[0151] Therefore, even when the addition period Tadd is inconsistent with the half-period Tr / 2 of the resonant period, the half-period Tr / 2 of the resonant period can be sandwiched between the addition period Tadd and the additional addition period Taddad. By adding the current detection value detected before the addition period Tadd with the current detection value detected before the additional addition period Taddad, the component of the resonant period Tr of the current detection value can be eliminated more effectively.

[0152] In this embodiment, as shown in the following formula, the current addition processing unit 34 adds the d-axis current detection value Idr calculated in the current current detection, the d-axis current detection value Idr calculated in the current detection before the addition period Tadd, and the d-axis current detection value Idr calculated in the current detection before the additional addition period Taddad by multiplying them by the gain Kad to calculate the d-axis current addition value SIdr. The d-axis current addition value SIdr is then multiplied by 1 / (2+Kad) to calculate the d-axis current detection value Idr* after current addition processing. As shown in the following formula, the current addition processing unit 34 adds the q-axis current detection value Iqr calculated in the current current detection, the q-axis current detection value Iqr calculated in the current detection before the addition period Tadd, and the q-axis current detection value Iqr calculated in the current detection before the additional addition period Taddad, multiplied by the gain Kad to calculate the q-axis current addition value SIqr. The q-axis current addition value SIqr is then multiplied by 1 / (2+Kad) to calculate the q-axis current detection value Iqr* after current addition processing. The gain Kad is adjusted to ensure good elimination effect.

[0153] SIdr(t) = Idr(t) + Idr(t - Tadd)

[0154] +Kad×Idr(t-Taddad)

[0155] Idr*(t)=1 / (2+Kad)×SIdr(t)…(6)

[0156] SIqr(t) = Iqr(t) + Iqr(t - Tadd)

[0157] +Kad×Iqr(t-Taddad)

[0158] Iqr*(t)=1 / (2+Kad)×SIqr(t)

[0159] In this embodiment, such as Figure 17 As shown, the current addition processing unit 34 includes a d-axis current delay unit 34j, a d-axis current addition delay unit 34k, a gain multiplier for the d-axis addition delay unit 34l, a first d-axis current adder 34m, a second d-axis current adder 34n, a gain multiplier for the d-axis current 34o, a q-axis current delay unit 34p, a q-axis current addition delay unit 34q, a gain multiplier for the q-axis addition delay unit 34r, a first q-axis current adder 34s, a second q-axis current adder 34t, and a gain multiplier for the q-axis current 34u. The current addition processing unit 34 is a discrete controller that operates according to each current detection cycle Tidt, and its operation cycle is the current detection cycle TIdt.

[0160] In the d-axis current delay 34j and the d-axis current additional delay 34k and the d-axis current first adder 34m, the d-axis current detection value Idr detected in the present operation cycle (current detection cycle TIdt) is input. The d-axis current delay 34j delays the input d-axis current detection value Idr by the addition cycle Tadd and outputs. The d-axis current delay 34j has a second natural number B number of delays, and outputs the d-axis current detection value Idr input before the second natural number B number of operation cycles (current detection cycle TIdt) in the present operation cycle. The d-axis current additional delay 34k delays the input d-axis current detection value Idr by the additional addition cycle Tadd and outputs. The d-axis current additional delay 34k has a third natural number C number of delays, and outputs the d-axis current detection value Idr input before the third natural number C number of operation cycles (current detection cycle TIdt) in the present operation cycle. The d-axis additional delay gain multiplier 34l multiplies the output value of the d-axis current additional delay 34k by the gain Kad. The d-axis current second adder 34n adds the output value of the d-axis current delay 34j and the output value of the d-axis additional delay gain multiplier 34l. The d-axis current first adder 34m adds the d-axis current detection value Idr detected in the present operation cycle and the output value of the d-axis current second adder 34n, and outputs the addition value SIdr of the d-axis current. The d-axis current gain multiplier 34o multiplies the addition value SIdr of the d-axis current by the gain 1 / (2+Kad), and outputs the d-axis current detection value Idr* after the current addition process.

[0161] In the q-axis current delay 34p and the d-axis current additional delay 34q and the q-axis current first adder 34s, the q-axis current detection value Iqr detected in the present operation cycle (current detection cycle TIdt) is input. The q-axis current delay 34p delays the input q-axis current detection value Iqr by the addition cycle Tadd and outputs. The q-axis current delay 34p has a second natural number B number of delays, and outputs the q-axis current detection value Iqr input before the second natural number B number of operation cycles (current detection cycle TIdt) in the present operation cycle. The q-axis current delay 34q delays the input q-axis current detection value Iqr by the additional addition cycle Taddad and outputs. The q-axis current additional delay 34q has a third natural number C number of delays, and outputs the q-axis current detection value Iqr input before the third natural number C number of operation cycles (current detection cycle TIdt) in the present operation cycle. The q-axis additional delay gain multiplier 34r multiplies the output value of the q-axis current additional delay 34q by the gain Kad. The q-axis current second adder 34t adds the output value of the q-axis current delay 34p and the output value of the q-axis additional delay gain multiplier 34r. The q-axis current first adder 34s adds the q-axis current detection value Iqr detected in the present operation cycle and the output value of the q-axis current second adder 34t, and outputs the addition value of the q-axis current SIqr. The q-axis current gain multiplier 34u multiplies the addition value of the q-axis current SIqr by the gain 1 / (2+Kad), and outputs the q-axis current detection value Iqr* after the current addition process.

[0162] <Noise reduction effect>

[0163] In the case where the addition cycle Tadd and the half cycle Tr / 2 of the resonance cycle do not coincide Figure 16 , it is described that the gain Kad is set to 0, and the current detection value detected before the addition cycle Tadd is added, but the current detection value detected before the additional addition cycle Taddad is not added. Figure 18 The transfer characteristics from the noise component included in the q-axis current detection value Iqr to the noise component included in the q-axis current obtained by converting the actual three-phase winding current to the coordinate system of the d-axis and the q-axis are shown. The horizontal axis indicates the frequency of the noise component, and the vertical axis indicates the gain. Figure 18The transfer characteristics in the case where the current addition processing is performed with Kad = 0, the transfer characteristics in the case where the current addition processing is not performed, and the transfer characteristics in the case where the response speed of the current feedback control is reduced without performing the current addition processing (the cutoff frequency is set to 1 / 10) are shown. In addition, the cutoff frequency of the current feedback control is 600 Hz in the case where the response speed is not reduced, and the cutoff frequency of the current feedback control is 60 Hz in the case where the response speed is reduced. Further, since the current detection period Tldt is 100 μs, the Nyquist frequency is 1 / 2 of the sampling frequency, that is, 5000 Hz.

[0164] In the case where the current addition processing is performed with Kad = 0, the gain is lower than that in the case where the current addition processing is not performed at a frequency of 1200 Hz or more, and the noise reduction performance is improved. In the case where the current addition processing is performed with Kad = 0, the gain starts to increase at a frequency of 2500 Hz or more, and the gain becomes 0.03 at a frequency of 3000 Hz, whereas the gain is almost 0 in the case where the response speed of the current feedback control is reduced. Therefore, the noise reduction performance in the case where the current addition processing is performed with Kad = 0 is worse than that in the case where the response speed of the current feedback control is reduced.

[0165] A case where the gain Kad is set to be greater than 0, and in addition to the addition of the current detection value detected before the addition period Tadd, the addition of the current detection value detected before the additional addition period Taddad is performed is described. Figure 19 The transfer characteristics in the case where the current addition processing is performed with Kad = 0.25, the transfer characteristics in the case where the current addition processing is not performed, and the transfer characteristics in the case where the response speed of the current feedback control is reduced without performing the current addition processing (the cutoff frequency is set to 1 / 10) are shown.

[0166] In the case where the current addition processing is performed with Kad = 0.25, the increase in the gain can be suppressed at a frequency of 2500 Hz or more, and the gain is almost 0 at a frequency of 3000 Hz, and the same noise reduction performance as that in the case where the response speed of the current feedback control is reduced can be achieved. Therefore, even in the case where the addition period Tadd and the half period Tr / 2 of the resonance period do not coincide with each other, by adding the current detection value detected before the additional addition period Taddad which is close to the half period Tr / 2 of the resonance period, the effect of reducing the component in the resonance period Tr of the current detection value can be improved.

[0167] If the gain Kad is made greater than 0.25, the noise reduction performance in the high frequency region is improved compared to the transfer characteristics in Figure 19 If the gain Kad is made smaller than 0.25, the noise reduction performance in the high frequency region is reduced compared to the transfer characteristics in Figure 19compared to the transfer characteristics in the above-described embodiment 1, the noise reduction performance in the low frequency domain is improved. Thus, the gain Kad can be adjusted according to the required noise reduction performance. Here, the cut-off frequency of the current feedback control is set to 600 Hz, but can be set to be lower than half of the resonance frequency. For example, when the resonance frequency is 3000 Hz (Tr = 330 μs), the cut-off frequency can be set to be lower than 1500 Hz, and when the resonance frequency is 2500 Hz (Tr = 400 μs), the cut-off frequency can be set to be lower than 1250 Hz.

[0168] 4. Embodiment 4

[0169] The control device 10 according to Embodiment 4 will be described. The same structural parts as those of the above-described Embodiment 1 will be omitted from the description. The basic structure of the alternating-current rotary electric machine 1 and the control device 10 according to this embodiment is the same as that of Embodiment 1, but the alternating-current rotary electric machine 1 and the control device 10 constitute an electric power steering device 100, which is different from Embodiment 1.

[0170] In the above-described Embodiments 1 to 3, the use of the alternating-current rotary electric machine 1 and the control device 10 is not particularly determined. In this embodiment, the alternating-current rotary electric machine 1 and the control device 10 constitute an electric power steering device 100. The electric power steering device 100 includes the control device 10 of the alternating-current rotary electric machine, the alternating-current rotary electric machine 1, and a drive force transmission mechanism 101 that transmits the drive force of the alternating-current rotary electric machine 1 to a steering device 102 of a vehicle.

[0171] Figure 20 A schematic structural diagram of the electric power steering device 100 according to this embodiment is shown in FIG. 10. The rotational shaft of the rotor of the alternating-current rotary electric machine 1 is connected to the steering device 102 of a wheel 103 through the drive force transmission mechanism 101. For example, the electric power steering device 100 includes a steering wheel 104 that is rotated to the left and right by a driver, a shaft 105 that is connected to the steering wheel 104 and transmits the steering torque of the steering wheel 104 to the steering device 102 of the wheel 103, a torque sensor 106 that is mounted on the shaft 105 and detects the steering torque Ts of the steering wheel 104, and the drive force transmission mechanism 101 such as a worm gear mechanism that connects the rotational shaft of the alternating-current rotary electric machine 1 to the shaft 105. The output signal of the torque sensor 106 is input to the control device 10 (input circuit 92).

[0172] The current command value calculation section 35 calculates the torque command value Tref based on the steering torque Ts detected from the output signal of the torque sensor 106. Then, the current command value calculation section 35 calculates the current command values Ido and Iqo of the d-axis and q-axis based on the torque command value Tref, the power supply voltage Vdc, and the rotational angular velocity ω, and so on, in the same manner as in Embodiment 1.

[0173] The mechanical resonance of the alternating-current rotary electric machine 1 occurs in the frame of the alternating-current rotary electric machine 1, or in a power pack in which the alternating-current rotary electric machine 1, the inverter 4, and the controller 6 are integrated, or in the electric power steering apparatus 100 that includes the frame or the power pack and the gear mechanism.

[0174] For example, in the electric power steering apparatus 100, the mechanical resonance period Tr of the alternating-current rotary electric machine 1 is in the range of 200 μs to 500 μs (the resonance frequency is in the range of 2 kHz to 5 kHz). Preferably, the resonance period Tr can be in the range of 300 μs to 400 μs (the resonance frequency is in the range of 2.5 kHz to 3.3 kHz). In this case, the cut-off frequency of the current feedback control can be set in the range of 100 Hz to 1250 Hz, and preferably, in the range of 200 Hz to 800 Hz. In this case, when the carrier period Tc is set to be 60 μs or less, the performance of the current feedback control can be maintained while the component of the resonance period Tr is well reduced from the current detection value by the current addition processing, and the noise generated by the alternating-current rotary electric machine 1 is reduced, so that the electric power steering apparatus 100 with low noise can be obtained.

[0175] [Other Embodiments]

[0176] Finally, other embodiments of the present application are described. In addition, the structures of the embodiments described below are not limited to be applied individually, but can be applied in combination with the structures of other embodiments as long as no contradiction occurs.

[0177] (1) In Embodiment 4 described above, the case where the alternating-current rotary electric machine 1 and the control device 10 are used for the electric power steering apparatus 100 is described as an example. However, the alternating-current rotary electric machine 1 and the control device 10 can be used for various uses, for example, for a power device for a wheel.

[0178] (2) In Embodiment 2 described above, the case where the current addition processing section 34 switches and outputs the current detection value after the current addition processing calculated by the same current addition processing as Embodiment 1 and the current detection value detected this time is described as an example. However, the current addition processing section 34 can be configured to switch and output the current detection value after the current addition processing calculated by the same current addition processing as Embodiment 3 and the current detection value detected this time.

[0179] (3) In the above embodiments, the modulation unit 363 was described using the modulation described in formula (1) as an example. However, the modulation unit 363 can perform various modulations other than formula (1), which can cause current detection errors to be generated by ringing. For example, the modulation unit 363 determines the minimum value Vmin and the maximum value Vmax of the coordinate-transformed three-phase voltage command values ​​Vuoc, Vvoc, and Vwoc, multiplies the sum of the minimum value Vmin and the maximum value Vmax by 0.5, calculates the bias voltage Voff, and subtracts the bias voltage Voff from the coordinate-transformed three-phase voltage command values ​​Vuoc, Vvoc, and Vwoc to calculate the three-phase voltage command values ​​Vuo, Vvo, and Vwo.

[0180] Vmin = MIN(Vuoc, Vvoc, Vwoc)

[0181] Vmax = MAX(Vuoc, Vvoc, Vwoc)

[0182] Voff=0.5×(Vmin+Vmax)…(7)

[0183] Vuo = Vuoc - Voff

[0184] Vvo = Vvoc - Voff

[0185] Vwo = Vwoc - Voff

[0186] The amplitude reduction modulation action of equation (7) is in Figure 21 The diagram above shows the coordinate-transformed three-phase voltage command values ​​Vuoc, Vvoc, and Vwoc. The coordinate-transformed three-phase voltage command values ​​Vuoc, Vvoc, and Vwoc exceed the range from -Vdc / 2 to +Vdc / 2, resulting in voltage saturation. On the other hand, the amplitude-reduced modulation of the three-phase voltage command values ​​Vuo, Vvo, and Vwo converges within the range of -Vdc / 2 to +Vdc / 2, preventing voltage saturation. Even with this modulation, when the voltage command value approaches +Vdc / 2 and the conduction period of the negative-side switching element becomes shorter, current detection errors occur due to ringing. Therefore, by performing current addition processing, the component of the resonant period Tr can be reduced from the current detection value, thereby reducing the noise of the AC rotating motor 1.

[0187] (4) In each of the above-described embodiments, the case where the current addition processing section 34 performs current addition processing on the current detection values Idr, Iqr of the d-axis and q-axis was described as an example. However, the current addition processing section 34 can also be configured to perform current addition processing on the three-phase current detection values Iur, Ivr, Iwr. For example, in a structure corresponding to Embodiment 1, the current addition processing section 34 performs current addition processing by adding the three-phase current detection values Iur, Ivr, Iwr detected in the present current detection and the three-phase current detection values Iur, Ivr, Iwr detected in the current detection before the addition period Tadd, respectively, and calculates the current addition-processed three-phase current detection values Iur*, Ivr*, Iwr*. In a structure corresponding to Embodiment 3, the current addition processing section 34 performs current addition processing by adding the three-phase current detection values Iur, Ivr, Iwr detected in the present current detection, the three-phase current detection values Iur, Ivr, Iwr detected in the current detection before the addition period Tadd, and the three-phase current detection values Iur, Ivr, Iwr detected before the additional addition period Taddad, respectively, and calculates the current addition-processed three-phase current detection values Iur*, Ivr*, Iwr*. Then, the current coordinate conversion section 33 performs three-phase / two-phase conversion and rotational coordinate conversion on the current addition-processed three-phase current detection values Iur*, Ivr*, Iwr* based on the magnetic pole position θ, converts them into the current addition-processed d-axis and q-axis current detection values Idr* and Iqr*, and outputs them to the voltage command value calculation section 36.

[0188] While various exemplary embodiments and examples are described in the present application, various features, modes and functions described in one or more embodiments are not limited to the application of the specific embodiments, and can be applied to the embodiments individually or in various combinations. Therefore, it can be considered that numerous modifications not exemplified are also included in the technical scope disclosed in the present application. For example, cases where at least one constituent element is modified, added or omitted, and cases where at least one constituent element is extracted and combined with the constituent elements of other embodiments are included.

[0189] Explanation of Reference Numerals

[0190] 1 alternating-current rotary electric machine, 3 direct-current power supply, 4 inverter, 5 current detection circuit, 6 controller, 10 control device of alternating-current rotary electric machine, 100 electric power steering device, 101 drive force transmission mechanism, 102 steering device, A first natural number, B second natural number, C third natural number, CA carrier wave, TIdt current detection period, T_th threshold value, Tadd addition period, Taddad additional addition period, Tc carrier wave period, Tr resonance period, Vdc power supply voltage.

Claims

1. A control device for an AC rotating motor, wherein the control device controls an AC rotating motor having three-phase windings, characterized in that, comprises: an inverter provided with three sets of series circuits corresponding to each phase of three phases, the series circuits connecting in series a positive side switching element connected to a positive side of a direct current power supply and a negative side switching element connected to a negative side of the direct current power supply, and connecting a connection point of the series connection to the winding of the corresponding phase; a current detection circuit having a resistance connected in series with the negative side switching element of at least two phases; and a controller that calculates three-phase voltage command values, performs on-off control of the switching elements by comparing each of the three-phase voltage command values with a carrier wave oscillating at a carrier period, detects a current flowing through the three-phase winding based on an output signal of the current detection circuit at each current detection timing, the current detection timing being set at a current detection period that is a first natural number times the carrier period, and the negative side switching element being turned on at the current detection timing, adds a current detection value detected at the current detection timing to a current detection value detected at a first past current detection timing to perform current addition processing, the first past current detection timing being earlier than the current detection timing by an addition period, the addition period being a second natural number times the current detection period, and calculates the three-phase voltage command values based on the current detection value after the current addition processing, the second natural number being set as a natural number at which the addition period is closest to a half period of a mechanical resonance period of the alternating current rotating electric machine.

2. The control device of the alternating current rotating electric machine according to claim 1, wherein for each phase, the controller turns on the negative side switching element when the carrier wave is higher than the voltage command value, and turns off the negative side switching element when the carrier wave is lower than the voltage command value, and a current is detected at each current detection timing at which the carrier wave is at an apex of a wave crest.

3. The control device of the alternating current rotating electric machine according to claim 2, wherein at each current detection timing, the controller converts a current detection value of the three-phase winding into d-axis and q-axis current detection values in a coordinate system of d-axis and q-axis that rotates in synchronization with a magnetic pole position of a rotor of the alternating current rotating electric machine, performs current addition processing by adding the d-axis and q-axis current detection values calculated at the current detection timing to the d-axis and q-axis current detection values calculated at the first past current detection timing, respectively, and calculates the three-phase voltage command values based on the d-axis and q-axis current detection values after the current addition processing.

4. The control device of the alternating current rotating electric machine according to claim 1, wherein ​ ​ ​ The controller converts the current detection values of the three-phase winding into d-axis and q-axis current detection values in a coordinate system that rotates in synchronization with the magnetic pole position of the rotor of the alternating-current rotary electric machine at each of the current detection timings, performs current addition processing by adding the d-axis and q-axis current detection values calculated at the current detection timing to the d-axis and q-axis current detection values calculated at the first past current detection timing, respectively, calculates current addition-processed d-axis and q-axis current detection values, and calculates the three-phase voltage command values based on the current addition-processed d-axis and q-axis current detection values.

5. The control device of an alternating-current rotary electric machine according to any one of claims 1 to 4, characterized in that, the controller performs current addition processing by adding the current detection value detected at the current detection timing, the current detection value detected at the first past current detection timing, and a current detection value detected at a second past current detection timing to which an additional addition period is added to the current detection timing, the additional addition period being a period that is a third natural number times the current detection period, the third natural number is set to a natural number at which the additional addition period secondly approximates to a half period of a mechanical resonance period of the alternating-current rotary electric machine.

6. The control device of an alternating-current rotary electric machine according to any one of claims 1 to 4, characterized in that, the controller calculates the three-phase voltage command values based on the current addition-processed current detection values when the conduction period within the carrier period of the negative-side switching element of any phase is lower than a threshold value set in advance, and calculates the three-phase voltage command values based on the current detection value detected at the current detection timing when the conduction period within the carrier period of the negative-side switching element of any phase is not lower than the threshold value.

7. The control device of an alternating-current rotary electric machine according to any one of claims 1 to 4, characterized in that, the controller shifts the three-phase voltage command values equally so that the average value of the three-phase voltage command values is smaller than the center of oscillation value of the carrier, and performs conduction cutoff control of the switching elements by comparing the shifted three-phase voltage command values with the carrier, respectively.

8. An electric power steering apparatus characterized by comprising: including: the control device of an alternating-current rotary electric machine according to any one of claims 1 to 7; the alternating-current rotary electric machine; and a drive force transmission mechanism that transmits a drive force of the alternating-current rotary electric machine to a steering device of a vehicle, the carrier period is set to be 60 μs or less, the mechanical resonance period of the alternating-current rotary electric machine is in the range of 200 μs to 500 μs.

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