Control system of bidirectional DC-DC converter based on CLLC structure

Abstract

The present invention discloses a control system of a bidirectional DC-DC converter based on a CLLC structure, which can be used for bidirectional energy conversion between 48V and 12V batteries in electric vehicles, with a maximum conversion power of up to one kilowatt. The converter consists of a power stage and a control stage, wherein the power stage includes batteries of different voltage levels, filter capacitors, power MOSFET tubes, resonant capacitors and inductors, and a transformer with an excitation inductor; the control stage includes a signal gating module, a control mode selection module, a frequency modulation module, a duty cycle modulation module, and a drive circuit module, and the power stage and the control stage are connected by isolation detection and isolation drive. The present invention proposes a control scheme of the converter and a specific design and implementation of an important control loop therein, so that the converter has the characteristics of automatic adjustment and control of the drive signal, high energy transmission efficiency, and high reliability compared with open-loop drive and traditional control schemes.

Description

Technical Field

[0001] The invention belongs to the technical field of switch power supply control, and in particular relates to a control system of a bidirectional DC-DC converter based on a CLLC structure. Background Art

[0002] To realize the clean energy in the transportation field, electric vehicles account for an increasing proportion. In addition to powering the basic function of driving the vehicle, the batteries of modern electric vehicles also need to power various applications such as vehicle control systems, lighting systems, and on-board charging systems. These applications require different DC voltage levels. Currently, electric vehicles are generally equipped with two groups of batteries, 48V and 12V, which are used to power high-voltage series applications (mainly driving the vehicle) and low-voltage series applications (mainly vehicle control systems). When charging and running electric vehicles, the batteries of these two voltage levels need to perform bidirectional high-power energy conversion according to the power load conditions on the high-voltage side and the low-voltage side.

[0003] In order to achieve kilowatt-level DC-DC conversion, it is necessary to adopt a power-level topology that can achieve voltage and current misalignment of the switch tube through self-resonance and thus achieve soft switching, such as Figure 1 The full-bridge CLLC topology shown has the characteristics of high stability and high conversion efficiency. Its disadvantages of requiring more power devices and high control complexity can also be tolerated by electric vehicle applications. At present, there are many studies on CLLC structure and its performance, mainly on the soft switching operating frequency range, but there are few complete studies on CLLC automatic control, and there are no related products yet.

[0004] Existing studies have shown that the natural frequency of CLLC is determined by the parameter value of the resonant device. When the actual operating frequency is slightly lower than the natural frequency, the time of each working cycle is slightly longer than the time required for resonance, which can leave a zero current time for the transformer to achieve soft switching of the power tube. Such an operating frequency is suitable for the CLLC power stage structure under the above application environment. The main goal of the control stage circuit is to automatically adjust the actual operating frequency to this appropriate range.

[0005] By performing an idealized equivalent circuit analysis on the power stage, it can be found that the load-side voltage is a function of the operating frequency. The traditional control scheme detects the load-side voltage and adaptively adjusts the frequency through a feedback loop. In the relevant technical field, the basic control method is to sample the load-side voltage and use PFM control to control the operating frequency. This basic control method can achieve adaptive control of the power-stage frequency, but it has the disadvantages that the duty cycle cannot be adaptively controlled and the power-stage conversion efficiency cannot be adaptively adjusted to the highest level.

[0006] To overcome this disadvantage, the relevant technology has developed to use PFM-PWM hybrid control. The latest control method currently published is a Chinese patent with publication number CN113422516A, which provides a method and system for PFM-PWM hybrid control of CLLC resonant converter, such as Figure 2 As shown, the system method collects the voltage and current information on the load side, and through calculation processing, jointly controls the operating frequency and duty cycle, first performs PFM modulation and then PWM modulation, which improves the system efficiency and is particularly suitable for applications in scenarios with a large range of load conditions. However, this patented technology has the problem that the voltage error on the power supply side cannot be adjusted directly through the feedback loop, and the problem that PWM is related to PFW signals and the two cannot be adjusted separately. Summary of the invention

[0007] In view of the above, the present invention provides a control system of a bidirectional DC-DC converter based on a CLLC structure, which adopts a scheme in which the primary and secondary sides jointly modulate the frequency and duty cycle, thereby realizing a more efficient and reliable control system.

[0008] A control system of a bidirectional DC-DC converter based on a CLLC structure, comprising:

[0009] An isolation detection input module is used to isolate and convert the collected high-side voltage VBH and low-side voltage VBL of the power stage (i.e., the bidirectional DC-DC converter power circuit) into analog voltages VCH and VCL based on the chip reference ground respectively;

[0010] The signal gating module selects VCH and VCL as the power supply side voltage VSUPPLY and the load side voltage VLOAD according to the externally given power level energy transfer direction;

[0011] The control mode selection module autonomously determines the current working mode of the power stage according to the changes of VSUPPLY and VLOAD, and outputs a working mode indication signal CM;

[0012] The frequency modulation module generates a voltage VFM for frequency modulation according to VSUPPLY and VLOAD, and modulates VFM into a pulse signal VPFW with a non-fixed frequency, and then performs edge integration processing on VPFW to obtain a trapezoidal wave CW;

[0013] The duty cycle modulation module uses the trapezoidal wave CW as the carrier wave, and performs error amplification processing on VLOAD to obtain two modulation waves MW1 and MW2, and then compares the carrier wave with MW1 and MW2 respectively, and then obtains the modulation signals VPWM1 and VPWM2 after PWM processing;

[0014] A driving module, used to generate 8 driving signals VC1 to VC8 after driving delay and protection of VPWM1 and VPWM2;

[0015] The isolated drive output module is used to convert the drive signals VC1-VC8 based on the chip reference ground into drive signals V1-V8 based on the source potential of each power switch device of the power stage, which correspond to the gate signals of each power switch device.

[0016] Further, if the power level energy transfer direction is from high voltage side to low voltage side, the signal selection module selects VCH as the power supply side voltage VSUPPLY and selects VCL as the load side voltage VLOAD; if the power level energy transfer direction is from low voltage side to high voltage side, the signal selection module selects VCL as the power supply side voltage VSUPPLY and selects VCH as the load side voltage VLOAD.

[0017] Furthermore, the judgment criteria of the control mode selection module are as follows:

[0018] When VLOAD<V2 and one of the following two conditions is met, the power stage is determined to be in the power-on mode, otherwise the power stage is determined to be in the normal working mode in all other cases;

[0019] Condition 1: VSUPPLY>V2;

[0020] Condition 2: V1<VSUPPLY≤V2 and VSUPPLY rises from below V1 to within this range; where: 0<V1<V2, V1 is close to 0, and V2 is close to and slightly less than the chip's operating voltage VDDC.

[0021] Furthermore, the expression of the voltage VFM generated by the frequency modulation module is as follows:

[0022] VFM=a*VSUPPLY+b*VLOAD+c*Vreff

[0023] Wherein: Vreff is the internal reference voltage of the chip, a, b, c are weight coefficients, 0<a<1, 0<c<1, when CM is low level, the power stage is in normal working mode, b=0; when CM is high level, the power stage is in power-on mode, -1<b<0.

[0024] Furthermore, the frequency of the pulse signal VPFW is positively correlated with VFM.

[0025] Furthermore, the modulation wave MW1 is positively correlated with VLOAD, and VDDC / 2<MW1<VDDC; the modulation wave MW2 is negatively correlated with VLOAD, and 0<MW2<VDDC / 2, and VDDC is the chip operating voltage.

[0026] Furthermore, the power stage, i.e., the bidirectional DC-DC converter power circuit, includes a high-voltage side full-bridge conversion circuit and a low-voltage side full-bridge conversion circuit. The DC side of the high-voltage side full-bridge conversion circuit is connected to a high-voltage battery, and the DC side of the low-voltage side full-bridge conversion circuit is connected to a low-voltage battery. The high-voltage side full-bridge conversion circuit and the low-voltage side full-bridge conversion circuit are coupled through a CLLC resonant device and an isolation transformer to achieve energy transmission; the high-voltage side full-bridge conversion circuit includes switch tubes Q1 to Q4, wherein Q1 and Q3 serve as upper tubes, and Q4 and Q2 correspond to lower tubes of Q1 and Q3; the low-voltage side full-bridge conversion circuit includes switch tubes Q5 to Q8, Among them, Q6 and Q8 serve as upper tubes, and Q7 and Q5 correspond to the lower tubes of Q6 and Q8; the lead wires of the bridge arm composed of Q1 and Q4 and the lead wires of the bridge arm composed of Q6 and Q7 are connected to a group of same-named ends of the isolation transformer after passing through the resonant device respectively, and the lead wires of the bridge arm composed of Q2 and Q3 and the lead wires of the bridge arm composed of Q5 and Q8 are connected to another group of same-named ends of the isolation transformer after passing through the resonant device respectively; the modulation signal VPWM1 is used to control the switching state of Q1, Q2, Q5, and Q6, and the modulation signal VPWM2 is used to control the switching state of Q3, Q4, Q7, and Q8.

[0027] The present invention proposes a novel PFM-PWM control method based on the traditional technology. By sampling the voltages on the power supply side and the load side, the method is divided into two working processes (power-on process and normal process). In the normal process, the working frequency is controlled by the power supply side voltage, and the duty cycle is controlled by the load side voltage. In the power-on process, the working frequency is controlled by the power supply voltage and the load side voltage, and the duty cycle is controlled by the load side voltage.

[0028] The control of the operating frequency in the present invention is achieved by adjusting the PWM carrier frequency, and the control of the duty cycle is adjusted by adjusting the PWM modulation wave amplitude, and the control method of first PFM and then PWM is also adopted. The innovation of the present invention lies in:

[0029] 1. Collecting the power supply side voltage for frequency adjustment can establish a frequency adjustment response to the power supply voltage change more quickly when the system is working normally. Frequency adjustment through the power supply side voltage can also make the operating frequency more stable, avoiding the frequency jitter caused by the modulation caused by the ripple voltage on the load side in normal working engineering.

[0030] 2. Using only the power supply side voltage for frequency regulation will cause the load side voltage to overcharge during the power-on process of the system. Based on this, the present invention proposes a power-on process that is different from the normal process. During the power-on process, the power supply side and load side voltages jointly modulate the operating frequency to solve the overcharging problem.

[0031] 3. Using a trapezoidal wave as the carrier and directly modulating the dead time (indirectly modulating the duty cycle) can minimize the dead time and improve the system conversion efficiency while ensuring safe commutation and soft switching of the power stage switch tube. BRIEF DESCRIPTION OF THE DRAWINGS

[0032] Figure 1 Schematic diagram of the topology of a bidirectional DC-DC converter with a full-bridge CLLC structure.

[0033] Figure 2 This is a structural block diagram of an existing control system based on a PFM-PWM hybrid method.

[0034] Figure 3 Schematic diagram of the working waveform of important signals at the power level.

[0035] Figure 4 It is a block diagram of the functional modules inside the control system of the present invention.

[0036] Figure 5 It is a schematic diagram of a specific circuit implementation structure of the control system of the present invention.

[0037] Figure 6 Schematic diagram of simulation results of the system according to an embodiment of the present invention.

[0038] Figure 7 Schematic diagram of waveforms of important signals in the system of an embodiment of the present invention. DETAILED DESCRIPTION

[0039] In order to describe the present invention more specifically, the technical solution of the present invention is described in detail below in conjunction with the accompanying drawings and specific implementation methods.

[0040] The bidirectional DC-DC converter targeted by the present invention is as follows Figure 1 As shown, its power level is composed of high voltage battery B H 、Low voltage battery B L and power supply and load (when high voltage side transfers energy to low voltage side, B H As the power supply, B L The power supply and the load are essentially DC batteries. Depending on the direction of energy transfer, two DC batteries with different voltage levels can be used as the power supply and the load respectively. Due to different working environments and the need for electrical isolation between different applications, the two DC batteries generally have different reference ground potentials. The high-voltage side and the low-voltage side in the CLLC structure are coupled together through the magnetic field generated by the intermediate transformer, thereby realizing energy transmission under electrical isolation conditions.

[0041] The ideal CLLC circuit has the following characteristics: Q1~Q8 are all ideal switching tubes, with on-resistance of 0, off-resistance of infinity, parasitic capacitance of 0, and opening and closing can be completed instantly; the resonant devices L1, L2, C1, and C2 are ideal capacitors and inductors, with parasitic resistance of 0; the excitation inductance Lm is significantly larger than L1, and the parasitic resistance is 0; the two coils of the transformer are fully coupled, and their respective inductance values ​​are large enough that magnetic saturation will not occur, and the transformer turns ratio n(LH):n(LL) is equal to the ratio of the rated operating voltages of the high-voltage side to the low-voltage side (48V:12V=4:1 in this embodiment). The basic working principle of the ideal CLLC circuit is to divide Q1~Q8 into two groups (Q1, Q2, Q5, Q6 as one group, and the rest as another group), and apply the same frequency, duty cycle of 50%, and phase complementary ideal driving pulses to these two groups of switch tubes. In this case, after Vin passes through Q1~Q4, it realizes ideal inversion and becomes a square wave with the same switching frequency. The square wave enters the resonant cavity (L1, C1 and L2, C2), and finally outputs the DC voltage to the filter capacitor and the load through the ideal rectifier bridge. The impedance of the resonant cavity changes with frequency. When a sine wave of a certain frequency makes the impedance of the resonant cavity 0 (the frequency at this time is called the natural resonant frequency fr), the partial pressure on the load is the largest. The above square wave can be regarded as the superposition of sine waves of different frequencies through Fourier decomposition, in which the frequency of the sine wave with the largest amplitude component is consistent with the square wave frequency (this frequency is called the working frequency fw), and this working frequency component plays a dominant role in the load voltage amplitude. When fw=fr, the entire circuit is in a resonant state, there is no voltage loss on the resonant cavity, and the power supply voltage is proportionally changed through the transformer and then acts on the load. This is the ideal working state of CLLC, which can be achieved by properly designing the resonant device value and the operating frequency.

[0042] The transformer turns ratio can be determined by the ratio of the voltage levels of the DC batteries on both sides. After that, the transformer can be regarded as an ideal transformer and the load-side devices can be equivalent to the power supply side. In a simple design, the corresponding resonant device values ​​after the equivalent are equal. The main limitation in actual circuit applications is that the switch tube has parasitic resistance and parasitic capacitance, switching loss and conduction loss, and a certain dead time is required in each switching cycle to realize the commutation between the bridge arms; the introduction of the resonant cavity can stagger the phase of the voltage and current on the switch tube through the capacitor and inductor, realize the soft switching of the switch tube, and reduce the switching loss, which is the largest loss. This is also the most important role of the CLLC structure. In practical applications, in order to pursue the commutation safety and high conversion efficiency of the entire system, it is necessary to make fw slightly smaller than fr to ensure that there is enough dead time in each cycle (the current on the transformer is 0) for commutation and soft switching. This is also the operating frequency point of the power stage module designed by the control system proposed in the present invention.

[0043] At this operating frequency point, the important open-loop operating waveforms of the power stage module in practical applications (considering only the non-ideal factors of the switch tube) are as follows: Figure 3 As shown in the figure, when complementary pulse waves are added to the gates of the two groups of switch tubes, the duty cycle is slightly less than 50% (leaving dead time S3 and S7), and the frequency fw of the pulse wave is slightly less than fr (leaving common resonance area S2 and S6), the entire switching cycle of the power stage module can be divided into 8 stages S1 to S8. When the first group of switch tubes (Q1, Q2, Q5, Q6) starts to conduct, the power stage module enters the S8 reverse current stage, at which time Lm and L1 are both negative (the actual current direction is the same as the current direction). Figure 1 The direction of the current on the power supply side is B. H (-)→GNDH→Q2→C1→Lm→L1→Q1→B H (+), the current direction of the battery on the power supply side is opposite to the voltage direction and absorbs energy in this stage. Since I(L1)-I(Lm)>0, there is a downward current on LH. At this time, energy is transferred from the primary side to the secondary side. In this stage, the energy stored in the resonant device and the excitation inductor is released. Part of it is reversely charged to the power supply, and the other part is used to supply power to the load.

[0044] Under the action of the connected power supply, the loop current in the S8 stage gradually decreases to 0 and then changes direction. After the current on L1 changes direction, the power stage module enters the S1 power transmission stage, and the current direction on the power supply side is B H (+)→Q1→L1→Lm→C1→Q2→GNDH→B H (-), since Lm is several times of L1, the change of I(Lm) is obviously slower than I(L1) and tends to be linear. There is still I(L1)-I(Lm)>0. There is a downward current on LH. At this time, energy is transferred from the primary side to the secondary side. From this stage, the power supply is mainly supplied to the load. The resonant device and the excitation inductor also have corresponding energy storage and release. This stage can be divided into two sub-stages, S1a and S1b, according to the change of the direction of I(Lm). The excitation inductor fully realizes the upper half cycle. After demagnetization, it will enter S1b and start forward excitation; when the excitation inductor starts forward excitation, Lm changes from releasing energy to absorbing energy, and the current on L1 starts to decrease from the maximum value until I(L1)=I(Lm). After the currents of the two are equal, Lm and L1 participate in the resonance together, and the power stage module enters the S2 common resonance stage. In this stage, the current on the power supply side is the same as the previous stage. In this stage, the current on LH is 0, and the current on LL is also 0, creating conditions for the soft shutdown of the load side switch tube.

[0045] After the first set of switch tubes are turned off, the power stage module enters the S3 dead zone stage. In this stage, no switch tube is turned on on the power supply side. The current on L1 and Lm needs to be freewheeled through the parasitic anti-parallel diode on the switch tube. The current direction on the power supply side is B.H (-)→GNDH→D4→L1→Lm→C1→D3→B H (+), this stage can be regarded as the reverse flow preparation stage; when the second group of switch tubes (Q3, Q4, Q7, Q8) start to conduct, the power stage module enters the S4 reverse flow stage, and the current direction angle on the power supply side remains unchanged in the previous stage, but because of the conduction of the switch tube, the reverse current is switched from D4 and D3 to Q4 and Q3. It should be supplemented that: ① The other half cycle is the same as the above half cycle process, except that it is switched to another group of bridge arms; ② The switch tube on the load side performs synchronous rectification with the power supply side; ③ D4 is the parasitic diode of Q4, and the rest are similar; ④ The above is the working process when energy is transmitted from the high voltage side to the low voltage side, and the working process when energy is transmitted in the reverse direction can be obtained by analogy.

[0046] Figure 4 This is a functional module structure block diagram for realizing the above control method proposed by the present invention. VBH and VBL are the detected high-voltage side and low-voltage side voltages of the power stage. The isolation detection input module isolates and converts VBH and VBL to analog voltages VCH and VCL based on the reference ground of the control system respectively. The signal selection module determines the correspondence between the high-voltage side and the low-voltage side and the load side and the power supply side according to the energy transfer direction indication signal given to the control system (generally implemented in the form of a chip) externally, and selects VCH and VCL to the power supply side voltage VSUPPLY and the load side voltage VLOAD accordingly.

[0047] Generally, during the charging process of the on-board DCDC converter, the 48V battery is directly connected to the external ACDC converter, and the 12V battery is converted from the 48V battery through the DCDC converter. At this time, the energy transmission direction is from the high-voltage side to the low-voltage side. The high-voltage side is the power side, and the low-voltage side is the load side. VCH is selected to VSUPPLY, and VCL is selected to VLAOD.

[0048] The control mode selection module autonomously determines whether the current power level is in the power-on mode or the normal working mode according to the changes of VSUPPLY and VLOAD, and gives a working mode indication signal CM.

[0049] When VLOAD<V2 and one of the following two conditions is met, the power stage is determined to be in the power-on mode, otherwise the power stage is determined to be in the normal working mode in all other cases;

[0050] Condition 1: VSUPPLY>V2;

[0051] Condition 2: V1<VSUPPLY≤V2 and VSUPPLY rises from below V1 to within this range; where: 0<V1<V2, V1 is close to 0, and V2 is close to and slightly less than the chip's operating voltage VDDC.

[0052] The frequency modulation module sets VFM=a*VSUPPLY+b*VLOAD+c*Vreff, where Vreff is the internal reference voltage, a and c are both greater than 0 and less than 1, when CM is at a low level, b=0; when CM is at a high level, -1<b<0; and then VFM is modulated into a pulse signal VPFW with a non-fixed frequency, the frequency of which is positively correlated with the voltage value of VFM, and then the pulse signal VPFW is edge-integrated to obtain a trapezoidal wave CW.

[0053] The duty cycle modulation module uses the above-mentioned trapezoidal wave CW as the carrier wave, and processes VLOAD separately (such as error amplification) to obtain the modulation waves MW1 and MW2, MW1 is positively correlated with VLOAD, MW2 is negatively correlated with VLOAD, VDDC / 2<MW1<VDDC, 0<MW2<VDDC / 2; then the carrier wave is compared with the modulation waves MW1 and MW2 respectively, and then the modulation signals VPWM1 and VPWM2 are obtained after PWM processing, which are respectively used as the driving signal prototypes of the first group of switch tubes (Q1, Q2, Q5, Q6) and the second group of switch tubes (Q3, Q4, Q7, Q8) of the power stage.

[0054] VPWM1 and VPWM2 are then passed through the driving circuit module to implement driving delay, driving protection and other functions to obtain driving signals VC1~VC8. After isolating and driving VC1~VC8, gate signals V1~V8 directly used to control the power stage switch tube can be obtained. Figure 4 It is a generalized control level structure diagram, which includes all functional modules except the power level, among which other modules except the isolated detection input and the isolated drive output constitute the overall structure of the control chip. In the overall structure of the control chip, except for the drive circuit module, it is the core circuit for realizing the control method of the present invention, which is also the innovation of the present invention.

[0055] exist Figure 5In the specific circuit implementation structure, the signal selection module controls the transmission gate array TGA1 to perform analog signal selection by the external direction indication signal DR to obtain VSUPPLY and VLOAD. VSUPPLY and VLOAD enter the frequency modulation module together, and after error amplification, they are summed with the frequency modulation reference voltage to obtain the voltage VFM directly used for frequency modulation. VFM obtains the frequency-modulated pulse wave VPFM through a frequency-voltage-frequency converter (a voltage-controlled oscillator VCO is selected in this example), and the rising and falling edges of VPFM are integrated to obtain the trapezoidal carrier CW that can be directly used for PWM comparison; VLOAD is respectively calculated with the two reference voltages inside the control chip to obtain the modulation wave voltages MW1 and MW2, which are respectively used to modulate the duty cycle of the two groups of switch tubes at the power stage, and the duty cycle of the two groups of switch tubes at the power stage is modulated. CW is compared with MW1 and MW2 for PWM respectively to obtain two groups of PWM waves VPWM1 and VPWM2; the control mode selection module compares VSUPPLY with two reference threshold voltages inside the control chip for hysteresis to determine the power-on situation starting from 0 on the power supply side, and compares VLOAD with another reference voltage to achieve the end control of the power-on process. The above two comparison information are logically "AND" operated to obtain a control mode selection signal CM, which is used to control a transmission gate TG1 in the frequency modulation module. When CM is high (the power-on process control mode is selected at this time), the transmission gate TG1 is turned on and VFM is modulated by VLOAD. When CM is low (the normal process control mode is selected at this time), TG1 is turned off and VFM is not modulated by VLOAD.

[0056] Next, we will explain some important working node waveforms in the embodiment system, where VSUPPLY and VLOAD are assumptions based on the situations that may occur in the actual system, and other waveforms are working waveforms that will appear under such assumptions. To vividly characterize the working characteristics of this embodiment, Figure 7 The changes in frequency and duty cycle are handled accordingly compared to the actual situation. Figure 7Five working time periods Z1 to Z5 are shown in the figure: Z1 shows the power-on process after the power stage is connected to the power supply, Z2 is the process of completing the power-on and entering the normal working state (at this time, the control chip has been controlled according to the normal process mode), Z3 is the normal process, Z4 is the working condition under the load condition, and Z5 is the working condition under the power supply voltage change. When VLOAD starts to rise from 0, CM is high level and the power-on mode is selected. When the power-on process is about to be completed, CM is low level and the normal mode is selected; in the power-on mode, VFM is affected by both VSUPPLY and VLOAD. The increase of VLOAD will cause VFM to decrease. The change of VLOAD in the normal mode will not affect VFM. At this time, VFM is only modulated by VSUPPLY. The lower VSUPPLY is, the lower VFM is. The frequency of VPFM is modulated linearly by VFM. The lower the VFM, the lower the frequency of VPFM. The rising and falling edges of VPFM are integrated respectively to convert the pulse wave into a trapezoidal wave CW. The difference between CW and VPFM is that every time the rising or falling edge is reached, the rising and falling process is linearly realized through a certain time to obtain a trapezoidal wave. MW1 and MW2 are obtained by operation and adjustment of VLOAD. MW1 is positively correlated with VLOAD, and MW2 is negatively correlated with VLOAD. MW1 is always greater than 1 / 2VDDC, and MW2 is always less than 1 / 2VDDC to ensure the existence of dead time. Generally, MW1 and MW2 are symmetrical about 1 / 2VDDC (VDDC is the power supply voltage of the control chip) so that the dead time of the two sets of power stage switches is consistent. When CW is higher than MW1, VPWM1 is at a high level, which generally means that the first group of switches in the power stage is turned on; when CW is lower than MW2, VPWM2 is at a high level, which generally means that the second group of switches in the power stage is turned on. MW1 is always higher than MW2, which ensures that VPWM1 and VPWM2 will not be high at the same time. There is a dead time DT between the two, and the two groups of switches will not be turned on at the same time.

[0057] from Figure 6It can be seen that when Vin (the power supply voltage of the power stage module) changes, Vout (the load side voltage of the power stage module) can fluctuate stably within a small range around the preset value of 12V, indicating that the basic function of stabilizing the output voltage can be achieved by adopting the control loop of the PFM and PWM hybrid control method, which reflects the feasibility of the PFM and PWM hybrid control method, system structure and core circuit of the present invention; by comparing Vout (the load side voltage under the scheme of jointly controlling the normal process and the power-on process described in the present invention) and Vout_contrast (the load side voltage under the control scheme of only the normal process), it can be found that the control method of the present invention can greatly reduce the power-on overcharge voltage of Vout close to one times the preset value, effectively alleviate the voltage pressure of the load side battery, and prevent the load side battery from breaking down and burning during the power-on process. The cost is that it takes longer time for the load side voltage to rise to the preset value than the traditional scheme, but the millisecond delay is completely acceptable in this application scenario, which reflects the effectiveness of the control scheme of the present invention. Figure 6 Only the simulation results for energy transfer from the high-pressure side to the low-pressure side are shown. Similar results can be obtained when energy is transferred in the opposite direction.

[0058] The above description of the embodiments is to facilitate the understanding and application of the present invention by those skilled in the art. It is obvious that those skilled in the art can easily make various modifications to the above embodiments and apply the general principles described herein to other embodiments without creative work. Therefore, the present invention is not limited to the above embodiments. Improvements and modifications made by those skilled in the art to the present invention based on the disclosure of the present invention should be within the protection scope of the present invention.

Claims

1. A control system for a bidirectional DC-DC converter based on a CLLC structure. It is characterized in that include: The isolation detection input module is used to isolate and convert the collected power level high-voltage side voltage VBH and low-voltage side voltage VBL into analog voltages VCH and VCL based on the chip reference ground respectively; The signal gating module selects VCH and VCL as the power supply side voltage VSUPPLY and the load side voltage VLOAD according to the externally given power level energy transfer direction; The control mode selection module autonomously determines the current working mode of the power stage according to the changes of VSUPPLY and VLOAD, and outputs a working mode indication signal CM; The frequency modulation module generates a voltage VFM for frequency modulation according to VSUPPLY and VLOAD through the following expression, and modulates VFM into a pulse signal VPFW with a non-fixed frequency, and then performs edge integration processing on VPFW to obtain a trapezoidal wave CW; VFM=a*VSUPPLY+b*VLOAD+c*Vreff Wherein: Vreff is the internal reference voltage of the chip, a, b, c are weight coefficients, 0<a<1, 0<c<1, when CM is at a low level, the power level is in the normal working mode, b=0; when CM is at a high level, the power level is in the power-on mode, -1<b<0; the control system is divided into two working processes, namely the power-on process and the normal process, by sampling the voltages on the power supply side and the load side. In the normal process, the working frequency is controlled by the power supply side voltage, and the duty cycle is controlled by the load side voltage; in the power-on process, the working frequency is controlled by the power supply voltage and the load side voltage, and the duty cycle is controlled by the load side voltage; The duty cycle modulation module uses the trapezoidal wave CW as the carrier wave, and performs error amplification processing on VLOAD to obtain two modulation waves MW1 and MW2, and then compares the carrier wave with MW1 and MW2 respectively, and then obtains the modulation signals VPWM1 and VPWM2 after PWM processing; A driving module, used to generate 8 driving signals VC1 to VC8 after driving delay and protection of VPWM1 and VPWM2; The isolated drive output module is used to convert the drive signals VC1-VC8 based on the chip reference ground into drive signals V1-V8 based on the source potential of each power switch device of the power stage, which correspond to the gate signals of each power switch device.

2. The control system according to claim 1, Features: If the power level energy transfer direction is from high voltage side to low voltage side, the signal selection module selects VCH as the power supply side voltage VSUPPLY and selects VCL as the load side voltage VLOAD; if the power level energy transfer direction is from low voltage side to high voltage side, the signal selection module selects VCL as the power supply side voltage VSUPPLY and selects VCH as the load side voltage VLOAD.

3. The control system according to claim 1, Features: The judgment criteria of the control mode selection module are as follows: When VLOAD<V th2 If one of the following two conditions is met, the power stage is determined to be in the power-on mode, otherwise the power stage is determined to be in the normal working mode; Condition 1: VSUPPLY>V th2 ; Condition 2: V th1 <VSUPPLY≤V th2 And VSUPPLY is from V th1 The following rises to this range; where: 0<V th1 <V th2 , V th1 Close to 0, V th2 Close to and slightly smaller than the chip's operating voltage VDDC.

4. The control system according to claim 1, Features: The frequency of the pulse signal VPFW is positively correlated with VFM.

5. The control system according to claim 1, Features: The modulation wave MW1 is positively correlated with VLOAD, and VDDC / 2<MW1<VDDC; the modulation wave MW2 is negatively correlated with VLOAD, and 0<MW2<VDDC / 2, where VDDC is the chip operating voltage.

6. The control system according to claim 1, Features: The power stage, i.e., the bidirectional DC-DC converter power circuit, includes a high-voltage side full-bridge conversion circuit and a low-voltage side full-bridge conversion circuit. The DC side of the high-voltage side full-bridge conversion circuit is connected to a high-voltage battery, and the DC side of the low-voltage side full-bridge conversion circuit is connected to a low-voltage battery. The high-voltage side full-bridge conversion circuit and the low-voltage side full-bridge conversion circuit are coupled through a CLLC resonant device and an isolation transformer to achieve energy transmission; the high-voltage side full-bridge conversion circuit includes switch tubes Q1 to Q4, wherein Q1 and Q3 are upper tubes, and Q4 and Q2 correspond to lower tubes of Q1 and Q3; the low-voltage side full-bridge conversion circuit includes switch tubes Q5 to Q8, wherein Q Q6 and Q8 serve as upper tubes, Q7 and Q5 correspond to the lower tubes of Q6 and Q8; the lead wires of the bridge arm composed of Q1 and Q4 and the lead wires of the bridge arm composed of Q6 and Q7 are connected to a group of same-named ends of the isolation transformer after passing through the resonant device respectively, and the lead wires of the bridge arm composed of Q2 and Q3 and the lead wires of the bridge arm composed of Q5 and Q8 are connected to another group of same-named ends of the isolation transformer after passing through the resonant device respectively; the modulation signal VPWM1 is used to control the switching state of Q1, Q2, Q5, and Q6, and the modulation signal VPWM2 is used to control the switching state of Q3, Q4, Q7, and Q8.

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