A current self-reduction laser radar optoelectronic analog front end

Abstract

The present invention discloses a current self-subtraction laser radar photoelectric analog front end, which includes a differential circuit, a feedback circuit and a transimpedance amplifier. One end of the differential circuit is connected to a photodetector for detecting laser echo signals, and the other end of the differential circuit is respectively connected to the input end of the feedback circuit and the input end of the transimpedance amplifier, and the output end of the feedback circuit is connected to the output end of the transimpedance amplifier; the differential circuit calculates the first-order derivative of the input waveform, the transimpedance amplifier is used to pre-amplify the photocurrent output by the differential circuit and output a voltage signal, and the feedback circuit realizes a shunting function when the photocurrent exceeds the detection range of the transimpedance amplifier; the advantages of the present invention are: simple circuit structure, low power consumption, and high engineering application value.

Description

Technical Field

[0001] The present invention relates to the field of integrated circuit design, and more specifically to a current self-subtraction laser radar optoelectronic analog front end. Background Art

[0002] With the continuous advancement of intelligent driving technology, the requirements for accurate and real-time vehicle detection of driving environment information have increased significantly in scenarios such as adaptive cruise control, automated parking, and active braking. LiDAR is an active detection system that uses lasers as a light source and features high sensitivity, high resolution, and strong interference resistance. The basic operating principle of a common LiDAR system can be described as the system transmitting laser light at a target and then calculating information such as the target's distance to the receiver based on the echo characteristics. Compared to most traditional measurement technologies such as millimeter-wave radar and machine vision, LiDAR technology avoids signal loss caused by dimensionality conversion, allowing the system to capture three-dimensional data natively. Furthermore, LiDAR significantly reduces the requirements for the dynamic range of target textures. Furthermore, LiDAR effectively improves the system's ability to capture discontinuously changing information. Due to its unique imaging principle, LiDAR is virtually unaffected by imaging conditions in practice, has a fast response time, and can achieve real-time imaging around the clock and in near-all-weather conditions. LiDAR enables accurate, real-time acquisition of target contour, distance, and orientation information, thereby enabling environmental awareness. These advantages have made it the preferred solution for environmental perception in current intelligent driving solutions.

[0003] To achieve long-range detection, lidars often implement some form of gain adjustment in their analog front-ends to achieve a wide dynamic range. For pulsed lidars, the analog front-end often uses multiple gain levels in the first-stage transimpedance amplifier to correspond to different target distances, thereby achieving a high dynamic range. This dynamic range expansion method places high demands on the gain control method, and traditional variable gain amplifier architectures struggle to meet the real-time performance requirements of lidars. Another common method for expanding dynamic range is to connect multiple transimpedance amplifiers in parallel. Several transimpedance amplifiers with different gains are connected in parallel at the input to pre-amplify photocurrents of varying magnitudes. This type of analog front-end meets the real-time performance requirements of lidars by operating the various transimpedance amplifiers simultaneously. However, the signal-to-noise ratio of the receive channel decreases due to the incorporation of new links, power consumption increases significantly, and the reduced circuit bandwidth introduced by this structure limits its application in frequency-modulated continuous-wave systems, reducing its engineering value. Summary of the Invention

[0004] The technical problem to be solved by the present invention is how to improve the dynamic range of the laser radar analog front-end input and reduce power consumption.

[0005] The present invention solves the above-mentioned technical problems through the following technical means: a current self-subtraction type laser radar photoelectric analog front end, including a differential circuit, a feedback circuit and a transimpedance amplifier, one end of the differential circuit is connected to a photodetector for detecting laser echo signals, the other end of the differential circuit is respectively connected to the input end of the feedback circuit and the input end of the transimpedance amplifier, and the output end of the feedback circuit is connected to the output end of the transimpedance amplifier; the differential circuit calculates the first-order derivative of the input waveform, the transimpedance amplifier is used to pre-amplify the photocurrent output by the differential circuit and output a voltage signal, and the feedback circuit realizes a shunting function when the photocurrent exceeds the detection range of the transimpedance amplifier.

[0006] The present invention does not require an external signal to control the gain change of the circuit. The feedback circuit realizes the shunting function when the photocurrent exceeds the detection range of the transimpedance amplifier. When the photocurrent is large, the feedback circuit is automatically turned on to realize the current shunting function. At this time, the transimpedance amplifier will not be saturated due to the excessive power of the echo laser. The input current value can be adjusted according to the laser echo power to expand the front-end dynamic range, thereby realizing the widening of the dynamic range of the input end. The circuit structure is simple, the power consumption is low, and the engineering application value is high.

[0007] Furthermore, the current self-subtraction type laser radar optoelectronic analog front end also includes a post-amplifier, and the input end of the post-amplifier is connected to the output end of the transimpedance amplifier.

[0008] Furthermore, the current self-subtraction type laser radar optoelectronic analog front end also includes a zero-crossing detection circuit for analyzing echo characteristics and a time resolution circuit for solving the distance of the target. The zero-crossing detection circuit is connected to the output end of the feedback circuit, and the zero-crossing detection circuit is also connected to the output end of the post-amplifier and the time resolution circuit.

[0009] Furthermore, the differential circuit includes an inductor L and a capacitor C, wherein the inductor L and the capacitor C are connected in series, and the series connection point serves as the output terminal of the differential circuit and is connected to the transimpedance amplifier and the feedback circuit. The non-series end of the inductor L is connected to ground, and the non-series end of the capacitor C, i.e., the input terminal of the differential circuit, is connected to the photodetector.

[0010] Furthermore, the feedback circuit includes a feedback resistor, a current self-subtraction branch, a current detection circuit and a first constant current bias circuit, the feedback resistor is connected to the current self-subtraction branch, the input end of the current self-subtraction branch is connected to the output end of the differential circuit, the current self-subtraction branch is connected to the current detection circuit and both are connected to the first constant current bias circuit.

[0011] Furthermore, the feedback resistor includes resistor RF1 and resistor RF2, the current self-reduction branch includes MOS transistor MF1 and MOS transistor MF2, and the current detection circuit includes resistor R dThe first constant current bias circuit includes a MOS transistor MF4 and a MOS transistor MF5. The gates of the MOS transistors MF1 and MF2 are short-circuited and connected to the output end of the differential circuit. The drain of the MOS transistor MF1 is connected to the power supply VDD, and the drain of the MOS transistor MF2 is connected to the drain of the MOS transistor MF4. The sources of the MOS transistors MF1 and MF2 are short-circuited and connected to a common node formed by the series connection of resistors RF1 and RF2. The source of the MOS transistors MF5 and MF4 are short-circuited and connected to the power supply VSS. The drain of the MOS transistor MF5, the gate of the MOS transistor MF5, and the gate of the MOS transistor MF4 are short-circuited and connected to the output node IBIAS of the on-chip bias current circuit to input a constant DC current. The gate VB of the MOS transistor MF3 is connected to the bias voltage circuit to ensure that the MOS transistor MF3 operates in the saturation region. The source of the MOS transistor MF3 is connected to the drain of the MOS transistor MF4, and the drain of the MOS transistor MF3 is connected to the resistor R d One end is connected, the drain of MOS tube MF3 is also the current flag output CRout, the resistor R d The other end is connected to the power supply VDD.

[0012] Furthermore, the transimpedance amplifier includes a first-stage amplifier circuit, a compensation capacitor, a second-stage amplifier circuit and a second constant current bias circuit. The first-stage amplifier circuit is connected to the second-stage amplifier circuit and both are connected to the second constant current bias circuit. A compensation capacitor is also connected between the first-stage amplifier circuit and the second-stage amplifier circuit.

[0013] Furthermore, the first-stage amplifier circuit includes sequentially numbered MOS transistors M1 to M8. The gate of the MOS transistor M1 is connected to the output end of the differential circuit. The source of the MOS transistor M1 is short-circuited with the source of the MOS transistor M2 and then connected to the second constant-current bias circuit. The drain of the MOS transistor M1 is connected to the source of the MOS transistor M3, the drain of the MOS transistor M2 is connected to the source of the MOS transistor M4, and the gate of the MOS transistor M3 is connected to the gate of the MOS transistor M4. The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M5, the drain of the MOS transistor M4 is connected to the drain of the MOS transistor M6, and the gate of the MOS transistor M5 is connected to the gate of the MOS transistor M6. The source of the MOS transistor M5 is connected to the drain of the MOS transistor M7, the source of the MOS transistor M6 is connected to the drain of the MOS transistor M8, the gate of the MOS transistor M7 is connected to the gate of the MOS transistor M8, and the connection node is connected to the drain of the MOS transistor M5. The source of the MOS transistor M7 is connected to the source of the MOS transistor M8 and is connected to the power supply VDD.

[0014] Furthermore, the secondary amplifier circuit includes sequentially numbered MOS transistors M9 to M12. The source of the MOS transistor M9 is short-circuited with the source of the MOS transistor M10 and then connected to the second constant current bias circuit. The gate of the MOS transistor M9 is connected to the drain of the MOS transistor M4. The drain of the MOS transistor M9 is connected to the drain of the MOS transistor M11. The drain of the MOS transistor M10 is connected to the drain of the MOS transistor M12. The gate of the MOS transistor M11 is connected to the gate of the MOS transistor M12, and the connection node is connected to the drain of the MOS transistor M10. The source of the MOS transistor M11 is connected to the source of the MOS transistor M12 and is connected to the power supply VDD. One end of the compensation capacitor Cc is connected to the drain of the MOS transistor M8, and the other end of the compensation capacitor Cc is connected to the drain of the MOS transistor M11.

[0015] Furthermore, the second constant current bias circuit includes a MOS transistor M13 and a MOS transistor M14. The drain of the MOS transistor M13 is connected to the source of the MOS transistor M1 and the source of the MOS transistor M2. The drain of the MOS transistor M14 is connected to the source of the MOS transistor M9 and the source of the MOS transistor M10. The gates of the MOS transistor M13 and the gates of the MOS transistor M14 are both connected to the power supply VB1. The sources of the MOS transistor M13 and the source of the MOS transistor M14 are connected and connected to the power supply VSS.

[0016] The advantages of the present invention are:

[0017] (1) The present invention does not require an external signal to control the gain change of the circuit. The feedback circuit realizes the shunting function when the photocurrent exceeds the detection range of the transimpedance amplifier. When the photocurrent is large, the feedback circuit automatically turns on to realize the current shunting function. At this time, the transimpedance amplifier will not be saturated due to the excessive power of the echo laser. The input current value can be adjusted according to the laser echo power to expand the front-end dynamic range, thereby realizing the widening of the input end dynamic range. The circuit structure is simple, the power consumption is low, and the engineering application value is high.

[0018] (2) In order to address the waveform feature distortion caused by nonlinear gain, the differential circuit is introduced to combine the characteristics of the first-order derivative of the Gaussian pulse with the back-end zero-crossing processing to obtain echo feature information including echo pulse width and amplitude. Compared with the full-waveform lidar processing circuit, the circuit complexity is greatly reduced, and the differential circuit can be implemented off-chip according to the application scenario, which greatly improves the versatility of the circuit.

[0019] (3) The analog front-end circuit of the present invention has a simple overall structure and low power consumption while meeting the requirements of real-time performance and a large input dynamic range. For high-beam laser radars, cascade matching can be used to achieve multi-channel expansion applications, greatly reducing design costs and complexity.

[0020] (4) The present invention has a simple structure and high loop stability, and can detect the self-reduction current. The working status of the circuit and the real-time echo optical power can be detected through the current flag. When the self-reduction branch is not working, the DC path is blocked, and the power consumption and noise are low.

[0021] (5) The transimpedance amplifier of the present invention has low input noise, high current detection sensitivity, and a small compensation capacitor, which saves circuit layout area. BRIEF DESCRIPTION OF THE DRAWINGS

[0022] Figure 1 This is a block diagram of the principle of a current-subtracting laser radar optoelectronic analog front end provided by an embodiment of the present invention;

[0023] Figure 2 A schematic diagram of a feedback circuit for a current-subtracting laser radar optoelectronic analog front end provided by an embodiment of the present invention;

[0024] Figure 3 This is a schematic diagram of a transimpedance amplifier for a current-subtracting laser radar optoelectronic analog front end provided by an embodiment of the present invention. DETAILED DESCRIPTION

[0025] To make the objectives, technical solutions, and advantages of the embodiments of the present invention more clear, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. All other embodiments obtained by ordinary technicians in this field based on the embodiments of the present invention without making any creative efforts shall fall within the scope of protection of the present invention.

[0026] like Figure 1 As shown, a current-subtracting laser radar optoelectronic analog front end includes a differential circuit 2, a feedback circuit 3, a transimpedance amplifier 4, and a post-amplifier 5. One end of the differential circuit 2 is connected to a photodetector 1 for detecting laser echo signals, and the other end of the differential circuit 2 is connected to the input of the feedback circuit 3 and the input of the transimpedance amplifier 4, respectively. The output of the feedback circuit 3 is connected to the output of the transimpedance amplifier 4, and the input of the post-amplifier 5 is connected to the output of the transimpedance amplifier 4. The differential circuit 2 calculates the first-order derivative of the input waveform, and the transimpedance amplifier 4 is used to pre-amplify the photocurrent output by the differential circuit 2 and output a voltage signal. The feedback circuit 3 implements a current shunting function when the photocurrent exceeds the detection range of the transimpedance amplifier 4.

[0027] In practical applications, the current-subtracting LiDAR optoelectronic analog front-end also includes a zero-crossing detection circuit 6 for analyzing echo characteristics and a time-resolver circuit 7 for determining target distance. The zero-crossing detection circuit 6 is connected to the output of the feedback circuit 3 and is also connected to the output of the post-amplifier 5 and the time-resolver circuit 7. The post-amplifier 5 can utilize the fixed-gain amplifier technology described in the document "Design of a Monolithic APD Front-End Readout Circuit for LiDAR Applications." This module further amplifies the output of the transimpedance amplifier 4 to improve the signal-to-noise ratio. The zero-crossing detection circuit 6 is implemented on an FPGA using a hysteresis comparator to reduce the running error introduced by the time-resolver circuit 7 and detect various echo information from the laser signal. The time-resolver circuit 7 can utilize an off-chip TDC to separately read the synchronization signal and the echo arrival time to achieve time-of-flight measurement. The system input is the photocurrent generated by the photodetector's response to the echoed laser light. The transimpedance amplifier 4 preprocesses the current signal and converts it into a voltage signal. At the same time, the strong echo signal is self-attenuated and a current flag is output to expand the link dynamic range while preserving as much original waveform information as possible. A post-amplifier 5 further amplifies the signal to improve the signal-to-noise ratio. An LC first-order differential circuit 2 transforms the output pulse waveform. A zero-crossing detection circuit 6 combines the inherent characteristics of Gaussian pulses to obtain various echo information. This assists in time resolution, specifically by subtracting the time of signal transmission from the time of echo reception to determine the time of flight.

[0028] Among them, the process of acquiring multiple echo information is as follows: since the laser radar echo information not only contains the distance information of the target, after normalizing the peak value, pulse width and other information of the waveform, the algorithm can obtain the intrinsic characteristics including the grayscale information, roughness, reflectivity and so on of the target surface. This requires that when the analog front end processes the echo information, it must not only accurately restore the rising edge of the echo signal, but also minimize the distortion of the signal through means similar to full waveform sampling. In pulse ranging, the laser signal is often modulated into a Gaussian pulse with a pulse width of about 10ns. The narrow pulse signal places high demands on the ADC sampling rate, which greatly increases the power consumption and cost of the system. The present invention utilizes the inherent characteristics of the Gaussian pulse, performs a first-order differential transformation on the signal based on its analytical expression, transforms the rising edge position and peak position of the signal to 0, detects the rising edge and peak value in the zero-crossing detection circuit through the self-subtraction branch and the output of the analog front end, and measures a variety of echo information including the original waveform amplitude and pulse width. The above process uses differential circuit 2 to add the maximum amplitude point of the original echo, that is, the waveform inflection point, that is, the first-order derivative zero point, as the judgment basis for time solution. This avoids the walking error problem caused by different echo powers when the traditional circuit uses the rising edge as the judgment criterion, thereby improving the distance solution accuracy.

[0029] The following is a detailed introduction to the principles of each circuit:

[0030] The differential circuit 2 includes an inductor L and a capacitor C, and the inductor L is connected in series with the capacitor C. The series connection point serves as the output end of the differential circuit 2 and is connected to the transimpedance amplifier 4 and the feedback circuit 3. The non-series end of the inductor L is connected to the ground, and the non-series end of the capacitor C, i.e., the input end of the differential circuit, is connected to the photodetector. The above-mentioned first-order LC circuit can be implemented off-chip or on-chip. The laser modulation mode emitted by the pulsed laser radar is Gaussian pulse modulation, and the pulse peak point corresponds to the zero point of its first-order derivative. Since a current self-reduction branch is introduced in the feedback loop of the back-end transimpedance amplifier 4 to improve the dynamic range, the echo information is greatly clipped. The introduction of the differential circuit 2 in conjunction with the zero-crossing detection can effectively restore some waveform features. Compared with the full waveform sampling analog front end, the present invention uses the inherent characteristics of the Gaussian pulse to analyze the waveform. Information including pulse width and echo amplitude can be obtained through the peripheral zero-crossing detection circuit 6, providing more dimensional information for back-end processing.

[0031] Considering the laser radar working scenario in a close-range blind spot, when the laser echo optical power exceeds its dynamic range, the transimpedance amplifier 4 enters a saturated state, causing many problems including pulse width broadening of the Gaussian pulse and transistor desaturation. At this time, the waveform is completely distorted and cannot be processed in the analog domain, which also poses a great challenge to the back-end digital processing. Therefore, it is considered to introduce a current self-reduction structure in the feedback circuit 3. The introduction of this mechanism protects the core operational amplifier of the transimpedance amplifier 4 from entering a saturated or deep saturated state to achieve a large dynamic range input. Figure 2 As shown, the feedback circuit 3 includes a feedback resistor, a current self-reduction branch, a current detection circuit and a first constant current bias circuit. The feedback resistor is connected to the current self-reduction branch. The input end of the current self-reduction branch is connected to the output end of the differential circuit 2. The current self-reduction branch is connected to the current detection circuit and both are connected to the first constant current bias circuit. Figure 2 As shown, resistors RF1 and RF2 are connected in series. The Iin node of resistor RF1 is connected to the Iin node of transimpedance amplifier 4, and the Vout node of resistor RF2 is connected to the Vout node of transimpedance amplifier 4. MOS transistors MF1 and MF2 form a current-subtracting branch. The gates of MOS transistors MF1 and MF2 are short-circuited and connected to Iin. The drain of MOS transistor MF1 is connected to VDD, and the drain of MOS transistor MF2 is connected to the drain of MOS transistor MF4. The sources of MOS transistors MF1 and MF2 are short-circuited and connected to the common node formed by the series connection of resistors RF1 and RF2.

[0032] Continue reading Figure 2MOS transistors MF5 and MF4 form a constant current bias circuit. The sources of these two MOS transistors are short-circuited and connected to GND, also known as VSS. The drain of MOS transistor MF5, the gate of MOS transistor MF5, and the gate of MOS transistor MF4 are short-circuited and connected to the output node IBIAS of the on-chip bias current circuit, inputting a constant DC current. The gate VB of MOS transistor MF3 is connected to the bias voltage circuit to ensure that MOS transistor MF3 operates in the saturation region. The source of MOS transistor MF3 is connected to the drain of MOS transistor MF4, and the source of MOS transistor MF3 is connected to one end of resistor Rd, which also serves as the current flag output CRout. In addition to being connected to the drain of MOS transistor MF3, resistor Rd also has its other end connected to VDD. VDD and VSS are the power supply and ground signals of the current-subtracting optoelectronic analog front end. Resistors RF1, RF2, and Rd are on-chip polysilicon resistors. Iin is the output of the differential shaping circuit, and Vout is the output of the transimpedance amplifier, which is connected to the post-amplifier module 5.

[0033] Continue reading Figure 2 Resistors RF1 and RF2 together form the feedback resistors of transimpedance amplifier 4, which determine the front-end gain. In practical implementations, the actual resistance of resistor RF1 is determined by the width-to-length ratio of MOS transistors MF1 and MF2. When the input current is low, MOS transistors MF1 and MF2 operate in the subthreshold region, and the leakage current of these transistors approaches zero. When the input current increases, the voltage drop across resistor RF1 increases, causing MOS transistors MF1 and MF2 to conduct, and the leakage current is input into the current detection circuit formed by resistor Rd and the MOS transistors. Assuming the current source formed by MOS transistors MF4 and MF5 has a constant DC current, the input self-reducing current fully loads Rd, causing the voltage at node CRout to change, thereby enabling current detection in the self-reducing current branch.

[0034] When the input photocurrent is low, the voltage difference across resistor RF1 is low, and the gate-source voltages of MOS transistors MF1 and MF2 are below the threshold voltage. At this point, the self-subtraction branch operates in the cutoff region. Even considering the secondary effects of the subthreshold region, the branch current is generally low and has no impact on the operation of transimpedance amplifier 4. When the input current increases, the voltage difference across resistor RF1 increases, turning on MOS transistors MF1 and MF2. Current is then shunted through the path from MOS transistor MF1 to MOS transistor MF2, achieving current self-subtraction and widening the dynamic range. When the current self-subtraction branch is operating, the current sense amplifier, consisting of resistor Rd, MOS transistors MF3, MOS transistors MF4, and MOS transistors MF5, synchronously reads the current magnitude of the self-subtraction branch and outputs it as a current flag CRout. Rd is the current sense resistor, MF3 is the amplifier transistor, and MF4 and MF5 provide the circuit bias. This structure not only detects the operating status of the self-subtraction branch, but also provides echo strength information for back-end digital processing. In fact, since the DC current of the current branch is almost zero under all working conditions, the noise introduced by it can be ignored to ensure the detection sensitivity of the receiver.

[0035] like Figure 3 As shown, the transimpedance amplifier 4 includes a primary amplifier circuit, a compensation capacitor, a secondary amplifier circuit and a second constant current bias circuit. The primary amplifier circuit is connected to the secondary amplifier circuit and both are connected to the second constant current bias circuit. A compensation capacitor is also connected between the primary amplifier circuit and the secondary amplifier circuit. The first-stage amplifier circuit includes MOS transistors M1 to M8, which are sequentially numbered. The gate of the MOS transistor M1 is connected to the output terminal Iin of the differential circuit 2. The source of the MOS transistor M1 is short-circuited with the source of the MOS transistor M2 and then connected to the drain of the MOS transistor M13. The drain of the MOS transistor M1 is connected to the source of the MOS transistor M3, the drain of the MOS transistor M2 is connected to the source of the MOS transistor M4, and the gate of the MOS transistor M3 is connected to the gate of the MOS transistor M4. The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M5, the drain of the MOS transistor M4 is connected to the drain of the MOS transistor M6, and the gate of the MOS transistor M5 is connected to the gate of the MOS transistor M6. The source of the MOS transistor M5 is connected to the drain of the MOS transistor M7, the source of the MOS transistor M6 is connected to the drain of the MOS transistor M8, the gate of the MOS transistor M7 is connected to the gate of the MOS transistor M8, and the connection node is connected to the drain of the MOS transistor M5. The source of the MOS transistor M7 is connected to the source of the MOS transistor M8 and is connected to the power supply VDD.

[0036] Continue reading Figure 3The secondary amplifier circuit includes sequentially numbered MOS transistors M9 to M12. The source of MOS transistor M9 is short-circuited with the source of MOS transistor M10 and then connected to the drain of MOS transistor M14. The gate of MOS transistor M9 is connected to the drain of MOS transistor M4. The drain of MOS transistor M9 is connected to the drain of MOS transistor M11. The drain of MOS transistor M10 is connected to the drain of MOS transistor M12. The gate of MOS transistor M11 is connected to the gate of MOS transistor M12, and the connection node is connected to the drain of MOS transistor M10. The source of MOS transistor M11 is connected to the source of MOS transistor M12 and is also connected to the power supply VDD. One end of the compensation capacitor Cc is connected to the drain of MOS transistor M8, and the other end of the compensation capacitor Cc is connected to the drain of MOS transistor M11.

[0037] Continue reading Figure 3 The second constant current bias circuit includes a MOS transistor M13 and a MOS transistor M14. The gate of the MOS transistor M13 and the gate of the MOS transistor M14 are both connected to the power supply VB1, and the source of the MOS transistor M13 and the source of the MOS transistor M14 are connected to the power supply VSS. Figure 3 VB1 to VB4 are the gate bias voltages of the corresponding connected MOS tubes.

[0038] Through the above technical solution, the present invention does not require an external signal to control the gain change of the circuit. The feedback circuit 3 realizes the shunt function when the photocurrent exceeds the detection range of the transimpedance amplifier 4. When the photocurrent is large, the feedback circuit 3 is automatically turned on to realize the current shunt function. At this time, the transimpedance amplifier 4 will not be saturated due to the excessively high echo laser power. The input current value can be adjusted according to the laser echo power to expand the front-end dynamic range, thereby realizing the widening of the input end dynamic range. The circuit structure is simple, the power consumption is low, and the engineering application value is high.

[0039] The above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit the same. Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the aforementioned embodiments, or make equivalent replacements for some of the technical features therein. However, these modifications or replacements do not deviate the essence of the corresponding technical solutions from the spirit and scope of the technical solutions of the various embodiments of the present invention.

Claims

1. A current-reduction laser radar optoelectronic analog front end, characterized in that: The invention comprises a differential circuit, a feedback circuit and a transimpedance amplifier, wherein one end of the differential circuit is connected to a photodetector for detecting laser echo signals, the other end of the differential circuit is respectively connected to the input end of the feedback circuit and the input end of the transimpedance amplifier, and the output end of the feedback circuit is connected to the output end of the transimpedance amplifier; the differential circuit calculates the first-order derivative of the input waveform, the transimpedance amplifier is used to pre-amplify the photocurrent output by the differential circuit and output a voltage signal, and the feedback circuit implements a shunting function when the photocurrent exceeds the detection range of the transimpedance amplifier; the feedback circuit comprises a feedback resistor, a current self-reduction branch, a current detection circuit and a first constant current bias circuit, the feedback resistor is connected to the current self-reduction branch, the input end of the current self-reduction branch is connected to the output end of the differential circuit, the current self-reduction branch is connected to the current detection circuit and both are connected to the first constant current bias circuit; the feedback resistor comprises a resistor RF1 and a resistor RF2, the current self-reduction branch comprises a MOS transistor MF1 and a MOS transistor MF2, and the current detection circuit comprises a resistor R d The first constant current bias circuit includes a MOS transistor MF4 and a MOS transistor MF5. The gates of the MOS transistors MF1 and MF2 are short-circuited and connected to the output end of the differential circuit. The drain of the MOS transistor MF1 is connected to the power supply VDD, and the drain of the MOS transistor MF2 is connected to the drain of the MOS transistor MF4. The sources of the MOS transistors MF1 and MF2 are short-circuited and connected to a common node formed by the series connection of resistors RF1 and RF2. The source of the MOS transistors MF5 and MF4 are short-circuited and connected to the power supply VSS. The drain of the MOS transistor MF5, the gate of the MOS transistor MF5, and the gate of the MOS transistor MF4 are short-circuited and connected to the output node IBIAS of the on-chip bias current circuit to input a constant DC current. The gate VB of the MOS transistor MF3 is connected to the bias voltage circuit to ensure that the MOS transistor MF3 operates in the saturation region. The source of the MOS transistor MF3 is connected to the drain of the MOS transistor MF4, and the drain of the MOS transistor MF3 is connected to the resistor R d One end is connected, the drain of MOS tube MF3 is also the current flag output CRout, the resistor R d The other end is connected to the power supply VDD.

2. The current self-subtraction laser radar photoelectric analog front end according to claim 1, characterized in that: A post-amplifier is also included, wherein the input end of the post-amplifier is connected to the output end of the transimpedance amplifier.

3. The current self-subtraction laser radar photoelectric analog front end according to claim 1, characterized in that: It also includes a zero-crossing detection circuit for analyzing echo characteristics and a time resolution circuit for solving the distance of the target. The zero-crossing detection circuit is connected to the output end of the feedback circuit, and the zero-crossing detection circuit is also connected to the output end of the post-amplifier and the time resolution circuit.

4. The current self-subtraction laser radar photoelectric analog front end according to claim 1, characterized in that: The differential circuit includes an inductor L and a capacitor C, wherein the inductor L and the capacitor C are connected in series, and the series connection point serves as the output end of the differential circuit and is connected to the transimpedance amplifier and the feedback circuit. The non-series end of the inductor L is connected to the ground, and the non-series end of the capacitor C, i.e., the input end of the differential circuit, is connected to the photodetector.

5. The current self-subtraction laser radar photoelectric analog front end according to claim 1, characterized in that: The transimpedance amplifier includes a primary amplifier circuit, a compensation capacitor, a secondary amplifier circuit and a second constant current bias circuit. The primary amplifier circuit is connected to the secondary amplifier circuit and both are connected to the second constant current bias circuit. A compensation capacitor is also connected between the primary amplifier circuit and the secondary amplifier circuit.

6. The current self-subtraction laser radar photoelectric analog front end according to claim 5, characterized in that: The first-stage amplifier circuit includes MOS transistors M1 to M8, which are sequentially numbered. The gate of the MOS transistor M1 is connected to the output end of the differential circuit. The source of the MOS transistor M1 is short-circuited with the source of the MOS transistor M2 and then connected to the second constant-current bias circuit. The drain of the MOS transistor M1 is connected to the source of the MOS transistor M3, the drain of the MOS transistor M2 is connected to the source of the MOS transistor M4, and the gate of the MOS transistor M3 is connected to the gate of the MOS transistor M4. The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M5, the drain of the MOS transistor M4 is connected to the drain of the MOS transistor M6, and the gate of the MOS transistor M5 is connected to the gate of the MOS transistor M6. The source of the MOS transistor M5 is connected to the drain of the MOS transistor M7, the source of the MOS transistor M6 is connected to the drain of the MOS transistor M8, the gate of the MOS transistor M7 is connected to the gate of the MOS transistor M8, and the connection node is connected to the drain of the MOS transistor M5. The source of the MOS transistor M7 is connected to the source of the MOS transistor M8 and is connected to the power supply VDD.

7. The current self-subtraction laser radar photoelectric analog front end according to claim 6, characterized in that: The secondary amplifier circuit includes sequentially numbered MOS transistors M9 to M12. The source of MOS transistor M9 is short-circuited with the source of MOS transistor M10 and then connected to the second constant current bias circuit. The gate of MOS transistor M9 is connected to the drain of MOS transistor M4. The drain of MOS transistor M9 is connected to the drain of MOS transistor M11. The drain of MOS transistor M10 is connected to the drain of MOS transistor M12. The gate of MOS transistor M11 is connected to the gate of MOS transistor M12, and the connection node is connected to the drain of MOS transistor M10. The source of MOS transistor M11 is connected to the source of MOS transistor M12 and is connected to the power supply VDD. One end of the compensation capacitor Cc is connected to the drain of MOS transistor M8, and the other end of the compensation capacitor Cc is connected to the drain of MOS transistor M11.

8. The current self-subtraction laser radar photoelectric analog front end according to claim 7, characterized in that: The second constant current bias circuit includes a MOS transistor M13 and a MOS transistor M14. The drain of the MOS transistor M13 is connected to the source of the MOS transistor M1 and the source of the MOS transistor M2. The drain of the MOS transistor M14 is connected to the source of the MOS transistor M9 and the source of the MOS transistor M10. The gates of the MOS transistor M13 and the gates of the MOS transistor M14 are both connected to the power supply VB1. The sources of the MOS transistor M13 and the source of the MOS transistor M14 are connected and connected to the power supply VSS.

Images