A deadbeat control method with delay compensation for a three-phase rectifier

Abstract

The present invention relates to a deadbeat control method with delay compensation for a three-phase rectifier, comprising the following steps: (1) performing closed-loop control on the three-phase rectifier using deadbeat control to achieve closed-loop control of DC voltage and AC current; (2) sampling k The three-phase grid voltage, three-phase current and DC side voltage at the moment; (3) The vector angle extrapolation method and current prediction calculation formula are used to obtain k +1 time three-phase grid voltage and three-phase current; (4) k Substitute the state variables at time +1 into the current prediction calculation formula and solve k +1 time, and convert it into a drive signal, that is, k Time output k The voltage value at the +1 moment is k +1 time converter to obtain k The method is beneficial to reduce the delay on the deadbeat control of the rectifier and improve the comprehensive performance of the rectifier.

Description

Technical Field

[0001] The present invention belongs to the field of power electronic converter control, and particularly relates to a deadbeat control method with delay compensation for a three-phase rectifier. Background Art

[0002] With the rapid development of power electronic technology, power conversion devices play an important role in industrial production, power transmission and distribution. Bidirectional rectifiers can realize bidirectional power flow conversion between AC-DC and DC-AC, and are widely used in high-voltage and high-power applications. However, with the increase in the voltage and power levels of power converters brought about by the rapid development of industry, two-level converters can no longer meet some specific applications. Compared with traditional two-level converters, three-level converters have the advantages of high voltage withstand, low output harmonic content, and an appropriate number of switching tubes. At the same time, three-level converters have fewer output levels and less modulation difficulty, so three-level converters have been widely used in high-voltage and high-power applications.

[0003] For the control scheme of rectifiers, traditional PI control with an outer DC voltage loop and PI control in the dq coordinate system of the inner current loop is usually adopted. However, the parameter tuning of the PI control scheme is relatively complex, and its dynamic response performance is poor. With the rapid development of predictive control, deadbeat control under its category has been studied by many scholars. The deadbeat control of rectifiers has the advantages of simple control principle and good dynamic response performance. Therefore, in order to achieve the functional control of rectifiers, a PI control with an outer DC voltage loop is adopted for the rectifier voltage outer loop, and a current deadbeat control is adopted for the current inner loop, so as to achieve voltage stable control of the rectifier while controlling reactive power.

[0004] When using a DSP to perform predictive control on a power electronic converter, due to different configuration methods of the EPWM module, there is a delay between the sampling moment of the control signal and the generation moment of the EPWM signal. This delay is caused by the execution time of the control program or a delay of one EPWM cycle. Summary of the Invention

[0005] The purpose of the present invention is to provide a deadbeat control method with delay compensation for a three-phase rectifier, which is beneficial to reducing the influence of delay on the deadbeat control of the rectifier and improving the comprehensive performance of the rectifier.

[0006] To achieve the above purpose, the technical solution adopted by the present invention is: a deadbeat control method with delay compensation for a three-phase rectifier, comprising the following steps:

[0007] (1) Adopt deadbeat control to perform closed-loop control on the three-phase rectifier to achieve closed-loop control of DC voltage and AC current;

[0008] (2) Sample the three-phase grid voltage, three-phase current, and DC-side voltage at time k;

[0009] (3) Obtain the three-phase grid voltage and three-phase current at time k+1 through the vector angle extrapolation method and the current prediction calculation formula;

[0010] (4) Substitute the state variables at time k+1 into the current prediction calculation formula, solve for the optimal voltage vector at time k+1, and convert it into a drive signal, that is, obtain the voltage value at time k+1 output at time k, and after a delay, obtain the voltage at time k+1 on the converter at time k+1.

[0011] Furthermore, a three-phase T-type NPC converter is used for the power conversion from three-phase alternating current to direct current; by adopting a DC voltage outer-loop PI control to stably control the DC voltage, and at the same time, the current inner-loop adopts a deadbeat control to stably control the alternating current, so as to realize the stable control of the active and reactive powers of the rectifier.

[0012] Furthermore, for the one-beat delay existing in the DSP implementation process of the deadbeat control, this method predicts time k+1 at time k by the method of predicting one beat in advance to eliminate the influence of the delay on the deadbeat control.

[0013] Furthermore, this method specifically includes the following steps:

[0014] Step 1: At time k, sample the three-phase grid voltage U a (k) gird 、U b (k) gird 、U c (k) gird , the three-phase grid current I a (k) rec 、I b (k) rec 、I c (k) rec and the DC voltage V dc ; at the same time, call the ideal converter output voltage V α * (k) rec 、V β * (k) rec calculated at the previous moment for subsequent control;

[0015] Step 2: Subtract and compare the true value V dc with the reference value V dc * , input it into the PI controller, and obtain the output value I d *(k + 1) rec ;

[0016] Step 3: Set the reference reactive power Q ref to zero, that is, the reference reactive current I q * (k + 1) rec is zero, and perform the inverse Park transformation on I d * (k) rec and I q * (k) rec to obtain the reference current I α * (k + 1) rec , I β * (k + 1) rec ;

[0017] Step 4: To calculate the converter output voltage at the (k + 1)-th moment at the k-th moment, perform the derivation and calculation of all variables for the next moment; through the coordinate transformation of the sampled state variables and the reference current value, obtain the grid voltages V α (k) gird , V β (k) gird and the current I α * (k + 1) rec , I β * (k + 1) rec , and perform vector angle extrapolation on these four quantities to obtain the grid voltages U α (k + 1) gird , U β (k + 1) gird and the reference current I α * (k + 2) rec , I β * (k + 2) rec ; Use the converter output voltage V α * (k) rec , V β * (k) rec to calculate the actual currents I α (k + 1) rec , I β (k + 1) rec ;

[0018] Step 5: Use the current prediction formula again to calculate the variables at the (k + 1)th moment, and obtain the output voltage V of the converter at the (k + 1)th moment α * (k + 1) rec 、V β * (k + 1) rec , and save the output voltage of the converter at the (k + 1)th moment for use in the calculation of the next control cycle; meanwhile, after the program execution at the kth moment is completed, convert the voltage value into the comparison value of EPWM, and wait until the EPWM module generates a driving signal at the (k + 1)th moment;

[0019] Step 6: At the (k + 1)th moment, after the driving signal is generated and the output voltage at the (k + 1)th moment is generated on the converter, the target current for control can be obtained at the (k + 2)th moment.

[0020] Compared with the prior art, the present invention has the following beneficial effects: This method adopts PI control for the DC voltage outer loop and deadbeat control for the current inner loop. It can not only solve the steady-state error problem through PI control, but also solve the dynamic response performance problem through deadbeat control, so as to achieve better comprehensive performance. This method adopts a one-beat delay compensation control scheme to eliminate the performance impact of delay on the rectifier control for the one-beat delay problem existing in the implementation process of DSP, thereby improving the comprehensive performance of the rectifier. Description of the Drawings

[0021] Figure 1 is the topological structure diagram of the three-level T-type NPC rectifier in the embodiment of the present invention;

[0022] Figure 2 is the schematic diagram of the influence of delay on the driving signal generated by DSP in the embodiment of the present invention;

[0023] Figure 3 is the schematic diagram of the deadbeat control of the rectifier without considering any delay in the embodiment of the present invention;

[0024] Figure 4 is the comparison schematic diagram of the deadbeat control of the rectifier under different delay conditions in the embodiment of the present invention;

[0025] Figure 5 is the schematic diagram of the deadbeat control of the rectifier considering one-beat delay compensation in the embodiment of the present invention;

[0026] Figure 6 is the principle diagram of the vector angle extrapolation method in the embodiment of the present invention;

[0027] Figure 7 is the control block diagram of the three-level T-type NPC rectifier with one-beat delay compensation control in the embodiment of the present invention. Detailed Implementation Modes

[0028] The present invention will be further described below in conjunction with the accompanying drawings and embodiments.

[0029] It should be noted that the following detailed description is exemplary and is intended to provide further illustration of the present application. Unless otherwise specified, all technical and scientific terms used herein have the same meaning as commonly understood by those of ordinary skill in the technical field to which the present application belongs.

[0030] It should be noted that the terms used herein are only for describing specific implementation modes and are not intended to limit the exemplary implementation modes according to the present application. As used herein, unless the context clearly indicates otherwise, the singular form is also intended to include the plural form. In addition, it should be understood that when the terms "comprise" and / or "include" are used in this specification, they indicate the presence of features, steps, operations, devices, components, and / or combinations thereof.

[0031] This embodiment provides a deadbeat control method with delay compensation for a three-phase rectifier, including the following steps:

[0032] (1) Adopt deadbeat control to perform closed-loop control on the three-phase rectifier to achieve closed-loop control of the DC voltage and AC current.

[0033] (2) Sample the three-phase grid voltage, three-phase current, and DC-side voltage at time k.

[0034] (3) Obtain the three-phase grid voltage and three-phase current at time k + 1 through the vector angle extrapolation method and the current prediction calculation formula.

[0035] (4) Substitute the state variables at time k + 1 into the current prediction calculation formula, solve for the optimal voltage vector at time k + 1, and convert it into a drive signal, that is, obtain the voltage value at time k + 1 output at time k, and obtain the voltage at time k + 1 on the converter at time k + 1 after delay.

[0036] This method can not only solve the delay problem of deadbeat control but also solve the delay problem of finite set model predictive control.

[0037] This method uses a three-phase T-type NPC converter to perform power conversion from three-phase alternating current to direct current; by adopting a DC voltage outer-loop PI control to stably control the DC voltage, and at the same time, the current inner-loop adopts deadbeat control to stably control the AC current, so as to realize the stable control of the active and reactive powers of the rectifier. This method aims at the one-beat delay existing in the DSP implementation process of deadbeat control, predicts time k + 1 at time k by the method of predicting one beat in advance, and eliminates the influence of the delay on deadbeat control.

[0038] As shown Figure 1 in the figure, it is the topological structure diagram of a three-level T-type NPC rectifier, which absorbs power from the three-phase power grid, passes through the filter inductor L rec , and then is rectified by the T-type NPC converter to obtain a stable voltage V dc on the DC bus, which is provided to the load. Among them, U a grid , U b grid , U c grid are the three-phase power grid voltages, L rec is the filter inductor of the three-phase rectifier, C1 and C2 are the DC bus voltage-dividing capacitors, and R dc load is the DC load resistance of the rectifier.

[0039] The control objective of the rectifier is to achieve a stable DC voltage output and at the same time control the reactive current part on the grid side. When the DC side load suddenly increases, the active power provided by the rectifier does not match the power of the DC side load, which will cause the load to absorb power from the DC bus, resulting in a decrease in the DC voltage. In order to maintain the stability of the DC bus voltage, voltage feedback control needs to be carried out after the DC voltage drops to increase the active power provided by the rectifier. Therefore, the rise or fall of the voltage on the DC bus represents the active power that the DC bus needs to reduce or increase. When the DC side load suddenly increases, the active power provided by the rectifier does not match the power of the DC side load, which will cause the load to absorb power from the DC bus, resulting in a decrease in the DC voltage. In order to maintain the stability of the DC bus voltage, voltage feedback control needs to be carried out after the DC voltage drops to increase the active power provided by the rectifier. Therefore, the rise or fall of the voltage on the DC bus represents the active power that the DC bus needs to reduce or increase.

[0040] This method uses a PI controller to perform closed-loop control on the DC voltage. When the DC voltage is lower than the rated voltage of the DC bus, the PI controller will perform PI control calculation on the difference between the two and transmit its output value to the current inner loop as the reference value of the current inner loop for current closed-loop control.

[0041]

[0042] In the formula, I d *rec is the reference active current of the rectifier, ΔV dc is the difference between the reference voltage value of the DC bus and the actual DC bus voltage, V dc * is the reference value of the DC voltage, k p and k iis the proportional-integral coefficient of the PI controller. However, the DC voltage closed-loop control can only reflect the change in active power. Therefore, the current reference value I of the reactive power needs to be added q *rec , so that the independent control of active and reactive power can be achieved.

[0043] The current closed-loop control based on predictive control is a current closed-loop control under the α-β coordinate system and belongs to the control of alternating quantities. The PI controller controls the DC quantity, and its output value is also a stable DC quantity at steady state. Therefore, it is necessary to convert the DC quantity into an alternating quantity. According to the knowledge of coordinate transformation, a three-phase rotating power grid can be equivalent to a rotating vector with a constant amplitude. By using the rotating coordinate transformation formula, the three-phase power grid can be transformed into a DC quantity. Similarly, by performing an inverse rotation transformation on the DC quantity, the active current I of the rectifier can be achieved d *rec , and the reactive reference current I q *rec The transformation of these two DC quantities to alternating quantities in the α-β coordinate system.

[0044]

[0045] Equation (2) is the Park inverse transformation formula, where x d and x q are the coordinate components of the dq coordinate system, and θ is the rotation angle of the rotating coordinate system. Through the Park inverse transformation, the components of the d-axis and q-axis can be converted into alternating quantities in the α-β coordinate system.

[0046] Therefore, substituting I d *rec and I q *rec into Equation (2) gives Equation (3).

[0047]

[0048] In the equation, I α * (k + 1) rec 、I β * (k + 1) rec are the components of the reference current in the α-β coordinate system, θ k is the power grid rotation angle at time k, T s is the control period, and T o is the fundamental wave period of the power grid.

[0049] So far, the reference value of the current in the α-β stationary coordinate system has been obtained, which prepares for the next-step current inner-loop predictive control. Next, the current predictive control of the rectifier is analyzed.

[0050] First, list the three-phase KVL equations of the rectifier as shown in Equation (4). Among them, V ao rec 、V bo rec 、V co rec are the three-phase output voltages of the rectifier-side converter, U a grid 、U b grid 、U c grid are the three-phase grid voltages, I a rec 、I b rec 、I c rec are the three-phase rectifier currents, and U no rec is the common-mode voltage of the rectifier.

[0051]

[0052] Perform a Clark transformation on the KVL equations of the three-phase rectifier to obtain the KVL equations in the α-β stationary coordinate system as shown in Equation (5).

[0053]

[0054] In the formula, V α rec 、V β rec are the components of the output voltage in the α-β coordinate system, I α inv 、I β inv are the components of the inverter output current in the α-β coordinate system, and U α grid 、U β grid are the components of the grid voltage in the α-β coordinate system. Discretize the expression (5) using the discrete formula to obtain the discrete voltage KVL equations as shown in Equation (6).

[0055]

[0056] Among them, the coefficient matrices A rec and B rec are listed as follows.

[0057]

[0058] It can be seen from this formula that due to the grid voltage U α (k)grid , U β (k) grid is known, and the coefficient matrix is known. Therefore, the reference current I α (k + 1) rec , I β (k + 1) rec at the next moment is in a one-to-one correspondence with the converter output voltages V α (k) rec , V β (k) rec . If it is desired to make the current at the (k + 1)-th moment reach the target value I α * (k + 1) rec , I β * (k + 1) rec , it is only necessary to substitute the target current at the (k + 1)-th moment into Equation (6) to inversely deduce the ideal output voltage V α * (k) rec , V β * (k) rec .

[0059] However, since predictive control is discrete control, there are three moments in the control implementation process: the controller samples the A / D signal; the control program ends and the EPWM comparison value is written; the EPWM value is loaded and the drive signal is generated. Due to different configuration methods of the EPWM, in the process from writing the comparison value into the EPWM module to the EPWM module generating the drive signal, it can be generated immediately or in the next EPWM cycle. However, both methods are affected by delay during predictive control. Among them, the immediate generation configuration method corresponds to only considering the computational delay, and the drive signal being generated at the start of the next EPWM control cycle corresponds to a one-beat delay.

[0060] To study more deeply the influence of delay on the DSP generating the drive signal, it is necessary to analyze the drive signal generated by the EPWM module under different conditions. The specific manifestation is as Figure 2 shown, where cases A, B, and C correspond to the cases of not considering delay, considering computational delay, and considering one-beat delay, respectively.

[0061] Since the time delay is not considered in Case A, the control program should be executed instantaneously and output to the EPWM module of the DSP to generate a driving signal. Therefore, in Case A, there is no time delay between the output signal moment and the sampling moment. However, in Case B, the calculation time delay is considered, and after the control program is executed, it will be immediately output to the EPWM module of the DSP to generate a driving signal. Therefore, in Case B, there is a time delay of one calculation amount between the output signal moment and the sampling moment. Although the calculation amount in Case C is the same as that in Case B, in Case C, the output value calculated in the previous cycle will generate a driving signal in the next cycle. In this case, the generation of the driving signal is different from the sampling time by a complete EPWM cycle T s . After analyzing the three DSP implementation cases, the current control for these cases will be analyzed in detail below.

[0062] As Figure 3 shown, it is a schematic diagram of the deadbeat control of a rectifier without considering any time delay, that is, in an ideal state. The blue dotted line in the figure is the reference current curve, and the red dotted line is the curve of the actual current. At time k, the deadbeat predictive control samples the control variables I α (k) rec , I β (k) rec , U α (k) grid , U β (k) grid , and calculates the output voltage V α * (k) rec , V β * (k) rec of the converter at time k according to the reference current I α * (k + 1) rec , I β * (k + 1) rec using the converter voltage calculation formula (3-1). The specific calculation formula is as follows. α * (k) rec , V β * (k) rec .

[0063]

[0064] As can be seen from the figure, without considering the time delay of program execution and the time delay of EPWM value loading, the A / D sampling process is completed at time k, and the program is executed at k' and the value is immediately output to the driving signal. Since it is in an ideal state, the calculation time Δt = k' - k can be ignored. Therefore, its output voltage V α * (k) rec , V β * (k)rec Output to the converter at time k. Then, in the ideal state, the current I at time k+1 α (k+1) rec 、I β (k+1) rec is equal to I α * (k+1) rec 、I β * (k+1) rec .

[0065] In the above part, the schematic diagram of deadbeat control in the ideal state has been completed. The core idea is to perform predictive calculations at time k and output the ideal voltage at time k, so as to obtain the reference current value at time k+1. However, due to the non-negligible calculation time of the controller and considering different configuration methods of EPWM, there is a certain delay in the system, which brings a certain error to the implementation process of control.

[0066] Figure 4 The following shows the comparison schematic diagram of deadbeat control of the rectifier under different conditions. In the figure, Group A, Group B, and Group C respectively represent the current control schematic diagrams without considering delay, considering computational delay, and considering one-beat delay.

[0067] Assume that at time k, the current values I(k) in the three cases are the same. Since the reference current I * (k+1) is also the same, in order to make the actual current value I(k+1) at the next moment equal to I * (k+1), the converter voltage calculation formula of Equation (8) is used for the current and grid voltage at the current moment to obtain the ideal converter output voltage V * (k). Since the current, grid voltage at time k and the reference current at time k+1 are the same in the three cases, the converter voltages V1 * (k) = V2 * (k) = V3 * (k) = V * (k).

[0068] However, as can be seen from Figure 4 , starting from time k, the three methods start to calculate. Since there is no computational delay in case A, the voltage is applied at time k. Without considering system errors, the current I(k+1) at time k+1 is equal to I *(k + 1), the control effect is achieved. However, in the case of Group B, considering the computational delay caused by program execution, before the program execution is completed, since the EPWM value is not updated, the output will be based on the voltage at the previous moment, that is, as shown by the yellow solid line in the figure, it will continue to decrease. When the program execution is completed, the EPWM output is immediately performed. However, at this time, a time of Δt = k' - k has passed, so the control effect cannot be achieved at the (k + 1) moment. In the case of Group C, the voltage value calculated at the k' moment will take effect at the (k + 1) moment, so the control effect cannot be achieved necessarily. Therefore, how to reduce the influence of the control effect deterioration caused by the delay is the problem to be solved in this chapter.

[0069] This method illustrates the influence of computational delay and one-beat delay on control. As can be seen from Figure 4 , the influence caused by one-beat delay on control is more serious. At the same time, the program execution time varies with the program size. Therefore, in the DSP, the loading configuration scheme for EPWM is usually that the comparison value calculated in the previous cycle is loaded and output at the next moment, and there is a one-beat delay in this configuration scheme. Therefore, this method conducts research on delay compensation control for the case of one-beat delay.

[0070] From the above analysis of the delay problem, it can be seen that if the one-beat delay in the actual control of the DSP is considered, it will cause the converter output voltage V α * (k) rec 、V β * (k) rec calculated at the k moment to act at the (k + 1) moment, resulting in a delay in the converter output voltage and causing a current error. To solve this problem, if the converter voltage V α (k + 1) rec 、V β (k + 1) rec can be calculated at the k moment, the problem caused by the delay can be eliminated.

[0071] As Figure 5 shown, it is a schematic diagram of deadbeat control for one-beat delay compensation control. At the k moment, the sampling of the state variables at the k moment is completed, and at the k' moment, the converter voltages V α (k + 1) rec 、V β (k + 1) rec at the (k + 1) moment are calculated and the voltage is generated at the (k + 1) moment. In this way, there is no delay between the generation of the drive signal and the control moment.

[0072] However, since only the state variables at time k can be sampled at time k, and the converter voltage at time k+1 cannot be directly predicted and calculated, the goal of this section is to sample at time k, derive the state variables at time k+1, and complete the predictive control calculation at time k+1.

[0073] To obtain the converter voltage at time k+1, the calculation formula for the converter voltage at time k+1 shown in Equation (9) is listed.

[0074]

[0075] As can be seen from Equation (9), if it is necessary to calculate V α (k+1) rec and V β (k+1) rec , the grid voltages U α (k+1) grid and U β (k+1) grid at time k+1, the currents I α (k+1) rec and I β (k+1) rec at time k+1, and the reference currents I α * (k+2) rec and I β * (k+2) rec at time k+2 are required. Therefore, it is necessary to derive the corresponding variables at time k+1 given the variables at time k.

[0076] The three-phase grid voltages U a grid , U b grid , and U c grid can be equivalent to a rotating vector with equal amplitude in the case of no significant change, and the three-phase reference currents I a *rec , I b *rec , and I c *rec are the same in the case of no change in the reference current amplitude. The grid voltages and reference currents at different times only correspond to different rotation angles θ in space.

[0077] For example Figure 6As shown, it is the schematic diagram of the vector angle extrapolation method. Since the grid voltage and the reference current at different times in space only differ in the rotating angle, given the grid voltage at time k and the reference current at time k + 1, the vector angle extrapolation method can be used to obtain the grid voltage at time k + 1 and the reference current at time k + 2.

[0078] The formula of the vector angle extrapolation method is shown in Equation (10).

[0079]

[0080] In addition, the true current at time k + 1 needs to be calculated. Since time k + 1 has not occurred yet, the current at time k + 1 cannot be sampled. Therefore, it is necessary to predict and estimate the current at time k + 1 based on the variables at time k. The current prediction formula is also used for current prediction calculation. Since the voltage to be applied at time k is calculated at time k - 1, the converter output voltage at time k is also known.

[0081]

[0082] Finally, the current I at time k + 1 is obtained through the current prediction formula shown in Equation (11). α (k + 1) rec 、I β (k + 1) rec 。 At the same time, the grid voltage U at time k + 1 obtained by the vector angle extrapolation method α (k + 1) rec 、U β (k + 1) rec , and the reference current I at time k + 2 α * (k + 2) rec 、I β * (k + 2) rec 。 Substitute them into Equation (9) to obtain the converter output voltage V at time k + 1 α * (k + 1) rec 、V β * (k + 1) recThus, through the sampling value at time k, the output voltage of the converter at time k, and the reference current at time k+1, it is extrapolated to time k+1, and the output voltage of the converter at time k+1 is obtained through calculation. At time k', the output voltage of the converter is output to the converter. After one beat delay of the digital control system, the output voltage of the converter at time k+1 calculated at time k passes through the EPWM module and realizes voltage output at time k+1. Through the one-beat delay compensation method, there is no delay between the voltage that the converter needs to output and the voltage generated in the actual situation.

[0083] This embodiment provides a deadbeat control method with delay compensation for a three-phase rectifier, and its specific implementation steps are as follows:

[0084] Step 1: At time k, sample the three-phase grid voltages U a (k) gird 、U b (k) gird 、U c (k) gird , the three-phase grid currents I a (k) rec 、I b (k) rec 、I c (k) rec and the DC voltage V dc . At the same time, call the ideal converter output voltages V α * (k) rec 、V β * (k) rec calculated at the previous moment for subsequent control preparation.

[0085] Step 2: Compare the difference between the true value of V dc and the reference value of V dc * , input it to the PI controller, and obtain the output value I d * (k+1) rec .

[0086] Step 3: Set the reference reactive power Q ref to zero, that is, the reference reactive current I q * (k+1) rec is zero, and perform Park inverse transformation on I d * (k) rec and I q * (k) rec to obtain the reference current I in the α-β coordinate systemα * (k + 1) rec 、I β * (k + 1) rec 。

[0087] Step 4: To calculate the converter output voltage at the (k + 1)-th moment at the k-th moment, the derivation calculation for the next moment is performed on all variables. By performing coordinate transformation on the sampled state variables and the reference current value, the grid voltage V α (k) gird 、V β (k) gird and the current I α * (k + 1) rec 、I β * (k + 1) rec are obtained, and vector angle extrapolation is performed on these four quantities to obtain the grid voltage U α (k + 1) gird 、U β (k + 1) gird and the reference current I α * (k + 2) rec 、I β * (k + 2) rec 。The true current I α * (k) rec 、V β * (k) rec at the (k + 1)-th moment is calculated using the converter output voltage V α (k + 1) rec 、I β (k + 1) rec 。

[0088] Step 5: Once again, use the current prediction formula to calculate the variables at the (k + 1)-th moment, obtaining the converter output voltage V α * (k + 1) rec 、V β * (k + 1) rec at the (k + 1)-th moment, and save the converter output voltage at the (k + 1)-th moment for use in the next control cycle calculation. At the same time, after the program execution at the k-th moment is completed, the voltage value is converted into the comparison value of EPWM, and the drive signal is generated by the EPWM module at the (k + 1)-th moment.

[0089] Step 6: At the (k + 1)-th moment, the driving signal generation is completed, and the output voltage at the (k + 1)-th moment is generated on the converter, and thus the target current to be controlled can be obtained at the (k + 2)-th moment.

[0090] So far, the analysis of the deadbeat control of the three-phase T-type NPC rectifier with one-beat delay compensation by this method is completed. The complete schematic diagram of the deadbeat control with one-beat delay compensation for the three-phase rectifier is as Figure 7 shown.

[0091] The above are only the preferred embodiments of the present invention, and are not intended to limit the present invention in other forms. Any person skilled in the art may use the disclosed technical content to make changes or modifications into equivalent embodiments with equivalent changes. However, any simple modifications, equivalent changes and modifications made to the above embodiments based on the technical essence of the present invention without departing from the technical solution content of the present invention still belong to the protection scope of the technical solution of the present invention.

Claims

1. A deadbeat control method with delay compensation for a three-phase rectifier, characterized in that, It includes the following steps: (1) Adopt deadbeat control to conduct closed-loop control on the three-phase rectifier to achieve closed-loop control of the DC voltage and AC current; (2) Sample the three-phase grid voltage, three-phase current, and DC-side voltage at the kth moment; (3) Obtain the three-phase grid voltage and three-phase current at the (k + 1)th moment through the vector angle extrapolation method and current prediction calculation formula; (4) Substitute the state variables at the (k + 1)th moment into the current prediction calculation formula, solve for the optimal voltage vector at the (k + 1)th moment, and convert it into a drive signal, that is, obtain the voltage value at the (k + 1)th moment output at the kth moment, and after delay, obtain the voltage at the (k + 1)th moment on the converter at the (k + 1)th moment; Adopt a three-phase T-type NPC converter to conduct power conversion from three-phase alternating current to direct current; by adopting DC voltage outer-loop PI control to stably control the DC voltage, and at the same time, the current inner loop adopts deadbeat control to stably control the AC current, so as to achieve stable control of the active and reactive powers of the rectifier; This method aims at the one-beat delay existing in the DSP implementation process of deadbeat control, and predicts the (k + 1)th moment at the kth moment by the method of predicting one beat in advance to eliminate the influence of the delay on deadbeat control; The deadbeat control method with delay compensation for the three-phase rectifier specifically includes the following steps: Step 1: At time k, sample the three-phase grid voltages U a (k) gird 、U b (k) gird 、U c (k) gird , the three-phase grid currents I a (k) rec 、I b (k) rec 、I c (k) rec and the DC voltage V dc ; Meanwhile, call the ideal converter output voltages V α * (k) rec 、V β * (k) rec calculated at the previous time for subsequent control preparation; Step 2: Subtract the true value V dc from the reference value V dc * for difference comparison, input it into the PI controller, and obtain the output value I d * (k + 1) rec ; Step 3: Set the reference reactive power Q ref to zero, i.e., the reference reactive current I q * (k + 1) rec is zero, and perform the inverse Park transformation on I d * (k) rec and I q * (k) rec to obtain the reference currents I α * (k + 1) rec and I β * (k + 1) rec ; Step 4: In order to calculate the converter output voltage at time k+1 at time k, all variables are derived and calculated at the next moment; by performing coordinate transformation on the sampled state variables and reference current values, the grid voltage V in the α-β coordinate system is obtained. α (k) gird 、V β (k) gird And the current I α * (k+1) rec ,I β * (k+1) rec , and perform vector angle extrapolation on these four quantities to obtain the grid voltage U at the next moment α (k+1) gird , U β (k+1) gird And the reference current I α * (k+2) rec ,I β * (k+2) rec ; Use the converter output voltage V at time k through the current prediction formula α * (k) rec 、V β * (k) rec Calculate the real current I at time k+1 α (k+1) rec ,I β (k+1) rec ; Step 5: Use the current prediction formula again to calculate the variables at the (k + 1)-th moment, and obtain the output voltage V of the converter at the (k + 1)-th moment α * (k + 1) rec and V β * (k + 1) rec , and save the output voltage of the converter at the (k + 1)-th moment for use in the calculation of the next control cycle; meanwhile, after the program execution at the k-th moment is completed, convert the voltage value into the comparison value of EPWM, and wait until the EPWM module generates a driving signal at the (k + 1)-th moment Step 6: At the (k + 1)th moment, after the generation of the drive signal is completed and the output voltage at the (k + 1)th moment is generated on the converter, the target current of the control can be obtained at the (k + 2)th moment.

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