An active noise reduction method based on analog circuit design

Abstract

The present invention provides an active noise reduction method based on analog circuit design, including a background current analysis module, an analog sampling feedback control unit, a power amplification unit, G1, G2, a signal conditioning circuit, a G3 current booster and an analog active noise reduction method equivalent circuit. In the power supply excitation stage, the background current analysis module inversely calculates the primary busbar current of the CTX to be calibrated through the secondary signal of CT0. G1 receives the CTX detection signal and forms a compensation current control signal. The compensation current control signal is transmitted to G2. The output end of the G2 circuit is signal-connected to the G3 current booster. The output end of the G3 current booster releases the output current and forms a closed loop. Moreover, G1 and the CTX detection signal are respectively inside the analog sampling feedback control unit, and G2 and the G3 current booster are respectively inside the power amplification unit.

Description

Technical Field

[0001] The present invention relates to the technical field of calibration of current transformers for metering, and particularly relates to an active noise reduction method based on analog circuit design. Background Art

[0002] Low-voltage current transformers for metering (hereinafter referred to as current transformers) are national mandatory inspection instruments and are widely used in power grids. They have long been emphasized by the state and the power industry. The inventory of current transformers in China is about 80 million units, with an annual increase of more than 3 million units. Their metering performance directly affects the fairness and justice of electric energy trade and is related to the national economy and people's livelihood. The full-range calibration of low-voltage current transformers for metering under live conditions is different from laboratory calibration. In addition to power frequency current, there are also harmonics and environmental noise in the primary current of low-voltage current transformers. There are various environmental noises in the working conditions of the primary current, making it impossible to distinguish power frequency current and noise current during the sampling process, and it is impossible to completely cancel the primary current during the inverting amplification process of the current signal, resulting in relatively large errors. Therefore, it is very necessary to introduce a corresponding signal noise reduction method in the system design to eliminate noise.

[0003] Among them, the Chinese patent "A current transformer capable of reducing current noise" with the publication number CN201721184877.6 relates to a current transformer capable of reducing current noise, including a protective cover, a skeleton, an insulating sleeve and a sound insulation layer. A first magnetic conductor and a second magnetic conductor are respectively arranged inside the protective cover, and a first connecting rod and a second connecting rod are sequentially arranged below the first magnetic conductor. The skeleton is located between the first magnetic conductor and the second magnetic conductor, and a wire is arranged outside the skeleton. The insulating sleeve is located outside the wire. A third connecting rod and a fourth connecting rod are sequentially arranged below the second magnetic conductor. The sound insulation layer is located on the inner surfaces of the protective cover, the first magnetic conductor, the insulating sleeve and the second magnetic conductor, and can effectively reduce noise.

[0004] This solution can effectively reduce noise, but it belongs to passive noise reduction and cannot dynamically cancel the background current. Summary of the Invention

[0005] In order to effectively distinguish the power frequency current and the noise current in the primary current of the current transformer and improve the accuracy of the current transformer calibration work, the present invention proposes an active noise reduction method based on analog circuit design, which solves the problem of noise reduction of the primary current during the current transformer calibration process.

[0006] The technical problem to be solved by the present invention is how to obtain an accurate primary power frequency current in the presence of background current.

[0007] In view of the above problems, the technical solution proposed by the present invention is:

[0008] An active noise reduction method based on analog circuit design, comprising:

[0009] The analog circuit includes a background current analysis module, an analog sampling feedback control unit, a power amplification unit, a signal conditioning circuit G1, a power amplification and signal conditioning circuit G2, a signal conditioning circuit, a current booster G3, and an analog active noise reduction method equivalent circuit, wherein,

[0010] In the power supply excitation stage, the background current analysis module inversely calculates the primary busbar current of the calibrated CTX through the CT0 secondary signal. The G1 receives the CTX detection signal, forms a compensation current control signal, and transmits the compensation current control signal to G2. The output end of the G2 circuit is signal-connected to the G3 current booster. The output end of the current booster G3 releases an output current and forms a closed loop for active noise reduction.

[0011] Optionally, the G1 includes an auxiliary CT2 and a current-voltage conversion circuit, which isolates and samples the secondary current of the CTx and then converts it into a voltage signal. The turns ratio of the auxiliary CT2 is K1, the output current is I1, and after passing through the resistor R1, it is converted into the voltage V fb = I1 * R1;

[0012] G2 consists of two stages. The first stage is an amplifier circuit based on negative feedback, which further amplifies the output voltage of the G1 unit. The amplification factor K2 = R4 / R8. The second stage is a power amplification circuit, which converts the voltage signal into a current output. R6 is the output current sampling resistor, which is connected to the feedback loop to control the output current I2, I2 = Vfb * K2 / R6. The current output I2 of G2 is the input of the G3 current booster.

[0013] Optionally, the G1 is located inside the analog sampling feedback control unit, and the G2 and the G3 current booster are respectively located inside the power amplification unit.

[0014] Optionally, (Ic - I0)G0 * G1 * G2 * G3 = Ic

[0015] Let the background residual current ΔI0’ = Ic - I0, Then

[0016]

[0017] G1 and G2 are small-signal processing circuits, whose transfer functions are easy to quantify and are controlled by broadband.

[0018] Optionally, the current booster G3 is a transformer. Among them, the voltage transfer function H ud and the current transfer function H id :

[0019]

[0020]

[0021] No-load voltage transfer function:

[0022]

[0023] When the secondary side of the transformer is short-circuited, the current transfer function is:

[0024]

[0025] Voltage transfer characteristics under load H ud With no-load voltage transmission characteristics H u The same zero frequency f h , where the resistance of the transformer load is R d ,but:

[0026]

[0027] C ps is a parasitic capacitor, and the resistive load does not change H u The pole frequency f u The size of f u Quality factor Q fu :

[0028]

[0029] When R d Very big makes Q fu >>1, the secondary side of the transformer is open, H ud and H u The characteristics are consistent with R d The decrease of Q fu Decreasing will increase the bandwidth frequency at fu.

[0030]

[0031] in:

[0032] Optionally, the current transfer characteristic H under load id With short-circuit current transmission characteristics H i The same resonant frequency f h If other resonant frequencies change, the load resistance of the transformer is R d ,but:

[0033]

[0034] In the low frequency band, Hid The resonant frequency f appears id1 :

[0035]

[0036] Optionally, if the load contains stray inductance and capacitance, the admittance of the load is set to:

[0037] The formula for the voltage transfer characteristic under the load is:

[0038]

[0039] The voltage transfer characteristic H under the load ud has the same zero - point frequency f as the no - load voltage transfer characteristic H u but the pole frequency changes h ,

[0040]

[0041] The pole frequency of H under the load ud is:

[0042]

[0043] The current transfer characteristic under the load is:

[0044]

[0045] The current transfer characteristic H under the load id has the same resonant frequency f as the short - circuit current transfer characteristic H i : h :

[0046]

[0047] Optionally, in G0(CTx), R 1ctx , L 1ctx are its primary - side impedance, N is the turns ratio, R 2ctx , L 2ctx are its secondary - side impedance, R mctx , L mctx are the exciting impedance, output the secondary current of CTx to G1

[0048] In G1, the secondary current of CTx changes to 1 / k of the original and is applied to R1, the voltage of R1 is UR1, and the output voltage V of the voltage - controlled voltage source fb =U R1 ,

[0049] In G2, Rid is the input impedance of the operational amplifier circuit, and the output current I1 of the voltage-controlled current source is I1 = V fb *K2 / R6,

[0050] In the G3 current booster, Lm and Rm are respectively the excitation impedances of the high-frequency transformer, n is the transformer turns ratio, and Ls and Rs are the leakage impedances.

[0051] Compared with the prior art, the beneficial effects of the present invention are:

[0052] During the calibration process of the current transformer, the power frequency current and the noise current in the primary current of the current transformer can be effectively distinguished. Through closed-loop control, the power frequency / harmonic power supply feedback control system uses the analog sampling feedback control unit to compensate the test input current. Through the error amplification and power amplification in the power amplification unit, the background elimination output current can follow the background current in real time and dynamically cancel the background current.

[0053] The above description is only an overview of the technical solution of the present invention. In order to be able to understand the technical means of the present invention more clearly, it can be implemented according to the content of the specification. And in order to make the above and other objects, features and advantages of the present invention more obvious and understandable, the following specifically describes the specific embodiments of the present invention. Brief Description of the Drawings

[0054] Figure 1 It is the feedback control principle block diagram of the active noise reduction method based on analog circuit design disclosed in the embodiment of the present invention;

[0055] Figure 2 It is the equivalent circuit diagram of G1 of the active noise reduction method based on analog circuit design disclosed in the embodiment of the present invention;

[0056] Figure 3 It is the equivalent circuit diagram of G2 of the active noise reduction method based on analog circuit design disclosed in the embodiment of the present invention;

[0057] Figure 4 It is the equivalent circuit diagram of G3 of the active noise reduction method based on analog circuit design disclosed in the embodiment of the present invention;

[0058] Figure 5 It is the equivalent circuit of the analog active noise reduction method of the active noise reduction method based on analog circuit design disclosed in the embodiment of the present invention. Detailed Embodiments

[0059] Next, the technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments.

[0060] Refer to the attachedFigures 1-5 As shown in Figures 1-5 , an active noise reduction method based on analog circuit design includes a background current analysis module, an analog sampling feedback control unit, a power amplification unit, G1, G2, a signal conditioning circuit, a G3 current booster, and an equivalent circuit of the analog active noise reduction method. During the power supply excitation stage, the background current analysis module uses the CT0 secondary signal (CT is the abbreviation of current transformer) to inversely deduce the primary busbar current of the calibrated CTX. G1 receives the CTX detection signal and forms a compensation current control signal, which is transmitted to G2. The output terminal of the G2 circuit is connected to the G3 current booster signal. The output terminal of the G3 current booster releases the output current and forms a closed loop. And G1 and the CTX detection signal are respectively inside the analog sampling feedback control unit, and G2 and the G3 current booster are respectively inside the power amplification unit. Among them,

[0061] The feedback control transfer function of background squeeze-out:

[0062] (Ic - I0)G0 * G1 * G2 * G3 = Ic

[0063] Wherein, Ic represents the output current, I0 represents the background current, and G0, G1, G2, G3 respectively represent the current transfer functions of the G0, G1, G2, G3 modules

[0064] Let the background residual current ΔI0’ = Ic - I0, Then

[0065]

[0066] G1 and G2 are small-signal processing circuits, whose transfer functions are easy to quantify and are controlled by broadband.

[0067] As an embodiment of the present invention, further, the signal conditioning circuit includes the following circuits,

[0068] G1 includes an auxiliary CT2 and a current-voltage conversion circuit, which isolates and collects the secondary current of CTx and then converts it into a voltage signal. The turns ratio of the auxiliary CT2 is K1, the output current is I1, and after passing through the resistor R1, it is converted into the voltage Vfb = I1 * R1;

[0069] G2 includes two stages. The first stage is an amplification circuit based on negative feedback, which further amplifies the output voltage of the G1 unit, and the amplification factor K2 = R4 / R8. The second stage is a power amplification circuit, which converts the voltage signal into a current output. R6 is the output current sampling resistor, which is connected to the feedback loop to control the output current I2, and I2 = Vfb * K2 / R6. The current output I2 of G2 is the input of the G3 current booster;

[0070] In G1, the transformation ratio of CT2 is fixed, the frequency of the bandwidth is greater than 2 kHz, the closed-loop feedback of the current-voltage conversion current is determined, and the bandwidth is greater than 10 kHz. In G2, the gain of the voltage feedback amplifier circuit is greater than 10 kHz, and the voltage-current conversion coefficient of the power output stage has a bandwidth greater than 2 kHz.

[0071] As an embodiment of the present invention, further, the G3 current booster includes the following circuit

[0072] The G3 current booster is a transformer. Among them, the voltage transfer function H of the transformer under load conditions ud and the current transfer function H id are as follows:

[0073]

[0074]

[0075] where Y is the Y-parameter matrix of the G3 model;

[0076]

[0077] In the matrix, y l and y m are the leakage reactance and magnetizing branch admittance in G3 respectively. y d is the load admittance, ω is the angular frequency, n is the transformation ratio of the transformer in G3, j represents the imaginary unit, d represents the load, y d is the load admittance, and Cps and Cp are stray capacitances.

[0078] No-load voltage transfer function:

[0079]

[0080] When the secondary side of the transformer is short-circuited, the current transfer function:

[0081]

[0082] The voltage transfer characteristic H ud under load has the same zero-frequency f u as the no-load voltage transfer characteristic H h , where the resistance value of the resistive load carried by the transformer is R d , then:

[0083]

[0084] And the resistive load will not change the magnitude of the pole frequency f u of H u , but will change the quality factor Q u of f fu :

[0085]

[0086] When R d Very big makes Q fu >>1, the secondary side of the transformer is open, H ud and H u The characteristics are consistent with R d The decrease of Q fu Decreasing will increase the bandwidth frequency at fu.

[0087]

[0088] in:

[0089] It can be seen that the current transfer characteristic H under load id With short-circuit current transmission characteristics H i The same resonant frequency f h , but other resonant frequencies change, and the load resistance of the transformer is R d ,but:

[0090]

[0091] At low frequencies, the parasitic capacitance C p and C ps The corresponding admittance is very small and can be ignored, H id It can be expressed as the following formula, in which case H id A new resonant frequency f appears id1 :

[0092]

[0093]

[0094] As the frequency increases, parasitic capacitance becomes the main factor affecting the current transmission characteristics. id It can be expressed as the following formula. At this time, H id Another new resonant frequency f appears id2 :

[0095]

[0096]

[0097] Compare f id1 、f id2 and f u have:

[0098]

[0099] Since the magnetizing inductance referred to the same side is greater than the leakage inductance, i.e., n 2 L m >> L s , from the above equation, there is f id1 < f u < f id2 , H id At a lower frequency band less than f u , i.e., the resonance frequency f id1 appears.

[0100] As an embodiment of the present invention, further, it can be seen that resistive loads of different magnitudes do not change the magnitudes of f id1 and f id2 , but will change the quality factors Q id1 and Q id2 near f f1 and Q f2 :

[0101]

[0102] Therefore, if R d is very small, Q f1 and Q f2 are very large, and the bandwidth frequencies of f id1 and f id2 are high; as R d increases, Q f1 and Q f2 decrease, and the bandwidths of f id1 and f id2 will become narrower.

[0103] The load contains stray inductance and capacitance. Let the admittance of the load be:

[0104] The formula for the voltage transfer characteristic under the load is:

[0105]

[0106] The voltage transfer characteristic H ud under the load has the same zero-point frequency f u as the no-load voltage transfer characteristic H h , but the pole frequency changes.

[0107]

[0108] The pole frequency of H ud under the load is:

[0109]

[0110] The current transfer characteristic under load is as follows:

[0111]

[0112] The current transfer characteristic H under load id has the same resonant frequency f as the short - circuit current transfer characteristic H i : h :

[0113]

[0114] (a) In the low - frequency band, the admittances corresponding to the parasitic capacitances Cp and Cps are very small, and H id can be expressed as:

[0115]

[0116] and form two new resonant frequencies:

[0117]

[0118]

[0119] (b) As the frequency is further increased, the parasitic capacitance becomes the main factor affecting the current transfer characteristic, and H id can be expressed as:

[0120]

[0121] and form another resonant frequency:

[0122]

[0123] As an embodiment of the present invention, further, simulate the equivalent circuit of the active noise - canceling method.

[0124] In G0(CTx), R 1ctx , L 1ctx are the primary - side impedances, N is the turns ratio, R 2ctx , L 2ctx are the secondary - side impedances, R mctx , L mctx are the exciting impedances. Output the secondary current of CTx to G1.

[0125] In G1, the secondary current of CTx changes to 1 / k of the original and is added to R1. The voltage of R1 is UR1, and the output voltage V of the voltage - controlled voltage source fb =U R1 ,

[0126] In G2, Rid is the input impedance of the operational - amplifier circuit, and the output current of the voltage - controlled current source I1 = Vfb *K2 / R6,

[0127] In the G3 current booster, Lm and Rm are the exciting impedances of the high-frequency transformer, n is the transformer turns ratio, and Ls and Rs are the leakage impedances.

[0128] The current transfer function is:

[0129]

[0130] There is a pole frequency in the current transfer function:

[0131]

[0132] The voltage across R1 is:

[0133]

[0134] The voltage transfer function is:

[0135]

[0136] From the transfer function, there is a zero frequency w1 and a pole frequency w2.

[0137]

[0138] From the resonant frequency of the voltage and current, it can be seen that when the load L increases, the resonant frequency decreases, and the voltage-current transfer function has the same resonant frequency w2.

[0139] Embodiment 2

[0140] In the present invention, by making the background cancellation output current change in real time following the background current, the effect of dynamically canceling the background current is achieved. The entire circuit is divided into four parts: the CT transfer function G0, the signal conditioning circuit G1, the power amplification and signal conditioning circuit G2, and the current booster G3. Among them, G1 includes an auxiliary CT2 and a current-voltage conversion circuit, whose function is to isolate and collect the secondary current of CTx and then convert it into a voltage signal. The turns ratio of the auxiliary CT2 is K1, the output current is I1, and after passing through the resistor R1, it is converted into the voltage Vfb = I1 * R1; G2 includes two stages. The first stage is an amplification circuit based on negative feedback, which further amplifies the output voltage of the G1 unit, and the amplification factor K2 = R4 / R8. The second stage is a power amplification circuit, which converts the voltage signal into a current output. R6 is the output current sampling resistor, which is connected to the feedback loop to control the output current I2, and I2 = Vfb * K2 / R6. The current output I2 of G2 is the input of the current booster G3 of G3. Furthermore, in the current booster G3, the current signal is amplified and the current is output, so that during the calibration process of the current transformer, the power frequency current and the noise current in the primary current of the current transformer can be effectively distinguished.

[0141] Obviously, those skilled in the art can make various changes and modifications to the present invention without departing from the spirit and scope of the present invention. Thus, if these modifications and variations of the present invention fall within the scope of the claims of the present invention and their equivalent technologies, the present invention is also intended to include these modifications and variations.

Claims

1. An active noise reduction method based on analog circuit design, characterized in that, Including: The analog circuit includes a background current analysis module, an analog sampling feedback control unit, a power amplification unit, a signal conditioning circuit G1, a power amplification and signal conditioning circuit G2, a signal conditioning circuit, a current booster G3, and an analog active noise cancellation equivalent circuit, where In the power supply excitation stage, the background current analysis module inversely calculates the primary busbar current of the CTX to be calibrated through the secondary signal of the current transformer CT0. G1 receives the CTX detection signal, forms a compensation current control signal, and transmits the compensation current control signal to G2. The output end of the G2 circuit is signal-connected to the current booster G3. The output end of the current booster G delivers an output current and forms a closed loop for active noise cancellation. Among them, the feedback control transfer function of background forcing out: (Ic - I0)G0*G1*G2*G3 = Ic (1) Among them, Ic represents the output current, I0 represents the background current, and G0, G1, G2, and G3 respectively represent the current transfer functions of the G0, G1, G2, and G3 modules. Let the background residual current ΔI0’ = Ic - I0, Then （2） The current booster G3 is a transformer. Among them, the voltage transfer function Hud and the current transfer function Hid under the condition that the transformer is loaded: （3） （4） Where Y is the Y-parameter matrix of the G3 model; In the matrix, yl and ym are the leakage magnetic flux and excitation branch admittances in G3 respectively, yd is the load admittance, ω is the angular frequency, n is the turns ratio of the transformer in G3, j represents the imaginary unit, d represents the load, yd is the load admittance, Cps and Cp are stray capacitances; No-load voltage transfer function: （5） When the secondary side of the transformer is short-circuited, the current transfer function: （6） The voltage transfer characteristic Hud under load has the same zero-frequency fh as the no-load voltage transfer characteristic Hu. Among them, the resistance value of the resistive load carried by the transformer is Rd, then Equation (3) can be written as: （7） Cps is the parasitic capacitance, and the resistive load does not change the magnitude of the pole frequency fu of Hu, but will change the quality factor Qfu of fu: （8） When Rd is very large such that Qfu >> 1, the secondary side of the transformer is open, and the characteristics of Hud and Hu are the same. As Rd decreases, Qfu decreases, which will cause the bandwidth frequency at fu to increase. （9） Wherein: ; Among them, in G0(CTx), R1ctx and L1ctx are its primary-side impedances, N is the turns ratio, R2ctx and L2ctx are its secondary-side impedances, Rmctx and Lmctx are the exciting impedances, and the secondary current of CTx is output to G1. In G1, the secondary current of CTx changes to 1 / k of the original and is added to R1. The voltage of R1 is UR1, and the output voltage of the voltage-controlled voltage source is Vfb = UR1. In G2, Rid is the input impedance of the operational amplifier circuit, and the voltage-controlled current source outputs a current I1 = Vfb*K2 / R6. In the current booster G3, Lm and Rm are the exciting impedances of the high-frequency transformer, n is the transformer turns ratio, and Ls and Rs are the leakage impedances.

2. The active noise cancellation method based on analog circuit design according to claim 1, characterized in that: The G1 includes an auxiliary CT2 and a current-voltage conversion circuit, which isolates and collects the secondary current of the CTx and then converts it into a voltage signal. The turns ratio of the auxiliary CT2 is K1, the output current is I1, and after passing through the resistor R1, it is converted into a voltage V fb =I1*R1; G2 consists of two stages. The first stage is an amplifier circuit based on negative feedback, which further amplifies the output voltage of the G1 unit with an amplification factor K2 = R4 / R8. The second stage is a power amplifier circuit that converts the voltage signal into a current output. R6 is the output current sampling resistor, which is connected to the feedback loop to control the output current I2, and I2 = Vfb*K2 / R6. The current output I2 of G2 is the input of the G3 current booster.

3. An active noise reduction method based on analog circuit design according to claim 1 or 2, characterized in that, The G1 is located inside the analog sampling feedback control unit, and the G2 and the G3 current booster are respectively located inside the power amplifier unit.

4. The method according to claim 1, wherein: Current transfer characteristic H under load id Having the same short-circuit current transfer characteristic H i Same resonant frequency f h , if other resonant frequencies change, the resistance value of the load carried by the transformer is R d , then Equation (9) can be written as: （10） At the low frequency band, H id has a resonance frequency f id1 : （11）。 5. The active noise reduction method based on analog circuit design according to claim 1, wherein: If the load contains stray inductance and capacitance, the admittance of the load is set to: The formula for the voltage transfer characteristic under load is: （15） Voltage transfer characteristic H under load ud having the same zero - point frequency f as the no - load voltage transfer characteristic H u but with the pole frequency changed h , （16） H under load ud The pole frequency is: （17） The current transfer characteristic under load is: （18） Current transfer characteristics H under load id Having the same short-circuit current transfer characteristics H i Same resonant frequency f h : （19）。

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