A Modeling Method for High-Precision Mathematical Model of a Dual-Three-Phase Permanent Magnet Synchronous Motor

Abstract

The present invention discloses a method for establishing a high-precision mathematical model of a dual-three-phase permanent magnet synchronous motor. The method calculates several working conditions of a single three-phase winding to establish an interpolation model between current and magnetic flux, and between current and torque, and combines mathematical methods such as inversion to realize the establishment of the mathematical model of the dual-three-phase motor. Compared with the existing mathematical model of the dual-three-phase permanent magnet synchronous motor, the present invention only uses the finite element calculation results of a single three-phase winding and combines the symmetry of the dual-three-phase motor to realize the establishment of the mathematical model of the dual-three-phase permanent magnet synchronous motor, reducing the calculation amount in the modeling process. The modeling method proposed by the present invention can fully consider the effects of electromagnetic coupling, magnetic saturation, harmonic magnetic field, and rotor position on the performance of the motor, and is a high-fidelity and high-precision mathematical model. When using this mathematical model for motor control strategy simulation, the use of the time-step finite element method is avoided, and the simulation time can be saved while ensuring the simulation accuracy.

Description

Technical Field

[0001] The present invention belongs to the field of motors, and relates to a high-precision modeling method for a dual three-phase permanent magnet synchronous motor, specifically to a method for establishing a high-precision mathematical model of a dual three-phase permanent magnet synchronous motor based on finite element calculation results and multi-dimensional interpolation. Background Art

[0002] Permanent magnet synchronous motors have been widely used in the fields of production and life due to their advantages such as small volume, high power density, simple structure, and stable operation. Among them, dual three-phase permanent magnet synchronous motors have attracted much attention because of their advantages such as low voltage and high power, low torque ripple, and fault tolerance performance. However, since the dual three-phase motor has two sets of three-phase windings, the electromagnetic coupling relationship inside it is extremely complex, which makes the mathematical modeling of the dual three-phase permanent magnet synchronous motor extremely complex. Currently, the commonly used modeling method is to regard the dual three-phase motor as the combination of two three-phase motors and model the two three-phase motors separately. However, this method ignores the electromagnetic coupling between the two sets of three-phase windings, resulting in a low-precision mathematical model of the dual three-phase motor and unable to achieve high-quality motor control. Summary of the Invention

[0003] The purpose of the present invention is to provide a method for modeling a high-precision mathematical model of a dual three-phase permanent magnet synchronous motor. This method calculates several working conditions of a single three-phase winding to establish an interpolation model between current and magnetic flux, and between current and torque. Combining mathematical methods such as inversion, the establishment of the mathematical model of the dual three-phase motor is realized. Compared with the existing mathematical model, this model can consider the influence of magnetic saturation, electromagnetic coupling, harmonic magnetic field, and rotor position, has higher precision and fidelity, can improve the control precision, and realize higher-quality motor control.

[0004] The purpose of the present invention is achieved through the following technical solutions:

[0005] A method for modeling a high-precision mathematical model of a dual three-phase permanent magnet synchronous motor includes the following steps:

[0006] Step 1: Perform finite element modeling on the target dual three-phase permanent magnet synchronous motor, apply multiple sets of d-axis currents i d1 and multiple sets of q-axis currents i q1 to the first set of three-phase windings A1B1C1, apply currents of i d2 = 0 and i q2 = 0 to the second set of three-phase windings A2B2C2, perform finite element calculation of the electromagnetic field, and solve the calculation results of magnetic flux and torque;

[0007] Step 2: Take i d1 , i q1 and the rotor electrical angle position θ as independent variables, and take the d-axis magnetic flux ψ d1, q-axis flux linkage ψ q1 and the d-axis flux linkage increment Δψ in the second set of three-phase windings A2B2C2 d2 , q-axis flux linkage increment Δψ q2 As dependent variables, establish the relationships between the four sets of flux linkages and currents ψ d1 (i d1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ), Δψ d2 (i d1 , i q1 , θ), Δψ q2 (i d1 , i q1 , θ);

[0008] Step 3: According to the symmetry and periodicity of the dual three-phase motor structure, shift ψ d1 (i d1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ), Δψ d2 (i d1 , i q1 , θ), Δψ q2 (i d1 , i q1 , θ) by 30°, and obtain ψ d2 (i d2 , i q2 , θ), ψ q2 (i d2 , i q2 , θ), Δψ d1 (i d2 , i q2 , θ), Δψ q1 (i d2 , i q2 , θ), that is:

[0009] ψ d2 (i d2 , i q2 , θ) = ψ d1 (i d1 , i q1 , θ - 30°);

[0010] ψ q2 (i d2 , i q2 , θ) = ψ q1 (i d1 , i q1 , θ - 30°);

[0011] Δψ d1 (i d2 ,i q2 ,θ) = Δψ d2 (i d1 ,i q1 ,θ - 30°);

[0012] Δψ q1 (i d2 ,i q2 ,θ) = Δψ q2 (i d1 ,i q1 ,θ - 30°);

[0013] During dual three - phase operation, the d - axis flux linkage ψ of the first three - phase winding A1B1C1 d1s and the q - axis flux linkage ψ q1s , and the d - axis flux linkage ψ of the second three - phase winding A2B2C2 d2s and the q - axis flux linkage ψ q2s are:

[0014] ψ d1s = ψ d1 (i d1 ,i q1 ,θ) + Δψ d1 (i d2 ,i q2 ,θ);

[0015] ψ q1s = ψ q1 (i d1 ,i q1 ,θ) + Δψ q1 (i d2 ,i q2 ,θ);

[0016] ψ d2s = ψ d2 (i d2 ,i q2 ,θ) + Δψ d2 (i d1 ,i q1 ,θ);

[0017] ψ q2s = ψ q2 (i d2 ,i q2 ,θ) + Δψ q2 (i d1 ,i q1 ,θ);

[0018] Step Four, for ψ d1 (id1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ) is inverted to obtain the current i d1 , i q1 and the flux linkage ψ d1 , ψ q1 and the relationship between the rotor position θ, i d1 (ψ d1 , ψ q1 , θ) and i q1 (ψ d1 , ψ q1 , θ); according to the symmetry and periodicity of the dual three-phase motor structure, i d1 (ψ d1 , ψ q1 , θ) and i q1 (ψ d1 , ψ q1 , θ) are phase-shifted by 30°, resulting in i d2 (ψ d2 , ψ q2 , θ) and i q2 (ψ d2 , ψ q2 , θ), that is:

[0019] i d2 (ψ d2 , ψ q2 , θ) = i d1 (ψ d1 , ψ q1 , θ - 30°);

[0020] i q2 (ψ d2 , ψ q2 , θ) = i q1 (ψ d1 , ψ q1 , θ - 30°);

[0021] Step Five: Taking i d1 , i q1 and the rotor electrical angle position θ as independent variables, and the calculated torque as the dependent variable, the relationship between the torque and the current and the rotor electrical angle position T e1 (i d1 , i q1 , θ) is obtained. According to the symmetry and periodicity of the dual three-phase motor structure, the torque generated when the second set of three-phase windings A2B2C2 operates is obtained by phase-shifting T e1 (i d1 , i q1 , θ) by 30°, that is:

[0022] T e2 (i d2 ,i q2 ,θ) = T e1 (i d1 ,i q1 ,θ - 30°);

[0023] Among them, T e1 and T e2 are both the torques during single - set winding operation and can be expressed as:

[0024] T e1 = 1.5p[ψ d1 (i d1 ,i q1 ,θ)i q1 - ψ q1 (i d1 ,i q1 ,θ)i d1 ;

[0025] T e2 = 1.5p[ψ d2 (i d2 ,i q2 ,θ)i q2 - ψ q2 (i d2 ,i q2 ,θ)i d2 ;

[0026] When the motor operates in a dual - three - phase state, its total torque is:

[0027] T s = 1.5p[ψ d1s i q1 - ψ q1s i d1+ ψ d2s i q2 - ψ q2s i d2 ;

[0028] = T e1 + T e2 + 1.5p[Δψ d1 i q1 - Δψ q1 i d1 + Δψ d2 i q2 - Δψ q2 i d2 ;

[0029] According to the torque, calculate the electrical angular velocity ω of the motor e :

[0030] ωe = ∫[(T s - T L ) / J] dt;

[0031] Wherein, T L is the load torque of the motor, and J is the moment of inertia of the motor.

[0032] According to the electrical angular velocity ω e of the motor, calculate the electrical angle θ of the motor rotor position:

[0033] θ = ∫ω e dt;

[0034] Step 6. Calculate the magnetic flux according to the input voltages u d1 , u q1 , u d2 , u q2 of the motor:

[0035] ψ d1s = ∫(u d1 - i d1 R + ω e ψ q1s ) dt;

[0036] ψ q1s = ∫(u q1 - i q1 R - ω e ψ d1s ) dt;

[0037] ψ d2s = ∫(u d2 - i d2 R + ω e ψ q2s ) dt;

[0038] ψ q2s = ∫(u q2 - i q2 R - ω e ψ d2s ) dt;

[0039] Wherein, R is the resistance value of the stator winding, and ω e is the electrical angular velocity of the motor.

[0040] Compared with the prior art, the present invention has the following advantages:

[0041] 1. Based on the finite element calculation results and combined with mathematical methods such as multi-dimensional interpolation and inversion, the present invention realizes the establishment of a high-precision mathematical model of a dual three-phase permanent magnet synchronous motor.

[0042] 2. Compared with the existing mathematical model of the dual-three-phase permanent magnet synchronous motor, the present invention only uses the finite element calculation results of a single three-phase winding, and combines the symmetry of the dual-three-phase motor to establish the mathematical model of the dual-three-phase permanent magnet synchronous motor, reducing the computational amount in the modeling process.

[0043] 3. The modeling method proposed by the present invention can fully consider the influence of electromagnetic coupling, magnetic saturation, harmonic magnetic field and rotor position on the motor performance, and is a high-fidelity and high-precision mathematical model.

[0044] 4. When using the mathematical model established by the present invention for motor control strategy simulation, the use of the time-step finite element method is avoided, and the simulation time can be saved while ensuring the simulation accuracy. Brief Description of the Drawings

[0045] Figure 1 is the structural diagram of the dual-three-phase permanent magnet synchronous motor, where: 1-1 is the permanent magnet, 1-2 is the motor rotor, 1-3 is the motor stator, 1-4 is the first set of three-phase windings A1B1C1, and 1-5 is the second set of three-phase windings A2B2C2;

[0046] Figure 2 is the relationship between the four groups of magnetic fluxes ψ d1 、ψ q1 、Δψ d2 、Δψ q2 and the currents i d1 、i q1 obtained by finite element calculation when the electrical angle θ of the rotor position is 0°;

[0047] Figure 3 is the mathematical model of the electromagnetic part of the motor established based on the finite element analysis calculation results of the motor, combined with mathematical methods such as multi-dimensional interpolation and inversion;

[0048] Figure 4 is the mathematical model of the motion part of the motor established based on the finite element analysis calculation results of the motor, combined with mathematical methods such as multi-dimensional interpolation and inversion;

[0049] Figure 5 is the comparison of the motor drive performance obtained by simulating using the mathematical model proposed by the present invention and the traditional motor mathematical model, where: (a) is the comparison of the motor torque waveforms, (b) is the comparison of the motor speed waveforms, and (c) is the comparison of the motor dq-axis current waveforms. Detailed Embodiment

[0050] The technical solutions of the present invention will be further described below in conjunction with the drawings, but are not limited thereto. Any modification or equivalent replacement of the technical solutions of the present invention without departing from the spirit and scope of the technical solutions of the present invention shall be covered by the protection scope of the present invention.

[0051] The present invention provides a high-precision mathematical model modeling method for a dual three-phase permanent magnet synchronous motor that can consider the effects of magnetic saturation, electromagnetic coupling, harmonic magnetic fields, and rotor position. The method includes the following steps:

[0052] 1) Perform finite element modeling on the target dual three-phase permanent magnet synchronous motor. Apply multiple sets of d-axis currents i d1 and multiple sets of q-axis currents i q1 to the first set of three-phase windings A1B1C1, and apply currents with i d2 = 0 and i q2 = 0 to the second set of three-phase windings A2B2C2, and perform finite element calculations of the electromagnetic field to solve the magnetic flux and torque calculation results.

[0053] 2) Take i d1 , i q1 , and the rotor electrical angle position θ as independent variables, and take the d-axis magnetic flux ψ d1 , q-axis magnetic flux ψ q1 in the first set of three-phase windings A1B1C1 and the d-axis magnetic flux increment Δψ d2 , q-axis magnetic flux increment Δψ q2 in the second set of three-phase windings A2B2C2 obtained after calculation as dependent variables, and then establish 4 groups of relationships between magnetic flux and current ψ d1 (i d1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ), Δψ d2 (i d1 , i q1 , θ), Δψ q2 (i d1 , i q1 , θ), where ψ d1 (i d1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ) reflects the dq-axis cross-coupling situation inside the three phases, while Δψ d2 (i d1 , i q1 , θ), Δψ q2 (i d1 , i q1 , θ) reflects the electromagnetic performance coupling between the two sets of three-phase windings.

[0054] 3) Due to the symmetry and periodicity of the dual three-phase motor structure, by using ψ d1 (i d1 , i q1, θ), ψ q1 (i d1 , i q1 , θ), Δψ d2 (i d1 , i q1 , θ), Δψ q2 (i d1 , i q1 , θ) is phase-shifted by 30° to obtain ψ d2 (i d2 , i q2 , θ), ψ q2 (i d2 , i q2 , θ), Δψ d1 (i d2 , i q2 , θ), Δψ q1 (i d2 , i q2 , θ), that is:

[0055] ψ d2 (i d2 , i q2 , θ) = ψ d1 (i d1 , i q1 , θ - 30°);

[0056] ψ q2 (i d2 , i q2 , θ) = ψ q1 (i d1 , i q1 , θ - 30°);

[0057] Δψ d1 (i d2 , i q2 , θ) = Δψ d2 (i d1 , i q1 , θ - 30°);

[0058] Δψ q1 (i d2 , i q2 , θ) = Δψ q2 (i d1 , i q1 , θ - 30°).

[0059] When operating in dual three-phase, the d-axis flux linkage ψ of the first set of three-phase windings A1B1C1 d1s and the q-axis flux linkage ψ q1s , the d-axis flux linkage ψ of the second set of three-phase windings A2B2C2 d2s and the q-axis flux linkage ψ q2s are:

[0060] ψ d1s = ψ d1 (i d1 , i q1 , θ) + Δψ d1 (i d2 , i q2 , θ);

[0061] ψ q1s = ψ q1 (i d1 , i q1 , θ) + Δψ q1 (i d2 , i q2 , θ);

[0062] ψ d2s = ψ d2 (i d2 , i q2 , θ) + Δψ d2 (i d1 , i q1 , θ);

[0063] ψ q2s = ψ q2 (i d2 , i q2 , θ) + Δψ q2 (i d1 , i q1 , θ).

[0064] According to the above relationship, the mathematical modeling of the flux linkage part of the two sets of three-phase windings A1B1C1 and A2B2C2 of the entire dual three-phase motor can be realized only by applying the excitation i d1 , i q1 to the first set of three-phase windings A1B1C1.

[0065] 4) Further, by taking the inverse of ψ d1 (i d1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ), the relationships i d1 , i q1 between the current i d1 , ψ q1 and the rotor position θ can be obtained, as well as i d1 (ψ d1 , ψ q1 , θ) and i q1 (ψ d1 , ψ q1, θ). Similarly, according to the symmetry of the dual-three-phase motor structure, by shifting the phases of i d1 (ψ d1 , ψ q1 , θ) and i q1 (ψ d1 , ψ q1 , θ) by 30°, we can obtain i d2 (ψ d2 , ψ q2 , θ) and i q2 (ψ d2 , ψ q2 , θ), that is:

[0066] i d2 (ψ d2 , ψ q2 , θ) = i d1 (ψ d1 , ψ q1 , θ - 30°);

[0067] i q2 (ψ d2 , ψ q2 , θ) = i q1 (ψ d1 , ψ q1 , θ - 30°).

[0068] 5) Taking i d1 , i q1 and the rotor electrical angle position θ as independent variables, and the calculated torque as the dependent variable, we can obtain the relationship between torque, current, and rotor electrical angle position T e1 (i d1 , i q1 , θ). According to the symmetry and periodicity of the dual-three-phase motor structure, the torque generated when the second set of three-phase windings A2B2C2 operates can be obtained by shifting the phase of T e1 (i d1 , i q1 , θ) by 30°, that is:

[0069] T e2 (i d2 , i q2 , θ) = T e1 (i d1 , i q1 , θ - 30°);

[0070] where T e1 and T e2 are both torques when a single set of windings operates and can be expressed as:

[0071] T e1 = 1.5p[ψ d1 (id1 , i q1 , θ) i q1 -ψ q1 (i d1 , i q1 , θ) i d1 ;

[0072] T e2 = 1.5p[ψ d2 (i d2 , i q2 , θ) i q2 -ψ q2 (i d2 , i q2 , θ) i d2 ;

[0073] Where p is the number of pole pairs of the permanent magnet.

[0074] When the motor operates in a dual-three-phase state, its total torque is:

[0075] T s = 1.5p[ψ d1s i q1 -ψ q1s i d1+ ψ d2s i q2 -ψ q2s i d2 ;

[0076] = T e1 + T e2 + 1.5p[Δψ d1 i q1 -Δψ q1 i d1 +Δψ d2 i q2 -Δψ q2 i d2 ;

[0077] According to the torque, the electrical angular velocity ω e of the motor can be calculated:

[0078] ω e = ∫[(T s - T L ) / J] dt;

[0079] Where T L is the load torque of the motor, and J is the moment of inertia of the motor.

[0080] Furthermore, the electrical angular position θ of the motor rotor can be calculated:

[0081] θ = ∫ω edt.

[0082] 6) Since the input to the motor is voltage u d1 , u q1 , u d2 , u q2 , it is necessary to calculate the flux linkage based on these four voltage inputs, i.e.:

[0083] ψ d1s = ∫(u d1 - i d1 R + ω e ψ q1s )dt;

[0084] ψ q1s = ∫(u q1 - i q1 R - ω e ψ d1s )dt;

[0085] ψ d2s = ∫(u d2 - i d2 R + ω e ψ q2s )dt;

[0086] ψ q2s = ∫(u q2 - i q2 R - ω e ψ d2s )dt;

[0087] Among them, R is the resistance of the stator winding, and ω e is the electrical angular velocity of the motor.

[0088] 7) In summary, the input of this mathematical model is voltage u d1 , u q1 , u d2 , u q2 . After step 6), four flux linkages ψ d1s , ψ q1s , ψ d2s , ψ q2s corresponding to two sets of three-phase windings are obtained. Then, each flux linkage is disassembled into two parts through step 3). On this basis, four currents i d1 , i q1 , i d2 , i q2 are solved through step 4). Furthermore, the torque is solved based on step 5). Combining with the motion equation of the motor, the electrical angular velocity ω e and the electrical angular position θ are solved. Thus, the establishment of the high-precision double three-phase permanent magnet synchronous motor mathematical model proposed by the present invention is realized.

[0089] Embodiment 1:

[0090] Figure 1 It is a finite element model of a 22 - pole 24 - slot double - layer winding dual - three - phase permanent magnet synchronous motor. The rated speed of the motor is 1800 r / min, the rated frequency is 330 Hz, and the control method of id = 0 is adopted for control. For Figure 1 the first set of three - phase windings A1B1C1 shown, apply multiple sets of d - axis current i d1 and multiple sets of q - axis current i q1 , with the range from - 20 A to 20 A. Apply i d2 = 0, i q2 = 0 current to the second set of three - phase windings A2B2C2, and conduct finite - element calculation of the electromagnetic field. Solve to obtain four sets of flux - linkage relationships ψ Figure 2 shown as ψ d1 (i d1 ,i q1 ,θ), ψ q1 (i d1 ,i q1 ,θ), Δψ d2 (i d1 ,i q1 ,θ), Δψ q2 (i d1 ,i q1 ,θ) and torque relationship T e1 (i d1 ,i q1 ,θ). Due to the symmetry of the two sets of three - phase windings of the dual - three - phase motor, perform phase - shift and transposition on ψ d1 (i d1 ,i q1 ,θ), ψ q1 (i d1 ,i q1 ,θ), Δψ d2 (i d1 ,i q1 ,θ), Δψ q2 (i d1 ,i q1 ,θ) and T e1 (i d1 ,i q1 ,θ), and then ψ d2 (i d2 ,i q2 ,θ), ψ q2 (i d2 ,i q2 ,θ), Δψ d1 (i d2 ,i q2 ,θ), Δψ q1 (i d2 ,iq2 , θ) and T e2 (i d1 , i q1 , θ). Based on these relationships, combined with the voltage equation and motion equation of the motor, the establishment of the motor mathematical model as shown in Figure 3 can be carried out. Using the motor mathematical model as shown in Figure 3 and 4 to calculate the dynamic characteristics of the motor with the traditional linear motor mathematical model, the obtained calculation results are as shown in Figure 5 . Since the mathematical model proposed in the present invention takes into account the effects of electromagnetic coupling, harmonic magnetic field and rotor position, the calculated dq-axis currents show regular fluctuations, and due to the symmetry of the two sets of windings, i d1 and i d2 , i q1 and i q2 have symmetric fluctuations with respect to each other, while the traditional linear model has no current fluctuations, and the waveforms of i d1 and i d2 , i q1 and i q2 completely coincide. Due to considering the influence of core saturation, the torque calculated by the mathematical model proposed in the present invention is slightly less than that of the linear model, and the time required for the speed to reach stability is slightly longer. These indicate that the mathematical model proposed in the present invention takes into account the effects of electromagnetic coupling, harmonic magnetic field, core saturation and rotor position of the motor, and has high precision and high fidelity.

Claims

1. A modeling method for a high-precision mathematical model of a dual-three-phase permanent magnet synchronous motor, characterized in that The method includes the following steps: Step 1: Perform finite element modeling on the target dual three-phase permanent magnet synchronous motor, apply multiple sets of d-axis currents i d1 and multiple sets of q-axis currents i q1 to the first set of three-phase windings A1B1C1, apply currents with i d2 = 0 and i q2 = 0 to the second set of three-phase windings A2B2C2, conduct finite element calculation of the electromagnetic field, and solve the calculation results of magnetic flux and torque; Step 2: Taking i d1 , i q1 , and the rotor electrical angle position θ as independent variables, and taking the d-axis flux linkage ψ d1 , q-axis flux linkage ψ q1 in the first set of three-phase windings A1B1C1 and the d-axis flux linkage increment Δψ d2 , q-axis flux linkage increment Δψ q2 in the second set of three-phase windings A2B2C2 as dependent variables, establish 4 groups of relationships between flux linkage and current ψ d1 (i d1 , i q1 , θ), ψ q1 (i d1 , i q1 , θ), Δψ d2 (i d1 , i q1 , θ), Δψ q2 (i d1 , i q1 , θ); Step 3: According to the symmetry and periodicity of the dual three-phase motor structure, shift the phase of ψ d1 (i d1 ,i q1 ,θ), ψ q1 (i d1 ,i q1 ,θ), Δψ d2 (i d1 ,i q1 ,θ), Δψ q2 (i d1 ,i q1 ,θ) by 30°, and obtain ψ d2 (i d2 ,i q2 ,θ), ψ q2 (i d2 ,i q2 ,θ), Δψ d1 (i d2 ,i q2 ,θ), Δψ q1 (i d2 ,i q2 ,θ), that is: ψ d2 (i d2 ,i q2 ,θ) = ψ d1 (i d1 ,i q1 ,θ - 30°); ψ q2 (i d2 ,i q2 ,θ) = ψ q1 (i d1 ,i q1 ,θ - 30°); Δψ d1 (i d2 ,i q2 ,θ) = Δψ d2 (i d1 ,i q1 ,θ - 30°); Δψ q1 (i d2 ,i q2 ,θ) = Δψ q2 (i d1 ,i q1 ,θ - 30°); During double three-phase operation, the d-axis flux linkage ψ of the first set of three-phase windings A1B1C1 d1s and the q-axis flux linkage ψ q1s , the d-axis flux linkage ψ of the second set of three-phase windings A2B2C2 d2s and the q-axis flux linkage ψ q2s are as follows: ψ d1s = ψ d1 (i d1 , i q1 , θ) + Δψ d1 (i d2 , i q2 , θ); ψ q1s = ψ q1 (i d1 , i q1 , θ) + Δψ q1 (i d2 , i q2 , θ); ψ d2s = ψ d2 (i d2 , i q2 , θ) + Δψ d2 (i d1 , i q1 , θ); ψ q2s = ψ q2 (i d2 , i q2 , θ) + Δψ q2 (i d1 , i q1 , θ); Step 4. Invert ψ d1 (i d1 ,i q1 ,θ), ψ q1 (i d1 ,i q1 ,θ) to obtain the relationships between the currents i d1 、i q1 and the fluxes ψ d1 、ψ q1 and the rotor electrical angle position θ, i d1 (ψ d1 ,ψ q1 ,θ) and i q1 (ψ d1 ,ψ q1 ,θ); According to the symmetry and periodicity of the dual-three-phase motor structure, shift i d1 (ψ d1 ,ψ q1 ,θ) and i q1 (ψ d1 ,ψ q1 ,θ) by 30°, to obtain i d2 (ψ d2 ,ψ q2 ,θ) and i q2 (ψ d2 ,ψ q2 ,θ), that is: i d2 (ψ d2 ,ψ q2 ,θ) = i d1 (ψ d1 ,ψ q1 ,θ - 30°); i q2 (ψ d2 ,ψ q2 ,θ) = i q1 (ψ d1 ,ψ q1 ,θ - 30°); Step Five: Taking i d1 , i q1 and the rotor electrical angle position θ as independent variables, and taking the calculated torque as the dependent variable, the relationship T e1 (i d1 , i q1 , θ) between the torque and the current and the rotor electrical angle position is obtained. According to the symmetry and periodicity of the dual three-phase motor structure, the torque generated when the second set of three-phase windings A2B2C2 operates is obtained by shifting T e1 (i d1 , i q1 , θ) by 30°, that is: T e2 (i d2 ,i q2 ,θ) = T e1 (i d1 ,i q1 ,θ - 30°); Among them, T e1 and T e2 are both torques during single set of winding operation; When the motor operates in a dual-three-phase state, its total torque is: T s = 1.5p[ψd 1s i q1 -ψ q1s id 1+ ψd 2s i q2 -ψ q2s id2] = T e1 + T e2 + 1.5p[Δψd1i q1 - Δψ q1 id1 + Δψd2i q2 - Δψ q2 id2]; where p is the number of pole pairs of the permanent magnet; Calculate the electrical angular velocity ω of the motor based on the torque e : ω e = ∫[(T s - T L ) / J] dt; where T L is the load torque of the motor, and J is the moment of inertia of the motor; According to the electrical angular velocity ω of the motor e , calculate the electrical angular position θ of the rotor: θ = ∫ω e dt; Step 6. Calculate the magnetic flux based on the input voltages u d1 , u q1 , u d2 , u q2 : ψ d1s = ∫(u d1 - i d1 R + ω e ψ q1s ) dt; ψ q1s = ∫(u q1 - i q1 R - ω e ψ d1s ) dt; ψ d2s = ∫(u d2 - i d2 R + ω e ψ q2s )dt; ψ q2s = ∫(u q2 - i q2 R - ω e ψ d2s )dt; where R is the resistance of the stator winding and ω e is the electrical angular velocity of the motor.

2. The high-precision mathematical model modeling method for a dual-three-phase permanent magnet synchronous motor according to claim 1, wherein The said T e1 and T e2 is expressed as: T e1 = 1.5p[ψ d1 (i d1 ,i q1 ,θ)i q1 -ψ q1 (i d1 ,i q1 ,θ)i d1 ; T e2 = 1.5p[ψ d2 (i d2 ,i q2 ,θ)i q2 -ψ q2 (i d2 ,i q2 ,θ)i d2 ; where p is the number of pole pairs of the permanent magnet.

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