Antenna array for high-frequency devices

Abstract

An antenna array for a high-frequency device, comprising a plurality of antenna elements (11) for a radar device and arranged in a two-dimensional array in a predetermined area. The plurality of antenna elements include grouped conducting elements (11a) and individual conducting elements (11c), having a specific distance for eliminating grating lobes, and each conducting element is electrically connected to a phase shifter (14). The conducting elements are arranged such that the density of the conducting elements at the central part in the two-dimensional array is high, and the density of the conducting elements at the four corners in the two-dimensional array is low.

Description

[0001] Cross - reference to related applications

[0002] This application is based on Japanese Patent Application No. 2021 - 100206 filed on Jun. 16, 2021, the disclosure of which is incorporated herein by reference. Technical field

[0003] The present disclosure relates to an antenna array for a high - frequency device. Background art

[0004] The technology of phased - array antennas for high - frequency devices has been developed (see, for example, US 2017 / 207547 A1). According to the phased - array antenna described in US2017 / 207547 A1, each array element is two - dimensionally arranged, and each array element is arranged, for example, in units of eight. Each array element is grouped into 4×2 and 8×1 square sub - arrays. A plurality of square sub - arrays are tiled in order to break the periodicity of the phase center, thereby reducing grating lobes. Summary of the invention

[0005] In the technology described in US 2017 / 207547 A1, the periodicity of the phase center is broken in order to suppress grating lobes. However, conversely, since the periodicity of the phase center is reduced, all phase centers are irregularly shifted from the element coordinates, which complicates the calculation of phase values and the calculation of tapering.

[0006] That is, when the off - grid increases in the vertical and horizontal directions at the phase - center position, the distance between adjacent elements changes from the ideal distance of 0.5λ, and there is no premise, which complicates the calculation of phase values. The inventors also found that by vertically or horizontally grouping each adjacent array element, the number of phase shifters can be reduced and the system can be simplified, and grating lobes are generated during scanning in the vertical or horizontal direction in the same direction as the grouping. On the other hand, for example, in a scanning radar sensor, in order to reduce costs and simplify the system, it is particularly necessary to reduce the number of phase shifters of the conductive elements electrically connected to the phased array.

[0007] In view of the foregoing difficulties, an object of the present disclosure is to provide an antenna array for a high - frequency device that can suppress the generation of grating lobes (and side lobes, etc.) while reducing the number of phase shifters.

[0008] According to one aspect of the present disclosure, an antenna array for a high - frequency device includes a plurality of antenna elements for a radar device and two - dimensionally arranged in a predetermined area. The plurality of antenna elements include conductive elements electrically connected to phase shifters. The conductive elements are arranged such that the density of the conductive elements at the central portion in the two - dimensional array is high, and the density of the conductive elements at the four corners in the two - dimensional array is low.

[0009] With the above-described antenna array, since the number of conductive elements can be reduced, the number of phase shifters electrically connected to the conductive elements can also be reduced. In addition, since the density of the conductive elements is high at the center and low at the four corners, unnecessary side lobes and the like can be suppressed from being generated. Description of the Drawings

[0010] The above and other objects, features, and advantages of the present invention will become more apparent from the following detailed description made with reference to the accompanying drawings, in which:

[0011] Figure 1 is a diagram for explaining real beams and virtual beams in a hybrid radar according to the first embodiment.

[0012] Figure 2 is an electrical configuration diagram showing a hybrid radar device according to the first embodiment.

[0013] Figure 3 is an electrical configuration diagram showing a plurality of receiver antenna arrays connected to phase shifters and downconverters according to the first embodiment.

[0014] Figure 4 is a diagram for explaining real beams and virtual beams according to the first embodiment.

[0015] Figure 5 is a layout diagram schematically showing a receiver antenna array having a phase shifter IC and a transceiver IC having a plurality of mixers in a hybrid radar structure according to the first embodiment.

[0016] Figure 6 is a connection diagram schematically showing the arrangement of conductive elements in an antenna array and the connection of phase shifters and downconverters according to the first embodiment.

[0017] Figure 7 is a dimensional diagram showing the arrangement of conductive elements according to the first embodiment.

[0018] Figure 8 is an explanatory diagram showing a theoretical calculation graph of the grating lobe angle and the scan angle with different d / λ values plotted simultaneously with the simulated grating lobe angle at several scan angles in an ideal uniform rectangular array (URA) having vertical grouping of adjacent elements.

[0019] Figure 9 is a simulated beam pattern of two types of null filters according to the first embodiment, the null filters being turned by 17.5° in the E-plane to show nulls at the same angle as the grating lobe, the simulated beam pattern being plotted simultaneously with the simulated beam pattern of the URA having vertical grouping of adjacent elements turned by 17.5° in the E-plane to show the grating lobe as a reference.

[0020] Figure 10 The simulated beam patterns of TX and RX (in this case TX = RX) of the antenna array according to the first embodiment, and the combined beam pattern of TX and RX steered at 17.5° in the E-plane, where the main lobe peak is normalized to 0 dB.

[0021] Figure 11 The simulated Rx beam pattern of the antenna array having three types of null filters in the antenna array 7, which is plotted simultaneously with the URA with adjacent elements vertically grouped according to the first embodiment when the main beam angle is steered at 5° in the E-plane.

[0022] Figure 12 The figure schematically shows the simulated RX beam pattern of the antenna array according to the first embodiment when the main beam angle is steered at 17.5° in the E-plane, which is plotted simultaneously with the URA with adjacent elements vertically grouped.

[0023] Figure 13 The figure shows the transition of the grating lobe generation angle when the main beam angle changes from 0° to 40° in the E-plane, and the simulation results of the null angles at several scan angles for two types of null filters used in the antenna array according to the first embodiment.

[0024] Figure 14 The figure schematically shows the arrangement of the conducting elements for the antenna array according to the second embodiment.

[0025] Figure 15 The figure schematically shows the arrangement of the conducting elements for the antenna array according to the third embodiment.

[0026] Figure 16 The figure schematically shows a partial arrangement of the conducting elements for the antenna array according to the third embodiment. Detailed Description of the Embodiment

[0027] Hereinafter, some embodiments of the antenna array for the high-frequency device for the radar device 1 will be described with reference to the drawings. In each of the following embodiments, the same or similar reference numerals are given to the same or similar configurations, and their descriptions are omitted as needed.

[0028] (First Embodiment)

[0029] The first embodiment will be described with reference to Figures 1 to 13 The radar device 1 is attached to the front end of the vehicle 40 as shown in Figure 1 and is used for long-range radar (LRR) applications to scan a predetermined range of several hundred meters in front of the vehicle. The radar device 1 can be attached to multiple positions on the front, rear, left, and right of the vehicle 40.

[0030] Figure 2The illustrated vehicle radar device 1 mainly includes a transceiver integrated circuit IC1 and a phase shifter integrated circuit IC2. The radar device 1 calculates the distance to a target, the presence angle, etc. by synthesizing signals of receiver (RX) channels. The number of Rx channels is 4. In the following example, the number of transmitter (TX) channels is 1, the number of RX channels is 4, and the RX channels are coded as Rx1, Rx2, Rx3, and Rx4. However, the number n of RX channels can be any number greater than 2.

[0031] The phase shifter integrated circuit IC2 includes RX phase shift units 10 for each of the RX channels Rx1 to Rx4. An antenna array 7 for a high-frequency device (hereinafter abbreviated as antenna array 7) is connected to each of the RX phase shift units 10. As Figure 3 shown, the antenna array 7 functions as a phased array antenna. The antenna array 7 is configured by combining the conductive elements 11a and 11c electrically connected to the phase shifter integrated circuit IC2 and the non-conductive elements 11b and virtual elements 11d not electrically connected to the phase shifter integrated circuit IC2. Details will be described later.

[0032] As Figure 2 shown, the RX phase shift unit 10 is connected to the IC pad 20. The conductive elements 11a and 11c constituting the antenna array 7 are connected to the corresponding IC pads 20 via PCB wiring. In addition, the RX phase shift unit 10 includes a variable gain amplifier 13, a phase shifter 14, and an amplifier 15 as a high-frequency unit 12.

[0033] In the RX phase shifter IC10, when a signal is received from the antenna array 7 through the IC pad 20, the variable gain amplifier 13 amplifies the signal received from the antenna array 7, and the phase shifter 14 shifts the phase of the amplified signal of the variable gain amplifier 13 by a phase shift value The amplifier 15 amplifies the phase-shifted signal of the phase shifter 14 and outputs the signal to the mixer 9. By configuring the variable gain amplifier 13 between the antenna array 7 and the phase shifter 14, the trade-off between NF and the distortion performance on the system of the radar device 1 can be improved according to the application. For example, a high-gain setting (minimum NF) improves the detection ability for long-distance targets, and a low-gain setting can reduce saturation when detecting short-distance targets.

[0034] In a configuration example that more specifically shows Figure 2 the connection Figure 3 the RX phase shift units 10 of the RX channels Rx1 to Rx4 process the signals received from the antenna array 7, and then synthesize the signals through nodes N1 to N5 and output them to the mixer 9. The node N1 synthesizes the received signals received from the two conductive elements 11a. The node N2 synthesizes the received signals received from the two conductive elements 11a and 11c.

[0035] Figure 3 The node N3 therein synthesizes the received signals received from the two conducting elements 11a and 11c. The node N4 synthesizes the received signals received from the two conducting elements 11a. At the node N5, the signals obtained through the nodes N1 to N4 are combined and output to the mixer 9. The line lengths from the conducting elements 11a and 11c to the mixer 9 can be configured as paths having equal lengths to each other.

[0036] On the other hand, as Figure 2 shown, the transceiver integrated circuit IC 1 is configured to control blocks in the control unit 2, the signal processing unit 3, the PLL 4, the TX unit 5, and the RX unit 6. The control unit 2 of the transceiver integrated circuit IC 1 performs various control functions such as the output frequency control unit 2a, the amplitude control unit 2b, and the phase control unit 2c by executing a predetermined control logic. The output frequency control unit 2a controls the output frequency of the PLL 4. The phase control unit 2c controls the phase shift value of the phase shifter 14 in the phase shifter integrated circuit IC2 The amplitude control unit 2b controls the amplitude of the variable gain amplifier 13 in the phase shifter integrated circuit IC2. The control unit 2 controls the phase shift value of the phase shifter 14 of each RX channel Rx1 to Rx4 by using the phase control unit 2c to control the RX beam scan angle of the RX channels Rx1 to Rx4.

[0037] The RX unit 6 includes an LO amplifier 8 and a mixer 9, and is connected to the RX phase shift unit 10 of the phase shifter integrated circuit IC2. The PLL 4 uses a reference clock CLK input from a reference oscillation circuit (not shown), and outputs a local signal (having, for example, 77 GHz) in the millimeter wave band having the same frequency to the mixers 9 in all the RX channels RX1 to RX4 by adjusting parameters such as multiples of the reference clock CLK. The mixer 9 can obtain an IF output having a frequency proportional to the distance by mixing the local signal and the signal received by reflecting the radio wave output from the TX unit 5 on the target. Although not described here, a multiplier can be provided to multiply the frequency to a desired frequency, and then the local signal can be output to each of the RX channels Rx1 to Rx4.

[0038] The LO amplifier 8 amplifies the local signal of the PLL 4 with a predetermined amplitude and outputs it to the mixers 9 in each of the RX channels Rx1 to Rx4. The mixers 9 of each of the RX channels Rx1 to Rx4 input and mix the output signals of the RX phase shift units 10 of each of the RX channels Rx1 to Rx4 and the local signal amplified by the LO amplifier 8 as the IF signals IF1 to IF4.

[0039] Since the same PLL 4 supplies the local signal to the mixers 9 of all the RX channels Rx1 to Rx4, the frequency change of the IF signal and the reference clock CLK and the change in frequency characteristics with respect to changes in the external environment are highly correlated.

[0040] In addition, the mixers 9 of each of the RX channels Rx1 to Rx4 output the output signals of each mixer 9 to the signal processing unit 3. The signal processing unit 3 includes a processor and predetermined electronic control logic, and can estimate the angle of a target present in a sector where the field of view is narrowed, such as by signal processing such as digital beamforming (DBF).

[0041] The signal processing unit 3 inputs the IF signal processed by the mixer 9 to the A / D converter 3a through an IF filter (not shown). The A / D converter 3a converts the IF signal into digital data through analog-to-digital conversion processing. The signal processing unit 3 performs predetermined digital signal processing through the FFT 3b, and as Figure 1 shown, measures the distance from the target vehicle 40 to another vehicle 41, the relative speed with respect to the vehicle 41, and the presence angle of the vehicle 41.

[0042] The signal processing unit 3 narrows the field of view to the Figure 1 shown sector Sb by using analog beamforming with the phase shifter 14. By performing signal processing through the DBF algorithm, the signal processing unit 3 forms a narrow virtual beam Sc in the sector Sb, as Figure 4 shown, and identifies the vehicle 41 as a scanning target with higher resolution. As a result, another vehicle 42 can be excluded from the scanning targets. In addition, multi-signal classification processing (multiple signal classification, i.e., MUSIC), etc. can be applied to multiple targets, which can obtain a higher resolution than the above-mentioned DBF.

[0043] For example, as Figure 1 and Figure 4 shown, the signal processing unit 3 uses the DBF algorithm to narrow the field of view to the sector Sb instead of the entire wide-angle field of view Sa, and obtains a virtual beam Sc for each sector Sb. Therefore, the vehicle 41 as a target can be identified with high resolution in the narrow sector Sb. Since the field of view can be narrowed to the sector Sb, the computational amount can be reduced compared with a conventional MIMO radar. Therefore, the hybrid method is an effective scanning method that simplifies the trade-off between shortening the scanning time and high-resolution capability.

[0044] Hereinafter, the structure of the antenna array 7 used in this radar device 1 will be described. Since the structures of the antenna arrays 7 for the TX unit 5 and the RX unit 6 are the same, the antenna array 7 connected to the RX unit 6 will be described below.

[0045] As Figure 5and Figure 6 As shown, the antenna array 7 of each RX channel Rx1 to RxX4 is configured by arranging elements 11a to 11d made of a metallic rectangular surface in regions separated by a grid pattern. The outer frame of the antenna array 7 is formed in a rectangular shape, and the rectangular elements 11a to 11d are arranged in regions of the grid-shaped vertices in the outer frame of the antenna array 7. In this embodiment, as Figure 5 or Figure 6 shown, the active elements are arranged in a two-dimensional array region divided into 16 rows and 12 columns.

[0046] In this embodiment, as Figure 5 or Figure 6 shown, in each antenna array 7, 18 elements 11a to 11d are arranged side by side along the long side in the Y direction in the grid-separated region, and 14 elements 11a to 11d are arranged side by side along the short side in the X direction in the grid-separated region. In addition, the distance between adjacent elements 11a to 11d is set to half of the radar wavelength λ, and the shape of each of the elements 11a to 11d is a rectangular shape. The antenna array 7 is arranged in the XY plane and emits a beam along the +Z axis direction orthogonal to the XY plane. As Figure 5 shown, the antenna array 7 having a 16×12 basic array is continuously arranged in the X-axis direction so as to be connected to four RX channels Rx1 to Rx4. In other words, in a hybrid system, for example, a 16×48 antenna array is divided into 16×12 antenna arrays 7, and N RX mixers are used to perform IF signal processing for N. In this embodiment, an example of N = 4 is described.

[0047] As described above, the antenna array 7 includes conductive elements 11a and 11c, non-conductive elements 11b, and dummy elements 11d. The conductive element 11a is the element among a pair of conductive elements 11a adjacent to each other in the Y direction that is electrically connected to the phase shifter integrated circuit IC2. The conductive element 11c is the element among a pair of conductive elements 11c separated from each other in the Y direction that is electrically connected to the phase shifter integrated circuit IC2. Therefore, the conductive elements 11a and 11c are shown with different reference numerals. Figure 5 and Figure 6 The filled regions in show the conductive element 11a, and the conductive element 11c is shown with hatched lines. The non-conductive element 11b is shown by a solid line frame, and the dummy element 11d is shown by a dashed line frame.

[0048] The dummy element 11d is arranged on the outermost periphery of the two-dimensional array of the antenna array 7. The dummy element 11d is not connected to the RX phase shifter unit 10 like the non-conductive element 11b. Since the dummy element 11d is arranged on the outermost periphery of the two-dimensional array, the quality of the TX and RX signals using the antenna array 7 can be improved.

[0049] In the configuration of this embodiment, if the conduction elements 11a are arranged at the vertices of the internal grid except for the virtual element 11d of the outermost frame, a total of 16×12 = 192 conduction elements 11a can be arranged. However, since phase shift control is complex when all the conduction elements 11a are controlled by the phase shifter integrated circuit IC2, it is not preferable to arrange the conduction elements 11a at all the vertices of the grid. Therefore, in this embodiment, by designing the two-dimensional arrangement of the conduction elements 11a and 11c and the disconnection elements 11b, the number of conduction elements 11a and 11c to be phase shift controlled is reduced, and the phase shift control is further simplified.

[0050] In the following description, as Figure 6 shown, the rows of each antenna array 7 are referred to as rows X1 to X12. In addition, both ends of the Y row in which the virtual element 11d is arranged are referred to as row Yd1 and Yd2, and the rows between them are referred to as rows Y1 to Y16. When showing the arrangement area of the elements 11a to 11d, it is represented by the symbol of coordinates (X, Y). In addition, for example, when the conduction elements 11a in row Y3 and the conduction elements 11a in row Y4 are electrically connected and grouped, the grouping is represented by a minus sign as in "Y3 - Y4".

[0051] As Figure 6 shown, a large number of IC pads 20 and a pair of conduction elements 11a and a pair of conduction elements 11c of the antenna array 7 are connected through the TX line 21 using a printed circuit board, whereby signals from the conduction elements 11a and 11c can be received. An example of the arrangement of the elements 11a to 11d will be described with reference to Figure 6 the following.

[0052] As Figure 6 shown, in the antenna array 7, the center of the row is between rows Y8 and Y9, and the center of the column is between columns X6 and X7. The conduction elements 11a are symmetrically arranged in the vertical direction with respect to the center of the row and symmetrically arranged in the horizontal direction with respect to the center of the column. In addition, the conduction elements 11a are arranged symmetrically with respect to the center point of the antenna array 7.

[0053] Specifically, in the antenna array 7, Figure 6 the conduction elements 11a in the left half region shown in

[0054] are symmetrically arranged in the vertical direction at:

[0055] coordinates (X1, Y3 - Y4) and coordinates (X1, Y13 - Y14),

[0056] coordinates (X2, Y2 - Y3) and coordinates (X2, Y14 - Y15),

[0057] Coordinates (X2, Y8 - Y9),

[0058] coordinates (X3, Y4 - Y5) and coordinates (X3, Y12 - Y13), and

[0059] coordinates (X3, Y7 - Y8) and coordinates (X3, Y9 - Y10).

[0060] In addition, the conducting elements 11a are symmetrically arranged in the vertical direction at:

[0061] coordinates (X4, Y1 - Y2) and coordinates (X4, Y15 - Y16),

[0062] coordinates (X4, Y3 - Y4) and coordinates (X4, Y13 - Y14),

[0063] coordinates (X4, Y6 - Y7) and coordinates (X4, Y10 - Y11),

[0064] coordinates (X5, Y4 - Y5) and coordinates (X5, Y12 - Y13),

[0065] coordinates (X5, Y6 - Y7) and coordinates (X5, Y10 - Y11),

[0066] coordinates (X5, Y8 - Y9),

[0067] coordinates (X6, Y2 - Y3) and coordinates (X6, Y14 - Y15), and

[0068] coordinates (X6, Y7 - Y8) and coordinates (X6, Y9 - Y10).

[0069] In addition, in the antenna array 7, Figure 6 the conducting elements 11a in the right - hand half region shown are symmetrically arranged in the vertical direction at:

[0070] coordinates (X12, Y3 - Y4) and coordinates (X12, Y13 - Y14),

[0071] coordinates (X11, Y2 - Y3) and coordinates (X11, Y14 - Y15),

[0072] coordinates (X11, Y5 - Y6) and coordinates (X11, Y11 - Y12),

[0073] coordinates (X11, Y8 - Y9),

[0074] coordinates (X10, Y4 - Y5) and coordinates (X10, Y12 - Y13),

[0075] coordinates (X10, Y7 - Y8) and coordinates (X10, Y9 - Y10).

[0076] In addition, the conduction elements 11a are symmetrically arranged in the vertical direction at:

[0077] coordinates (X9, Y1 - Y2) and coordinates (X9, Y15 - Y16),

[0078] coordinates (X9, Y3 - Y4) and coordinates (X9, Y13 - Y14),

[0079] coordinates (X9, Y6 - Y7) and coordinates (X9, Y10 - Y11),

[0080] coordinates (X8, Y4 - Y5) and coordinates (X8, Y12 - Y13),

[0081] coordinates (X8, Y6 - Y7) and coordinates (X8, Y10 - Y11),

[0082] coordinates (X8, Y8 - Y9),

[0083] coordinates (X7, Y2 - Y3) and coordinates (X7, Y14 - Y15), and

[0084] coordinates (X7, Y7 - Y8) and coordinates (X7, Y9 - Y10).

[0085] The grouping direction of the conduction elements 11a is the Y direction and not the X direction. Therefore, the conduction elements 11a can be arranged without generating grating lobes in the X direction, which would tend to occur if grouping were performed because the horizontal conduction elements are placed at an ideal half - λ spacing without any grouping.

[0086] The connection center part connecting a pair of conduction elements 11a is defined as the Rx feed point of the transmission line 21, and the phase center of the grouped pair of conduction elements 11a is located at the connection center part. The transmission line 21 is configured by using the wiring arranged on the printed circuit board. The line lengths of the transmission line 21 connecting the IC pad 20 and each pair of conduction elements 11a can be equal to each other or have a relationship of p×λ (where p is an integer) with each other in order to align the phases of all channels. This configuration makes it easy to design the arrangement of the conduction elements 11a in the antenna array 7.

[0087] In addition, as Figure 6 shown, the conduction elements 11c are also arranged in the antenna array 7, but each conduction element 11c is sandwiched between the disconnect elements 11b in the Y direction, thus being separated from each other. Two single conduction elements 11c are arranged separately from each other in the same direction as the Y direction in which the conduction elements 11a are grouped, and are symmetrically arranged with respect to the center line of the row.

[0088] Conduction elements 11c are arranged in pairs at coordinates (X6, Y5) and coordinates (X6, Y12). Conduction elements 11c are arranged in pairs at coordinates (X7, Y5) and coordinates (X7, Y12). The distance between the centers of the conduction elements 11c in row Y5 and the centers of the conduction elements 11c in row Y12 is 3.5λ. These conduction elements 11c separated by 3.5λ are used as a first null filter (steerable null filter).

[0089] Conduction elements 11c are arranged in pairs at coordinates (X5, Y1) and coordinates (X5, Y16). Similarly, conduction elements 11c are arranged in pairs at coordinates (X8, Y1) and coordinates (X8, Y16). The distance between the centers of the conduction elements 11c in row Y1 and the centers of the conduction elements 11c in row Y16 is 7.5λ. These conduction elements 11c separated by 7.5λ are used as a second null filter.

[0090] Conduction elements 11c are arranged in pairs at coordinates (X3, Y2) and coordinates (X3, Y15). Similarly, conduction elements 11c are arranged in pairs at coordinates (X10, Y2) and coordinates (X10, Y15). Since the row-to-row distance or column-to-column distance between adjacent elements 11a to 11d is 0.5λ, the distance between the centers of the conduction elements 11c in row Y3 and the centers of the conduction elements 11c in row Y15 is 6.5λ. These conduction elements 11c separated by 6.5λ are used as a third null filter.

[0091] As described above, the conduction elements 11c are arranged as separate individual elements from each other, and the distance between the centers of the conduction elements 11c is set to a specific distance (0.5 + m)λ (where m is an integer). That is, the conduction elements 11c are arranged symmetrically along a line at intervals of (0.5 + m)λ (m = 1, 2,...) from the center of the rows of the antenna array 7. In addition, in order to increase the density of the conduction elements 11a and 11c in the central portion, it is desirable to set m to be equal to or greater than 3. By setting m to be equal to or greater than 3, the effective elements in the central portion of the antenna array 7 can be made dense to obtain better sidelobe performance, and the measurement of the sidelobes can be performed. Note, Figure 6 Examples of m = 3, 6, and 7 are shown. In addition, in order to change the characteristics of the null filter, it is desirable to provide multiple sets of conduction elements 11c in the antenna array 7 that satisfy different values of m. Since the grating lobe also has an angular width, the grating lobe with an angular width can be suppressed by superimposing null filters with different damping characteristics near the angle where the grating lobe is generated.

[0092] Figure 7 The element arrangement of the extracted columns X6 and X7 is shown. The conduction elements 11a in rows Y7 - Y8 adjacent to each other in the Y direction are controlled so that the phase shift value achieved by the phase shifter 14 The same. Therefore, the phase center of the conducting elements 11a in rows Y7 - Y8 is at the mid - position between rows Y7 - Y8. Since the same signal is given to the conducting elements 11a in rows Y9 - Y10, the phase center of the conducting elements 11a in rows Y9 - Y10 is at the mid - position between rows Y9 - Y10.

[0093] Since the distance between the elements in rows Y7 - Y8 and rows Y9 - Y10 is λ / 2, the phase - center distance d between the conducting elements 11a in rows Y7 - Y8 and rows Y9 - Y10 is λ, which is twice of λ / 2. Figure 8 The theoretical calculations of the grating - lobe angle versus the scan angle at different d / λ values are plotted, and are plotted simultaneously with the simulated grating - lobe angles at several scan angles in an ideal URA (Uniform Rectangular Array) with adjacent elements vertically grouped. As Figure 8 shown, the relationship of the grating - lobe generation angle between the phase - center spacing d and the radar wavelength λ is the same as that in the case of a d = 1λ design.

[0094] As a result, it is difficult in principle to strongly generate grating lobes. This is a phenomenon caused by grouping adjacent elements in the Y - direction to reduce the number of phase shifters 14. However, as described above, by arranging a single element 11c as Figure 7 shown, the phase - center distance d can be formed to be approximately 1.25λ, and the λ - periodicity of the phase - center spacing d can be broken. As a result, the grating lobes can be reduced by several dB. In addition, the conducting element 11c has an attenuation characteristic as a steerable null filter and can follow and suppress the grating lobes.

[0095] In other words, a pair of adjacent conducting elements 11a are arranged symmetrically in the Y - direction in the antenna array 7, and a single conducting element 11c is arranged by being sandwiched between the disconnecting elements 11b on both sides in the Y - direction. Therefore, even if the adjacent conducting elements 11a are grouped into a pair, the periodicity of the phase center can be broken.

[0096] In addition, the arrangement position of the single conducting element 11c will be further explained and described. The conducting element 11c is arranged symmetrically at the vertices of a two - dimensional quadrilateral with respect to the center point of the antenna array 7. The vertices of the quadrilateral represent, for example:

[0097] a pair of coordinates (X6, Y5) and coordinates (X6, Y12), and a pair of coordinates (X7, Y5) and coordinates (X7, Y12);

[0098] a pair of coordinates (X5, Y1) and coordinates (X5, Y16), and a pair of coordinates (X8, Y1) and coordinates (X8, Y16); and

[0099] A pair of coordinates (X3, Y2) and coordinate (X3, Y15), and a pair of coordinates (X10, Y2) and coordinate (X10, Y15).

[0100] By adopting such an arrangement, symmetry with respect to the X direction and the Y direction can be maintained. In addition, since the single conduction element 11c is arranged separately from a pair of conduction elements 11a adjacent to each other along the Y direction, the uniformity of the phase center interval can be reduced when the conduction elements 11a are grouped. In this way, grating lobes can be suppressed, and the sidelobe level can be followed and suppressed.

[0101] In addition, the conduction elements 11a and 11c in the left half region and the conduction elements 11a and 11c in the right half region are symmetrically arranged. In the configuration of the present embodiment, in the arrangement of the conduction elements 11a and the disconnection elements 11b, the density occupied by the conduction elements 11a and 11c in the central part of the antenna array 7 increases, and the density at its four corners decreases. If it is determined that N = the number, the reference for the central part of the occupancy density is (N - 2) / 3 + 2 = (16 - 2) / 3 + 2 ≈ 6 elements, and the reference for the four corners of the occupancy density is (N - 2) / 3 = (16 - 2) / 3 ≈ 4 elements.

[0102] Specifically, in the 6×6 square region in the central part of the antenna array 7, the occupancy density of the conduction elements 11a and 11c in the 3×3 square region is between 7 / 9 and 9 / 9, that is, greater than 75%. On the other hand, the occupancy density of the conduction elements 11a and 11c at the four corners of the antenna array 7 is 4 / 9, that is, approximately 44%. The rectangular antenna array 7 has an occupancy density of 5 / 9, that is, the conduction elements 11a and 11c at the centers of both ends of the four sides are approximately 56%. From the perspective of tapering, the design of eliminating the conduction elements 11a and 11c at the four corners is effective. This is because the distance from the central part of the antenna array 7 is large, so that Figure 2 the variable gain amplifier 13 in the phase shifter IC 2 requires a large amount of attenuation in order to achieve tapering.

[0103] As a general comparative example, it can be conceived to randomly arrange the conduction elements 11a. In this case, the occupancy density of the conduction elements 11a and 11c is basically constant throughout the region. However, assuming that the occupancy density near the center is the average value, the occupancy density is as low as 4 / 9 to 5 / 9, that is, 44% to 56%, so there is a concern that the sidelobe level may deteriorate. According to the configuration of the present embodiment, since the occupancy density near the central part is higher than the occupancy density at the four corners, the deterioration of the sidelobe level can be suppressed while maintaining the number of the arranged conduction elements 11a and 11c.

[0104] In addition, when the conduction element 11a is arranged at the above position of the antenna array 7, the occupancy rate of the conduction element 11a with respect to the antenna array 7 becomes 60.4%. In addition, by grouping two conduction elements 11a in the Y direction (vertical direction), the occupancy rate of the conduction element 11a that requires phase shift control can be reduced to approximately half. In fact, considering the ungrouped conduction element 11c, it is designed to be 33%. This means that controlling 192 × 33% = 64 channels is sufficient. For example, when the phase shifter IC2 is used for 16 channels, only four phase shifter IC2s can be used to control the antenna array 7.

[0105] The simulation results will be described below. The inventors have simulated the structure of the antenna array 7 in which the conduction elements 11a and 11c are arranged as described above. Figure 9 The simulated beam patterns of two types of null filters with d = 3.5λ and d = 7.5λ are shown, which are turned by 17.5° in the E-plane to show the nulls at the same angle as the grating lobes according to the first embodiment, and are plotted simultaneously with the simulated beam pattern for the URA with vertical grouping of adjacent elements turned by 17.5° in the E-plane to show the grating lobes as a reference.

[0106] By configuring the first or second null filter, compared with the case of random placement, the loss of the main beam angle can be minimized, the side lobe level and the grating lobe level can be suppressed, and the stepped / grating lobe angle can be followed and suppressed.

[0107] In addition, Figure 10 The simulated beam patterns of Tx and Rx (in this case Tx = Rx) of the antenna array according to the first embodiment and the combined beam pattern of Tx and Rx turned by 17.5° in the E-plane (where the main lobe peak is normalized to 0 dB) are shown. As Figure 10 shown, in the spectrum after the combination of Tx and Rx, during the 17.5° vertical scan, the grating lobe generated near -43° can be suppressed to -40 dBc or less. In addition, the side lobe level can be suppressed to approximately -35 dBc.

[0108] In addition, Figure 11 The simulated Rx beam pattern of the antenna array with a null filter when the main beam angle is turned by 5° in the E-plane is shown, which is plotted simultaneously with a conventional array with vertical grouping of adjacent elements according to the first embodiment. Figure 12 A diagram schematically showing the simulated Rx beam pattern of the antenna array according to the first embodiment when the main beam angle is turned by 17.5° in the E-plane is shown. Compared with this case of random arrangement, Figure 11 all of these cases in can suppress the side lobe level and the grating lobe level.

[0109] If the grating lobe level remains strong, when the vehicle uses the radar device 1 and the main beam to be detected is adjusted in the forward direction, it will be strongly affected by the road surface reflection in the vertical direction at the installation position of the radar device 1, and the reflection will interfere with the received signal. Therefore, by suppressing the grating lobe level, even when applied to the radar device 1, the influence from the road surface reflection can be suppressed, and false detection can be prevented.

[0110] Figure 13 Shows the transition of the grating lobe generation angle and the simulation result of the grating lobe level when the main beam angle changes from 0° to 40°. When the phase shift value of the phase shifter 14 in the RX phase shift unit 10 is adjusted to continuously change the main beam angle from 0° to 40°, the grating lobe generation angle also changes. However, due to the influence of the zero-value filter embedded in the two-dimensional array, the grating lobe can be suppressed by following the grating lobe angle.

[0111] (Conclusion)

[0112] According to this embodiment, since the occupancy density of the conducting elements 11a near the center part of the antenna array 7 is higher than that of the four corners, this configuration can reduce the number of the arranged conducting elements 11a compared with the random arrangement, and at the same time, the deterioration of the sidelobe level can be suppressed. Therefore, this configuration can suppress the generation of the grating lobe while reducing the number of the arranged conducting elements 11a.

[0113] In addition, according to this embodiment, the conducting element 11c sandwiched by the disconnected elements 11b is arranged so that the periodicity of the phase center can be reduced after the conducting elements 11a are grouped, and the conducting elements 11c are arranged symmetrically by line and point symmetrically at a specific interval. With this configuration, a zero-value filter can be configured, and the grating lobe can be followed and suppressed.

[0114] According to this embodiment, in the design of the conducting elements 11a and the disconnected elements 11b, the density of the conducting elements 11a in the center part increases, and the density of the four corners decreases. With this configuration, since the phase shift control can be further simplified, the number of the phase shifters 14 in the circuit IC2 can be reduced, and the sidelobe level can be reduced. In addition, by grouping adjacent conducting elements 11a, this configuration can commonly control a plurality of conducting elements 11a corresponding to the same phase shifter 14, and the number of the installed phase shifters 14 can be reduced to about half. The grating lobe generated at that time can be suppressed at all required scanning angles.

[0115] (Second Embodiment)

[0116] Reference will be made to Figure 14 describe the second embodiment. As Figure 14As shown, the shapes of the conductive elements 11a and 11c can be formed into shapes other than a quadrilateral, for example, a polygonal shape such as an octagon. Figure 14 Only the conductive elements 11a and 11c arranged in an octagonal shape are shown, and the disconnecting elements 11b and the dummy elements 11d are not shown. The second embodiment provides an effect similar to the above-described embodiment. In addition, the shapes of the conductive elements 11a and 11c can be different from each other.

[0117] (Third Embodiment)

[0118] Reference will be made to Figure 15 and Figure 16 to describe the third embodiment. Figure 15 It is shown that the conductive elements 11a and 11c are configured in an octagonal shape as in the second embodiment. As Figure 15 shown, the conductive element 11a has coordinate centers arranged at least in part in a two-dimensional grid lattice in a predetermined rule. Desirably, the coordinate centers are arranged to be two-dimensionally shifted from the positions of the grid points to the top, bottom, left, or right of the center of the grid lattice. It is desirable that the conductive element 11C is fixedly arranged in the grid lattice.

[0119] Figure 15 It shows the desired directions for shifting the conductive element 11a from the center of the grid lattice. Desirably, the conductive element 11a arranged on the center side in the X direction is shifted outward along the X direction by a predetermined interval less than 0.5λ grid point interval. In addition, it is desirable that the conductive element 11a arranged on the center side in the Y direction is shifted along the Y direction by a predetermined interval less than 0.5λ grid point interval so as to be guided toward the conductive element 11c.

[0120] In addition, as Figure 15 shown by the arrow directions in, it is desirable that the conductive element 11a in the diagonal direction of XY is shifted toward the center direction by a predetermined interval less than 0.5λ grid point interval. It is desirable that these shift intervals are set to be line-symmetric in the X and Y directions, that is, point-symmetric at the center position at the same interval. It can be conceived that the conductive element 11a will be slightly offset to an angle filling the area of the disconnecting element 11b in order to disrupt the periodicity of the phase center. As a result, it is expected that this configuration can suppress grating lobes.

[0121] In addition, Figure 16 It shows the arrangement pitch of the conductive elements 11a and 11c in four rows X5 to X8 on the center side in the X direction. The pitch of the conductive elements 11c in rows X6 and X7 in the Y direction is fixed at 3.5λ. The pitch of the conductive elements 11c in rows X5 and X8 in the Y direction is fixed at 7.5λ. As a result, the characteristics of the conductive element 11c as a null filter can be maintained.

[0122] (Other Embodiments)

[0123] The present disclosure is not limited to the above-described embodiments, but can be implemented in various variations and can be applied to various embodiments without departing from its gist. For example, the present disclosure can be modified as follows.

[0124] The two conduction elements 11a are grouped in the Y direction (i.e., in the vertical direction), however, the present disclosure is not limited thereto. The two conduction elements 11a can be grouped in the X direction (i.e., in the horizontal direction). Although an embodiment in which two single conduction elements 11c are arranged to be separated in the Y direction (i.e., in the vertical direction) has been described, the present disclosure is not limited thereto. Four or more conduction elements 11c can be arranged in a separated state in the Y direction.

[0125] The present invention has been described based on the above-described embodiments. However, it should be understood that the present invention is not limited to the embodiment and the structure. The present disclosure encompasses various modifications and variations within an equivalent range. Additionally, various modes / combinations, and one or more elements added to / removed from them can also be considered as the present disclosure and be understood as its technical idea.

Claims

1. An antenna array for a high-frequency device, comprising: a plurality of antenna elements for a radar device and arranged in a two-dimensional array in a predetermined area, wherein the plurality of antenna elements include conducting elements electrically connected to a phase shifter, and the conducting elements are arranged such that the density of the conducting elements at the central portion of the two-dimensional array is high, and the density of the conducting elements at the four corner portions of the two-dimensional array is low, wherein the conducting elements adjacent to each other in the vertical or horizontal direction in the two-dimensional array are defined as first conducting elements, the adjacent first conducting elements are grouped and controlled by the phase shifter.

2. The antenna array according to claim 1, wherein the plurality of antenna elements include disconnecting elements not electrically connected to the phase shifter, a single conducting element electrically connected to a single phase shifter is defined as a second conducting element, and the second conducting elements are arranged to be separated from each other in the same direction as the direction in which the first conducting elements are grouped.

3. The antenna array according to claim 2, wherein the second conducting elements are set as zero-value filters and arranged to be linearly symmetrically arranged with respect to the center of the antenna array at intervals of (0.5 + m)λ (assuming m = 1, 2,...).

4. The antenna array according to claim 3, wherein, in the interval of (0.5 + m)λ, the value of m is equal to or greater than 3.

5. The antenna array according to claim 3 or 4, wherein multiple groups of second conducting elements are provided, and in each of the multiple groups of second conducting elements, the value of m is different in the interval of (0.5 + m)λ.

6. The antenna array according to claim 3 or 4, wherein the second conducting elements are arranged point-symmetrically with respect to the center of the antenna array at the vertices of a two-dimensional quadrilateral.

7. The antenna array according to any one of claims 1 to 4, wherein the conducting elements have a polygonal shape other than a quadrilateral.

8. The antenna array according to any one of claims 1 to 4, wherein virtual elements are arranged on the outermost periphery of the two-dimensional array.

9. The antenna array according to any one of claims 1 to 4, wherein the line lengths of the transmission lines connecting the IC pads including the phase shifter and the conducting elements are equal to each other or have a relationship of p×λ (where p is an integer) with each other, so as to align the phases for all channels.

10. The antenna array according to any one of claims 1 to 4, wherein the antenna array is provided by a hybrid radar structure having a plurality of mixers.

11. The antenna array according to any one of claims 1 to 4, wherein the antenna array is used as a phased array antenna for the transmitter unit and the receiver unit of the radar device.

12. The antenna array according to claim 5, wherein the antenna array is used as a phased array antenna for the transmitter unit and the receiver unit of the radar device.

13. The antenna array according to claim 1, wherein two of the first conducting elements in the vertical or horizontal direction in the two-dimensional array are grouped, and The two grouped first conduction elements are controlled by the same phase shifter.

14. An antenna array for a high-frequency device, comprising: a plurality of antenna elements for a radar device and arranged in a two-dimensional array in a predetermined area, wherein the plurality of antenna elements include conduction elements electrically connected to phase shifters, and the conduction elements are arranged such that the density of the conduction elements at the central portion of the two-dimensional array is high, and the density of the conduction elements at the four corners of the two-dimensional array is low, wherein the conduction elements adjacent to each other in the vertical or horizontal direction of the two-dimensional array are defined as first conduction elements, the first conduction elements have coordinate centers arranged in at least a part of a two-dimensional grid lattice according to a predetermined rule, and the coordinate centers are shifted from the grid points to the top side, bottom side, left side or right side.

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