Control Method for a Current-Source Dual-Active-Bridge Converter with Bipolar Output

Abstract

The present application is applicable to the field of power electronics technology, and provides a control method for a current source type dual active bridge converter with bipolar output. The control method includes: collecting the input current of the primary side of the current source type dual active bridge converter, the capacitance voltage of the clamping capacitor, the target output voltage of the target output port in the secondary side, and the total output voltage of the secondary side; calculating the first duty cycle of the switching tubes in the primary side of the converter according to the capacitance voltage; calculating the phase shift angle between the primary and secondary sides of the converter according to the total output voltage and the input current; calculating the second duty cycle of the switching tubes in the secondary side of the converter according to the total output voltage and the target output voltage; generating a driving signal for controlling the switching tubes in the converter according to the first duty cycle, the phase shift angle between the primary and secondary sides, and the second duty cycle; and controlling the on / off of the switching tubes in the converter by using the driving signal. The present application can achieve bipolar voltage balance and soft switching of the bipolar output converter, and reduce the switching loss.

Description

Technical Field

[0001] This application belongs to the field of power electronics technology, and particularly relates to a control method for a current-source type dual-active bridge converter with bipolar output. Background Art

[0002] With the progress of science and technology and the demand for environmental protection and sustainable development, the development of renewable energy and distributed energy supply technologies has become a research hotspot in recent years due to their cleanliness, high efficiency, and ability to improve the operation of the power grid. However, distributed power sources such as photovoltaic and wind power are vulnerable to environmental impacts and have volatility. They are uncontrollable power sources relative to the power grid and bring a series of problems such as high single-unit grid connection costs and difficult control. Therefore, relevant researchers have proposed the concept of a DC microgrid to reduce the impact of distributed power sources on the power grid, which is more conducive to the access of distributed power sources and can reduce the investment in reactive power compensation equipment compared with an AC power grid. Among them, the structure of the DC microgrid can be divided into a single-pole DC bus and a bipolar DC bus. In contrast, the bipolar DC bus system is more reliable because when one DC bus fails, the other DC bus can still operate normally and can provide three different voltage levels, making it more flexible to use.

[0003] However, there is a key problem in the bipolar DC microgrid, that is, the bipolar voltage imbalance. This is because loads with different powers or characteristics are connected to the bipolar DC bus, causing the bipolar voltage to deviate from the originally symmetric voltage. In existing power electronic devices, in order to generate a bipolar voltage port and alleviate the problem of voltage imbalance, a measure of cascading a voltage balancer after a traditional DC-DC converter is usually taken. Although the bipolar voltage can be balanced, due to the influence of unbalanced loads, it is difficult for the converter to achieve soft switching, resulting in an increase in switching losses and a large bipolar voltage deviation. Summary of the Invention

[0004] The embodiment of this application provides a control method for a current-source type dual-active bridge converter with bipolar output, which can solve the problems that it is difficult for the converter to achieve soft switching, resulting in an increase in switching losses and a large bipolar voltage deviation.

[0005] The embodiment of this application provides a control method for a current-source type dual-active bridge converter with bipolar output. The output ports on the secondary side of the current-source type dual-active bridge converter include a first output port and a second output port. The control method includes:

[0006] Collect the input current on the primary side of the current-source type dual-active bridge converter, the capacitance voltage of the clamping capacitor in the primary side of the current-source type dual-active bridge converter, the target output voltage of the target output port in the secondary side of the current-source type dual-active bridge converter, and the total output voltage of the secondary side of the current-source type dual-active bridge converter;

[0007] Calculate the first duty cycle of the switching tube in the primary side of the current-source dual-active-bridge converter according to the capacitor voltage;

[0008] Calculate the phase-shift angle between the primary and secondary sides of the current-source dual-active-bridge converter according to the total output voltage and the input current;

[0009] Calculate the second duty cycle of the switching tube in the secondary side of the current-source dual-active-bridge converter according to the total output voltage and the target output voltage;

[0010] Generate the driving signal for controlling the switching tube in the current-source dual-active-bridge converter according to the first duty cycle, the phase-shift angle between the primary and secondary sides, and the second duty cycle;

[0011] Control the on-off of the switching tube in the current-source dual-active-bridge converter by using the driving signal to balance the voltages of the first output port and the second output port.

[0012] Wherein, the target output port is the first output port or the second output port.

[0013] Optionally, calculating the first duty cycle of the switching tube in the primary side of the current-source dual-active-bridge converter according to the capacitor voltage includes:

[0014] Calculate the capacitor voltage difference between the capacitor voltage and the preset capacitor voltage threshold;

[0015] Process the capacitor voltage difference by using a voltage closed-loop PI regulator to output a first control quantity;

[0016] Process the first control quantity output by the voltage closed-loop PI regulator by using a limiter to obtain the first duty cycle of the switching tube in the primary side of the current-source dual-active-bridge converter.

[0017] Optionally, the first duty cycle D1 of the switching tube in the primary side of the current-source dual-active-bridge converter should satisfy the following formula:

[0018]

[0019] Wherein, V in represents the input voltage of the primary side of the current-source dual-active-bridge converter, K represents the transformer turns ratio of the current-source dual-active-bridge converter, V o represents the total output voltage of the secondary side of the current-source dual-active-bridge converter, represents the capacitor voltage.

[0020] Optionally, calculating the phase-shift angle between the primary and secondary sides of the current-source dual-active-bridge converter according to the total output voltage and the input current includes:

[0021] Calculate the output total voltage difference between the total output voltage and the preset total output voltage threshold;

[0022] The output total voltage difference is processed by a voltage closed-loop PI regulator to output a second control quantity;

[0023] The second control quantity output by the voltage closed-loop PI regulator is processed by a limiter;

[0024] The current difference between the data output by the limiter and the input current is calculated;

[0025] The current difference is processed by a current closed-loop PI regulator to output a third control quantity;

[0026] The third control quantity output by the current closed-loop PI regulator is processed by a limiter to obtain the phase shift angle between the primary and secondary sides of the current-source dual-active-bridge converter.

[0027] Optionally, according to the total output voltage and the target output voltage, the second duty ratio of the switching tube in the secondary side of the current-source dual-active-bridge converter is calculated, including:

[0028] The voltage difference between the target output voltage and half of the total output voltage is calculated;

[0029] The voltage difference is processed by a voltage closed-loop PI regulator to output a fourth control quantity;

[0030] The fourth control quantity output by the voltage closed-loop PI regulator is processed by a limiter to obtain the second duty ratio of the switching tube in the secondary side of the current-source dual-active-bridge converter.

[0031] Optionally, the current-source dual-active-bridge converter includes a primary side, a secondary side, and a transformer;

[0032] The secondary side includes a first switching tube, a second switching tube, a first capacitor, a second capacitor, a first load, and a second load;

[0033] The primary side includes an input power supply, a first DC inductor, a second DC inductor, a leakage inductor, the clamping capacitor, a third switching tube, a fourth switching tube, a fifth switching tube, and a sixth switching tube;

[0034] The transformer includes a primary winding and a secondary winding;

[0035] The positive pole of the input power supply is respectively connected to the first end of the first DC inductor and the first end of the second DC inductor. The negative pole of the input power supply is respectively connected to the source electrodes of the third switching tube, the fourth switching tube, and the negative pole of the clamping capacitor. The positive pole of the clamping capacitor is respectively connected to the drain electrodes of the fifth switching tube and the sixth switching tube. The first end of the leakage inductor is respectively connected to the drain electrode of the third switching tube, the source electrode of the fifth switching tube, and the second end of the first DC inductor. The second end of the leakage inductor is connected to the positive pole of the primary winding. The negative pole of the primary winding is respectively connected to the drain electrode of the fourth switching tube, the source electrode of the sixth switching tube, and the second end of the second DC inductor;

[0036] The positive pole of the secondary winding is respectively connected to the source pole of the first switching tube and the drain pole of the second switching tube, the negative pole of the secondary winding is respectively connected to the negative pole of the first capacitor and the positive pole of the second capacitor, the drain pole of the first switching tube is connected to the positive pole of the first capacitor, and the source pole of the second switching tube is connected to the negative pole of the second capacitor;

[0037] The first end of the first load is connected to the positive pole of the first capacitor, the second end of the first load is connected to the negative pole of the first capacitor, and both ends of the first load are the first output port;

[0038] The first end of the second load is connected to the positive pole of the second capacitor, the second end of the second load is connected to the negative pole of the second capacitor, and both ends of the second load are the second output port.

[0039] Optionally, the duty ratios of the third switching tube and the fourth switching tube are both the first duty ratio; the duty ratio of the first switching tube is the second duty ratio; the drive signals of the third switching tube and the fifth switching tube are complementary; the drive signals of the fourth switching tube and the sixth switching tube are complementary; the phase shift between the drive signals of the third switching tube and the fourth switching tube is 180 degrees; the phase shift between the drive signal of the fourth switching tube and the drive signal of the first switching tube is the primary-secondary phase shift angle; the drive signals of the first switching tube and the second switching tube are complementary.

[0040] Optionally, generating the drive signals of the switching tubes in the current source type dual active bridge converter according to the first duty ratio, the primary-secondary phase shift angle and the second duty ratio includes:

[0041] Generating the drive signal for controlling the third switching tube according to the first duty ratio;

[0042] Generating the drive signal of the fifth switching tube according to the drive signal of the third switching tube and the complementary relationship between the drive signals of the third switching tube and the fifth switching tube;

[0043] Generating the drive signal of the fourth switching tube according to the drive signal of the third switching tube and the phase shift between the drive signals of the third switching tube and the fourth switching tube;

[0044] Generating the drive signal of the sixth switching tube according to the drive signal of the fourth switching tube and the complementary relationship between the drive signals of the fourth switching tube and the sixth switching tube;

[0045] Generating the drive signal of the first switching tube according to the second duty ratio, the primary-secondary phase shift angle and the drive signal of the fourth switching tube;

[0046] Generating the drive signal of the second switching tube according to the drive signal of the first switching tube and the complementary relationship between the drive signals of the first switching tube and the second switching tube.

[0047] Optionally, the inductance value of the excitation inductor in the transformer of the current source type dual active bridge converter is 160 to 320 μH.

[0048] The above solution of this application has the following beneficial effects:

[0049] In the embodiment of this application, by collecting the input current on the primary side of the current source type dual active bridge converter, the capacitor voltage of the clamping capacitor, the target output voltage of the target output port on the secondary side, and the total output voltage, and generating the duty cycle of the switching tubes on the primary and secondary sides of the current source type dual active bridge converter and the phase shift angle between the primary and secondary sides according to the collected current and voltage, then generating a driving signal for controlling the switching tubes based on the duty cycle of the switching tubes on the primary and secondary sides and the phase shift angle between the primary and secondary sides, and using the driving signal to control the on and off of the switching tubes, thereby enabling the converter to achieve bipolar voltage balance and soft switching, and reducing the switching loss.

[0050] Other beneficial effects of this application will be described in detail in the subsequent specific implementation section. Description of the Drawings

[0051] In order to more clearly illustrate the technical solutions in the embodiments of this application, the following will briefly introduce the drawings required for use in the embodiments or the description of the prior art. Obviously, the drawings in the following description are only some embodiments of this application. For those of ordinary skill in the art, without creative efforts, other drawings can also be obtained based on these drawings.

[0052] Figure 1 It is a flowchart of the control method for a current source type dual active bridge converter with bipolar output provided by an embodiment of this application;

[0053] Figure 2 It is a schematic diagram of the topology structure of a current source type dual active bridge converter with bipolar output provided by an embodiment of this application;

[0054] Figure 3 It is a comparison diagram of the soft switching range of the converter provided by an embodiment of this application;

[0055] Figure 4a It is a steady-state waveform diagram of the forward working mode of a current source type dual active bridge converter with bipolar output provided by an embodiment of this application;

[0056] Figure 4b It is a steady-state waveform diagram of the reverse working mode of a current source type dual active bridge converter with bipolar output provided by an embodiment of this application;

[0057] Figure 5a It is an equivalent circuit diagram of the secondary side when the first switching tube S3 is conducting and the second switching tube S4 is off in a current source type dual active bridge converter with bipolar output provided by an embodiment of this application;

[0058] Figure 5b This is the equivalent circuit diagram of the secondary side when the second switching transistor S4 of the current source type dual-active-bridge converter with bipolar output provided by an embodiment of the present application is turned on and the first switching transistor S3 is turned off. Detailed implementation manners

[0059] In the following description, specific details such as specific system structures and technologies are presented for the purpose of illustration rather than limitation, so as to thoroughly understand the embodiments of the present application. However, those skilled in the art should clearly understand that the present application can also be implemented in other embodiments without these specific details. In other cases, detailed descriptions of well-known systems, devices, circuits, and methods are omitted to avoid unnecessary details from interfering with the description of the present application.

[0060] It should be understood that when used in the specification and appended claims of the present application, the term "comprising" indicates the presence of the described features, wholes, steps, operations, elements, and / or components, but does not exclude the presence or addition of one or more other features, wholes, steps, operations, elements, components, and / or their combinations.

[0061] It should also be understood that the term "and / or" as used in the specification and appended claims of the present application refers to any combination and all possible combinations of one or more of the associated listed items, and includes these combinations.

[0062] As used in the specification and appended claims of the present application, the term "if" can be interpreted as "when", "once", "in response to determining", or "in response to detecting" according to the context. Similarly, the phrase "if determined" or "if [the described condition or event] is detected" can be interpreted as meaning "once determined", "in response to determining", "once [the described condition or event] is detected", or "in response to detecting [the described condition or event]" according to the context.

[0063] In addition, in the description of the specification and appended claims of the present application, the terms "first", "second", "third", etc. are only used for distinguishing descriptions and cannot be understood as indicating or implying relative importance.

[0064] References to "one embodiment" or "some embodiments" etc. described in the specification of this application mean that specific features, structures, or characteristics described in connection with that embodiment are included in one or more embodiments of this application. Thus, statements such as "in one embodiment", "in some embodiments", "in other some embodiments", "in still other embodiments", etc. that appear in different places in this specification do not necessarily all refer to the same embodiment, but mean "one or more but not all embodiments", unless otherwise specifically emphasized in another way. The terms "comprising", "including", "having" and their variants all mean "including but not limited to", unless otherwise specifically emphasized in another way.

[0065] In existing power electronic devices, in order to generate a bipolar voltage port and alleviate the problem of voltage imbalance, a measure is taken to cascade a voltage balancer after a traditional DC-DC converter. Although the bipolar voltage can be balanced, due to the influence of unbalanced loads, it is difficult for the converter to achieve soft switching, resulting in an increase in switching losses and a large bipolar voltage deviation.

[0066] In view of the above problems, the embodiments of this application provide a control method for a current-source type dual-active-bridge converter with bipolar output. This control method collects the input current of the primary side of the current-source type dual-active-bridge converter, the capacitance voltage of the clamping capacitor, the target output voltage of the target output port of the secondary side, and the total output voltage, and generates the duty cycles of the switching tubes in the primary and secondary sides of the current-source type dual-active-bridge converter and the phase-shift angle between the primary and secondary sides according to the collected current and voltage. Then, based on the duty cycles of the switching tubes in the primary and secondary sides and the phase-shift angle between the primary and secondary sides, drive signals for controlling the switching tubes are generated, and the on-off of the switching tubes is controlled by using the drive signals, so that the converter achieves bipolar voltage balance and soft switching, and reduces the switching losses.

[0067] It should be noted that the above control method for the current-source type dual-active-bridge converter can be executed by a controller. Exemplarily, the above controller can be a (DSP, Digital Signal Processing) processor.

[0068] The following gives an exemplary description of a control method for a current-source type dual-active-bridge converter with bipolar output provided by this application.

[0069] The above current-source type dual-active-bridge converter with bipolar output is a bipolar output converter, that is, the secondary side of this current-source type dual-active-bridge converter has two output ports, specifically a first output port and a second output port. It should be noted that the current-source type dual-active-bridge converter in the following text is all this current-source type dual-active-bridge converter with bipolar output.

[0070] Such as Figure 1As shown in the figure, a control method for a current-source type dual-active bridge converter with bipolar output provided by an embodiment of the present application includes the following steps:

[0071] Step 11, collect the input current of the primary side of the current-source type dual-active bridge converter, the capacitance voltage of the clamping capacitor in the primary side of the current-source type dual-active bridge converter, the target output voltage of the target output port in the secondary side of the current-source type dual-active bridge converter, and the total output voltage of the secondary side of the current-source type dual-active bridge converter.

[0072] The above total output voltage refers to the sum of the output voltage of the first output port and the output voltage of the second output port; the above target output port refers to the first output port or the second output port; the above target output voltage refers to the output voltage of the first output port or the output voltage of the second output port.

[0073] In some embodiments of the present application, the above capacitance voltage, target output voltage, and total output voltage can be collected by a voltage sensor and output to the controller; the above input current can be collected by a current sensor and output to the controller.

[0074] Step 12, calculate the first duty cycle of the switching tube in the primary side of the current-source type dual-active bridge converter according to the capacitance voltage.

[0075] Calculating the first duty cycle here is for facilitating the subsequent generation of the driving signal for controlling the switching tube in the primary side of the current-source type dual-active bridge converter.

[0076] Step 13, calculate the primary-secondary phase shift angle of the current-source type dual-active bridge converter according to the total output voltage and the input current.

[0077] Calculating the primary-secondary phase shift angle of the current-source type dual-active bridge converter here facilitates the subsequent generation of the driving signal for controlling the switching tube in the secondary side of the current-source type dual-active bridge converter according to the phase shift relationship between the switching tube in the primary side and the switching tube in the secondary side of the current-source type dual-active bridge converter.

[0078] Step 14, calculate the second duty cycle of the switching tube in the secondary side of the current-source type dual-active bridge converter according to the total output voltage and the target output voltage.

[0079] Calculating the second duty cycle here is for generating the driving signal for controlling the switching tube in the secondary side of the current-source type dual-active bridge converter.

[0080] Step 15, generate the driving signal for controlling the switching tube in the current-source type dual-active bridge converter according to the first duty cycle, the primary-secondary phase shift angle, and the second duty cycle.

[0081] The driving signal generated here can control the on-off of the switching tubes in the primary side and the secondary side of the current-source type dual-active bridge converter, thereby achieving the balance of the bipolar voltage.

[0082] Step 16: Use the drive signal to control the on / off of the switching tubes in the current-source dual-active-bridge converter, so as to balance the voltages of the first output port and the second output port.

[0083] The above drive signal can be a pulse-width modulation (PWM) drive signal. Exemplarily, after generating the PWM drive signal for controlling the switching tubes in the current-source dual-active-bridge converter, the controller can output the PWM drive signal to the gates of the switching tubes in the current-source dual-active-bridge converter through the drive circuit to control the on / off of the switching tubes.

[0084] It should be noted that in some embodiments of the present application, the balance of the voltages of the first output port and the second output port means that the difference between the bipolar voltages (i.e., the voltages of the first output port and the second output port) is within 0 to 2%.

[0085] It is worth mentioning that the above control method collects the input current on the primary side of the current-source dual-active-bridge converter, the capacitor voltage of the clamping capacitor, the target output voltage of the target output port on the secondary side, and the total output voltage, and generates the duty cycles of the switching tubes in the primary and secondary sides of the current-source dual-active-bridge converter and the phase-shift angle between the primary and secondary sides according to the collected current and voltage. Then, based on the duty cycles of the switching tubes in the primary and secondary sides and the phase-shift angle between the primary and secondary sides, a drive signal for controlling the switching tubes is generated, and the drive signal is used to control the on / off of the switching tubes, so that the converter realizes soft switching and reduces the switching loss.

[0086] In some embodiments of the present application, the inductance value of the magnetizing inductance in the transformer of the current-source dual-active-bridge converter can be 160 to 320 μH. In this way, the ripple of the magnetizing inductance current can be appropriately increased to assist in realizing soft switching.

[0087] Next, an exemplary description of the specific implementation process of step 12 is given.

[0088] Step 12.1: Calculate the difference between the capacitor voltage and the preset capacitor voltage threshold.

[0089] According to the capacitor voltage value of the clamping capacitor in the primary side of the current-source dual-active-bridge converter and the preset value of the capacitor voltage of the clamping capacitor, a capacitor voltage deviation is formed, and the capacitor voltage deviation is input into the voltage closed-loop PI regulator.

[0090] Step 12.2: Process the difference between the capacitor voltages by using the voltage closed-loop PI regulator and output a first control quantity.

[0091] The voltage closed-loop PI regulator forms a first control quantity through the linear combination of the proportion and integral of the capacitor voltage deviation, and outputs the first control quantity to the limiter. It should be noted that the first control quantity output by the voltage closed-loop PI regulator is actually a duty cycle.

[0092] Step 12.3: Use the limiter to process the first control quantity output by the voltage closed-loop PI regulator to obtain the first duty cycle of the switching tube in the primary side of the current-source dual-active-bridge converter.

[0093] The limiter performs a limiting process on the input first control quantity, limits the duty cycle between 0.5 and 0.72, and outputs the first duty cycle.

[0094] It should be noted that the above first duty cycle D1 should satisfy the following formula:

[0095]

[0096] where V in represents the input voltage of the primary side of the current-source dual-active-bridge converter, K represents the transformer turns ratio of the current-source dual-active-bridge converter, V o represents the total output voltage of the secondary side of the current-source dual-active-bridge converter, represents the capacitor voltage.

[0097] According to the above formula, the capacitor voltage of the clamping capacitor can be controlled by adjusting the value of the duty cycle D1 of the switching tube in the primary side of the current-source dual-active-bridge converter, so that the voltage matching between the primary and secondary sides of the current-source dual-active-bridge converter can be realized, thereby reducing the circulating current of the circuit and improving the efficiency and power density of the current-source dual-active-bridge converter.

[0098] Next, an exemplary description of the specific implementation steps of step 13 is given.

[0099] Step 13.1: Calculate the output total voltage difference between the total output voltage and the preset total output voltage threshold.

[0100] According to the total output voltage deviation formed by the total output voltage of the secondary side of the current-source dual-active-bridge converter and the total output voltage preset value, the total output voltage deviation is input to the voltage closed-loop PI regulator.

[0101] Step 13.2: Use the voltage closed-loop PI regulator to process the output total voltage difference and output a second control quantity.

[0102] The voltage closed-loop PI regulator forms a second control quantity through the linear combination of the proportion and integral of the total output voltage deviation, and outputs the second control quantity to the limiter.

[0103] Step 13.3, process the second control quantity output by the voltage closed-loop PI regulator by using a limiter.

[0104] It should be noted that the second control quantity output by the voltage closed-loop PI regulator here is actually a current. The limiter performs a limiting process on the input second control quantity and outputs it.

[0105] Step 13.4, calculate the current difference between the data output by the limiter and the input current.

[0106] According to the current deviation formed by the data output by the limiter and the input current, input the current deviation into the current closed-loop PI regulator.

[0107] Step 13.5, process the current difference by using the current closed-loop PI regulator to output a third control quantity.

[0108] The voltage closed-loop PI regulator forms the third control quantity by linearly combining the proportion and integral of the current deviation and outputs the third control quantity to the limiter.

[0109] Step 13.6, process the third control quantity output by the current closed-loop PI regulator by using a limiter to obtain the phase shift angle between the primary and secondary sides of the current-source dual-active-bridge converter.

[0110] It should be noted that the third control quantity output by the current closed-loop PI regulator here is actually a phase shift angle. The limiter performs a limiting process on the input third control quantity and outputs the final phase shift angle between the primary and secondary sides.

[0111] Next, an exemplary description is given for the specific implementation steps of Step 14.

[0112] Step 14.1, calculate the voltage difference between the target output voltage and half of the total output voltage.

[0113] According to the voltage value of half of the total output voltage and the target output voltage, form an output voltage deviation, and input the output voltage deviation into the voltage closed-loop PI regulator.

[0114] Step 14.2, process the output voltage difference by using the voltage closed-loop PI regulator to output a fourth control quantity.

[0115] The voltage closed-loop PI regulator forms the fourth control quantity by linearly combining the proportion and integral of the output voltage difference and outputs the fourth control quantity to the limiter. It should be noted that the fourth control quantity output by the voltage closed-loop PI regulator is actually a duty cycle.

[0116] Step 14.3, process the fourth control quantity output by the voltage closed-loop PI regulator by using a limiter to obtain the second duty cycle of the switching tube in the secondary side of the current-source dual-active-bridge converter.

[0117] The limiter performs a limiting process on the fourth control quantity input and outputs the second duty cycle.

[0118] It is worth mentioning that the duty cycle of the switching tubes on the secondary side of the fine-tuning current source type dual active bridge converter is around 0.5, which can accurately control the bipolar voltage and eliminate the steady-state error.

[0119] It should be noted that the above current closed-loop PI regulator and voltage closed-loop PI regulator are both commonly used regulators, so their principles will not be elaborated too much here.

[0120] Next, an exemplary description will be given of the structure of the current source type dual active bridge converter with bipolar output provided by this application.

[0121] The above current source type dual active bridge converter includes a primary side, a secondary side, and a transformer.

[0122] As Figure 2 shown, the secondary side of the above current source type dual active bridge converter includes a first switching tube (such as Figure 2 S3 in Figure 2 ), a second switching tube (such as Figure 2 S4 in a ), a first capacitor (such as Figure 2 C in b ), a second capacitor (such as Figure 2 C in Figure 2 ), a first load (such as R1 in Figure 2 ), a second load (such as Figure 2 R2 in Figure 2 ).

[0123] The primary side of the above current source type dual active bridge converter includes an input power supply (such as Figure 2 V in in ), a first DC inductor (such as Figure 2 L1 in Figure 2 ), a second DC inductor (such as Figure 2 L2 in Figure 2 ), a leakage inductor (such as Figure 2 L in r ), a clamping capacitor (such as Figure 2 C in c ), a third switching tube (such as Figure 2 Q1 in Figure 2 ), a fourth switching tube (such as Figure 2 Q2 in Figure 2 ), a fifth switching tube (such as Figure 2 Q in 1a ), a sixth switching tube (such as Figure 2 Q in 2a ).

[0124] The above transformer includes a primary winding (such as Figure 2 N1 in Figure 2 ) and a secondary winding (such as Figure 2 N2 in Figure 2 ).

[0125] Among them, the positive pole of the input power supply is respectively connected to the first ends of the first DC inductor and the second DC inductor, the negative pole of the input power supply is respectively connected to the source electrodes of the third switching tube, the fourth switching tube, and the negative pole of the clamping capacitor, the positive pole of the clamping capacitor is respectively connected to the drain electrodes of the fifth switching tube and the sixth switching tube, the first end of the leakage inductor is respectively connected to the drain electrode of the third switching tube, the source electrode of the fifth switching tube, and the second end of the first DC inductor, the second end of the leakage inductor is connected to the positive pole of the primary winding, and the negative pole of the primary winding is respectively connected to the drain electrode of the fourth switching tube, the source electrode of the sixth switching tube, and the second end of the second DC inductor.

[0126] The positive pole of the secondary winding is respectively connected to the source electrode of the first switching tube and the drain electrode of the second switching tube, the negative pole of the secondary winding is respectively connected to the negative pole of the first capacitor and the positive pole of the second capacitor, the drain electrode of the first switching tube is connected to the positive pole of the first capacitor, and the source electrode of the second switching tube is connected to the negative pole of the second capacitor.

[0127] The first end of the above-mentioned first load is connected to the positive pole of the first capacitor, the second end of the first load is connected to the negative pole of the first capacitor, and the two ends of the first load are the first output port; the first end of the above-mentioned second load is connected to the positive pole of the second capacitor, the second end of the second load is connected to the negative pole of the second capacitor, and the two ends of the second load are the second output port.

[0128] Among them, the relationship of the driving signals between the above-mentioned switching tubes (the first switching tube S3, the second switching tube S4, the third switching tube Q1, the fourth switching tube Q2, the fifth switching tube Q 1a and the sixth switching tube Q 2a ) is as follows: the driving signals of the third switching tube Q1 and the fifth switching tube Q 1a are complementary; the driving signals of the fourth switching tube Q2 and the sixth switching tube Q 2a are complementary; the phase shift between the driving signals of the third switching tube Q1 and the fourth switching tube Q2 is 180 degrees; the phase shift between the driving signal of the fourth switching tube Q2 and the driving signal of the first switching tube S3 is the primary-secondary phase shift angle; the driving signals of the first switching tube S3 and the second switching tube S4 are complementary.

[0129] The following makes an exemplary description of the specific steps of step 15.

[0130] Step 15.1, generate a driving signal for controlling the third switching tube according to the first duty ratio.

[0131] Among them, the duty ratios of the above-mentioned third switching tube and the fourth switching tube are both the first duty ratio. Specifically, after obtaining the first duty ratio, the PWM driving signal for controlling the third switching tube can be generated through the unipolar carrier modulation method and the PWM generation circuit.

[0132] Step 15.2: Generate the driving signal of the fifth switching transistor based on the driving signal of the third switching transistor and the complementary relationship between the driving signal of the third switching transistor and the driving signal of the fifth switching transistor.

[0133] In some embodiments of the present application, since the driving signal of the third switching transistor and the driving signal of the fifth switching transistor are complementary, after obtaining the PWM driving signal of the third switching transistor, the driving signal of the fifth switching transistor can be obtained based on the complementary relationship.

[0134] Step 15.3: Generate the driving signal of the fourth switching transistor based on the driving signal of the third switching transistor and the phase shift between the driving signal of the third switching transistor and the driving signal of the fourth switching transistor.

[0135] In some embodiments of the present application, after obtaining the first duty cycle, the PWM driving signal for controlling the fourth switching transistor can be generated by the unipolar carrier modulation method and the PWM generation circuit. At the same time, since the phase shift between the driving signal of the third switching transistor Q1 and the driving signal of the fourth switching transistor Q2 is 180 degrees, therefore, it is also necessary to adjust the PWM driving signal of the fourth switching transistor based on the driving signal of the third switching transistor to obtain the driving signal for controlling the fourth switching transistor.

[0136] Step 15.4: Generate the driving signal of the sixth switching transistor based on the driving signal of the fourth switching transistor and the complementary relationship between the driving signal of the fourth switching transistor and the driving signal of the sixth switching transistor.

[0137] In some embodiments of the present application, since the driving signal of the fourth switching transistor and the driving signal of the sixth switching transistor are complementary, after obtaining the driving signal of the fourth switching transistor, the driving signal of the sixth switching transistor can be obtained based on the complementary relationship.

[0138] Step 15.5: Generate the driving signal of the first switching transistor based on the second duty cycle, the primary-secondary phase shift angle, and the driving signal of the fourth switching transistor.

[0139] Wherein, the duty cycle of the first switching transistor is the second duty cycle. Specifically, after obtaining the second duty cycle, the PWM driving signal for controlling the first switching transistor can be generated by the unipolar carrier modulation method and the PWM generation circuit. At the same time, since the phase shift between the driving signal of the first switching transistor and the driving signal of the fourth switching transistor is the primary-secondary phase shift angle, therefore, it is also necessary to adjust the PWM driving signal of the first switching transistor based on the driving signal of the fourth switching transistor to obtain the driving signal for controlling the first switching transistor.

[0140] Step 15.6: Generate the driving signal of the second switching transistor based on the driving signal of the first switching transistor and the complementary relationship between the driving signal of the first switching transistor and the driving signal of the second switching transistor.

[0141] In some embodiments of the present application, since the driving signal of the second switching transistor and the driving signal of the first switching transistor are complementary, after obtaining the driving signal of the first switching transistor, the driving signal of the second switching transistor can be obtained based on the complementary relationship.

[0142] To facilitate understanding of the technical solution provided by the present application, the control method of the current source type dual active bridge converter with bipolar output provided by the present application will be exemplarily described below with reference to specific embodiments.

[0143] Exemplarily, as shown in Figure 2 , two switching transistors (S3, S4) and two capacitors (C a , C b ) form a half-bridge network. The primary and secondary sides of the converter are connected by a high-frequency transformer for electrical isolation of the input and output terminals. The leakage inductance L r of the transformer acts as an energy storage element and plays a key role in the bidirectional transmission of the converter. I1 represents the input current, represents the inductor current of the first DC inductor L1, represents the inductor current of the second DC inductor L2, represents the inductor current of the leakage inductance L r , L m represents the magnetizing inductance (L m is the inductance presented by the secondary winding N2 when the load side of the transformer is open. ), V out1 represents the output voltage of the first output port, V out2 represents the output voltage of the second output port, V o represents the total output voltage of the secondary side of the converter.

[0144] When the bipolar DC ports of the current source type dual active bridge converter are connected to an unbalanced load (such as R1 > R2), initially the output voltage V out1 of the first output port will be greater than the output voltage V out2 of the second output port. In order to achieve the balance of the bipolar voltage, the current source type dual active bridge converter will generate a DC bias current I bias on its secondary side to compensate for the power at the R2 end and make V out2 as equal as possible to V out1 . However, due to the generation of I bias , the switching current changes when the conduction signals of the first switching transistor S3 and the second switching transistor S4 come (when R1 > R2, the switching current before the first switching transistor S3 turns on will decrease), making it more difficult for the first switching transistor S3 to achieve soft switching (ZVS, Zero Voltage Switch), especially when V out1It is more difficult to achieve soft switching under the condition of open-circuit at the end connection. Moreover, for the secondary side of the current-source dual-active-bridge converter, unbalanced loads will bring about an asymmetric ZVS implementation situation. For example, when R1 > R2, the parasitic capacitance of the first switching tube S3 discharges slowly, and the parasitic capacitance of the second switching tube S4 discharges quickly, deforming the switching current and further exacerbating the problem of bipolar voltage imbalance.

[0145] The above problems can be solved by reasonably designing the exciting inductance. Specifically, in some embodiments of the present application, the exciting inductance L of the current-source dual-active-bridge converter is designed. m The value range of is 160 - 320 μH. The current ripple of the exciting inductance can be appropriately increased, so that the exciting inductance can assist in achieving soft switching within the full load range of the converter, reducing the switching loss, and alleviating the bipolar voltage imbalance problem caused by unbalanced loads. As Figure 3 shown, the dotted line represents the ZVS boundary of the traditional converter ignoring the exciting inductance, and the solid line represents the ZVS boundary when the exciting inductance of the current-source dual-active-bridge converter is set to 265 μH. The ordinate P* represents the per-unit value of the unbalanced load, and the abscissa d represents the duty cycle. It is not difficult to see that the ZVS range of the current-source dual-active-bridge converter has been extended. Through further design improvement, ZVS within the full load range of the current-source dual-active-bridge converter can be achieved.

[0146] On the primary side of the current-source dual-active-bridge converter, the drive signals of the third switching tube Q1 and the fifth switching tube Q 1a are complementary, and the drive signals of the fourth switching tube Q2 and the sixth switching tube Q 2a are complementary. There is a certain dead zone between the complementary drive signals. The phase shift between the third switching tube Q1 and the fourth switching tube Q2 is 180 degrees, and the duty cycles of the third switching tube Q1 and the fourth switching tube Q2 are the first duty cycle D1; on the secondary side of the current-source dual-active-bridge converter, the drive signals of the first switching tube S3 and the second switching tube S4 are complementary, and there is a certain dead zone between them. The phase shift between the fourth switching tube Q2 and the first switching tube S3 is the phase shift angle

[0147] When the current-source dual-active-bridge converter operates in the forward direction (energy flows from the low-voltage side to the high-voltage side), the drive signal of the fourth switching tube Q2 should lead the drive signal of the first switching tube S3. At this time, the phase shift angle is positive, as Figure 4a shown, where V ab represents Figure 2 the voltage between points ab in, V cd represents Figure 2 the voltage between points cd in, and the abscissa represents the unit time. T s represents the switching period of the current-source dual-active-bridge converter.Figure 4b Similarly, when the current-source dual-active-bridge converter operates in the reverse direction (energy flows from the high-voltage side to the low-voltage side), the driving signal of the fourth switch Q2 should lag behind the driving signal of the first switch S3, and the phase-shift angle is negative, as Figure 4b shown.

[0148] Affected by parameters such as dead time and circuit parasitic resistance (on-resistance of the switch and resistance of the secondary winding N2), the current-source dual-active-bridge converter will have problems such as bipolar voltage steady-state error and slow dynamic response. Specifically, combining Figure 5a 、 Figure 5b the schematic diagram of the parasitic elements of the current-source dual-active-bridge converter shown, according to the volt-second balance principle of the magnetizing inductor, the following expressions can be obtained:

[0149]

[0150]

[0151] where, T s is the switching period of the current-source dual-active-bridge converter, is the voltage across the magnetizing inductor, R ds is the on-resistance of the first switch S3 and the second switch S4, R L is the resistance of the secondary winding N2 of the current-source dual-active-bridge converter, i s is the current passing through the secondary side of the current-source dual-active-bridge converter, T1 is the time when the voltage v cd at the midpoint of the secondary bridge arm of the current-source dual-active-bridge converter is positive, T2 is the time when the voltage v cd at the midpoint of the secondary bridge arm of the current-source dual-active-bridge converter is negative, and D2 represents the duty cycle of the first switch S3. Ideally, T1 = T2 = Ts / 2, but in fact, affected by dead time and asymmetric discharge of the parasitic capacitance of the switch, it only satisfies T1 + T2 = T s , and T1 ≠ T2. When the secondary duty cycle is fixed at 0.5, there will be a situation where V out1 ≠ V out2 ≠ V o / 2. Therefore, it is necessary to finely adjust the duty cycle of the switch in the secondary side of the current-source dual-active-bridge converter to achieve the purpose of accurately controlling the bipolar voltage and eliminating the steady-state error. Specifically, adjust the duty cycle of the first switch S3 to about 0.5 according to the voltage value of the target output port.

[0152] Generate the driving signal for controlling the third switch according to the first duty cycle D1 of the third switch Q1, and then according to the driving signal of the third switch Q1 and the fifth switch Q 1aThe relationship between the driving signals (the driving signal of the third switching transistor Q1 and the driving signal of the fifth switching transistor Q 1a are complementary) to generate the driving signal for controlling the fifth switching transistor Q 1a According to the relationship between the driving signal of the third switching transistor Q1 and the driving signal of the fourth switching transistor Q2 (the driving signals of the third switching transistor Q1 and the fourth switching transistor Q2 are phase-shifted by 180 degrees), generate the driving signal for controlling the fourth switching transistor Q2. According to the relationship between the driving signal of the fourth switching transistor Q2 and the driving signal of the sixth switching transistor Q 2a The relationship between the driving signals (the driving signal of the fourth switching transistor Q2 and the driving signal of the sixth switching transistor Q 2a are complementary) to generate the driving signal for controlling the sixth switching transistor Q 2a According to the second duty cycle D2, the primary-secondary phase-shift angle and the driving signal of the fourth switching transistor Q2 to generate the driving signal of the first switching transistor S3. According to the relationship between the driving signal of the first switching transistor S3 and the driving signal of the second switching transistor S4 (the driving signals of the first switching transistor S3 and the second switching transistor S4 are complementary), generate the driving signal of the second switching transistor S4.

[0153] Finally, after generating the driving signals for controlling the switching transistors in the current-source dual-active-bridge converter, the controller can output the driving signals to the gates of the corresponding switching transistors in the current-source dual-active-bridge converter through the driving circuit, control the on-off of the corresponding switching transistors, realize soft switching within the full load range of the converter, reduce the switching losses, and alleviate the bipolar voltage imbalance problem caused by unbalanced loads.

[0154] The current-source dual-active-bridge converter and control method provided by the above embodiments of the present application have the following advantages: 1. Only a few switching devices are required to automatically achieve bipolar voltage balance. Since there is no need to cascade an additional voltage balancer, by further improving the design of magnetic components, the number of magnetic components can be reduced; 2. Due to the adoption of the topology structure of the current-source dual-active bridge and the corresponding control method, bidirectional power flow and adjustable input voltage can be achieved, which is more flexible in practical applications. And because of the existence of the DC inductor, the input current ripple is small, which is suitable for the connection between energy storage devices and bipolar DC microgrids; 3. By using a reasonable design of the magnetizing inductor, soft switching under extremely unbalanced loads can be achieved, and by using the method of fine-tuning the duty cycle of the secondary switching transistors, the bipolar voltage steady-state error caused by circuit parasitic components can be further reduced. At the same time, the decoupled duty-cycle closed-loop control and phase-shift angle closed-loop control are adopted, and the control is relatively simple.

[0155] The above are the preferred embodiments of the present application. It should be noted that for those of ordinary skill in the art, without departing from the principle described in the present application, several improvements and refinements can be made, and these improvements and refinements should also be regarded as the protection scope of the present application.

Claims

1. A control method for a current-source type dual-active bridge converter with bipolar output, characterized in that, The output ports on the secondary side of the current-source dual-active-bridge converter include a first output port and a second output port, and the control method includes: Collecting the input current on the primary side of the current-source dual-active-bridge converter, the capacitance voltage of the clamping capacitor in the primary side of the current-source dual-active-bridge converter, the target output voltage of the target output port in the secondary side of the current-source dual-active-bridge converter, and the total output voltage of the secondary side of the current-source dual-active-bridge converter; the target output port is the first output port or the second output port; Calculating a first duty cycle of the switching tubes in the primary side of the current-source dual-active-bridge converter according to the capacitance voltage; Calculating the phase-shift angle between the primary and secondary sides of the current-source dual-active-bridge converter according to the total output voltage and the input current; Calculating a second duty cycle of the switching tubes in the secondary side of the current-source dual-active-bridge converter according to the total output voltage and the target output voltage; Generating a driving signal for controlling the switching tubes in the current-source dual-active-bridge converter according to the first duty cycle, the phase-shift angle between the primary and secondary sides, and the second duty cycle; Controlling the on and off of the switching tubes in the current-source dual-active-bridge converter by using the driving signal to balance the voltages of the first output port and the second output port; The current-source dual-active-bridge converter includes a primary side, a secondary side, and a transformer; The secondary side includes a first switching tube, a second switching tube, a first capacitor, a second capacitor, a first load, and a second load; The primary side includes an input power supply, a first DC inductor, a second DC inductor, a leakage inductance, the clamping capacitor, a third switching tube, a fourth switching tube, a fifth switching tube, and a sixth switching tube; The transformer includes a primary winding and a secondary winding; The positive pole of the input power supply is respectively connected to the first end of the first DC inductor and the first end of the second DC inductor, the negative pole of the input power supply is respectively connected to the source of the third switching tube, the source of the fourth switching tube, and the negative pole of the clamping capacitor, the positive pole of the clamping capacitor is respectively connected to the drain of the fifth switching tube and the drain of the sixth switching tube, the first end of the leakage inductance is respectively connected to the drain of the third switching tube, the source of the fifth switching tube, and the second end of the first DC inductor, the second end of the leakage inductance is connected to the positive pole of the primary winding, and the negative pole of the primary winding is respectively connected to the drain of the fourth switching tube, the source of the sixth switching tube, and the second end of the second DC inductor; The positive pole of the secondary winding is respectively connected to the source of the first switching tube and the drain of the second switching tube, the negative pole of the secondary winding is respectively connected to the negative pole of the first capacitor and the positive pole of the second capacitor, the drain of the first switching tube is connected to the positive pole of the first capacitor, and the source of the second switching tube is connected to the negative pole of the second capacitor; The first end of the first load is connected to the positive pole of the first capacitor, the second end of the first load is connected to the negative pole of the first capacitor, and both ends of the first load are the first output port; The first end of the second load is connected to the positive electrode of the second capacitor, and the second end of the second load is connected to the negative electrode of the second capacitor. Both ends of the second load are the second output ports.

2. The control method according to claim 1, characterized in that, Calculating the first duty cycle of the switching tube in the primary side of the current-source dual-active-bridge converter according to the capacitor voltage includes: Calculating the difference between the capacitor voltage and a preset capacitor voltage threshold; Processing the difference between the capacitor voltages by a voltage closed-loop PI regulator to output a first control quantity; Processing the first control quantity output by the voltage closed-loop PI regulator by a limiter to obtain the first duty cycle of the switching tube in the primary side of the current-source dual-active-bridge converter.

3. The control method according to claim 2, characterized in that, The first duty cycle D1 of the switching tube in the primary side of the current-source dual-active-bridge converter satisfies: Among them, V in represents the input voltage of the primary side of the current-source dual-active-bridge converter, K represents the transformer turns ratio of the current-source dual-active-bridge converter, and V o represents the total output voltage of the secondary side of the current-source dual-active-bridge converter, represents the capacitor voltage.

4. The control method according to claim 1, characterized in that, Calculating the primary-secondary phase shift angle of the current-source dual-active-bridge converter according to the total output voltage and the input current includes: Calculating the difference between the total output voltage and a preset total output voltage threshold; Processing the difference between the total output voltages by a voltage closed-loop PI regulator to output a second control quantity; Processing the second control quantity output by the voltage closed-loop PI regulator by a limiter; Calculating the difference between the data output by the limiter and the input current; Processing the current difference by a current closed-loop PI regulator to output a third control quantity; Processing the third control quantity output by the current closed-loop PI regulator by a limiter to obtain the primary-secondary phase shift angle of the current-source dual-active-bridge converter.

5. The control method according to claim 1, characterized in that, Calculating the second duty cycle of the switching tube in the secondary side of the current-source dual-active-bridge converter according to the total output voltage and the target output voltage includes: Calculating the difference between the target output voltage and half of the total output voltage; Processing the voltage difference by a voltage closed-loop PI regulator to output a fourth control quantity; Processing the fourth control quantity output by the voltage closed-loop PI regulator by a limiter to obtain the second duty cycle of the switching tube in the secondary side of the current-source dual-active-bridge converter.

6. The control method according to claim 1, wherein The duty cycles of the third switching tube and the fourth switching tube are both the first duty cycle; the duty cycle of the first switching tube is the second duty cycle; the drive signals of the third switching tube and the fifth switching tube are complementary; the drive signals of the fourth switching tube and the sixth switching tube are complementary; the phase shift between the drive signals of the third switching tube and the fourth switching tube is 180 degrees; the phase shift between the drive signal of the fourth switching tube and the drive signal of the first switching tube is the primary-secondary phase shift angle; the drive signals of the first switching tube and the second switching tube are complementary.

7. The control method according to claim 6, wherein Generating drive signals for controlling the switching tubes in the current-source dual-active-bridge converter according to the first duty cycle, the primary-secondary phase shift angle, and the second duty cycle includes: Generating a drive signal for controlling the third switching tube according to the first duty cycle; Generate the driving signal of the fifth switching tube according to the driving signal of the third switching tube and the complementary relationship between the driving signal of the third switching tube and the driving signal of the fifth switching tube; Generate the driving signal of the fourth switching tube according to the driving signal of the third switching tube and the phase shift between the driving signal of the third switching tube and the driving signal of the fourth switching tube; Generate the driving signal of the sixth switching tube according to the driving signal of the fourth switching tube and the complementary relationship between the driving signal of the fourth switching tube and the driving signal of the sixth switching tube; Generate the driving signal of the first switching tube according to the second duty ratio, the primary-secondary phase shift angle, and the driving signal of the fourth switching tube; Generate the driving signal of the second switching tube according to the driving signal of the first switching tube and the complementary relationship between the driving signal of the first switching tube and the driving signal of the second switching tube.

8. The control method according to claim 1, wherein The inductance value of the exciting inductor in the transformer of the current source type dual active bridge converter is 160 to 320 μH.

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