Control device for an alternating current rotating electric machine and electric power steering device

Abstract

Provided is a control device for an alternating-current rotary electric machine that can reduce noise and suppress an increase in noise of the alternating-current rotary electric machine by taking into account a mechanical resonance period of the alternating-current rotary electric machine, and an electric power steering device. The control device (10) for the alternating-current rotary electric machine switches n zero-phase candidate voltage values in order every switching period (Tv) to calculate a zero-phase voltage value (Vz), turns on and off a switching element based on a voltage command value corrected based on the zero-phase voltage value (Vz), sets a deviation between the zero-phase voltage value (Vz) and a value that is a j multiple of the zero-phase voltage value (Vz) delayed by the switching period (Tv) as a zero-phase time delay deviation value (Vzdif), j is a natural number that minimizes a difference between a half period of a mechanical resonance period (Tm) of the alternating-current rotary electric machine and a j multiple of the switching period (Tv), the n zero-phase candidate voltage values are set in advance so that a root mean square value (Vzdif_rms) of the zero-phase time delay deviation value is smaller than an alternating-current component root mean square value (VzAC_rms) of the zero-phase voltage value.

Description

TECHNICAL FIELD

[0001] The present application relates to a control device for an alternating-current rotary electric machine and an electric power steering device. BACKGROUND

[0002] Patent Document 1 discloses a technology for providing an alternating-current rotary electric machine with less electrical noise, preparing a plurality of carriers of different frequencies, randomly selecting one of the prepared plurality of carriers every prescribed time, and driving the alternating-current rotary electric machine using the selected carrier for PWM control.

[0003] Patent Document 2 discloses a technology as a method for reducing noise (EMI) caused by dithering of a switching frequency, including a step of providing a control device that signals with a motor using a control device including at least one register to control the motor based on an action parameter of the control device, a step of selecting a first clock frequency, a step of selecting a first switching frequency, a step of initializing the at least one register to set at least one determined number, a step of selecting a first jump period, and a step of randomly modulating a switching frequency of pulse modulation based on at least one determined number of the at least one register, the first clock frequency, and the first jump frequency.

[0004] Patent Document 3 discloses a technology including an offset voltage operation section that operates an offset voltage, and a modified three-phase voltage command value operation section that outputs a modified three-phase voltage command value by adding the offset voltage equally to three-phase voltage command values respectively, the offset voltage operation section switching and outputting n offset candidate voltages of different values as the offset voltage at every set time, thereby making the on and off timings of the applied voltage to the three-phase winding vary equally to reduce noise (EMI).

[0005] PRIOR ART DOCUMENTS

[0006] PATENT DOCUMENTS

[0007] Patent Document 1: Japanese Patent Laid-Open No. 2001-346393

[0008] Patent Document 2: Japanese Patent No. 4629938

[0009] Patent Document 3: Japanese Patent No. 6644172 SUMMARY

[0010] PROBLEMS TO BE SOLVED BY THE INVENTION

[0011] The technologies of Patent Document 1 and Patent Document 2 have the following problems (1) and (2).

[0012] (1) In a method of preparing a plurality of frequencies of carriers and selecting 1 from the plurality of frequencies, a plurality of carriers need to be stored, PWM control needs to be performed corresponding to the carriers of the plurality of frequencies, and thus the memory capacity and the processing load increase, making it difficult to install to a cheap microcomputer (CPU).

[0013] (2) It is difficult to install to a microcomputer (CPU) so as to be able to arbitrarily change the carriers.

[0014] Therefore, in Patent Literature 3, for the problems of (1) and (2), instead of changing the PWM control itself, addition of an offset voltage, which is easier to handle, is used to reduce noise (EMI).

[0015] However, in the technology of Patent Literature 3, the characteristics of the mechanical resonance period of the alternating-current rotating electric machine are not taken into consideration, and due to the addition of the random offset voltage, which is sequentially selected from n offset candidate voltages having different values, the component of the mechanical resonance period included in the applied voltage increases, sometimes resulting in an increase in noise of the alternating-current rotating electric machine. In Patent Literature 3, how to set the n offset candidate voltages so that the noise of the alternating-current rotating electric machine does not increase due to the mechanical resonance period of the alternating-current rotating electric machine is not described.

[0016] Therefore, an object of the present application is to provide an alternating-current rotating electric machine control device and an electric power steering device that can reduce noise (EMI) and suppress an increase in noise of an alternating-current rotating electric machine, taking into consideration the mechanical resonance period of the alternating-current rotating electric machine.

[0017] Technical means for solving the technical problem

[0018] The alternating-current rotating electric machine control device according to the present application, which controls an alternating-current rotating electric machine having a stator with a three-phase winding wound therearound and a rotor, via an inverter having a plurality of switching elements, includes:

[0019] a voltage command calculation section that calculates a command voltage vector that represents a three-phase voltage command value applied to the three-phase winding, or a voltage applied to the three-phase winding, in a fixed coordinate system of 2 axes associated with the three-phase winding;

[0020] a zero-phase voltage value calculation section that sequentially switches n (n is a natural number of 2 or more) zero-phase candidate voltage values having different values per switching period and calculates a zero-phase voltage value as the zero-phase candidate voltage value;

[0021] a voltage command correction section that corrects the three-phase voltage command value or the command voltage vector based on the zero-phase voltage value; and

[0022] a PWM control section that performs on-off control of a plurality of switching elements based on the corrected three-phase voltage command value or the corrected command voltage vector obtained by the voltage command correction section,

[0023] The control device of the alternating-current rotary electric machine sets a deviation between the zero-phase voltage value and a time-delayed zero-phase voltage value obtained by delaying the zero-phase voltage value by j times the switching period (j is a natural number of 1 or more) to a zero-phase time-delay deviation value,

[0024] j is a natural number that minimizes a difference between a half period of a mechanical resonance period of the alternating-current rotary electric machine and j times the switching period,

[0025] The control device of the alternating-current rotary electric machine sets the n zero-phase candidate voltage values in advance so that an effective value of the zero-phase time-delay deviation value is smaller than an effective value of an alternating-current component of the zero-phase voltage value.

[0026] The electric power steering device according to the present application includes:

[0027] The control device of the alternating-current rotary electric machine;

[0028] The inverter;

[0029] The alternating-current rotary electric machine; and

[0030] A drive force transmission mechanism that transmits a drive force of the alternating-current rotary electric machine to a steering device of a vehicle,

[0031] A PWM period of the PWM control section is set to 60 μs or less,

[0032] The mechanical resonance period of the alternating-current rotary electric machine is in a range of 200 μs or more and 500 μs or less.

[0033] Effects of the Invention

[0034] By setting the n zero-phase candidate voltage values in advance as described above, a component of the mechanical resonance period included in the zero-phase voltage value and the zero-phase time-delay deviation value can be reduced, and an increase in noise of the alternating-current rotary electric machine caused by resonance can be suppressed. Thus, according to the control device of the alternating-current rotary electric machine and the electric power steering device according to the present application, noise (EMI) can be reduced, and an increase in noise of the alternating-current rotary electric machine can be suppressed. BRIEF DESCRIPTION OF DRAWINGS

[0035] Figure 1 is a schematic configuration diagram of the alternating-current rotary electric machine, the inverter, and the control device according to Embodiment 1.

[0036] Figure 2is a block diagram of the control device according to Embodiment 1.

[0037] Figure 3 is a hardware configuration diagram of the control device according to Embodiment 1.

[0038] Figure 4 is a graph showing a setting example of the zero-phase candidate voltage value n = 80 according to Embodiment 1.

[0039] Figure 5 is a timing chart illustrating PWM control according to Embodiment 1.

[0040] Figure 6 is a timing chart illustrating PWM control after addition of the zero-phase voltage value according to Embodiment 1.

[0041] Figure 7 is a graph showing a frequency analysis result of the applied voltage according to a comparative example in which the zero-phase voltage value is not added.

[0042] Figure 8 is a graph showing a frequency analysis result of the applied voltage in the case of n = 80 according to Embodiment 1.

[0043] Figure 9 is a graph showing a sensitivity characteristic of the noise of the alternating-current rotating electric machine with respect to the winding current according to Embodiment 1.

[0044] Figure 10 is a timing chart of the zero-phase voltage value and the zero-phase time delay deviation value in the case of n = 80 according to Embodiment 1.

[0045] Figure 11 is a graph showing a frequency analysis result of the zero-phase voltage value and the zero-phase time delay deviation value in the case of n = 80 according to Embodiment 1.

[0046] Figure 12 is a timing chart of the zero-phase voltage value and the zero-phase time delay deviation value according to a comparative example.

[0047] Figure 13 is a graph showing a frequency analysis result of the zero-phase voltage value and the zero-phase time delay deviation value according to the comparative example.

[0048] Figure 14 is a graph showing a measured result of the noise level in the case where the setting of the zero-phase voltage value according to Embodiment 1 is changed.

[0049] Figure 15 is a timing chart of the zero-phase voltage value and the zero-phase time delay deviation value in the case of n = 2 according to Embodiment 1.

[0050] Figure 16Fig. 6 is a graph showing the frequency analysis result of the zero-phase voltage value and the zero-phase time delay deviation value in the case where n = 2, which is related to Embodiment 1.

[0051] Figure 17 Fig. 7 is a graph showing the frequency analysis result of the applied voltage in the case where n = 2, which is related to Embodiment 1.

[0052] Figure 18 Fig. 8 is a graph showing a setting example of the zero-phase candidate voltage value in the case where n = 10, which is related to Embodiment 1.

[0053] Figure 19 Fig. 9 is a time chart of the zero-phase voltage value and the zero-phase time delay deviation value in the case where n = 10, which is related to Embodiment 1.

[0054] Figure 20 Fig. 10 is a graph showing the frequency analysis result of the zero-phase voltage value and the zero-phase time delay deviation value in the case where n = 10, which is related to Embodiment 1.

[0055] Figure 21 Fig. 11 is a graph showing the frequency analysis result of the applied voltage in the case where n = 10, which is related to Embodiment 1.

[0056] Figure 22 Fig. 12 is a graph showing a setting example of the zero-phase candidate voltage value in the case where n = 40, which is related to Embodiment 1.

[0057] Figure 23 Fig. 13 is a time chart of the zero-phase voltage value and the zero-phase time delay deviation value in the case where n = 40, which is related to Embodiment 1.

[0058] Figure 24 Fig. 14 is a graph showing the frequency analysis result of the zero-phase voltage value and the zero-phase time delay deviation value in the case where n = 40, which is related to Embodiment 1.

[0059] Figure 25 Fig. 15 is a graph showing the frequency analysis result of the applied voltage in the case where n = 40, which is related to Embodiment 1.

[0060] Figure 26 Fig. 16 is a graph showing the frequency analysis result of the applied voltage in the case where the amplitude is set to twice in n = 10, which is related to Embodiment 1.

[0061] Figure 27 Fig. 17 is a schematic configuration diagram of an alternating-current rotating electric machine, an inverter, and a control device, which is related to Embodiment 2.

[0062] Figure 28 Fig. 18 is a time chart illustrating PWM control and current detection, which is related to Embodiment 2.

[0063] Figure 29Fig. 2 is a diagram showing an equivalent circuit of a series circuit corresponding to one phase in a case where no resistance is provided, according to Embodiment 2.

[0064] Figure 30 Fig. 3 is a diagram showing an equivalent circuit of a series circuit corresponding to one phase in a case where a resistance is provided, according to Embodiment 2.

[0065] Figure 31 Fig. 4 is a diagram showing a measured result of a noise level in a case where a setting of a zero-phase voltage value is changed in a state where a winding current is small, according to Embodiment 2.

[0066] Figure 32 Fig. 5 is a diagram showing a measured result of a noise level in a case where a setting of a zero-phase voltage value is changed in a state where a winding current is large, according to Embodiment 2.

[0067] Figure 33 Fig. 6 is a diagram illustrating voltage vectors according to Embodiment 3.

[0068] Figure 34 Fig. 7 is a diagram illustrating eight conduction-off modes and voltage vectors according to Embodiment 3.

[0069] Figure 35 Fig. 8 is a timing chart illustrating PWM control according to Embodiment 3. DETAILED DESCRIPTION

[0070] 1. Embodiment 1

[0071] A control device 10 (hereinafter referred to as control device 10) according to Embodiment 1 will be described with reference to the accompanying drawings. Figure 1 Fig. 1 is a diagram showing a brief configuration of an alternating-current rotary electric machine 1, an inverter 4, and the control device 10 according to the present embodiment. In the present embodiment, the alternating-current rotary electric machine 1 serves as a driving power source of an electric power steering device 100, and the alternating-current rotary electric machine 1, the inverter 4, and the control device 10 constitute the electric power steering device 100.

[0072] 1-1. Alternating-current rotary electric machine 1

[0073] The alternating-current rotary electric machine 1 has three-phase windings Cu, Cv, and Cw of U phase, V phase, and W phase. The alternating-current rotary electric machine 1 includes a stator and a rotor arranged on a radially inner side of the stator. The stator is wound with the three-phase windings Cu, Cv, and Cw. In the present embodiment, the rotor is provided with a permanent magnet, and a synchronous rotary electric machine of a permanent magnet type is adopted. The three-phase windings can be star-connected or delta-connected.

[0074] The rotor includes a rotation detection circuit 2 for detecting a rotation angle of the rotor. The rotation detection circuit 2 uses a resolver, an encoder, an MR sensor, or the like. An output signal of the rotation detection circuit 2 is input to the control device 10.

[0075] 1-2. Inverter 4

[0076] The inverter 4 is provided with three sets of series circuits (branches) corresponding to the three phases, respectively, and the series circuits are connected in series with a positive-side switching element SP connected to the positive side of the direct-current power supply 3 and a negative-side switching element SN connected to the negative side of the direct-current power supply 3. Then, the connection points of the two switching elements in the series circuit of each phase are connected to the winding of the corresponding phase.

[0077] Specifically, in the series circuit of the U phase, the U-phase positive-side switching element SPu and the U-phase negative-side switching element SNu are connected in series, and the connection point of the two switching elements is connected to the winding Cu of the U phase. In the series circuit of the V phase, the V-phase positive-side switching element SPv and the V-phase negative-side switching element SNv are connected in series, and the connection point of the two switching elements is connected to the winding Cv of the V phase. In the series circuit of the W phase, the W-phase positive-side switching element SPw and the W-phase negative-side switching element SNw are connected in series, and the connection point of the two switching elements is connected to the winding Cw of the W phase. The smoothing capacitor 5 is connected between the positive side and the negative side of the direct-current power supply 3.

[0078] As the switching elements, IGBTs (Insulated Gate Bipolar Transistors) connected in reverse parallel with diodes, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors), bipolar transistors connected in reverse parallel with diodes, or the like are used. The gate terminals of the switching elements are connected to the control device 10 via a gate drive circuit or the like. Each switching element is turned on or off by a switching signal GPu to GNw output from the control device 10.

[0079] The direct-current power supply 3 outputs a direct-current voltage Vdc to the inverter 4. In the present embodiment, the direct-current voltage Vdc is set to 12 V. As the direct-current power supply 3, any device that outputs the direct-current voltage Vdc, such as a battery, a DC-DC converter, a diode rectifier, a PWM rectifier, or the like, can be used. The direct-current power supply 3 is provided with a voltage sensor that detects the direct-current voltage Vdc, and the output signal of the voltage sensor can be input to the control device 10. The control device 10 can perform control using the detected direct-current voltage Vdc.

[0080] 1-3. Electric Power Steering Apparatus 100

[0081] The electric power steering system 100 includes a control device 10 for an AC rotary motor, an inverter 4, an AC rotary motor 1, and a drive force transmission mechanism 101 that transmits the driving force of the AC rotary motor 1 to the steering system 102 of the vehicle.

[0082] The rotor shaft of the AC rotary motor 1 is connected to the steering device 102 of the wheel 103 via a drive force transmission mechanism 101. For example, the electric power steering device 100 includes a steering wheel 104 that the driver rotates left and right, a shaft 105 connected to the steering wheel 104 and transmitting the steering torque of the steering wheel 104 to the wheel 103 via the steering device 102, a torque sensor 106 mounted on the shaft 105 and detecting the steering torque Ts of the steering wheel 104, and a worm gear mechanism connecting the rotor shaft of the AC rotary motor 1 to the shaft 105, etc., as part of the drive force transmission mechanism 101. The output signal of the torque sensor 106 is input to the control device 10 (input circuit 92).

[0083] 1-4. Control device 10

[0084] The control device 10 controls the AC rotating motor 1 via the inverter 4. For example... Figure 2 As shown, the control device 10 includes a rotation detection unit 31, a voltage command calculation unit 32, a zero-phase voltage value calculation unit 33, a voltage command correction unit 34, and a PWM control unit 35. Each function of the control device 10 is implemented by the processing circuitry provided by the control device 10. Specifically, the control device 10 is as follows... Figure 3 As shown, the processing circuit includes: an arithmetic processing unit 90 (computer) such as a CPU (Central Processing Unit); a storage device 91 for exchanging data with the arithmetic processing unit 90; an input circuit 92 for inputting external signals to the arithmetic processing unit 90; and an output circuit 93 for outputting signals from the arithmetic processing unit 90 to the outside.

[0085] As the arithmetic processing device 90, an ASIC (Application Specific Integrated Circuit), an IC (Integrated Circuit), a DSP (Digital Signal Processor), an FPGA (Field Programmable Gate Array), various logic circuits, various signal processing circuits, and the like can be provided. Further, as the arithmetic processing device 90, a plurality of arithmetic processing devices of the same kind or different kinds can be provided to share the execution of each processing. As the storage device 91, a RAM (Random Access Memory) configured to be able to read and write data from and to the arithmetic processing device 90, a ROM (Read Only Memory) configured to be able to read data from the arithmetic processing device 90, and the like can be provided. The input circuit 92 is connected to the rotation detection circuit 2, various sensors such as the torque sensor 106, switches, and the like, and is provided with an A / D converter and the like that input output signals of these sensors and switches to the arithmetic processing device 90. The output circuit 93 is connected to electric loads such as a gate drive circuit that performs on-off driving of a switching element, and is provided with a drive circuit and the like that outputs control signals from the arithmetic processing device 90 to these electric loads.

[0086] Further, each function of each control section 31 to 35 and the like provided in the control device 10 is realized by the arithmetic processing device 90 executing software (program) stored in the storage device 91 such as a ROM in cooperation with other hardware of the control device 10 such as the storage device 91, the input circuit 92, and the output circuit 93. In addition, n zero-phase candidate voltage values Vzc_1 to Vzc_n, switching period Tv, and the like used by each control section 31 to 35 and the like are stored in the storage device 91 such as a ROM as a part of the software (program). Hereinafter, each function of the control device 10 will be described in detail.

[0087] <Rotation detection section 31>

[0088] The rotation detection section 31 detects the magnetic pole position θ of the rotor (the rotation angle θ of the rotor) and the rotation angular velocity ω on the electrical angle. In the present embodiment, the rotation detection section 31 detects the magnetic pole position θ of the rotor (the rotation angle θ) and the rotation angular velocity ω on the basis of the output signal of the rotation detection circuit 2. The magnetic pole position θ is set to the direction of the N pole of the permanent magnet provided on the rotor. In addition, the rotation detection section 31 can be configured to estimate the rotation angle (magnetic pole position) on the basis of current information and the like obtained by superimposing a high-order harmonic component on a current command value, without using a rotation sensor (so-called sensorless system).

[0089] Voltage command calculation section 32

[0090] Voltage command calculation section 32 calculates three-phase voltage command values Vub, Vvb, Vwb applied to the three-phase winding.

[0091] In the present embodiment, voltage command calculation section 32 detects the steering torque Ts of the driver based on the output signal of torque sensor 106. Voltage command calculation section 32 calculates the three-phase voltage command values Vub, Vvb, Vwb based on the steering torque Ts and the magnetic pole position θ. Details are described below.

[0092] Voltage command calculation section 32 sets the current command value Iqo of the q-axis based on the steering torque Ts as shown in the following equation, and sets the current command value Ido of the d-axis to 0.

[0093] Iqo = Ka x Ts

[0094] Ido = 0 ··· (1-1)

[0095] Here, Ka is a constant, but can also vary depending on the steering torque Ts and the running speed of the vehicle, etc. In addition, the current command value Iqo of the q-axis can be set based on a known compensation control corresponding to the steering condition. Furthermore, the current command values Ido, Iqo of the d-axis and the q-axis can be determined by a known vector control method such as maximum torque current control, field weakening control, etc. In addition, the d-axis is determined in the direction of the magnetic pole position θ (N-pole), and the q-axis is determined in the direction that advances 90 degrees in the electrical angle from the d-axis.

[0096] Voltage command calculation section 32 converts the current command values Ido, Iqo of the d-axis and the q-axis into the voltage command value Vdo of the d-axis and the voltage command value Vqo of the q-axis based on the specifications of alternating current rotating electric machine 1 (resistance value R of the winding, d-axis inductance Ld, q-axis inductance Lq, and the magnetic flux linkage ψ of the permanent magnet) and the rotational angular velocity ω as shown in the following equation.

[0097] Vdo = R x Ido - ω x Lq x Iqo

[0098] Vqo = R x Iqo - ω x (Ld x Ido + ψ) ··· (1-2)

[0099] Then, voltage command calculation section 32 performs fixed coordinate conversion and two-phase three-phase conversion on the voltage command values Vdo, Vqo of the d-axis and the q-axis based on the magnetic pole position θ and converts into the three-phase voltage command values Vub, Vvb, Vwb as shown in the following equation.

[0100] [Mathematical expression 1]

[0101]

[0102] In addition, the voltage command calculation section 32 can apply a known modulation such as superposition of a third harmonic to the three-phase voltage command values Vub, Vvb, Vwb.

[0103] <Zero-phase voltage value calculation section 33>

[0104] The zero-phase voltage value calculation section 33 calculates n (n is a natural number of 2 or more) zero-phase candidate voltage values Vzc_l, Vzc_2,..., Vzc_n, which have different values, in order by each switching period Tv and as a zero-phase voltage value Vz. In addition, the set values of the n zero-phase candidate voltage values Vzc_l to Vzc_n are described later.

[0105] For example, Figure 4 A set example of 80 zero-phase candidate voltage values Vzc_l to Vzc_80 in the case where n = 80 is shown in Table 1. The zero-phase voltage value calculation section 33 refers to a zero-phase candidate voltage value table data in which 80 zero-phase candidate voltage values Vzc_l to Vzc_80 are set in advance, selects the first zero-phase candidate voltage value Vzc_l to the 80th zero-phase candidate voltage value Vzc_80 in order in each switching period Tv, and calculates the selected zero-phase candidate voltage value Vzc as a zero-phase voltage value Vz. In addition, after the 80th zero-phase candidate voltage value Vzc_80, the first zero-phase candidate voltage value Vzc_l is selected. Thus, n times the value of the switching period Tv becomes one round in which the n zero-phase candidate voltage values Vzc_l to Vzc_n are set as the zero-phase voltage value Vz for the one round period Tvn (= n x Tv).

[0106] <Voltage command correction section 34>

[0107] The voltage command correction section 34 corrects the three-phase voltage command values Vub, Vvb, Vwb based on the zero-phase voltage value Vz. In the present embodiment, the voltage command correction section 34 adds the zero-phase voltage value Vz to each of the three-phase voltage command values Vub, Vvb, Vwb to calculate corrected three-phase voltage command values Vuc, Vvc, Vwc as shown in the following equations.

[0108] Vuc = Vub + Vz

[0109] Vvc = Vvb + Vz... (1-4)

[0110] Vwc = Vwb + Vz

[0111] <PWM control section 35>

[0112] The PWM control unit 35 performs on / off control on multiple switching elements of the inverter 4 based on the corrected three-phase voltage command values ​​Vuc, Vvc, and Vwc obtained by the voltage command correction unit 34. The PWM control unit 35 compares the carrier wave CA oscillating in the PWM period Tc with each of the corrected three-phase voltage command values ​​Vuc, Vvc, and Vwc, and performs on / off control on multiple switching elements based on the comparison results.

[0113] like Figure 5 As shown, the carrier CA is a triangular wave that oscillates around 0 with an amplitude of half the DC voltage, Vdc / 2, during the PWM period Tc. For each phase, when the carrier CA is lower than the voltage command value, the PWM control unit 35 turns on the switching signal GP of the positive-side switching element (1 in this example) and turns on the positive-side switching element. When the carrier CA exceeds the voltage command value, the PWM control unit 35 turns off the switching signal GP of the positive-side switching element (0 in this example) and turns off the positive-side switching element. On the other hand, for each phase, when the carrier CA is lower than the voltage command value, the PWM control unit 35 turns off the switching signal GN of the negative-side switching element (0 in this example) and turns off the negative-side switching element. When the negative-side switching element is turned off and the carrier CA exceeds the voltage command value, the PWM control unit 35 turns on the switching signal GN of the negative-side switching element (1 in this example) and turns on the negative-side switching element. In addition, for each phase, a short-circuit prevention period (dead time) can be set between the conduction period of the switching element on the positive side and the conduction period of the switching element on the negative side to turn off both the switching elements on the positive and negative sides.

[0114] <Example of PWM control after summing zero-phase voltage values ​​Vz>

[0115] Figure 6 The image shows an example of PWM control after summing the zero-phase voltage values ​​Vz. The switching period Tv of the zero-phase voltage value Vz is set to be the same as the PWM period Tc, and the zero-phase voltage value Vz is switched at the timing of the peak of the trough of the carrier wave CA. The switching period Tv can be set differently from the PWM period Tc. For example, the switching period Tv can be set to a natural multiple of the PWM period Tc, or it can be set to a natural multiple of half a period of the PWM period Tc.

[0116] For each switching cycle Tv, the zero-phase voltage value Vz is set sequentially in the order of the first zero-phase candidate voltage value Vzc_1, the second zero-phase candidate voltage value Vzc_2, the third zero-phase candidate voltage value Vzc_3, the fourth zero-phase candidate voltage value Vzc_4, ... Figure 6In this case, for the sake of simplicity, the three-phase voltage command values Vub, Vvb, Vwb are set to 0. Therefore, the three-phase voltage command values Vuc, Vvc, Vwc after the addition of the zero-phase voltage value Vz all become the same value.

[0117] As described above, the three-phase voltage command values Vuc, Vvc, Vwc after the addition are compared with the carrier CA, and switching signals of the positive side and the negative side of the three phases are generated. As a result, in the case where the switching signal of the positive side of the three phases is on, the direct-current voltage Vdc is applied to the three-phase winding, and in the case where the switching element of the positive side of the three phases is off, 0 V is applied to the three-phase winding. Figure 6 In this case, the applied voltages Vu_PWM, Vv_PWM, Vw_PWM applied to the three-phase winding are shown. Regardless of whether the three-phase voltage command values Vub, Vvb, Vwb are constant values or not, the periods Δton_1, Δton_2, Δton_3 during which the applied voltage Vu_PWM, Vv_PWM, Vw_PWM of each phase is switched from 0 V to the direct-current voltage Vdc with respect to the reference timing tb_1, tb_2, tb_3 of the vertex of the trough of the carrier CA, and the periods Δtoff_1, Δtoff_2, Δtoff_3 during which the applied voltage Vu_PWM, Vv_PWM, Vw_PWM of each phase is switched from the direct-current voltage Vdc to 0 V, vary equally in the three phases by adding the zero-phase voltage value Vz.

[0118] <Harmonic Component Overlapped to Applied Voltage Based on PWM Control>

[0119] By the PWM control, the applied voltages Vu_PWM, Vv_PWM, Vw_PWM of the three-phase winding become a rectangular wave of the PWM period Tc, and contain a large number of harmonic components in addition to the components of the three-phase voltage command values originally intended to be applied. Due to the harmonic components, noise (EMI) is conducted or radiated from the inverter 4 to the peripheral equipment, and becomes a main cause of malfunction of the peripheral equipment. Therefore, it is preferable to reduce the harmonic components contained in the applied voltages Vu_PWM, Vv_PWM, Vw_PWM of the three-phase winding as much as possible.

[0120] Figure 7 is a frequency analysis result (analysis frequency range: 150 kHz to 3000 kHz) of the applied voltage Vu_PWM of the U-phase winding in the case where the zero-phase voltage value Vz is always set to 0, that is, in the case where the zero-phase voltage value Vz is not added. In the same figure, the dotted line is the level of the harmonic component connecting the levels of the components. In order to reduce the noise generated by the inverter 4, it is necessary to reduce the level of this high-frequency component.

[0121] <Reduction of Harmonic Component Caused by Addition of Zero-Phase Voltage Value Vz>

[0122] Figure 8 In the figure, the case where the zero-phase voltage value Vz is added to the 80 zero-phase candidate voltage values based on the Figure 4 frequency analysis result of the time voltage Vu_PWM of the U-phase winding in the case where the zero-phase voltage value Vz is added is shown. In addition, Tc = Tv = 50 μs is set. Figure 8 In the figure, the level of the harmonic component in the case where the zero-phase voltage value Vz shown is not added is shown by a broken line. According to the figure, by the addition of the zero-phase voltage value Vz, the on and off timings of the applied voltage vary equally in all phases, and therefore, in a wide frequency band of 150 kHz to 3000 kHz, the harmonic component is reduced, and the noise (EMI) generated by the inverter 4 can be reduced. Figure 7

[0123] <Zero-phase voltage value Vz containing mechanical resonance component and necessity of reduction>

[0124] In order to reduce the noise generated by the alternating-current rotating electric machine 1 including the inverter 4, it is important to reduce as much as possible the component of the mechanical resonance period Tm of the alternating-current rotating electric machine 1 contained in the zero-phase voltage value Vz. In particular, in the electric power steering apparatus 100, quietness is required, and therefore, this point is very important.

[0125] Here, the reason why the noise is generated in the alternating-current rotating electric machine 1 by the addition of the zero-phase voltage value Vz for noise reduction is described. The zero-phase voltage value Vz is added equally to the three-phase voltage command values Vub, Vvb, Vwb in the three phases, and therefore, it corresponds to the "zero-phase voltage" in the symmetrical coordinate method. Therefore, if the alternating-current rotating electric machine 1 is three-phase three-wire type, by the addition of the zero-phase voltage value Vz, the line voltage between the two phase voltages does not change, and the three-phase winding currents Iu, Iv, Iw do not change. However, in reality, due to the difference in the on-resistance between the switching elements on the positive side and the switching elements on the negative side of the inverter 4, the voltage difference between the drain and the source (in the case of MOSFET), and the voltage difference between the collector and the emitter (in the case of IGBT and bipolar transistor), by the addition of the zero-phase voltage value Vz, the three-phase winding currents Iu, Iv, Iw change. Thereby, depending on the frequency band, sometimes the noise is generated from the alternating-current rotating electric machine 1.

[0126] Figure 9 is a measurement result of the sensitivity characteristic of the noise with respect to the winding current measured in the alternating-current rotating electric machine 1 for the electric power steering apparatus. The greater the gain of the vertical axis, the greater the noise of the alternating-current rotating electric machine 1 with respect to the amplitude of the noise component of the current. In this alternating-current rotating electric machine 1, the peak value exists near 2500 Hz. This is because the mechanical resonance frequency of the frame of the alternating-current rotating electric machine 1 is near 2500 Hz (the resonance period Tm is 400 μs). ​

[0127] In addition, the mechanical resonance of the alternating-current rotary electric machine 1 occurs in the frame of the alternating-current rotary electric machine 1, or in a power source box in which the alternating-current rotary electric machine 1, the inverter 4, and the control device 10 are integrated, or in a power plant including the frame or the power source box and a gear mechanism, and the like.

[0128] Among the various frequencies included in the variation component of the winding current generated by the addition of the zero-phase voltage value Vz, the frequency component close to the mechanical resonance frequency of the alternating-current rotary electric machine 1 easily becomes the noise of the alternating-current rotary electric machine 1. Thus, it is desirable to reduce the component of the mechanical resonance period Tm of the alternating-current rotary electric machine 1 included in the zero-phase voltage value Vz.

[0129] <Setting of n zero-phase candidate voltage values satisfying the idea A>

[0130] Next, a method of setting n zero-phase candidate voltage values for reducing the component of the mechanical resonance period Tm included in the zero-phase voltage value Vz will be described.

[0131] The setting of the n zero-phase candidate voltage values of the present application is based on the following idea A.

[0132] Idea A:

[0133] As shown in the following equation, n zero-phase candidate voltage values Vzc_1 to Vzc_n are set in advance so that the root mean square value Vzdif_rms of the zero-phase time delay deviation value Vzdif is smaller than the root mean square value VzAC_rms of the alternating-current component VzAC of the zero-phase voltage value Vz.

[0134] VzAC_rms > Vzdif_rms ··· (1-5)

[0135] Here, the alternating-current component VzAC of the zero-phase voltage value Vz becomes a value obtained by subtracting the direct-current component (average value of one cycle Tvn (= n x Tv) of the zero-phase voltage value Vz) from the zero-phase voltage value Vz, as shown in the following equation. The root mean square value VzAC_rms of the alternating-current component VzAC of the zero-phase voltage value becomes the square root of a value obtained by averaging the square value of the alternating-current component VzAC of the zero-phase voltage value in one cycle Tvn, as shown in the following equation. Here, t is time.

[0136] [Mathematical equation 2]

[0137]

[0138]

[0139] Furthermore, the zero-phase time delay deviation value Vzdif, as shown in the following formula, is the deviation between the zero-phase voltage value Vz and the zero-phase voltage value after a time delay of j times the switching period Tv (j is a natural number greater than or equal to 1). Here, j is set to a natural number that minimizes the difference between half-period Tm / 2 of the mechanical resonance period of the AC rotating motor and j times the switching period Tv.

[0140] [Mathematical Expression 3]

[0141] Vzdif(t)=Vz(t)-Vz(tj×Tv)…1-7)

[0142] The effective value of the zero-phase time delay deviation value Vzdif, Vzdif_rms, is shown in the following formula. It is the square root of the value obtained by averaging the squares of the zero-phase time delay deviation value Vzdif over one cycle Tvn.

[0143] [Mathematical Expression 4]

[0144]

[0145] The above is idea A.

[0146] <Example of setting n zero-phase candidate voltage values>

[0147] Next, as Figure 9 As shown, when the mechanical resonant period frequency of the AC rotating motor 1 is 2500Hz (mechanical resonant period Tm = 400μs) and the switching period of the zero-phase voltage value Vz is Tv = 100μs, the j that minimizes the difference between the half-period Tm / 2 of the resonant period and the j-fold value of the switching period Tv is j = 2. Figure 10 The upper part shows the sequential output during the switching cycle Tv. Figure 4 The zero-phase voltage value Vz(t) is obtained from the 80 zero-phase candidate voltage values ​​shown. Figure 10 The lower part is shown in Equation (1-7), which shows the zero-phase time delay deviation value Vzdif(t) obtained by subtracting a multiple of the zero-phase voltage value Vz(t) from the zero-phase voltage value Vz(t) after delaying the switching period Tv by a factor of j (in this example, 100μs×2=200μs).

[0148] If we use equation (1-6) to calculate Figure 10 The effective value of the AC component of the zero-phase voltage Vz(t) in the upper half is VzAC_rms, then VzAC_rms = 0.3445V. On the other hand, if equation (1-8) is used to calculate... Figure 10the effective value Vzdif_rms of the zero-phase time delay deviation value Vzdif(t) of the lower half of FIG. 9 is Vzdif_rms = 0.329 V. Thus, as in Expression (1-5), 80 zero-phase candidate voltage values are set so that VzAC_rms > Vzdif_rms, satisfying the above Idea A.

[0149] Figure 11 The upper half of FIG. 10 shows the frequency analysis result of the zero-phase voltage value Vz of the upper half of FIG. 9, Figure 10 The upper half of FIG. 10 shows the frequency analysis result of the zero-phase voltage value Vz of the upper half of FIG. 9, Figure 11 The lower half of FIG. 10 shows the frequency analysis result of the zero-phase time delay deviation value Vzdif of the lower half of FIG. 9. In the case where Idea A is satisfied, as shown in Figure 10 The lower half of FIG. 10 shows the frequency analysis result of the zero-phase time delay deviation value Vzdif of the lower half of FIG. 9. In the case where Idea A is satisfied, as shown in Figure 11 the components of the mechanical resonance frequency (around 2500 Hz) included in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif become small. Thus, the components of the mechanical resonance period Tm of the alternating-current rotating electric machine 1 included in the zero-phase voltage value Vz can be reduced, and the noise of the alternating-current rotating electric machine 1 can be reduced.

[0150] Next, as a setting example of n zero-phase candidate voltage values that do not satisfy Idea A, the seven zero-phase candidate voltage values (Vzc_1 = -0.75 V, Vzc_2 = -0.5 V, Vzc_3 = 0 V, Vzc_4 = 0.75 V, Vzc_5 = 0.5 V, Vzc_6 = -0.25 V, Vzc_7 = 0.25 V) described in Patent Literature 3 (paragraphs 0036 to 0038) are used for explanation.

[0151] Figure 12 The upper half of FIG. 11 shows the zero-phase voltage value Vz(t) in the case where the seven zero-phase candidate voltage values of Patent Literature 3 are used, Figure 12 The lower half of FIG. 11 shows the zero-phase time delay deviation value Vzdif(t) obtained by subtracting the time delay zero-phase voltage value Vz(t - 200 μs) obtained by delaying the zero-phase voltage value Vz(t) by 200 μs from the zero-phase voltage value Vz(t). If the effective value VzAC_rms of the alternating-current component of the zero-phase voltage value Vz(t) of Patent Literature 3 of the upper half of FIG. 11 is calculated, Figure 12 The lower half of FIG. 11 shows the zero-phase time delay deviation value Vzdif(t) obtained by subtracting the time delay zero-phase voltage value Vz(t - 200 μs) obtained by delaying the zero-phase voltage value Vz(t) by 200 μs from the zero-phase voltage value Vz(t). If the effective value VzAC_rms of the alternating-current component of the zero-phase voltage value Vz(t) of Patent Literature 3 of the upper half of FIG. 11 is calculated, Figure 12 The effective value Vzdif_rms of the zero-phase time delay deviation value Vzdif(t) of Patent Literature 3 of the lower half of FIG. 11 is Vzdif_rms = 0.7789 V. Thus, VzAC_rms < Vzdif_rms, and Idea A is not satisfied.

[0152] Figure 13 The upper half of FIG. 12 shows the frequency analysis result of the zero-phase voltage value Vz of Patent Literature 3 of the upper half of FIG. 11, Figure 12 The upper half of FIG. 12 shows the frequency analysis result of the zero-phase voltage value Vz of Patent Literature 3 of the upper half of FIG. 11,Figure 11 The lower half shows Figure 12 The frequency analysis results of the zero-phase time delay deviation value Vzdif from the lower half of patent document 3. In cases where idea A is not satisfied, such as... Figure 13 As shown, the mechanical resonant frequency component (around 2500Hz) contained in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif increases. Consequently, the mechanical resonant period Tm component of the AC rotary motor 1 contained in the zero-phase voltage value Vz is larger, and the noise of the AC rotary motor 1 increases.

[0153] Figure 14 In, it is shown that has Figure 9 The measured results of the noise level in the AC rotary motor 1 used in the electric power steering system with the sensitivity characteristics changed under the condition of changing the setting of the zero phase voltage value Vz. Figure 14 The left side represents the case where the zero-phase voltage value Vz(t) = 0. Figure 14 The central case is based on the zero-phase voltage value Vz(t) of the seven zero-phase candidate voltage values ​​in Patent Document 3. Figure 14 The right side is based on Figure 4 The case of the zero-phase voltage value Vz(t) of the 80 zero-phase candidate voltage values. Figure 14 In the case of Patent Document 3, which does not satisfy idea A in the center, the noise deteriorates compared to the case where the zero-phase voltage value Vz = 0 on the left (especially at 2.8 kHz, close to the mechanical resonant frequency). On the other hand, in Figure 14 In the case of this application satisfying idea A on the right, the noise is reduced overall compared to the case of zero-phase voltage value Vz = 0 on the left, and the noise does not increase but decreases even at the mechanical resonant frequency.

[0154] Therefore, by setting n zero-phase candidate voltage values ​​to satisfy idea A, and adding the zero-phase voltage value Vz to the three-phase voltage command value, the noise of the AC rotating motor 1 can be reduced in many frequency bands, including the mechanical resonant frequency band of the AC rotating motor 1.

[0155] <Example 1 of variations of n zero-phase candidate voltage values>

[0156] A variation of n candidate zero-phase voltage values ​​is explained. The n candidate zero-phase voltage values ​​are set to satisfy idea A, and are not limited to this. Figure 4 The 80 zero-phase candidate voltage values ​​are provided below. Specific examples are given below.

[0157] The following explanation is provided for the case where n=2, the first zero-phase candidate voltage value Vzc_1=-0.5V, and the second zero-phase candidate voltage value Vzc_1=0.5V. Figure 15The upper part shows the zero-phase voltage value Vz(t) after outputting two zero-phase candidate voltage values ​​Vzc_1 and Vzc_2 sequentially according to the switching period Tv = 100μs. Figure 16 The lower half shows the zero-phase time delay deviation value Vzdif(t) obtained by setting j=2 in order to satisfy idea A and subtracting a time delay zero-phase voltage value Vz(t-200μs) that delays the switching period Tv of the zero-phase voltage value Vz(t) (in this example, 100μs×2=200μs) from the zero-phase voltage value Vz(t).

[0158] Figure 15 The effective value of the AC component of the zero-phase voltage Vz(t) in the upper half is VzAC_rms = 0.500V. Figure 15 The effective value of the zero-phase time delay deviation value Vzdif(t) in the lower half is Vzdif_rms=0V, VzAC_rms>Vzdif_rms, which satisfies idea A.

[0159] Figure 16 The upper part shows Figure 15 The frequency analysis results of the zero-phase voltage value Vz in the upper half, Figure 16 The lower half shows Figure 15 The frequency analysis results of the zero-phase time delay deviation value Vzdif in the lower half. Under the condition of satisfying idea A, such as... Figure 16 As shown, the mechanical resonant frequency component (around 2500Hz) contained in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif becomes smaller. Therefore, the component of the mechanical resonant period Tm of the AC rotary motor 1 contained in the zero-phase voltage value Vz can be reduced, thereby reducing the noise of the AC rotary motor 1.

[0160] Figure 17 The table shows the frequency analysis results of the time voltage Vu_PWM of the U-phase winding with the zero-phase voltage value Vz based on two zero-phase candidate voltage values. Figure 17 In the middle, dashed lines indicate the absence of [something]. Figure 7 The diagram shows the harmonic component levels under the zero-phase voltage value Vz. According to this diagram, by adding the zero-phase voltage values ​​Vz, the on-time and off-time of the applied voltage vary equally throughout the entire phase. Therefore, harmonic components are reduced in a large frequency band from 150kHz to 3000kHz, thus reducing the noise generated by inverter 4. Since idea A is satisfied, the noise of AC rotating motor 1 can be reduced in many frequency bands, including the mechanical resonant frequency band of AC rotating motor 1.

[0161] <Example 2 of the variation of n zero-phase candidate voltage values>

[0162] Figure 18In the middle, a setting example of the 10 zero-phase candidate voltage values Vzc_l to Vzc_10 in the case of n = 10 is shown. The zero-phase voltage value calculation part 33 selects the 1st zero-phase candidate voltage value Vzc_l to the 10th zero-phase candidate voltage value Vzc_10 in order in each switching period Tv, and calculates the selected zero-phase candidate voltage value Vzc as the zero-phase voltage value Vz. Further, after the 10th zero-phase candidate voltage value Vzc_10, the 1st zero-phase candidate voltage value Vzc_l is selected.

[0163] Figure 19 The upper half of FIG. 12 shows the zero-phase voltage value Vz(t) after the 10 zero-phase candidate voltage values Vzc_l to Vzc_10 are output in order in the switching period Tv = 100 μs. Figure 19 The lower half of FIG. 12 shows the zero-phase time delay deviation value Vzdif(t) obtained by setting j = 2 in order to satisfy the idea A, and subtracting the value obtained by delaying the zero-phase voltage value Vz(t) by j times the switching period Tv (in this example, 100 μs x 2 = 200 μs) from the zero-phase voltage value Vz(t).

[0164] Figure 19 The effective value VzAC_rms of the alternating component of the zero-phase voltage value Vz(t) of the upper half of FIG. 12 is 5.07 V, Figure 19 The effective value Vzdif_rms of the zero-phase time delay deviation value Vzdif(t) of the lower half of FIG. 12 is 2.53 V, and VzAC_rms > Vzdif_rms, so the idea A is satisfied.

[0165] Figure 20 The upper half of FIG. 13 shows Figure 19 The frequency analysis result of the zero-phase voltage value Vz of the upper half of FIG. 13, Figure 20 The lower half of FIG. 13 shows Figure 19 The frequency analysis result of the zero-phase time delay deviation value Vzdif of the lower half of FIG. 13. In the case where the idea A is satisfied, as shown in Figure 20 the components of the mechanical resonance frequency (around 2500 Hz) included in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif become small. Thus, the components of the mechanical resonance period Tm of the alternating-current rotating electrical machine 1 included in the zero-phase voltage value Vz can be reduced, and the noise of the alternating-current rotating electrical machine 1 can be reduced.

[0166] Figure 21 In FIG. 14, the frequency analysis result of the time voltage Vu_PWM of the U-phase winding in the case where the zero-phase voltage value Vz based on the 10 zero-phase candidate voltage values is added is shown. Figure 21 In FIG. 15, the frequency analysis result of the time voltage Vu_PWM of the U-phase winding in the case where the zero-phase voltage value Vz based on the 10 zero-phase candidate voltage values is not added is shown by a broken line. Figure 7The diagram shows the harmonic component levels under the zero-phase voltage value Vz. According to this diagram, by adding the zero-phase voltage values ​​Vz, the on-time and off-time of the applied voltage vary equally throughout the entire phase. Therefore, harmonic components are reduced in a large frequency band from 150kHz to 3000kHz, thus reducing the noise generated by inverter 4. Furthermore, since idea A is satisfied, the noise of AC rotating motor 1 can be reduced in many frequency bands, including the mechanical resonant frequency band of AC rotating motor 1.

[0167] <Example 3 of variations of n zero-phase candidate voltage values>

[0168] Figure 22 The example shown illustrates a configuration of 40 candidate zero-phase voltage values ​​Vzc_1 to Vzc_40 when n=40. In each switching cycle Tv, the zero-phase voltage calculation unit 33 sequentially selects the first candidate zero-phase voltage value Vzc_1 to the 40th candidate zero-phase voltage value Vzc_40, and calculates the selected candidate zero-phase voltage value Vzc as the zero-phase voltage value Vz. Furthermore, after the 40th candidate zero-phase voltage value Vzc_40, the first candidate zero-phase voltage value Vzc_1 is selected.

[0169] Figure 23 The upper part shows the zero-phase voltage value Vz(t) after outputting 40 zero-phase candidate voltage values ​​Vzc_1 to Vzc_40 sequentially according to the switching period Tv = 100μs. Figure 23 The lower half shows the zero-phase time delay deviation value Vzdif(t) obtained by setting j=2 in order to satisfy idea A and subtracting a time delay zero-phase voltage value Vz(t-200μs) that delays the switching period Tv of the zero-phase voltage value Vz(t) (in this example, 100μs×2=200μs) from the zero-phase voltage value Vz(t).

[0170] Figure 23 The effective value of the AC component of the zero-phase voltage Vz(t) in the upper half is VzAC_rms = 3.94V. Figure 23 The effective value of the zero-phase time delay deviation value Vzdif(t) in the lower half is Vzdif_rms = 3.18V, and VzAC_rms > Vzdif_rms, which satisfies idea A.

[0171] Figure 24 The upper part shows Figure 23 The frequency analysis results of the zero-phase voltage value Vz in the upper half, Figure 24 The lower half shows Figure 23 The frequency analysis results of the zero-phase time delay deviation value Vzdif in the lower half. Under the condition of satisfying idea A, such as... Figure 24As shown, the component of the mechanical resonance frequency (around 2500 Hz) included in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif is reduced. Thus, the component of the mechanical resonance period Tm of the AC rotating electrical machine 1 included in the zero-phase voltage value Vz is reduced, and the noise of the AC rotating electrical machine 1 is reduced.

[0172] Figure 25 In the case of the zero-phase voltage value Vz, the frequency analysis result of the applied voltage Vu_PWM of the U-phase winding is shown. Figure 25 In the case of the zero-phase voltage value Vz, the frequency analysis result of the applied voltage Vu_PWM of the U-phase winding is shown. Figure 7 As shown, the component of the mechanical resonance frequency (around 2500 Hz) included in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif is reduced. Thus, the component of the mechanical resonance period Tm of the AC rotating electrical machine 1 included in the zero-phase voltage value Vz is reduced, and the noise of the AC rotating electrical machine 1 is reduced.

[0173] <SUMMARY>

[0174] The above describes a plurality of specific examples. The present application aims at reducing the noise (EMI) generated by the inverter 4 and reducing the noise of the AC rotating electrical machine 1. To this end, in the present application, by setting n zero-phase candidate voltage values so as to satisfy the idea A, the component of the mechanical resonance period Tm of the AC rotating electrical machine 1 included in the zero-phase voltage value Vz and the zero-phase time delay deviation value Vzdif is reduced, and the noise of the AC rotating electrical machine 1 is reduced. According to the plurality of specific examples, the larger n is, the more the component of 150 kHz to 3000 kHz included in the applied voltage of the winding tends to be reduced.

[0175] Thus, in the case of implementing the present application, first, n zero-phase candidate voltage values are determined so as to satisfy the idea A, and the component of the mechanical resonance period Tm of the AC rotating electrical machine 1 included in the zero-phase voltage value Vz is reduced. On this basis, the component of 150 kHz to 3000 kHz included in the applied voltage of the winding satisfies the specification, and in the case of not satisfying the specification, n is increased and it is confirmed again whether the specification is satisfied. However, n needs to be at least 2.

[0176] <Adjustment of the amplitudes of the n zero-phase candidate voltage values>

[0177] Further, in addition to increasing n, it is also effective to adjust the amplitudes of the n zero-phase candidate voltage values. Figure 26 In the case of the zero-phase voltage value Vz, the frequency analysis result of the applied voltage Vu_PWM of the U-phase winding is shown. Figure 18Frequency analysis results of the applied voltage Vu_PWM of the U-phase winding in the case where the 10 zero-phase candidate voltage values are all set to 2 times for use. The frequency analysis results of the applied voltage Vu_PWM of the U-phase winding in the case where the 10 zero-phase candidate voltage values are not set to 2 times Figure 21 In contrast, the 10 zero-phase candidate voltage values are set to 2 times Figure 26 In the case where the 10 zero-phase candidate voltage values are set to 2 times, the harmonic components around approximately 300 kHz to 500 kHz are reduced. Thus, adjusting the amplitudes of the n zero-phase candidate voltage values is effective for adjusting the noise level.

[0178] As a general standard, in the case where it is desired to sufficiently reduce the LW band (150 kHz to 500 kHz), the n zero-phase candidate voltage values can be set in advance such that the difference between the maximum value among the n zero-phase candidate voltage values and the minimum value among the n zero-phase candidate voltage values is 10% or more of the direct-current voltage Vdc supplied to the inverter.

[0179] In the case where it is desired to sufficiently reduce the AM band (500 kHz to 1730 kHz), the n zero-phase candidate voltage values can be set in advance such that the difference between the maximum value among the n zero-phase candidate voltage values and the minimum value among the n zero-phase candidate voltage values is 5% or more of the direct-current voltage Vdc supplied to the inverter.

[0180] <Setting of each period>

[0181] The mechanical resonance period Tm of the alternating-current rotary electric machine 1 for an electric power steering device according to the present application is in the range of 200 μs or more and 500 μs or less (the mechanical resonance frequency is 2 kHz or more and 5 kHz or less). More preferably, the mechanical resonance period Tm is in the range of 300 μs or more and 400 μs or less (the mechanical resonance frequency is 2.5 kHz or more and 3.3 kHz or less). Further, in the case where the PWM period Tc is 60 μs or less and 6.67 μs or more, the electric power steering device 100 is suitable for reducing noise and reducing the noise of the alternating-current rotary electric machine 1. If the PWM period Tc is 60 μs or less and 6.67 μs or more, the PWM frequency (1 / Tc) is 16 kHz or more and 150 kHz or less, and in the case where it is 150 kHz or more, the noise reduction effect is reduced, and in the case where it is less than 16 kHz, the sound of the PWM frequency component of the alternating-current rotary electric machine 1 is noticeable, and thus the noise in the audible range of humans is deteriorated.

[0182] 2. Embodiment 2

[0183] The AC rotating electric machine 1, the inverter 4, and the control device 10 according to Embodiment 2 will be described. The same structure as that of Embodiment 1 will be omitted from the description. The basic structure of the AC rotating electric machine 1, the inverter 4, and the control device 10 according to the present embodiment is the same as that of Embodiment 1, but the inverter 4 is provided with a current sensor, and the control device 10 sets the voltage command value based on the current detection, which is different from Embodiment 1. Figure 27 FIG. 1 is a schematic configuration diagram of the AC rotating electric machine 1, the inverter 4, and the control device 10 according to the present embodiment.

[0184] The three-phase series circuit of the inverter 4 has the resistance Ru of the U phase, the resistance Rv of the V phase, and the resistance Rw of the W phase connected in series with the switching element SN on the negative side of each phase. The resistances Ru, Rv, and Rw of each phase are connected in series with the negative side of the switching element SN on the negative side. In the present embodiment, the three-phase resistances Ru, Rv, and Rw are provided as shunt resistances for current detection, and the potential difference across each phase resistance is detected by the amplifiers 21, 22, and 23, and is input to the control device 10. Thus, the inverter 4 of the present embodiment is a so-called lower arm 3 shunt type inverter.

[0185] In addition, the resistances of each phase can be connected in series with the switching element SP on the positive side. Alternatively, the bus bar on the positive side or the bus bar on the negative side connecting the inverter 4 and the DC power supply 3 can be connected in series with a resistance.

[0186] In the present embodiment, the voltage command calculation section 32 detects the currents Iud, Ivd, and Iwd flowing through the windings of each phase based on the potential difference across the resistances Ru, Rv, and Rw of each phase. Then, the voltage command calculation section 32 performs three-phase to two-phase conversion and rotational coordinate conversion on the three-phase current detection values Iud, Ivd, and Iwd based on the magnetic pole position θ, and converts them into the d-axis current detection value Idd and the q-axis current detection value Iqd, as shown in the following equation.

[0187] [Equation 5]

[0188]

[0189] In addition, in the case where a resistance is connected in series to the bus bar on the positive side or the bus bar on the negative side connecting the inverter 4 and the DC power supply 3, the voltage command calculation section 32 detects the bus bar current based on the potential difference across the resistance, and detects the winding currents Iud, Ivd, and Iwd of each phase based on the on-off pattern of the switching elements of each phase at the time when the bus bar current is detected and the bus bar current, using a publicly known method.

[0190] As with Embodiment 1, the voltage command calculation section 32 detects the steering torque Ts of the driver on the basis of the output signal of the torque sensor 106. The voltage command calculation section 32 sets the current command value Iqo of the q-axis on the basis of the steering torque Ts as shown in Expression (1-1), and sets the current command value Ido of the d-axis to 0.

[0191] Then, the voltage command calculation section 32 changes the three-phase voltage command values Vub, Vvb, Vwb so that the current detection values Idd, Iqd of the d-axis and the q-axis approach the current command values Ido, Iqo of the d-axis and the q-axis. The voltage command calculation section 32 changes the voltage command value Vdo of the d-axis so that the current detection value Idd of the d-axis approaches the current command value Ido of the d-axis by PI control as shown in the following expression, and changes the voltage command value Vqo of the q-axis so that the current detection value Iqd of the q-axis approaches the current command value Iqo of the q-axis by PI control. In addition, feedforward control for non-interference of the d-axis current and the q-axis current or the like can be performed.

[0192] [Expression 6]

[0193]

[0194]

[0195] Here, Kd, Kq are proportional gains, Td, Tq are integral time constants, and s is a Laplace operator.

[0196] Then, as with Embodiment 1, the voltage command calculation section 32 performs fixed coordinate conversion and two-phase three-phase conversion on the voltage command values Vdo, Vqo of the d-axis and the q-axis on the basis of the magnetic pole position θ as shown in Expression (1-3), and converts them into the three-phase voltage command values Vub, Vvb, Vwb.

[0197] As with Embodiment 1, the zero-phase voltage value calculation section 33 calculates the zero-phase voltage value Vz by switching the n (n is a natural number of 2 or more) zero-phase candidate voltage values Vzc_1 to Vzc_n, which have different values, in order for each switching period Tv. The n zero-phase candidate voltage values Vzc_1 to Vzc_n are set in advance so as to satisfy the idea A.

[0198] As with Embodiment 1, the voltage command correction section 34 corrects the three-phase voltage command values Vub, Vvb, Vwb on the basis of the zero-phase voltage value Vz. The voltage command correction section 34 adds the zero-phase voltage value Vz to the three-phase voltage command values Vub, Vvb, Vwb, respectively, as shown in Expression (1-4), to calculate the corrected three-phase voltage command values Vuc, Vvc, Vwc.

[0199] As with Embodiment 1, the PWM control section 35 performs on-off control of the plurality of switching elements of the inverter 4 based on the corrected three-phase voltage command values Vuc, Vvc, Vwc.

[0200] <Current detection timing>

[0201] The current detection timing based on the potential difference across the resistance is described. In the series circuit of the U phase, when the switching element SNu on the negative side is on, current flows through the resistance Ru of the U phase, and the current flowing through the resistance is equal to the current Iu flowing through the winding Cu of the U phase. Thus, if the switching element SNu on the negative side of the U phase is on, the detection of the winding current Iu of the U phase can be performed at an arbitrary timing. Similarly, if the switching element SNv on the negative side of the V phase is on, the detection of the winding current Iv of the V phase can be performed at an arbitrary timing. If the switching element SNw on the negative side of the W phase is on, the detection of the winding current Iw of the W phase can be performed at an arbitrary timing.

[0202] However, in the present embodiment, as shown in Figure 28 the voltage command calculation section 32 detects the currents Iud, Ivd, Iwd flowing through the windings of the respective phases at the timing of the peak of the wave of the carrier based on the potential difference across the resistances Ru, Rv, Rw of the respective phases. In addition, in the case where the resistances Ru, Rv, Rw of the respective phases are connected in series with the switching elements on the positive side, the voltage command calculation section 32 is configured to detect the currents Iud, Ivd, Iwd flowing through the windings of the respective phases at the timing of the peak of the wave of the carrier based on the potential difference across the resistances Ru, Rv, Rw of the respective phases.

[0203] Hereinafter, the effects thereof are described. Figure 28 The control operation corresponding to 3 periods of the PWM period Tc is shown. As with Embodiment 1, the switching period Tv of the zero-phase voltage value Vz is set to be the same as the PWM period Tc, and the zero-phase voltage value Vz is switched at the timing of the peak of the wave of the carrier CA. Figure 28 In the present embodiment, the 1st zero-phase candidate voltage value Vzc_1, the 2nd zero-phase candidate voltage value Vzc_2, and the 3rd zero-phase candidate voltage value Vzc_3 are sequentially set to the zero-phase voltage value Vz for each switching period Tv. The three-phase voltage command values Vub, Vvb, Vwb are set to different values. Thus, the three-phase voltage command values Vuc, Vvc, Vwc after the addition of the zero-phase voltage value Vz become different values, and vary according to the change in the zero-phase voltage value Vz for each switching period Tv.

[0204] Figure 28In the drawing, the current Iu flowing through the winding of the U phase, the current Iv flowing through the winding of the V phase, and the current Iw flowing through the winding of the W phase are shown. Further, the moving average Iu_ave, Iv_ave, Iw_ave of the currents Iu, Iv, Iw of the respective phases over the PWM period Tc is shown. The winding currents Iu, Iv, Iw of the respective phases include a ripple component with respect to the average Iu_ave, Iv_ave, Iw_ave of the winding currents of the respective phases. However, at the peak of the crest of the carrier CA (or the peak of the trough), the winding currents Iu, Iv, Iw of the respective phases coincide with the average Iu_ave, Iv_ave, Iw_ave of the winding currents of the respective phases. Thus, by setting the current detection timing to the peak of the crest of the carrier CA, it is possible to detect the fundamental component of the rotation period from the winding current including the ripple component.

[0205] In particular, in the control of the AC rotary electric machine 1 for the electric power steering apparatus, it is required to detect the winding current with high precision. If an error occurs in the detected value of the winding current with respect to the true value of the winding current, the control is performed so that the detected value of the winding current having the error with respect to the true value coincides with the current command value, and torque pulsation occurs in the output torque of the AC rotary electric machine 1. The torque pulsation is transmitted to the steering wheel 104 via the shaft 105, and the steering feeling of the driver deteriorates.

[0206] By setting the current detection timing to the peak of the crest of the carrier CA, it is possible to detect the fundamental component of the rotation period from the winding current including the ripple component. Thus, it is possible to detect the winding current with high precision, to suppress the occurrence of torque pulsation caused by the current detection error, and to obtain a good steering feeling of the steering wheel of the driver.

[0207] Next, the advantages of connecting the resistance R in series to the negative side and setting the n zero-phase candidate voltage values according to the idea A will be described.

[0208] Effect on winding current when resistance R is not provided

[0209] Figure 29 In the drawing, the equivalent circuit of the series circuit of the inverter 4 corresponding to one phase when the resistance R is not provided is shown. The positive-side switching element SP and the negative-side switching element SN are represented by switches, and the on-resistance is ignored. Between the positive-side switching element SP and the negative-side switching element SN, there is an output terminal connected to the winding of the corresponding phase, and the potential Vout of the output terminal becomes the applied voltage of the winding.

[0210] Here, if the ratio of the on-period of the positive-side switching element SP in the PWM period Tc is D, the average Vout_ave of the potential Vout of the output terminal in the PWM period Tc can be represented by the following equation.

[0211] Vout_ave = D x Vdc... (2-3)

[0212] Similarly, if the proportion of the on period of the switching element SPu on the positive side of the U phase is set as Du, and the proportion of the on period of the switching element SPv on the positive side of the V phase is set as Dv, the average value Voutu_ave of the output terminal potential of the series circuit of the U phase and the average value Voutv_ave of the output terminal potential of the series circuit of the V phase can be expressed by the following equations.

[0213] Voutu_ave = Du x Vdc

[0214] Voutv_ave = Dv x Vdc... (2-4)

[0215] Here, in the case where the zero phase voltage value Vz is added to the state of equation (2-4), if the variation amount of the proportion of the on period of each phase resulting from the addition of the zero phase voltage value Vz is set as ΔD, it can be expressed by the following equation.

[0216] Voutu_ave = (Du + ΔD) x Vdc

[0217] Voutv_ave = (Dv + ΔD) x Vdc... (2-5)

[0218] According to equation (2-5), the line-to-line voltage ΔVout_uv between the U phase and the V phase can be expressed by the following equation.

[0219] ΔVout_uv = Voutu_ave - Voutv_ave

[0220] = (Du - Dv) x Vdc... (2-6)

[0221] From this equation, it is understood that no influence resulting from the addition of the zero phase voltage value Vz is generated in the line-to-line voltage. In the case where no resistance R is provided, the winding current of the alternating current rotating electric machine 1 flows based on the line-to-line voltage, and therefore the influence of the zero phase voltage value Vz is not reflected in the winding current.

[0222] <Influence on Winding Current in the Case Where Resistance R is Provided>

[0223] Figure 30 In FIG. 6, an equivalent circuit of the series circuit of the inverter 4 corresponding to one phase in the case where the resistance R is provided is shown. Except for the resistance R connected in series to the negative side of the switching element SN, the equivalent circuit is the same as that shown in FIG. 5. Figure 29

[0224] ​The average value Vout_ave of the potential Vout of the output terminal in the PWM period Tc can be expressed by the following equation. Here, R denotes the resistance value of the resistance R.

[0225] Vout_ave = D x Vdc + (1 - D) x (-R x I)

[0226] (2-7)

[0227] Similarly, if the proportion of the on period of the switching element SPu on the positive side of the U phase is set as Du, the proportion of the on period of the switching element SPv on the positive side of the V phase is set as Dv, the winding current of the U phase is set as Iu, and the winding current of the V phase is set as Iv, the average value Voutu_ave of the output terminal potential of the series circuit of the U phase and the average value Voutv_ave of the output terminal potential of the series circuit of the V phase can be expressed by the following equations.

[0228] Voutu_ave = Du x Vdc + (1 - Du) x (-R x Iu)

[0229] Voutv_ave = Dv x Vdc + (1 - Dv) x (-R x Iv)

[0230] (2-8)

[0231] Here, in the case where the zero-phase voltage value Vz is added to the state of equation (2-8), if the variation amount of the proportion of the on period of each phase resulting from the addition of the zero-phase voltage value Vz is set as ΔD, it can be expressed by the following equation.

[0232] Voutu_ave = (Du + ΔD) x Vdc + (1 - (Du + ΔD)) x (-R x Iu)

[0233] Voutv_ave = (Dv + ΔD) x Vdc + (1 - (Dv + ΔD)) x (-R x Iv)

[0234] (2-9)

[0235] According to equation (2-9), the line-to-line voltage ΔVout_uv between the U phase and the V phase can be expressed by the following equation.

[0236] ΔVout_uv = Voutu_ave - Voutv_ave

[0237] = (Du - Dv) x Vdc + ΔD x R x (Iu - Iv)

[0238] (2-10)

[0239] Compared to equation (2-6) without resistor R, equation (2-10) with resistor R has "ΔD×R×(Iu-Iv)" added to the second term on the right. Therefore, with resistor R, the zero-phase voltage Vz is added to the line voltage, and this effect is reflected in the winding current.

[0240] Therefore, in inverters with a lower arm 3-shunting method as in this embodiment, or inverters with an upper arm 3-shunting method, or inverters with a bus 1-shunting method, the effect of the sum of the zero-phase voltage values ​​Vz is reflected in the line voltage and winding current in inverters where the resistance of the switching element SN on the negative side is different from the resistance of the switching element SP on the positive side.

[0241] <Noise reduction based on idea A>

[0242] The following illustrates the situation where, by setting resistor R in inverter 4, even when the variation in winding current increases due to the addition of zero-phase voltage value Vz, the noise of AC rotating motor 1 can be reduced by satisfying the addition of zero-phase voltage value Vz of idea A.

[0243] Figure 31 and Figure 32 The image shows an inverter with a 3-splitter lower arm according to this embodiment, which has... Figure 9 The measured results of the noise level in the AC rotary motor 1 used in the electric power steering system with the sensitivity characteristics of the motor were obtained under the condition that the zero phase voltage value Vz was changed. Figure 31 This is the case where the effective value of the winding current is 10 Arms. Figure 32 This is the case where the effective value of the winding current is 70 Arms.

[0244] Figure 31 and Figure 32 The left side represents the case where the zero-phase voltage value Vz(t) = 0. Figure 31 and Figure 32 The central case is based on the zero-phase voltage value Vz(t) of the seven zero-phase candidate voltage values ​​in Patent Document 3. Figure 31 and Figure 32 The right side is based on Figure 4 The case of the zero-phase voltage value Vz(t) of the 80 zero-phase candidate voltage values. Figure 31 and Figure 32 In the case of the central non-satisfaction of idea A in patent document 3, the noise deteriorates compared to the case where the zero-phase voltage value Vz = 0 on the left (especially close to the mechanical resonant frequency of 2.8 kHz), and the effective value of the winding current is smaller. Figure 31 In comparison, the effective value of the winding current is larger. Figure 32 The noise level worsens even more.

[0245] On the other hand, in Figure 31 and Figure 32 In the case of the present application satisfying the idea A on the right side, the noise is reduced as a whole compared to the case of the zero-phase voltage value Vz = 0 on the left side, and the noise does not increase but decrease even at the mechanical resonance frequency. Thus, the noise reduction caused by the addition of the zero-phase voltage value Vz satisfying the idea A exceeds the noise increase caused by the setting of the resistance R to the inverter 4 and the addition of the zero-phase voltage value Vz. That is, as in the inverter of the lower arm 3 shunt method, the upper arm 3 shunt method, and the bus 1 shunt method, even in the case of the setting of the resistance R to the inverter 4, by setting n zero-phase candidate voltage values so as to satisfy the idea A and adding the zero-phase voltage value Vz to the three-phase voltage command value, the noise of the alternating-current rotating electric machine 1 can be reduced in many frequency bands including the mechanical resonance band of the alternating-current rotating electric machine 1.

[0246] 3. Embodiment 3

[0247] The alternating-current rotating electric machine 1, the inverter 4, and the control device 10 according to Embodiment 3 will be described. The same structure parts as those of Embodiment 1 described above will be omitted from the description. The basic structure of the alternating-current rotating electric machine 1, the inverter 4, and the control device 10 according to the present embodiment is the same as that of Embodiments 1 and 2, but the control device 10 sets a command voltage vector, corrects the command voltage vector based on the zero-phase voltage value Vz, and performs the on-off control of the switching elements based on the corrected command voltage vector, which is different from Embodiments 1 and 2.

[0248] < Voltage command calculation unit 32 >

[0249] In the present embodiment, the voltage command calculation unit 32 calculates a command voltage command Vo obtained after the voltage applied to the windings of the three phases is expressed by a 2-axis fixed coordinate system associated with the windings of the three phases. In the present embodiment, as shown in FIG. 2, the 2-axis fixed coordinate system is composed of an α-axis determined in the direction of the winding Cu of the U phase and a β-axis determined in the direction advanced by 90° in electrical angle with respect to the α-axis. The pole position θ is the angle of the d-axis with respect to the α-axis. Figure 33

[0250] The voltage command calculation unit 32 calculates the voltage command value Vdo of the d-axis and the voltage command value Vqo of the q-axis using the same method as that of Embodiments 1 or 2.

[0251] ​In this embodiment, the voltage command calculation section 32 performs fixed coordinate conversion on the d-axis voltage command value Vdo and the q-axis voltage command value Vqo based on the magnetic pole position θ to convert them into the α-axis voltage command value Vα and the β-axis voltage command value Vβ as shown in the following equation. The voltage vector represented by the α-axis voltage command value Vα and the β-axis voltage command value Vβ becomes the command voltage vector Vo.

[0252] [Equation 7]

[0253]

[0254] The voltage command calculation section 32 determines four conduction and non-conduction patterns of the plurality of switching elements of the inverter 4 set between the PWM periods Tc based on the command voltage vector Vo, and determines the periods of the four conduction and non-conduction patterns in the PWM period Tc. The determined four conduction and non-conduction patterns include two conduction and non-conduction patterns corresponding to the zero voltage vector. This process is the same as the known space vector PWM.

[0255] As shown in Figure 34 , there are eight conduction and non-conduction patterns of the plurality of switching elements of the inverter 4 that can be set. Here, "1" indicates that the switching element is on, and "0" indicates that the switching element is off. The eight conduction and non-conduction patterns correspond to the eight basic voltage vectors V0 to V7 shown in Figure 33 .

[0256] The 0th basic voltage vector V0 is a zero voltage vector. That is, in the 0th basic voltage vector V0, the three-phase positive-side switching elements SPu, SPv, and SPw are all off, and the three-phase negative-side switching elements SNu, SNv, and SNw are all on, and thus the direct-current voltage Vdc is not applied to the three-phase windings. The 7th basic voltage vector V7 is a zero voltage vector. That is, in the 7th basic voltage vector V7, the three-phase positive-side switching elements SPu, SPv, and SPw are all on, and the three-phase negative-side switching elements SNu, SNv, and SNw are all off, and thus the direct-current voltage Vdc is not applied to the three-phase windings.

[0257] The 1st basic voltage vector V1 becomes a vector in the direction of the U-phase winding, the 2nd basic voltage vector V2 becomes a vector in the opposite direction of the W-phase winding, the 3rd basic voltage vector V3 becomes a vector in the direction of the V-phase winding, the 4th basic voltage vector V4 becomes a vector in the opposite direction of the U-phase winding, the 5th basic voltage vector V5 becomes a vector in the direction of the W-phase winding, and the 6th basic voltage vector V6 becomes a vector in the opposite direction of the V-phase winding.

[0258] The voltage command calculation section 32 decides two fundamental voltage vectors (hereinafter, referred to as approaching fundamental voltage vectors Vn1, Vn2) that approach the command voltage vector Vo, from the six fundamental voltage vectors V1 to V6 excluding the 0th and 7th fundamental voltage vectors V0, V7 that are zero voltage vectors. Then, the voltage command calculation section 32 decides the 0th and 7th fundamental voltage vectors V0, V7 as four fundamental voltage vectors (hereinafter, referred to as set fundamental voltage vectors) set between the PWM periods Tc, from the two approaching fundamental voltage vectors Vn1, Vn2 decided.

[0259] Figure 33 In the example of FIG. 3, the 1st and 2nd fundamental voltage vectors V1, V2 that approach the command voltage vector Vo are decided as the two approaching fundamental voltage vectors Vn1, Vn2.

[0260] Then, the voltage command calculation section 32 decomposes the command voltage vector Vo into components Vn1c, Vn2c of the two approaching fundamental voltage vectors Vn1, Vn2. Then, the voltage command calculation section 32 calculates the ratios Dn1, Dn2 of the set periods of the two approaching fundamental voltage vectors Vn1, Vn2 in the PWM period Tc, based on the magnitudes of the components Vn1c, Vn2c of the two approaching fundamental voltage vectors Vn1, Vn2 with respect to the DC voltage Vdc, respectively, as shown in the following equation.

[0261] [Equation 8]

[0262]

[0263]

[0264] Figure 33 In the example of FIG. 3, the magnitudes of the components Vn1c, Vn2c of the two approaching fundamental voltage vectors Vn1, Vn2 are calculated based on the voltage command value Vα of the α-axis and the voltage command value Vβ of the β-axis, as shown in the following equation.

[0265] [Equation 9]

[0266]

[0267]

[0268] Then, the voltage command calculation section 32 sets so that the ratios D0, D7 of the set periods of the 0th and 7th fundamental voltage vectors V0, V7 that are zero voltage vectors in the PWM period Tc satisfy the following equation.

[0269] D0 + D7 = 1 - (Dn1 + Dn2) ··· (3-4)

[0270] That is, the periods other than the set periods of the two approximate basic voltage vectors Vn1, Vn2 in the PWM period Tc can be arbitrarily allocated to the zero voltage vectors, i.e., V0, V7 to be set. For example, the ratios D0, D7 of the set periods of V0, V7 are set to be equal as follows.

[0271] D0 = D7 = {1 - (Dn1 + Dn2)} / 2 ··· (3-5)

[0272] In addition, the periods after the ratios Dn1, Dn2, D0, D7 of the set periods of the four set basic voltage vectors are multiplied by the PWM period Tc become the set periods of the four set basic voltage vectors.

[0273] <Zero Phase Voltage Value Calculation Section 33>

[0274] As with Embodiment 1, the zero phase voltage value calculation section 33 switches the n (n is a natural number of 2 or more) zero phase candidate voltage values Vzc_1 to Vzc_n having different values in order by each switching period Tv and calculates as the zero phase voltage value Vz. The n zero phase candidate voltage values Vzc_1 to Vzc_n are set in advance so as to satisfy the Idea A.

[0275] <Voltage Command Correction Section 34>

[0276] In the present embodiment, the voltage command correction section 34 corrects the command voltage vector Vo based on the zero phase voltage value Vz. The voltage command correction section 34 changes the periods of the two on-off modes corresponding to the zero voltage vectors (in the present example, the set periods of the 0th and 7th basic voltage vectors V0, V7) according to the zero phase voltage value Vz so that the two total periods do not change.

[0277] The voltage command correction section 34 calculates the zero phase voltage correction period ratio Dz by dividing the zero phase voltage value Vz by the direct current voltage Vdc as follows.

[0278] Dz = Vz / Vdc ··· (3-6)

[0279] Then, the voltage command correction section 34 calculates the ratio D0c of the set period of the 0th basic voltage vector Vo after correction by adding the zero phase voltage correction period ratio Dz to the ratio D0 of the set period of the 0th basic voltage vector Vo as shown below and calculates the ratio D7c of the set period of the 7th basic voltage vector V7 after correction by subtracting the zero phase voltage correction period ratio Dz from the ratio D7 of the set period of the 7th basic voltage vector V7.

[0280] D0c = D0 + Dz

[0281] D7c = D7 - Dz ··· (3-7)

[0282] In addition, the ratios D0c, D7c of the set periods of the corrected 0th and 7th basic voltage vectors are multiplied by the period Tc after the PWM period, and the periods become the set periods of the 0th and 7th basic voltage vectors, respectively. The ratios Dn1, Dn2 of the set periods of the two approximate basic voltage vectors Vn1, Vn2 do not change due to the zero-phase voltage value Vz.

[0283] PWM control section 35

[0284] The PWM control section 35 performs on-off control of the plurality of switching elements based on the four on-off patterns (two approximate basic voltage vectors Vn1, Vn2 and 0th and 7th basic voltage vectors V0, V7 in this example) determined by the voltage command calculation section 32, and the ratios (Dn1, Dn2, D0c, D7c in this example) of the periods of the four on-off patterns determined and corrected by the voltage command calculation section 32 and the voltage command correction section 34.

[0285] The PWM control section 35 generates the switching signals GPu to GNw of the respective switching elements based on the four set basic voltage vectors (Vn1, Vn2, V0, V7) set in the PWM period Tc and the ratios (Dn1, Dn2, D0c, D7c) of the set periods of the four set basic voltage vectors in the PWM period Tc. For example, as shown in FIG. 6, Figure 33 Figure 35 The switching signals GPu to GNw of the respective switching elements are generated.

[0286] Here, the voltage after the switching signals GPu, GPv, GPw of the switching elements on the positive side of each phase are multiplied by the direct-current voltage Vdc corresponds to the applied voltage Vu_PWM, Vv_PWM, Vw_PWM of each phase winding.

[0287] ​Thus, by the zero-phase voltage value Vz, only the setting periods of the 0th and 7th basic voltage vectors, which are zero voltage vectors, vary, and the setting periods of the two proximate basic voltage vectors Vnl, Vn2, which are not zero voltage vectors, do not vary, and thus the average value of the applied voltage to the winding in the PWM period Tc does not vary. Thus, by the correction of the zero-phase voltage value Vz, the average value of the line-to-line voltage in the PWM period Tc does not vary, and thus the winding current does not vary. On the other hand, by the zero-phase voltage value Vz, the setting periods of the 0th and 7th basic voltage vectors, which are zero voltage vectors, are varied, and thus the on and off timings of the applied voltages Vu_PWM, Vv_PWM, Vw_PWM of the respective phase windings are equally varied in all phases. Thus, as in Embodiments 1 and 2, in a wide frequency band of 150 kHz to 3,000 kHz, the harmonic components of the applied voltage of the respective phase windings can be reduced, and the noise generated by the inverter 4 can be reduced. Furthermore, the zero-phase voltage value Vz satisfies the idea A, and thus the noise of the alternating-current rotating electrical machine 1 can be reduced in a wide frequency band including the mechanical resonance frequency band of the alternating-current rotating electrical machine 1.

[0288] [Other Embodiments]

[0289] Finally, other embodiments of the present application will be described. In addition, the structures of the embodiments described below are not limited to being applied individually, and can be applied in combination with the structures of other embodiments, as long as no contradiction arises.

[0290] (1) The zero-phase voltage value calculating section 33 can vary the magnitude of the zero-phase voltage value Vz in accordance with the amplitude of the applied voltage to the three-phase winding. For example, in Embodiments 1 and 2, the amplitude of the applied voltage to the three-phase winding is set to the amplitude of the fundamental component of the three-phase voltage command values Vub, Vvb, Vwb.

[0291] In Embodiment 3, the amplitude of the applied voltage to the three-phase winding is set to the magnitude of the command voltage vector Vo.

[0292] Then, the zero-phase voltage value calculating section 33 sets the zero-phase voltage value Vz by setting the voltage value, which is set by sequentially switching the n zero-phase candidate voltage values Vzc_l to Vzc_n for each switching period Tv, multiplied by a gain in accordance with the amplitude of the applied voltage to the three-phase winding. For example, the zero-phase voltage value calculating section 33 decreases the gain as the amplitude of the applied voltage to the three-phase winding increases. Furthermore, the zero-phase voltage value calculating section 33 decreases the gain as the rotational speed of the alternating-current rotating electrical machine 1 increases.

[0293] (2) The zero-phase voltage value calculation section 33 can vary the magnitude of the zero-phase voltage value Vz in accordance with the direct-current voltage Vdc. The zero-phase voltage value calculation section 33 sets the gain in accordance with the direct-current voltage Vdc, multiplies the voltage value set by sequentially switching the n zero-phase candidate voltage values Vzc_l to Vzc_n for each switching period Tv by the gain, and sets the zero-phase voltage value Vz. For example, the zero-phase voltage value calculation section 33 increases the gain as the direct-current voltage Vdc increases.

[0294] (3) The alternating-current rotary electric machine 1 can be a rotary electric machine provided with three-phase windings, and for example, can be a synchronous rotary electric machine of an excitation winding type in which a rotor is provided with an electromagnet, or an induction motor in which a rotor is not provided with a permanent magnet. In addition, the alternating-current rotary electric machine 1 can also be a rotary electric machine provided with a plurality of sets of three-phase windings. The structure of the present application can be applied to each set of three-phase windings.

[0295] (4) In the embodiment 2, a current sensor can be provided on the electric wire connecting the inverter 4 and the three-phase windings.

[0296] (5) The alternating-current rotary electric machine 1 can be a driving force source of various devices other than the electric power steering device 100. For example, the alternating-current rotary electric machine 1 can be provided as a driving force source of a wheel.

[0297] Although various exemplary embodiments and examples are described in the present application, the various features, modes, and functions described in one or more embodiments are not limited to the application of the specific embodiments, and can be applied to the embodiments individually or in various combinations. Therefore, it can be considered that an infinite number of modifications not exemplified are also included in the technical scope disclosed in the present application specification. For example, cases in which at least one constituent element is modified, added, or omitted, and cases in which at least one constituent element is extracted and combined with the constituent elements of other embodiments are included.

[0298] Explanation of Reference Signs

[0299] 1 alternating-current rotary electric machine

[0300] 4 inverter

[0301] 10 control device of alternating-current rotary electric machine

[0302] 32 voltage command calculation section

[0303] 33 zero-phase voltage value calculation section

[0304] 34 voltage command correction section

[0305] 35 PWM control section

[0306] 100 electric power steering device

[0307] CA carrier wave

[0308] R resistance

[0309] Tc PWM period

[0310] Tm resonance period

[0311] Tv switching period

[0312] Vo command voltage vector

[0313] Vz zero-phase voltage value

[0314] VzAC AC component of the zero-phase voltage value

[0315] VzAC_rms RMS value of the AC component of the zero-phase voltage value

[0316] Vzc zero-phase candidate voltage value

[0317] Vzdif zero-phase time delay deviation value

[0318] Vzdif_rms RMS value of the zero-phase time delay deviation value.

Claims

1. A control device of an alternating-current rotary electric machine that controls an alternating-current rotary electric machine having a stator with a three-phase winding wound therearound and a rotor, via an inverter having a plurality of switching elements, characterized by comprising: a voltage command calculation section that calculates a command voltage vector that represents three-phase voltage command values applied to the three-phase winding, or a voltage applied to the three-phase winding, in a fixed coordinate system of 2-axes associated with the three-phase winding; a zero-phase voltage value calculation section that calculates, as a zero-phase voltage value, n zero-phase candidate voltage values that are sequentially switched in each switching period with different values, where n is a natural number of 2 or more; a voltage command correction section that corrects the three-phase voltage command values or the command voltage vector based on the zero-phase voltage value; and a PWM control section that performs on-off control of the plurality of switching elements based on the corrected three-phase voltage command values or the corrected command voltage vector obtained by the voltage command correction section, wherein the control device of the alternating-current rotary electric machine sets a deviation between the zero-phase voltage value and a time-delayed zero-phase voltage value obtained by delaying the zero-phase voltage value by j times the switching period, where j is a natural number of 1 or more, as a zero-phase time-delay deviation value, j is a natural number that minimizes a difference between a half period of a mechanical resonance period of the alternating-current rotary electric machine and j times the switching period, and the control device of the alternating-current rotary electric machine sets the n zero-phase candidate voltage values in advance so that an effective value of the zero-phase time-delay deviation value is smaller than an effective value of an alternating-current component of the zero-phase voltage value.

2. The control device of the alternating-current rotary electric machine according to claim 1, wherein the voltage command calculation section calculates the three-phase voltage command values, the voltage command correction section adds the zero-phase voltage value to each of the three-phase voltage command values to calculate the corrected three-phase voltage command values, and the PWM control section compares a carrier that vibrates in a PWM period with each of the corrected three-phase voltage command values, and performs on-off control of the plurality of switching elements based on a comparison result.

3. The control device of the alternating-current rotary electric machine according to claim 1, wherein the voltage command calculation section calculates the command voltage vector, determines four on-off modes of the plurality of switching elements including two on-off modes corresponding to a zero voltage vector, set between PWM periods, based on the command voltage vector, and determines periods of the four on-off modes in the PWM periods, and the voltage command correction section changes periods of the two on-off modes corresponding to the zero voltage vector according to the zero-phase voltage value so that a total of two periods does not change. ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ The PWM control section performs on-off control of the plurality of switching elements based on the four on-off patterns determined by the voltage command calculation section and the four on-off patterns determined and corrected by the voltage command calculation section and the voltage command correction section.

4. The control device of an alternating-current rotary electric machine according to any one of claims 1 to 3, characterized in that The n zero-phase candidate voltage values are set in advance so that a difference between a maximum value within the n zero-phase candidate voltage values and a minimum value within the n zero-phase candidate voltage values is 5% or more of a direct-current voltage supplied to the inverter.

5. The control device of an alternating-current rotary electric machine according to any one of claims 1 to 3, characterized in that The n zero-phase candidate voltage values are set in advance so that a difference between a maximum value within the n zero-phase candidate voltage values and a minimum value within the n zero-phase candidate voltage values is 10% or more of a direct-current voltage supplied to the inverter.

6. The control device of an alternating-current rotary electric machine according to any one of claims 1 to 3, characterized in that The inverter is provided with three sets of series circuits corresponding to three phases, respectively, in which a switching element connected to a positive side of a positive side of a direct-current power supply and a switching element connected to a negative side of a negative side of the direct-current power supply are connected in series, and a connection point of the series connection is connected to the winding of the corresponding phase, and the series circuit of at least one phase has a resistance connected in series with the switching element of the positive side or the switching element of the negative side.

7. The control device of an alternating-current rotary electric machine according to claim 6, characterized in that The n zero-phase candidate voltage values are set in advance so that a difference between a maximum value within the n zero-phase candidate voltage values and a minimum value within the n zero-phase candidate voltage values is 5% or more of a direct-current voltage supplied to the inverter.

8. The control device of an alternating-current rotary electric machine according to claim 6, characterized in that The n zero-phase candidate voltage values are set in advance so that a difference between a maximum value within the n zero-phase candidate voltage values and a minimum value within the n zero-phase candidate voltage values is 10% or more of a direct-current voltage supplied to the inverter.

9. The control device of an alternating-current rotary electric machine according to claim 6, characterized in that The voltage command calculation section detects a current flowing through the winding based on a potential difference across the resistance, and calculates the three-phase voltage command values or the command voltage vector based on the detected value of the current.

10. The control device of an alternating-current rotary electric machine according to claim 9, characterized in that The n zero-phase candidate voltage values are set in advance so that a difference between a maximum value within the n zero-phase candidate voltage values and a minimum value within the n zero-phase candidate voltage values is 5% or more of a direct-current voltage supplied to the inverter.

11. The control device of an alternating-current rotary electric machine according to claim 9, characterized in that The n zero-phase candidate voltage values are set in advance so that a difference between a maximum value in the n zero-phase candidate voltage values and a minimum value in the n zero-phase candidate voltage values is 10% or more of a DC voltage supplied to the inverter.

12. An electric power assisted steering apparatus characterised in that Comprise: The control device of the alternating-current rotary electric machine according to any one of claims 1 to 11; The inverter; The alternating-current rotary electric machine; And A drive force transmission mechanism that transmits a drive force of the alternating-current rotary electric machine to a steering device of a vehicle, A PWM period of the PWM control section is set to 60 μs or less, The resonance period of the alternating-current rotary electric machine is mechanically in a range of 200 μs or more and 500 μs or less.

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